LT3481 36V, 2A, 2.8MHz Step-Down Switching Regulator with 50µA Quiescent Current FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ DESCRIPTION Wide Input Range: 3.6V to 34V Operating, 36V Maximum 2A Maximum Output Current Low Ripple Burst Mode® Operation 50μA IQ at 12VIN to 3.3VOUT Output Ripple < 15mV Adjustable Switching Frequency: 300kHz to 2.8MHz Low Shutdown Current: IQ < 1μA Integrated Boost Diode Power Good Flag Saturating Switch Design: 0.18Ω On-Resistance 1.265V Feedback Reference Voltage Output Voltage: 1.265V to 20V Soft-Start Capability Synchronizable Between 275kHz to 475kHz Small 10-Pin Thermally Enhanced MSOP and (3mm x 3mm) DFN Packages APPLICATIONS ■ ■ ■ ■ ■ The LT®3481 is an adjustable frequency (300kHz to 2.8MHz) monolithic buck switching regulator that accepts input voltages up to 34V (36V maximum). A high efficiency 0.18Ω switch is included on the die along with a boost Schottky diode and the necessary oscillator, control, and logic circuitry. Current mode topology is used for fast transient response and good loop stability. Low ripple Burst Mode operation maintains high efficiency at low output currents while keeping output ripple below 15mV in a typical application. In addition, the LT3481 can further enhance low output current efficiency by drawing bias current from the output when VOUT is above 3V. Shutdown reduces input supply current to less than 1μA while a resistor and capacitor on the RUN/SS pin provide a controlled output voltage ramp (soft-start). A power good flag signals when VOUT reaches 90% of the programmed output voltage. The LT3481 is available in 10-Pin MSOP and 3mm x 3mm DFN packages with exposed pads for low thermal resistance. , LT, LTC and LTM are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Automotive Battery Regulation Power for Portable Products Distributed Supply Regulation Industrial Supplies Wall Transformer Regulation TYPICAL APPLICATION 3.3V Step-Down Converter VIN 4.5V TO 34V VIN BD RUN/SS EFFICIENCY (%) 0.47μF 4.7μH VC LT3481 SW RT 330pF 80 1000.0 70 100.0 60 10.0 50 1.0 BIAS PG 60.4k 10000.0 BOOST 16.2k 4.7μF 90 324k GND 40 FB 22μF 200k 3481 TA01 VIN = 12V VOUT = 3.3V L = 4.7μ F = 800 kHz 30 0.0001 0.001 0.01 0.1 LOAD CURRENT (A) 1 POWER LOSS (mW) OFF ON Efficiency VOUT 3.3V 2A 0.1 0.01 10 3481 TA01b 3481fb 1 LT3481 ABSOLUTE MAXIMUM RATINGS (Note 1) VIN, RUN/SS Voltage .................................................36V BOOST Pin Voltage ...................................................56V BOOST Pin Above SW Pin.........................................30V FB, RT, VC Voltage .......................................................5V BIAS, PG, BD Voltage ................................................30V Maximum Junction Temperature........................... 125°C LT3481E, LT3481I ............................................. 125°C LT3481H ........................................................... 150°C Operating Temperature Range (Note 2) LT3481E............................................... –40°C to 85°C LT3481I.............................................. –40°C to 125°C LT3481H ............................................ –40°C to 150°C Storage Temperature Range................... –65°C to 150°C Lead Temperature (Soldering, 10 sec) (MSE Only) ....................................................... 300°C PIN CONFIGURATION TOP VIEW TOP VIEW BD 1 10 RT BOOST 2 9 VC SW 3 VIN 4 7 BIAS RUN/SS 5 6 PG 11 BD BOOST SW VIN RUN/SS 8 FB 1 2 3 4 5 11 10 9 8 7 6 RT VC FB BIAS PG MSE PACKAGE 10-LEAD PLASTIC MSOP DD PACKAGE 10-LEAD (3mm s 3mm) PLASTIC DFN EXPOSED PAD (PIN 11) IS GND MUST BE CONNECTED TO GND EXPOSED PAD (PIN 11) IS GND MUST BE CONNECTED TO GND θJA = 40°C/W θJA = 43°C/W ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING PACKAGE DESCRIPTION TEMPERATURE RANGE LT3481EDD#PBF LT3481EDD#TRPBF LBVS 10-Lead (3mm × 3mm) Plastic DFN –40°C to 85°C LT3481IDD#PBF LT3481IDD#TRPBF LBVV 10-Lead (3mm × 3mm) Plastic DFN –40°C to 125°C LT3481HDD#PBF LT3481HDD#TRPBF LDPT 10-Lead (3mm × 3mm) Plastic DFN –40°C to 150°C LT3481EMSE#PBF LT3481EMSE#TRPBF LTBVT 10-Lead Plastic MSOP –40°C to 85°C LT3481IMSE#PBF LT3481IMSE#TRPBF LTBVW 10-Lead Plastic MSOP –40°C to 125°C LT3481HMSE#PBF LT3481HMSE#TRPBF LTDPV 10-Lead Plastic MSOP –40°C to 150°C LEAD BASED FINISH TAPE AND REEL PART MARKING PACKAGE DESCRIPTION TEMPERATURE RANGE LT3481EDD LT3481EDD#TR LBVS 10-Lead (3mm × 3mm) Plastic DFN –40°C to 85°C LT3481IDD LT3481IDD#TR LBVV 10-Lead (3mm × 3mm) Plastic DFN –40°C to 125°C LT3481HDD LT3481HDD#TR LDPT 10-Lead (3mm × 3mm) Plastic DFN –40°C to 150°C LT3481EMSE LT3481EMSE#TR LTBVT 10-Lead Plastic MSOP –40°C to 85°C LT3481IMSE LT3481IMSE#TR LTBVW 10-Lead Plastic MSOP –40°C to 125°C LT3481HMSE LT3481HMSE#TR LTDPV 10-Lead Plastic MSOP –40°C to 150°C Consult LTC Marketing for parts specified with wider operating temperature ranges. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ 3481fb 2 LT3481 ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 10V, VRUNS/SS = 10V VBOOST = 15V, VBIAS = 3.3V unless otherwise noted. (Note 2) PARAMETER CONDITIONS MIN ● Minimum Input Voltage Quiescent Current from VIN VRUN/SS = 0.2V VBIAS = 3V, Not Switching ● VBIAS = 0, Not Switching Quiescent Current from BIAS VRUN/SS = 0.2V VBIAS = 3V, Not Switching ● VBIAS = 0, Not Switching Minimum Bias Voltage Feedback Voltage ● FB Pin Bias Current (Note 3) VFB = 1.25V, VC = 0.4V FB Voltage Line Regulation 4V < VIN < 34V 1.25 1.24 ● TYP MAX UNITS 3 3.6 V 0.01 0.5 μA 22 60 μA 75 120 μA 0.01 0.5 μA 50 120 μA 0 5 μA 2.7 3 V 1.265 1.265 1.29 1.3 V V 30 100 nA 0.002 0.02 %/V Error Amp GM 330 Error Amp Gain 800 VC Source Current 65 μMho μA VC Sink Current 85 μA VC Pin to Switch Current Gain 3.5 A/V VC Clamp Voltage Switching Frequency 2 RT = 8.66k RT = 29.4k RT = 187k 2.5 1.25 250 ● Minimum Switch Off-Time 3.2 V 2.8 1.4 300 3.1 1.55 350 MHz MHz kHz 130 200 nS 3.8 4.4 A Switch Current Limit Duty Cycle = 5% Switch VCESAT ISW = 2A 360 Boost Schottky Reverse Leakage VSW = 10V, VBIAS = 0V 0.02 2 μA 1.5 2.1 V ● Minimum Boost Voltage (Note 4) mV BOOST Pin Current ISW = 1A 18 35 mA RUN/SS Pin Current VRUN/SS = 2.5V 5 10 μA 0.2 V RUN/SS Input Voltage High 2.5 V RUN/SS Input Voltage Low PG Threshold Offset from Feedback Voltage VFB Rising 122 PG Leakage VPG = 5V 0.1 PG Sink Current VPG = 3V PG Hysteresis mV 5 Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LT3481E is guaranteed to meet performance specifications from 0°C to 85°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. The LT3481I specifications are ● 100 mV 1 μA 600 μA guaranteed over the –40°C to 125°C temperature range. The LT3481H specifications are guaranteed over the –40°C to 150°C operating temperature range. High junction temperatures degrade operating lifetimes. Operating lifetime is derated at junction temperatures greater than 125°C. Note 3: Bias current flows into the FB pin. Note 4: This is the minimum voltage across the boost capacitor needed to guarantee full saturation of the switch. 3481fb 3 LT3481 TYPICAL PERFORMANCE CHARACTERISTICS Efficiency (VOUT = 3.3V) Efficiency (VOUT = 5.0V) 90 VIN = 12V 80 EFFICIENCY (%) 60 50 40 60 20 0 0.0001 0.001 0.01 0.1 LOAD CURRENT (A) 1 VIN = 24V 40 20 L: NEC PLC-0745-4R7 f: 800kHz 0 0.0001 0.001 0.01 0.1 LOAD CURRENT (A) 1 30 20 10 VIN = 12V VOUT = 3.3V 3.5 250 INCREASED SUPPLY CURRENT DUE TO CATCH DIODE LEAKAGE AT HIGH TEMPERATURE 200 150 25 20 15 INPUT VOLTAGE (V) 30 0 25 50 75 100 TEMPERATURE (°C) 3.5 LOAD CURRENT (A) 2.5 MINIMUM 2.0 VOUT = 5.0V TA = 25 °C L = 4.7μ f = 800 kHz 20 15 INPUT VOLTAGE (V) 25 30 3481 G07 10 20 15 INPUT VOLTAGE (V) 25 30 3481 G06 Switch Current Limit DUTY CYCLE = 10 % 4.0 3.0 2.5 2.0 3.0 2.5 DUTY CYCLE = 90 % 2.0 1.5 1.0 0.5 1.0 10 5 4.5 1.5 1.0 VOUT = 3.3V TA = 25 °C L = 4.7μ f = 800 kHz 1.0 125 150 SWITCH CURRENT LIMIT (A) 3.5 SWITCH CURRENT LIMIT(A) 4.0 3.0 MINIMUM 2.0 Switch Current Limit Maximum Load Current 5 2.5 3481 G05 4.0 1.5 3.0 1.5 3481 G04 TYPICAL TYPICAL 100 0 –50 –25 35 3 Maximum Load Current 50 FRONT PAGE APPLICATION 10 1 2 2.5 1.5 SWITCHING FREQUENCY (MHz) 4.0 LOAD CURRENT (A) 300 SUPPLY CURRENT (μA) SUPPLY CURRENT (μA) 60 40 0.5 3481 G03 CATCH DIODE: DIODES, INC. PDS360 350 50 VOUT = 3.3V L = 10μH LOAD = 1A 0 No Load Supply Current TA = 25°C 5 65 50 10 400 70 0 70 3481 G02 No Load Supply Current 0 75 55 L: NEC PLC-0745-4R7 f: 800kHz 3481 G01 80 VIN = 24V 60 10 10 VIN = 12V 50 30 10 80 70 30 VIN = 12V 85 VIN = 7V 80 VIN = 24V 70 90 EFFICIENCY (%) 90 EFFICIENCY (%) Efficiency 100 100 0 20 60 40 DUTY CYCLE (%) 80 100 3481 G08 0 –50 –25 0 25 50 75 TEMPERATURE (°C) 100 125 3481 G09 3481fb 4 LT3481 TYPICAL PERFORMANCE CHARACTERISTICS Switch Voltage Drop 500 400 300 200 100 90 1.290 80 1.285 70 FEEDBACK VOLTAGE (V) BOOST PIN CURRENT (mA) VOLTAGE DROP (mV) 600 60 50 40 30 20 500 0 1000 1500 2000 2500 3000 3500 SWITCH CURRENT (mA) 500 1000 1500 2000 2500 3000 3500 SWITCH CURRENT (mA) 1.260 RT = 45.3k 125 150 Minimum Switch On-Time 140 1.00 0.95 0.90 1000 800 600 400 200 0.85 0 200 400 600 800 1000 1200 1400 FB PIN VOLTAGE (mV) 4381 G13 Soft Start 80 60 40 20 RUN/SS Pin Current 2.5 2.0 1.5 1.0 Boost Diode 1.4 10 8 6 4 3 3.5 3481 G16 1.0 0.8 0.6 0.2 0 0 1.2 0.4 2 0.5 25 50 75 100 125 150 TEMPERATURE (˚C) 1.6 BOOST DIODE Vf (V) RUN/SS PIN CURRENT (μA) 3.5 3.0 0 3481 G15 12 2.5 2 1.5 RUN/SS PIN VOLTAGE (V) 100 3481 G14 4.0 1 120 0 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 0 MINIMUM SWITCH ON TIME (ns) SWITCHING FREQUENCY (kHz) 1.05 0.5 25 50 75 100 TEMPERATURE (°C) RT = 45.3k 1.10 0 0 4381 G12 1200 1.15 SWITCH CURRENT LIMIT (A) 1.265 Frequency Foldback Switching Frequency 0.80 –50 –25 1.270 3481 G11 3481 G10 1.20 1.275 1.250 –50 –25 0 0 1.280 1.255 10 0 FREQUENCY (MHz) Feedback Voltage Boost Pin Current 700 0 5 20 30 15 25 10 RUN/SS PIN VOLTAGE (V) 35 3481 G17 0 0 1.0 0.5 1.5 BOOST DIODE CURRENT (A) 2.0 3481 G18 3481fb 5 LT3481 TYPICAL PERFORMANCE CHARACTERISTICS Error Amp Output Current Minimum Input Voltage Minimum Input Voltage 100 4.5 6.5 4.0 6.0 INPUT VOLTAGE (V) VC PIN CURRENT (μA) 60 40 20 0 –20 –40 INPUT VOLTAGE (V) 80 3.5 3.0 VOUT = 3.3V TA = 25 °C L = 4.7μ f = 800kHz 2.5 –60 –80 1.065 1?.265 1.165 1.365 FB PIN VOLTAGE (V) 2.0 0.001 1.465 5.0 4.5 0.1 0.01 1 LOAD CURRENT (A) 4.0 0.001 10 2.50 THRESHOLD VOLTAGE (V) 1.50 1.00 SWITCHING THRESHOLD 0.50 10 Switching Waveforms; Burst Mode 1.200 CURRENT LIMIT CLAMP 0.1 0.01 1 LOAD CURRENT (A) 3481 G21 Power Good Threshold VC Voltages 2.00 VOUT = 5.0V TA = 25 °C L = 4.7μ f = 800kHz 3481 G20 3481 G19 THRESHOLD VOLTAGE (V) 5.5 VIN = 12V; FRONT PAGE APPLICATION ILOAD = 10mA IL 0.5A/DIV 1.180 1.160 VSW 5V/DIV 1.140 VOUT 10mV/DIV 1.120 PG RISING 0 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 1.100 –50 –25 3481 G22 0 25 50 75 100 125 150 TEMPERATURE (°C) 5μs/DIV 3481 G24 3481 G23 Switching Waveforms; Transition from Burst Mode to Full Frequency Switching Waveforms; Full Frequency Continuous Operation IL 0.5A/DIV IL 0.5A/DIV VRUN/SS 5V/DIV VRUN/SS 5V/DIV VOUT 10mV/DIV VOUT 10mV/DIV VIN = 12V; FRONT PAGE APPLICATION ILOAD = 1A VIN = 12V; FRONT PAGE APPLICATION ILOAD = 140mA 1μs/DIV 3481 G25 1μs/DIV 3481 G26 3481fb 6 LT3481 PIN FUNCTIONS BD (Pin 1): This pin connects to the anode of the boost Schottky diode. BOOST (Pin 2): This pin is used to provide a drive voltage, higher than the input voltage, to the internal bipolar NPN power switch. SW (Pin 3): The SW pin is the output of the internal power switch. Connect this pin to the inductor, catch diode and boost capacitor. VIN (Pin 4): The VIN pin supplies current to the LT3481’s internal regulator and to the internal power switch. This pin must be locally bypassed. RUN/SS (Pin 5): The RUN/SS pin is used to put the LT3481 in shutdown mode. Tie to ground to shut down the LT3481. Tie to 2.3V or more for normal operation. If the shutdown feature is not used, tie this pin to the VIN pin. RUN/SS also provides a soft-start function; see the Applications Information section. PG (Pin 6): The PG pin is the open collector output of an internal comparator. PG remains low until the FB pin is within 10% of the final regulation voltage. PG output is valid when VIN is above 3.5V and RUN/SS is high. BIAS (Pin 7): The BIAS pin supplies the current to the LT3481’s internal regulator. Tie this pin to the lowest available voltage source above 3V (typically VOUT). This architecture increases efficiency especially when the input voltage is much higher than the output. FB (Pin 8): The LT3481 regulates the FB pin to 1.265V. Connect the feedback resistor divider tap to this pin. VC (Pin 9): The VC pin is the output of the internal error amplifier. The voltage on this pin controls the peak switch current. Tie an RC network from this pin to ground to compensate the control loop. RT (Pin 10): Oscillator Resistor Input. Connecting a resistor to ground from this pin sets the switching frequency. Exposed Pad (Pin 11): Ground. The Exposed Pad must be soldered to PCB. BLOCK DIAGRAM VIN 4 VIN C1 7 5 10 BIAS – + INTERNAL 1.265V REF RUN/SS 3 SLOPE COMP BD SWITCH LATCH BOOST 2 C3 R RT OSCILLATOR 300kHz–2.8MHz Q S SW RT DISABLE SOFT-START 6 1 L1 VOUT 3 C2 D1 BurstMode DETECT PG ERROR AMP + – + – 1.12V GND 11 FB VC CLAMP VC 9 CC RC CF 8 R2 R1 3481 BD 3481fb 7 LT3481 OPERATION The LT3481 is a constant frequency, current mode stepdown regulator. An oscillator, with frequency set by RT, enables an RS flip-flop, turning on the internal power switch. An amplifier and comparator monitor the current flowing between the VIN and SW pins, turning the switch off when this current reaches a level determined by the voltage at VC. An error amplifier measures the output voltage through an external resistor divider tied to the FB pin and servos the VC pin. If the error amplifier’s output increases, more current is delivered to the output; if it decreases, less current is delivered. An active clamp on the VC pin provides current limit. The VC pin is also clamped to the voltage on the RUN/SS pin; soft-start is implemented by generating a voltage ramp at the RUN/SS pin using an external resistor and capacitor. An internal regulator provides power to the control circuitry. The bias regulator normally draws power from the VIN pin, but if the BIAS pin is connected to an external voltage higher than 3V bias power will be drawn from the external source (typically the regulated output voltage). This improves efficiency. The RUN/SS pin is used to place the LT3481 in shutdown, disconnecting the output and reducing the input current to less than 1μA. The switch driver operates from either the input or from the BOOST pin. An external capacitor and diode are used to generate a voltage at the BOOST pin that is higher than the input supply. This allows the driver to fully saturate the internal bipolar NPN power switch for efficient operation. To further optimize efficiency, the LT3481 automatically switches to Burst Mode operation in light load situations. Between bursts, all circuitry associated with controlling the output switch is shut down reducing the input supply current to 50μA in a typical application. The oscillator reduces the LT3481’s operating frequency when the voltage at the FB pin is low. This frequency foldback helps to control the output current during startup and overload. The LT3481 contains a power good comparator which trips when the FB pin is at 91% of its regulated value. The PG output is an open-collector transistor that is off when the output is in regulation, allowing an external resistor to pull the PG pin high. Power good is valid when the LT3481 is enabled and VIN is above 3.6V. 3481fb 8 LT3481 APPLICATIONS INFORMATION FB Resistor Network The output voltage is programmed with a resistor divider between the output and the FB pin. Choose the 1% resistors according to: ⎛V ⎞ R1= R2 ⎜ OUT – 1⎟ ⎝ 1.265 ⎠ Reference designators refer to the Block Diagram. Setting the Switching Frequency The LT3481 uses a constant frequency PWM architecture that can be programmed to switch from 300kHz to 2.8MHz by using a resistor tied from the RT pin to ground. A table showing the necessary RT value for a desired switching frequency is in Figure 1. SWITCHING FREQUENCY (MHz) RT VALUE (kΩ) 0.2 0.3 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 2.6 2.8 267 187 133 84.5 60.4 45.3 36.5 29.4 23.7 20.5 16.9 14.3 12.1 10.2 8.66 Figure 1. Switching Frequency vs. RT Value Operating Frequency Tradeoffs Selection of the operating frequency is a tradeoff between efficiency, component size, minimum dropout voltage, and maximum input voltage. The advantage of high frequency operation is that smaller inductor and capacitor values may be used. The disadvantages are lower efficiency, lower maximum input voltage, and higher dropout voltage. The highest acceptable switching frequency (fSW(MAX)) for a given application can be calculated as follows: fSW(MAX ) = VD + VOUT tON(MIN) ( VD + VIN – VSW ) where VIN is the typical input voltage, VOUT is the output voltage, is the catch diode drop (~0.5V), VSW is the internal switch drop (~0.5V at max load). This equation shows that slower switching frequency is necessary to safely accommodate high VIN/VOUT ratio. Also, as shown in the next section, lower frequency allows a lower dropout voltage. The reason input voltage range depends on the switching frequency is because the LT3481 switch has finite minimum on and off times. The switch can turn on for a minimum of ~150ns and turn off for a minimum of ~150ns. This means that the minimum and maximum duty cycles are: DCMIN = fSW tON(MIN) DCMAX = 1– fSW tOFF(MIN) where fSW is the switching frequency, the tON(MIN) is the minimum switch on time (~150ns), and the tOFF(MIN) is the minimum switch off time (~150ns). These equations show that duty cycle range increases when switching frequency is decreased. A good choice of switching frequency should allow adequate input voltage range (see next section) and keep the inductor and capacitor values small. Input Voltage Range The maximum input voltage for LT3481 applications depends on switching frequency, the Absolute Maximum Ratings on VIN and BOOST pins, and on operating mode. If the output is in start-up or short-circuit operating modes, then VIN must be below 34V and below the result of the following equation: VIN(MAX ) = VOUT + VD –V +V fSW tON(MIN) D SW where VIN(MAX) is the maximum operating input voltage, VOUT is the output voltage, VD is the catch diode drop (~0.5V), VSW is the internal switch drop (~0.5V at max load), fSW is the switching frequency (set by RT), and tON(MIN) is the minimum switch on time (~150ns). Note that a higher switching frequency will depress the maximum operating input voltage. Conversely, a lower switching 3481fb 9 LT3481 APPLICATIONS INFORMATION frequency will be necessary to achieve safe operation at high input voltages. If the output is in regulation and no short-circuit or start-up events are expected, then input voltage transients of up to 36V are acceptable regardless of the switching frequency. In this mode, the LT3481 may enter pulse skipping operation where some switching pulses are skipped to maintain output regulation. In this mode the output voltage ripple and inductor current ripple will be higher than in normal operation. The minimum input voltage is determined by either the LT3481’s minimum operating voltage of ~3.6V or by its maximum duty cycle (see equation in previous section). The minimum input voltage due to duty cycle is: VIN(MIN) = VOUT + VD –V +V 1– fSW tOFF(MIN) D SW where VIN(MIN) is the minimum input voltage, and tOFF(MIN) is the minimum switch off time (150ns). Note that higher switching frequency will increase the minimum input voltage. If a lower dropout voltage is desired, a lower switching frequency should be used. Inductor Selection For a given input and output voltage, the inductor value and switching frequency will determine the ripple current. The ripple current ΔIL increases with higher VIN or VOUT and decreases with higher inductance and faster switching frequency. A reasonable starting point for selecting the ripple current is: ΔIL = 0.4((IOUT(MAX)) where IOUT(MAX) is the maximum output load current. To guarantee sufficient output current, peak inductor current must be lower than the LT3481’s switch current limit (ILIM). The peak inductor current is: IL(PEAK) = IOUT(MAX) + ΔIL/2 where IL(PEAK) is the peak inductor current, IOUT(MAX) is the maximum output load current, and ΔIL is the inductor ripple current. The LT3481’s switch current limit (ILIM) is at least 3.5A at low duty cycles and decreases linearly to 2.5A at DC = 0.8. The maximum output current is a function of the inductor ripple current: IOUT(MAX) = ILIM – ΔIL/2 Be sure to pick an inductor ripple current that provides sufficient maximum output current (IOUT(MAX)). The largest inductor ripple current occurs at the highest VIN. To guarantee that the ripple current stays below the specified maximum, the inductor value should be chosen according to the following equation: ⎛ VOUT + VD ⎞ ⎛ VOUT + VD ⎞ L=⎜ ⎜ 1– ⎟ VIN(MAX ) ⎟⎠ ⎝ fΔIL ⎟⎠ ⎜⎝ where VD is the voltage drop of the catch diode (~0.4V), VIN(MAX) is the maximum input voltage, VOUT is the output voltage, fSW is the switching frequency (set by RT), and L is in the inductor value. The inductor’s RMS current rating must be greater than the maximum load current and its saturation current should be about 30% higher. For robust operation in fault conditions (start-up or short circuit) and high input voltage (>30V), the saturation current should be above 3.5A. To keep the efficiency high, the series resistance (DCR) should be less than 0.1Ω, and the core material should be intended for high frequency applications. Table 1 lists several vendors and suitable types. Table 1. Inductor Vendors VENDOR URL PART SERIES Murata www.murata.com LQH55D Open TDK www.componenttdk.com SLF7045 SLF10145 Shielded Shielded Toko www.toko.com D62CB Shielded D63CB Shielded D75C Shielded D75F Open CR54 Open CDRH74 Shielded CDRH6D38 Shielded CR75 Open Sumida www.sumida.com TYPE 3481fb 10 LT3481 APPLICATIONS INFORMATION Of course, such a simple design guide will not always result in the optimum inductor for your application. A larger value inductor provides a slightly higher maximum load current and will reduce the output voltage ripple. If your load is lower than 2A, then you can decrease the value of the inductor and operate with higher ripple current. This allows you to use a physically smaller inductor, or one with a lower DCR resulting in higher efficiency. There are several graphs in the Typical Performance Characteristics section of this data sheet that show the maximum load current as a function of input voltage and inductor value for several popular output voltages. Low inductance may result in discontinuous mode operation, which is okay but further reduces maximum load current. For details of maximum output current and discontinuous mode operation, see Linear Technology Application Note 44. Finally, for duty cycles greater than 50% (VOUT/VIN > 0.5), there is a minimum inductance required to avoid subharmonic oscillations. See AN19. Input Capacitor Bypass the input of the LT3481 circuit with a ceramic capacitor of X7R or X5R type. Y5V types have poor performance over temperature and applied voltage, and should not be used. A 4.7μF to 10μF ceramic capacitor is adequate to bypass the LT3481 and will easily handle the ripple current. Note that larger input capacitance is required when a lower switching frequency is used. If the input power source has high impedance, or there is significant inductance due to long wires or cables, additional bulk capacitance may be necessary. This can be provided with a low performance electrolytic capacitor. Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input capacitor is required to reduce the resulting voltage ripple at the LT3481 and to force this very high frequency switching current into a tight local loop, minimizing EMI. A 4.7μF capacitor is capable of this task, but only if it is placed close to the LT3481 and the catch diode (see the PCB Layout section). A second precaution regarding the ceramic input capacitor concerns the maximum input voltage rating of the LT3481. A ceramic input capacitor combined with trace or cable inductance forms a high quality (under damped) tank circuit. If the LT3481 circuit is plugged into a live supply, the input voltage can ring to twice its nominal value, possibly exceeding the LT3481’s voltage rating. This situation is easily avoided (see the Hot Plugging Safety section). For space sensitive applications, a 2.2μF ceramic capacitor can be used for local bypassing of the LT3481 input. However, the lower input capacitance will result in increased input current ripple and input voltage ripple, and may couple noise into other circuitry. Also, the increased voltage ripple will raise the minimum operating voltage of the LT3481 to ~3.7V. Output Capacitor and Output Ripple The output capacitor has two essential functions. Along with the inductor, it filters the square wave generated by the LT3481 to produce the DC output. In this role it determines the output ripple, and low impedance at the switching frequency is important. The second function is to store energy in order to satisfy transient loads and stabilize the LT3481’s control loop. Ceramic capacitors have very low equivalent series resistance (ESR) and provide the best ripple performance. A good starting value is: COUT = 100 VOUT fSW where fSW is in MHz, and COUT is the recommended output capacitance in μF. Use X5R or X7R types. This choice will provide low output ripple and good transient response. Transient performance can be improved with a higher value capacitor if the compensation network is also adjusted to maintain the loop bandwidth. A lower value of output capacitor can be used to save space and cost but transient performance will suffer. See the Frequency Compensation section to choose an appropriate compensation network. 3481fb 11 LT3481 APPLICATIONS INFORMATION Table 2. Capacitor Vendors VENDOR PHONE URL PART SERIES Panasonic (714) 373-7366 www.panasonic.com Ceramic, COMMANDS Polymer, EEF Series Tantalum Kemet (864) 963-6300 www.kemet.com Ceramic, Tantalum Sanyo (408) 749-9714 www.sanyovideo.com T494, T495 Ceramic, Polymer, POSCAP Tantalum Murata (408) 436-1300 AVX www.murata.com Ceramic www.avxcorp.com Ceramic, Tantalum Taiyo Yuden (864) 963-6300 www.taiyo-yuden.com When choosing a capacitor, look carefully through the data sheet to find out what the actual capacitance is under operating conditions (applied voltage and temperature). A physically larger capacitor, or one with a higher voltage rating, may be required. High performance tantalum or electrolytic capacitors can be used for the output capacitor. Low ESR is important, so choose one that is intended for use in switching regulators. The ESR should be specified by the supplier, and should be 0.05Ω or less. Such a capacitor will be larger than a ceramic capacitor and will have a larger capacitance, because the capacitor must be large to achieve low ESR. Table 2 lists several capacitor vendors. TPS Series Ceramic Table 3. Diode Vendors PART NUMBER VR (V) IAVE (A) VF AT 1A (mV) VF AT 2A (mV) On Semicnductor MBRM120E MBRM140 20 40 1 1 530 550 595 Diodes Inc. B120 B130 B220 B230 DFLS240L 20 30 20 30 40 1 1 2 2 2 500 500 International Rectifier 10BQ030 20BQ030 30 30 1 2 420 500 500 500 470 470 Catch Diode Ceramic Capacitors The catch diode conducts current only during switch off time. Average forward current in normal operation can be calculated from: Ceramic capacitors are small, robust and have very low ESR. However, ceramic capacitors can cause problems when used with the LT3481 due to their piezoelectric nature. When in Burst Mode operation, the LT3481’s switching frequency depends on the load current, and at very light loads the LT3481 can excite the ceramic capacitor at audio frequencies, generating audible noise. Since the LT3481 operates at a lower current limit during Burst Mode operation, the noise is typically very quiet to a casual ear. If this is unacceptable, use a high performance tantalum or electrolytic capacitor at the output. ID(AVG) = IOUT (VIN – VOUT)/VIN where IOUT is the output load current. The only reason to consider a diode with a larger current rating than necessary for nominal operation is for the worst-case condition of shorted output. The diode current will then increase to the typical peak switch current. Peak reverse voltage is equal to the regulator input voltage. Use a diode with a reverse voltage rating greater than the input voltage. Table 3 lists several Schottky diodes and their manufacturers. 3481fb 12 LT3481 APPLICATIONS INFORMATION A final precaution regarding ceramic capacitors concerns the maximum input voltage rating of the LT3481. A ceramic input capacitor combined with trace or cable inductance forms a high quality (under damped) tank circuit. If the LT3481 circuit is plugged into a live supply, the input voltage can ring to twice its nominal value, possibly exceeding the LT3481’s rating. This situation is easily avoided (see the Hot Plugging Safely section). cases a zero is required and comes from either the output capacitor ESR or from a resistor RC in series with CC. This simple model works well as long as the value of the inductor is not too high and the loop crossover frequency is much lower than the switching frequency. A phase lead capacitor (CPL) across the feedback divider may improve the transient response. Figure 3 shows the transient response when the load current is stepped from 500mA to 1500mA and back to 500mA. Frequency Compensation CURRENT MODE POWER STAGE gm = 3.5mho SW ERROR AMPLIFIER OUTPUT R1 FB + Loop compensation determines the stability and transient performance. Designing the compensation network is a bit complicated and the best values depend on the application and in particular the type of output capacitor. A practical approach is to start with one of the circuits in this data sheet that is similar to your application and tune the compensation network to optimize the performance. Stability should then be checked across all operating conditions, including load current, input voltage and temperature. The LT1375 data sheet contains a more thorough discussion of loop compensation and describes how to test the stability using a transient load. Figure 2 shows an equivalent circuit for the LT3481 control loop. The error amplifier is a transconductance amplifier with finite output impedance. The power section, consisting of the modulator, power switch and inductor, is modeled as a transconductance amplifier generating an output current proportional to the voltage at the VC pin. Note that the output capacitor integrates this current, and that the capacitor on the VC pin (CC) integrates the error amplifier output current, resulting in two poles in the loop. In most LT3481 – The LT3481 uses current mode control to regulate the output. This simplifies loop compensation. In particular, the LT3481 does not require the ESR of the output capacitor for stability, so you are free to use ceramic capacitors to achieve low output ripple and small circuit size. Frequency compensation is provided by the components tied to the VC pin, as shown in Figure 2. Generally a capacitor (CC) and a resistor (RC) in series to ground are used. In addition, there may be lower value capacitor in parallel. This capacitor (CF) is not part of the loop compensation but is used to filter noise at the switching frequency, and is required only if a phase-lead capacitor is used or if the output capacitor has high ESR. 1.265V CPL gm = 330μmho ESR C1 + 3Meg C1 VC CF POLYMER OR TANTALUM GND RC CERAMIC R2 CC 3481 F02 Figure 2. Model for Loop Response VOUT = 12V; FRONT PAGE APPLICATION IL 1A/DIV VOUT 100mV/DIV 10μs/DIV 3481 F03 Figure 3. Transient Load Response of the LT3481 Front Page Application as the Load Current is Stepped from 500mA to 1500mA. VOUT = 3.3V 3481fb 13 LT3481 APPLICATIONS INFORMATION Burst Mode Operation To enhance efficiency at light loads, the LT3481 automatically switches to Burst Mode operation which keeps the output capacitor charged to the proper voltage while minimizing the input quiescent current. During Burst Mode operation, the LT3481 delivers single cycle bursts of current to the output capacitor followed by sleep periods where the output power is delivered to the load by the output capacitor. In addition, VIN and BIAS quiescent currents are reduced to typically 20μA and 50μA respectively during the sleep time. As the load current decreases towards a no load condition, the percentage of time that the LT3481 operates in sleep mode increases and the average input current is greatly reduced resulting in higher efficiency. See Figure 4. boost diode can be tied to the input (Figure 5c), or to another supply greater than 2.8V. The circuit in Figure 5a is more efficient because the BOOST pin current and BIAS pin quiescent current comes from a lower voltage source. You must also be sure that the maximum voltage ratings of the BOOST and BIAS pins are not exceeded. VOUT BD BOOST VIN VIN LT3481 GND 4.7μF C3 SW (5a) For VOUT > 2.8V VOUT VIN = 12V; FRONT PAGE APPLICATION ILOAD = 10mA D2 BD IL 0.5A/DIV BOOST VIN VIN GND 4.7μF VSW 5V/DIV VOUT 10mV/DIV LT3481 C3 SW (5b) For 2.5V < VOUT < 2.8V 5μs/DIV VOUT 3481 F04 BD BOOST Figure 4. Burst Mode Operation VIN BOOST and BIAS Pin Considerations Capacitor C3 and the internal boost Schottky diode (see the Block Diagram) are used to generate a boost voltage that is higher than the input voltage. In most cases a 0.22μF capacitor will work well. Figure 2 shows three ways to arrange the boost circuit. The BOOST pin must be more than 2.3V above the SW pin for best efficiency. For outputs of 3V and above, the standard circuit (Figure 5a) is best. For outputs between 2.8V and 3V, use a 1μF boost capacitor. A 2.5V output presents a special case because it is marginally adequate to support the boosted drive stage while using the internal boost diode. For reliable BOOST pin operation with 2.5V outputs use a good external Schottky diode (such as the ON Semi MBR0540), and a 1μF boost capacitor (see Figure 5b). For lower output voltages the 4.7μF VIN LT3481 GND C3 SW (5c) For VOUT < 2.5V 3481 FO5 Figure 5. Three Circuits For Generating The Boost Voltage The minimum operating voltage of an LT3481 application is limited by the minimum input voltage (3.6V) and by the maximum duty cycle as outlined in a previous section. For proper startup, the minimum input voltage is also limited by the boost circuit. If the input voltage is ramped slowly, or the LT3481 is turned on with its RUN/SS pin when the output is already in regulation, then the boost capacitor 3481fb 14 LT3481 APPLICATIONS INFORMATION may not be fully charged. Because the boost capacitor is charged with the energy stored in the inductor, the circuit will rely on some minimum load current to get the boost circuit running properly. This minimum load will depend on input and output voltages, and on the arrangement of the boost circuit. The minimum load generally goes to zero once the circuit has started. Figure 6 shows a plot of minimum load to start and to run as a function of input voltage. In many cases the discharged output capacitor will present a load to the switcher, which will allow it to start. The plots show the worst-case situation where VIN is ramping very slowly. For lower start-up voltage, the boost diode can be tied to VIN; however, this restricts the input range to one-half of the absolute maximum rating of the BOOST pin. At light loads, the inductor current becomes discontinuous and the effective duty cycle can be very high. This reduces the minimum input voltage to approximately 300mV above VOUT. At higher load currents, the inductor current is continuous and the duty cycle is limited by the maximum duty cycle of the LT3481, requiring a higher input voltage to maintain regulation. Soft-Start The RUN/SS pin can be used to soft-start the LT3481, reducing the maximum input current during start-up. The RUN/SS pin is driven through an external RC filter to create a voltage ramp at this pin. Figure 7 shows the startup and shut-down waveforms with the soft-start circuit. By choosing a large RC time constant, the peak start-up current can be reduced to the current that is required to regulate the output, with no overshoot. Choose the value of the resistor so that it can supply 20μA when the RUN/SS pin reaches 2.3V. IL 1A/DIV RUN 15k RUN/SS 0.22μF VRUN/SS 2V/DIV GND VOUT 2V/DIV 2ms/DIV 3481 F07 Figure 7. To Soft-Start the LT3481, Add a Resisitor and Capacitor to the RUN/SS Pin 6.0 TO START Synchronization INPUT VOLTAGE (V) 5.5 5.0 4.5 4.0 TO RUN 3.5 VOUT = 3.3V TA = 25°C L = 4.7μ f = 800 kHz 3.0 2.5 2.0 0.001 0.1 0.01 1 LOAD CURRENT (A) 10 The LT3481 should not be synchronized until its output is near regulation as indicated by the PG flag. This can be done with the system microcontroller/microprocessor or with a discrete circuit by using the PG output. If a sync signal is applied while the PG is low, the LT3481 may exhibit erratic operation. See Typical Applications 8.0 TO START INPUT VOLTAGE (V) 7.0 6.0 5.0 TO RUN 4.0 3.0 2.0 0.001 VOUT = 5.0V TA = 25°C L = 4.7μ f = 800 kHz 0.1 0.01 1 LOAD CURRENT (A) The internal oscillator of the LT3481 can be synchronized to an external 275kHz to 475kHz clock by using a 5pF to 20pF capacitor to connect the clock signal to the RT pin. The resistor tying the RT pin to ground should be chosen such that the LT3481 oscillates 20% lower than the intended synchronization frequency (see Setting the Switching Frequency section). 10 3481 F06 Figure 6. The Minimum Input Voltage Depends on Output Voltage, Load Current and Boost Circuit When applying a sync signal, positive clock transitions reset LT3481’s internal clock and negative transitions initiate a switch cycle. The amplitude of the sync signal must be at least 2V. The sync signal duty cycle can range 3481fb 15 LT3481 APPLICATIONS INFORMATION from 5% up to a maximum value given by the following equation: ⎛ VOUT + VD ⎞ DCSYNC(MAX ) = ⎜ 1 – – f • 600ns VIN – VSW + VD ⎟⎠ SW ⎝ where VOUT is the programmed output voltage, VD is the diode forward drop, VIN is the typical input voltage, VSW is the switch drop, and fSW is the desired switching frequency. For example, a 24V input to 5V output at 300kHz can be synchronized to a square wave with a maximum duty cycle of 60%. For some applications, such as 12VIN to 5VOUT at 350kHz, the maximum allowable sync duty cycle will be less than 50%. If a low duty cycle clock cannot be obtained from the system, then a one-shot should be used between the sync signal and the LT3481. See Typical Applications. The value of the coupling capacitor which connects the clock signal to the RT pin should be chosen based on the clock signal amplitude. Good starting values for 3.3V and 5V clock signals are 10pF and 5pF, respectively. These values should be tested and adjusted for each individual application to assure reliable operation. Caution should be used when synchronizing more than 50% above the initial switching frequency (as set by the RT resistor) because at higher clock frequencies the amplitude of the internal slope compensation used to prevent subharmonic switching is reduced. This type of subharmonic switching only occurs at input voltages less than twice output voltage. Higher inductor values will tend to reduce this problem. Shorted and Reversed Input Protection If the inductor is chosen so that it won’t saturate excessively, an LT3481 buck regulator will tolerate a shorted output. There is another situation to consider in systems where the output will be held high when the input to the LT3481 is absent. This may occur in battery charging applications or in battery backup systems where a battery or some other supply is diode OR-ed with the LT3481’s output. If the VIN pin is allowed to float and the RUN/SS pin is held high (either by a logic signal or because it is tied to VIN), then the LT3481’s internal circuitry will pull its quiescent current through its SW pin. This is fine if your system can tolerate a few mA in this state. If you ground the RUN/SS pin, the SW pin current will drop to essentially zero. However, if the VIN pin is grounded while the output is held high, then parasitic diodes inside the LT3481 can pull large currents from the output through the SW pin and the VIN pin. Figure 8 shows a circuit that will run only when the input voltage is present and that protects against a shorted or reversed input. D4 MBRS140 VIN VIN BOOST LT3481 RUN/SS VOUT SW VC GND FB BACKUP 3481 F08 Figure 8. Diode D4 Prevents a Shorted Input from Discharging a Backup Battery Tied to the Output. It Also Protects the Circuit from a Reversed Input. The LT3481 Runs Only When the Input is Present PCB Layout For proper operation and minimum EMI, care must be taken during printed circuit board layout. Figure 9 shows the recommended component placement with trace, ground plane and via locations. Note that large, switched currents flow in the LT3481’s VIN and SW pins, the catch diode (D1) and the input capacitor (C1). The loop formed by these components should be as small as possible. These components, along with the inductor and output capacitor, should be placed on the same side of the circuit board, and their connections should be made on that layer. Place a local, unbroken ground plane below these components. The SW and BOOST nodes should be as small as possible. Finally, keep the FB and VC nodes small so that the ground traces will shield them from the SW and BOOST nodes. The Exposed Pad on the bottom of the package must be soldered to ground so that the pad acts as a heat sink. To keep thermal resistance low, extend the ground plane as much as possible, and add thermal vias under and near the LT3481 to additional ground planes within the circuit board and on the bottom side. 3481fb 16 LT3481 APPLICATIONS INFORMATION Hot Plugging Safely L1 C2 VOUT CC RRT RC R2 R1 D1 C1 RPG GND 3481 F09 VIAS TO LOCAL GROUND PLANE VIAS TO RUN/SS VIAS TO VOUT VIAS TO PG VIAS TO VIN OUTLINE OF LOCAL GROUND PLANE Figure 9. A Good PCB Layout Ensures Proper, Low EMI Operation CLOSING SWITCH SIMULATES HOT PLUG IIN VIN The small size, robustness and low impedance of ceramic capacitors make them an attractive option for the input bypass capacitor of LT3481 circuits. However, these capacitors can cause problems if the LT3481 is plugged into a live supply (see Linear Technology Application Note 88 for a complete discussion). The low loss ceramic capacitor, combined with stray inductance in series with the power source, forms an under damped tank circuit, and the voltage at the VIN pin of the LT3481 can ring to twice the nominal input voltage, possibly exceeding the LT3481’s rating and damaging the part. If the input supply is poorly controlled or the user will be plugging the LT3481 into an energized supply, the input network should be designed to prevent this overshoot. Figure 10 shows the waveforms that result when an LT3481 circuit is connected to a 24V supply through six feet of 24-gauge twisted pair. The DANGER VIN 20V/DIV RINGING VIN MAY EXCEED ABSOLUTE MAXIMUM RATING LT3481 + 4.7μF LOW IMPEDANCE ENERGIZED 24V SUPPLY IIN 10A/DIV STRAY INDUCTANCE DUE TO 6 FEET (2 METERS) OF TWISTED PAIR 20μs/DIV (10a) 0.7Ω LT3481 VIN 20V/DIV + 0.1μF 4.7μF IIN 10A/DIV (10b) LT3481 + 22μF 35V AI.EI. 20μs/DIV VIN 20V/DIV + 4.7μF IIN 10A/DIV (10c) 20μs/DIV 3481 F10 Figure 10. A Well Chosen Input Network Prevents Input Voltage Overshoot and Ensures Reliable Operation when the LT3481 is Connected to a Live Supply 3481fb 17 LT3481 APPLICATIONS INFORMATION first plot is the response with a 4.7μF ceramic capacitor at the input. The input voltage rings as high as 50V and the input current peaks at 26A. A good solution is shown in Figure 10b. A 0.7Ω resistor is added in series with the input to eliminate the voltage overshoot (it also reduces the peak input current). A 0.1μF capacitor improves high frequency filtering. For high input voltages its impact on efficiency is minor, reducing efficiency by 1.5 percent for a 5V output at full load operating from 24V. High Temperature Considerations The PCB must provide heat sinking to keep the LT3481 cool. The Exposed Pad on the bottom of the package must be soldered to a ground plane. This ground should be tied to large copper layers below with thermal vias; these layers will spread the heat dissipated by the LT3481. Place additional vias can reduce thermal resistance further. With these steps, the thermal resistance from die (or junction) to ambient can be reduced to θJA = 35°C/W or less. With 100 LFPM airflow, this resistance can fall by another 25%. Further increases in airflow will lead to lower thermal resistance. Because of the large output current capability of the LT3481, it is possible to dissipate enough heat to raise the junction temperature beyond the absolute maximum of 125°C (150°C for the H grade). When operating at high ambient temperatures, the maximum load current should be derated as the ambient temperature approaches 125°C (150°C for the H grade). Power dissipation within the LT3481 can be estimated by calculating the total power loss from an efficiency measurement and subtracting the catch diode loss. The die temperature is calculated by multiplying the LT3481 power dissipation by the thermal resistance from junction to ambient. Other Linear Technology Publications Application Notes 19, 35 and 44 contain more detailed descriptions and design information for buck regulators and other switching regulators. The LT1376 data sheet has a more extensive discussion of output ripple, loop compensation and stability testing. Design Note 100 shows how to generate a bipolar output supply using a buck regulator. TYPICAL APPLICATIONS 5V Step-Down Converter VOUT 5V 2A VIN 6.3V TO 34V BD VIN ON OFF RUN/SS BOOST 0.47μF VC 4.7μF LT3481 SW D RT 20k L 6.8μH BIAS PG 590k 60.4k GND 330pF f = 800kHz FB 22μF 200k 3481 TA02 D: DIODES INC. DFLS240L L: TAIYO YUDEN NP06DZB6R8M 3481fb 18 LT3481 TYPICAL APPLICATIONS 3.3V Step-Down Converter VIN 4.4V TO 34V VIN BD RUN/SS ON OFF VOUT 3.3V 2A BOOST L 4.7μH 0.47μF VC 4.7μF SW LT3481 D RT 16.2k PG BIAS 324k 60.4k FB GND 330pF 22μF 200k f = 800kHz 3481 TA03 D: DIODES INC. DFLS240L L: TAIYO YUDEN NP06DZB4R7M 2.5V Step-Down Converter VOUT 2.5V 2A VIN 4V TO 34V BD VIN RUN/SS ON OFF D2 BOOST L 4.7μH 1μF VC 4.7μF SW LT3481 D1 RT 22.1k PG BIAS 196k 84.5k FB GND 220pF 47μF 200k f = 600kHz 3481 TA04 D1: DIODES INC. DFLS240L D2: MBR0540 L: TAIYO YUDEN NP06DZB4R7M 5V, 2MHz Step-Down Converter VIN 8.6V TO 22V TRANSIENT TO 36V VIN ON OFF VOUT 5V 2A BD RUN/SS BOOST 0.47μF VC 2.2μF LT3481 SW D RT 20k L 2.2μH PG BIAS 590k 16.9k GND 330pF f = 2MHz FB 10μF 200k 3481 TA05 D: DIODES INC. DFLS240L L: SUMIDA CDRM4D22/HP-2R2 3481fb 19 LT3481 TYPICAL APPLICATIONS 12V Step-Down Converter VOUT 12V 2A VIN 15V TO 34V VIN BD RUN/SS ON OFF BOOST 0.47μF VC 10μF SW LT3481 D RT 30k L 10μH PG BIAS 845k 60.4k FB GND 330pF 22μF 100k f = 800kHz 3481 TA06 D: DIODES INC. DFLS240L L: NEC/TOKIN PLC-0755-100 5V Step-Down Converter with Sync Input VIN 20V TO 34V BD VIN 4.7μF NOTE: DO NOT APPLY SYNC SIGNAL UNTIL PGOOD GOES HIGH RUN/SS ON OFF VOUT 5V 2A BOOST L 8.2μH 0.47μF VC 8.2pF RT SYNC IN SW D LT3481 3.3V SQ WAVE 300kHz TO 375kHz PGOOD BIAS PG 11.8k 75pF 100k 29.4k 226k 1000pF VOUT f = 250kHz GND FB 47μF 10k 3481 TA07 D: DIODES INC. DFLS240L L: NEC/TOKIN PLC-0755-8R2 3481fb 20 LT3481 TYPICAL APPLICATIONS 5V Step-Down Converter with Sync and One-Shot VOUT 5V 2A VIN 8V TO 34V BD VIN 4.7μF RUN/SS ON OFF BOOST L 15μH 0.47μF VC 15pF 1k SYNC IN SW RT 3V SQ WAVE Hz TO 450kHz LT3481 BIAS PG 25k 11.8k 75pF 100k Q1 29.4k 133k 25k D AND 50pF 1000pF FB GND VOUT 47μF 10k f = 300kHz 3481 TA08 D: DIODES INC. DFLS240L L: NEC/TOKIN PLC-0755-150 Q1: ON SEMI MMBT3904 1.8V Step-Down Converter VOUT 1.8V 2A VIN 3.5V TO 27V VIN ON OFF BD RUN/SS BOOST 0.47μF VC 4.7μF LT3481 SW D RT 15.4k L 3.3μH PG BIAS 84.5k 105k GND 330pF f = 500kHz FB 47μF 200k 3481 TA09 D: DIODES INC. DFLS240L L: TAIYO YUDEN NP06DZB3R3M 3481fb 21 LT3481 PACKAGE DESCRIPTION DD Package 10-Lead Plastic DFN (3mm × 3mm) (Reference LTC DWG # 05-08-1699) 0.675 p0.05 3.50 p0.05 1.65 p0.05 2.15 p0.05 (2 SIDES) PACKAGE OUTLINE 0.25 p 0.05 0.50 BSC 2.38 p0.05 (2 SIDES) RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS R = 0.115 TYP 6 3.00 p0.10 (4 SIDES) 0.38 p 0.10 10 1.65 p 0.10 (2 SIDES) PIN 1 TOP MARK (SEE NOTE 6) (DD10) DFN 1103 5 0.200 REF 1 0.25 p 0.05 0.50 BSC 0.75 p0.05 0.00 – 0.05 2.38 p0.10 (2 SIDES) BOTTOM VIEW—EXPOSED PAD NOTE: 1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-2). CHECK THE LTC WEBSITE DATA SHEET FOR CURRENT STATUS OF VARIATION ASSIGNMENT 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE 3481fb 22 LT3481 PACKAGE DESCRIPTION MSE Package 10-Lead Plastic MSOP (Reference LTC DWG # 05-08-1663) BOTTOM VIEW OF EXPOSED PAD OPTION 2.794 p 0.102 (.110 p .004) 5.23 (.206) MIN 0.889 p 0.127 (.035 p .005) 1 2.06 p 0.102 (.081 p .004) 1.83 p 0.102 (.072 p .004) 2.083 p 0.102 3.20 – 3.45 (.082 p .004) (.126 – .136) 10 0.50 0.305 p 0.038 (.0197) (.0120 p .0015) BSC TYP RECOMMENDED SOLDER PAD LAYOUT 3.00 p 0.102 (.118 p .004) (NOTE 3) 10 9 8 7 6 3.00 p 0.102 (.118 p .004) (NOTE 4) 4.90 p 0.152 (.193 p .006) 0.254 (.010) DETAIL “A” 0o – 6o TYP 1 2 3 4 5 GAUGE PLANE 0.53 p 0.152 (.021 p .006) DETAIL “A” 0.18 (.007) 0.497 p 0.076 (.0196 p .003) REF SEATING PLANE 0.86 (.034) REF 1.10 (.043) MAX 0.17 – 0.27 (.007 – .011) TYP 0.50 (.0197) BSC 0.127 p 0.076 (.005 p .003) MSOP (MSE) 0603 NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX 3481fb Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 23 LT3481 TYPICAL APPLICATION 1.265V Step-Down Converter VOUT 1.265V 2A VIN 3.6V TO 27V BD VIN RUN/SS ON OFF BOOST 0.47μF VC 4.7μF LT3481 SW D RT 13k BIAS PG 105k L 3.3μH GND FB 330pF 47μF f = 500kHz 3481 TA10 D: DIODES INC. DFLS240L L: TAIYO YUDEN NP06DZB3R3M RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1933 500mA (IOUT), 500kHz Step-Down Switching Regulator in SOT-23 VIN: 3.6V to 36V, VOUT(MIN) = 1.2V, IQ = 1.6mA, ISD <1μA, ThinSOT Package LT3437 60V, 400mA (IOUT), MicroPower Step-Down DC/DC Converter with Burst Mode VIN: 3.3V to 80V, VOUT(MIN) = 1.25V, IQ = 100μA, ISD <1μA, DFN Package LT1936 36V, 1.4A (IOUT), 500kHz High Efficiency Step-Down DC/DC Converter VIN: 3.6V to 36V, VOUT(MIN) = 1.2V, IQ = 1.9mA, ISD <1μA, MS8E Package LT3493 36V, 1.2A (IOUT), 750kHz High Efficiency Step-Down DC/DC Converter VIN: 3.6V to 40V, VOUT(MIN) = 0.8V, IQ = 1.9mA, ISD <1μA, DFN Package LT1976/LT1977 60V, 1.2A (IOUT), 200kHz/500kHz, High Efficiency Step-Down DC/DC Converter with Burst Mode VIN: 3.3V to 60V, VOUT(MIN) = 1.20V, IQ = 100μA, ISD <1μA, TSSOP16E Package LT1767 25V, 1.2A (IOUT), 1.1MHz, High Efficiency Step-Down DC/DC Converter VIN: 3.0V to 25V, VOUT(MIN) = 1.20V, IQ = 1mA, ISD <6μA, MS8E Package LT1940 Dual 25V, 1.4A (IOUT), 1.1MHz, High Efficiency Step-Down DC/DC Converter VIN: 3.6V to 25V, VOUT(MIN) = 1.20V, IQ = 3.8mA, ISD <30μA, TSSOP16E Package LT1766 60V, 1.2A (IOUT), 200kHz, High Efficiency Step-Down DC/DC Converter VIN: 5.5V to 60V, VOUT(MIN) = 1.20V, IQ = 2.5mA, ISD = 25μA, TSSOP16E Package LT3434/LT3435 60V, 2.4A (IOUT), 200/500kHz, High Efficiency Step-Down DC/DC Converter with Burst Mode VIN: 3.3V to 60V, VOUT(MIN) = 1.20V, IQ = 100μA, ISD <1μA, TSSOP16E Package 3481fb 24 Linear Technology Corporation LT 1008 REV B • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2006