IXBD4410
IXBD4411
ISOSMARTTM Half Bridge Driver Chipset
Type
Description
IXBD4410PI
IXBD4411PI
Full-Feature Low-Side Driver 16-Pin P-DIP
Full-Feature High-Side Driver 16-Pin P-DIP
-40 to +85°C
-40 to +85°C
IXBD4410SI
IXBD4411SI
Full-Feature Low-Side Driver 16-Pin SO
Full-Feature High-Side Driver 16-Pin SO
-40 to +85°C
-40 to +85°C
The IXBD4410/IXBD4411 ISOSMART
chipset is designed to control the gates
of two Power MOSFETs or Power
IGBTs that are connected in a halfbridge (phase-leg) configuration for
driving multiple-phase motors, or used
in applications that require half-bridge
power circuits. The IXBD4410/
IXBD4411 is a full-feature chipset
consisting of two 16-Pin DIP or SO
devices interfaced and isolated by two
small-signal ferrite pulse transformers.
The small-signal transformers provide
greater than 1200 V isolation.
Even with commutating noise ambients
greater than ±50 V/ns and up to 1200 V
potentials, this chipset establishes
error-free two-way communications
between the system ground-reference
IXBD4410 and the inverter output-
Package
Temperature Range
reference IXBD4411. They incorporate
undervoltage VDD or VEE lockout and
overcurrent or desaturation shutdown
to protect the IGBT or Power MOSFET
devices from damage.
The chipset provides the necessary
gate drive signals to fully control the
grounded-source low-side power
device as well as the floating-source
high-side power device. Additionally,
the IXBD4410/4411 chipset provides a
negative-going, off-state gate drive
signal for improved turn-off of IGBTs or
Power MOSFETs and a system logiccompatible status fault output FLT to
indicate overcurrent or desaturation,
and undervoltage VDD or VEE. During a
status fault, both chipset keep their
respective gate drive outputs off; at
VEE.
Features
z
z
z
z
z
z
z
z
z
z
z
z
z
z
1200 V or greater low-to-high side
isolation.
Drives Power Systems Operating on
up to 575 V AC mains
dv/dt immunity of greater than
±50V/ns
Proprietary low-to-high side level
translation and communication
On-chip negative gate-drive supply
to ensure Power MOSFET or IGBT
turn-off and to prevent gate noise
interference
5 V logic compatible HCMOS inputs
with hysteresis
Available in either the 16-Pin DIP or
the 16-Pin wide-body, small-outline
plastic package
20 ns switching time with 1000 pF
load; 100 ns switching time with
10,000 pF load
100 ns propagation delay time
2 A peak output drive capability
Self shut-down of output in response
to over-current or short-circuit
Under-voltage and over-voltage VDD
lockout protection
Protection from cross conduction of
the half bridge
Logic compatible fault indication
from both low and high-side driver
Applications
540 Vz
z
z
z
z
z
1- or 3-Phase Motor Controls
Switch Mode Power Supplies
(SMPS)
Uninterruptible Power Supplies
(UPS)
Induction Heating and Welding
Systems
Switching Amplifiers
General Power Conversion Circuits
IXYS reserves the right to change limits, test conditions and dimensions.
© 2004 IXYS All rights reserved
1
IXBD4410
IXBD4411
Symbol
Definition
Maximum Ratings
VDD/VEE
Vin
Supply Voltage
Input Voltage (INH, INL)
-0.5 ... 24
-0.5...VDD +0.5
V
V
Iin
Io (rev)
Input Current (INL, INH, IM)
Peak Reverse Output Current (OUT)
±10
2
A
A
PD
PD
Maximum Power Dissipation (TA = 25° C)
TC = 25° C (16-Pin SOIC)
600
10
mW
W
TA
TJM
Tstg
TL
Operating Ambient Temperature
Maximum Junction Temperature
Storage Temperature Range
Lead Soldering Temperature for 10 s
RthJA
RthJC
(16-Pin SOIC)
-40 ... 85
150
-55 ... 150
300
1.67
10
Dimensions in inch (1" = 25.4 mm)
16-Pin SOIC
°C
°C
°C
°C
K/W
K/W
Die substrate
connected to tab
Recommended Operating Conditions
VDD/VEE
VDD/LG
LGh/LGl
Supply Voltage
10 ... 20
V
10 ... 16.5
V
±50
V/ns
Symbol
Definition/Condition
Characteristic Values
(TA = 25°C, VDD = 15 V, unless otherwise specified)
min.
typ.
max.
Maximum Common Mode dv/dt
INL, INH Inputs (referred to LG)
Vt+
VtVih
Iin
Cin
Positive-Going Threshold
Negative-Going Threshold
Input Hysteresis
Input Leakage Current/Vin=VDD or LG
Input Capacitance
3.65
1
1
-1
1
10
V
V
V
µA
pF
16-Pin Plastic DIP
Open Drain Fault Output (referred to LG)
Voh
Vol
HI Output/Rpu = 10 kΩ to VDD
LO Output/Io = 4 mA
VDD-0.05
0.3
0.5
V
V
OUT Output (referred to LG)
Voh
Vol
Ro
Ro
Ipk
HI Output/Io = -5 mA
LO Output/Io = 5 mA
Output HI Res./Io = -0.1 A
Output LO Res./Io = 0.1 A
Peak Output Current/CL = 10 nF
VDD-0.05
1.5
VEE+0.05
3
3
2
5
4
V
V
Ω
Ω
A
IM Input (referred to KG)
Vt+
Cin
Rs
Positive-Going Threshold
Input Capacitance
Shorting Device Output Resistance
0.24
50
0.3
10
75
0.45
-6.5
-25
600
-7.5
100
V
pF
Ω
End view
VEE Supply (referred to LG)
VEE
Iout
finv
VEEF
Output Voltage/Io = 1 mA, Co = 1 µF
Output Current/Vout = 0.70 • VEE
Inverting Frequency
Undervoltage Fault Indication
-5
-20
-3
-4.8
V
mA
kHz
V
© 2004 IXYS All rights reserved
IXBD4410
IXBD4411
Symbol
Definition/Condition
Characteristic Values
(TA = 25°C, VDD = 15 V, unless otherwise specified)
min.
typ.
max.
VDD Undervoltage Lockout
Vuv
Vuh
Drop Out
Hysteresis
9.5
0.1
10.5
0.15
11.5
0.3
V
V
Quiescent Power Supply Current
IDD
VDD Current/Vin=VDD or LG, Io = 0
20
mA
INL and INH Inputs (Fig. 1a - 1c)
td(on)
Turn-on delay time;
4410
CL =1nF
110
175
ns
tr
Rise time;
CL =10 nF
CL =1 nF
70
15
100
20
ns
ns
td(off)
Turn-off delay time
4410
CL =1nF
70
150
ns
tf
Fall time
CL =10nF
CL =1nF
70
15
150
20
ns
ns
tdlh(off)
4410/4412
Turn-on delay time vs.
CL =1nF
60
150
ns
CL =1nF
60
150
ns
200
300
ns
200
300
ns
4411
Turn-off delay time
tdlh(on)
4410
Turn-on delay time vs.
4411
Turn-off delay time
Fault Output Delay for any Fault Conditions (4410/4411)
tFLT
FLT Delay/Rpu = 2 kΩ CL = 20 pF
Overcurrent Protection Delay
toc
Driver-Off delay time
CL = 1 nF
1N5817
1N5817
Fig. 1a: IXBD4410/4411 Switching time test circuit
© 2004 IXYS All rights reserved
Fig. 1b: Output signal waveform
3
IXBD4410
IXBD4411
Chipset Overview
This ISOSMARTTM chipset is a pair of
integrated circuits providing isolated
high- and low-side drivers for phaseleg motor controls, or any other
application which utilizes a half bridge,
2- or 3-phase drive configuration. They
consist of two drive control inputs (INL
and INH) for two Power-MOSFET/IGBT
gate-drive outputs. Both inputs operate
from a common ground and are
activated by HCMOS compatible logic
levels. The low-side output operates
near input ground, while the high-side
output operates from a floating ground
that is nominally the source connection
of the high-side phase-leg power
device. Both outputs typically provide 2
A of transient current drive for fast
switching of the phase-leg power
device.
IXBD4410/IXBD4411
The full featured ISOSMARTTM driver
chipset incorporates a IXBD4410 as
the low-side driver (Fig. 3) and a
IXBD4411 as the high-side driver (Fig.
2). When input "INL" is set to a positive
logic level, the low-side gate output
goes high (turns on); when "INH" is set
to a positive logic level, the high-side
gate drive output goes high. The highside IC is isolated from the low-side IC
by a magnetic barrier, across which the
turn on/off signal is transmitted to the
high-side gate drive. The IXBD4411
fault signal is also transmitted back to
the IXBD4410 driver via these
transformers. This isolation only
depends on the low cost communications transformer, which is designed to
withstand 1200 V or more.
There are two magnetic transmission
channels between the low- and highside IC's for bi-directional communication. One sends a signal from the
low-side IXBD4410 IC up to the highside IXBD4411 IC and the other sends
a signal back from the high-side to the
low-side IC. The signal that is sent up
controls the IXBD4411 gate-drive
output. The signal sent from the
IXBD4411 back to the IXBD4410
indicates a high-side fault has occurred
(overcurrent, or under-voltage of the
high-side +power supplies). This is
detected at the IXBD4410 driver and
sets "FLT" pin low, to indicate the highside fault.
The fault signal that is returned from
15
VDD
UVDD
UVEE
14
VEE
13
12
Fig. 2: IXBD4411, high-side driver block diagram
15
VDD
UVDD
UVEE
14
VEE
13
12
Fig. 3: IXBD4410, low-side driver block diagram
Fig. 4: Logic representation of IXBD4410 FLT signal
© 2004 IXYS All rights reserved
IXBD4410
IXBD4411
the IXBD4411 is strictly for status only.
Any gate-drive shutdown because of a
high-side fault is done locally within the
high-side IXBD4411. The IXBD4411
gate-drive will turn-off the power device
whenever an overcurrent or under
voltage condition arises. The overcurrent
sensing is active only while the gate
driver output is "high" (on). The
overcurrent fault condition is latched and
is reset on the next INH gate input
positive transition. The FLT (pin 8) of the
IXBD4411 is not used and should be
grounded.
The low-side IXBD4410 driver provides
an output pin 8 (FLT) to indicate a highside (IXBD4411) or a low-side
(IXBD4410) fault. This output pin is an
"open-drain" output. The IXBD4410 lowside driver fault indications are similar to
the IXBD4411 high-side driver indications
as outlined above. A "graphic" logic
diagram of the chipset's FLT function is
presented in Fig.4. Note that this diagram
presents the logic of this function at the
"low-side" IXBD4410 driver and is not the
actual circuit. It describes the combined
logic of the "fault logic" and "hi-side fault
sense" blocks in both the IXBD4410 and
IXBD4411 as shown in Fig. 2 and 3.
The most efficient method of providing
power for the high-side driver is by
bootstrapping. This method is illustrated
in the functional drawing on page 4 and
in the application example (Fig. 6 and 9)
by diode D1 and capacitor C1. Using this
method, the power is drawn through a
high-voltage diode onto a reservoir
capacitor whenever the floating high-side
ground returns to near the real ground of
the low-side driver. When the high-side
gate is turned on and the floating ground
moves towards a higher potential, the
bootstrapping diode back-biases, and the
high-side driver draws its power solely
from the reservoir capacitor. Power may
also be provided via any isolated power
supply (usually an extra secondary on
the system housekeeping supply
switching transformer).
Both the IXBD4410 and IXBD4411
contain on-board negative charge pumps
to provide negative gate drive, which
ensures turn-off of the high- or low-side
power device in the presence of currents
induced by power device Miller capacitance or from inductive ground transients.
These charge pumps provide -5 V
relative to the local driver ground when
© 2004 IXYS All rights reserved
VDD is at +15 V, and at rated average
currents of 25 mA. The charge pump
requires two external capacitors, C7 and
C11 in Fig. 6. The charge pump
frequency is nominally 600 kHz. The
charge pump clock is turned off
whenever the difference between the VDD
and VEE supplies exceed 20 V, to prevent
exceeding the breakdown rating of the
IC.
Both the IXBD4410 and IXBD4411
drivers possess two local grounds each,
a common logic ground, and "Kelvin"
ground. The Kelvin ground and logic
grounds are first connected directly to
each other, and then to the Kelvin-source
of the power device for accurate overcurrent measurement in the presence of
inductive transients on the power device
source terminal.
Power MOSFET or IGBT overcurrent
sensing utilizes an on-chip comparator
with a typical 300 mV threshold. In a
typical application, the current mirror pin
of the Power MOSFET or IGBT is connected to a grounded, low-value resistor,
and to the overcurrent comparator input
on the high- or low-side driver. The
comparator will respond typically within
150 ns to an overcurrent condition to
shutdown the driver output. The power
switches could be protected also by
desa-turation detection (see Fig. 6, 7 and
9).
To assure maximum protection for the
phase-leg power devices, the chipset
incorporates the following Power
MOSFET and IGBT protection circuits:
z
Power device overcurrent or desaturation protection. The IXBD4410/
4411will turn off the driven device
within 150 ns of sensing an output
overcurrent or desaturation condition.
z
Gate-drive lockout circuitry to prevent
cross conduction (simultaneous
conduction of the low- and high-side
phase-leg power devices), either
under normal operating conditions or
when a fault occurs.
z
During power-up, the chipset's gatedrive outputs will be low (off), until the
voltage reaches the under-voltage trip
point.
z
Under-voltage gate-drive lockout on
the low- and/or high- side driver whenever the respective positive power
supply falls below 9.5 V typically.
z
Under-voltage gate-drive lockout on
the low- and high- side driver
whenever the respective negative
power supply rises above -3 V
typically.
5
IXBD4410
IXBD4411
Pin Description
IXBD4410 (Low-Side Driver)
Pin Description
IXBD4411 (High-Side Driver)
VDD
Sym. Pin Description of IXBD 4410/4411
VDD 1 Positive power supply.
16
INL
NC
2
Logic input signal referenced to
LG (logic ground). In the
IXBD4410. A "high" to this pin
turns on its gate drive output and
resets its fault logic. A "low" to
this pin turns off the gate drive
output. In the IXBD4411 this pin
is not used and should be
connected to its ground (LG).
No Connection (IXBD 4411)
INH
NC
3
Logic input signal referenced to
LG (logic ground). In the
IXBD4410, this signal is
transmitted to the IXBD4411
"high-side" driver through pins 4
and 5 (T- and T+). A "high" to
this pin turns on the IXBD4411
gate drive output and resets its
fault logic. A "low" to this pin
turns off the IXBD4411 gate
drive output. In the IXBD4411
this pin is not used and should
be connected to its ground (LG).
No Connection (IXBD 4411)
TT+
4
5
Transmitter output complementary drive signals. Direct drive of
the low signal transformer, which
is connected to the receiver of
the chipset's companion device.
In the IXBD4410, this signal
transmits the on/off command to
its companion IXBD4411. In the
IXBD4411, this signal transmits
the fault indication to its
companion IXBD4410 driver.
RR+
6
7
Receiver input complementary
signal. Directly connected to the
low signal transformer, which is
driven by the chipset's companion device. In the IXBD4410,
this input receives the fault
indication from its companion
IXBD4411 driver. In the
IXBD4411, this input receives
the on/off command from its
companion IXBD4410 driver.
VEE
T Connected to
A Pin 14 (substrate)
B
VDD
NC
OUT
NC
VEE
T-
CA
T+
CB
R-
LG
R+
KG
NC
IM
Sym. Pin Description of IXBD 4410/4411
FLT
NC
8
Low/high side fault output. In the
IXBD4410, this output indicates
a fault condition of either device
of the chipset. A "high" indicates
no fault, A "low" indicates that
either overcurrent,VDD or VEE
under-voltage occurred. In case
of overcurrent, this output will
remain active "low" until the next
input cycle of the respective
driver. In case of under-voltage,
this output will remain "low" until
the proper voltage is restored.
The IXBD4411 does not have a
FLT output,and its pin 8 should
be tied to LG
No Connection (IXBD 4411)
IM
9
Current sense or desaturation
detection input. This input is
active only while the OUT pin is
"high" (on). When the OUT pin is
"low" (off) this input is pulled to
ground through a 70 Ω resistor.
Any voltage at this pin above the
threshold of .3 V typical, will turn
the output (pin 15) off. This pin is
used for power device
overcurrent protection.
KG
10 Kelvin ground. This ground is
used as Kelvin connection for
overcurrent or desaturation
sensing.
LG
11 Logic and power ground.
CB
12 Capacitor terminals for negative
charge pump (VEE); "+"
13 terminal is CB (pin 12).
CA
VEE 14 Negative supply terminal(substrate)
OUT 15 Gate drive output. In the
IXBD4410 this output responds
to the INL signal. A "high" at INL
will turn it on ("high"), a "low" will
turn it off ("low"). In the
IXBD4411, this output responds
to the transmitted signal from the
companion IXBD4410. A "high"
at INH of the IXBD4410 drives
will turn it on ("high"). A "low"
will turn it off ("low"). This output
will turn off ("low") also in
response to any fault condition.
© 2004 IXYS All rights reserved
IXBD4410
IXBD4411
Application
The IXBD4410/4411 chipset devices
are specifically designed as MOSgated transistor drivers in half-bridge
power converters, 1- and 3-phase
motor controls, and UPS applications.
The phase-leg PWM command is
normally generated by previous (user
provided) circuitry. It must be
decomposed into two separate logic
signals, one for the high-side and one
for the low-side power transistors, with
appropriate deadtime for each state
transition. The deadtime insures nonoverlapping conduction even if the
turn-on and turn-off delay times of the
power devices are unequal. The
minimum deadtime should be greater
than tdlh. A separate circuit, or an IC
device like the IXYS deadtime
generator IXDP630, can be used to
perform this function. The
ISOSMART™ chipset family of
devices do not generate deadtime,
although there is an internal lockout
that prohibits one device form being
commanded "on" before the other is
commanded "off". This simplifies startup and shutdown protection circuitry,
preventing logic error during power-up
from turning on both high-and low-side
transistors simultaneously.
Negative VEE Charge Pump Circuit
Design
The on-chip VEE generator provided in
the IXBD4410/4411 generates a negative power supply, regulated at 20 V
below the positive VDD rail. If VDD is +10
V, VEE will be -10 V. If VDD is +15 V, VEE
will be -5 V. This negative drive
potential in the off-state is either
desirable or required in many instances. When switching a clamped
inductive load (Fig. 5), the turn-on of
Q2 will commutate the freewheeling
diode around Q1. Whether this diode is
intrinsic (as in a MOSFET) or extrinsic
(IGBT or bipolar), its reverse recovery
is critical to proper circuit operation.
At high turn-on di/dt in Q2 and near its
rated voltage, the recovery of D1 can
get quite "snappy" (the di/dt in the
second half of the recovery process,
after the diode has begun to recover its
blocking capability, can get very large),
creating a very high dv/dt across Q1.
This dv/dt is impressed across the
Miller capacitance of Q1, forcing a
large current to flow out the gate
© 2004 IXYS All rights reserved
reduces the voltage required to create
a failure, this problem is even more
likely to occur. In an industrial module
package (e.g.: a 150 A/1200 V IGBT
phase-leg module), the series inductance contributed by the long gate
leads and connectors further complicate the design.
Fig. 5: Switching a clamped inductive
load
terminal of the device. If this current
pulse causes a high enough voltage
drop across the output impedance of
the gate drive circuit, Rout, Q1 will be
turned on.
The Q1 conduction in every instance
Q2 is turned on (and vice versa), aside
from degrading efficiency, can lead to
catastrophic failure of both power
transistors. At high temperature, where
the -6 to -7 mV/°C temperature
coefficient of IGBT/MOSFET threshold
In a heavily snubbered converter, or in
a power supply design with low
transformer leakage inductance, the
design problem is relatively simple and
negative drive is seldom required.
However, in a modern snubberless or
lightly snubbered converter design, it is
important to keep the gate drive
impedance high enough during
transistor turnoff to limit the reapplied
dv/dt (the transistor is its own 'active'
snubber). This is always important for
EMI control, and in the case of IGBT
may be required to achieve the
necessary RBSOA. At the same time, it
is mandatory to keep the off-state gate
drive impedance very low to assure the
transistor remain off during induced
dv/dt (including diode recovery dv/dt).
In some instances, it is simply not
possible to satisfy both criteria with 0 V
applied in the off-state. In these cases
the IXBD4410/4411 with VEE negative
bias generator must be used.
The internal VEE generator is a charge
pump circuit. Referring to Fig. 6, an
external charge pump capacitor is
required between the CA and CB
1N5817
1N5817
Fig. 6: IXBD4410/4411 Detailed one phase circuit with dead time generator IXDP 630
7
IXBD4410
IXBD4411
terminals (C7, C11), and an output
reservoir capacitor between VEE and
GND (C10, C14). A 0.1 µF charge
pump capacitor (C7, C11) is recommended. The voltage regulation
method used in the IXBD4410/4411
allows a 1 to 2 V ripple frequency
depends on the size of the VEE output
reservoir capacitor (C10, C14) and the
average load current. The minimum
recommended output reservoir (C10,
C14) is 4.7 µF tantalum, or 10 µF if
aluminium electrolytic construction is
chosen. Note that this reservoir
capacitor is in addition to a good
quality high frequency bypass capacitor (0.1 µF) that should be placed from
VEE to GND (C9, C13).
A small resistor in series with the
charge pump capacitor, (R7, R8)
reduces the peak charging currents of
the charge pump. A value or 68 Ω or
greater is recommended, as illustrated
in the applications example in Fig. 6.
Current Sense / Desaturation
Detection Circuit
All members of the ISOSMART™
driver family provide a very flexible
overcurrent/short circuit protection
capability that works with both standard
three-terminal power transistors, and
with 4- and 5-terminal current sensing
power devices. Overcurrent detection
is accomplished as illustrated in Fig. 7a
(for a current mirror power device) and
Fig. 7b (for a standard three terminal
power transistor). Desaturation
detection is accomplished with the
same internal circuits by measuring the
voltage across the power transistor in
the on-state with an external resistor
divider (Fig. 7c).
The IM input trip point VTIM, typically
300 mV, is referenced to the Kelvin
ground pin KG.
Current Mirror MOSFET and IGBT allow
good control of peak let-through
currents and excellent short circuit
protection when combined with the
ISOSMART™ driver family of devices.
The sense resistor is chosen to
develop 300 mV at the desired peak
transistor current, assuming a mirror
ration of 1400:1, and a trip point of 30
A is desired:
Rs = 300 mV • 1400/30 A = 14 Ω
(use 15 Ω CC).
It is important to realize that Coss per
a
b
IM
c
IM
Rs
KG
GND
With Current Mirror
IM
Rs
KG
GND
Rs
KG
GND
With Standard
MOSFET/IGBT
Desaturation Detection
with Standard
MOSFET/IGBT
Fig. 7: Alternative overcurrent protection circuits
unit area of the mirror cells is much
larger that Coss per unit area of the bulk
of the chip due to periphery effects.
This causes a large transient current
pulse at the mirror output whenever the
transistor switches (C • dv/dt currents),
which can cause false overcurrent
trigger. The RC filter indicated in Fig.
7a will eliminate this problem.
Standard three-terminal MOSFET and
IGBT devices (in discrete as well as
modern industrial single transistor and
phase-leg modules) can also be
protected from short circuit with the
ISOSMART™ driver family devices. In
discrete device designs, where the
source/emitter terminal is available,
overcurrent protection with an external
power resistor can be implemented.
The resistor is placed in series with the
device emitter, with the full device
current flowing through it (Fig. 7b). The
sense resistor is again selected to
develop 300 mV at the desired peak
transistor current, assuming a trip point
of 30 A is desired:
Rs = 300 mV / 30 A = 10 mΩ
(use 10 mΩ, noninductive
current sense resistor).
It is important to recognize that
"noninductive" is a relative term,
especially when applied to current
sense resistor construction and
characterization. There is always
significant series inductance inserted
with the sense resistor, and L • di/dt
voltage transients can cause false
overcurrent trigger.
The RC filter indicated in Figure 7b will
eliminate this problem. Choosing the
RC pole at the current sense resistor
RL zero should exactly compensate for
series inductance. Because the exact
value is not normally known (and can
vary depending on PC layout and
component lead dress) this is not
normally a good idea. Usually, the RC
time constant should be two to ten
times longer than the suspected RL
time constant.
Desaturation detection as in Figure 7c
is probably the most common method
of short circuit protection in use today.
While not strictly an "overcurrent"
detector, if the power transistor gain,
and consequently short circuit letthrough current, is well controlled (as
with modern MOSFET and IGBT) this
methodology offers very effective
protection.
The IXBD4410/4411 half-bridge circuits
in Fig. 6 uses desaturation detection. In
Fig. 6, the voltage across the two
power MOSFET devices (or IGBTs) are
monitored by two sets of voltagedivider networks, R10 and R11 for the
high-side gate driver, and R13 and R14
for the low-side gate driver. The
dividers are set to trip the IM input
comparators when either Power
MOSFET device VDS exceeds a
reasonable value, perhaps 50 V
(usually a value of 10 % of the nominal
DC bus voltage works well). R10 or
R13 are chosen to tolerate the applied
steady state DC bus voltage at an
acceptable power dissipation.
Dielectric withstand capability, power
handling, temperature rise, and PC
board creep and strike spacings, must
all be carefully considered in the
design of the voltage-divider networks.
In the off-state, the voltage across the
Power MOSFET device may go as high
as the DC bus potential. To keep this
normal condition from setting the
internal fault flip-flop of the IXBD4410
or the IXBD4411, an internal CMOS
switch is turned on and placed across
lM and KG pins shorting them together.
This effectively discharges C8 or C12
in Fig. 6 and maintains zero potential
© 2004 IXYS All rights reserved
IXBD4410
IXBD4411
with respect to KG at IM.
When the command arrives to switch
on the Power MOSFET device, the
CMOS switch shorting IM to KG is
turned off. The driven Power MOSFET
device is switched on approximately
100 ns to 1 µs later, and with typical
load conditions, its drain-to-source
potential, VDS, may take an additional
10 µs of delay to collapse to the normal
on-state voltage level. To prevent false
triggering due to this, C8 or C12 in
parallel combination with R10 and R11,
or R13 and R14, delays the IM input
signal. During this turn-on interval, the
voltage across C8 or C12 will rise until
the Power MOSFET device finally
comes on and pulls the voltage across
C8 or C12 back down. If the MOSFET
device load circuit is shorted, its VDS
voltage cannot collapse at turn-on. In
this case, the voltage across C8 or C12
rises rapidly until it reaches 300 mV,
tripping the fault flip-flop and shutting
down the driver output. At the same
time, C8 or C12 must be kept small
enough that the added delay does not
slow down the detection of a short
circuit event so much that the Power
MOSFET device fails before the driver
realizes that it is in trouble.
Three Phase Motor Controls
Fig. 8 is a block diagram of a typical 3phase PWM voltage-source inverter
motor control. The power circuit
consists of six power switching transistors with freewheeling diodes around
each of them. The control function may
be performed digitally by a microprocessor, microcontroller, DSP chip, or user
custom IC; or it may be performed by a
PC board full of random logic and
analog circuits. In any of these cases,
the PWM command for all six power
transistors is generated in one circuit,
and this circuit is usually referred to
system ground potential - the bottom
terminal of the power bridge.
The ISOSMART™ family of drivers is
the interface between the world of
control logic and the world of power, 5 V
input logic commands precisely control
actions at high voltage and current
(1200 V and 100 A in a typical application). Fig. 6 is a detailed schematic of
one phase of three 3-phase motor
control, showing the interconnection of
the IXBD4410/4411 and its associated
circuitry.
© 2004 IXYS All rights reserved
Fig. 8: Typical 3-phase motor control system block diagram
PCB Layout Considerations
The IXBD4410/4411 is intended to be
used in high voltage, high speed, high
dv/dt applications.
To ensure proper operation, great care
must be taken in laying out the printed
circuit board. The layout critical areas
include the communication links, current
sense, gate drive, and supply bypassing.
The communication path should be as
short as possible. Added inductance
disturbs the frequency response of the
signal path, and these distortions may
cause false triggering in the receiver.
The transformer should be placed
between the two ICs with the orientation
of one IC reversed (Fig. 9).
Capacitance between the high- and lowside should be minimized. No signal
trace should run underneath the
communication path, and high- and lowside traces should be separated on the
PCB. The dv/dt of the high-side during
power stage switching may cause false
logic transitions in low-side circuits due
to capacitive coupling.
The low signal pulse transformer
provides the isolation between high-and
low-side circuits. For 460 V~ line
operation, a spacing of 4 mm is recommended between low- and high-side
circuits, and a transformer HIPOT
specification of at least 1500 V~ is
required. This creep spacing is usually
adequate to control leakage currents on
the PCB with up to 1200 V~ applied
after 10 to 15 years of accumulated dust
and particulates in a standard industrial
environment. In other environments, or
at other line voltages, this spacing
should be appropriately modified.
The current sense/desaturation detect
input is noise sensitive. The 300 mV trip
point is referred to the KG (Kelvin
ground) pin, and the applied signal must
be kept as clean as possible, A filter is
recommended, preferably a monolithic
ceramic capacitor placed as close to the
IC as possible directly between IM and
KG. To preserve maximum noise
immunity, the KG pin should first be
connected directly to the LG pin, and the
pair then sent directly to the power
transistor source/emitter terminal, or (if a
desaturation detection circuit is used) to
the bottom of the divider resistor chain.
All supply pins must be bypassed with a
low impedance capacitor (preferably
monolithic ceramic construction) with
minimum lead length. The output driver
stage draws 2 A (typical) currents during
transitions at di/dt values in excess of
100 A/µs. Supply line inductance will
cause supply and ground bounce on the
chip that can cause problems (logic
oscillations and, in severe cases,
possible latch-up failure) without proper
bypassing. These bypass elements are
in addition to the reservoir capacitors
required for the negative Vee supply and
the high-side bootstrapped supply if
these features are used.
Power Circuit Noise Considerations
In a typical transistor inverter, the output
MOSFET may switch on or off with di/dt
>500 A/µs. Referring to Fig.10 and
assuming that the MOSFET source
terminal has a one inch path on the PCB
to system ground, a voltage as high as
V = 27 nH • 500 A/µs = 13.5 V can be
developed.
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IXBD4410
IXBD4411
If the MOSFET switched 25 A, the
transient will last as long as (25/500) µs
or 50 ns, which is more than the typical
6 or 7 ns propagations or of a 74HC
series gate.
Fig. 10 illustrates an example layout
problem. The power circuit consists of
three power transistors (MOSFETs in
this example). With the ISOSMART™
gate driver chipset grounded as in
option (b) in Fig. 10, the communication
path from the IXDP630 will operate
without errors. The PC trace induced
voltages are not common with the digital
drive output follows (after its propagation delay) and the MOSFET starts to
conduct. The voltage transient induced
across LS1 (V = LS1 • di/dt) raises the
local ground (point a) until it exceeds
Voh (630) - Vil and the driver (after its
propagation delay) turns the MOSFET
off. Now the MOSFET current falls,
V(LS1) drops, point (a) drops to system
ground (or slightly below), and the driver
detects a "1" at its input. After its
propagation delay, it again turns the
conversion equipment due to their very
high common mode dv/dt rejection
capabilities.
Transformer Considerations
The transformer is the communication
link and isolation barrier between the
high- and low-side ICs. The high-side
gate and fault signals are transmitted
through the transformer while maintaining the proper isolation. The
transmitter signal is in the form of a
Fig. 9: Suggested IC Orientation
path, so the input of the gate driver will
not see or respond to them.
Unfortunately, the MOSFET will not
operate properly. The voltage induced
across LS1 when Q1 is turned on, acts
as source degeneration, modifying the
turn-on behavior of the MOSFET. If
LS1= 27 nH, and VCC is 15 V (assuming
the gate plateau of the MOSFET is 6 V),
the di/dt at turn-on will be regulated by
the driver/MOSFET/LS1 loop to about
200 A/µs; quite a surprise when your
circuit requires 500 A/µs to operate
correctly.
It is possible to make use of this
behavior to create a turn-on or turn-off
di/dt limiter (perhaps to snub the upper
free wheeling diode reverse recovery).
While possible, this is normally not
desirable or practical where two or more
transistors are controlled. Equalizing the
parasitic impedances of three traces
while positioning the transistors next to
their heat sink and meeting UL/VDE
voltage spacings is just too difficult.
Grounding the gate driver as in option
(a) in Fig. 10 solves the MOSFET turn
on problem by eliminating LS1 from the
source feedback loop. Now, unfortunately, the gate driver will oscillate
every time it is turned on or off. As the
IXDP630 output goes "high", the gate
Fig. 10: Potential layout problems that create functional problems
MOSFET on, continuing the oscillation
for one more cycle.
To eliminate this problem, a ground
level transformation circuit must be
added, that rejects this common mode
transient. The simplest is a de-coupling
circuit, also illustrated in Fig. 10. The
capacitor voltage on Cd remains
constant while the transient voltage is
dropped across Rd and the driver
detects no input transition, eliminating
the oscillation. This circuit does add
significantly to turn-on and turn-off delay
time, and cannot be used if the transient
lasts longer than the allowable delays.
Delay times must be considered in
selection of system dead time.
The most complex (and most effective)
method of eliminating the effects of
transients between grounds is isolation.
Optocouplers and pulse transformers
are the most commonly used isolation
techniques, and work very well in this
case. The IXDP630/631 has been
specifically designed to directly drive a
high speed optocoupler like the Hewlett
Packard HCPL22XX family or the
General Instrument 740L60XX optologic
family. These optos are especially well
suited to motor control and power
square wave, but the receiver responds
only to the logic edges. This allows for
much smaller transformer designs,
since a 10 kHz switching frequency
does not require a 10 kHz pulse
transformer.
The recommended transformer for this
ISOSMART™ driver chipset is
fabricated using a very small ferrite
shield bead (see Fig. 11), onto which a
six-turn primary and a two-turn
secondary winding of 36 AWG magnet
wire are made. The two windings are
segment wound to achieve primary-tosecondary isolation of up to 2500 V~.
The six-turn primaries are connected to
the respective IXBD4410/4411
transmitter outputs and the two-turn
secondaries are connected to their
respective receiver inputs.
© 2004 IXYS All rights reserved
IXBD4410
IXBD4411
The nominal electrical specifications of
the transformer are as follows:
z
z
z
z
z
z
Open circuit inductance
(100 kHz; 20 mV):
3 µH
Interwinding capacitance:
2 pF
Primary leakage inductance: 0.1 µH
Turns ratio:
6:2
Primary-to-secondary isolation
(1min):
1500 V~
Core permeability (µi):
125
The recommended ferrite bead is Fair
Rite Products part number 2661000101,
which is manufactured by:
Fair-Rite Products Corp.
Wallkill, NY
Phone: (800) 836-0427
Web site: www.fair-rite.com
As seen in the application drawings
(Fig. 6, 9 and 12) a coupling capacitor
(22 nF) and a damping resistor (22 Ω)
are added in series with the primary
side of the transformer. The capacitor
will control the small amount of energy
needed to transfer the signal to the
companion driver. The resistor will
control the damping of the signal and
limit the peak transmitter output current.
The receiver is designed to operate
Fig. 11: Ferrite bead dimensions
over a wide common mode input range.
To reduce noise pickup, the receiver
has ±250 mV of input hysteresis.
If the signal is being distorted at the
transmitter, the transmitter is probably
running into current limit. A decrease in
the coupling capacitance or an increase
in the damping resistance should solve
this problem. The receiver operates
over a wide input range. The minimum
amplitude for one side of the receiver is
about 1 V and a maximum of about 3 V.
It is critical that there be no overshoot
on the transformer secondary waveform. Each signal should be slightly
overdamped. If significant overshoot
exists, the received signal may be
logically inverted. An increase of the
damping resistor will solve this problem.
Fig. 12: Transmitter/Receiver Waveforms
DS96528E(05/04)
© 2004 IXYS All rights reserved
11