TI UCD7231

UCD7231
www.ti.com
SLUS997 – NOVEMBER 2009 – REVISED JANUARY 2010
Digital Control Compatible Synchronous-Buck Gate Driver
With Current Sense and Fault Protection
Check for Samples: UCD7231
FEATURES
1
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•
•
•
•
•
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Dual High Current Drivers.
Full Compatibility with TI Fusion Digital Power
Supply Controllers, such as UCD91xx and
UCD92xx Families
Operational to 2 MHz Switching Frequency
High-Side FET and Output Current Limit
Protection with Independently Adjustable
Thresholds
Fast High-Side Over-Current Sense Circuit
with Fault Flag Output – Prevents Catastrophic
Current Levels on a Cycle by Cycle Basis
Differential High-Gain Current Sense Amplifier
Voltage Proportional to Load Current Monitor
Output
Wide Input Voltage Range: 4.7 V to 15 V
Operation to 2.2 V Input Supported with an
External 4.5-6.5 V Bias Supply
Onboard Regulated Supplies for Gate Drive
and Internal Circuits
Integrated Thermal Shutdown
Selectable Operation Modes:
– PWM plus Synchronous Rectifier Enable
(SRE) with Automatic Dead-Time Control
– Direct High-Gate and Low-Gate Inputs for
Direct FET Control
3-State PWM Input for Power Stage Shutdown
UVLO Housekeeping Circuit
Rated from –40°C to +125°C Junction
Temperature
Pb (Lead) Free Packaging
APPLICATIONS
•
•
Digitally-Controlled Synchronous-Buck Power
Stages for Single- and Multi-Phase
Applications
Digitally-Controlled Power Modules
DESCRIPTION
The UCD7231 high current driver is specifically
designed
for
digitally-controlled,
point-of-load,
synchronous buck switching power supplies. Two
driver circuits provide high charge and discharge
current for the high-side NMOS switch and the
low-side NMOS synchronous rectifier in a
synchronous buck circuit. The MOSFET gates are
driven to +6 V by an internally regulated VGG supply.
The internal VGG regulator can be disabled to permit
the user to supply their own gate drive voltage. This
flexibility allows a wide power conversion input
voltage range of 2.2 to 15 V. Internal under voltage
lockout (UVLO) logic insures VGG is good before
allowing chip operation.
A drive logic block allows operation in one of two
modes selected by the SRE Mode pin. In
Synchronous Mode, the logic block uses the PWM
signal to control both the high-side and low-side gate
drive signals. Dead time is automatically adjusted to
prevent cross conduction. The Synchronous Rectifier
Enable (SRE) pin controls whether or not the low-side
FET is turned on when the PWM signal is low. In
Independent Mode, the PWM and SRE pins control
the high-side and low-side gates directly. No
anti-cross-conduction logic is used in this mode.
On-board comparators monitor the voltage across the
high side switch and the voltage across an external
current sense element to safeguard the power stage
from sudden high current loads. Blanking delay is set
for the high side comparator by a single resistor in
order to avoid false reports coincident with switching
edge noise. In the event of a high-side fault or an
over-current fault, the high-side FET turned off and
the Fault Flag (FLT) is asserted to alert the digital
controller. The fault thresholds are independently set
by the HS Sense and ILIM pins.
Output current is measured and monitored by a
precision, high gain, switched capacitor differential
amplifier that processes the voltage present across
an external current sense element. The amplified
signal is available for use by the digital controller on
the IMON pin. The current sense amplifier has output
offset of 0.5 V so that both positive (sourcing) and
negative (sinking) current can be sensed.
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2009–2010, Texas Instruments Incorporated
UCD7231
SLUS997 – NOVEMBER 2009 – REVISED JANUARY 2010
www.ti.com
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
DESCRIPTION (CONTINUED)
An on-chip temperature sense monitors the die temperature. If it exceeds approximately 165°C, the temperature
sensor will initiate a thermal shutdown that halts output switching and sets the FLT flag. The temperature fault
automatically clears when the die temperatures falls by approximately 20°.
FUNCTIONAL BLOCK DIAGRAM
9
VGG DIS
4
3
2
Vin
HS Sense
Bias + VGG
Generator
13
10
16
BP3
UCD7231
PWM
BST
SRE
1
18
UVLO
SRE Mode
HS Gate
19
Digital Control
FLT
HS Fault
SW
RDLY
11
TSD
VGG
OC Fault
Blanking
Control
LS Gate
20
17
15
Thermal
Sense
PGND
5
ILIM
CSP
0.5 V
6
IMON
G = 50
12
AGND
PP
CSN
14
8
7
PAD
Figure 1. UCD7231 Block Diagram
2
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UCD7231
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SLUS997 – NOVEMBER 2009 – REVISED JANUARY 2010
SIMPLIFIED APPLICATION DIAGRAM
Vin
16
Vin
10 PWM
HS Sense 1
From
Controller
BST 18
4 SRE
HS Gate 19
2 FLT
To
Controller
Vout
SW 20
6 IMON
UCD7231
VGG 17
3 SRE Mode
LS Gate 15
9 BP3
GND
PGND 14
5 ILIM
11 RDLY
CSP 8
13 VGG DIS
AGND
12
CSN 7
PAD
PP
Figure 2. Typical Synchronous Buck Power Stage
SW
HS Gate
BST
VG G
Vin
CONNECTION DIAGRAM
20
19
18
17
16
HS Sense 1
15 LS Gate
FLT 2
14 PGND
UCD7231
(QFN - RTJ)
(4x4, 0.50)
SRE Mode 3
13 VGG DIS
6
7
8
9
10
BP3
PWM
11 RDLY
CSP
ILIM 5
CSN
12 AGND
IMO N
SRE 4
ORDERING INFORMATION
TEMPERATURE RANGE
–40°C to +125°C
PACKAGE
Plastic QFN-20 (RTJ)
TAPE AND REEL QTY
PART NUMBER
250
UCD7231RTJT
2500
UCD7231RTJR
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3
UCD7231
SLUS997 – NOVEMBER 2009 – REVISED JANUARY 2010
www.ti.com
ABSOLUTE MAXIMUM RATINGS (1)
over operating free-air temperature range (unless otherwise noted)
SYMBOL
VALUE
UNIT
VIN
Supply voltage
PARAMETER
Vin
–0.3 to 16
V
VBST
Bootstrap voltage
VBST DC
–0.3 to 23
V
VBST Pulse (VSW@20V < 400ns)
–0.3 to 27
VBST Pulse (VSW@22V < 64ns)
–0.3 to 29
VGG
Gate drive supply voltage
Vo
Output gate drive voltage
VSW
–0.3 to 37
VGG (Externally supplied)
–0.3 to 7.0
HS Gate – SW
Switch node voltage
V
–0.3 to 7.0
LS Gate
PGND – 0.3 to VGG+0.3
VSW DC
–1 to 16
VSW Pulse < 400 ns, E = 20 µJ
–2 to 20
VSW Pulse < 64 ns
–5 to 22
V
V
VSW Pulse < 16 ns
–10 to 30
CSP, CSN, RDLY
–0.3 to 5.6
ILIM
–0.3 to 3.6
HS Sense
–0.3 to 16
PWM, SRE, SRE Mode
–0.3 to 5.6
VGG DIS
–0.3 to 3.6
Analog outputs
IMON
–0.3 to 3.6
V
Digital outputs
FLT
–0.3 to 3.6
V
Human body model
HBM
2000
Charged device model
CDM
500
Analog inputs
Digital inputs
ESD Rating
VBST Pulse (VSW@30V < 16ns)
V
V
V
TA
Operating ambient temperature
–40 to 125
°C
TJ
Operating junction temperature
–40 to 150
°C
TSTG
Storage temperature
–65 to 150
°C
300
°C
Lead temperature
(1)
Soldering, 10 seconds
Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only and functional operation of the device at these or any other condition beyond those indicated is not implied. Exposure to
absolute-maximum-rated conditions for extended periods may affect device reliability. All voltages are with respect to AGND. Currents
are positive into, negative out of the specified terminal. Consult company packaging information for thermal limitations and
considerations of packages.
DISSIPATION RATINGS (1)
(1)
PACKAGE
TA ≤ 40°C
POWER RATING
DERATING FACTOR
TA > 40°C
qJA
RTJ
1.89 W
22 mW/°C
45°C/W
Based on a thermal resistance of 45°C/W using JEDEC High-K board and the exposed die pad is connected to a Cu pad on the board.
This pad is connected to the ground plane by a 3x3 via matrix.
RECOMMENDED OPERATING CONDITIONS
over operating free-air temperature range (unless otherwise noted)
MIN
TYP
MAX
UNIT
VIN
Power Input Voltage (Internally generated VGG)
4.7
12.0
15.0
V
VIN
Power Input Voltage (Externally supplied VGG)
2.2
–
15.0
V
VGG
Externally supplied gate drive voltage
4.5
6.0
6.5
V
TJ
Operating junction temperature range
–40
–
125
°C
4
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ELECTRICAL CHARACTERISTICS
VIN = 12V, 4.7 µF from VGG to PGND, 1.0 µF from BP3 to AGND, 0.22 µF from BST to SW, TA = TJ = –40°C to +125°C,
RDLY = 8.06kΩ, SRE Mode = 3.3V, VGG DIS tied to AGND (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
SUPPLY SECTION
Supply current
Outputs not switching, VIN = 5 V, PWM = LOW
8
10
mA
Supply current
Outputs not switching, VIN = 15 V, PWM = LOW
8
10
mA
4.4
4.5
V
GATE DRIVE UNDER-VOLTAGE LOCKOUT
VGG
UVLO OFF
VGG rising
VGG
UVLO ON
VGG falling
VGG
UVLO hysteresis
4.1
4.3
V
80
mV
VGG SUPPLY GENERATOR
VGG
VIN > 7 V, I_VGG < 100 mA
Dropout
6.0
6.4
VIN > 7 V, I_VGG < 100 mA
7.0
V
200
mV
2.0
V
DIGITAL INPUT SIGNALS (PWM, SRE)
VIH_PWM
Positive-going input threshold voltage
VIL_PWM
Negative-going input threshold voltage
PWM
Input voltage hysteresis, (VIH – VIL)
VIH_SRE
Positive-going input threshold voltage
VIL_SRE
Negative-going input threshold voltage
SRE
Input voltage hysteresis, (VIH – VIL)
1.8
0.80
ISRE
V
0.90
1.5
0.9
VPWM = 5.0 V
IPWM
0.90
Input current
V
1.00
V
0.45
V
140
VPWM = 3.3 V
Input current
V
1.7
70
VPWM = 0 V
–63
VSRE = 5.0 V
190
VSRE = 3.3 V
µA
12
VSRE = 0 V
µA
–330
tHLD_R
3-state hold-off time
VPWM transition from 0 V to 1.65 V,
Time until VLS Gate falls to 0 V
450
600
750
ns
tHLD_R
3-state recovery time
VPWM transition from 1.65 V to 0 V,
Time until VLS Gate rises to VGG
150
330
500
ns
Tmin
PWM minimum pulse to force HS gate pulse
CL = 3 nF at HS gate, VPWM = 3.3 V
PWM frequency
QgHS + QgLS < 46 nC, VGG = 6.4 V
50
ns
2
MHz
OUTPUT CURRENT LIMIT (ILIM)
ILIM Input impedance
250
ILIM set point range
0.5
2.7
kΩ
3
3.3
V
FLT output high level
Iload = –2 mA
V
FLT output low level
Iload = 2 mA
0.1
0.6
V
tFAULT_HS
Fault detection time. Delay until HS Gate
falling.
V(ILIM) = 1.50 V, (CSP – CSN) = 20.0 mV,
CSN = 1.80 V
100
150
ns
tFAULT_LS
Fault detection time. Delay until LS Gate
rising.
V(ILIM) = 1.50 V, (CSP – CSN) = 20.0 mV,
CSN = 1.80 V
150
200
ns
tFAULT_FLT
Fault detection time. Delay until FLT
asserted
V(ILIM) = 1.50 V, (CSP – CSN) = 20.0 mV,
CSN = 1.80 V
85
170
ns
Propagation delay from PWM to reset FLT
PWM falling to FLT falling after a current limit event is
cleared. PWM pulse width ≥100 ns.
85
200
ns
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UCD7231
SLUS997 – NOVEMBER 2009 – REVISED JANUARY 2010
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ELECTRICAL CHARACTERISTICS (continued)
VIN = 12V, 4.7 µF from VGG to PGND, 1.0 µF from BP3 to AGND, 0.22 µF from BST to SW, TA = TJ = –40°C to +125°C,
RDLY = 8.06kΩ, SRE Mode = 3.3V, VGG DIS tied to AGND (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
CURRENT SENSE BLANKING (RDLY, HS Sense)
IRDLY
RDLY source current
8.06 kΩ resistor from RDLY to AGND
RDLY resistance range
80
90
100
µA
7.5
8.06
10
kΩ
80
100
120
ns
tBLANK
HS blanking time
RDLY = 8.06 kΩ. From SW rising to HS fault comparator
enabled
IHS Sense
HS Sense sink current
RHS
100
µA
tHSFAULT_HS
HS fault detection time. Delay after tBLANK
until HS Gate falling
RDLY = 8.06 kΩ, RHS Sense = 2.0 kΩ to VIN, VIN = 12 V,
VIN – VSW = 220 mV
20
ns
tHSFAULT_LS
HS fault detection time. Delay after tBLANK
until LS Gate falling
RDLY = 8.06 kΩ, RHS Sense = 2.0 kΩ to VIN, VIN = 12 V,
VIN – VSW = 220 mV
30
ns
Sense
= 2.0 kΩ to VIN, VIN = 12 V
CURRENT SENSE AMPLIFER (IMON, CSP, CSN)
VCM
V(IMON) at no load
CSP = CSN = 1.8 V
Closed loop DC gain
CSP – CSN = 10 mV; 0.5 V ≤ CSN ≤ 3.3 V
Gain with 2.49k resistors in series with CSP, CSN
Input impedance
Differential, CSP – CSN
Input common mode voltage range
VCM(max) is limited to (VGG – 1.2 V)
0.490
0.500
0.510
V
48.5
50
51.5
V/V
45.3
47.6
49.6
V/V
100
–0.3
V(IMON)MIN
CSP = 1.2 V; CSN = 1.3 V; I(IMON) = –250 µA
V(IMON)MAX
CSP = 1.3 V; CSN = 1.2 V; I(IMON) = 500 µA
3
Sampling Rate
kΩ
5.6
V
0.1
0.15
V
3.2
3.3
5
V
Msps
LOW-SIDE OUTPUT DRIVER (LS Gate)
Peak Source Current (1)
Peak Sink Current
(1)
VGG = 6.2 V, PWM = Low, LS Gate = 3 V
6
VGG = 6.2 V, PWM = High, LS Gate = 3 V
6
tRL
Rise Time
CL = 6 nF, VIN = 12 V, VGG = 6.2 V
30
tFL
Fall Time
CL = 6 nF, VIN = 12 V, VGG = 6.2 V
20
Output with VGG <UVLO
VGG = 1.0 V, Isink = 10 mA
Propagation Delay from PWM to LS Gate
CL = 3 nF, PWM falling SW = 0 V, VGG = 6.2 V
0
A
A
ns
ns
0.5
V
46
ns
HIGH-SIDE OUTPUT DRIVER (HS Gate)
Source current (1)
VIN = 12 V, BST = 6.2 V, PWM = High,
HS Gate = 3 V
4
A
Sink current (1)
VIN = 12 V, BST = 6.2 V, PWM = Low,
HS Gate = 3 V
4
A
tRH
Rise time
CL = 3 nF HS Gate to SW, VGG = 6.2 V
27
ns
tFH
Fall time
CL = 3 nF HS Gate to SW, VGG = 6.2 V
21
ns
Propagation delay from PWM to HS Gate
CL = 3 nF HS Gate to SW, PWM rising,
SW = 0 V, VGG = 6.2 V
50
ns
SWITCHING TIME
tDLH
HS gate turn-off propagation delay
CL = 3 nF
16
ns
tDLL
LS gate turn-off propagation delay
CL = 3 nF
15
ns
tDTH
Dead time LS; Gate off to HS; Gate on
CL = 3 nF
12
ns
tDTL
Dead time HS; Gate off to LS; Gate on
CL = 3 nF
15
ns
Forward bias current 100 mA
0.4
V
BOOTSTRAP DIODE
VF
Forward voltage
THERMAL SHUTDOWN
Rising threshold (1)
155
165
175
°C
Falling threshold (1)
135
145
155
°C
Hysteresis
(1)
6
20
°C
As designed and characterized. Not 100% tested in production.
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PIN FUNCTIONS
PIN
QFN-20
NAME
I/O
FUNCTION
1
HS Sense
I
High-side current fault threshold set pin. A resistor is connected from this pin directly to the drain of the
high-side FET. The voltage drop across this resistor sets the maximum voltage drop allowed across the
high-side FET after the blanking time set by RDLY. Exceeding this threshold will assert FLT and truncate
the HS Gate pulse. This pin sinks a constant 100µA of current.
2
FLT
O
Fault Flag. The FLT signal is a 3.3v digital output which is asserted high when an over-current,
over-temperature, or UVLO fault is detected. After an over-current event is detected, the flag is reset low on
the falling edge of the next PWM pin, provided the over-current condition is no longer detected during the
on-time of the PWM signal. For UVLO and over-temperature faults, the flag is reset when the fault condition
is no longer present.
3
SRE Mode
I
Synchronous Rectifier Enable Mode select pin. When high, the high-side and low-side gate drive timing is
controlled by the PWM pin. Anti-cross-conduction logic prevents simultaneous application of high-side and
low-side gate drive. When low, independent operation of the high-side and low-side gate is selected. The
high-side gate is directly controlled by the PWM signal. The low-side gate is directly controlled by the SRE
signal. No anti-cross-conduction circuitry is active in this mode. This pin should not be left floating.
4
SRE
I
Synchronous Rectifier Enable or Low-Side Input. This pin is a digital input capable of accepting 3.3V or 5V
logic level signals. A Schmitt trigger input comparator desensitizes this pin from external noise. When SRE
Mode is high, this signal, when low, disables the synchronous rectifier FET. The LS Gate signal is held off.
When SRE Mode is high, this signal, when high, allows the LS Gate signal to function according to the state
of the PWM pin. When SRE Mode is low, this pin is a direct input to the LS Gate driver.
5
ILIM
I
Output current limit threshold set pin. The voltage on this pin sets the fault threshold voltage on the IMON pin.
The nominal threshold voltage range is 0.5 V to 3.0 V. When V(IMON) exceeds V(ILIM), the FLT pin is
asserted and the HS Gate pulse is truncated.
6
IMON
O
Current Sense Linear Amplifier Output. The output voltage level on this pin represents the average output
current. V(IMON) = 0.5 V + 48(V(CSP) – V(CSN)).
7
CSN
I
Inverting input of the output current sense amplifier and current limit comparator.
8
CSP
I
Non-inverting input of the output current sense amplifier and current limit comparator.
9
BP3
O
Bypass capacitor for internal 3.3V supply. Connect a 1µF (minimum) ceramic capacitor from this pin to
AGND.
10
PWM
I
PWM input. This pin is a digital input capable of accepting 3.3 V or 5 V logic level signals. A Schmitt trigger
input comparator desensitizes this pin from external noise. When SRE Mode is high, this pin controls both
gate drivers. When SRE Mode is low, this pin only controls the high-side driver. This pin can detect when
the input drive signal has switched to a high impedance (3-state) mode. When the high impedance mode is
detected, both the HS Gate and LS Gate signals are held low.
11
RDLY
I
Requires a resistor to AGND for setting the Current Sense blanking time for the high-side current sense
comparator and output current limit circuitry.
12
AGND
–
Analog ground return for all circuits except the LS Gate driver.
13
VGG DIS
I
VGG Disable pin. When pulled high, the on-chip VGG linear regulator is disabled. When disabled, an
externally supplied gate voltage must be connected to the VGG pin. Connect this pin to AGND to use the
on-chip regulator.
14
PGND
–
Power Ground pin. This pin provides a return path for the low-side gate driver.
15
LS Gate
O
The Low-Side high-current driver output. Drives the gate of the low-side synchronous MOSFET between
VGG and PGND.
16
Vin
I
Input Voltage to the buck power stage and driver circuitry
17
VGG
I/O Gate Drive voltage supply. When VGG DIS is low, VGG is generated by an on-chip linear regulator. Nominal
output voltage is 6.4 V. When VGG DIS is high, an externally supplied gate voltage can be applied to this
pin. Connect a 4.7 µF capacitor from this pin to PGND.
18
BST
I/O Floating bootstrap supply for high side driver. Connect the bootstrap capacitor between this pin and the SW
node. The bootstrap capacitor provides the charge to turn on the high-side MOSFET.
19
HS Gate
O
20
SW
I/O Switching node connection to buck inductor. This pin provides a return path for the high-side gate driver.
PP
PAD
–
The High-Side high-current driver output. Drives the gate of the high side buck MOSFET between BST and
SW.
Power Pad. Connect directly to AGND for better thermal performance and EMI reduction.
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DETAILED DESCRIPTION
GENERAL
The UCD7231 is designed primarily to be a synchronous buck driver with current measurement and fault
detection capabilities that make it an ideal partner with digital power controllers. This device incorporates two
high-current gate drive stages and sophisticated current measurement circuitry that allows for the monitoring and
reporting of output load current. Two separate fault detection blocks protect the power stage from excessive load
current or short circuits. On-chip thermal shutdown protects the device in case of severe over-temperature
conditions. Detected faults immediately truncate the power conversion cycle in progress, without controller
intervention, and assert a digital fault flag (FLT). Gate drive voltage is supplied by an on-chip linear regulator. If
desired, this regulator can be disabled and an external gate drive voltage can be supplied. Mode selection pins
allow the device to be used in synchronous mode or independent mode. In synchronous mode, the high-side and
low-side gate timing is controlled by a single PWM input. Anti-cross-conduction dead-time intervals are applied
automatically to the gate drives. In independent mode, the high-side and low-side gate drive signals are
controlled directly by the PWM and SRE pins. The automatic dead-time logic is disabled in this mode. When
operating in synchronous mode, the use of the low-side FET can be disabled under the control of the SRE pin.
This feature facilitates start-up into a pre-bias voltage and is also used in some applications to reduce power
consumption at light loads.
PWM INPUT
The PWM input pin accepts the digital signal from the controller that represents the desired high-side FET
on-time duration. This input is designed to accept 3.3V logic levels, but is also tolerant of 5V input levels. The
SRE Mode pin sets the behavior of the PWM pin. When the SRE Mode pin is asserted high, the device is placed
in synchronous mode. In this mode, the timing duration of the high-side gate drive and the low-side gate drive
are both controlled PWM input signal. When PWM is high, the high-side gate drive (HS Gate) is on and the
low-side gate drive (LS Gate) is off. When PWM is low, the high-side gate drive is off and the low-side gate drive
is on. Automatic anti-cross-conduction logic monitors the gate to source voltage of the FETs to verify that the
proper FET is turned off before the other FET is turned on. When the SRE Mode pin is asserted low, the device
is placed in independent mode. In this mode the PWM input only controls the high-side gate drive. When PWM is
high, the high-side gate drive is on. The low side FET in independent mode is directly controlled by the SRE pin.
No anti-cross-conduction logic is active in independent mode. The user must insure that the PWM and SRE
signals do not overlap.
The PWM input supports a 3-state detection feature. It can detect if the PWM input signal has entered a 3-state
mode. When 3-state mode is detected, both the high-side and low-side gate drive signals are held off. To support
this mode, the PMW input pin has an internal pull-up resistor of approximately 50kΩ to 3.3V. It also has a 50kΩ
pull-down resistor to ground. During normal operation, the PWM input signal swings below 0.8V and above 2.5V.
If the source driving the PWM pin enters a 3-state or high impedance state, the internal pull-up/pull-down
resistors will tend to pull the voltage on the PWM pin to 1.65V. If the voltage on the PWM pin remains within the
0.8V to 2.5V 3-state detection band for longer than the tHLD_R 3-state detection hold-off time, then the device
enters 3-state mode and turns both gate drives off. This behavior occurs regardless of the state of the SRE Mode
and SRE pins. When exiting 3-state mode, PWM should first be asserted low. This will insure that the bootstrap
capacitor is recharged before attempting to turn on the high-side FET.
The logic threshold of this pin typically exhibits 900mV of hysteresis to provide noise immunity and insure
glitch-free operation of the gate drivers.
SRE INPUT
The SRE (Synchronous Rectifier Enable) pin is a digital input with an internal 10kΩ pull-up resistor to 3.3V. It is
designed to accept 3.3V logic levels, but is also tolerant of 5V levels. The SRE Mode pin sets the behavior of the
SRE pin. When the SRE Mode pin is asserted high, the device is placed in synchronous mode. In this mode, the
input, when asserted high, enables the operation of the low-side synchronous rectifier FET. The state of the
low-side gate drive signal is governed by the PWM input. When SRE is asserted low while in synchronous mode,
the low-side FET gate drive is continuously held low, keeping the FET off. While held off, current flow in the
low-side FET is restricted to its intrinsic body diode. When the SRE Mode pin is asserted low, the device is
placed in independent mode. In this mode, the state of the low-side gate drive signal follows the state of the SRE
signal. It is completely independent of the state of the PWM signal. No anti-cross-conduction logic is active in
independent mode. The user must insure that the PWM and SRE signals do not overlap.
8
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The logic threshold of this pin typically exhibits 450mV of hysteresis to provide noise immunity and insure
glitch-free operation of the low-side gate driver.
SRE MODE
The SRE Mode pin is a digital input designed to accept 3.3V logic levels, but is also tolerant of levels up to 5V.
This pin sets the operational mode on the device. When asserted high, the device will be placed in synchronous
mode. In this mode the behavior of both the high-side and low-side gate drive signals are under the control of the
PWM input. When asserted low, this pin configures the device for independent mode. In this mode the high-side
FET is under the control of the PWM pin. The low-side FET is under the control of the SRE pin. The SRE Mode
pin is designed to be permanently tied high or low depending on the power architecture being implemented. It is
not intended to be switched dynamically while the device is in operation. This pin can be tied to the BP3 pin to
always select synchronous mode.
Vin
Vin supplies power to the internal circuits of the device. The input power is conditioned by an internal linear
regulator that provides the VGG gate drive voltage. A second regulator that operates off of the VGG rail produces
an internal 3.3V supply that powers the internal analog and digital functional blocks. The BP3 pin provides
access for a high frequency bypass capacitor on this internal rail. The VGG regulator produces a nominal output
of 6.4V. The output of the VGG regulator is monitored by the Under-Voltage Lock-Out (UVLO) circuitry. The
device will not attempt to produce gate drive pulses until the VGG voltage is above the UVLO threshold. This
insures that there is sufficient voltage available to drive the power FETs into saturation when switching activity
begins. To use the internal VGG regulator, the voltage on Vin should be at least 4.7V.
When performing power conversion with less than 4.7V on the Vin pin, the gate drive voltage must be supplied
externally. (See VGG and VGG DIS sections for details.)
VGG
The VGG pin is the gate drive voltage for the high current gate drivers stages. The voltage on this pin can be
supplied internally by the on-chip regulator, or it can be externally supplied by the user. When using the internal
regulator, the VGG DIS pin should be tied low. When an external source of VGG is to be used, the VGG DIS pin
must be tied high. Current is drawn from the VGG supply in fast, high-current pulses. A 4.7µF ceramic capacitor
should be connected from the VGG pin to the PGND pin as close as possible to the package.
Whether internally or externally supplied, the voltage on the VGG pin is monitored by the UVLO circuitry. The
voltage must be higher than the UVLO threshold before power conversion can occur. Note that the FLT pin is
asserted high when VGG is below the UVLO threshold.
The average current drawn from the VGG supply is dependant on the switching frequency and the total gate
charge of the power FETs connected to the driver. This current can be significant and is a major contributor to
the overall power dissipation of the driver. The total gate charge (Qg) is a function of the value of VGG and the
power FET construction. A value for Qg can be obtained from the FET manufacturer’s data sheet. A graph of Qg
vs VGS is usually supplied. Use the value of VGG as the VGS value and read the corresponding value of Qg. A
value of Qg should be obtained for both the high-side and low-side FETs.
To keep the current draw from the VGG supply within its capability, the switching frequency of the power stage
should be limited to the following:
92000
Fsw(max) =
Qg HS + QgLS
(1)
Where Fsw(max) is the maximum switching frequency in kHz, QgHS is the gate charge of the high-side FET
measured at VGS = 6.4V, and QgLS is the total gate charge of the low-side FET(s) measured at VGS = 6.4V, both
specified in nanocoulombs (nC). Selecting FETs with lower gate charge will permit higher operating frequencies.
The formula above allows for a maximum of 92mA of total gate drive current. An additional 8mA is consumed by
the remaining circuitry within the device.
The average gate drive current, in mA, can be calculated from the following equation (with switching frequency in
kHz and charge in nC):
IGATE_AVE = (QgHS + QgLS) × Fsw × 1000
(2)
(2)
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Assuming VGG = 6.4V and Fsw = 500kHz, a typical Qg for a low-side FET is 50nC. A typical high-side FET Qg is
13nC. This combination creates an IGATE_AVE of 31.5mA. If the switching frequency was doubled, the current
draw would double to 63mA. If VIN = 12V and the internal VGG linear regulator is being used, the power
dissipation in the VGG regulator, for this case, at 500kHz operation, is 176mW. At 1MHz, it increases to 353mW.
Keep in mind that this is not the total power dissipation of the driver, only the portion dissipated in the VGG
regulator. Good thermal layout techniques are required for this device.
VGG DIS
This pin, when asserted high, disables the on-chip VGG linear regulator. When tied low, the VGG linear regulator is
used to derive VGG from VIN. This pin is designed to be permanently tied high or low depending on the power
architecture being implemented. It is not intended to be switched dynamically while the device is in operation.
SW
The SW pin connects to the switching node of the power conversion stage. It acts as the return path for the
high-side gate driver. When configured as a synchronous buck stage, the voltage swing on SW normally
traverses from below ground to well above VIN. A power Schottky diode should be connected from this pin to
PGND to clamp the negative voltage swing on this pin to less than 1V. A series 1Ω resistor connects this pin to
the actual switching node. It acts as a current limiting resistor when the Schottky diode is clamping negative
voltage swings. The diode should be rated for at least 0.5A of current and exhibit a breakdown voltage of at least
30V. Small-signal Schottky diodes should not be used.
Parasitic inductance in the high-side FET and the output capacitance (Coss) of both power FETs form a resonant
circuit that can produce high frequency (>100MHz) ringing on this node. The voltage peak of this ringing, if not
controlled, can exceed twice VIN. Care must be taken to not allow the peak ringing amplitude to exceed twice the
value of the input voltage, even if that voltage amplitude is within the Absolute Maximum rating limit for the pin. In
many cases, a series resistor and capacitor snubber network connected from the switching node to PGND can
be helpful in damping the ringing and decreasing the peak amplitude. It is recommended that provisions for
snubber network components be provided during the layout of the printed circuit board. If testing reveals that the
ringing amplitude at the SW pin exceeds twice VIN, then the snubber components need to be populated.
BST
The BST pin provides the drive voltage for the high-side FET. A bootstrap capacitor is connected from this pin to
the SW node. Internally, a diode connects the BST pin to the VGG supply. In normal operation, when the
high-side FET is off and the low-side FET is on, the SW node is pulled to ground and, thus, holds one side of the
bootstrap capacitor at ground potential. The other side of the bootstrap capacitor is clamped by the internal diode
to VGG. The voltage across the bootstrap capacitor at this point is the magnitude of the gate drive voltage
available to switch-on the high-side FET. The bootstrap capacitor should be a low ESR ceramic type, with a
recommended minimum value of 0.22µF. A minimum voltage rating of 16V or higher is recommended.
HS GATE
The HS Gate signal directly drives the gate of the high-side power FET. It provides high current drive to charge
the gate capacitance of the FET rapidly to insure that it makes the transition from off to on as quickly as possible
to minimize switching losses. When commanded on, the HS Gate is driven to the BST pin potential. As the FET
begins to turn on, the SW will quickly rise to the VIN potential. This voltage swing is coupled by the bootstrap
capacitor to the BST pin. The net result is that the BST pin voltage, and thus the HS Gate voltage, is always
equal to VSW + VGG. As the FET gate charges, the current return path for the driver is provided by the SW pin.
When the HS Gate is commanded off, the driver pulls the pin to the SW potential. As the FET turns off, the SW
pin will swing quickly to slightly below ground. Once again, this voltage swing is coupled to the BST pin by the
bootstrap cap. The HS Gate circuitry is referenced to the SW pin and floats with the SW signal swing. The
circuitry loop from the HS Gate pin to the gate of the FET and from the source of the high-side FET to the SW
pin should kept as small and tight as possible to limit stray inductance. Likewise, the loop from the BST pin to the
bootstrap capacitor and back to the SW pin should be kept small and tight.
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LS GATE
The LS Gate signal directly drives the gate of the low-side power FET. It provides high current drive to quickly
charge the gate capacitance of the FET, which is often considerably larger than the high-side FET. When
commanded on, the LS Gate is driven to the VGG pin potential. The current return path for the driver is provided
by the PGND pin. When commanded off, the LS Gate pin is driven to the PGND potential. The traces from the
LS Gate and the PGND pins to the low-side FET gate and source pins should be short and wide to minimize
parasitic inductance and resistance.
CSP, CSN
These pins are the input to the differential current sense amplifier. The Current Sense Positive (CSP) pin
connects to the non-inverting input, the Current Sense Negative (CSN) connects to the inverting input. This
amplifier provides the means to monitor and measure the output current of the power stage. The circuitry can be
used with a discrete, low value, series current sense resistor, or can make use of the popular inductor DCR
sense method.
The DCR method is illustrated in Figure 3. A series resistor and capacitor network is added across the buck
stage power inductor. It can be shown that when the value of L/DCR is equal to RC, then the voltage developed
across the capacitor, C, is a replica of the voltage waveform the ideal current would induce in the dc resistance
(DCR) of the inductor. This method does not detect changes in current due to changes in inductance value
caused by saturation effects. The value used for C should be in the 0.1µF to 2.2µF range. This keeps the
impedance of the sense network low, which reduces its susceptibility to noise pickup from the switching node.
The trace lengths of the CSP and CSN signals should be kept short and parallel. To aid in rejection of high
frequency common-mode noise, a series 2.49k resistor should be added to both the CSP and CSN signal paths,
with the resistors being placed close to the pins at the package. This small amount of additional resistance
slightly lowers the current sense gain.
Power inductors are selected for the lowest possible DCR to minimize losses. Typical DCR values range from
0.5mΩ to 5mΩ. With a load current of 20A, the voltage presented across the CSP and CSN pins is only in the
range of 10mV to 100mV. Keep in mind that this small differential signal is riding on a large common mode signal
that is the dc output voltage. This makes the current sense signal challenging to process.
L
DCR
SW
Vout
C
R
2.49kW
CSP
2.49kW
CSN
Figure 3. DCR Current Sense
The UCD7231 uses switched capacitor technology to perform the differential to single-ended conversion of the
sensed current signal. This technique offers excellent common mode rejection. The differential CSP-CSN signal
is amplified by a factor of 48 and then a fixed 500mV pedestal voltage is added to the result. This signal is
presented to the IMON pin.
When using inductors with DCR values of 2.0mΩ or higher, it may be necessary to attenuate the input signal to
prevent saturation of the current sense amplifier. This is easily accomplished through the addition of resistor R2
as shown in Figure 4.
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L
DCR
SW
Vout
C
R1
R2
2.49 kW
CSP
2.49 kW
CSN
Figure 4. Attenuating the DCR Sense Signal
The amount of attenuation is equal to R2/(R1 + R2). The equivalent resistance value to use in the L/DCR = RC
formula is the parallel combination of R1 and R2. Thus, when using the circuit of Figure 4, L/DCR =
C×R1×R2/(R1+R2).
IMON
The IMON signal is a voltage proportional to the output current delivered by the power stage. The voltage
magnitude obeys the following equation when using the circuit of Figure 3. This equation takes into account the
gain reduction caused by the series 2.49k resistors.
V(Iout) = 0.5 + 48 × DCR × Iload
(3)
(3)
If the calculated value of V(IMON) exceeds the range of the analog-to-digital converter (ADC) or, if used, the
maximum fault comparator threshold limit of a controller monitoring this voltage, then the circuit of Figure 4
should be used. When using the circuit of Figure 4, the voltage on IMON obeys this modified equation:
æ R2 ö
V(Iout) = 0.5 + 48 ´ DCR ´ Iload ´ ç
÷
è R1 + R2 ø
(4)
In either case, the output voltage is 500mV at no load. Current that is sourced to the load causes the IMON
voltage to rise above 500mV. Current that is forced into the power stage (sinking current) is considered
“negative” current and will cause the IMON voltage to fall below 500mV. The usable dynamic range of the IMON
signal is approximately 100mV to 3.1V. Keep in mind that this signal swing could exceed not just the maximum
range of an analog to digital converter (ADC) that may be used to read or monitor the IMON signal, but also the
maximum programmable limit for the fault OC threshold. For example, the UCD92xx family of digital controllers
has maximum limit of 2.5V for the ADC converter and 2.0V for the fault OC threshold, even though the input pin
can tolerate voltages up to 3.3V.
The IMON voltage is internally fed to the non-inverting input of the output over-current fault comparator. Good
practice dictates that the over-current threshold should be set at approximately 150% of the rated power stage
output current plus one half of the peak-to-peak inductor ripple current. This mandates that the IMON signal should
remain within its linear dynamic range at this threshold load current level. This requirement may force the use of
the attenuation circuit of Figure 4. Note that the IMON voltage (that goes to the output over-current fault
comparator) is held during the blanking interval set by the resistor on the RDLY pin. This means that the IMON pin
will not reflect output current changes during the blanking interval, and that a fault will not be flagged until the
blanking interval terminates.
ILIM
The ILIM pin feeds the inverting input of the output over-current fault comparator. The voltage applied to this pin
sets the over-current fault threshold. When the voltage on the IMON pin exceeds the voltage on this pin, a fault is
flagged. The voltage on this pin can be set by a voltage divider, a DAC, or by a filtered PWM output. The usable
voltage range of the ILIM pin is approximately 0.6V to 3.1V. This represents the linear range of the IMON signal for
sourced output current. When using a voltage divider to set the threshold, a (0.01µF) capacitor to BP3 can be
added to improve noise immunity.
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RDLY
The RDLY pin sets the blanking time of the high-side fault detection comparator. A resistor to AGND sets the
blanking time according to the following formula, where tBLANK is in nanoseconds and RDLY is in kΩ. Values of
RDLY of greater than 25kΩ should not be used.
RDLY = 0.11 × (tBLANK – 27)
(5)
(5)
To calculate the nominal blanking time for a given value of resistance, use the formula below.
tBLANK = 9.13 × RDLY + 27
(6)
(6)
The blanking interval begins on the rising edge of SW. During the blanking time the high-side fault comparator is
held off. A high-side fault is flagged when the voltage drop across the high-side FET exceeds the threshold set
by the HS Sense pin. Blanking is required because the high amplitude ringing that occurs on the rising edge of
SW would otherwise cause false triggering of the fault comparator. The required amount of blanking time is a
function of the high-side FET, the PCB layout, and whether or not a snubber network is being used. A value of
100ns is a typical starting point. An RDLY of 8.06kΩ will provide 100ns of blanking. The blanking interval should
be kept as short as possible, consistent with reliable fault detection. The blanking interval sets the minimum duty
cycle pulse width where high-side fault detection is possible. When the duty cycle of the PWM pulses are
narrower than the blanking time, the high-side fault detection comparator is held off for the entire on-time and is,
therefore, blind to any high-side faults.
Internally, the RDLY pin is fed by a 90µA current source. When using the default value of 8.06kΩ, the voltage
observed on the RDLY pin will be approximately 725mV.
HS SENSE
A resistor from the HS Sense pin to the drain of the high-side FET sets the high-side fault detection threshold.
When the high-side FET is on, the current flow in the FET produces a voltage drop across the device. The
magnitude of this voltage is equal to the RDS(ON) times the current through the FET. An absolute maximum
current level can be set during the design stage and the resultant voltage drop across the FET can be calculated.
This maximum voltage drop, ΔVMAX, sets the high-side fault threshold.
Internally, a high speed comparator monitors the voltage between the SW pin and the HS Sense pin when the
high-side FET is on. Whenever the voltage on the SW pin is lower than the voltage on the HS Sense pin, a fault
is flagged. To prevent false tripping during the ringing that accompanies the rising edge of SW, the output of the
comparator is held off (blanked) for a time interval set by the RDLY pin. The voltage on the HS Sense pin is set
by a resistor connected from the pin to the high-side FET drain. The HS Sense resistor value is calculated from
the following formula, where ΔVMAX is in mV, and RHS Sense is in kilohms.
RHS Sense = ΔVMAX / 100
(7)
(7)
For example, if ΔVMAX is 100mV, then RHS Sense is 1.0kΩ.
The equation can be restated as follows, with RHS Sense in kilohms, RDS(ON) in milliohms, and IMAX in amps:
RHS Sense = ®DS(ON)HOT × IMAX) / 100
(8)
(8)
As a general rule of thumb, the value of IMAX should be set to about 150% of the expected maximum steady-state
current. This allows some headroom to avoid nuisance fault events due to transient load currents and the
inductor ripple current. Also, keep in mind that the RDS(ON) of a FET has a large positive temperature coefficient
of approximately 4000ppm/°C. The junction temperature of the FET will be elevated when operating at currents
near the IMAX threshold. In the equation above, use a value of RDS(ON)HOT that is approximately 140% of its typical
room temperature value. When using the internal VGG gate drive supply, the FET, when turned on, is driven to a
VGS enhancement voltage of approximately 6V. Most FET data sheets provide RDS(ON) values for VGS values of
4.5V and 10V. Do not use the VGS = 10V value for the room temperature RDS(ON) value. Some manufacturers
provide a graph of RDS(ON) vs VGS. If provided, use the VGS = 6V value for the room temperature RDS(ON) value.
A 100µA current sink pulls current through RHS Sense. This sets up a reference voltage drop equal to ΔVMAX. It is
important to connect the far end of the RHS Sense resistor directly to the drain of the high-side FET. This should be
made with a separate, non-current-carrying trace. This insures that only the RDS(ON) of the FET influences the
fault threshold and not the resistance of the pc board traces.
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FLT
The Fault Flag (FLT) is a digital output pin that is asserted when a significant fault is detected. It is meant to alert
the host controller to an event that has interrupted power conversion. The FLT pin is held low in normal
operation. When a fault is detected it is asserted high (3.3V). There are four events that can trigger the FLT
signal: output over-current, high-side over-current, UVLO and thermal shutdown. The operation of the device
during fault conditions is described in the Fault Behavior section. When asserted in response to an over-current
fault, the FLT signal is reset low upon the falling edge of a subsequent PWM pulse, provided no faults are
detected during the on-time of the pulse. If the fault is still present, the flag will remain asserted. When asserted
in response to an UVLO or thermal shutdown event, the FLT pin will automatically de-assert itself when the
UVLO or thermal event has passed. If the on-time of the PWM pulse is less than 100ns, then more than one
pulse may be required to reset the flag.
BP3
The BP3 pin provides a connection point for a bypass capacitor that quiets the internal 3.3V voltage rail. Connect
a 1µF (or greater) ceramic capacitor from this pin to analog ground. Do not draw current from this pin. It is not
intended to be a significant source of 3.3V. It can, however, be used to as a source of 3.3V for an ILIM voltage
divider and a tie point for the SRE Mode pin. Current draw should be limited to 100µA or less.
FAULT BEHAVIOR
When faults are detected, the device reacts immediately to minimize power dissipation in the FETs and protect
the system. The type of fault influences the behavior of the gate drive signals.
When a thermal shutdown fault occurs, both HS Gate and LS Gate are immediately forced low. They will stay
low, regardless of the state of PWM and SRE, for the duration of the thermal shutdown.
A UVLO fault occurs when the voltage on the VGG pin is less than the UVLO threshold. During this time both the
HS Gate and LS Gate are driven low, regardless of the state of PWM and SRE. The fault is automatically cleared
when the VGG voltage rises above the UVLO threshold.
When either a high-side fault or an output over-current fault is detected, the FLT pin is asserted high, and both
gate signals are immediately pulled low. During a high-side fault, a high-side gate pulse will be issued with each
incoming PWM pulse. If the fault is still present, the HS Gate signal will again be truncated. This behavior
repeats on a cycle-by-cycle basis until the fault is gone or the PWM input is held low. This behavior is illustrated
in Figure 5.
PWM
Fault Detected
FLT
HS
Gate
LS
Gate
Figure 5. High-Side Over-Current Fault Response
When a high-side fault and output over-current fault are detected concurrently, then both FET drives are
immediately turned off and held off. If the output over-current fault is still present at the next PWM rising edge,
then no HS Gate pulse will be issued and both gates will continue to be held off. Unlike the high-side fault
detection circuitry, the output over-current fault circuitry is not reset on a cycle-by-cycle basis. The output current
must fall below the over-current threshold before switching will resume.
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FLT RESET
With the exception of a UVLO fault or a thermal shutdown fault, the FLT flag, once asserted, is cleared by
subsequent PWM pulses. The FLT flag will be cleared on the falling edge of the next PWM pulse, provided a
fault condition is not asserted during the entire on-time of the PWM pulse. If a fault is present or detected during
the on-time interval, the FLT pin will remain asserted. This behavior is illustrated in Figure 6.
PWM
Fault Detected
Internal
Fault
Signal
Fault Still Present
FLT
No fault present during
entire PWM high
interval . FLT reset on
PWM falling edge
Figure 6. FLT Reset Sequence
Whenever the voltage on the VGG pin is below the UVLO falling threshold, as at the time of initial power-up, for
example, the FLT pin will be asserted. When the voltage on the VGG pin rises above the UVLO rising threshold,
the FLT pin will be cleared automatically. This permits the FLT pin to be used as a “Power Not Good” signal at
initial power-up to signify that there is insufficient gate drive voltage available to permit proper power conversion.
When FLT goes low, it is an indication of “Gate Drive Power Good” and power conversion can commence. After
initial power-up, the assertion of the FLT flag should be interpreted that power conversion has stopped or has
been limited by a fault condition.
THERMAL SHUTDOWN
If the junction temperature exceeds approximately 165°C, the device will enter thermal shutdown. This will assert
the FLT pin and both gate drivers will be turned off. When the junction temperature cools by approximately 20°C,
the device will exit thermal shutdown. The FLT flag is reset upon exiting thermal shutdown.
Gate driver temperature will be strongly influenced by the switching frequency being used, the value of VIN and
VGG, and the total capacitive load on the HS Gate and LS Gate pins. The driver junction temperature is not
normally strongly affected by load current. However, a rise in the PCB substrate temperature due to load current
induced power dissipation in nearby components will raise the junction temperature and contribute to a possible
thermal shutdown event.
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APPLICATION INFORMATION
EXAMPLE 20A POWER STAGE
A partial schematic of a 20A power conversion stage designed for 500kHz operation is shown in Figure 7.
Vin
(6V - 14V)
C11
0.1mF
16
Vin HS Sense 1
10 PWM
From
controller
BST 18
4 SRE
HS Gate 19
2 FLT
To
controller
6 I MON
U1
UCD7231
3 SRE Mode
R5
10.0kW
C12
1mF
R6
31.6kW
R7
9 BP3
5 ILIM
SW 20
VGG 17
C3
0.22mF
R9
0W (Opt )
Q1
CSD16322Q5
L1
1mH, 1.2mW
R11
1W
Vout
D1
B0540W
C4
4.7uF
C5
PGND 14
CSP 8
13 VGGDIS
CSN 7
AGND PAD
12
PP
R1
768W R2
2200pF
LS Gate 15
11 RDLY
8.06kW
C1
22mF
C2
22mF
R12
1.65kW
R8
3.01W
0.5W
R10
0 W (Opt)
Opt
C7
C8
C9
47mF 47mF 330mF
GND
Q2
CSD16401Q5
R3
2.49kW
C6
1mF
R4
2.49kW
Figure 7. Example 20A Power Stage
This power stage has been designed to operate with a nominal input voltage of 12V. It will perform well with input
voltages from 6V to 14V. The output voltage range is assumed to be 3.3V or lower. It has been configured to use
the internal VGG supply and operate strictly in synchronous mode. The controller and voltage feedback
components are not shown. This design works well with any of the UCD92xx family of Digital Power Controllers.
The first step in designing the power stage is selecting a nominal operating frequency. Lower switching
frequencies will reduce FET switching losses and driver gate currents, but will require higher inductor values to
keep inductor ripple current within reasonable values. Higher switching frequencies allow for smaller inductor
values, which likely reduces their physical size and DCR, but higher FET switching losses and gate drive power
may offset the efficiency gains achieved from reduced inductor DCR. 500kHz is a good starting point for power
stages in the 15A to 25A range.
INDUCTOR SELECTION
Once a switching frequency has been selected, an appropriate inductance value can now be selected. Ripple
current and saturation current are the two key parameters that drive inductor value selection. Ripple current is
the ac variation of the current through the inductor. It is superimposed on the average dc (load) current flowing
through the inductor. High values of ripple current cause increased core losses in the inductor, and require more
low ESR capacitance to keep the output ripple voltage to acceptable levels. A general rule of thumb is to limit the
inductor ripple current to approximately 30% of the rated dc load current. The peak-to-peak ripple current in an
inductor determined by the voltage across the inductor, the time duration of that applied voltage, and the value of
the inductor.
ΔIPP = VL × Δt / L
(9)
(9)
In a switching regulator, this equation can be rewritten to use the duty-cycle and switching frequency of the
high-side FET to calculate the ripple current.
16
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ΔIPP = [(Vin – Vout) × Vout] / (Vin × FSW × L)
(10)
(10)
For a synchronous buck regulator, the ripple current is highest at 50% duty cycle, or when Vout is one half of
Vin. At higher or lower duty cycles, the ripple current decreases.
For this design, the maximum output current is targeted to be 20A. If the 30% ripple current rule is applied, the
maximum allowable ripple current is 6APP. The previous equation can be rearranged to use this value to compute
a minimum inductance value that will meet our criteria.
LMIN = [(Vin – Vout) × Vout] / (Vin × FSW × ΔIMAX)
(11)
(11)
For this design, the maximum ripple occurs when Vin = 14V and Vout is at the highest targeted output voltage of
3.3V. This produces a value for LMIN of 0.84µH. This value is rounded up to 1µH, which is a very popular value
that is widely available from inductor vendors.
Now that the inductance value has been determined, the current handling capacity of the inductor drives the next
step in the selection process. The inductor saturation limit and DCR heating limit are two key parameters. At full
load, the peak current in the inductor is equal to the load current plus one half of the ΔIPP value. For this design,
the peak inductor current is approximately 23A. The inductor must have a saturation current rating, ISAT, of at
least 23A. The inductor saturation rating is the current level at which the inductance value falls by 20 or 30%
(depending on the vendor) from its no-load value. As current increases above this value, the inductance value
may fall sharply, depending on the core material and construction of the inductor. Operating an inductor in its
saturated region causes the current through it to increase rapidly, causing potentially damaging levels of current
to flow in the high-side FET. Good engineering practice dictates that there be should be 15% or more headroom
in the inductor saturation limit to allow for transient currents and surges that will be encountered in normal
operation. For this design, an ISAT rating of at least 1.15 × 23A = 26.5A would be required.
For highest efficiency, an inductor with the lowest DCR will always have the lowest I2R losses. However, low
resistance requires wire with a large cross section. This forces the inductor to be physically larger than a higher
DCR device. The DCR of the inductor will limit its current handling capacity due to the heating it will cause when
current flows through it. Inductor manufacturers typically give a maximum current rating for an inductor based on
the current that produces a 40°C rise in the device temperature. Keep in mind that in an 85°C ambient
environment, a 40° rise will result in a device temperature of 125°C. Every inductor has two maximum current
ratings: one is the 40°C rise rating, the other is the ISAT rating. The maximum usable current rating for the
inductor is the lower of the two values. In a well designed inductor, the 40°C rise rating and ISAT are
approximately equal. The 40° rise rating should be at least equal to the maximum steady state load current of the
power stage. Headroom above the steady state 40°C rise rating is not required. Momentary surge currents above
the rating value will not cause a significant temperature rise due to the thermal mass of the part.
The last key inductor consideration is the choice of core material. Core material affects cost, power dissipation
due to core loss, and saturation characteristics. There are three popular core materials used in power inductors:
powdered iron, ferrite, and powdered alloy. Powder iron is inexpensive and has a desirable soft saturation
characteristic that makes it very tolerant of surge and transient currents. However, at high values of ripple current
and higher switching frequencies (500kHz and up), core losses become quite large. The heating due to core loss
is in addition to the I2R heating due to the winding DCR. Excessive core loss can cause the core temperature to
rise dramatically. In some cases, this can lead to permanent degradation of the core. Powdered iron cores are
best used at switching frequencies at or below 350kHz. Ferrite has the lowest core losses, making it ideal for
higher switching frequencies. Ferrite saturates easily, so ferrite based inductors are produced with some form of
air gap that lowers their effective permeability and extends their saturation limit. However, once the core reaches
saturation, the falloff in inductance is quite steep. This dictates the selection of a device that has some extra ISAT
headroom to allow for transient current surges. Ferrite is also the most costly core material. Powdered alloy
cores are an improved version of powdered iron cores. By using more exotic metal mixtures in the core, alloy
cores exhibit lower core loss at high frequencies and ripple currents compared to powered iron. In some cases,
they approach the performance of ferrite. The powdered alloy cores retain the desirable soft saturation
characteristic of powdered iron cores. Cost wise, powdered alloy usually falls between powdered iron and ferrite.
Now that the key inductor requirements are known, a device can be selected. In this design, a BI Technologies
HM00-08822LFTR device, for example, meets the requirements. This is a 0.95µH device, with 1.2mΩ DCR. It
uses a ferrite core with an ISAT rating of 29A.
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CALCULATING THE DCR CURRENT SENSE COMPONENTS
With an inductor selected, the next step is to calculate the value of the DCR current sensing components. While
the inductor has a nominal room temperature resistance of 1.2mΩ, when in use, the winding temperature will be
elevated. Copper has a positive temperature coefficient of 3800ppm/°C. If we assume a typical temperature rise
of 20°, then the winding resistance will increase by 7.6% to approximately 1.3mΩ. This DCR value will be used in
the following calculations.
With 20A of load current through the inductor, the voltage drop due to the DCR will be 1.3 × 20 = 26mV. This will
be amplified by a factor of 48 by the current sense amplifier within the UCD7231. This will boost the signal to
1.25V. The internal circuitry then adds a 0.5V pedestal to the amplified signal which results in 1.75V at the IMON
pin. This voltage is within the 2.0V dynamic range of the current measurement and fault detection circuitry of the
controller, so the design can make use of the current sense network shown in Figure 3. No attenuation of the
signal is necessary. R2 in Figure 7 is not required and does not have to be loaded. (If a higher DCR inductor
were selected, attenuation of the current sense signal might be required, and, in that case, R2 would be
populated.)
The values for the current sense RC network (R1 and C6) around the inductor can now be calculated. The
requirement is L/DCR = RC. Let C = 1µF. Using 1µH for L and the warm DCR value of 1.3mΩ for DCR, the
calculated value for R is 769Ω. The nearest standard 1% value is 768Ω. Thus, C6 = 1µF and R1 = 768Ω.
The CSP and CSN pins are sensitive to noise pickup. Signal traces to these pins should be kept short and away
from the switching node and the gate drive traces. They should be shielded by ground planes and adjacent
ground fingers if possible. Series 2.49kΩ resistors R3 and R4 are added close to the CSP and CSN pins to help
attenuate noise. Further reduction in noise can be achieved by placing the current sense capacitor, C6, close to
R3 and R4.
FET SELECTION
At a minimum, the FETs used in the power stage must have a VDS breakdown rating of at least 1.5 times the
maximum input voltage. This headroom is required since the peak voltage on the switching node is always higher
than the input voltage due to ringing caused by energy storage in the parasitic inductance of the FETs and the
PCB traces. With good layout practices and the use of a snubber network, the peak voltage on the FETs can be
limited to 1.5 times Vin. In this example, a minimum VDS rating of 21V is required to accommodate a 14V input
voltage.
The high-side FET should be selected to handle current pulses equal to twice the steady state current rating of
the power stage. This allows headroom for ripple current, load transients, and brief over-current events. Note that
this is a pulsed current requirement, not a continuous current requirement. The average current in the high-side
FET is roughly equal to the load current times the duty cycle. For this example, an ID peak current rating of 40A
or higher is the target. The average current in the FET will be highest at full load, at the lowest input voltage and
highest output voltage. In this example, Vin(min) is 6V and Vout(max) is 3.3V. At full load, the average FET
current will be 11A. Adding a 20% safety margin to this value produces a 13.2A steady state drain current
requirement.
When converting power from input voltages of approximately 8V and higher, switching losses begin to dominate
over conduction losses in the high-side FET. That means RDS(ON) is not the primary specification that drives
high-side FET selection. Low gate charge (Qg), low gate-to-drain charge (Qgd), and low gate resistance (Rg)
become more important parameters. One of the most useful figures of merit is the product of on-resistance and
gate charge (Qg × RDS(ON)). The lower the number, the better the FET.
FETs are characterized at several standard gate enhancement voltages. The most popular are VGS voltages are
4.5V and 10V. Since our design is using approximately 6V of gate drive, the datasheet values of RDS(ON) at
4.5VGS will be of greatest interest. Be cautious of FETs that are characterized at 2.5VGS. These are low-threshold
FETs that are useful when converting power at input voltages below 6V. However, due to subtle, but serious,
side effects of the low threshold voltage, they are best avoided when converting power at voltages above 6V.
For this design the TI CSD16322Q5 is an excellent choice for the high-side FET (Q1). It has low charge, an
impressive figure of merit, and low Qgd. It exhibits very low switching losses. It is produced in an industry
standard, thermally enhanced, 5 × 6mm package. It has more than enough current handling capability for this
20A design.
18
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Low-side FET selection is driven primarily by RDS(ON). The lower the value, the higher the efficiency. Lower
RDS(ON) requires a larger die size, which increases total gate charge and device cost. For a given RDS(ON) value,
the part with the lowest Qg is likely to be the best choice. At higher input voltages and narrower duty cycles, the
low-side FET is conducting current for the majority of switching cycle. A thermally enhanced package is a must.
The continuous current rating of the FET should at least be equal to the current rating of the power stage.
The TI CSD16401Q5 is used as the low-side FET (Q2) in this design. It has an RDS(ON) of 1.5mΩ, with only 21nC
of Qg at 4.5V. It has more than enough current handling capacity. Its 25V minimum BVDSS rating beats our
minimum voltage criteria. It comes in the same 5 × 6mm package as Q1.
In rare instances, the addition of a series gate resistor can be of some benefit when dealing with high amplitude
ringing. Usually, however, the addition of series gate resistance increases switching losses and increases the
risk of cross-conduction between the high-side and low-side FETs. A very tight, low stray inductance PCB layout,
or a snubber network are the preferred methods for reducing ringing. Resistors R9 and R10 are shown as
placeholders in Figure 7. They can be added to the PCB layout to allow for the possibility that series gate
resistance may be needed. In most cases they are not required and can be considered optional. If they are
added to the design, the default value of 0Ω should initially be used.
SW NODE CLAMP
At higher output currents, the switching node can momentarily swing more than a 1V below ground. This
condition can interfere with the proper operation of the chip. To prevent the SW pin from being subjected to
excessive negative voltage swings, a Schottky diode clamp and current limiting resistor, D1 and R11, are
inserted between the actual switching node and the SW pin (pin 20). Diode D1 should be a power Schottky
device rated at a minimum of 0.5A of current and at least 30V breakdown voltage. The device shown in Figure 7
is a 0.5A, 40V device in a SOD123 package. The diode should be placed as close as possible to the UCD7231
and be connected between the SW pin and PGND pin by short, wide traces. Small-signal Schottky diodes should
not be used. Their forward voltage drop at higher currents is too high to provide effective clamping. Use a value
of 1Ω for R11. Larger values will interfere with the anti-cross conduction logic used to control the turn-on and
turn-off of the high-side FET, Q1.
SNUBBER NETWORK
Energy stored in the parasitic inductance in the source and drain leads of the power FETs is released when the
FETs abruptly turn on and off. The parasitic inductance interacts with the output capacitance ©OSS) of the FETs
to form a resonant circuit. The end result is high amplitude, high frequency ringing on the switching node that is
most prominent just after the high-side FET is turned on. The frequency of the ringing is commonly in the
100MHz range. Its peak amplitude can be as much as twice the input voltage. If nothing is done to damp the
ringing, it can cause avalanche breakdown of the low-side FET, increase radiated EMI levels, and, most
important for this discussion, interfere with the detection of an over-current condition. When left undamped, the
ringing on the switching node can take several hundreds of nanoseconds to die out.
A simple series RC network connected to the switching node is commonly used to dampen or “snub” the ringing.
The capacitor couples the high frequency content to the resistor, and the resistor dissipates the energy. With the
correct values, the ringing can be made to decay to negligible levels in 100ns or less. C5 (2200pF) and R8
(3.01Ω) perform this function in the example circuit. R8 must be capable of dissipating several hundred milliwatts
of power. The amount of power dissipated in R8 is proportional to the switching frequency and the value of C5.
With the values shown, R8 will dissipate approximately 125mW at 500kHz. This will double if the switching
frequency is increased to 1MHz. It is recommended that a 500mW rated resistor be used for R8. The optimum
values of the snubber R and C are device and layout dependant. Some experimentation may be needed to
achieve the optimum trade-off between damping time and power lost in the damping resistor. In most cases, the
value of R is between 1Ω and 10Ω, and C is between 1000pF and 4700pF. Higher values of C cause more
current to flow in R, thus increasing the power dissipated in it.
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COMPUTING VALUES FOR RDLY and RHS Sense
RDLY sets the amount of blanking time for the high speed comparator that monitors the voltage drop across the
high-side FET during its on-time. This comparator fires when the high-side FET is conducting too much current.
Because of the time it takes for ringing to decay on the switching node, the comparator “decision” should be
delayed for a short amount of time after the high-side FET is turned on. With a proper snubber network, a delay
time of 100ns should be sufficient to allow for proper over-current detection. Using the formula in the RDLY
section, value of 8.03kΩ produces a 100ns delay. The nearest standard value is 8.06kΩ, so this is the value
used for R7.
Note that when the duty cycle is of shorter duration than the blanking time, the high-side fault sensing circuit is
blanked for the entire time. Thus there is no high-side FET protection when the duty cycle duration is shorter
than the RDLY blanking time. This condition commonly occurs during soft-start, when the output voltage is being
ramped up from zero, or when operating at high switching frequencies and attempting to produce low output
voltages from high input voltages. Keep this in mind when setting operating frequency and input to output voltage
ratios.
The first step in selecting a value for RHS Sense, is to determine what is the maximum allowable voltage drop
across the high-side FET. This is calculated from the RDS(ON) of the FETs, taking into account its likely junction
temperature when operating at the maximum current point, and by the maximum allowable FET current. The
RDS(ON) value on the data sheet is specified at 25°C and at a particular VGS voltage, typically 4.5V and 10V. In
this case, neither value is correct, since this design will applying approximately 6.2V to the gate. Additionally,
FET RDS(ON) has a high, positive temperature coefficient of typically 4000ppm/°C. This means for a 100°C rise in
junction temperature, the on-resistance will go up by 40%. For a fault condition, using a 125°C junction
temperature is a reasonable assumption. A valid estimate of the RDS(ON) with 6V of enhancement at 125°C of the
CSD16322Q5 is 5.0mΩ.
The second step is to calculate a maximum current value for the high-side FET. A good rule of thumb is to use
150% of the rated output current value, plus one half of the peak-to-peak inductor ripple current. For this
example this gives a value of 1.5 × 20 × ½ × 5 = 32.5A. This level of current provides headroom for transients,
start-up surge currents, and the increase in inductor ripple current as the inductance falls with increasing current.
The maximum allowable voltage drop can now be calculated as just the product of the maximum current value
and the “hot” RDS(ON). This produces a value of 162.5mV for this design. This should not be regarded as a
precision value. Keep in mind that the high-side FET protection is meant to be “last resort” protection for the
power stage to prevent catastrophic damage to the power train. The maximum voltage drop value should be set
high enough to prevent nuisance trips of the protection circuitry under normal operation.
The value for RHS Sense can now be calculated. Its resistance, in kΩ, is equal to the maximum high-side voltage
drop, in mV, divided by 100. In our example this produces a value of 1.63kΩ. Rounding this up to the nearest
standard value of 1.65kΩ gives the value for R12.
SETTING THE ILIM THRESHOLD
The primary fault protection mechanism in the UCD7231 is the output current detection circuitry. An internal
comparator monitors the voltage on the ILIM and IMON pins. When the voltage on IMON exceeds the voltage on
ILIM, the FLT pin is asserted and power conversion stops. If a UCD92xx controller is used to drive the power
stage, it can also monitor the voltage on IMON and detect an over current condition. The threshold for the fault trip
point is easily set by firmware, making it very flexible. The maximum current sense input voltage that can be
correctly digitally sampled by a UCD92xx controller is 2.5V. (The maximum programmable limit for the fast OC
threshold is 2.0V.) For this design it was decided to use this 2.5V level as the threshold for the ILIM comparator.
In this way the digitally programmable controller will detect a slowly changing OC fault, and the ILIM comparator
in the driver will protect the system from a sudden increase in current. This corresponds to an output current of
approximately 31A. All that needs to be done is to set up a voltage divider that will produce 2.5V at the ILIM pin.
The BP3 pin provides a convenient source of clean, regulated 3.3V. The value of R5 was arbitrarily set to
10.0kΩ. Simple math produces a value of 31.6kΩ for R6. These values produce the desired 2.5V on ILIM. The
voltage divider only draws 80mA from the BP3 pin, which is within the allowable limits.
20
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INPUT AND OUTPUT CAPACITORS
At the drain of the high-side FET, current is drawn in fast, brief, rectangular pulses. It is very important to provide
low impedance, high frequency energy storage right at the drain of the FET. For this 20A power stage, two 22µF,
16V or 25V ceramic capacitors are recommended. C1 and C2 should be placed as close to the drain of Q1 as
possible. The ground side of the capacitors should be connected as close as possible to the source lead of Q2. If
designing a multiphase power supply, these capacitors should be present at each power stage. Bulk input
bypass capacitance may also be required to minimize voltage variations during transient loads. This bulk
capacitance is not shown on Figure 7, but it is typically required. Bulk capacitance can be shared among multiple
power stages.
The inductor ripple current must be absorbed by the output capacitors. The ripple current is triangular in shape
and contains significant energy at the switching frequency and its harmonics. To keep the ripple voltage
amplitude to a minimum, low ESR and low ESL capacitors must be used. Multilayer ceramic capacitors are ideal
devices. While bulk capacitance is also required to provide energy storage during transient events, the bulk
capacitors do not typically handle much ripple current because their higher ESL and ESR make them look
inductive at the ripple frequencies.
The output ripple voltage is directly proportional to the inductor ripple current. The inductor ripple current varies
widely with input voltage and duty cycle. That makes it difficult to come up with a one-size-fits-all
recommendation for the proper amount of ceramic output capacitance. A good starting point is approximately
100µF. In this design two 47µF capacitors are used (C7 and C8). These capacitors should be placed close to the
inductor, L1, and the ground side of these caps should be connected as close as possible to the source lead of
the low-side FET, Q2.
Bulk capacitance is used not only for short term transient energy storage, but also as a frequency response
tailoring element in the power supply feedback loop. Several hundred microfarads, at a minimum, are commonly
used in a power stage of this current capability. In this example, 330µF is being used (C9). More capacitance
may be required depending on the transient response requirements of the load.
BYPASS AND BOOTSTRAP CAPACITORS
In this design, the bypass capacitors on BP3 (C12), VGG (C4), and the bootstrap capacitor (C3) use the
recommended values. A high frequency 0.1µF bypass capacitor, C11, has also been added at the Vin pin of the
UCD7231. This cap attenuates the very high frequency noise that is present on the Vin rail. It should be placed
as close as possible to pin 16 and connect to analog ground with a short, direct trace.
LAYOUT RECOMMENDATIONS
Proper component placement and trace routing can have a significant impact on overall power stage efficiency
and reduce noise coupling into nearby circuits. The following are some key layout considerations.
• Locate the driver as close as possible to the power FETs, but do not place it directly under either FET. The
driver is a power device and needs its own thermal cooling path. Clustering multiple hot parts too close
together can increase the risk of excessive temperature rise and potentially cause a thermal shutdown event.
• Locate the VGG bypass and bootstrap capacitors as close as possible to the driver.
• Pay special attention to the GND trace. The ground side of the input bypass capacitors, the ground side of
the output capacitors, the low-side FET source leads, and the PGND connection to the driver should
connected together in a tight “single point” ground, using wide, low inductance traces and few, if any vias.
Use of a ground plane is strongly encouraged.
• Connect the power-pad on the bottom of the driver to analog ground. The power-pad is not intended to be a
high current carrying connection. The analog ground and power ground should be connected together at one
point, near the AGND pin. Care should be taken to insure that heavy currents are not pulled through the
analog ground traces.
• The switching node trace should be kept short and compact. This is the noisiest node in the system with high
dV/dt slew rates.
• Use wide traces for the HS Gate and LS Gate signals closely following the associated switching node and
ground traces. Use 0.050” to 0.080” (1.27 to 2.03 mm) wide traces if possible. Use at least two vias if the gate
drive trace has to be routed from one layer to another.
• Keep the low level input and output traces away from the switching node. The high dV/dt signal present there
can induce significant noise into the relatively high impedance nodes. Pay particular attention to the routing of
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the CSP and CSN traces.
INDUCTOR CURRENT SENSE TRACE LAYOUT
Since matching of the L/DCR to RC time constants is important to obtain an accurate replica of the inductor
current, the PCB layout must be done correctly to insure that the voltage drop across the inductor is sensed
properly. For best results, the current sensing connections should be made by separate, non-current-carrying
traces that connect directly to the inductor solder pads. The sensing connections should not be made to current
carrying traces that lead to the switching node or the output capacitors. An example of a correct and incorrect
layout is given in Figure 8.
Inductor PCB pads
Inductor PCB pads
To current sense
circuitry
To current sense
circuitry
Right!
Wrong!
Figure 8. Inductor Current Sense Trace Layout
The current carrying traces have finite resistance that exhibit an additional voltage drop which will contaminate
the sensed readings. It represents an additional DCR that is not taken into account in the current sensing
equations. The trace resistance varies with the thickness of the PCB copper used on the board. This thickness
can vary from batch to batch of pc boards, so the additional resistance of the traces is not a tightly controlled
value. Even a short length of PCB trace can introduce a significant amount of added resistance. Remember,
milliohms matter. By making a Kelvin connection to the inductor pads, the effects of PCB trace resistance can be
minimized.
LIMITATIONS OF DCR CURRENT SENSING
The accuracy of the DCR current sense method is limited by the stability of the DCR and L values of the power
inductor. In practice, the inductance value of the power inductor decreases with increasing load current. Most
inductors will exhibit a 20% to 30% reduction in inductance as load current changes from no load to full rated
current. The DCR sense method cannot detect inductor saturation or a cracked core, both of which cause greatly
increased ac current to flow in the inductor.
The resistance of the inductor windings is strongly affected by temperature. Most inductors use copper wire, and
copper has a resistance temperature coefficient of approximately +3800ppm/°C. This means that if the winding
temperature of the inductor rises by 40°C, its DCR will increase by 15.2%. This will cause the sensed voltage at
CSP and CSN to increase by 15.2% as well for the same current flow. If high accuracy of measured current is
important, then some form of temperature correction needs to be applied to the DCR sensed reading. This
requires some form of temperature sensing and a method to correlate the sensed temperature to the actual
winding temperature.
Since it is impractical to place a temperature sensor inside the inductor to sense the winding temperature, a
practical alternative is to sense the high-side FET device temperature. Tests have shown that a small analogoutput temperature sensor placed under the high-side FET on the back side of the board works well as a
substitute. Its temperature output correlates strongly to the inductor winding temperature. The voltage
proportional to temperature can be fed to the Temp input of the UCD92xx family of Digital Power Controllers. The
firmware internal to the controller can use the temperature reading to correct for the temperature effects on the
DCR current sense readings.
RELATED PRODUCTS
DEVICE
22
DESCRIPTION
LITERATURE NUMBER
UCD9240
Digital Point of Load System Controller
SLUS766C
UCD9220
Digital PWM System Controller
SLUS904
UCD9112
Digital Dual-Phase Synchronous Buck Controller
SLVS711C
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RELATED LITERATURE
DESCRIPTION
LITERATURE NUMBER
QFN/SON PCB Attachment
SLUA271A
Quad Flatpack No-Lead Logic Packages
SCBA017D
Reducing Ringing Through PCB Layout Techniques
SLPA005
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REVISION HISTORY
Changes from Original (November 2009) to Revision A
Page
•
Added additional parameters for VBST values; DC, Pulse 400ns, Pulse 64ns, and Pulse 16ns ........................................... 4
•
Changed VSW Pulse < 64 ns max value from 20 to 22 ......................................................................................................... 4
•
Changed VSW Pulse < 16 ns max value from 20 to 30 ......................................................................................................... 4
24
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Product Folder Link(s): UCD7231
PACKAGE OPTION ADDENDUM
www.ti.com
19-Jan-2011
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package
Drawing
Pins
Package Qty
Eco Plan
(2)
Lead/
Ball Finish
MSL Peak Temp
(3)
Samples
(Requires Login)
UCD7231RTJR
ACTIVE
QFN
RTJ
20
3000
Green (RoHS
& no Sb/Br)
CU NIPDAU Level-2-260C-1 YEAR
Purchase Samples
UCD7231RTJT
ACTIVE
QFN
RTJ
20
250
Green (RoHS
& no Sb/Br)
CU NIPDAU Level-2-260C-1 YEAR
Request Free Samples
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
14-Jul-2012
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
Diameter Width (mm)
(mm) W1 (mm)
B0
(mm)
K0
(mm)
P1
(mm)
W
Pin1
(mm) Quadrant
UCD7231RTJR
QFN
RTJ
20
3000
330.0
12.4
4.25
4.25
1.15
8.0
12.0
Q2
UCD7231RTJR
QFN
RTJ
20
3000
330.0
12.4
4.25
4.25
1.15
8.0
12.0
Q2
UCD7231RTJT
QFN
RTJ
20
250
180.0
12.4
4.25
4.25
1.15
8.0
12.0
Q2
UCD7231RTJT
QFN
RTJ
20
250
180.0
12.4
4.25
4.25
1.15
8.0
12.0
Q2
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
14-Jul-2012
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
UCD7231RTJR
QFN
RTJ
20
3000
367.0
367.0
35.0
UCD7231RTJR
QFN
RTJ
20
3000
367.0
367.0
35.0
UCD7231RTJT
QFN
RTJ
20
250
210.0
185.0
35.0
UCD7231RTJT
QFN
RTJ
20
250
210.0
185.0
35.0
Pack Materials-Page 2
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