G FEATURINperature Range July 2000 Tem ommercial Extended C 70˚C to C 0˚ -2 ment dheld Equip an H e bl ta or for P ML4863* High Efficiency Flyback Controller GENERAL DESCRIPTION FEATURES The ML4863 is a flyback controller designed for use in multi-cell battery powered systems such as PDAs and notebook computers. The flyback topology is ideal for systems where the battery voltage can be either above or below the output voltage, and where multiple output voltages are required. ■ Variable frequency current mode control and synchronous rectification for high efficiency ■ Minimum external components ■ Guaranteed start-up and operation over a wide input voltage range (3.15V to 15V) ■ High frequency operation (>200kHz) minimizes the size of the magnetics ■ Flyback topology allows multiple outputs in addition to the regulated 5V ■ Built-in overvoltage and current limit protection The ML4863 uses the output voltage as the feedback control signal to the current mode variable frequency flyback controller. In addition, a synchronous rectifier control output is supplied to provide the highest possible conversion efficiency (greater than 85% efficiency over a 1mA to 1A load range). The ML4863 has been designed to operate with a minimum number of external components to optimize space and cost. *Some Packages Are Obsolete BLOCK DIAGRAM VCC SHDN 3 1 4 BIAS & UVLO VIN VFB VCC 4.5V LDO 5 VFB VFB – + I COMP – GND 8 VCC CURRENT COMPARATOR + VREF SWITCHING CONTROL OUT 1 A1 6 18mV Rgm 18mV VCC RECTIFIER COMPARATOR OUT 2 – COMP + CROSS-CONDUCTION PROTECTION BLANKING A2 7 SENSE 2 1 ML4863 PIN CONFIGURATION ML4863 8-Pin SOIC (S08) VIN 1 8 GND SENSE 2 7 OUT 2 SHDN 3 6 OUT 1 VFB 4 5 VCC TOP VIEW PIN DESCRIPTION PIN NAME FUNCTION 1 VIN Battery input voltage 2 SENSE Secondary side current sense 3 SHDN Pulling this pin high initiates a shutdown mode to minimize battery drain 4 2 VFB Feedback input from transformer secondary, and supply voltage when VOUT > 4.5V PIN NAME FUNCTION 5 VCC Internal power supply node for connection of a bypass capacitor 6 OUT 1 Flyback primary switch MOSFET driver output 7 OUT 2 Flyback synchronous rectifier MOSFET driver output 8 GND Analog signal ground ML4863 ABSOLUTE MAXIMUM RATINGS Absolute maximum ratings are those values beyond which the device could be permanently damaged. Absolute maximum ratings are stress ratings only and functional device operation is not implied. Lead Temperature (Soldering 10 Sec.) ..................... 260ºC Thermal Resistance (qJA) .................................... 160ºC/W VIN ................................................................. GND – 0.3V to 18V Voltage on any other pin ........................... GND – 0.3V to 7V Source or Sink Current (OUT1 & OUT2) ...................... 1A Junction Temperature .............................................. 150ºC Storage Temperature Range...................... –65ºC to 150ºC Temperature Range ML4863CS ................................................. 0ºC to 70ºC ML4863ES ............................................. –20ºC to 70ºC ML4863IS .............................................. –40ºC to 85ºC VIN Operating Range ................................... 3.15V to 15V OPERATING CONDITIONS ELECTRICAL CHARACTERISTICS Unless otherwise specified, VIN = 12V, TA = Operating Temperature Range (Note 1) SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS C Suffix 2.1 2.5 2.8 µs E/I Suffix 2.1 2.5 2.95 µs VFB = 0V 450 650 850 ns Line, Load, & Temp 4.85 5 5.15 V OSCILLATOR tON ON Time Minimum Off Time VFB REGULATION Total Variation OUTPUT DRIVERS OUT1 Rise Time CLOAD = 3nF, 20% to 90% of VCC 60 70 ns OUT1 Fall Time CLOAD = 3nF, 90% to 20% of VCC 60 70 ns OUT2 Rise Time CLOAD = 3nF, 20% to 90% of VCC 60 70 ns OUT2 Fall Time Continuous Mode, CLOAD = 3nF, 90% to 20% of VCC 60 70 ns Discontinuous Mode, CLOAD = 3nF, 90% to 20% of VCC 125 150 ns SHDN Input High Voltage 2.0 V Input Low Voltage Input Bias Current SHDN = 5V SENSE Threshold – Full Load VIN = 5V, VFB = VFB (No Load) – 100mV SENSE Threshold – Short Circuit VFB = 0V 0.8 V 5 10 µA 150 160 mV 235 mV SENSE 130 CIRCUIT PROTECTION Undervoltage Lockout Start-up Threshold 3.0 3.15 V Undervoltage Lockout Hysteresis 0.5 0.6 V 3 ML4863 ELECTRICAL CHARACTERISTICS SYMBOL (Continued) PARAMETER CONDITIONS MIN TYP MAX UNITS 100 150 µA SHDN = 5V 20 25 µA SHDN = 5V, VIN < 6V 5 10 µA SUPPLY IFB VFB Quiescent Current IIN VIN Shutdown Current VCC Note 1: 4 VCC Output Voltage VFB = 0V, VIN = 15V, CVCC = 0.1µF 4.5 5.5 V VFB = 0V, VIN = 6V, CVCC = 0.1µF 4.0 5.0 V VFB = 0V, VIN = 3.15V, CVCC = 0.1µF 2.8 VFB = 5V 4.5 Limits are guaranteed by 100% testing, sampling, or correlation with worst case test conditions. V 5 5.15 V ML4863 FUNCTIONAL DESCRIPTION TRANSCONDUCTANCE AMPLIFIER The ML4863 utilizes a flyback topology with constant ontime control. The circuit determines the length of the offtime by waiting for the inductor current to drop to a level determined by the feedback voltage (VFB). Consequently, the current programming is somewhat unconventional because the valley of the current ripple is programmed instead of the peak. The controller automatically enters burst mode when the programmed current falls below zero. Constant on-time control therefore features a transition into and out of burst mode which does not require additional control circuitry. The control circuit is made up of four distinctive blocks; the constant on-time oscillator, the current programming comparator, the feedback transconductance amplifier, and the synchronous rectifier controller. A simplified circuit diagram is shown in Figure 1. OSCILLATOR & COMPARATOR The oscillator has a constant on-time and a minimum offtime. The off-time is extended as long as the output of the current programming comparator is low. Note that in constant on-time control, a discharge (off-time) cycle is needed for the inductor current to be sensed. The minimum off-time is required to account for the finite circuit delays in sensing the inductor output current. The feedback transconductance amplifier generates a current from the voltage difference between the output and the reference. This current produces a voltage across Rgm that adds to the negative voltage on the current sense resistor, RSENSE. When the current level in the inductor drops low enough to cause the voltage at the non-inverting input of the current programming comparator to go positive, the comparator trips and the converter starts a new on-cycle. The current programming comparator controls the length of the off-time by waiting until the current in the secondary decreases to the value specified by the feedback transconductance amplifier. In this way, the feedback transconductance amplifier‘s output current steers the current level in the inductor. When the output voltage drops due to a load increase, it will increase the output current of the feedback amplifier and generate a larger voltage across Rgm which in turn raises the secondary current trip level. However, when the output voltage is too high, the feedback amplifier’s output current will eventually become negative. Because the output current of the inductor can never go negative by virtue of the diode, the non-inverting input of the comparator will also stay negative. This causes the converter to stop operation until the output voltage drops enough to increase the output current of the feedback transconductance amplifier above zero. VOUT IS VIN RESR 4 LP FEEDBACK TRANSCONDUCTANCE AMPLIFIER RP + CP VREF CURRENT PROGRAMMING COMPARATOR CONSTANT ON-TIME MINIMUM OFF-TIME OSCILLATOR C ONE SHOT tON 2.5µs + – 1:1 LOAD VFB COMP – OUT 1 6 ONE SHOT tOFF 500ns Rgm RECTIFIER COMPARATOR – COMP + OUT 2 BLANKING 7 A2 SENSE ML4863 2 RSENSE Figure 1. Schematic of the ML4863 Controller and Power Stage 5 ML4863 FUNCTIONAL DESCRIPTION (Continued) SYNCHRONOUS RECTIFIER CONTROL where h = converter efficiency. The control circuitry for the synchronous rectifier does not influence the operation of the main controller. The synchronous rectifier is turned on during the minimum off time, or whenever the SENSE pin is less than –18mV. During transitions where the primary switch is turned on before the voltage on the SENSE pin goes above –18mV, the gate of the synchronous rectifier is discharged softly to avoid accidently triggering the current-mode comparator with the gate discharge spike on the sense resistor. Once RSENSE has been determined, LP can be found: The part will also operate with a Schottky diode in place of the synchronous rectifier, but the conversion efficiency will suffer. The normal operating range and current limit point are determined by the current programming comparator. They are dependent on the value of the synchronous rectifier current sense resistor (RSENSE), the nominal transformer primary inductance (LP), and the input voltage. RSENSE can be calculated by: 6 VIN0 MIN5 VOUT + VIN ´ 150mV V I 0 5 + 20 ´ V 0 0 5 ´ I5 IN MIN OUT MAX IN MAX 0 OUT MAX (2) Three operational modes are defined by the voltage at the SENSE pin at the end of the off-time: discontinuous mode, continuous mode, and current limit. The following values can be used to determine the current levels of each mode: VSENSE < 0V: discontinuous mode 0V < VSENSE < 160mV: continuous mode 160mV < VSENSE < 235mV: current limit CURRENT LIMIT AND MODES OF OPERATION R SENSE = LP = (25 × 10 −6 ) × VIN0MAX5 × R SENSE ´h 5 (1) Inserting the maximum value of VSENSE for each operational mode into the following equation will determine the maximum current levels for each operational mode: IOUT = VIN V t × VIN × SENSE + ON ×η VOUT + VIN R SENSE 2 × LP (3) ML4863 DESIGN CONSIDERATIONS DESIGN PROCEDURE See Table 1 for suggested component manufacturers. A typical design can be implemented by using the following procedure. Component Manufacturer 1. Sense Resistors Dale IRC LRC Series WSL Series (402) 563-6506 (512) 992-7900 The maximum input voltage (VIN(MAX)) The mainimum input voltage (VIN(MIN)) The maximum output current (IOUT(MAX)) The maximum output ripple (DVOUT) Inductors Coilcraft R4999 (847) 639-6400 As a general design rule, the output ripple should be kept below 100mV to ensure stability. Capacitors Specify the application by defining: MOSFETs 2. Select a sense resistor, RSENSE, using equation 1. 3a. Determine the inductance required for the optimum output ripple using equation 2. 3b. Determine the minimum inductor current rating required. The peak inductor current is calculated using the following formula: IL PEAK = . 10 -6 ) 235mV VIN ( MAX) (25 + R SENSE LP (4) 3c. Specify the inductor's DC winding resistance. A good rule of thumb is to allow 5mW, or less, of resistance per µH of inductance. For minimum core loss, choose a high frequency core material such as Kool-Mu, ferrite, or MPP. 3d. Specify the coupled inductor's turns ratio: C = IOUT ( MAX) 4b. + VIN ( MAX) VOUT 25. 10 DV (207) 282-5111 Sprague 593D Series (207) 324-4140 National NDS94XX NDS99XX (800) 272-9954 Motorola MMDF Series MMSF Series (602) 897-5056 Siliconix Littlefoot Series (408) 988-8000 Select the sense resistor, RSENSE, using Equation 1: 4 150mV 4V × + × 0.85 5+ 4 500mA 20 × 6 × 0.5 (1a) Determine the inductance required using equation 2. LP = (25 × 10 −6 ) × 6 × 0.12 = 18µH 3b. (2a) Determine the minimum inductor current rating required. (6) As a final design check, evaluate the system stability (no compensation, single pole response) by using the following equation: R ! TPS series 2. (5) 150mV ∆VOUT ≤ (6 × 10 −6 ) × AVX Specify the application by defining: VIN(MAX) = 6V VIN(MIN) = 4V IOUT(MAX) = 500mA DVOUT = 100mV 3a. DVOUT R SENSE LPE-6562 Series (605) 665-9301 LPT-4545 series RSENSE = 138mW @ 120mW -6 OUT OCTA-PAC Series (561)241-7876 Dale 1. R SENSE = Establish the maximum allowable ESR for the ouput capacitor: RESR < 5. OUT Coiltronics DESIGN EXAMPLE Calculate the minimum output capacitance required using: V Phone Table 1. Component Suppliers Np : Ns = 1:1 4a. Part Number SENSE × (VOUT + VIN (MIN) ) LP "# $ IL PEAK 235mV 6 × (25 . × 10 −6 ) = + = 2.79A 120mΩ 18 × 10 –6 (4a) (7) where RSENSE and LP are the actual values to be used. 7 ML4863 DESIGN CONSIDERATIONS (Continued) 3c. 3d. Specify the inductor’s DC winding resistance: LAYOUT LDCR = 90mW Good PC board layout practices will ensure the proper operation of the ML4863. Important layout considerations follow: Specify the coupled inductor's turn ratio: Np : Ns = 1:1 4a. • The connection from the current sense resistor to the SENSE pin of the ML4863 should be made by a separate trace and connected right at the sense resistor lead. Calculate the minimum output capacitance required using equation 5. C = 0.50 × 4b. 5 + 6 × 25. × 10 5 0.1 −6 = 55µF (5a) Establish the maximum ESR for the output capacitor using equation 6. RESR < • Trace lengths from the capacitors to the inductor, and from the inductor to the FET should be as short as possible to minimize noise and ground bounce. 0.1× 0.12 = 80mW 150mV (6a) Based on these calculations, the design should use two 100µF capacitors, with an ESR of 100mW each, in parallel to meet the capacitance and ESR requirements. 5. • The VCC bypass capacitor needs to be located close to the ML4863 for adequate filtering of the IC's internal bias voltage. • Power and ground planes must be large enough to handle the current the converter has been designed for. See Figure 5 for a sample PC board layout. As a final design check, evaluate the system stability using equation 7. 100mV ≤ (6 × 10 −6 ) × 0.12 × (5 + 4) "# = 360mV (7a) ! 18 × 10 $ –6 Since the inequality is met, the circuit should be stable. Some typical application circuits are shown in Figures 2, 3, and 4. VOUT 5V, 1A 400µF Coiltronics CTX20-4 VIN 47µF VIN 100µF ML4863 VIN ML4863 NDS9955 GND VIN GND SENSE OUT 2 SENSE OUT 2 SHDN OUT 1 SHDN OUT 1 VFB VFB VCC 1µF VCC NDS9410 NDS9410 1µF 100mΩ Figure 2. 5V, 1A Circuit 8 VOUT 5V, 2A 800µF Dale LPE6562 50mΩ Figure 3. 5V, 2A Circuit ML4863 12V C4 33µF 20V C5 33µF 20V 5V C6 100µF 6.3V T1 DALE LPE-6562-A145 7 9 8 2 3 C7 100µF 6.3V C8 100µF 6.3V C9 100µF 6.3V C10 100µF 6.3V C11 100µF 6.3V C12 100µF 6.3V 3.3V 1,5 C13 100µF 6.3V 6,10 NDS9955 Q1A Q1B 4 Q2A Q2B MMDF3N03 ML4863 VIN SHDN C1 33µF 20V C2 33µF 20V VIN R1 120mΩ GND SENSE OUT 2 R2 30mΩ SHDN OUT 1 VFB VCC C3 1µF 50V R3 60mΩ Figure 4. 5W Multiple Output DC-DC Converter Figure 5. Typical PC Board Layout 9 ML4863 PHYSICAL DIMENSIONS inches (millimeters) Package: S08 8-Pin SOIC 0.189 - 0.199 (4.80 - 5.06) 8 PIN 1 ID 0.148 - 0.158 0.228 - 0.244 (3.76 - 4.01) (5.79 - 6.20) 1 0.017 - 0.027 (0.43 - 0.69) (4 PLACES) 0.050 BSC (1.27 BSC) 0.059 - 0.069 (1.49 - 1.75) 0º - 8º 0.055 - 0.061 (1.40 - 1.55) 0.012 - 0.020 (0.30 - 0.51) 0.004 - 0.010 (0.10 - 0.26) 0.015 - 0.035 (0.38 - 0.89) 0.006 - 0.010 (0.15 - 0.26) SEATING PLANE ORDERING INFORMATION PART NUMBER TEMPERATURE RANGE PACKAGE ML4863CS ML4863ES ML4863IS (Obsolete) 0ºC to 70ºC –20ºC to 70ºC –40ºC to 85ºC 8-Pin SOIC (S08) 8-Pin SOIC (S08) 8-Pin SOIC (S08) © Micro Linear 1997. is a registered trademark of Micro Linear Corporation. All other trademarks are the property of their respective owners. Products described herein may be covered by one or more of the following U.S. patents: 4,897,611; 4,964,026; 5,027,116; 5,281,862; 5,283,483; 5,418,502; 5,508,570; 5,510,727; 5,523,940; 5,546,017; 5,559,470; 5,565,761; 5,592,128; 5,594,376; 5,652,479; 5,661,427; 5,663,874; 5,672,959; 5,689,167. Japan: 2,598,946; 2,619,299; 2,704,176. Other patents are pending. Micro Linear reserves the right to make changes to any product herein to improve reliability, function or design. Micro Linear does not assume any liability arising out of the application or use of any product described herein, neither does it convey any license under its patent right nor the rights of others. The circuits contained in this data sheet are offered as possible applications only. Micro Linear makes no warranties or representations as to whether the illustrated circuits infringe any intellectual property rights of others, and will accept no responsibility or liability for use of any application herein. The customer is urged to consult with appropriate legal counsel before deciding on a particular application. 10 2092 Concourse Drive San Jose, CA 95131 Tel: 408/433-5200 Fax: 408/432-0295 www.microlinear.com DS4863-01