MICROSEMI LX1684CD

LX1684
Voltage-Mode PWM Controller
®
TM
P RODUCTION D ATA S HEET
KEY FEATURES
DESCRIPTION
Current is sensed using the voltage
drop across the RDS(ON) of the MOSFET
this sensing is delayed for 1µs to
eliminate MOSFET ringing errors.
Hiccup-mode fault protection reduces
average power to the power elements
during short-circuit conditions.
Under-voltage lockout and soft-start
are provided for optimal start-up
performance. Pulling the soft-start pin to
ground can disable the LX1684.
The small 14-pin SOIC packaging,
combined with low profile, low ESR
capacitors, and TO-252 packaged FETs
results in a high efficiency converter in a
small board area. In most cases PCB
copper can accomplish necessary heat
sinking and no bulky additional heat
sinks are required.
If a low profile design is not required
small electrolytic capacitors can be used
reducing the overall converter cost.
ƒ Fixed 175kHz Switching
Frequency
ƒ Constant Frequency VoltageMode Control Requires No
External Compensation
ƒ Hiccup-Mode Over-Current
Protection
ƒ High Efficiency
ƒ Output Voltage Set By
Resistor Divider
ƒ Under-Voltage Lockout
ƒ Soft-Start And Enable
ƒ Synchronous Rectification
ƒ Small, 14-pin Surface Mount
Package
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The LX1684 is a monolithic, voltagemode pulse-width modulator controller.
It is designed to implement a flexible,
low cost buck (step-down) regulator
supply with a minimal of external
components.
The LX1684 has a synchronous
driver for higher efficiency and is
optimized to provide 12V to 3.3V or
12V to 2.5V regulation. It also can be
used to convert 5V or 3.3V to voltages
as low as 1.25V.
Switching frequency is fixed at
175kHz for optimal cost and space.
Short-circuit current limiting can be
implemented without expensive current
sense resistors.
Similar to the LX1682 in function but
with the positive side of the current
sense comparator brought out (CSP) to
allow it to be referenced to the topside
FET.
APPLICATIONS
ƒ 12V to 3.3V Or Less Buck
Regulators
ƒ 3.3/5V to 2.5V Or Less Buck
Regulators
ƒ Hard Disk Drives
ƒ Computer Add-on Cards
IMPORTANT: For the most current data, consult MICROSEMI’s website: http://www.microsemi.com
PRODUCT HIGHLIGHT - TYPICAL 12V TO 3.3V APPLICATION
12V
10%
5V
5%
C5
1µF
R1
165 Ohm
12V
1
LM78L05
or equivalent
2
5V
TO
VCC
3
4
R2
100 Ohm
Css
0.1µF
5
6
Optional 12V only circuit
7
FB
VCC
NC
NC
SS
CS
CSP
NC
GND
VC1
PGND
BDR
NC
TDR
C1
0.1µF
14
RSET
12 3.3K
3x
820µF,
16V
D1
13
C4
5817SMT
11
Q1
SUB45N
05-20L
10
9
L1 is a powdered
iron toroid
8
Q2
SUB45N0520L
L1
10µH
4x
1500µF
6.3V
VOUT
3.3V
10A
C3
LX1684
PACKAGE ORDER INFO
TA (°C)
OUTPUT
0 to 70
Synchronous
D
Plastic SOIC
14-PIN
RoHS Compliant / Pb-free Transition DC: 0440
LX1684CD
Note: Available in Tape & Reel. Append the letters “TR” to the part number. (i.e. LX1684CD-TR)
Copyright © 2000
Rev. 1.0c, 2005-03-08
Microsemi
Integrated Products Division
11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570
Page 1
LX1684
Voltage-Mode PWM Controller
®
TM
P RODUCTION D ATA S HEET
Supply Voltage (VC1) ...................................................................................... 18V
Supply Voltage (VCC)........................................................................................ 7V
Input Voltage (CSP Pin) ................................................................................. 14V
Output Drive Peak Current Source (500ns) ................................................... 1.0A
Output Drive Peak Current Sink (500ns) ....................................................... 1.0A
Input Voltage (SS/ENABLE Pin) .......................................................... -0.3 to 6V
Operating Junction Temperature.................................................................. 150°C
Storage Temperature ................................................................... -65°C to +150°C
Peak Package Solder Reflow Temp. (40 second max. exposure) ...260°C (+0, -5)
Note:
PACKAGE PIN OUT
FB
1
14
V CC
N/C
2
13
N/C
SS
3
12
CS
VCSP
4
11
N/C
GND
5
10
V C1
PGND
6
9
N/C
BDRV
7
8
TDRV
Exceeding these ratings could cause damage to the device. All voltages are with respect to
Ground. Currents are positive into, negative out of specified terminal.
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ABSOLUTE MAXIMUM RATINGS (NOTE 1)
14-P IN SOIC
(Top View)
RoHS / Pb-free 100% Matte Tin Lead Finish
THERMAL DATA
D
PACKAGE
THERMAL RESISTANCE-JUNCTION TO AMBIENT, θJA
165°C/W
Junction Temperature Calculation: TJ = TA + (PD x θJA).
The θJA numbers are guidelines for the thermal performance of the device/pc-board
system. All of the above assume no ambient airflow.
FUNCTIONAL PIN DESCRIPTION
PIN NAME
DESCRIPTION
VFB
Voltage Feedback. A 1.25V reference is connected to a resistor divider to set desired output voltage.
SS
Soft-Start And Hiccup Capacitor Pin. During start up the voltage of this pin controls the output voltage. An
internal 22kΩ resistor and the external capacitor set the time constant for soft-startup. Soft-start does not begin
until the supply voltage exceeds the UVLO threshold. When over-current occurs, this capacitor is used for timing
hiccup. The PWM can be disabled by pulling the SS pin below 0.3V
VCSP
Positive Over-Current Threshold Input
GND
Analog ground for SS, FB, CS and VCC.
MOSFET driver power ground
TDRV
Gate Drive For Upper MOSFET.
BDRV
Gate Drive For Lower MOSFET.
VC1
Separate Supply For MOSFET Gate Drives. Connect to gate drive voltage.
CS
Over-Current Set. Connect resistor between CS pin and the source of the upper MOSFET to set current-limit
point.
VCC
IC Supply Voltage (nominal 5V).
Copyright © 2000
Rev. 1.0c, 2005-03-08
Microsemi
Integrated Products Division
11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570
PACKAGE DATA
PGND
Page 2
LX1684
Voltage-Mode PWM Controller
®
TM
P RODUCTION D ATA S HEET
Unless otherwise specified, the following specifications apply over the operating ambient temperature 0°C ≤ TA ≤ 70°C except where
otherwise noted. Test conditions: VCC=5V, VC1=16V
Parameter
`
`
VFB
Max
VOUT =VFB, TA=25°C
1.237
1.25
1.262
V
VOUT =VFB, 0°C < TA < 70°C
1.231
1.269
V
200
kHz
Frequency
FOSC
Ramp Amplitude
VRAMP
135
RIN
VOUT = VFB
1.25
VPP
20
kΩ
ISET
VCS = VCC –0.4V
40
45
µA
300
380
CURRENT SENSE
VTRIP
Current Sense Delayed
`
TCSD
420
mV
1.0
µsec
50
nS
14
V
OUTPUT DRIVERS
Drive Rise Time, Fall Time
TRF
CL = 3000pF
Drive High
VDH
ISOURCE = 20mA, VC1=16V
Drive Low
VDL
ISINK = 20mA, VC1 =16V
VST
VC1 > 4.0V
13
0.1
0.2
V
4.25
4.5
V
UVLO AND SOFT-START (SS)
VCC5 Start-Up Threshold
4.0
Hysteresis
SS Resistor
RSS
SS Output Enable
VEN
Hiccup Duty Cycle
DCHIC
0.25
0.10
V
22
kΩ
0.3
0.35
V
CSS = 0.1µF, FREQ =100Hz
12
14
%
SUPPLY CURRENT
VC1 Dynamic Supply Current
ICD
Out Freq = 175kHz, CL=3000pF, Synch., VSS
> 0.3V
30
40
mA
Static Supply Current VC1
IVC1
VSS < 0.3V ; outputs low (disable), VC1 = 16V
7
9
mA
5V
IVCC
VSS > 0.3V ; outputs low (disable)
9
12
mA
Copyright © 2000
Rev. 1.0c, 2005-03-08
Microsemi
Integrated Products Division
11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570
Page 3
ELECTRICALS
`
175
ERROR AMPLIFIER
Current Set
`
Units
OSCILLATOR
Input Resistance
`
LX1684
Typ
Test Conditions
REFERENCE
Reference Voltage
`
Min
Symbol
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ELECTRICAL CHARACTERISTICS
LX1684
Voltage-Mode PWM Controller
®
TM
P RODUCTION D ATA S HEET
R SET
I SET
CS
12
+17.5V
CS Com p
I RESET
VCSP
4
V TRIP
10
PW M
+
I SET
R
Q
S
Q
V FB
-
+
+
V RESET
-
R1
V CO RE
TDRV
Error Com p
Am plifier/
Com pensation
C IN
8
Set
1
R2
V C1
320k
20k
V IN (+12V)
L
7
ESR
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BLOCK DIAGRAM
C O UT
BDRV
V REF
6
PG ND
R SS
5
Hiccup Hiccup
G ND
+5V
UVLO
Ramp
O scillator
UVLO
14
V CC
3
SS/ENABLE
C SS
BLOCK DIAGRAM
Copyright © 2000
Rev. 1.0c, 2005-03-08
Microsemi
Integrated Products Division
11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570
Page 4
LX1684
Voltage-Mode PWM Controller
®
TM
P RODUCTION D ATA S HEET
GENERAL DESCRIPTION
The LX1684 is a voltage-mode pulse-width modulation
controller integrated circuit. The internal oscillator and ramp
generator frequency is fixed at 175kHz. The devices have internal
compensation, so that no external compensation is required.
POWER UP AND INITIALIZATION
At power up, the LX1684 monitors the supply voltage to both
the +5V and the VC1 pins (there is no special requirement for the
sequence of the two supplies). Before both supplies reach their
under-voltage lock-out (UVLO) thresholds, the soft-start (SS) pin
is held low to prevent soft-start from beginning; the oscillator
control is disabled and the top MOSFET is kept OFF.
SOFT-START
Once the supplies are above the UVLO threshold, the soft-start
capacitor begins to be charged up by the reference through a 20k
internal resistor. The capacitor voltage at the SS pin rises as a
simple RC circuit. The SS pin is connected to the amplifier's noninverting input that controls the output voltage. The output voltage
will follow the SS pin voltage if sufficient charging current is
provided to the output capacitor. The simple RC soft-start allows
the output to rise faster at the beginning and slower at the end of
the soft-start interval. Thus, the required charging current into the
output capacitor is less at the end of the soft-start interval so
decreasing the possibility of an over-current. A comparator
monitors the SS pin voltage and indicates the end of soft-start
when SS pin voltage reaches 95% of VREF .
OVER-CURRENT PROTECTION (OCP) AND HICCUP
The LX1684 uses the RDS(ON) of the upper MOSFET,
together with a resistor (RSET) to set the actual current limit point.
Unlike the LX1681/2 controllers the LX1684 includes the positive
current sense comparator input providing true Kelvin sensing
which is more accurate and offers more noise immunity.
This
Kelvin sensing also simplifies the PCB layout for current sense.
The CSP pin has a useful common mode input range to about 14V.
The comparator senses the current 1µs after the top MOSFET is
switched on.
Experiments have shown that the MOSFET drain voltage will
ring for 200-500ns after the gate is turned on. In order to reduce
inaccuracies due to ringing, a 1µs blanking delay after gate turn-on
is built into the current sense comparator. This 1us delay reduces
the effectiveness of the current sense comparator when the output
pulse width is below 1us. This can be problem when the set
application output voltage is less than 3.0V with a 12V input.
Under this condition the output current limit protection will not
function properly. This is usually not true with a short circuit
current condition, which causes the on time to be greater than the
blanking delay. In this circumstance the current limit comparator
would turn off the top FET driver.
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THEORY OF OPERATION
The comparator draws a current (ISET), whose magnitude is
45µA. The set resistor is selected to set the current limit for the
application. When the sensed voltage across the RDS(ON) plus the
set resistor exceeds the 400mV VTRIP threshold, the OCP
comparator outputs a signal to reset the PWM latch and to start
hiccup mode. The soft-start capacitor (CSS) is discharged slowly
(10 times slower than when being charged up by RSS). When the
voltage on the SS/ENABLE pin reaches a 0.3V threshold, hiccup
finishes and the circuit soft-starts again. During hiccup, the top
MOSFET is OFF and the bottom MOSFET remains ON. Hiccup is
disabled during the soft-start interval, allowing the circuit to start
up with the maximum current. If the rise speed of the output
voltage is too fast, the required charging current to the output
capacitor may be higher than the limit-current. In this case, the
peak MOSFET current is regulated to the limit-current by the
current-sense comparator. If the MOSFET current still reaches its
limit after the soft-start finishes, the hiccup is triggered again. The
hiccup ensures the average heat generation on both MOSFET’s and
the average current to be much less than that in normal operation,
if the output has a short circuit. Over-current protection can also be
implemented using a sense resistor, instead of using the RDS(ON)
of the upper MOSFET, for greater set-point accuracy. See
Application Information section.
OSCILLATOR FREQUENCY
An internal oscillator sets the switching frequency at about 175
kHz.
DESCRIPTION
Copyright © 2000
Rev. 1.0c, 2005-03-08
Microsemi
Integrated Products Division
11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570
Page 5
LX1684
Voltage-Mode PWM Controller
®
TM
P RODUCTION D ATA S HEET
OUTPUT INDUCTOR
OUTPUT CAPACITOR (continued)
The output inductor should be selected to meet the requirements
of the output voltage ripple in steady-state operation and the
inductor current slew-rate during transient. The peak-to-peak
output voltage ripple is:
Electrolytic capacitors can be used for the output capacitor, but
are less stable with age than tantalum capacitors. As they age, their
ESR degrades, reducing the system performance and increasing the
risk of failure. It is recommended that multiple parallel capacitors
be used, so that, as ESR increases with age, overall performance
will still meet the processor’s requirements.
There is frequently strong pressure to use the least expensive
components possible, however, this could lead to degraded longterm reliability, especially in the case of filter capacitors.
Linfinity’s demonstration boards use Sanyo MV-GX filter
capacitors, which are aluminum electrolytic, and have
demonstrated reliability. The Oscon series from Sanyo generally
provides the very best performance in terms of long term ESR
stability and general reliability, but at a substantial cost penalty.
The MV-GX series provides excellent ESR performance at a
reasonable cost. Beware of off-brand, very low-cost filter
capacitors, which have been shown to degrade in both ESR and
general electrolytic characteristics over time.
VRIPPLE = ESR × IRIPPLE
where
IRIPPLE =
(VIN - VOUT ) × VOUT
fSW × L
VIN
IRIPPLE is the inductor ripple current, L is the output inductor
value and ESR is the Effective Series Resistance of the output
capacitor.
IRIPPLE should typically be in the range of 20% to 40% of the
maximum output current. Higher inductance results in lower
output voltage ripple, allowing slightly higher ESR to satisfy the
transient specification. Higher inductance also slows the inductor
current slew rate in response to the load-current step change, ∆I,
resulting in more output-capacitor voltage droop. The inductorcurrent rise and fall times are:
TRISE =
L × ∆I
(VIN − VOUT )
and
TFALL =
L × ∆I
VOUT
INPUT CAPACITOR
The input capacitor and the input inductor are to filter the
pulsating current generated by the buck converter to reduce
interference to other circuits connected to the same 12V rail. In
addition, the input capacitor provides local de-coupling the buck
converter. The capacitor should be rated to handle the RMS current
requirement. The RMS current is:
IRMS = IL d (1 − d )
When using electrolytic capacitors, the capacitor voltage droop
is usually negligible, due to the large capacitance.
OUTPUT CAPACITOR
ESR × (IRIPPLE + ∆I ) < VEX
where IRIPPLE is the inductor ripple current, ∆I is the maximum
load current step change, and VEX is the allowed output voltage
excursion in the transient.
where IL is the inductor current and the d is the duty cycle. The
maximum value, when d = 50%, IRMS = 0.5IL . For 12V input and
output in the range of 3V, the required RMS current is very close
to 0.43IL.
SOFT-START CAPACITOR
The value of the soft-start capacitor determines how fast the
output voltage rises and how large the inductor current is required
to charge the output capacitor. The output voltage will follow the
voltage at SS pin if the required inductor current does not exceed
the maximum current in the inductor. The SS pin voltage can be
expressed as:
VSS = VSET (1 − e −t / RSSCSS )
where VSET is the reference voltage. RSS and CSS are soft start
resistor and capacitor. The required inductor current for the output
capacitor to follow the SS-pin voltage equals the required capacitor
current plus the load current. The soft-start capacitor should be
selected so that the overall inductor current does not exceed its
maximum.
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Integrated Products Division
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Page 6
APPLICATIONS
The output capacitor is sized to meet ripple and transient
performance specifications. Effective Series Resistance (ESR) is a
critical parameter. When a step load current occurs, the output
voltage will have a step that equals the product of the ESR and the
current step, ∆I. In an advanced microprocessor power supply, the
output capacitor is usually selected for ESR instead of capacitance
or RMS current capability. A capacitor that satisfies the ESR
requirement usually has a larger capacitance and current capability
than strictly needed. The allowed ESR can be found by:
Copyright © 2000
Rev. 1.0c, 2005-03-08
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APPLICATION INFORMATION
LX1684
Voltage-Mode PWM Controller
®
TM
P RODUCTION D ATA S HEET
SOFT-START CAPACITOR (continued)
OUTPUT ENABLE
The capacitor current to follow the SS-pin voltage is:
The LX1684 FET driver outputs are driven to ground by pulling
the soft-start pin below 0.3V.
ICOUT = COUT
dV COUT −(t / RSSCSS )
=
×e
dt
CSS
PROGRAMMING THE OUTPUT VOLTAGE
where COUT is the output capacitance. The typical value of CSS
should be in the range of 0.1 to 0.2µF.
During the soft-start interval the load current from a
microprocessor is negligible; therefore, the capacitor current is
approximately the required inductor current.
The output voltage is sensed by the feedback pin (V FB ) which
has a 1.25V reference. The output voltage can be set to any voltage
above 1.25V (and lower than the input voltage) by means of a
resistor divider (see Product Highlight).
VOUT = VREF (1 +
OVER-CURRENT PROTECTION
Current limiting occurs at current level ICL , when the voltage
detected by the current sense comparator is greater than the current
sense comparator threshold, VTRIP (400mV).
ICL × RDS (ON ) + ISET × RSET = VTRIP
So,
RSET =
VTRIP − ICL × RDS (ON )
ISET
RSET =
400mV − ICL × RDS (ON )
45µA
Example:
For 10A current limit, using SUB45N05-20L MOSFET (20mΩ
RDS(ON) ):
RSET =
0.4 − 10 × 0.020
= 4.42kΩ
45 × 10 −6
Current Sensing Using Sense Resistor
The method of current sensing using the RDS(ON) of the upper
MOSFET is economical, but can have a large tolerance, since the
RDS(ON) can vary with temperature, etc. A more accurate alternative
is to use an external sense resistor (RSENSE ). The sense resistor
could be a PCB trace (for construction details, see Application
Note AN-10 or LX1668 data sheet). The over-current trip point is
calculated as in the equations above, replacing RDS(ON) with RSENSE .
RSET
VTRIP − ( ICL × RSENSE )
=
ISET
RSET =
Copyright © 2000
Rev. 1.0c, 2005-03-08
0.4 − 10 × 0.005
= 7.8kΩ
45 × 10 −6
Note: Keep R 1 and R 2 close to 100Ω (order of magnitude).
FET SELECTION
To insure reliable operation, the operating junction temperature
of the FET switches must be kept below certain limits. The Intel
specification states that 115°C maximum junction temperature
should be maintained with an ambient of 50°C. This is achieved by
properly derating the part, and by adequate heat sinking. One of the
most critical parameters for FET selection is the RDS(ON) resistance.
This parameter directly contributes to the power dissipation of the
FET devices, and thus impacts heat sink design, mechanical layout,
and reliability. In general, the larger the current handling capability
of the FET, the lower the R DS(ON) will be, since more die area is
available.
TABLE 1 - FET Selection Guide
This table gives selection of suitable FETs from VISHAY
RDS(ON)
RDS(ON)
VDS
Device
@4.5V
@10V
(V)
(mΩ)
(mΩ)
2
D PAK and TO-220
SUB70N03-09BP
30
13
9
SUB45N03-13L
30
20
13
SUB45N05-20L
40
20
18
SUB70N04-10
40
14
10
SO-8
Si4810DY
30
20
13.5
Si4812DY
30
28
18
Gate
Charge
typ(nC)
15.5
45
43
50
20
16
Heat Dissipated In Upper MOSFET
The heat dissipated in the top MOSFET will be:
PD = ( I 2 × RDS (ON ) × Duty Cycle ) + (0.5 × I × VIN × tSW × fS )
Where tSW is switching transition line for body diode (~100ns) and
fS is the switching frequency.
For the SUB70N03-09BP (13mΩ RDS(ON) ), converting 12V to
3.3V at 15A will result in typical heat dissipation of 2.6W.
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Integrated Products Division
11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570
Page 7
APPLICATIONS
Example:
For 10A current limit, using a 5mΩ sense resistor:
R1
)
R2
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APPLICATION INFORMATION
LX1684
Voltage-Mode PWM Controller
®
TM
P RODUCTION D ATA S HEET
FET SELECTION (continued)
Synchronous Rectification – Lower MOSFET
The lower pass element can be either a MOSFET or a Schottky
diode. The use of a MOSFET (synchronous rectification) will
result in higher efficiency, but at higher cost than using a Schottky
diode (non-synchronous). Power dissipated in the bottom
MOSFET will be:
If the FET switches have been carefully selected, external
heatsinking is generally not required. However, this means that
copper trace on the PC board must now be used. This is a potential
trouble spot; as much copper area as possible must be dedicated to
heatsinking the FET switches, and the diode as well if a nonsynchronous solution is used..
+12V Input
PD = I 2 × RDS (ON ) × [1 − Duty Cycle ] = 3.26W
[SUB45N03-13L or 2.12W for the SUB70N03-09BP]
Non-Synchronous Operation - Schottky Diode
A typical Schottky diode, with a forward drop of 0.6V will
dissipate 0.6 * 15 * [1 – 3.3/12] = 6.5W (compared to the 2.1 to
4.2W dissipated by a MOSFET under the same conditions). This
power loss becomes much more significant at lower duty cycles.
The use of a dual Schottky diode in a single TO-220 package (e.g.
the MBR2535) helps improve thermal dissipation.
LAYOUT GUIDELINES - THERMAL DESIGN
A great deal of time and effort were spent optimizing the
thermal design of the demonstration boards. Any user who intends
to implement an embedded motherboard would be well advised to
carefully read and follow these guidelines.
Copyright © 2000
Rev. 1.0c, 2005-03-08
Output
LX1684
GND
FIGURE 2 — Key Power PCB Traces
General Notes
As always, be sure to provide local capacitive decoupling close
to the chip. Be sure use ground plane construction for all highfrequency work. Use low ESR capacitors where justified, but be
alert for damping and ringing problems. High-frequency designs
demand careful routing and layout, and may require several
iterations to achieve desired performance levels.
Power Traces
To reduce power losses due to ohmic resistance, careful consideration should be given to the layout of traces that carry high
currents. The main paths to consider are:
ƒ
Input power from 12V supply to drain of top MOSFET.
ƒ
Trace between top MOSFET and lower MOSFET or
Schottky diode.
ƒ
Trace between lower MOSFET or Schottky diode and
ground.
ƒ
Trace between source of top MOSFET and inductor and
load.
All of these traces should be made as wide and thick as possible,
in order to minimize resistance and hence power losses. It is also
recommended that, whenever possible, the ground, input and
output power signals should be on separate planes (PCB layers).
See Figure 2 – bold traces are power traces.
Microsemi
Integrated Products Division
11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570
Page 8
APPLICATIONS
Boost Operation
The LX1684 needs a secondary supply voltage (VC1) to provide
sufficient drive to the upper MOSFET. The top FET must be a
logic level power MOSFET such as SUB45N03-13L. It must be able
to turn on to a low RDS(ON) with VGS of 4.5V or higher. VC1 can be
generated using a bootstrap (charge pump) circuit, as shown in the
Product Highlight on page 1. The capacitor, (C1) is alternatively
charged up from 5V via the Schottky diode, (D1), and then boosted
up when the FET is turned on. Under any circumstance the voltage
at VC1 should not be more than 18V for more than 300nS and must
not be greater than 19V for more than 50nS. Lab evaluation and
module production test should be the final arbiter to verify the
proper operation. For application with a large MOSFET, the
maximum voltage at VC1 should be kept lower due to thermal
dissipation in the FET driver section. It is inherent in a higher
current power supply that the parasitic inductance and capacitance
on PCB board and Power MOSFET device induces ringing at the
gate drive. The extra thermal dissipation and the higher peak
voltage generated by gate ringing should be taken in account
during final design. The temperature rise due to gate drive thermal
dissipation can be reduced by extra heat sinking. A resistor in
series with the gate in order of 10ohm or snubber circuitry can
reduce the gate ringing. The voltage must provide sufficient gate
drive to the external MOSFET in order to get a low RDS(ON) and
MUST be lower than maximum voltage rating of 18V.
Note that using the bootstrap circuit in synchronous rectification
mode is likely to result in faster turn-on than in non-synchronous
mode.
WWW . Microsemi .C OM
APPLICATION INFORMATION
LX1684
Voltage-Mode PWM Controller
®
TM
P RODUCTION D ATA S HEET
10.8V - 13.2V
5V
C5
1µF
R1
300 Ohm
R2
100 Ohm
Css
0.1µF
FB
NC
VCC
NC
SS
CSP
CS
NC
GND
VC1
PGND
BDR
2x
820µF,
16V
RSET
2K
Q1
L1
10µH
C6
1µF
NC
TDR
MBR2545
Figure 3
– 12V to 5V with P-MOSFET.
3.3V
10%
C5
1µF
2
3
4
R2
50K Ohm
Css
0.1µF
5
6
7
VCC
NC
NC
SS
CSP
CS
NC
GND
VC1
PGND
BDR
NC
TDR
C1
0.1µF
14
13
4x
1500
µF
No Charge Pump is Needed.
R1
150 Ohm
FB
VOUT
L1 is powder
iron Toroid
5V
5%
1
WWW . Microsemi .C OM
TYPICAL APPLICATION
RSET
12 3.9K
2x
150µF,
6.3V
D1
C4
5817SMT
11
10
9
Q1
SUB45N
05-20L
L1 is a powdered
iron toroid
8
L1
5µH
2x
270µF
2V
APPLICATIONS
Q2
SUB45N0520L
VOUT
1.25V
10A
C3
C3 Panasonic EEF-UE0D271R 270uF 3A 15mOhm ESR
C4 Panasonic EEF-UE0J151R 150uF 3A 18mOhm ESR
Figure 4
Copyright © 2000
Rev. 1.0c, 2005-03-08
– 3.3V to 1.25V with Charge Pump.
Microsemi
Integrated Products Division
11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570
Page 9
LX1684
Voltage-Mode PWM Controller
®
TM
P RODUCTION D ATA S HEET
OUTPUT VOLTAGE VS. LOAD
EFFICENCY VS. OUTPUT CURRENT
1.00
3.320
Output Voltage (V)
0.95
Efficieny
WWW . Microsemi .C OM
CHARACTERIZATION CHARTS
SEE PRODUCT HIGHLIGHT SCHEMATIC
Typical 12V to 3.3V application
0.90
0.85
0.80
3.315
3.310
3.305
0.75
0.70
3.300
0
1
2
3
4
5
6
7
8
9
10
0
2
Output Current (A)
4
6
8
10
12
Output Current (A)
OUTPUT STARTUP
OUTPUT VOLTAGE VS. TEMPERATURE
3.34
VOUT = 3.3V
Output Voltage(V)
3.33
2V /
DIV
3.32
3.31
3.30
VCC = 5VDC
3.29
2V /
DIV
3.28
20
30
40
50
60
70
80
90
Temperature (°C)
1ms / DIV
OUTPUT VOLTAGE VS. TEMPERATURE
OUTPUT VOLTAGE VS. TEMPERATURE
VOUT = 2VDC
100mV
/ DIV
100mV
/ DIV
IOUT = 10A
CHARTS
IOUT = 5A
25A /
DIV
10µs / DIV
100µs / DIV
Copyright © 2000
Rev. 1.0c, 2005-03-08
Microsemi
Integrated Products Division
11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570
Page 10
LX1684
Voltage-Mode PWM Controller
®
TM
P RODUCTION D ATA S HEET
D
WWW . Microsemi .C OM
PACKAGE DIMENSIONS
14-Pin Plastic SOIC
F
B
1
2
P
3
D
G
A
L
C
K
Dim
M
INCHES
MIN
MAX
0.336 0.344
0.150 0.155
0.053 0.069
0.013 0.020
0.030
0.050 BSC
0.007 0.010
0.004 0.010
0.189 0.205
0
8
0.228 0.244
0.004
MECHANICALS
A
B
C
D
F
G
J
K
L
M
P
*LC
MILLIMETERS
MIN
MAX
8.54
8.74
3.81
3.94
1.35
1.75
0.33
0.51
0.77
1.27 BSC
0.19
0.25
0.10
0.25
4.82
5.21
0
8
5.79
6.20
0.10
J
*Lead Coplanarity
Note:
1. Dimensions do not include mold flash or protrusions; these shall not exceed 0.155mm(.006”) on any side. Lead dimension shall not
include solder coverage.
Copyright © 2000
Rev. 1.0c, 2005-03-08
Microsemi
Integrated Products Division
11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570
Page 11
LX1684
TM
Voltage-Mode PWM Controller
®
P RODUCTION D ATA S HEET
WWW . Microsemi .C OM
NOTES
NOTES
PRODUCTION DATA – Information contained in this document is proprietary to
Microsemi and is current as of publication date. This document may not be modified in
any way without the express written consent of Microsemi. Product processing does not
necessarily include testing of all parameters. Microsemi reserves the right to change the
configuration and performance of the product and to discontinue product at any time.
Copyright © 2000
Rev. 1.0c, 2005-03-08
Microsemi
Integrated Products Division
11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570
Page 12