ANALOGICTECH CBC2518T

AAT2513
Dual 600mA Step-Down
Converter with Synchronization
General Description
Features
The AAT2513 is a high efficiency dual synchronous
step-down converter for applications where power
efficiency, thermal performance and solution size
are critical. Input voltage ranges from 2.7V to 5.5V,
making it ideal for systems powered by single-cell
lithium-ion/polymer batteries.
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Each converter is capable of 600mA output current
and has its own enable pin. Efficiency of the converters is optimized over full load range. Total no
load quiescent current is 60µA, allowing high efficiency even under light load conditions.
The integrated power switches are controlled by
pulse width modulation (PWM) with a 1.7MHz typical switching frequency at full load, which minimizes
the size of external components. Fixed frequency,
low noise operation can be forced by a logic signal
on the MODE pin. Furthermore, an external clock
can be used to synchronize the switching frequency
of both converters.
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•
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•
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•
•
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SystemPower™
VIN Range: 2.7V to 5.5V
Output Current:
— Channel 1: 600mA
— Channel 2: 600mA
96% Efficient Step-Down Converter
Low No Load Quiescent Current
— 60µA Total for Both Converters
Integrated Power Switches
100% Duty Cycle
1.7MHz Switching Frequency
Optional Fixed Frequency or External SYNC
Logic Selectable 180° Phase Shift Between
the Two Converters
Current Limit Protection
Automatic Soft-Start
Over-Temperature Protection
QFN33-16 Package
-40°C to +85°C Temperature Range
A phase shift pin (PS) is available to operate the
www.DataSheet4U.com Applications
two converters 180° out of phase at heavy load
to
achieve low input ripple.
• Cellular Phones / Smart Phones
The AAT2513 is available in a Pb-free, thermally
• Digital Cameras
enhanced 16-pin QFN33 package and is specified
• Handheld Instruments
for operation over the -40°C to +85°C temperature
• Micro Hard Disc Drives
range.
• Microprocessor / DSP Core / IO Power
• PDAs and Handheld Computers
Typical Application
Input:
2.7V to 5.5V
CIN
1μF
L1
VIN1
VIN2
VOUT1
LX1
2μH
R1
VCC
FB1
AAT2513
L2
VOUT2
R2
LX2
2μH
MODE/SYNC
PS
C1
4.7μF
R4
AGND
2513.2007.04.1.1
R3
FB2
EN1
EN2
C2
4.7μF
PGND1 PGND2
1
AAT2513
Dual 600mA Step-Down
Converter with Synchronization
Pin Descriptions
Pin #
Symbol
1
PS
2
AGND
4, 3
FB1, FB2
5,
6,
7,
8,
16
15
14
13
VIN1, VIN2
N/C
LX1, LX2
PGND1, PGND2
10, 9
EN1, EN2
11
12
VCC
MODE/SYNC
EP
Function
Phase shift pin. Logic high enables the PS feature which forces the two converters
to operate 180° out of phase when both are in forced PWM mode.
Analog ground. Return the feedback resistive divider to this ground. See section on
PCB layout guidelines and evaluation board layout diagram.
Feedback input pins. An external resistive divider ties to each and programs the
respective output voltage to the desired value.
Input supply voltage pins. Must be closely decoupled to the respective PGND.
Not connected
Output switching nodes that connect to the respective output inductor.
Main power ground return. Connect to the input and output capacitor return. See
section on PCB layout guidelines and evaluation board layout diagram.
Converter enable input pins. A logic high enables the converter channel. A logic low
forces the channel into shutdown mode, reducing the channel supply current to less
than 1µA. This pin should not be left floating. When not actively controlled, this pin
can be tied directly to VIN and/or VCC.
Control circuit power supply. Connect to the higher voltage of VIN1 or VIN2.
Logic low enables automatic light load mode for optimized efficiency throughout the
entire load range. Logic high forces low noise PWM operation under all operating
conditions. Connect to an external clock for synchronization (PWM only).
Exposed paddle (bottom). Use properly sized vias for thermal coupling to the
ground plane. See section on PCB layout guidelines.
Pin Configuration
QFN33-16
(Top View)
PGND2
LX2
N/C
VIN2
13
14
15
16
PS
AGND
FB2
FB1
1
12
2
11
3
10
4
9
MODE/SYNC
VCC
EN1
EN2
8
7
6
5
PGND1
LX1
N/C
VIN1
2
2513.2007.04.1.1
AAT2513
Dual 600mA Step-Down
Converter with Synchronization
Absolute Maximum Ratings1
TA = 25°C unless otherwise noted.
Symbol
Description
Value
Units
VIN1/2
GND, PGND1/2
EN1/2, SYNC,
LX1/2, FB1/2, PS
TJ
TS
TLEAD
Input Voltage
Ground Pins
-0.3 to 6.0
-0.3 to +0.3
V
V
-0.3 to VCC + 0.3
V
-40 to 150
-65 to 150
300
°C
°C
°C
Value
Units
50
2
°C/W
W
Maximum Rating
Operating Temperature Range
Storage Temperature Range
Maximum Soldering Temperature (at leads, 10 sec)
Thermal Information
Symbol
θJA
PD
Description
Thermal Resistance
Maximum Power Dissipation
1. Stresses above those listed in Absolute Maximum Ratings may cause permanent damage to the device. Functional operation at conditions other than the operating conditions specified is not implied. Only one Absolute Maximum rating should be applied at any one time.
2513.2007.04.1.1
3
AAT2513
Dual 600mA Step-Down
Converter with Synchronization
Electrical Characteristics1
VIN = VCC = 3.6V, TA = -40°C to +85°C, unless noted otherwise. Typical values are at TA = 25°C.
Symbol
Description
Conditions
Power Supply
VCC,
Input Voltage
VIN1, VIN2
UVLO
Under-Voltage Lockout
IQ
Quiescent Current
ISHDN
Shutdown Current
Each Converter
VFB
Feedback Voltage Tolerance
VOUT
Output Voltage Range
LX Reverse Leakage Current
(Fixed)
LX Leakage Current
Feedback Leakage
P-Channel Current Limit
High Side Switch On Resistance
Low Side Switch On Resistance
ILX_LEAK
ILX_LEAK
IFB
ILIM
RDS(ON)H
RDS(ON)L
ΔVOUT/
VOUT/ΔIOUT
ΔVOUT/
VOUT/ΔVIN
VFB
FOSC
TS
Logic
TSD
THYS
VIL
VIH
IEN,
IMODE/SYNC,
IPS
Min
Typ
2.7
VCC Rising
VCC Falling
VEN1 = VEN2 = VCC, No Load
EN1 = EN2 = GND
Max
Units
5.5
V
2.7
2.35
60
V
120
1.0
µA
µA
-3.0
-3.0
%
0.6
VIN
V
VIN Open, VLX = 5.5V, EN = GND
1.0
µA
VIN = 5.5V, VLX = 0 to VIN
VFB = 1.0V
Each Converter
1.0
0.2
1.0
0.45
0.40
µA
µA
A
Ω
Ω
IOUT = 0 to 600mA, VIN = 2.9 to 5.5V
IOUT = 0 to 450mA, VIN = 2.7 to 5.5V
Load Regulation
ILOAD = 0 to 600 mA
0.002
%/mA
Line Regulation
VIN = 2.7 to 5.5V, ILOAD = 100 mA
0.125
%/V
0.591 0.600 0.609
V
1.7
MHz
150
µs
140
°C
15
°C
Feedback Threshold Voltage
Accuracy
Oscillator Frequency
Start-Up Time
No Load, TA = 25°C
From Enable to Output Regulation;
Both Channels
Over-Temperature Shutdown
Threshold
Over-Temperature Shutdown
Hysteresis
EN, MODE/SYNC, PS Logic
Low Threshold
EN, MODE/SYNC, PS Logic
High Threshold
Logic Input Current
0.6
1.4
VIN = VFB = 5.5V
-1.0
V
V
1.0
µA
1. The AAT2513 guaranteed to meet performance specifications over the -40°C to +85°C operating temperature range and is assured
by design, characterization and correlation with statistical process controls.
4
2513.2007.04.1.1
AAT2513
Dual 600mA Step-Down
Converter with Synchronization
Electrical Characteristics
Efficiency vs. Load
DC Regulation
(VOUT = 3.3V; L = 4.7µH; LL Mode)
(VIN = 5.0V; VOUT = 3.3V; L = 4.7µH; LL Mode)
100
1.00
VIN = 3.6V
0.75
Output Error (%)
Efficiency (%)
90
80
VIN = 4.2V
70
VIN = 5.0V
60
50
40
0.50
0.25
0.00
-0.25
-0.50
-0.75
30
0.1
1
10
100
-1.00
0.1
1000
1
Efficiency vs. Load
DC Regulation
(VIN = 3.3V to 5.5V; VOUT = 2.5V; L = 3.3µH; LL Mode)
2.0
VIN = 2.7V
1.5
Output Error (%)
Efficiency (%)
1000
(VOUT = 2.5V; L = 3.3µH; LL Mode)
100
80
VIN = 3.6V
70
VIN = 4.2V
60
50
VIN = 5.0V
40
1.0
0.5
0.0
-0.5
-1.0
-1.5
-2.0
0.1
30
0.1
1
10
100
1000
1
Output Current (mA)
10
100
1000
Output Current (mA)
Efficiency vs. Load
DC Regulation
(VOUT = 1.8V; L = 2.2µH; LL Mode)
(VOUT = 1.8V; L = 2.2µH; LL Mode)
1.0
100
90
0.8
VIN = 2.7V
Output Error (%)
Efficiency (%)
100
Output Current (mA)
Output Current (mA)
90
10
80
70
VIN = 3.6V
60
VIN = 4.2V
50
40
VIN = 5.0V
30
VIN = 5.0V
0.6
0.4
VIN = 4.2V
0.2
0.0
-0.2
-0.4
VIN = 3.3V
-0.6
-0.8
20
-1.0
0.1
1
10
Output Current (mA)
2513.2007.04.1.1
100
1000
0.1
1
10
100
1000
Output Current (mA)
5
AAT2513
Dual 600mA Step-Down
Converter with Synchronization
Electrical Characteristics
Efficiency vs. Load
DC Regulation
(VOUT = 1.5V; L = 2.2µH; LL Mode)
(VOUT = 1.5V; L = 2.2µH; LL Mode)
1.0
100
0.8
VIN = 2.7V
80
Output Error (%)
Efficiency (%)
90
70
60
VIN = 3.6V
50
VIN = 4.2V
40
30
VIN = 3.3V
0.6
0.4
VIN = 4.2V
0.2
0.0
VIN = 5.0V
-0.2
-0.4
-0.6
-0.8
-1.0
20
0.1
1
10
100
0.1
1000
1
Output Current (mA)
10
100
1000
Output Current (mA)
Switching Frequency vs. Temperature
Switching Frequency vs. Input Voltage
4
Frequency Variation (%)
Switching Frequency (MHz)
(IOUT = 600mA; 25°C)
1.90
VIN = 4.2V
1.85
1.80
1.75
1.70
VIN = 3.6V
1.65
1.60
1.55
-40
3
2
VOUT = 1.5V
1
VOUT = 1.8V
0
-1
-2
VIN = 3.3V
VIN = 2.5V
-3
-4
-20
0
20
40
60
80
100
120
2.7
3.1
3.5
Temperature (°°C)
3.9
4.3
4.7
5.1
5.5
Input Voltage (V)
Output Voltage Error Vs. Temperature
No Load Quiescent Current vs. Input Voltage
(VOUT = 2.5V; IOUT = 600mA)
70
VIN = 3.6V
0.25
0.20
0.15
0.10
VIN = 4.2V
0.05
0.00
-40
-20
0
20
40
60
Temperature (°°C)
6
Input Current (µA)
Output Voltage Error (%)
0.30
80
100
120
65
60
55
85°C
25°C
-40°C
50
45
2.5
3
3.5
4
4.5
5
5.5
6
Input Voltage (V)
2513.2007.04.1.1
AAT2513
Dual 600mA Step-Down
Converter with Synchronization
Electrical Characteristics
P-Channel RDS(ON) vs. Input Voltage
VIH vs. Input Voltage
1000
1.3
1.2
120°C
100°C
800
1.1
85°C
VIH (V)
RDS(ON) (mΩ
Ω)
900
700
600
0.9
0.8
400
0.7
25°C
25°C
1.0
500
300
2.5
-40°C
85°C
0.6
3
3.5
4
4.5
5
5.5
6
2.5
Input Voltage (V)
3.0
3.5
4.0
4.5
5.0
5.5
6.0
Input Voltage (V)
VIL vs. Input Voltage
Soft Start
(VIN = 3.6V; VOUT = 1.8V; IOUT = 600mA)
Enable Voltage (top) (V)
Output Voltage (middle) (V)
VIL (mV)
1.1
1.0
25°C
-40°C
0.9
0.8
85°C
0.7
0.6
2.5
3.0
3.5
4.0
4.5
5.0
5.5
4
3
2
1
0
0.4
0.2
0.0
-0.2
6.0
Time (50µs/div)
Input Voltage (V)
Load Transient
(1mA to 450mA; VIN = 3.6V; VOUT = 1.8V; COUT = 10µF; CFF = 100pF)
1.8
450mA
1mA
0.5
0
Time (20µs/div)
2513.2007.04.1.1
2.0
1.8
1.6
450mA
1mA
0.5
0.0
-0.5
Load Current (middle) (A)
Inductor Current (bottom) (A)
2.0
Output Voltage (AC) (top) (V)
Load Transient
(1mA to 450mA; VIN = 3.6V; VOUT = 1.8V; COUT = 4.7µF)
Load Current (middle) (A)
Inductor Current (bottom) (A)
Output Voltage (top) (V)
0.6
Inductor Current (bottom) (A)
1.2
Time (20µs/div)
7
AAT2513
Dual 600mA Step-Down
Converter with Synchronization
Electrical Characteristics
Load Transient
Load Transient
(5mA to 600mA; VIN = 3.6V; VOUT = 1.8V; COUT = 4.7µF)
(1mA to 600mA; VIN = 3.6V; VOUT = 1.8V; COUT = 10µF; CFF = 100pF)
1.3
600mA
5mA
1.0
0.5
0.0
-0.5
Output Voltage (top) (V)
Output Voltage (top) (V)
1.8
2.0
1.8
1.6
600mA
1mA
0.5
0
Time (40µs/div)
Load Current (middle) (A)
Inductor Current (bottom) (A)
2.3
Load Current (middle) (A)
Inductor Current (bottom) (A)
2.8
Time (40µs/div)
Load Transient
(450mA to 600mA; VIN = 3.6V; VOUT = 1.8V; COUT = 10µF; CFF = 100pF
Output Voltage (top) (V)
1.6
600mA
450mA
0.6
0.4
0.2
Time (20µs/div)
2.0
1.9
1.8
1.7
600mA
450mA
0.6
0.4
0.2
Load Current (middle) (A)
Output Current (bottom) (A)
1.8
Load Current (middle) (A)
Inductor Current (bottom) (A)
2.0
Output Voltage (AC) (top) (V)
Load Transient
(450mA to 600mA; VIN = 3.6V; VOUT = 1.8V; COUT = 4.7µF)
Time (20µs/div)
Line Transient
(VIN = 3.6V to 4.2V; VOUT = 1.8V; IOUT = 600mA; COUT = 4.7µF)
Input Voltage (top) (V)
4
3
2
1.84
1
1.82
1.80
1.78
1.76
1.74
Output Voltage (bottom) (V)
5
Time (40µs/div)
8
2513.2007.04.1.1
AAT2513
Dual 600mA Step-Down
Converter with Synchronization
Electrical Characteristics
Line Regulation
Line Regulation
(VOUT = 1.8V; L = 2.2µH)
(VOUT = 1.5V; L = 2.2µH)
2.0
1.0
1.5
IOUT = 0.1mA to 100mA
Accuracy (%)
Accuracy (%)
0.5
0.0
-0.5
IOUT = 400mA
-1.0
-1.5
-2.0
2.5
IOUT = 0.1mA to 100mA
1.0
0.5
0.0
-0.5
IOUT = 400mA
-1.0
-1.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
-2.0
2.5
4.0
4.5
5.0
5.5
Output Voltage Ripple
(VOUT = 1.8V; VIN = 3.6V; Load = 600mA)
6.0
1.80
1.75
0.2
0.1
0.0
1.82
1.80
1.78
0.7
0.6
0.5
0.4
Time (10µs/div)
Time (0.2µs/div)
Input Ripple
Input Ripple
(CIN = 2 x 10µF; VIN = 3.6V; VOUT1 = 1.8V; VOUT2 = 2.5V;
IOUT1,2 = 600mA; 0°° Phase Shift; PS = Low)
(CIN = 2 x 10µF; VIN = 3.6V; VOUT1 = 1.8V;
VOUT2 = 2.5V; IOUT1,2 = 600mA; 180°° Phase Shift)
3.59
LX2
4
LX1
2
0
3.60
3.59
LX2
4
LX1
0
-2
-2
Time (0.2µs/div)
2
Switching Voltage
LX1,LX2 (V)
3.60
Switching Voltage
LX1,LX2 (V)
3.61
Input Voltage (top) (V)
3.61
3.62
2513.2007.04.1.1
Inductor Current (bottom) (A)
1.85
Output Voltage (top) (V)
Output Voltage Ripple
(VOUT = 1.8V; VIN = 3.6V; Load = 1mA)
-0.1
Input Voltage (top) (V)
3.5
Input Voltage (V)
Inductor Current (bottom) (A)
Output Voltage (top) (V)
Input Voltage (V)
3.0
Time (0.2µs/div)
9
AAT2513
Dual 600mA Step-Down
Converter with Synchronization
Functional Block Diagram
FB1
VIN1
VCC
DH
Comp
Err.
Amp.
LX1
Logic
Voltage
Reference
DL
Control
Logic
EN1
PGND1
AGND
VIN2
Oscillator
MODE/SYNC
PS
FB
Err.
Amp.
DH
Comp.
LX2
Logic
Voltage
Reference
EN2
Functional Description
The AAT2513 is a peak current mode pulse width
modulated (PWM) converter with internal compensation. Each channel has independent input,
enable, feedback, and ground pins with a 1.7MHz
clock. Both converters operate in either a fixed frequency (PWM) mode or a more efficient light load
(LL) mode. A phase shift pin programs the converters to operate in phase or 180° out of phase. The
converter can also be synchronized to an external
clock during PWM operation.
The input voltage range is 2.7V to 5.5V. An external resistive divider as shown in Figure 1 programs
the output voltage up to the input voltage. The converter MOSFET power stage is sized for 600mA
load capability with up to 96% efficiency. Light load
efficiency is up to 90% at a 1mA load.
10
Control
Logic
DL
PGND2
Soft Start / Enable
The AAT2513 soft start control prevents output voltage overshoot and limits inrush current when either
the input power or the enable input is applied.
When pulled low, the enable input forces the converter into a low power non-switching state with a
bias current of less than 1µA.
Low Dropout Operation
For conditions where the input voltage drops to the
output voltage level, the converter duty cycle
increases to 100%. As the converter approaches
the 100% duty cycle, the minimum off time initially
forces the high side on time to exceed the 1.7MHz
clock cycle and reduce the effective switching frequency. Once the input drops below the level
where the converter can regulate the output, the
high side P-channel MOSFET is enabled continuously for 100% duty cycle. At 100% duty cycle the
output voltage tracks the input voltage minus the
I*R drop of the high side P-channel MOSFET.
2513.2007.04.1.1
AAT2513
Dual 600mA Step-Down
Converter with Synchronization
VIN
U1
AAT2513
C3
10μF
5
10
1.8V
11
L1
7
2.2uH
4
R1
118k
6
2
C1
4.7μF
R2
59.0k
8
VIN1
VIN2
EN1
EN2
VCC
PS
LX1
LX2
FB1
FB2
N/C
MODE/SYNC
AGND
PGND1
N/C
PGND2
16
9
2.5V
1
L2
14
2.2μH
3
R3
187k
12
15
13
C2
4.7μF
R4
59.0k
Figure 1: AAT2513 Typical Schematic.
Low Supply UVLO
Applications Information
Under-voltage lockout (UVLO) guarantees sufficient VIN bias and proper operation of all internal
circuitry prior to activation.
Inductor Selection
Fault Protection
For overload conditions, the peak inductor current
is limited. Thermal protection disables the converter when the internal dissipation or ambient temperature becomes excessive. The over-temperature
threshold for the junction temperature is 140°C with
15°C of hysteresis.
The step down converter uses peak current mode
control with slope compensation to maintain stability
for duty cycles greater than 50%. The output inductor value must be selected so the inductor current
down slope meets the internal slope compensation
requirements. The internal slope compensation for
the adjustable and low voltage fixed versions of the
AAT2513 is 0.6A/µsec. This equates to a slope compensation that is 75% of the inductor current down
slope for a 1.8V output and 2.2µH inductor.
PWM/LL Operation
For fixed frequency, with minimum ripple under
light load conditions, the MODE/SYNC pin should
be tied to a logic high. For more efficient operation
under light load conditions the MODE/SYNC pin
should be tied to a logic low level.
Clock Phase and Frequency
A logic high on the PS pin while in PWM mode
forces both converters to operate 180° out of phase
thus reducing the input ripple by roughly half. A
logic low on the PS pin synchronizes both converters in phase.
2513.2007.04.1.1
m=
0.75 ⋅ VO 0.75 ⋅ 1.8V
A
=
= 0.6
L
2.2µH
µsec
L=
0.75 ⋅ VO 0.75V ⋅ VO
µs
≈ 1.2 A ⋅ VO
=
A
m
0.6 µs
= 1.2
µs
2.5V = 3.1µH
A
In this case a standard 3.3µH value is selected.
11
AAT2513
Dual 600mA Step-Down
Converter with Synchronization
Table 1 displays the suggested inductor values for
the AAT2513.
This equation provides an estimate for the input
capacitor required for a single channel.
Manufacturer's specifications list both the inductor
DC current rating, which is a thermal limitation, and
the peak current rating, which is determined by the
inductor's saturation characteristics. The inductor
should not show any appreciable saturation under
all normal load conditions. Some inductors may
meet the peak and average current ratings yet
result in excessive losses due to a high DCR.
Always consider the losses associated with the
DCR and its effect on the total converter efficiency
when selecting an inductor.
The equation below solves for the input capacitor
size for both channels. It makes the worst case
assumption that both converters are operating at
50% duty cycle with in phase synchronization.
The 2.2uH CDRH2D11 series inductor selected
from Sumida has a 98mΩ DCR and a 1.27A DC
current rating. At full load the inductor DC loss is
35mW which corresponds to a 3.2% loss in efficiency for a 600mA, 1.8V output.
Input Capacitor
A key feature of the AAT2513 is that the fundamental switching frequency ripple at the input can
be reduced by operating the two converters 180°
out of phase. This reduces the input ripple by
roughly half, reducing the required input capacitance. An X5R ceramic input capacitor as small as
1µF is often sufficient. To estimate the required
input capacitor size, determine the acceptable
input ripple level (VPP) and solve for C. The calculated value varies with input voltage and is a maximum when VIN is double the output voltage.
CIN =
CIN =
1
⎛ VPP
⎞
- ESR · 4 · FS
⎝ IO1 + IO2
⎠
Because the AAT2513 channels will generally
operate at different duty cycles the actual ripple will
vary and be less than the ripple (VPP) used to solve
for the input capacitor in the above equation.
Always examine the ceramic capacitor DC voltage
coefficient characteristics when selecting the proper value. For example, the capacitance of a 10µF
6.3V X5R ceramic capacitor with 5V DC applied is
actually about 6µF.
The maximum input capacitor RMS current is:
IRMS = IO1 · ⎛
⎝
VO1 ⎛
V ⎞
· 1 - O1 ⎞ + IO2 · ⎛
VIN ⎝
VIN ⎠ ⎠
⎝
VO2 ⎛
V ⎞
· 1 - O2 ⎞
VIN ⎝
VIN ⎠ ⎠
The input capacitor RMS ripple current varies with
the input and output voltage and will always be less
than or equal to half of the total DC load current of
both converters combined.
VO ⎛
V ⎞
⋅ 1- O
VIN ⎝
VIN ⎠
IRMS(MAX) =
IO1(MAX) + IO2(MAX)
2
⎛ VPP
⎞
- ESR ⋅ FS
⎝ IO
⎠
Configuration
Output Voltage
Inductor
Slope Compensation
0.6V adjustable
with external
resistive divider
0.6V-2.0V
2.5V
3.3V
2.2µH
3.3µH
4.7µH
0.6A/µs
Table 1: Inductor Values.
12
2513.2007.04.1.1
AAT2513
Dual 600mA Step-Down
Converter with Synchronization
This equation also makes the worst-case assumption that both converters are operating at 50% duty
cycle synchronized.
VO
⎛
V ⎞
· 1- O
The term VIN ⎝ VIN ⎠ appears in both the input
voltage ripple and input capacitor RMS current
equations. It is at maximum when VO is twice VIN.
This is why the input voltage ripple and the input
capacitor RMS current ripple are a maximum at
50% duty cycle.
VO ⎛
V ⎞
· 1 - O = D ⋅ (1 - D) = 0.52 = 0.25
VIN ⎝
VIN ⎠
The input capacitor provides a low impedance loop
for the edges of pulsed current drawn by the
AAT2513. Low ESR/ESL X7R and X5R ceramic
capacitors are ideal for this function. To minimize
the stray inductance, the capacitor should be
placed as close as possible to the IC. This keeps
the high frequency content of the input current
localized, minimizing EMI and input voltage ripple.
The proper placement of the input capacitor (C3 and
C9) can be seen in the evaluation board layout in
Figures 3 and 4. Since decoupling must be as close
to the input pins as possible it is necessary to use
two decoupling capacitors, one for each converter.
A Laboratory test set-up typically consists of two
long wires running from the bench power supply to
the evaluation board input voltage pins. The inductance of these wires along with the low ESR ceramic input capacitor can create a high Q network that
may effect the converter performance.
This problem often becomes apparent in the form
of excessive ringing in the output voltage during
load transients. Errors in the loop phase and gain
measurements can also result.
Since the inductance of a short printed circuit board
trace feeding the input voltage is significantly lower
than the power leads from the bench power supply,
most applications do not exhibit this problem.
In applications where the input power source lead
inductance cannot be reduced to a level that does
not effect the converter performance, a high ESR
tantalum or aluminum electrolytic (C10 of Figure 2)
2513.2007.04.1.1
should be placed in parallel with the low ESR, ESL
bypass ceramic. This dampens the high Q network
and stabilizes the system.
Output Capacitor
The output capacitor limits the output ripple and
provides holdup during large load transitions. A
4.7µF to 10µF X5R or X7R ceramic capacitor typically provides sufficient bulk capacitance to stabilize the output during large load transitions and has
the ESR and ESL characteristics necessary for low
output ripple.
The output voltage droop due to a load transient is
dominated by the capacitance of the ceramic output capacitor. During a step increase in load current the ceramic output capacitor alone supplies
the load current until the loop responds. As the loop
responds the inductor current increases to match
the load current demand. This typically takes two
to three switching cycles and can be estimated by:
COUT =
3 · ΔILOAD
VDROOP · FS
Once the average inductor current increases to the
DC load level, the output voltage recovers. The
above equation establishes a limit on the minimum
value for the output capacitor with respect to load
transients.
The internal voltage loop compensation also limits
the minimum output capacitor value to 4.7µF. This
is due to its effect on the loop crossover frequency
(bandwidth), phase margin, and gain margin.
Increased output capacitance will reduce the
crossover frequency with greater phase margin.
The maximum output capacitor RMS ripple current
is given by:
IRMS(MAX) =
1
VOUT · (VIN(MAX) - VOUT)
L · F · VIN(MAX)
2· 3
·
Dissipation due to the RMS current in the ceramic
output capacitor ESR is typically minimal, resulting in
less than a few degrees rise in hot spot temperature.
13
AAT2513
Dual 600mA Step-Down
Converter with Synchronization
Adjustable Output Resistor Selection
Resistors R1 through R4 of Figure 1 program the output to regulate at a voltage higher than 0.6V. To limit
the bias current required for the external feedback
resistor string, the minimum suggested value for R2
and R4 is 59kΩ. Although a larger value will reduce
the quiescent current, it will also increase the impedance of the feedback node, making it more sensitive
to external noise and interference. Table 2 summarizes the resistor values for various output voltages
with R2 and R4 set to either 59kΩ for good noise
immunity or 221kΩ for reduced no load input current.
⎛ VOUT ⎞
⎛ 1.5V ⎞
R1 = V
-1 · R2 = 0.6V - 1 · 59kΩ = 88.5kΩ
⎝ REF ⎠
⎝
⎠
With an external feedforward capacitor (C4 and C5 of
Figure 2) the AAT2513 delivers enhanced transient
response for extreme pulsed load applications. The
addition of the feedforward capacitor typically requires
a larger output capacitor (C1 and C2) for stability.
VOUT (V)
R2, R4 = 59kΩ
R1, R3 (kΩ)
R2, R4 = 221kΩ
R1, R3 (kΩ)
0.8
0.9
1.0
1.1
1.2
1.3
1.4
1.5
1.8
1.85
2.0
2.5
3.3
19.6
29.4
39.2
49.9
59.0
68.1
78.7
88.7
118
124
137
187
265
75
113
150
187
221
261
301
332
442
464
523
715
1000
Table 2: Feedback Resistor Values.
Thermal Calculations
There are three types of losses associated with the
AAT2513 converter: switching losses, conduction
losses, and quiescent current losses. The conduction
14
losses are associated with the RDS(ON) characteristics
of the power output switching devices. The switching
losses are dominated by the gate charge of the
power output switching devices. At full load, assuming continuous conduction mode (CCM), a simplified
form of the dual converter losses is given by:
PTOTAL =
+
IO12 · (RDSON(HS) · VO1 + RDSON(LS) · [VIN -VO1])
VIN
IO22 · (RDSON(HS) · VO2 + RDSON(LS) · [VIN -VO2])
VIN
+ (tsw · F · [IO1 + IO2] + 2 · IQ) · VIN
IQ is the AAT2513 quiescent current for one channel and tSW is used to estimate the full load switching losses.
For the condition where channel one is in dropout
at 100% duty cycle the total device dissipation
reduces to:
PTOTAL = IO12 · RDSON(HS)
+
IO22 · (RDSON(HS) · VO2 + RDSON(LS) · [VIN -VO2])
VIN
+ (tsw · F · IO2 + 2 · IQ) · VIN
Since RDS(ON), quiescent current, and switching
losses all vary with input voltage, the total losses
should be investigated over the complete input
voltage range.
Given the total losses, the maximum junction temperature can be derived from the θJA for the
QFN33-12 package which is 28°C/W to 50°C/W
minimum.
TJ(MAX) = PTOTAL · ΘJA + TAMB
2513.2007.04.1.1
AAT2513
Dual 600mA Step-Down
Converter with Synchronization
PCB Layout
Use the following guidelines to insure a proper layout:
1. Due to the pin placement of VIN for both converters, proper decoupling is not possible with just
one input capacitor. The input capacitors C3 and
C9 should connect as closely as possible to the
respective VIN and GND as shown in Figure 3.
2. Connect the output capacitor and inductor as
closely as possible. The connection of the inductor
to the LX pin should also be as short as possible.
2513.2007.04.1.1
3. The feedback trace should be separate from any
power trace and connect as close as possible to
the load point. Sensing along a high-current load
trace will degrade DC load regulation. Place the
external feedback resistors as close as possible
to the FB pin. This prevents noise from being
coupled into the high impedance feedback node.
4. Keep the resistance of the trace from the load
return to GND to a minimum. This minimizes any
error in DC regulation due to potential differences of the internal signal ground and the
power ground.
5. For good thermal coupling, PCB vias are
required from the pad for the QFN paddle to the
ground plane. The via diameter should be 0.3mm
to 0.33mm and positioned on a 1.2 mm grid.
15
AAT2513
Dual 600mA Step-Down
Converter with Synchronization
Design Example
Specifications
VO1
2.5V @ 600mA (adjustable using 0.6V version), pulsed load ΔILOAD = 300mA
VO2
1.8V @ 600mA (adjustable using 0.6V version), pulsed load ΔILOAD = 300mA
VIN
2.7V to 4.2V (3.6V nominal)
FS
1.7 MHz
TAMB
85°C
1.8V VO1 Output Inductor
L1 = 1.2
µs
µs
⋅ VO1 = 1.2
⋅ 1.8V = 2.2µH (see table 1).
A
A
For Sumida CDRH2D11 2.2µH DCR = 98mΩ.
ΔI1 =
⎛ 2.5V⎞
VO1 ⎛ VO1 ⎞
2.5V
⋅ 1=
⋅ 1= 230mA
L ⋅ F ⎝ VIN ⎠
3.3µH ⋅ 1.7MHz ⎝ 4.2V⎠
IPK1 = IO1 +
ΔI1
= 0.4A + 0.115A = 0.515A
2
PL1 = IO12 ⋅ DCR = 0.6A2 ⋅ 123mΩ = 44mW
2.5V VO2 Output Inductor
L1 = 1.2
µs
µs
⋅ VO1 = 1.2
⋅ 2.5V = 3.3µH (see table 1).
A
A
For Sumida inductor CDRH2D11 3.3µH DCR = 123mΩ.
ΔI2 =
⎛ 2.5V⎞
VO2 ⎛ VO2 ⎞
2.5V
⋅ 1=
⋅ ⎝1 = 230mA
L ⋅ F ⎝ VIN ⎠
3.3µH ⋅ 1.7MHz
4.2V⎠
IPK2 = IO2 +
ΔI2
= 0.4A + 0.115A = 0.515A
2
PL2 = IO22 ⋅ DCR = 0.6A2 ⋅ 123mΩ = 44mW
16
2513.2007.04.1.1
AAT2513
Dual 600mA Step-Down
Converter with Synchronization
1.8V Output Capacitor
COUT =
3 · ΔILOAD
3 · 0.3A
=
= 4.8µF
0.2V · 1.7MHz
VDROOP · FS
IRMS(MAX) =
(VOUT) · (VIN(MAX) - VOUT)
1
1.8V · (4.2V - 1.8V)
·
= 31mArms
=
L · F · VIN(MAX)
2 · 3 2.2µH · 1.7MHz · 4.2V
2· 3
1
·
Pesr = esr · IRMS2 = 5mΩ · (31mA)2 = 4.8µW
2.5V Output Capacitor
COUT =
3 · ΔILOAD
3 · 0.3A
=
= 4.8µF
0.2V · 1.7MHz
VDROOP · FS
IRMS(MAX) =
(VOUT) · (VIN(MAX) - VOUT)
1
2.5V · (4.2V - 2.5V)
·
= 67mArms
=
L · F · VIN(MAX)
2 · 3 3.3µH · 1.7MHz · 4.2V
2· 3
1
·
Pesr = esr · IRMS2 = 5mΩ · (67mA)2 = 22µW
Input Capacitor
Input Ripple VPP = 25mV.
CIN =
1
1
=
= 10µF
⎛ VPP
⎞
⎛ 25mV
⎞
- ESR · 4 · FS
- 5mΩ · 4 · 1.7MHz
⎝ IO1 + IO2
⎠
⎝ 1.2A
⎠
IRMS(MAX) =
IO1 + IO2
= 0.6Arms
2
P = esr · IRMS2 = 5mΩ · (0.6A)2 = 0.8mW
2513.2007.04.1.1
17
AAT2513
Dual 600mA Step-Down
Converter with Synchronization
AAT2513 Losses
The maximum dissipation occurs at dropout where VIN = 2.7V. All values assume an 85°C ambient and a 120°C
junction temperature.
PTOTAL =
IO12 · (RDSON(HS) · VO1 + RDSON(LS) · (VIN -VO1)) + IO22 · (RDSON(HS) · VO2 + RDSON(LS) · (VIN -VO2))
VIN
+ (tsw · F · IO2 + 2 · IQ) · VIN
=
0.62 · (0.725Ω · 2.5V + 0.7Ω · (2.7V - 2.5V)) + 0.62 · (0.725Ω · 1.8V + 0.7Ω · (2.7V - 1.8V))
2.7V
+ (5ns · 1.7MHz · 0.6A + 60µA) · 2.7V = 533mW
TJ(MAX) = TAMB + ΘJA · PLOSS = 85°C + (50°C/W) · 533mW = 111°C
TJ(MAX) = TAMB + ΘJA · PLOSS = 85°C + (28°C/W) · 533mW = 100°C
Phase Shift
3
2
1
L1, L2 CDRH2D11
C1, C2 4.7μF 10V 0805 X5R
U1
AAT2513
VIN
6
C4
100pF
VIN
C5
100pF
16
1
R1
187k
R3
88.7k
2
3
4
C10
120μF
5
R2
59.0k
R4
59.0k
7
C9
10μF
LX2
N/C
N/C
VIN2
LX2
PS
AGND
PGND2
MODE/SYNC
FB2
VCC
FB1
EN1
VIN1
LX1
EN2
PGND1
VO2
VIN
15
L2
Sync
14
C7
1μF
13
C2
12
11
1
2
3
VCC
LX1
10
C8
0.1μF
9
L1
VO1
8
C6
1μF
C1
C3
10μF
GND
GND
R5
10
GND
VCC
VIN
3 2 1
3 2 1
Enable 2
Enable 1
GND
GND
Figure 2: AAT2513 Evaluation Board Schematic1.
1. For enhanced transient configuration C5, C4 = 100pF and C1, C2 = 10µF.
18
2513.2007.04.1.1
AAT2513
Dual 600mA Step-Down
Converter with Synchronization
Adjustable Version
(0.6V device)
R2, R4 = 59kΩ
R2, R4 = 221kΩ1
VOUT (V)
R1, R3 (kΩ)
R1, R3 (kΩ)
L1, L2 (µH)
0.8
0.9
1.0
1.1
1.2
1.3
1.4
1.5
1.8
1.85
2.0
2.5
3.3
19.6
29.4
39.2
49.9
59.0
68.1
78.7
88.7
118
124
137
187
265
75.0
113
150
187
221
261
301
332
442
464
523
715
1000
1.0 - 1.5
1.0 - 1.5
1.0 - 1.5
1.0 - 1.5
1.0 - 1.5
1.0 - 1.5
2.2
2.2
2.2
2.2
3.3
3.3
4.7
Fixed Version
R2, R4 not used
VOUT (V)
R1, R3 (kΩ)
L1, L2 (µH)
0.6-3.3V
zero
2.2
Table 5: Evaluation Board Component Values.
Figure 3: AAT2513 Evaluation Board
Top Side.
Figure 4: AAT2513 Evaluation Board
Bottom Side.
1. For reduced quiescent current, R2 and R4 = 221kΩ.
2513.2007.04.1.1
19
AAT2513
Dual 600mA Step-Down
Converter with Synchronization
Manufacturer
Part
Number
Sumida
Sumida
Sumida
Sumida
Taiyo Yuden
Taiyo Yuden
Taiyo Yuden
Taiyo Yuden
CDRH2D11
CDRH2D11
CDRH2D11
CDRH2D11
CBC2518T
CBC2518T
CBC2518T
CBC2016T
Inductance
(µH)
Max DC
Current (A)
DCR
(Ω)
Size (mm)
LxWxH
Type
1.5
2.2
3.3
4.7
1.0
2.2
4.7
2.2
1.48
1.27
1.02
0.88
1.2
1.1
0.92
0.83
0.068
0.098
0.123
0.170
0.08
0.13
0.2
0.2
3.2x3.2x1.2
3.2x3.2x1.2
3.2x3.2x1.2
3.2x3.2x1.2
2.5x1.8x1.8
2.5x1.8x1.8
2.5x1.8x1.8
2.0x1.6x1.6
Shielded
Shielded
Shielded
Shielded
Wire Wound Chip
Wire Wound Chip
Wire Wound Chip
Wire Wound Chip
Table 3: Typical Surface Mount Inductors.
Manufacturer
Part Number
Value
Voltage
Temp. Co.
Case
Murata
Murata
Murata
GRM219R61A475KE19
GRM21BR60J106KE19
GRM21BR60J226ME39
4.7µF
10µF
22µF
10V
6.3V
6.3V
X5R
X5R
X5R
0805
0805
0805
Table 4: Surface Mount Capacitors.
20
2513.2007.04.1.1
AAT2513
Dual 600mA Step-Down
Converter with Synchronization
Ordering Information
Voltage
Package
Channel 1
Channel 2
Marking1
Part Number (Tape and Reel)2
QFN33-16
0.6V
0.6V
UFXYY
AAT2513IVN-AA-T1
All AnalogicTech products are offered in Pb-free packaging. The term “Pb-free” means
semiconductor products that are in compliance with current RoHS standards, including
the requirement that lead not exceed 0.1% by weight in homogeneous materials. For more
information, please visit our website at http://www.analogictech.com/pbfree.
Legend
Voltage
Adjustable
(0.6V)
1.5
1.8
1.9
2.5
2.6
2.7
2.8
2.85
2.9
3.0
3.3
Code
A
G
I
Y
N
O
P
Q
R
S
T
W
1. XYY = assembly and date code.
2. Sample stock is generally held on part numbers listed in BOLD.
2513.2007.04.1.1
21
AAT2513
Dual 600mA Step-Down
Converter with Synchronization
Package Information1
QFN33-16
0.230 ± 0.05
Pin 1 Identification
1
0.400 ± 0.100
1.70 ± 0.05
3.000 ± 0.05
13
9
0.500 ± 0.05
Top View
0.025 ± 0.025
Bottom View
0.214 ± 0.036
0.900 ± 0.100
Pin 1 Dot By Marking
3.000 ± 0.05
5
C0.3
Side View
All dimensions in millimeters.
1. The leadless package family, which includes QFN, TQFN, DFN, TDFN and STDFN, has exposed copper (unplated) at the end of the
lead terminals due to the manufacturing process. A solder fillet at the exposed copper edge cannot be guaranteed and is not required
to ensure a proper bottom solder connection.
© Advanced Analogic Technologies, Inc.
AnalogicTech cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in an AnalogicTech product. No circuit patent licenses, copyrights, mask work
rights, or other intellectual property rights are implied. AnalogicTech reserves the right to make changes to their products or specifications or to discontinue any product or service without notice. Except as provided in AnalogicTech’s terms and conditions of sale, AnalogicTech assumes no liability whatsoever, and AnalogicTech disclaims any express or implied warranty relating to the sale and/or use of AnalogicTech products including liability or warranties relating to fitness for a particular purpose, merchantability, or infringement of any patent,
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Specific testing of all parameters of each device is not necessarily performed. AnalogicTech and the AnalogicTech logo are trademarks of Advanced Analogic Technologies Incorporated.
All other brand and product names appearing in this document are registered trademarks or trademarks of their respective holders.
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Phone (408) 737- 4600
Fax (408) 737- 4611
22
2513.2007.04.1.1