LINER LT1167_02

LT1167
Single Resistor Gain
Programmable, Precision
Instrumentation Amplifier
U
FEATURES
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DESCRIPTIO
The LT ®1167 is a low power, precision instrumentation
amplifier that requires only one external resistor to set gains
of 1 to 10,000. The low voltage noise of 7.5nV/√Hz (at 1kHz)
is not compromised by low power dissipation (0.9mA typical
for ±2.3V to ±15V supplies).
The part’s high accuracy (10ppm maximum nonlinearity,
0.08% max gain error (G = 10)) is not degraded even for load
resistors as low as 2k (previous monolithic instrumentation
amps used 10k for their nonlinearity specifications). The
LT1167 is laser trimmed for very low input offset voltage
(40µV max), drift (0.3µV/°C), high CMRR (90dB, G = 1) and
PSRR (105dB, G = 1). Low input bias currents of 350pA max
are achieved with the use of superbeta processing. The
output can handle capacitive loads up to 1000pF in any gain
configuration while the inputs are ESD protected up to 13kV
(human body). The LT1167 with two external 5k resistors
passes the IEC 1000-4-2 level 4 specification.
The LT1167, offered in 8-pin PDIP and SO packages, requires
significantly less PC board area than discrete multi op amp
and resistor designs. These advantages make the LT1167 the
most cost effective solution for precision instrumentation
amplifier applications.
Single Gain Set Resistor: G = 1 to 10,000
Gain Error: G = 10, 0.08% Max
Input Offset Voltage Drift: 0.3µV/°C Max
Meets IEC 1000-4-2 Level 4 ESD Tests with
Two External 5k Resistors
Gain Nonlinearity: G = 10, 10ppm Max
Input Offset Voltage: G = 10, 60µV Max
Input Bias Current: 350pA Max
PSRR at G = 1: 105dB Min
CMRR at G = 1: 90dB Min
Supply Current: 1.3mA Max
Wide Supply Range: ±2.3V to ±18V
1kHz Voltage Noise: 7.5nV/√Hz
0.1Hz to 10Hz Noise: 0.28µVP-P
Available in 8-Pin PDIP and SO Packages
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APPLICATIO S
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Bridge Amplifiers
Strain Gauge Amplifiers
Thermocouple Amplifiers
Differential to Single-Ended Converters
Medical Instrumentation
, LTC and LT are registered trademarks of Linear Technology Corporation.
U
TYPICAL APPLICATIO
Single Supply Barometer
VS
Gain Nonlinearity
3
2
LT1634CCZ-1.25
2
8
+
1/2
LT1490
1
LUCAS NOVA SENOR
NPC-1220-015-A-3L
–
4
1
–
4
5k
R6
1k
5
6
R8
100k
5k
2
+
–
7
R2
12Ω
5k
+
3
6
LT1167
G = 60
8
3
5
TO
4-DIGIT
DVM
+
4
5
1/2
LT1490
–
2
1
R1
825Ω
RSET
R3
50k
1
5k
6
OFFSET R4
ADJUST 50k
VS
NONLINEARITY (100ppm/DIV)
R5
392k
7
R7
50k
0.2% ACCURACY AT 25°C
1.2% ACCURACY AT 0°C TO 60°C
VS = 8V TO 30V
VOLTS
2.800
3.000
3.200
INCHES Hg
28.00
30.00
32.00
1167 TA02
1167 TA01
G = 1000
RL = 1k
VOUT = ±10V
OUTPUT VOLTAGE (2V/DIV)
1
LT1167
W W
W
AXI U
U
ABSOLUTE
RATI GS
U
U
W
PACKAGE/ORDER I FOR ATIO
(Note 1)
Supply Voltage ...................................................... ±20V
Differential Input Voltage (Within the
Supply Voltage) ..................................................... ±40V
Input Voltage (Equal to Supply Voltage) ................ ±20V
Input Current (Note 3) ........................................ ±20mA
Output Short-Circuit Duration .......................... Indefinite
Operating Temperature Range ................ – 40°C to 85°C
Specified Temperature Range
LT1167AC/LT1167C (Note 4) .................. 0°C to 70°C
LT1167AI/LT1167I ............................. – 40°C to 85°C
Storage Temperature Range ................. – 65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
ORDER PART
NUMBER
TOP VIEW
RG 1
8
RG
–IN 2
–
7
+VS
+IN 3
+
6
OUTPUT
5
REF
–VS 4
LT1167ACN8
LT1167ACS8
LT1167AIN8
LT1167AIS8
LT1167CN8
LT1167CS8
LT1167IN8
LT1167IS8
N8 PACKAGE
8-LEAD PDIP
S8 PACKAGE
8-LEAD PLASTIC SO
TJMAX = 150°C, θJA = 130°C/ W (N8)
TJMAX = 150°C, θJA = 190°C/ W (S8)
S8 PART MARKING
1167A
1167AI
1167
1167I
Consult factory for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
VS = ±15V, VCM = 0V, TA = 25°C, R L = 2k, unless otherwise noted.
SYMBOL PARAMETER
CONDITIONS (Note 7)
G
Gain Range
G = 1 + (49.4k/RG)
Gain Error
G=1
G = 10 (Note 2)
G = 100 (Note 2)
G = 1000 (Note 2)
Gain Nonlinearity (Note 5)
LT1167AC/LT1167AI
MIN
TYP
MAX
1
10k
LT1167C/LT1167I
MIN
TYP
MAX
1
UNITS
10k
0.008
0.010
0.025
0.040
0.02
0.08
0.08
0.10
0.015
0.020
0.030
0.040
0.03
0.10
0.10
0.10
%
%
%
%
VO = ±10V, G = 1
VO = ±10V, G = 10 and 100
VO = ±10V, G = 1000
1
2
15
6
10
40
1.5
3
20
10
15
60
ppm
ppm
ppm
VO = ±10V, G = 1, RL = 600
VO = ±10V, G = 10 and 100,
RL = 600
VO = ±10V, G = 1000,
RL = 600
5
6
12
15
6
7
15
20
ppm
ppm
20
65
25
80
ppm
VOST
Total Input Referred Offset Voltage
VOST = VOSI + VOSO/G
VOSI
Input Offset Voltage
G = 1000, VS = ±5V to ±15V
15
40
20
60
µV
VOSO
Output Offset Voltage
G = 1, VS = ±5V to ±15V
40
200
50
300
µV
IOS
Input Offset Current
90
320
100
450
pA
IB
Input Bias Current
50
350
80
500
pA
en
Input Noise Voltage (Note 8)
µVP-P
µVP-P
µVP-P
0.1Hz to 10Hz, G = 1
0.1Hz to 10Hz, G = 10
0.1Hz to 10Hz, G = 100 and 1000
2.00
0.50
0.28
2.00
0.50
0.28
fO = 1kHz
7.5
12
7.5
12
nV/√Hz
67
90
67
90
nV/√Hz
Total RTI Noise = √eni 2 + (eno /G)2 (Note 8)
eni
Input Noise Voltage Density (Note 8)
eno
Output Noise Voltage Density (Note 8) fO = 1kHz (Note 3)
2
LT1167
ELECTRICAL CHARACTERISTICS
VS = ±15V, VCM = 0V, TA = 25°C, RL = 2k, unless otherwise noted.
LT1167AC/LT1167AI
MIN
TYP
MAX
LT1167C/LT1167I
MIN
TYP
MAX
UNITS
fO = 0.1Hz to 10Hz
10
10
pAP-P
Input Noise Current Density
fO = 10Hz
124
124
fA/√Hz
RIN
Input Resistance
VIN = ±10V
1000
GΩ
CIN(DIFF)
Differential Input Capacitance
fO = 100kHz
1.6
1.6
pF
CIN(CM)
Common Mode Input
Capacitance
fO = 100kHz
1.6
1.6
pF
VCM
Input Voltage Range
G = 1, Other Input Grounded
VS = ±2.3V to ±5V
VS = ±5V to ±18V
SYMBOL PARAMETER
CONDITIONS (Note 7)
in
Input Noise Current
CMRR
1000
– VS + 1.9
– VS + 1.9
200
+ VS – 1.2 – VS + 1.9
+ VS – 1.4 – VS + 1.9
+ VS – 1.2
+ VS – 1.4
V
V
1k Source Imbalance,
VCM = 0V to ±10V
G=1
G = 10
G = 100
G = 1000
90
106
120
126
95
115
125
140
85
100
110
120
95
115
125
140
dB
dB
dB
dB
Power Supply
Rejection Ratio
VS = ±2.3 to ±18V
G=1
G = 10
G = 100
G = 1000
105
125
131
135
120
135
140
150
100
120
126
130
120
135
140
150
dB
dB
dB
dB
IS
Supply Current
VS = ±2.3V to ±18V
VOUT
Output Voltage Swing
RL = 10k
VS = ±2.3V to ±5V
VS = ±5V to ±18V
PSRR
Common Mode
Rejection Ratio
200
IOUT
Output Current
BW
Bandwidth
G=1
G = 10
G = 100
G = 1000
SR
Slew Rate
G = 1, VOUT = ±10V
Settling Time to 0.01%
10V Step
G = 1 to 100
G = 1000
RREFIN
Reference Input Resistance
IREFIN
Reference Input Current
VREF
Reference Voltage Range
AVREF
Reference Gain to Output
0.9
– VS + 1.1
– VS + 1.2
20
1.3
0.9
+ VS – 1.2 – VS + 1.1
+ VS – 1.3 – VS + 1.2
27
20
– VS + 1.6
V
V
27
mA
kHz
kHz
kHz
kHz
1.2
V/µs
14
130
14
130
µs
µs
20
20
kΩ
50
50
µA
1.2
VREF = 0V
+ VS – 1.2
+ VS – 1.3
mA
1000
800
120
12
1000
800
120
12
0.75
1.3
0.75
+ VS – 1.6 – VS + 1.6
+ VS – 1.6
1 ± 0.0001
1 ± 0.0001
V
3
LT1167
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VS = ±15V, VCM = 0V, 0°C ≤ TA ≤ 70°C, RL = 2k, unless otherwise noted.
SYMBOL PARAMETER
CONDITIONS (Note 7)
MIN
LT1167AC
TYP
MAX
MIN
LT1167C
TYP
MAX
UNITS
Gain Error
G=1
G = 10 (Note 2)
G = 100 (Note 2)
G = 1000 (Note 2)
●
●
●
●
0.01
0.08
0.09
0.14
0.03
0.30
0.30
0.33
0.012
0.100
0.120
0.140
0.04
0.33
0.33
0.35
Gain Nonlinearity
VOUT = ±10V, G = 1
VOUT = ±10V, G = 10 and 100
VOUT = ±10V, G = 1000
●
●
●
1.5
3
20
10
15
60
2
4
25
15
20
80
ppm
ppm
ppm
G/T
Gain vs Temperature
G < 1000 (Note 2)
●
20
50
20
50
ppm/°C
VOST
Total Input Referred
Offset Voltage
VOST = VOSI + VOSO/G
VOSI
Input Offset Voltage
VS = ±5V to ±15V
●
18
60
23
80
µV
VOSIH
Input Offset Voltage Hysteresis
(Notes 3, 6)
VOSO
Output Offset Voltage
VS = ±5V to ±15V
VOSOH
Output Offset Voltage Hysteresis
(Notes 3, 6)
VOSI/T
Input Offset Drift (Note 8)
(Note 3)
●
0.05
0.3
0.06
0.4
µV/°C
VOSO/T
Output Offset Drift
(Note 3)
●
0.7
3
0.8
4
µV/°C
IOS
Input Offset Current
●
100
400
120
550
IOS/T
Input Offset Current Drift
●
0.3
IB
Input Bias Current
●
75
IB/T
Input Bias Current Drift
●
0.4
VCM
Input Voltage Range
CMRR
PSRR
Common Mode
Rejection Ratio
Power Supply Rejection Ratio
3.0
60
●
380
70
500
0.4
450
105
µV
µV
30
pA
pA/°C
600
0.4
pA
pA/°C
G = 1, Other Input Grounded
VS = ±2.3V to ±5V
VS = ±5V to ±18V
●
●
– VS + 2.1
– VS + 2.1
1k Source Imbalance,
VCM = 0V to ±10V
G=1
G = 10
G = 100
G = 1000
●
●
●
●
88
100
115
117
92
110
120
135
83
97
113
114
92
110
120
135
dB
dB
dB
dB
VS = ±2.3V to ±18V
G=1
G = 10
G = 100
G = 1000
●
●
●
●
103
123
127
129
115
130
135
145
98
118
124
126
115
130
135
145
dB
dB
dB
dB
+ VS – 1.3
+ VS – 1.4
IS
Supply Current
VS = ±2.3V to ±18V
●
VOUT
Output Voltage Swing
RL = 10k
VS = ±2.3V to ±5V
VS = ±5V to ±18V
●
●
– VS + 1.4
– VS + 1.6
●
16
21
1.1
1.0
IOUT
Output Current
SR
Slew Rate
G = 1, VOUT = ±10V
●
0.65
VREF
REF Voltage Range
(Note 3)
●
– VS + 1.6
4
µV
3.0
30
%
%
%
%
– VS + 2.1
– VS + 2.1
1.5
+ VS – 1.3
+ VS – 1.5
+ VS – 1.6
+ VS – 1.3
+ VS – 1.4
1.0
– VS + 1.4
– VS + 1.6
+ VS –1.3
+ VS – 1.5
16
21
0.65
1.1
– VS + 1.6
1.5
V
V
mA
V
V
mA
V/µs
+ VS – 1.6
V
LT1167
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C.
VS = ±15V, VCM = 0V, – 40°C ≤ TA ≤ 85°C, RL = 2k, unless otherwise noted. (Note 4)
SYMBOL PARAMETER
CONDITIONS (Note 7)
MIN
LT1167AI
TYP
MAX
MIN
LT1167I
TYP
MAX
UNITS
Gain Error
G=1
G = 10 (Note 2)
G = 100 (Note 2)
G = 1000 (Note 2)
●
●
●
●
0.014
0.130
0.140
0.160
0.04
0.40
0.40
0.40
0.015
0.140
0.150
0.180
0.05
0.42
0.42
0.45
%
%
%
%
GN
Gain Nonlinearity (Notes 2, 4)
VO = ±10V, G = 1
VO = ±10V, G = 10 and 100
VO = ±10V, G = 1000
●
●
●
2
5
26
15
20
70
3
6
30
20
30
100
ppm
ppm
ppm
G/T
Gain vs Temperature
G < 1000 (Note 2)
●
20
50
20
50
ppm/°C
VOST
Total Input Referred Offset Voltage
VOST = VOSI + VOSO/G
VOSI
Input Offset Voltage
20
75
25
100
VOSIH
Input Offset Voltage Hysteresis
VOSO
Output Offset Voltage
VOSOH
Output Offset Voltage Hysteresis
(Notes 3, 6)
VOSI /T
Input Offset Drift (Note 8)
(Note 3)
●
VOSO/T
Output Offset Drift
(Note 3)
●
0.8
5
IOS
Input Offset Current
●
110
550
IOS/T
Input Offset Current Drift
●
0.3
IB
Input Bias Current
●
180
600
220
IB/T
Input Bias Current Drift
VCM
Input Voltage Range
VS = ±2.3V to ±5V
VS = ±5V to ±18V
●
●
– VS + 2.1
– VS + 2.1
CMRR
Common Mode Rejection Ratio
1k Source Imbalance,
VCM = 0V to ±10V
G=1
G = 10
G = 100
G = 1000
●
●
●
●
86
98
114
116
90
105
118
133
81
95
112
112
90
105
118
133
dB
dB
dB
dB
VS = ±2.3V to ±18V
G=1
G = 10
G = 100
G = 1000
●
●
●
●
100
120
125
128
112
125
132
140
95
115
120
125
112
125
132
140
dB
dB
dB
dB
PSRR
Power Supply Rejection Ratio
●
(Notes 3, 6)
3.0
180
●
0.05
Supply Current
VOUT
Output Voltage Swing
IOUT
Output Current
SR
Slew Rate
G = 1, VOUT = ±10V
VREF
REF Voltage Range
(Note 3)
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be imparied.
Note 2: Does not include the effect of the external gain resistor RG.
Note 3: This parameter is not 100% tested.
Note 4: The LT1167AC/LT1167C are designed, characterized and expected
to meet the industrial temperature limits, but are not tested at – 40°C and
85°C. I-grade parts are guaranteed.
Note 5: This parameter is measured in a high speed automatic tester that
does not measure the thermal effects with longer time constants. The
magnitude of these thermal effects are dependent on the package used,
heat sinking and air flow conditions.
●
●
– VS + 1.4
– VS + 1.6
●
15
20
●
0.55
0.95
●
– VS + 1.6
600
0.3
0.4
µV/°C
1
6
µV/°C
120
700
0.06
0.3
pA
pA/°C
+VS – 1.3
+ VS – 1.4
1.1
+ VS – 1.3 – VS + 1.4
+ VS – 1.5 – VS + 1.6
1.6
V
V
mA
+ VS – 1.3
+ VS – 1.5
15
20
0.55
0.95
+ VS – 1.6 – VS + 1.6
pA
pA/°C
800
0.6
1.6
µV
µV
30
+ VS – 1.3 – VS + 2.1
+ VS – 1.4 – VS + 2.1
1.1
●
VS = ±2.3V to ±5V
VS = ±5V to ±18V
200
0.5
●
IS
500
30
µV
µV
3.0
V
V
mA
V/µs
+ VS – 1.6
V
Note 6: Hysteresis in offset voltage is created by package stress that
differs depending on whether the IC was previously at a higher or lower
temperature. Offset voltage hysteresis is always measured at 25°C, but
the IC is cycled to 85°C I-grade (or 70°C C-grade) or – 40°C I-grade
(0°C C-grade) before successive measurement. 60% of the parts will
pass the typical limit on the data sheet.
Note 7: Typical parameters are defined as the 60% of the yield parameter
distribution.
Note 8: Referred to input.
5
LT1167
U W
TYPICAL PERFOR A CE CHARACTERISTICS
Gain Nonlinearity, G = 100
NONLINEARITY (1ppm/DIV)
1167 G01
Gain Nonlinearity, G = 1000
NONLINEARITY (100ppm/DIV)
NONLINEARITY (ppm)
70
1167 G03
G = 100 OUTPUT VOLTAGE (2V/DIV)
RL = 2k
VOUT = ±10V
Gain Error vs Temperature
Gain Nonlinearity vs Temperature
80
G = 1000 OUTPUT VOLTAGE (2V/DIV)
RL = 2k
VOUT = ±10V
1167 G02
OUTPUT VOLTAGE (2V/DIV)
G = 10
RL = 2k
VOUT = ±10V
0.20
VS = ± 15V
VOUT = – 10V TO 10V
RL = 2k
0.15
0.10
60
GAIN ERROR (%)
OUTPUT VOLTAGE (2V/DIV)
G=1
RL = 2k
VOUT = ±10V
NONLINEARITY (10ppm/DIV)
Gain Nonlinearity, G = 10
NONLINEARITY (10ppm/DIV)
Gain Nonlinearity, G = 1
50
40
30
G = 1000
0.05
G=1
0
– 0.05
– 0.10
20
1167 G04
G = 1, 10
10
– 0.15
G = 100
0
– 50 – 25
75
50
25
TEMPERATURE (°C)
0
100
150
VS = ±15V
G = 10*
VOUT = ±10V
RL = 2k
G = 100*
*DOES NOT INCLUDE
G = 1000*
TEMPERATURE EFFECTS
OF RG
– 0.20
– 50
– 25
0
25
50
TEMPERATURE (°C)
75
1167 G05
Distribution of Input
Offset Voltage, TA = – 40°C
30
25
20
15
10
VS = ±15V
G = 1000
25
60
20
15
10
1167 G40
0
– 60
40
137 N8 (2 LOTS)
165 S8 (3 LOTS)
302 TOTAL PARTS
5
5
6
30
137 N8 (2 LOTS)
165 S8 (3 LOTS)
302 TOTAL PARTS
0
0
– 80 – 60 – 40 – 20
20 40
INPUT OFFSET VOLTAGE (µV)
Distribution of Input
Offset Voltage, TA = 85°C
Distribution of Input
Offset Voltage, TA = 25°C
PERCENT OF UNITS (%)
PERCENT OF UNITS (%)
35
VS = ±15V
G = 1000
1167 G06
35
PERCENT OF UNITS (%)
40
100
VS = ±15V
G = 1000
137 N8 (2 LOTS)
165 S8 (3 LOTS)
302 TOTAL PARTS
30
25
20
15
10
5
– 40 – 20
0
20
40
INPUT OFFSET VOLTAGE (µV)
60
1167 G41
0
0
– 80 – 60 – 40 – 20
20 40
INPUT OFFSET VOLTAGE (µV)
60
1167 G42
LT1167
U W
TYPICAL PERFOR A CE CHARACTERISTICS
Distribution of Output
Offset Voltage, TA = – 40°C
30
VS = ±15V
G=1
137 N8 (2 LOTS)
165 S8 (3 LOTS)
25 302 TOTAL PARTS
PERCENT OF UNITS (%)
PERCENT OF UNITS (%)
137 N8 (2 LOTS)
35 165 S8 (3 LOTS)
302 TOTAL PARTS
30
25
20
15
10
Distribution of Output
Offset Voltage, TA = 85°C
40
VS = ±15V
G=1
20
15
10
5
0
Distribution of Input Offset
Voltage Drift
15
10
VS = ±15V
TA = – 40°C TO 85°C
G=1
35
5
137 N8 (2 LOTS)
165 S8 (3 LOTS)
302 TOTAL PARTS
30
25
20
15
10
0
0
0.1 0.2 0.3
– 0.4 – 0.3 – 0.2 – 0.1 0
INPUT OFFSET VOLTAGE DRIFT (µV/°C)
8
N8
6
4
2
0
0
500
270 S8
122 N8
392 TOTAL PARTS
30
20
100
1167 G10
0
– 100
5
Input Bias and Offset Current
vs Temperature
10
– 60
20
60
– 20
INPUT BIAS CURRENT (pA)
1
2
3
4
TIME AFTER POWER ON (MINUTES)
1167 G09
– 60
20
60
– 20
INPUT OFFSET CURRENT (pA)
100
1167 G11
INPUT BIAS AND OFFSET CURRENT (pA)
10
VS = ±15V
TA = 25°C
40
PERCENT OF UNITS (%)
PERCENT OF UNITS (%)
50
270 S8
122 N8
392 TOTAL PARTS
20
S8
10
Input Offset Current
30
VS = ± 15V
TA = 25°C
G=1
12
1167 G47
Input Bias Current
0
– 100
14
–5 –4 –3 –2 –1 0 1 2 3 4 5
OUTPUT OFFSET VOLTAGE DRIFT (µV/°C)
1167 G46
40
1167 G45
Warm-Up Drift
5
VS = ±15V
TA = 25°C
10
–400 –300 –200 –100 0 100 200 300 400
OUTPUT OFFSET VOLTAGE (µV)
CHANGE IN OFFSET VOLTAGE (µV)
40
137 N8 (2 LOTS)
165 S8 (3 LOTS)
302 TOTAL PARTS
20
50
15
Distribution of Output Offset
Voltage Drift
PERCENT OF UNITS (%)
PERCENT OF UNITS (%)
25
20
1167 G44
1167 G43
VS = ±15V
TA = – 40°C TO 85°C
G = 1000
25
0
– 200 –150 –100 –50 0
50 100 150 200
OUTPUT OFFSET VOLTAGE (µV)
–400 –300 –200 –100 0 100 200 300 400
OUTPUT OFFSET VOLTAGE (µV)
30
30
5
5
0
VS = ±15V
G=1
137 N8 (2 LOTS)
35 165 S8 (3 LOTS)
302 TOTAL PARTS
PERCENT OF UNITS (%)
40
Distribution of Output
Offset Voltage, TA = 25°C
400
VS = ±15V
VCM = 0V
300
200
100
IOS
0
IB
– 100
– 200
– 300
– 400
– 500
–75 – 50 –25 0
25 50 75
TEMPERATURE (°C)
100 125
1167 G12
7
LT1167
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TYPICAL PERFOR A CE CHARACTERISTICS
COMMON MODE REJECTION RATIO (dB)
160
INPUT BIAS CURRENT (pA)
400
300
200
100
70°C
85°C
–100
– 200
0°C
– 300
– 40°C
25°C
– 400
140
G = 100
120
G = 10
100
G=1
80
60
40
20
0
0.1
– 500
–15 –12 – 9 – 6 – 3 0 3 6 9 12 15
COMMON MODE INPUT VOLTAGE (V)
VS = ±15V
TA = 25°C
1k SOURCE
IMBALANCE
G = 1000
1
10
1k
100
FREQUENCY (Hz)
10k
V – = – 15V
TA = 25°C
80
60
–10
10k
100k
G = 10
10
20
1k
100
FREQUENCY (Hz)
G = 100
20
0
10
G=1
– 20
0.01
0.1
1
10
FREQUENCY (kHz)
100
1000
60
40
20
0
0.1
GAIN = 10
1/fCORNER = 7Hz
GAIN = 100, 1000
1k
100
FREQUENCY (Hz)
10k
100k
85°C
25°C
1.00
– 40°C
0.75
0.50
10
15
5
SUPPLY VOLTAGE (± V)
0
20
1167 G18
VS = ±15V
TA = 25°C
NOISE VOLTAGE (0.2µV/DIV)
1/fCORNER = 9Hz
10
1.25
VS = ±15V
TA = 25°C
GAIN = 1
1
0.1Hz to 10Hz Noise Voltage,
Referred to Input, G = 1000
NOISE VOLTAGE (2µV/DIV)
VOLTAGE NOISE DENSITY (nV√Hz)
VS = ±15V
TA = 25°C
10
80
0.1Hz to 10Hz Noise Voltage,
G=1
1/fCORNER = 10Hz
G = 1000
1167 G17
Voltage Noise Density
vs Frequency
100
G=1
VS = ± 15V
TA = 25°C
1167 G16
1000
100
Supply Current vs Supply Voltage
30
40
1
G = 10
1.50
SUPPLY CURRENT (mA)
G=1
0
0.1
120
V + = 15V
TA = 25°C
1167 G15
G = 1000
40
G = 100
100
G = 100
50
G = 1000
G = 10
120
140
Gain vs Frequency
60
GAIN (dB)
POSITIVE POWER SUPPLY REJECTION RATIO (dB)
Positive Power Supply Rejection
Ratio vs Frequency
140
160
1167 G14
1167 G13
160
100k
NEGATIVE POWER SUPPLY REJECTION RATIO (dB)
500
0
Negative Power Supply Rejection
Ratio vs Frequency
Common Mode Rejection Ratio
vs Frequency
Input Bias Current
vs Common Mode Input Voltage
BW LIMIT
GAIN = 1000
0
1
10
100
1k
FREQUENCY (Hz)
10k
100k
1167 G19
8
0
1
2
3
4 5 6
TIME (SEC)
7
8
9
10
1167 G20
0
1
2
3
4 5 6
TIME (SEC)
7
8
9
10
1167 G21
LT1167
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TYPICAL PERFOR A CE CHARACTERISTICS
Current Noise Density
vs Frequency
0.1Hz to 10Hz Current Noise
Short-Circuit Current vs Time
50
VS = ±15V
TA = 25°C
VS = ±15V
TA = 25°C
OUTPUT CURRENT (mA)
(SINK)
(SOURCE)
100
RS
VS = ±15V
40
CURRENT NOISE (5pA/DIV)
CURRENT NOISE DENSITY (fA/√Hz)
1000
TA = – 40°C
30
TA = 25°C
20
TA = 85°C
10
0
– 10
TA = 85°C
– 20
– 30
TA = – 40°C
– 40
TA = 25°C
– 50
10
10
100
FREQUENCY (Hz)
1
1000
1
0
2
3
4 5 6
TIME (SEC)
7
1167 G22
8
9
10
2
1
0
3
TIME FROM OUTPUT SHORT TO GROUND (MINUTES)
1167 G24
1167 G23
Overshoot vs Capacitive Load
Large-Signal Transient Response
Small-Signal Transient Response
100
70
5V/DIV
OVERSHOOT (%)
80
20mV/DIV
VS = ±15V
VOUT = ± 50mV
RL = ∞
90
60
50
AV = 1
40
30
AV = 10
20
10
G=1
VS = ±15V
RL = 2k
CL = 60pF
AV ≥ 100
0
10
100
1000
CAPACITIVE LOAD (pF)
10000
10µs/DIV
1167 G28
G=1
VS = ±15V
RL = 2k
CL = 60pF
10µs/DIV
1167 G29
1167 G25
Output Impedance vs Frequency
Large-Signal Transient Response
Small-Signal Transient Response
VS = ± 15V
TA = 25°C
G = 1 TO 1000
20mV/DIV
100
5V/DIV
OUTPUT IMPEDANCE (Ω)
1000
10
1
0.1
1
10
100
FREQUENCY (kHz)
1000
G = 10
VS = ±15V
RL = 2k
CL = 60pF
10µs/DIV
1167 G31
G = 10
VS = ±15V
RL = 2k
CL = 60pF
10µs/DIV
1167 G32
1167 G26
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LT1167
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TYPICAL PERFOR A CE CHARACTERISTICS
Undistorted Output Swing
vs Frequency
Small-Signal Transient Response
Large-Signal Transient Response
VS = ± 15V
TA = 25°C
5V/DIV
30 G = 10, 100, 1000
G=1
25
20mV/DIV
PEAK-TO-PEAK OUTPUT SWING (V)
35
20
15
10
5
0
10
100
FREQUENCY (kHz)
1
1000
1167 G34
10µs/DIV
G = 100
VS = ±15V
RL = 2k
CL = 60pF
G = 100
VS = ±15V
RL = 2k
CL = 60pF
10µs/DIV
1167 G35
1167 G27
Settling Time vs Gain
Large-Signal Transient Response
Small-Signal Transient Response
100
20mV/DIV
VS = ± 15V
TA = 25°C
∆VOUT = 10V
1mV = 0.01%
5V/DIV
SETTLING TIME (µs)
1000
10
1
1
10
100
1000
1167 G37
50µs/DIV
G = 1000
VS = ±15V
RL = 2k
CL = 60pF
G = 1000
VS = ±15V
RL = 2k
CL = 60pF
50µs/DIV
1167 G38
GAIN (dB)
1167 G30
VS = ±15
G=1
TA = 25°C
CL = 30pF
RL = 1k
8
OUTPUT STEP (V)
6
4
2
1.8
TO 0.1%
1.6
0V
VOUT
0
0V
–2
VOUT
–4
TO 0.01%
–6
–8
–10
+ VS
VS = ± 15V
VOUT = ±10V
G=1
TO 0.01%
SLEW RATE (V/µs)
10
Output Voltage Swing
vs Load Current
Slew Rate vs Temperature
OUTPUT VOLTAGE SWING (V)
(REFERRED TO SUPPLY VOLTAGE)
Settling Time vs Step Size
1.4
+ SLEW
1.2
– SLEW
1.0
TO 0.1%
2
3
4
5 6 7 8 9 10 11 12
SETTLING TIME (µs)
1167 G33
10
0.8
– 50 –25
50
25
0
75
TEMPERATURE (°C)
100
125
1167 G36
VS = ± 15V
85°C
25°C
– 40°C
+ VS – 0.5
+ VS – 1.0
+ VS – 1.5
SOURCE
+ VS – 2.0
– VS + 2.0
– VS + 1.5
SINK
– VS + 1.0
– VS + 0.5
– VS
0.01
0.1
1
10
OUTPUT CURRENT (mA)
100
1167 G39
LT1167
W
BLOCK DIAGRA
V+
VB
R5
10k
+
R6
10k
6 OUTPUT
A1
–
R3
400Ω
–IN
C1
2
Q1
R1
24.7k
–
V–
+
A3
RG 1
RG 8
V–
VB
V+
+
R7
10k
R8
10k
5 REF
A2
–
R4
400Ω
+IN
3
C2
V–
Q2
R2
24.7k
7
V+
4
V–
V–
PREAMP STAGE
DIFFERENCE AMPLIFIER STAGE
1167 F01
Figure 1. Block Diagram
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THEORY OF OPERATIO
The LT1167 is a modified version of the three op amp
instrumentation amplifier. Laser trimming and monolithic construction allow tight matching and tracking of
circuit parameters over the specified temperature range.
Refer to the block diagram (Figure 1) to understand the
following circuit description. The collector currents in Q1
and Q2 are trimmed to minimize offset voltage drift, thus
assuring a high level of performance. R1 and R2 are
trimmed to an absolute value of 24.7k to assure that the
gain can be set accurately (0.05% at G = 100) with only
one external resistor RG. The value of RG determines the
transconductance of the preamp stage. As RG is reduced
for larger programmed gains, the transconductance of
the input preamp stage increases to that of the input
transistors Q1 and Q2. This increases the open-loop gain
when the programmed gain is increased, reducing the
input referred gain related errors and noise. The input
voltage noise at gains greater than 50 is determined only
by Q1 and Q2. At lower gains the noise of the difference
amplifier and preamp gain setting resistors increase the
noise. The gain bandwidth product is determined by C1,
C2 and the preamp transconductance which increases
with programmed gain. Therefore, the bandwidth does
not drop proportionally to gain.
The input transistors Q1 and Q2 offer excellent matching,
which is inherent in NPN bipolar transistors, as well as
picoampere input bias current due to superbeta processing. The collector currents in Q1 and Q2 are held constant
due to the feedback through the Q1-A1-R1 loop and
Q2-A2-R2 loop which in turn impresses the differential
input voltage across the external gain set resistor RG.
Since the current that flows through RG also flows through
R1 and R2, the ratios provide a gained-up differential voltage,G = (R1 + R2)/RG, to the unity-gain difference amplifier
A3. The common mode voltage is removed by A3, resulting in a single-ended output voltage referenced to the
voltage on the REF pin. The resulting gain equation is:
VOUT – VREF = G(VIN+ – VIN–)
where:
G = (49.4kΩ / RG) + 1
solving for the gain set resistor gives:
RG = 49.4kΩ /(G – 1)
11
LT1167
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THEORY OF OPERATIO
Input and Output Offset Voltage
Output Offset Trimming
The offset voltage of the LT1167 has two components: the
output offset and the input offset. The total offset voltage
referred to the input (RTI) is found by dividing the output
offset by the programmed gain (G) and adding it to the
input offset. At high gains the input offset voltage dominates, whereas at low gains the output offset voltage
dominates. The total offset voltage is:
The LT1167 is laser trimmed for low offset voltage so that
no external offset trimming is required for most applications. In the event that the offset needs to be adjusted, the
circuit in Figure 2 is an example of an optional offset adjust
circuit. The op amp buffer provides a low impedance to the
REF pin where resistance must be kept to minimum for
best CMRR and lowest gain error.
2
–IN
–
1
Total output offset voltage (RTO)
= (input offset • G) + output offset
RG
3
+
V+
5
–
The reference terminal is one end of one of the four 10k
resistors around the difference amplifier. The output voltage of the LT1167 (Pin 6) is referenced to the voltage on
the reference terminal (Pin 5). Resistance in series with
the REF pin must be minimized for best common mode
rejection. For example, a 2Ω resistance from the REF pin
to ground will not only increase the gain error by 0.02%
but will lower the CMRR to 80dB.
+IN
OUTPUT
REF
8
Reference Terminal
6
LT1167
1
±10mV
ADJUSTMENT RANGE
1/2
LT1112
+
Total input offset voltage (RTI)
= input offset + (output offset/G)
2
10mV
100Ω
3
10k
100Ω
–10mV
V–
1167 F02
Figure 2. Optional Trimming of Output Offset Voltage
Single Supply Operation
Input Bias Current Return Path
For single supply operation, the REF pin can be at the same
potential as the negative supply (Pin 4) provided the
output of the instrumentation amplifier remains inside the
specified operating range and that one of the inputs is at
least 2.5V above ground. The barometer application on the
front page of this data sheet is an example that satisfies
these conditions. The resistance Rb from the bridge transducer to ground sets the operating current for the bridge
and also has the effect of raising the input common mode
voltage. The output of the LT1167 is always inside the
specified range since the barometric pressure rarely goes
low enough to cause the output to rail (30.00 inches of Hg
corresponds to 3.000V). For applications that require the
output to swing at or below the REF potential, the voltage
on the REF pin can be level shifted. An op amp is used to
buffer the voltage on the REF pin since a parasitic series
resistance will degrade the CMRR. The application in the
back of this data sheet, Four Digit Pressure Sensor, is an
example.
The low input bias current of the LT1167 (350pA) and the
high input impedance (200GΩ) allow the use of high
impedance sources without introducing additional offset
voltage errors, even when the full common mode range is
required. However, a path must be provided for the input
bias currents of both inputs when a purely differential
signal is being amplified. Without this path the inputs will
float to either rail and exceed the input common mode
range of the LT1167, resulting in a saturated input stage.
Figure 3 shows three examples of an input bias current
path. The first example is of a purely differential signal
source with a 10kΩ input current path to ground. Since the
impedance of the signal source is low, only one resistor is
needed. Two matching resistors are needed for higher
impedance signal sources as shown in the second
example. Balancing the input impedance improves both
common mode rejection and DC offset. The need for input
resistors is eliminated if a center tap is present as shown
in the third example.
12
LT1167
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THEORY OF OPERATIO
–
THERMOCOUPLE
RG
–
MICROPHONE,
HYDROPHONE,
ETC
LT1167
RG
+
–
LT1167
+
200k
10k
RG
LT1167
+
200k
CENTER-TAP PROVIDES
BIAS CURRENT RETURN
1167 F03
Figure 3. Providing an Input Common Mode Current Path
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APPLICATIO S I FOR ATIO
The LT1167 is a low power precision instrumentation
amplifier that requires only one external resistor to accurately set the gain anywhere from 1 to 1000. The output
can handle capacitive loads up to 1000pF in any gain
configuration and the inputs are protected against ESD
strikes up to 13kV (human body).
Input Protection
The LT1167 can safely handle up to ±20mA of input
current in an overload condition. Adding an external 5k
input resistor in series with each input allows DC input
fault voltages up to ±100V and improves the ESD immunity to 8kV (contact) and 15kV (air discharge), which is the
IEC 1000-4-2 level 4 specification. If lower value input
resistors are needed, a clamp diode from the positive
supply to each input will maintain the IEC 1000-4-2
specification to level 4 for both air and contact discharge.
VCC
VCC
J1
2N4393
J2
2N4393
RIN
OPTIONAL FOR HIGHEST
ESD PROTECTION
+
RG
RIN
VCC
LT1167
OUT
REF
–
VEE
Figure 4. Input Protection
1167 F04
A 2N4393 drain/source to gate is a good low leakage diode
for use with 1k resistors, see Figure 4. The input resistors
should be carbon and not metal film or carbon film.
RFI Reduction
In many industrial and data acquisition applications,
instrumentation amplifiers are used to accurately amplify
small signals in the presence of large common mode
voltages or high levels of noise. Typically, the sources of
these very small signals (on the order of microvolts or
millivolts) are sensors that can be a significant distance
from the signal conditioning circuit. Although these sensors may be connected to signal conditioning circuitry,
using shielded or unshielded twisted-pair cabling, the cabling may act as antennae, conveying very high frequency
interference directly into the input stage of the LT1167.
The amplitude and frequency of the interference can have
an adverse effect on an instrumentation amplifier’s input
stage by causing an unwanted DC shift in the amplifier’s
input offset voltage. This well known effect is called RFI
rectification and is produced when out-of-band interference is coupled (inductively, capacitively or via radiation)
and rectified by the instrumentation amplifier’s input transistors. These transistors act as high frequency signal
detectors, in the same way diodes were used as RF
envelope detectors in early radio designs. Regardless of
the type of interference or the method by which it is
coupled into the circuit, an out-of-band error signal appears in series with the instrumentation amplifier’s inputs.
13
LT1167
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APPLICATIO S I FOR ATIO
To significantly reduce the effect of these out-of-band
signals on the input offset voltage of instrumentation
amplifiers, simple lowpass filters can be used at the
inputs. These filters should be located very close to the
input pins of the circuit. An effective filter configuration is
illustrated in Figure 5, where three capacitors have been
added to the inputs of the LT1167. Capacitors CXCM1 and
CXCM2 form lowpass filters with the external series resistors RS1, 2 to any out-of-band signal appearing on each of
the input traces. Capacitor CXD forms a filter to reduce any
unwanted signal that would appear across the input traces.
An added benefit to using CXD is that the circuit’s AC
common mode rejection is not degraded due to common
mode capacitive imbalance. The differential mode and
common mode time constants associated with the capacitors are:
tDM(LPF) = (2)(RS)(CXD)
tCM(LPF) = (RS1, 2)(CXCM1, 2)
Setting the time constants requires a knowledge of the
frequency, or frequencies of the interference. Once this
frequency is known, the common mode time constants
can be set followed by the differential mode time constant.
To avoid any possibility of inadvertently affecting the
signal to be processed, set the common mode time
constant an order of magnitude (or more) larger than the
differential mode time constant. Set the common mode
IN +
CXD
0.1µF
IN –
V+
RS1 CXCM1
1.6k 0.001µF
RS2
1.6k
+
RG
LT1167
VOUT
–
CXCM2
0.001µF
V–
f– 3dB ≈ 500Hz
1167 F05
EXTERNAL RFI
FILTER
Figure 5. Adding a Simple RC Filter at the Inputs to an
Instrumentation Amplifier is Effective in Reducing Rectification
of High Frequency Out-of-Band Signals
14
time constants such that they do not degrade the LT1167’s
inherent AC CMR. Then the differential mode time constant can be set for the bandwidth required for the application. Setting the differential mode time constant close to
the sensor’s BW also minimizes any noise pickup along
the leads. To avoid any possibility of common mode to
differential mode signal conversion, match the common
mode time constants to 1% or better. If the sensor is an
RTD or a resistive strain gauge, then the series resistors
RS1, 2 can be omitted, if the sensor is in proximity to the
instrumentation amplifier.
“Roll Your Own”—Discrete vs Monolithic LT1167
Error Budget Analysis
The LT1167 offers performance superior to that of “roll
your own” three op amp discrete designs. A typical application that amplifies and buffers a bridge transducer’s
differential output is shown in Figure 6. The amplifier, with
its gain set to 100, amplifies a differential, full-scale output
voltage of 20mV over the industrial temperature range. To
make the comparison challenging, the low cost version of
the LT1167 will be compared to a discrete instrumentation
amp made with the A grade of one of the best precision
quad op amps, the LT1114A. The LT1167C outperforms
the discrete amplifier that has lower VOS, lower IB and
comparable VOS drift. The error budget comparison in
Table 1 shows how various errors are calculated and how
each error affects the total error budget. The table shows
the greatest differences between the discrete solution and
the LT1167 are input offset voltage and CMRR. Note that
for the discrete solution, the noise voltage specification is
multiplied by √2 which is the RMS sum of the uncorelated
noise of the two input amplifiers. Each of the amplifier
errors is referenced to a full-scale bridge differential
voltage of 20mV. The common mode range of the bridge
is 5V. The LT1114 data sheet provides offset voltage,
offset voltage drift and offset current specifications for the
matched op amp pairs used in the error-budget table. Even
with an excellent matched op amp like the LT1114, the
discrete solution’s total error is significantly higher than
the LT1167’s total error. The LT1167 has additional advantages over the discrete design, including lower component cost and smaller size.
LT1167
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APPLICATIO S I FOR ATIO
+
+
10V
350Ω
–
10k**
–
10k**
+
350Ω
RG
499Ω
350Ω
10k*
10k*
1/4
LT1114A
LT1167C
202Ω**
REF
350Ω
–
PRECISION BRIDGE TRANSDUCER
1/4
LT1114A
–
1/4
LT1114A
10k*
10k*
+
LT1167 MONOLITHIC
INSTRUMENTATION AMPLIFIER
G = 100, RG = ±10ppm TC
SUPPLY CURRENT = 1.3mA MAX
“ROLL YOUR OWN” INST AMP, G = 100
* 0.02% RESISTOR MATCH, 3ppm/°C TRACKING
** DISCRETE 1% RESISTOR, ±100ppm/°C TC
100ppm TRACKING
SUPPLY CURRENT = 1.35mA FOR 3 AMPLIFIERS
1167 F06
Figure 6. “Roll Your Own” vs LT1167
Table 1. “Roll Your Own” vs LT1167 Error Budget
ERROR, ppm OF FULL SCALE
ERROR SOURCE
LT1167C CIRCUIT CALCULATION
“ROLL YOUR OWN”’ CIRCUIT
CALCULATION
Absolute Accuracy at TA = 25°C
Input Offset Voltage, µV
Output Offset Voltage, µV
Input Offset Current, nA
CMR, dB
60µV/20mV
(300µV/100)/20mV
[(450pA)(350/2)Ω]/20mV
110dB→[(3.16ppm)(5V)]/20mV
100µV/20mV
[(60µV)(2)/100]/20mV
[(450pA)(350Ω)/2]/20mV
[(0.02% Match)(5V)]/20mV
3000
150
4
790
5000
60
4
500
Total Absolute Error
3944
5564
(100ppm/°C Track)(60°C)
[(1.6µV/°C)(60°C)]/20mV
[(1.1µV/°C)(2)(60°C)]/100/20mV
3600
1200
180
6000
4800
66
Total Drift Error
4980
10866
10ppm
(0.3µVP-P)(√ 2)/20mV
15
14
10
21
Total Resolution Error
Grand Total Error
29
8953
31
16461
Drift to 85°C
Gain Drift, ppm/°C
Input Offset Voltage Drift, µV/°C
Output Offset Voltage Drift, µV/°C
Resolution
Gain Nonlinearity, ppm of Full Scale
Typ 0.1Hz to 10Hz Voltage Noise, µVP-P
(50ppm + 10ppm)(60°C)
[(0.4µV/°C)(60°C)]/20mV
[6µV/°C)(60°C)]/100/20mV
15ppm
0.28µVP-P/20mV
LT1167C “ROLL YOUR OWN”
G = 100, VS = ±15V
All errors are min/max and referred to input.
Current Source
Figure 7 shows a simple, accurate, low power programmable current source. The differential voltage across Pins
2 and 3 is mirrored across RG. The voltage across RG is
amplified and applied across RX, defining the output
current. The 50µA bias current flowing from Pin 5 is
buffered by the LT1464 JFET operational amplifier. This
has the effect of improving the resolution of the current
source to 3pA, which is the maximum IB of the LT1464A.
Replacing RG with a programmable resistor greatly
increases the range of available output currents.
15
LT1167
U
W
U U
APPLICATIO S I FOR ATIO
8
VS
+
RG
LT1167
2
REF
VX
5
–
4
–V S
1
1/2
LT1464
+
V
[(+IN) – (–IN)]G
IL = X =
RX
RX
G=
RX
6
1
–IN
important, R6 and C2 make up a 0.3Hz highpass filter.
The AC signal at LT1112’s Pin 5 is amplified by a gain of
101 set by (R7/R8) +1. The parallel combination of C3
and R7 form a lowpass filter that decreases this gain at
frequencies above 1kHz. The ability to operate at ±3V on
0.9mA of supply current makes the LT1167 ideal for
battery-powered applications. Total supply current for
this application is 1.7mA. Proper safeguards, such as
isolation, must be added to this circuit to protect the
patient from possible harm.
7
–
3
+IN
IL
2
3
LOAD
49.4kΩ
+1
RG
Low IB Favors High Impedance Bridges,
Lowers Dissipation
1167 F07
Figure 7. Precision Voltage-to-Current Converter
Nerve Impulse Amplifier
The LT1167’s low current noise makes it ideal for high
source impedance EMG monitors. Demonstrating the
LT1167’s ability to amplify low level signals, the circuit in
Figure 8 takes advantage of the amplifier’s high gain and
low noise operation. This circuit amplifies the low level
nerve impulse signals received from a patient at Pins 2
and 3. RG and the parallel combination of R3 and R4 set
a gain of ten. The potential on LT1112’s Pin 1 creates a
ground for the common mode signal. C1 was chosen to
maintain the stability of the patient ground. The LT1167’s
high CMRR ensures that the desired differential signal is
amplified and unwanted common mode signals are attenuated. Since the DC portion of the signal is not
3V
PATIENT/CIRCUIT
PROTECTION/ISOLATION
3
8
+IN
C1
0.01µF
R3
30k
R1
12k
R2
1M
R4
30k
–
PATIENT
GROUND
1
1/2
LT1112
+
–IN
+
0.3Hz
HIGHPASS
7
RG
6k
6
LT1167
G = 10
1
5
–
5
R6
1M
4
6
+
8
1/2
LT1112
–
4
–3V
–3V
3
AV = 101
POLE AT 1kHz
Figure 8. Nerve Impulse Amplifier
16
3V
C2
0.47µF
2
2
The LT1167’s low supply current, low supply voltage
operation and low input bias currents optimize it for
battery-powered applications. Low overall power dissipation necessitates using higher impedance bridges. The
single supply pressure monitor application (Figure 9)
shows the LT1167 connected to the differential output of
a 3.5k bridge. The bridge’s impedance is almost an order
of magnitude higher than that of the bridge used in the
error-budget table. The picoampere input bias currents
keep the error caused by offset current to a negligible
level. The LT1112 level shifts the LT1167’s reference pin
and the ADC’s analog ground pins above ground. The
LT1167’s and LT1112’s combined power dissipation is
still less than the bridge’s. This circuit’s total supply
current is just 2.8mA.
7
OUTPUT
1V/mV
R7
10k
R8
100Ω
C3
15nF
1167 F08
LT1167
U
W
U U
APPLICATIO S I FOR ATIO
BI TECHNOLOGIES
67-8-3 R40KQ
(0.02% RATIO MATCH)
5V
1
3
8
3.5k
+
40k
7
3.5k
REF
G = 200
249Ω
3.5k
6
LT1167
3.5k
1
2
IN
5
–
ADC
LTC®1286
20k
3
4
+
2
1
1/2
LT1112
40k
DIGITAL
DATA
OUTPUT
AGND
–
1167 F09
Figure 9. Single Supply Bridge Amplifier
U
TYPICAL APPLICATIO
AC Coupled Instrumentation Amplifier
1
–
RG
8
REF
3
5
+
OUTPUT
R1
500k
C1
0.3µF
–
+IN
6
LT1167
1
1/2
LT1112
+
2
–IN
2
3
f –3dB =
1
(2π)(R1)(C1)
= 1.06Hz
1167 TA04
17
LT1167
U
PACKAGE DESCRIPTION
Dimensions in inches (millimeters) unless otherwise noted.
N8 Package
8-Lead PDIP (Narrow 0.300)
(LTC DWG # 05-08-1510)
0.400*
(10.160)
MAX
8
7
6
5
1
2
3
4
0.255 ± 0.015*
(6.477 ± 0.381)
0.300 – 0.325
(7.620 – 8.255)
0.009 – 0.015
(0.229 – 0.381)
(
+0.035
0.325 –0.015
8.255
+0.889
–0.381
)
0.045 – 0.065
(1.143 – 1.651)
0.065
(1.651)
TYP
0.100
(2.54)
BSC
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm)
18
0.130 ± 0.005
(3.302 ± 0.127)
0.125
(3.175) 0.020
MIN
(0.508)
MIN
0.018 ± 0.003
(0.457 ± 0.076)
N8 1098
LT1167
U
PACKAGE DESCRIPTION
Dimensions in inches (millimeters) unless otherwise noted.
S8 Package
8-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
0.189 – 0.197*
(4.801 – 5.004)
8
7
6
5
0.150 – 0.157**
(3.810 – 3.988)
0.228 – 0.244
(5.791 – 6.197)
1
0.010 – 0.020
× 45°
(0.254 – 0.508)
0.008 – 0.010
(0.203 – 0.254)
0.053 – 0.069
(1.346 – 1.752)
0°– 8° TYP
0.016 – 0.050
(0.406 – 1.270)
0.014 – 0.019
(0.355 – 0.483)
TYP
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
2
3
4
0.004 – 0.010
(0.101 – 0.254)
0.050
(1.270)
BSC
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
SO8 1298
19
LT1167
U
TYPICAL APPLICATIO
4-Digit Pressure Sensor
9V
R8
392k
3
+
2
2
1
1/4
LT1114
1
LT1634CCZ-1.25
LUCAS NOVA SENOR
NPC-1220-015A-3L
4
–
11
–
4
5k
R9
1k
2
9V
R1
825Ω
+
RSET
7
6
LT1167
G = 60
R2
12Ω
5k
6
–
1
5k
5k
2
1
8
5
3
3
+
4
10
+
12
0.2% ACCURACY AT ROOM TEMP
1.2% ACCURACY AT 0°C TO 60°C
VOLTS
2.800
3.000
3.200
INCHES Hg
28.00
30.00
32.00
13
+
–
14
1/4
LT1114
8
1/4
LT1114
5
9
TO
4-DIGIT
DVM
CALIBRATION
ADJUST
–
R4
100k
R5
100k
R3
51k
R6
50k
R7
180k
C1
1µF
1167 TA03
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC1100
Precision Chopper-Stabilized Instrumentation Amplifier
Best DC Accuracy
LT1101
Precision, Micropower, Single Supply Instrumentation Amplifier
Fixed Gain of 10 or 100, IS < 105µA
LT1102
High Speed, JFET Instrumentation Amplifier
Fixed Gain of 10 or 100, 30V/µs Slew Rate
LT1168
Low Power, Single Resistor Programmable Instrumentation Amplifier
ISUPPLY = 530µA Max
LTC 1418
14-Bit, Low Power, 200ksps ADC with Serial and Parallel I/O
Single Supply 5V or ±5V Operation, ±1.5LSB INL
and ±1LSB DNL Max
LT1460
Precision Series Reference
Micropower; 2.5V, 5V, 10V Versions; High Precision
LT1468
16-Bit Accurate Op Amp, Low Noise Fast Settling
16-Bit Accuracy at Low and High Frequencies, 90MHz GBW,
22V/µs, 900ns Settling
LTC1562
Active RC Filter
Lowpass, Bandpass, Highpass Responses; Low Noise,
Low Distortion, Four 2nd Order Filter Sections
LTC1605
16-Bit, 100ksps, Sampling ADC
Single 5V Supply, Bipolar Input Range: ±10V,
Power Dissipation: 55mW Typ
®
20
Linear Technology Corporation
1167fa LT/TP 0301 2K REV A • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com
 LINEAR TECHNOLOGY CORPORATION 1998