LM2696 3A, Constant On Time Buck Regulator General Description Features The LM2696 is a pulse width modulation (PWM) buck regulator capable of delivering up to 3A into a load. The control loop utilizes a constant on-time control scheme with input voltage feed forward. This provides a topology that has excellent transient response without the need for compensation. The input voltage feed forward ensures that a constant switching frequency is maintained across the entire VIN range. n n n n n n n n n The LM2696 is capable of switching frequencies in the range of 100 kHz to 500 kHz. Combined with an integrated 130 mΩ high side NMOS switch the LM2696 can utilize small sized external components and provide high efficiency. An internal soft-start and power-good flag are also provided to allow for simple sequencing between multiple regulators. The LM2696 is available with an adjustable output in an exposed pad TSSOP-16 package. Input voltage range of 4.5V–24V Constant On-Time No compensation needed Maximum Load Current of 3A Switching frequency of 100 kHz–500 kHz Constant frequency across input range TTL compatible shutdown thresholds Low standby current of 12 µA 130 mΩ internal MOSFET switch Applications n High efficiency step-down switching regulators n LCD Monitors n Set-Top Boxes Typical Application Circuit 20153401 © 2005 National Semiconductor Corporation DS201534 www.national.com LM2696 3A, Constant On Time Buck Regulator October 2005 LM2696 Connection Diagram Top View 20153402 eTSSOP-16 Package Pin Descriptions Pin # Name 1, 2, 3 SW 4 CBOOT 5 AVIN 6 EXTVCC 7 FB Function Switching node Bootstrap capacitor input Analog voltage input Output of internal regulator for decoupling Feedback signal from output 8 N/C No connect 9 GND Ground 10 SS 11 PGOOD 12 RON 13 SD 14, 15, 16 PVIN - Exposed Pad Soft-start pin Power-good flag, open drain output Sets the switch on-time dependent on current Shutdown pin Power voltage input Must be connected to ground Ordering Information Order Number Package Type NSC Package Drawing Supplied As LM2696MXA eTSSOP-16 MXA16A 92 units/rail LM2696MXAX eTSSOP-16 MXA16A 2,500 Units Tape and Reel www.national.com 2 Junction Temperature If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. Lead Temperature (Soldering, 10 sec.) 260˚C Minimum ESD Rating 1.5 kV +150˚C Voltages from the indicated pins to GND AVIN −0.3V to +26V PVIN −0.3V to (AVIN+0.3V) CBOOT Operating Range Junction Temperature −0.3V to +33V CBOOT to SW −0.3V to +7V FB, SD, SS, PGOOD −0.3V to +7V Storage Temperature Range −65˚C to +150˚C −40˚C to +125˚C AVIN to GND 4.5V to 24V PVIN 4.5V to 24V Electrical Characteristics Specifications with standard typeface are for TJ = 25˚C, and those in boldface type apply over the full Operating Temperature Range (TJ = −40˚C to +125˚C). Minimum and Maximum limits are guaranteed through test, design or statistical correlation. Typical values represent the most likely parametric norm at TJ = 25˚C and are provided for reference purposes only. Unless otherwise specified VIN = 12V. Symbol Parameter Condition VFB Feedback Pin Voltage VIN = 4.5V to 24V ISW = 0A to 3A Min Typ Max Units 1.225 1.254 1.282 V ICL Switch Current Limit VCBOOT = VSW + 5V 4.9 6.4 A RDS_ON Switch On Resistance ISW = 3A 0.13 0.22 Ω IQ Operating Quiescent Current VFB = 1.5V 1.3 2 mA Rising VIN VUVLO AVIN Under Voltage Lockout VUVLO HYS AVIN Under Voltage Lockout Hysteresis ISD Shutdown Quiescent Current VSD = 0V kON Switch On-Time Constant ION = 50 µA to 100 µA VD ON RON Voltage TOFF_MIN Minimum Off Time TON MIN Minimum On-time VEXTVCC EXTVCC Voltage ∆VEXTVCC EXTVCC Load Regulation IEXTVCC = 0 µA to 50 µA VPWRGD PGOOD Threshold (PGOOD Transition from Low to High) With respect to VFB VPG_HYS PGOOD Hysteresis 3.6 3.9 4.125 4.3 V 60 120 mV 12 25 µA 50 66 82 µA µs 0.35 0.65 0.95 V 165 12 250 30 ns µs FB = 1.24V FB = 0V 400 3.30 91.5 ns 3.65 4.00 V 0.03 0.5 % 93.5 95.5 % 1 2.1 % IOL PGOOD Low Sink Current IOH PGOOD High Leakage Current IFB Feedback Pin Bias Current VFB = 1.2V ISS_SOURCE Soft-Start Pin Source Current VSS = 0V ISS SINK Soft-Start Pin Sink Current VSS = 1.2V VSD = 0V 15 ISD Shutdown Pull-Up Current VSD = 0V 1 VIH SD Pin Minimum High Input Level VIL SD Pin Maximum Low Input Level θJ-A Thermal Resistance VPGOOD = 0.4V 0.5 0.7 2 mA 50 nA 0 nA 1 1.4 µA mA 3 1.8 µA V 0.6 35.1 V ˚C/W Note 1: Absolute Maximum Ratings indicate limits beyond which damage may occur to the device. Operating Ratings indicate conditions for which the device is intended to be functional, but do not guarantee specific performance limits. For guaranteed specifications, see Electrical Characteristics. Note 2: Without PCB copper enhancements. The maximum power dissipation must be derated at elevated temperatures and is limited by TJMAX (maximum junction temperature), θJ-A (junction to ambient thermal resistance) and TA (ambient temperature). The maximum power dissipation at any temperature is: PDissMAX = (TJMAX - TA) /θJ-A up to the value listed in the Absolute Maximum Ratings. θJ-A for TSSOP-16 package is 38.1˚C/W, TJMAX = 125˚C. 3 www.national.com LM2696 Absolute Maximum Ratings (Note 1) LM2696 Typical Performance Characteristics IQ vs Temp IQ vs VIN 20153403 20153404 IQ in Shutdown vs Temp IQ vs VIN in Shutdown 20153405 20153406 Shutdown Thresholds vs Temp EXTVCC vs Temp 20153408 20153407 www.national.com 4 LM2696 Typical Performance Characteristics (Continued) EXTVCC vs VIN EXTVCC vs Load Current 20153410 20153409 Feedback Threshold Voltage vs Temp kON vs Temp 20153412 20153411 Switch ON Time vs RON Pin Current Min Off-Time vs Temp 20153414 20153413 5 www.national.com LM2696 Typical Performance Characteristics (Continued) Max and Min Duty-Cycle vs Freq (Min TON = 400 ns, Min TOFF = 200 ns) FET Resistance vs Temp 20153415 20153416 RON Pin Voltage vs Temp Current Limit vs Temp 20153418 20153417 www.national.com 6 LM2696 Block Diagram 20153419 hysteretic or constant on-time, require a minimum ESR. A minimum ESR is required so that the control signal will be dominated by ripple that is in phase with the switchpin. Using a control signal dominated by voltage ripple that is in phase with the switchpin eliminates the need for compensation, thus reducing parts count and simplifying design. Alternatively, an RC feed forward scheme may be used to eliminate the need for a minimum ESR. Application Information CONSTANT ON-TIME CONTROL OVERVIEW The LM2696 buck DC-DC regulator is based on the constant on-time control scheme. This topology relies on a fixed switch on-time to regulate the output. The on-time can be set manually by adjusting the size of an external resistor (RON). The LM2696 automatically adjusts the on-time inversely with the input voltage (AVIN) to maintain a constant frequency. In continuous conduction mode (CCM) the frequency depends only on duty cycle and on-time. This is in contrast to hysteretic regulators where the switching frequency is determined by the output inductor and capacitor. In discontinuous conduction mode (DCM), experienced at light loads, the frequency will vary according to the load. This leads to high efficiency and excellent transient response. The on-time will remain constant for a given VIN unless an over-current or over-voltage event is encountered. If these conditions are encountered the FET will turn off for a minimum pre-determined time period. This minimum TOFF (250 ns) is internally set and cannot be adjusted. After the TOFF period has expired the FET will remain off until the comparator trip-point has been reached. Upon passing this trip-point the FET will turn back on, and the process will repeat. Switchers that regulate using an internal comparator to sense the output voltage for switching decisions, such as INTERNAL OPERATION UNDER-VOLTAGE COMPARATOR An internal comparator is used to monitor the feedback pin for sensing under-voltage output events. If the output voltage drops below the UVP threshold the power-good flag will fall. ON-TIME GENERATOR SHUTDOWN The on-time for the LM2696 is inversely proportional to the input voltage. This scheme of on-time control maintains a constant frequency over the input voltage range. The ontime can be adjusted by using an external resistor connected between the PVIN and RON pins. CURRENT LIMIT The LM2696 contains an intelligent current limit off-timer. If the peak current in the internal FET exceeds 4.9A the present on-time is terminated; this is a cycle-by-cycle current 7 www.national.com LM2696 Application Information flowing into the RON pin and is directly related to the on-time pulse. Connecting a resistor from this pin to PVIN allows the switching frequency to remain constant as the input voltage changes. In normal operation this pin is approximately 0.65V above GND. In shutdown, this pin becomes a high impedance node to prevent current flow. (Continued) limit. Following the termination of the on-time, a nonresetable extended off timer is initiated. The length of the off-time is proportional to the feedback voltage. When FB = 0V the off-time is preset to 20 µs. This condition is often a result of in short circuit operation when a maximum amount of off-time is required. This amount of time ensures safe short circuit operation up to the maximum input voltage of 24V. In cases of overload (not complete short circuit, FB > 0V) the current limit off-time is reduced. Reduction of the off-time during smaller overloads reduces the amount of fold back. This also reduces the initial startup time. The on time may be exoressed as: Where VIN is the voltage at the high side of the RON resistor (typically PVIN), VD is the diode voltage present at the RON pin (0.65V typical), RON is in kΩ, and kON is a constant value set internally (66 µA • µs nominal). This equation can be re-arranged such that RON is a function of switching frequency: N-CHANNEL HIGH SIDE SWITCH AND DRIVER The LM2696 utilizes an integrated N-Channel high side switch and associated floating high voltage gate driver. This gate driver circuit works in conjunction with an external bootstrap capacitor and an internal diode. The minimum off-time (165 ns) is set to ensure that the bootstrap capacitor has sufficient time to charge. THERMAL SHUTDOWN Where fSW is in kHz. In CCM the frequency may be determined using the relationship: An internal thermal sensor is incorporated to monitor the die temperature. If the die temp exceeds 165oC then the sensor will trip causing the part to stop switching. Soft-start will restart after the temperature falls below 155oC. COMPONENT SELECTION As with any DC-DC converter, numerous trade-offs are present that allow the designer to optimize a design for efficiency, size and performance. These trade-offs are taken into consideration throughout this section. The first calculation for any buck converter is duty cycle. Ignoring voltage drops associated with parasitic resistances and non-ideal components, the duty cycle may be expressed as: (TON is in µs) Which is typically used to set the switching frequency. Under no condition should a bypass capacitor be connected to the RON pin. Doing so couples any AC perturbations into the pin and prevents proper operation. INDUCTOR SELECTION Selecting an inductor is a process that may require several iterations. The reason for this is that the size of the inductor influences the amount of ripple present at the output that is critical to the stability of an adaptive on-time circuit. Typically, an inductor is selected such that the maximum peak-to-peak ripple current is equal to 30% of the maximum load current. The inductor current ripple (∆IL) may be expressed as: A duty cycle relationship that considers the voltage drop across the internal FET and voltage drop across the external catch diode may be expressed as: Therefore, L can be initially set to the following by applying the 30% guideline: Where VD is the forward voltage of the external catch diode (DCATCH) and VSW is the voltage drop across the internal FET. FREQUENCY SELECTION Switching frequency affects the selection of the output inductor, capacitor, and overall efficiency. The trade-offs in frequency selection may be summarized as; higher switching frequencies permit use of smaller inductors possibly saving board space at the trade-off of lower efficiency. It is recommended that a nominal frequency of 300 kHz should be used in the initial stages of design and iterated if necessary. The switching frequency of the LM2696 is set by the resistor connected to the RON pin. This resistor controls the current www.national.com The other features of the inductor that should be taken into account are saturation current and core material. A shielded inductor or low profile unshielded inductor is recommended to reduce EMI. 8 (Continued) The value of Rff should be large in order to prevent any potential offset in VOUT. Typically the value of Rff is on the order of 1 MΩ and the value of RFB1 should be less than 10 kΩ. The large difference in resistor values minimizes output voltage offset errors in DCM. The value of the capacitor may be selected using the following relationship: OUTPUT CAPACITOR The output capacitor size and ESR have a direct affect on the stability of the loop. This is because the adaptive on-time control scheme works by sensing the output voltage ripple and switching appropriately. The output voltage ripple on a buck converter can be approximated by assuming that the AC inductor ripple current flows entirely into the output capacitor and the ESR of the capacitor creates the voltage ripple. This is expressed as: ∆VOUT≈ ∆IL • RESR Where the on-time (TON_MIN) is in µs, and the resistance (Rff) is in MΩ. To ensure stability, two constraints need to be met. These constraints are the voltage ripple at the feedback pin must be greater than some minimum value and the voltage ripple must be in phase with the switch pin. The ripple voltage necessary at the feedback pin may be estimated using the following relationship: ∆VFB > −0.057 • fSW + 35 Where fSW is in kHz and ∆VFB is in mV. FEEDBACK RESISTORS The feedback resistors are used to scale the output voltage to the internal reference value such that the loop can be regulated. The feedback resistors should not be made arbitrarily large as this creates a high impedance node at the feedback pin that is more susceptible to noise. Typically, RFB2 is on the order of 1 kΩ. To calculate the value of RFB1, one may use the relationship: This minimum ripple voltage is necessary in order for the comparator to initiate switching. The voltage ripple at the feedback pin must be in-phase with the switch. Because the ripple due to the capacitor charging and capacitor ESR are out of phase, the ripple due to capacitor ESR must dominate. The ripple at the output may be calculated by multiplying the feedback ripple voltage by the gain seen through the feedback resistors. This gain H may be expressed as: Where VFB is the internal reference voltage that can be found in the electrical characteristics table (1.254V typical). The output voltage value can be set in a precise manner by taking into account the fact that the reference voltage is regulating the bottom of the output ripple as opposed to the average value. This relationship is shown in Figure 2. To simplify design and eliminate the need for high ESR output capacitors, an RC network may be used to feed forward a signal from the switchpin to the feedback (FB) pin. See the ‘Ripple Feed Forward’ section for more details. Typically, the best performance is obtained using POSCAPs, SP CAPs, tantalum, Niobium Oxide, or similar chemistry type capacitors. Low ESR ceramic capacitors may be used in conjunction with the RC feed forward scheme; however, the feed forward voltage at the feedback pin must be greater than 30 mV. RIPPLE FEED FORWARD An RC network may be used to eliminate the need for high ESR capacitors. Such a network is connected as shown in Figure 1. 20153428 FIGURE 1. RC Feed Forward Network 9 www.national.com LM2696 Application Information LM2696 Application Information (Continued) 20153431 FIGURE 2. Average and Ripple Output Voltages AVIN CAPACITOR AVIN is the analog bias rail of the device. It should be bypassed externally with a small (1 µF) ceramic capacitor to prevent unwanted noise from entering the device. In a shutdown state the current needed by AVIN will drop to approximately 12 µA, providing a low power sleep state. In most cases of operation, AVIN is connected to PVIN; however, it is possible to have split rail operation where AVIN is at a higher voltage than PVIN. AVIN should never be lower than PVIN. Splitting the rails allows the power conversion to occur from a lower rail than the AVIN operating range. It can be seen that the average output voltage is higher than the gained up reference by exactly half the output voltage ripple. The output voltage may then be appended according to the voltage ripple. The appended VOUT term may be expressed using the relationship: One should note that for high output voltages ( > 5V), a load of approximately 15 mA may be required for the output voltage to reach the desired value. SOFT-START CAPACITOR The SS capacitor is used to slowly ramp the reference from 0V to its final value of 1.25V (during shutdown this pin will be discharged to 0V). This controlled startup ability eliminates large in-rush currents in an attempt to charge up the output capacitor. By changing the value of this capacitor, the duration of the startup may be changed accordingly. The startup time may be calculated using the following relationship: INPUT CAPACITOR Because PVIN is the power rail from which the output voltage is derived, the input capacitor is typically selected according to the load current. In general, package size and ESR determine the current capacity of a capacitor. If these criteria are met, there should be enough capacitance to prevent impedance interactions with the source. In general, it is recommended to use a low ESR, high capacitance electrolytic and ceramic capacitor in parallel. Using two capacitors in parallel ensures adequate capacitance and low ESR over the operating range. The Sanyo MV-WX series electrolytic capacitors and a ceramic capacitor with X5R or X7R dielectric are an excellent combination. To calculate the input capacitor RMS, one may use the following relationship: Where ISS is the soft-start pin source current (1 µA typical) that may be found in the electrical characteristics table. While the CSS capacitor can be sized to meet the startup requirements, there are limitations to its size. If the capacitor is too small, the soft-start will have little effect as the reference voltage is rising faster than the output capacitor can be charged causing the part to go into current limit. Therefore a minimum soft-start time should be taken into account. This can be determined by: that can be approximated by, Typical values are 470 µF for the electrolytic capacitor and 0.1 µF for the ceramic capacitor. www.national.com 10 turned on. The recommendation is to use a 10 kΩ–100 kΩ resistor. This range of values is a compromise between rise time and power dissipation. (Continued) While COUT and VOUT control the slew rate of the output voltage, the total amount of time the LM2696 takes to startup is dependent on two other terms. See the “Startup” section for more information. CATCH DIODE The catch or freewheeling diode acts as the bottom switch in a non-synchronous buck switcher. Because of this, the diode has to handle the full output current whenever the FET is not conducting. Therefore, it must be sized appropriately to handle the current. The average current through the diode can be calculated by the equation: ID_AVG = IOUT • (1–D) EXTVCC CAPACITOR External VCC is a 3.65V rail generated by an internal subregulator that powers the parts internal circuitry. This rail should be bypassed with a 1 µF ceramic capacitor (X5R or equivalent dielectric). Although EXTVCC is for internal use, it can be used as an external rail for extremely light loads ( < 50 µA). If EXTVCC is accidentally shorted to GND the part is protected by a 5 mA current limit. This rail also has an under-voltage lockout that will prevent the part from switching if the EXTVCC voltage drops. Care should also be taken to ensure that the reverse voltage rating of the diode is adequate. Whenever the FET is conducting the voltage across the diode will be approximately equal to VIN. It is recommended to have a reverse rating that is equal to 120% of VIN to ensure adequate guard banding against any ringing that could occur on the switch node. SHUTDOWN Selection of the catch diode is critical to overall switcher performance. To obtain the optimal performance, a Schottky diode should be used due to their low forward voltage drop and fast recovery. The state of the shutdown pin enables the device or places it in a sleep state. This pin has an internal pull-up and may be left floating or connected to a high logic level. Connecting this pin to GND will shutdown the part. Shutting down the part will prevent the part from switching and reduce the quiescent current drawn by the part. This pin must be bypassed with a 1 nF ceramic capacitor (X5R or Y5V) to ensure proper logic thresholds. BYPASS CAPACITOR A bypass capacitor must be used on the AVIN line to help decouple any noise that could interfere with the analog circuitry. Typically, a small (1 µF) ceramic capacitor is placed as close as possible to the AVIN pin. CBOOT CAPACITOR The purpose of an external bootstrap capacitor is to turn the FET on by using the SW node as a pedestal. This allows the voltage on the CBOOT pin to be greater than VIN. Whenever the catch diode is conducting and the SW node is at GND, an internal diode will conduct that charges the CBOOT capacitor to approximately 4V. When the SW node rises, the CBOOT pin will rise to approximately 4V above the SW node. For optimal performance, a 0.1 µF ceramic capacitor (X5R or equivalent dielectric) should be used. EXTERNAL OPERATION STARTUP The total startup time, from the initial VIN rise to the time VOUT reaches its nominal value is determined by three separate steps. Upon the rise of VIN, the first step to occur is that the EXTVCC voltage has to reach its nominal output voltage of 3.65V before the internal circuitry is active. This time is dictated by the output capacitance (1 µF) and the current limit of the regulator (5 mA typical), which will always be on the order of 730 µs. Upon reaching its steady state value, an internal delay of 200 µs will occur to ensure stable operation. Upon completion the LM2696 will begin switching and the output will rise. The rise time of the output will be governed by the soft-start capacitor. To highlight these three steps a timing diagram please refer to Figure 3. PGOOD RESISTOR The PGOOD resistor is used to pull the PGOOD pin high whenever a steady state operating range is achieved. This resistor needs to be sized to prevent excessive current from flowing into the PGOOD pin whenever the open drain FET is 11 www.national.com LM2696 Application Information LM2696 Application Information (Continued) 20153437 FIGURE 3. Startup Timing Diagram UNDER- & OVER-VOLTAGE CONDITIONS The LM2696 has a built in under-voltage comparator that controls PGOOD. Whenever the output voltage drops below the set threshold, the PGOOD open drain FET will turn on pulling the pin to ground. For an over-voltage event, there is no separate comparator to control PGOOD. However, the loop responds to prevent this event from occurring because the error comparator is essentially sensing an OVP event. If the output is above the feedback threshold then the part will not switch back on; therefore, the worst-case condition is one on-time pulse. short. This is to prevent the output from dropping or any fold back from occurring if a momentary short occurred because of a transient or load glitch. If a short circuit were present, the off-time would extend to approximately 12 µs. This ensures that the inductor current will reach a low value (approximately 0A) before the next switching cycle occurs. The extended off-time prevents runaway conditions caused by hard shorts and high side blanking times. If the part is in an over current condition, the output voltage will begin to drop as shown in Figure 4. If the output voltage is dropping and the current is below the current limit threshold, (I1), the part will assert a pulse (t2) after a minimum off-time (t1). This is in an attempt to raise the output voltage. If the part is in an over current condition and the output voltage is below the regulation value (VL) as shown in Figure 4, the part will assert a pulse of minimal width (t4) and extend the off-time (t5). In the event that the voltage is below the regulation value (VL) and the current is below the current limit value, the part will assert two (or more) pulses separated by some minimal off-time (t1). CURRENT LIMIT The LM2696 utilizes a peak-detect current limit that senses the current through the FET when conducting and will immediately terminate the on-pulse whenever the peak current exceeds the threshold (4.9A typical). In addition to terminating the present on-pulse, it enforces a mandatory off-time that is related to the feedback voltage. If current limit trips and the feedback voltage is close to its nominal value of 1.25V, the off-time imposed will be relatively www.national.com 12 LM2696 Application Information (Continued) 20153438 FIGURE 4. Fault Condition Timing MODES OF OPERATION Legend: t1: Min off-time (165 ns typical) t2: On-time (set by the user) t3: Min off-time (165 ns typical) t4: Blanking time (165 ns typical) t5: Extended off-time (12 µs typical) VL: UVP threshold The last benefit of this scheme is when the short circuit is removed, and full load is re-applied, the part will automatically recover into the load. The variation in the off-time removes the constraints of other frequency fold back systems where the load would typically have to be reduced. Since the LM2696 utilizes a catch diode, whenever the load current is reduced to a point where the inductor ripple is greater than two times the load current, the part will enter discontinuous operation. This is because the diode does not permit the inductor current to reverse direction. The point at which this occurs is the critical conduction boundary and can be calculated by the following equation: One advantage of the adaptive on-time control scheme is that during discontinuous conduction mode the frequency will gradually decrease as the load current decreases. In DCM the switching frequency may be determined using the relationship: It can be seen that there will always be some minimum switching frequency. The minimum switching frequency is determined by the parameters above and the minimum load presented by the feedback resistors. If there is some minimum frequency of operation the feedback resistors may be sized accordingly. The adaptive on-time control scheme is effectively a pulseskipping mode, but since it is not tied directly to an internal clock, its pulse will only occur when needed. This is in contrast to schemes that synchronize to a reference clock frequency. The constant on-time pulse-skipping/DCM mode minimizes output voltage ripple and maximizes efficiency. 20153439 FIGURE 5. Normalized Output Voltage Versus Load Current 13 www.national.com LM2696 Application Information (Continued) Several diagrams are shown in Figure 6 illustrating continuous conduction mode (CCM), discontinuous conduction mode (DCM), and the boundary condition. 20153442 20153443 20153444 20153445 FIGURE 6. Modes of Operation It can be seen that in DCM, whenever the inductor runs dry the SW node will become high impedance. Ringing will occur as a result of the LC tank circuit formed by the inductor and the parasitic capacitance at the SW node. www.national.com 14 (Continued) LINE REGULATION The LM2696 regulates to the lowest point of the output voltage (VL in Figure 8 ). This is to say that the output voltage may be represented by a waveform that is some average voltage with ripple. The LM2696 will regulate to the trough of the ripple. 20153446 FIGURE 7. Parasitic Tank Circuit at the Switchpin 20153447 FIGURE 8. Average Output Voltage and Regulation Point PGC = AVIN + VGS • QGS • fSW PSW = 0.5 • VIN · IOUT • (tr + tf) • fSW The output voltage is given by the following relationship: Typical values are: RDS_ON = 130 mΩ VGS = 4V QGS = 13.3 nC tr = 3.8 ns tf = 4.5 ns as discussed in the Feedback Resistor section of this document. TRANSIENT RESPONSE Constant on-time architectures have inherently excellent transient line and load response. This is because the control loop is extremely fast. Any change in the line or load conditions will result in a nearly instantaneous response in the PWM off time. If one considers the switcher response to be nearly cycleby-cycle, and amount of energy contained in a single PWM pulse, there will be very little change in the output for a given change in the line or load. Power loss due to catch diode: PD = (1-D) • (IOUT • Vf) Power loss due to DCR and ESR: PDCR = IOUT2 • RDCR PESR_OUTPUT = IRIPPLE2/√12 • RESR_OUTPUT PESR_INPUT = IOUT2(D(1-D)) • RESR_INPUT EFFICIENCY The constant on-time architecture features high efficiency even at light loads. The ability to achieve high efficiency at light loads is due to the fact that the off-time will become necessarily long at light loads. Having extended the off-time, there is little mechanism for loss during this interval. The efficiency is easily estimated using the following relationships: Power loss due to FET: PFET = PC + PGC + PSW Where: PC = D • (IOUT2 • RDS_ON) Power loss due to Controller: PCONT = VIN • IQ IQ is typically 1.3 mA The efficiency may be calculated as shown below: Total power loss = PFET + PD + PDCR + PESR_OUTPUT + PESR_INPUT + PCONT Power Out = IOUT • VOUT 15 www.national.com LM2696 Application Information LM2696 Application Information PIN is the input power in Watts (PIN = VIN·IIN) θJA is the thermal coefficient of the LM2696 TA is the ambient temperature in oC (Continued) LAYOUT CONSIDERATIONS The LM2696 regulation and under-voltage comparators are very fast and will respond to short duration noise pulses. Layout considerations are therefore critical for optimum performance. The components at pins 5, 6, 7, 12 and 13 should be as physically close as possible to the IC, thereby minimizing noise pickup in the PC traces. If the internal dissipation of the LM2696 produces excessive junction temperatures during normal operation, good use of the PC board’s ground plane can help considerably to dissipate heat. The exposed pad on the bottom of the TSSOP-16 package can be soldered to a ground plane on the PC board, and that plane should extend out from beneath the IC to help dissipate the heat. Use of several vias beneath the part is also an effective method of conducting heat. Additionally, the use of wide PC board traces, where possible, can also help conduct heat away from the IC. Judicious positioning of the PC board within the end product, along with use of any available air flow (forced or natural convection) can help reduce the junction temperatures. Traces in the power plane (Figure 9) should be short and wide to minimize the trace impedance; they should also occupy the smallest area that is reasonable to minimize EMI. Sizing the power plane traces is a tradeoff between current capacity, inductance, and thermal dissipation. For more information on layout considerations, please refer to National Semiconductor Application Note AN-1229. PRE-BIAS LOAD STARTUP Should the LM2696 start into a pre-biased load the output will not be pulled low. This is because the part is asynchronous and cannot sink current. The part will respond to a pre-biased load by simply enabling PWM high or extending the off-time until regulation is achieved. This is to say that if the output voltage is greater than the regulation voltage the off-time will extend until the voltage discharges through the feedback resistors. If the load voltage is greater than the regulation voltage, a series of pulses will charge the output capacitor to its regulation voltage. THERMAL CONSIDERATIONS The thermal characteristics of the LM2696 are specified using the parameter θJA, which relates the junction temperature to the ambient temperature. While the value of θJA is specific to a given set of test parameters (including board thickness, number of layers, orientation, etc), it provides the user with a common point of reference. To obtain an estimate of a devices junction temperature, one may use the following relationship: TJ = PIN (1-Efficiency) x θJA + TA Where: TJ is the junction temperature in oC 20153450 FIGURE 9. Bold Traces Are In The Power Plane www.national.com 16 LM2696 Application Information (Continued) 20153451 FIGURE 10. 5V-to-2.5V Voltage Applications Circuit Bill of Materials (Figure 10: Medium Voltage Board, 5V-to-2.5V conversion, fsw = 300 kHz) Designator Function Description Vendor Part Number CIN Input Cap 470 µF Sanyo 10MV470WX CBY Bypass Cap 0.1 µF Vishay VJ0805Y104KXAM CSS Soft-Start Cap 0.01 µF Vishay VJ080JY103KXX CEXT EXTVCC 1 µF Vishay VJ0805Y105JXACW1BC CBOOT Boot 0.1 µF Vishay VJ0805Y104KXAM VJ0805Y105JXACW1BC CAVIN Analog VIN 1 µF Vishay COUT Output Cap 47 µF AVX TPSW476M010R0150 CSD Shutdown Cap 1 nF Vishay VJ0805Y102KXXA RFB1 High Side FB Res 1 kΩ Vishay CRCW08051001F RFB2 Low Side RB Res 1 kΩ Vishay CRCW08051001F RON On Time Res 143 kΩ Vishay CRCW08051433F DCATCH Boot Diode 40V @ 3A Diode Central Semi CMSH3-40M-NST L Output Inductor 6.8 uH, 4.9A ISAT Coilcraft MSS1260-682MX 20153452 FIGURE 11. 12V-to 3.3V Voltage Applications Circuit 17 www.national.com LM2696 Application Information (Continued) Bill of Materials (Figure 11: Medium Voltage Board, 12V-to-3.3V conversion, fsw = 300 kHz) Designator Function Description Vendor Part Number CIN Input Cap 560 µF Sanyo 35MV560WX CBY Bypass Cap 0.1 µF Vishay VJ0805Y104KXAM CSS Soft-Start Cap 0.01 µF Vishay VJ080JY103KXX CEXT EXTVCC 1 µF Vishay VJ0805Y105JXACW1BC CBOOT Boot 0.1 µF Vishay VJ0805Y104KXAM CAVIN Analog VIN 1 µF Vishay VJ0805Y105JXACW1BC COUT Output Cap 100 µF Sanyo 6SVPC100M CSD Shutdown Cap 1 nF Vishay VJ0805Y102KXXA Cff Feedforward Cap 560 pF Vishay VJ0805A561KXXA Rff Feedforward Res 1 MΩ Vishay CRCW08051004F RFB1 High Side FB Res 1.62 kΩ Vishay CRCW08051621F RFB2 Low Side RB Res 1 kΩ Vishay CRCW08051001F RON On Time Res 143 kΩ Vishay CRCW08051433F DCATCH Boot Diode 40V @ 3A Diode Central Semi CMSH3-40M-NST L Output Inductor 10 uH, 5.4A ISAT Coilcraft MSS1278-103MX www.national.com 18 LM2696 3A, Constant On Time Buck Regulator Physical Dimensions inches (millimeters) unless otherwise noted eTSSOP-16 Package Order Number LM2696MXA or LM2696MXAX NS Package Number MXA16A National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves the right at any time without notice to change said circuitry and specifications. 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