LM3402/LM3402HV 0.5A Constant Current Buck Regulator for Driving High Power LEDs General Description Features The LM3402/02HV are monolithic switching regulators designed to deliver constant currents to high power LEDs. Ideal for automotive, industrial, and general lighting applications, they contain a high-side N-channel MOSFET switch with a current limit of 735 mA (typical) for step-down (Buck) regulators. Hysteretic control with controlled on-time coupled with an external resistor allow the converter output voltage to adjust as needed to deliver a constant current to series and series - parallel connected arrays of LEDs of varying number and type, LED dimming by pulse width modulation (PWM), broken/open LED protection, low-power shutdown and thermal shutdown complete the feature set. Integrated 0.5A N-channel MOSFET VIN Range from 6V to 42V (LM3402) VIN Range from 6V to 75V (LM3402HV) 500 mA Output Current Over Temperature Cycle-by-Cycle Current Limit No Control Loop Compensation Required Separate PWM Dimming and Low Power Shutdown Supports all-ceramic output capacitors and capacitor-less outputs n Thermal shutdown protection n MSOP-8 Package n n n n n n n n Applications n n n n n LED Driver Constant Current Source Automotive Lighting General Illumination Industrial Lighting Typical Application 20192101 © 2006 National Semiconductor Corporation DS201921 www.national.com LM3402/LM3402HV 0.5A Constant Current Buck Regulator for Driving High Power LEDs October 2006 LM3402/LM3402HV Connection Diagram 20192102 8-Lead Plastic MSOP-8 Package NS Package Number MUA08A Ordering Information Order Number Package Type NSC Package Drawing LM3402MM Supplied As 1000 units on tape and reel LM3402MMX MSOP-8 LM3402HVMM MUA08A LM3402HVMMX 3500 units on tape and reel 1000 units on tape and reel 3500 units on tape and reel Pin Descriptions Pin(s) Name Description 1 SW Switch pin 2 BOOT MOSFET drive bootstrap pin 3 DIM Input for PWM dimming 4 GND Ground pin 5 CS Current sense feedback pin 6 RON On-time control pin 7 VCC 8 VIN www.national.com Application Information Connect this pin to the output inductor and Schottky diode. Connect a 10 nF ceramic capacitor from this pin to SW. Connect a logic-level PWM signal to this pin to enable/disable the power FET and reduce the average light output of the LED array. Connect this pin to system ground. Set the current through the LED array by connecting a resistor from this pin to ground. A resistor connected from this pin to VIN sets the regulator controlled on-time. Output of the internal 7V linear Bypass this pin to ground with a minimum 0.1 µF ceramic capacitor regulator with X5R or X7R dielectric. Input voltage pin Nominal operating input range is 6V to 42V (LM3402) or 6V to 75V (LM3402HV). 2 Absolute Maximum Ratings (LM3402HV)(Note 1) If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. VIN to GND -0.3V to 45V VIN to GND -0.3V to 76V BOOT to GND -0.3V to 59V BOOT to GND -0.3V to 90V SW to GND -1.5V SW to GND -1.5V BOOT to VCC -0.3V to 45V BOOT to VCC -0.3V to 76V BOOT to SW -0.3V to 14V BOOT to SW -0.3V to 14V VCC to GND -0.3V to 14V VCC to GND -0.3V to 14V DIM to GND -0.3V to 7V DIM to GND -0.3V to 7V CS to GND -0.3V to 7V CS to GND -0.3V to 7V RON to GND -0.3V to 7V RON to GND -0.3V to 7V Junction Temperature 150˚C Junction Temperature 150˚C Storage Temp. Range -65˚C to 125˚C Storage Temp. Range -65˚C to 125˚C ESD Rating (Note 2) 2kV ESD Rating (Note 2) Soldering Information Lead Temperature (Soldering, 10sec) Infrared/Convection Reflow (15sec) Lead Temperature (Soldering, 10sec) 260˚C 235˚C Infrared/Convection Reflow (15sec) Operating Ratings (LM3402) (Note 1) VIN Junction Temperature Range Thermal Resistance θJA (Note 3) 2kV Soldering Information 260˚C 235˚C Operating Ratings (LM3402HV) (Note 1) VIN 6V to 42V −40˚C to +125˚C Junction Temperature Range Thermal Resistance θJA (Note 3) 200˚C/W 3 6V to 75V −40˚C to +125˚C 200˚C/W www.national.com LM3402/LM3402HV Absolute Maximum Ratings (LM3402)(Note 1) LM3402/LM3402HV Electrical Characteristics VIN = 24V unless otherwise indicated. Typicals and limits appearing in plain type apply for TA = TJ = +25˚C. (Note 4) Limits appearing in boldface type apply over full Operating Temperature Range. Datasheet min/max specification limits are guaranteed by design, test, or statistical analysis. LM3402 Symbol Parameter Conditions Min Typ Max Units SYSTEM PARAMETERS tON-1 On-time 1 VIN = 10V, RON = 200 kΩ 2.1 2.75 3.4 µs tON-2 On-time 2 VIN = 40V, RON = 200 kΩ 490 650 810 ns Min Typ Max Units LM3402HV Symbol Parameter Conditions SYSTEM PARAMETERS tON-1 On-time 1 VIN = 10V, RON = 200 kΩ 2.1 2.75 3.4 µs tON-2 On-time 2 VIN = 70V, RON = 200 kΩ 290 380 470 ns Typ Max Units 200 206 LM3402/LM3402HV Symbol Parameter Conditions Min REGULATION AND OVER-VOLTAGE COMPARATORS VREF-REG CS Regulation Threshold CS Decreasing, SW turns on VREF-0V CS Over-voltage Threshold CS Increasing, SW turns off 300 mV ICS CS Bias Current CS = 0V 0.1 µA VSD-TH Shutdown Threshold RON / SD Increasing VSD-HYS Shutdown Hysteresis RON / SD Decreasing 40 mV Minimum Off-time CS = 0V 300 ns 194 mV SHUTDOWN 0.3 0.7 1.05 V OFF TIMER tOFF-MIN INTERNAL REGULATOR VCC-REG VCC Regulated Output 6.6 7 7.4 V VIN-DO VIN - VCC Dropout ICC = 5 mA, 6.0V < VIN < 8.0V 300 VCC-BP-TH VCC Bypass Threshold VIN Increasing 8.8 V VCC-BP-HYS VCC Bypass Hysteresis VIN Decreasing 225 mV VCC-Z-6 VCC Output Impedance (0 mA < ICC < 5 mA) VIN = 6V 55 Ω VIN = 8V 50 VIN = 24V 0.4 VCC-LIM VCC Current Limit (Note 3) VIN = 24V, VCC = 0V 16 mA VCC-UV-TH VCC Under-voltage Lock-out Threshold VCC Increasing 5.25 V VCC-UV-HYS VCC Under-voltage Lock-out Hysteresis VCC Decreasing 150 mV VCC-UV-DLY VCC Under-voltage Lock-out Filter Delay 100 mV Overdrive 3 µs IIN-OP IIN Operating Current Non-switching, CS = 0V 600 900 µA IIN-SD IIN Shutdown Current RON / SD = 0V 90 180 µA 735 940 mA VCC-Z-8 VCC-Z-24 mV CURRENT LIMIT ILIM Current Limit Threshold 530 DIM COMPARATOR VIH Logic High DIM Increasing VIL Logic Low DIM Decreasing IDIM-PU DIM Pull-up Current DIM = 1.5V www.national.com 4 2.2 V 0.8 75 V µA LM3402/LM3402HV Symbol (Continued) Parameter Conditions Min Typ Max Units 0.7 1.5 Ω 3 4 V N-MOSFET AND DRIVER RDS-ON Buck Switch On Resistance ISW = 200mA, BOOT-SW = 6.3V VDR-UVLO BOOT Under-voltage Lock-out Threshold BOOT–SW Increasing VDR-HYS BOOT Under-voltage Lock-out Hysteresis BOOT–SW Decreasing 1.7 400 mV THERMAL SHUTDOWN TSD Thermal Shutdown Threshold 165 ˚C TSD-HYS Thermal Shutdown Hysteresis 25 ˚C 200 ˚C/W THERMAL RESISTANCE θJA Junction to Ambient MUA Package Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is intended to be functional, but specific performance is not guaranteed. For guaranteed specifications and the test conditions, see Electrical Characteristics. Note 2: The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin. Note 3: VCC provides self bias for the internal gate drive and control circuits. Device thermal limitations limit external loading. Note 4: Typical specifications represent the most likely parametric norm at 25˚C operation. 5 www.national.com LM3402/LM3402HV Electrical Characteristics VIN = 24V unless otherwise indicated. Typicals and limits appearing in plain type apply for TA = TJ = +25˚C. (Note 4) Limits appearing in boldface type apply over full Operating Temperature Range. Datasheet min/max specification limits are guaranteed by design, test, or statistical analysis. (Continued) LM3402/LM3402HV Typical Performance Characteristics VREF vs Temperature (VIN = 24V) VREF vs VIN, LM3402 (TA = 25˚C) 20192129 20192130 VREF vs VIN, LM3402HV (TA = 25˚C) Current Limit vs Temperature (VIN = 24V) 20192131 20192132 Current Limit vs VIN, LM3402 (TA = 25˚C) Current Limit vs VIN, LM3402HV (TA = 25˚C) 20192133 www.national.com 20192134 6 LM3402/LM3402HV Typical Performance Characteristics (Continued) TON vs VIN, RON = 100 kΩ (TA = 25˚C) TON vs VIN, (TA = 25˚C) 20192135 20192136 TON vs RON, LM3402 (TA = 25˚C) TON vs VIN, (TA = 25˚C) 20192137 20192144 VCC vs VIN (TA = 25˚C) TON vs RON, LM3402HV (TA = 25˚C) 20192138 20192139 7 www.national.com LM3402/LM3402HV Typical Performance Characteristics (Continued) VO-MAX vs fSW, LM3402 (TA = 25˚C) VO-MIN vs fSW, LM3402 (TA = 25˚C) 20192140 20192141 VO-MIN vs fSW, LM3402HV (TA = 25˚C) VO-MAX vs fSW, LM3402HV (TA = 25˚C) 20192142 www.national.com 20192143 8 LM3402/LM3402HV Block Diagram 20192103 resistor, RSNS, to ground. VSNS is fed back to the CS pin, where it is compared against a 200 mV reference, VREF. The on-comparator turns on the power MOSFET when VSNS falls below VREF. The power MOSFET conducts for a controlled on-time, tON, set by an external resistor, RON, and by the input voltage, VIN. On-time is governed by the following equation: Application Information THEORY OF OPERATION The LM3402 and LM3402HV are buck regulators with a wide input voltage range, low voltage reference, and a fast output enable/disable function. These features combine to make them ideal for use as a constant current source for LEDs with forward currents as high as 500 mA. The controlled on-time (COT) architecture is a combination of hysteretic mode control and a one-shot on-timer that varies inversely with input voltage. Hysteretic operation eliminates the need for smallsignal control loop compensation. When the converter runs in continuous conduction mode (CCM) the controlled on-time maintains a constant switching frequency over the range of input voltage. Fast transient response, PWM dimming, a low power shutdown mode, and simple output overvoltage protection round out the functions of the LM3402/02HV. At the conclusion of tON the power MOSFET turns off for a minimum off-time, tOFF-MIN, of 300 ns. Once tOFF-MIN is complete the CS comparator compares VSNS and VREF again, waiting to begin the next cycle. CONTROLLED ON-TIME OVERVIEW Figure 1 shows the feedback system used to control the current through an array of LEDs. A voltage signal, VSNS, is created as the LED current flows through the current setting 9 www.national.com LM3402/LM3402HV Application Information (Continued) 20192105 FIGURE 1. Comparator and One-Shot The LM3402/02HV regulators should be operated in continuous conduction mode (CCM), where inductor current stays positive throughout the switching cycle. During steady-state operationin the CCM, the converter maintains a constant switching frequency, which can be selected using the following equation: MAXIMUM OUTPUT VOLTAGE The 300 ns minimum off-time limits on the maximum duty cycle of the converter, DMAX, and in turn ,the maximum output voltage VO(MAX) is determined by the following equations: VF = forward voltage of each LED, n = number of LEDs in series The maximum number of LEDs, nMAX, that can be placed in a single series string is governed by VO(MAX) and the maximum forward voltage of the LEDs used, VF(MAX), using the expression: AVERAGE LED CURRENT ACCURACY The COT architecture regulates the valley of ∆VSNS, the AC portion of VSNS. To determine the average LED current (which is also the average inductor current) the valley inductor current is calculated using the following expression: At low switching frequency the maximum duty cycle and output voltage are higher, allowing the LM3402/02HV to regulate output voltages that are nearly equal to input voltage. The following equation relates switching frequency to maximum output voltage. In this equation tSNS represents the propagation delay of the CS comparator, and is approximately 220 ns. The average inductor/LED current is equal to IL-MIN plus one-half of the inductor current ripple, ∆iL: IF = IL = IL-MIN + ∆iL / 2 Detailed information for the calculation of ∆iL is given in the Design Considerations section. www.national.com 10 the DIM pin to the response of the internal power MOSFET. In general, fDIM should be at least one order of magnitude lower than the steady state switching frequency in order to prevent aliasing. (Continued) MINIMUM OUTPUT VOLTAGE The minimum recommended on-time for the LM3402/02HV is 300 ns. This lower limit for tON determines the minimum duty cycle and output voltage that can be regulated based on input voltage and switching frequency. The relationship is determined by the following equation: PEAK CURRENT LIMIT The current limit comparator of the LM3402/02HV will engage whenever the power MOSFET current (equal to the inductor current while the MOSFET is on) exceeds 735 mA (typical). The power MOSFET is disabled for a cool-down time that is 10x the steady-state on-time. At the conclusion of this cool-down time the system re-starts. If the current limit condition persists the cycle of cool-down time and restarting will continue, creating a low-power hiccup mode, minimizing thermal stress on the LM3402/02HV and the external circuit components. HIGH VOLTAGE BIAS REGULATOR The LM3402/02HV contains an internal linear regulator with a 7V output, connected between the VIN and the VCC pins. The VCC pin should be bypassed to the GND pin with a 0.1 µF ceramic capacitor connected as close as possible to the pins of the IC. VCC tracks VIN until VIN reaches 8.8V (typical) and then regulates at 7V as VIN increases. Operation begins when VCC crosses 5.25V. OVER-VOLTAGE/OVER-CURRENT COMPARATOR The CS pin includes an output over-voltage/over-current comparator that will disable the power MOSFET whenever VSNS exceeds 300 mV. This threshold provides a hard limit for the output current. Output current overshoot is limited to 300 mV / RSNS by this comparator during transients. The OVP/OCP comparator can also be used to prevent the output voltage from rising to VO(MAX) in the event of an output open-circuit. This is the most common failure mode for LEDs, due to breaking of the bond wires. In a current regulator an output open circuit causes VSNS to fall to zero, commanding maximum duty cycle. Figure 2 shows a method using a zener diode, Z1, and zener limiting resistor, RZ, to limit output voltage to the reverse breakdown voltage of Z1 plus 200 mV. The zener diode reverse breakdown voltage, VZ, must be greater than the maximum combined VF of all LEDs in the array. The maximum recommended value for RZ is 1 kΩ. As discussed in the Maximum Output Voltage section, there is a limit to how high VO can rise during an output opencircuit that is always less than VIN. If no output capacitor is used, the output stage of the LM3402/02HV is capable of withstanding VO(MAX) indefinitely, however the voltage at the output end of the inductor will oscillate and can go above VIN or below 0V. A small (typically 10 nF) capacitor across the LED array dampens this oscillation. For circuits that use an output capacitor, the system can still withstand VO(MAX) indefinitely as long as CO is rated to handle VIN. The high current paths are blocked in output open-circuit and the risk of thermal stress is minimal, hence the user may opt to allow the output voltage to rise in the case of an open-circuit LED failure. INTERNAL MOSFET AND DRIVER The LM3402/02HV features an internal power MOSFET as well as a floating driver connected from the SW pin to the BOOT pin. Both rise time and fall time are 20 ns each (typical) and the approximate gate charge is 3 nC. The high-side rail for the driver circuitry uses a bootstrap circuit consisting of an internal high-voltage diode and an external 10 nF capacitor, CB. VCC charges CB through the internal diode while the power MOSFET is off. When the MOSFET turns on, the internal diode reverse biases. This creates a floating supply equal to the VCC voltage minus the diode drop to drive the MOSFET when its source voltage is equal to VIN. FAST SHUTDOWN FOR PWM DIMMING The DIM pin of the LM3402/02HV is a TTL logic compatible input for low frequency PWM dimming of the LED. A logic low (below 0.8V) at DIM will disable the internal MOSFET and shut off the current flow to the LED array. While the DIM pin is in a logic low state the support circuitry (driver, bandgap, VCC) remains active in order to minimize the time needed to turn the LED array back on when the DIM pin sees a logic high (above 2.2V). A 75 µA (typical) pull-up current ensures that the LM3402/02HV is on when DIM pin is open circuited, eliminating the need for a pull-up resistor. Dimming frequency, fDIM, and duty cycle, DDIM, are limited by the LED current rise time and fall time and the delay from activation of 11 www.national.com LM3402/LM3402HV Application Information LM3402/LM3402HV Application Information (Continued) 20192112 FIGURE 2. Output Open Circuit Protection LOW POWER SHUTDOWN The LM3402/02HV can be switched to a low power state (IIN-SD = 90 µA) by grounding the RON pin with a signal-level MOSFET as shown in Figure 3. Low power MOSFETs like the 2N7000, 2N3904, or equivalent are recommended devices for putting the LM3402/02HV into low power shutdown. Logic gates can also be used to shut down the LM3402/ 02HV as long as the logic low voltage is below the over temperature minimum threshold of 0.3V. Noise filter circuitry on the RON pin can cause a few pulses with a longer on-time than normal after RON is grounded or released. In these cases the OVP/OCP comparator will ensure that the peak inductor or LED current does not exceed 300 mV / RSNS. 20192113 FIGURE 3. Low Power Shutdown exceeded. The threshold for thermal shutdown is 165˚C with a 25˚C hysteresis (both values typical). During thermal shutdown the MOSFET and driver are disabled. THERMAL SHUTDOWN Internal thermal shutdown circuitry is provided to protect the IC in the event that the maximum junction temperature is www.national.com 12 SWITCHING FREQUENCY Switching frequency is selected based on the tradeoffs between efficiency (better at low frequency), solution size/cost (smaller at high frequency), and the range of output voltage that can be regulated (wider at lower frequency.) Many applications place limits on switching frequency due to EMI sensitivity. The on-time of the LM3402/02HV can be programmed for switching frequencies ranging from the 10’s of kHz to over 1 MHz. The maximum switching frequency is limited only by the minimum on-time requirement. Figure 4 shows the equivalent impedances presented to the inductor current ripple when an output capacitor, CO, and its equivalent series resistance (ESR) are placed in parallel with the LED array. The entire inductor ripple current flows through RSNS to provide the required 25 mV of ripple voltage for proper operation of the CS comparator. LED RIPPLE CURRENT Selection of the ripple current, ∆iF, through the LED array is analogous to the selection of output ripple voltage in a standard voltage regulator. Where the output ripple in a voltage regulator is commonly ± 1% to ± 5% of the DC output voltage, LED manufacturers generally recommend values for ∆iF ranging from ± 5% to ± 20% of IF. Higher LED ripple current allows the use of smaller inductors, smaller output capacitors, or no output capacitors at all. The advantages of higher ripple current are reduction in the solution size and cost. Lower ripple current requires more output inductance, higher switching frequency, or additional output capacitance. The advantages of lower ripple current are a reduction in heating in the LED itself and greater range of the average LED current before the current limit of the LED or the driving circuitry is reached. BUCK CONVERTERS WITHOUT OUTPUT CAPACITORS The buck converter is unique among non-isolated topologies because of the direct connection of the inductor to the load during the entire switching cycle. By definition an inductor will control the rate of change of current that flows through it, and this control over current ripple forms the basis for component selection in both voltage regulators and current regulators. A current regulator such as the LED driver for which the LM3402/02HV was designed focuses on the control of the current through the load, not the voltage across it. A constant current regulator is free of load current transients, and has no need of output capacitance to supply the load and maintain output voltage. Referring to the Typical Application circuit on the front page of this datasheet, the inductor and LED can form a single series chain, sharing the same current. When no output capacitor is used, the same equations that govern inductor ripple current, ∆iL, also apply to the LED ripple current, ∆iF. For a controlled on-time converter such as LM3402/02HV the ripple current is described by the following expression: 20192115 FIGURE 4. LED and CO Ripple Current To calculate the respective ripple currents the LED array is represented as a dynamic resistance, rD. LED dynamic resistance is not always specified on the manufacturer’s datasheet, but it can be calculated as the inverse slope of the LED’s VF vs. IF curve. Note that dividing VF by IF will give an incorrect value that is 5x to 10x too high. Total dynamic resistance for a string of n LEDs connected in series can be calculated as the rD of one device multiplied by n. Inductor ripple current is still calculated with the expression from Buck Regulators without Output Capacitors. The following equations can then be used to estimate ∆iF when using a parallel capacitor: A minimum ripple voltage of 25 mV is recommended at the CS pin to provide good signal-to-noise ratio (SNR). The CS pin ripple voltage, ∆VSNS, is described by the following: The calculation for ZC assumes that the shape of the inductor ripple current is approximately sinusoidal. Small values of CO that do not significantly reduce ∆iF can also be used to control EMI generated by the switching action of the LM3402/02HV. EMI reduction becomes more important as the length of the connections between the LED and the rest of the circuit increase. ∆VSNS = ∆iF x RSNS BUCK CONVERTERS WITH OUTPUT CAPACITORS A capacitor placed in parallel with the LED or array of LEDs can be used to reduce the LED current ripple while keeping the same average current through both the inductor and the LED array. This technique is demonstrated in Design Ex13 www.national.com LM3402/LM3402HV ample 1. With this topology the output inductance can be lowered, making the magnetics smaller and less expensive. Alternatively, the circuit could be run at lower frequency but keep the same inductor value, improving the efficiency and expanding the range of output voltage that can be regulated. Both the peak current limit and the OVP/OCP comparator still monitor peak inductor current, placing a limit on how large ∆iL can be even if ∆iF is made very small. A parallel output capacitor is also useful in applications where the inductor or input voltage tolerance is poor. Adding a capacitor that reduces ∆iF to well below the target provides headroom for changes in inductance or VIN that might otherwise push the peak LED ripple current too high. Design Considerations LM3402/LM3402HV Design Considerations RECIRCULATING DIODE The LM3402/02HV is a non-synchronous buck regulator that requires a recirculating diode D1 (see the Typical Application circuit) to carrying the inductor current during the MOSFET off-time. The most efficient choice for D1 is a Schottky diode due to low forward drop and near-zero reverse recovery time. D1 must be rated to handle the maximum input voltage plus any switching node ringing when the MOSFET is on. In practice all switching converters have some ringing at the switching node due to the diode parasitic capacitance and the lead inductance. D1 must also be rated to handle the average current, ID, calculated as: (Continued) INPUT CAPACITORS Input capacitors at the VIN pin of the LM3402/02HV are selected using requirements for minimum capacitance and rms ripple current. The input capacitors supply pulses of current approximately equal to IF while the power MOSFET is on, and are charged up by the input voltage while the power MOSFET is off. Switching converters such as the LM3402/02HV have a negative input impedance due to the decrease in input current as input voltage increases. This inverse proportionality of input current to input voltage can cause oscillations (sometimes called ‘power supply interaction’) if the magnitude of the negative input impedance is greater the the input filter impedance. Minimum capacitance can be selected by comparing the input impedance to the converter’s negative resistance; however this requires accurate calculation of the input voltage source inductance and resistance, quantities which can be difficult to determine. An alternative method to select the minimum input capacitance, CIN(MIN), is to select the maximum voltage ripple which can be tolerated. This value,∆vIN(MAX), is equal to the change in voltage across CIN during the converter on-time, when CIN supplies the load current. CIN(MIN) can be selected with the following: ID = (1 – D) x IF This calculation should be done at the maximum expected input voltage. The overall converter efficiency becomes more dependent on the selection of D1 at low duty cycles, where the recirculating diode carries the load current for an increasing percentage of the time. This power dissipation can be calculated by checking the typical diode forward voltage, VD, from the I-V curve on the product datasheet and then multiplying it by ID. Diode datasheets will also provide a typical junction-to-ambient thermal resistance, θJA, which can be used to estimate the operating die temperature of the Schottky. Multiplying the power dissipation (PD = ID x VD) by θJA gives the temperature rise. The diode case size can then be selected to maintain the Schottky diode temperature below the operational maximum. A good starting point for selection of CIN is to use an input voltage ripple of 5% to 10% of VIN. A minimum input capacitance of 2x the CIN(MIN) value is recommended for all LM3402/02HV circuits. To determine the rms current rating, the following formula can be used: Design Example 1: LM3402 The first example circuit will guide the user through component selection for an architectural accent lighting application. A regulated DC voltage input of 24V ± 10% will power a single 1W white LED at a forward current of 350 mA ± 5%. The typical forward voltage of a 1W InGaN LED is 3.5V, hence the estimated average output voltage will be 3.7V. The objective of this application is to place the complete current regulator and LED in the compact space formerly occupied by an MR16 halogen light bulb. (The LED will be on a separate metal-core PCB.) Switching frequency will be as fast as the 300 ns tON limit allows, with the emphasis on space savings over efficiency. Efficiency cannot be ignored, however, as the confined space with little air-flow requires a maximum temperature rise of 40˚C in each circuit component. A complete bill of materials can be found in Table 1 at the end of this datasheet. Ceramic capacitors are the best choice for the input to the LM3402/02HV due to their high ripple current rating, low ESR, low cost, and small size compared to other types. When selecting a ceramic capacitor, special attention must be paid to the operating conditions of the application. Ceramic capacitors can lose one-half or more of their capacitance at their rated DC voltage bias and also lose capacitance with extremes in temperature. A DC voltage rating equal to twice the expected maximum input voltage is recommended. In addition, the minimum quality dielectric which is suitable for switching power supply inputs is X5R, while X7R or better is preferred. www.national.com 14 LM3402/LM3402HV Design Example 1: LM3402 (Continued) 20192119 FIGURE 5. Schematic for Design Example 1 RON and tON To select RON the expression relating tON to input voltage from the Controlled On-time Overview section can be rewritten as: LMIN = [(26.4 – 3.7) x 300 x 10-9] / (0.6 x 0.35) = 32.4 µH The closest standard inductor value is 33 µH. Off-the-shelf inductors rated at 33 µH are available from many magnetics manufacturers. Inductor datasheets should contain three specifications which are used to select the inductor. The first of these is the average current rating, which for a buck regulator is equal to the average load current, or IF. The average current rating is given by a specified temperature rise in the inductor, normally 40˚C. For this example, the average current rating should be greater than 350 mA to ensure that heat from the inductor does not reduce the lifetime of the LED or cause the LM3402 to enter thermal shutdown. The second specification is the tolerance of the inductance itself, typically ± 10% to ± 30% of the rated inductance. In this example an inductor with a tolerance of ± 20% will be used. With this tolerance the typical, minimum, and maximum inductor current ripples can be calculated: Minimum on-time occurs at the maximum VIN, which is 24V x 110% = 26.4V. RON is therefore calculated as: RON = (300 x 10-9 x 26.4) / 1.34 x 10-10 = 59105 Ω The closest 1% tolerance resistor is 59.0 kΩ. The switching frequency of the circuit can then be found using the equation relating RON to fSW: fSW = 3.7 / (59000 x 1.34 x 10-10) = 468 kHz USING AN OUTPUT CAPACITOR The inductor will be the largest component used in this design. Because the application does not require any PWM dimming, an output capacitor can be used to greatly reduce the inductance needed without worry of slowing the potential PWM dimming frequency. The total solution size will be reduced by using an output capacitor and small inductor as opposed to one large inductor. ∆iL(TYP) = [(26.4 – 3.7) x 300 x 10-9] / 33 x 10-6 = 206 mAP-P ∆iL(MIN) = [(26.4 – 3.7) x 300 x 10-9] / 39.6 x 10-6 = 172 mAP-P OUTPUT INDUCTOR Knowing that an output capacitor will be used, the inductor can be selected for a larger current ripple. The desired maximum value for ∆iL is ± 30%, or 0.6 x 350 mA = 210 mAP-P. Minimum inductance is selected at the maximum input voltage. Re-arranging the equation for current ripple selection yields the following: ∆iL(MAX) = [(26.4 – 3.7) x 300 x 10-9] / 26.4 x 10-6 = 258 mAP-P The third specification for an inductor is the peak current rating, normally given as the point at which the inductance drops off by a given percentage due to saturation of the core. The worst-case peak current occurs at maximum input voltage and at minimum inductance, and can be determined with the equation from the Design Considerations section: 15 www.national.com LM3402/LM3402HV Design Example 1: LM3402 rD of 1.0Ω at 350 mA. The required capacitor impedance to reduce the worst-case inductor ripple current of 258 mAP-P is therefore: (Continued) ZC = [0.035 / (0.258 - 0.035] x 1.0 = 0.157Ω A ceramic capacitor will be used and the required capacitance is selected based on the impedance at 468 kHz: IL(PEAK) = 0.35 + 0.258 / 2 = 479 mA For this example the peak current rating of the inductor should be greater than 479 mA. In the case of a short circuit across the LED array, the LM3402 will continue to deliver rated current through the short but will reduce the output voltage to equal the CS pin voltage of 200 mV. Worst-case peak current in this condition is equal to: CO = 1/(2 x π x 0.157 x 4.68 x 105) = 2.18 µF This calculation assumes that impedance due to the equivalent series resistance (ESR) and equivalent series inductance (ESL) of CO is negligible. The closest 10% tolerance capacitor value is 2.2 µF. The capacitor used should be rated to 10V or more and have an X7R dielectric. Several manufacturers produce ceramic capacitors with these specifications in the 0805 case size. A typical value for ESR of 1 mΩ can be read from the curve of impedance vs. frequency in the product datasheet. ∆iL(LED-SHORT) = [(26.4 – 0.2) x 300 x 10-9] / 26.4 x 10-6 = 298 mAP-P IL(PEAK) = 0.35 + 0.149 = 499 mA In the case of a short at the switch node, the output, or from the CS pin to ground the short circuit current limit will engage at a typical peak current of 735 mA. In order to prevent inductor saturation during these short circuits the inductor’s peak current rating must be above 735 mA. The device selected is an off-the-shelf inductor rated 33 µH ± 20% with a DCR of 96 mΩ and a peak current rating of 0.82A. The physical dimensions of this inductor are 7.0 x 7.0 x 4.5 mm. INPUT CAPACITOR Following the calculations from the Input Capacitor section, ∆vIN(MAX) will be 1%P-P = 240 mV. The minimum required capacitance is: CIN(MIN) = (0.35 x 300 x 10-9) / 0.24 = 438 nF In expectation that more capacitance will be needed to prevent power supply interaction a 1.0 µF ceramic capacitor rated to 50V with X7R dielectric in a 1206 case size will be used. From the Design Considerations section, input rms current is: RSNS The current sensing resistor value can be determined by re-arranging the expression for average LED current from the LED Current Accuracy section: IIN-RMS = 0.35 x Sqrt(0.154 x 0.846) = 126 mA Ripple current ratings for 1206 size ceramic capacitors are typically higher than 1A, more than enough for this design. RSNS = 0.74Ω, tSNS = 220 ns RECIRCULATING DIODE The first parameter for D1 which must be determined is the reverse voltage rating. Schottky diodes are available at reverse ratings of 30V and 40V, often in the same package, with the same forward current rating. To account for ringing a 40V Schottky will be used. The next parameters to be determined are the forward current rating and case size. In this example the low duty cycle (D = 3.7 / 24 = 15%) requires the recirculating diode D1 to carry the load current much longer than the internal power MOSFET of the LM3402. The estimated average diode current is: Sub-1Ω resistors are available in both 1% and 5% tolerance. A 1%, 0.75Ω resistor will give the best accuracy of the average LED current. To determine the resistor size the power dissipation can be calculated as: PSNS = (IF)2 x RSNS PSNS = 0.352 x 0.75 = 92 mW Standard 0805 size resistors are rated to 125 mW and will be suitable for this application. To select the proper output capacitor the equation from Buck Regulators with Output Capacitors is re-arranged to yield the following: ID = 0.35 x 0.85 = 298 mA Schottky diodes are available at forward current ratings of 0.5A, however the current rating often assumes a 25˚C ambient temperature and does not take into account the application restrictions on temperature rise. A diode rated for higher current may be needed to keep the temperature rise below 40˚C.To determine the proper case size, the dissipation and temperature rise in D1 can be calculated as shown in the Design Considerations section. VD for a small case The target tolerance for LED ripple current is ± 5% or 10%P-P = 35 mAP-P, and the LED datasheet gives a typical value for www.national.com 16 Recirculating diode loss, PD = 119 mW Current Sense Resistor Loss, PSNS = 92 mW Electrical efficiency, η = PO / (PO + Sum of all loss terms) = 1.295 / (1.295 + 0.377) = 77% (Continued) size such as SOD-123 in a 40V, 0.5A Schottky diode at 350 mA is approximately 0.4V and the θJA is 206˚C/W. Power dissipation and temperature rise can be calculated as: DIE TEMPERATURE TLM3402 = (PC + PG + PS) x θJA TLM3402 = (0.028 + 0.05 + 0.078) x 200 = 31˚C PD = 0.298 x 0.4 = 119 mW TRISE = 0.119 x 206 = 24.5˚C Design Example 2: LM3402HV According to these calculations the SOD-123 diode will meet the requirements. Heating and dissipation are among the factors most difficult to predict in converter design. If possible, a footprint should be used that is capable of accepting both SOD-123 and a larger case size, such as SMA. A larger diode with a higher forward current rating will generally have a lower forward voltage, reducing dissipation, as well as having a lower θJA, reducing temperature rise. The second example application is an RGB backlight for a flat screen monitor. A separate boost regulator provides a 60V ± 5% DC input rail that feeds three LM3402HV current regulators to drive one series array each of red, green, and blue 1W LEDs. The target for average LED current is 350 mA ± 5% in each string. The monitor will adjust the color temperature dynamically, requiring fast PWM dimming of each string with external, parallel MOSFETs. 1W green and blue InGaN LEDs have a typical forward voltage of 3.5V, however red LEDs use AlInGaP technology with a typical forward voltage of 2.9V. In order to match color properly the design requires 14 green LEDs, twice as many as needed for the red and blue LEDs. This example will follow the design for the green LED array, providing the necessary information to repeat the exercise for the blue and red LED arrays. The circuit schematic for Design Example 2 is the same as the Typical Application on the front page. The bill of materials (green array only) can be found in Table 2 at the end of this datasheet. CB and CF The bootstrap capacitor CB should always be a 10 nF ceramic capacitor with X7R dielectric. A 25V rating is appropriate for all application circuits. The linear regulator filter capacitor CF should always be a 100 nF ceramic capacitor, also with X7R dielectric and a 25V rating. EFFICIENCY To estimate the electrical efficiency of this example the power dissipation in each current carrying element can be calculated and summed. This term should not be confused with the optical efficacy of the circuit, which depends upon the LEDs themselves. Total output power, PO, is calculated as: OUTPUT VOLTAGE Green Array: VO(G) = 14 x 3.5 + 0.2 = 49.2V Blue Array: VO(B) = 7 x 3.5 + 0.2 = 24.7V Red Array: VO(R) = 7 x 2.9 + 0.2 = 20.5V PO = IF x VO = 0.35 x 3.7 = 1.295W RON and tON A compromise in switching frequency is needed in this application to balance the requirements of magnetics size and efficiency. The high duty cycle translates into large conduction losses and high temperature rise in the IC. For best response to a PWM dimming signal this circuit will not use an output capacitor; hence a moderate switching frequency of 300 kHz will keep the inductance from becoming so large that a custom-wound inductor is needed. This design will use only surface mount components, and the selection of off-theshelf SMT inductors for switching regulators is poor at 1000 µH and above. RON is selected from the equation for switching frequency as follows: Conduction loss, PC, in the internal MOSFET: PC = (IF2 x RDSON) x D = (0.352 x 1.5) x 0.154 = 28 mW Gate charging and VCC loss, PG, in the gate drive and linear regulator: PG = (IIN-OP + fSW x QG) x VIN PG = (600 x 10-6 + 468000 x 3 x 10-9) x 24 = 48 mW Switching loss, PS, in the internal MOSFET: PS = 0.5 x VIN x IF x (tR + tF) x fSW PS = 0.5 x 24 x 0.35 x (40 x 10-9) x 468000 = 78 mW AC rms current loss, PCIN, in the input capacitor: RON = 49.2 / (1.34 x 10-10 x 3 x 105) = 1224 kΩ PCIN = IIN(rms)2 x ESR = (0.126)2 x 0.006 = 0.1 mW (negligible) The closest 1% tolerance resistor is 1.21 MΩ. The switching frequency and on-time of the circuit can then be found using the equations relating RON and tON to fSW: DCR loss, PL, in the inductor PL = IF2 x DCR = 0.352 x 0.096 = 11.8 mW fSW = 49.2 / (1210000 x 1.34 x 10-10) = 303 kHz 17 www.national.com LM3402/LM3402HV Design Example 1: LM3402 LM3402/LM3402HV Design Example 2: LM3402HV short but will reduce the output voltage to equal the CS pin voltage of 200 mV. Worst-case peak current in this condition would be equal to: (Continued) tON = (1.34 x 10-10 x 1210000) / 60 = 2.7 µs ∆iF(LED-SHORT) = [(63 – 0.2) x 2.7 x 10-6] / 544 x 10-6 = 314 mAP-P IF(PEAK) = 0.35 + 0.156 = 506 mA USING AN OUTPUT CAPACITOR This application is dominated by the need for fast PWM dimming, requiring a circuit without any output capacitance. In the case of a short at the switch node, the output, or from the CS pin to ground the short circuit current limit will engage at a typical peak current of 735 mA. In order to prevent inductor saturation during these fault conditions the inductor’s peak current rating must be above 735 mA. A 680 µH off-the shelf inductor rated to 1.2A (peak) and 0.72A (average) with a DCR of 1.1Ω will be used for the green LED array. RSNS A preliminary value for RSNS was determined in selecting ∆iL. This value should be re-evaluated based on the calculations for ∆iF: OUTPUT INDUCTOR In this example the ripple current through the LED array and the inductor are equal. Inductance is selected to give the smallest ripple current possible while still providing enough ∆vSNS signal for the CS comparator to operate correctly. Designing to a desired ∆vSNS of 25 mV and assuming that the average inductor current will equal the desired average LED current of 350 mA yields the target current ripple in the inductor and LEDs: ∆iF = ∆iL = ∆vSNS / RSNS, RSNS = VSNS / IF ∆iF = 0.025 / 0.57 = 43.8 mA With the target ripple current determined the inductance can be chosen: Sub-1Ω resistors are available in both 1% and 5% tolerance. A 1%, 0.56Ω device is the closest value, and a 0.125W, 0805 size device will handle the power dissipation of 69 mW. With the resistance selected, the average value of LED current is re-calculated to ensure that current is within the ± 5% tolerance requirement. From the expression for LED current accuracy: LMIN = [(60 – 49.2) x 2.7 x 10-6] / (0.044) = 663 µH IF = 0.19 / 0.56 + 0.043 / 2 = 361 mA, 3% above 350 mA The closest standard inductor value is 680 µH. As with the previous example, the average current rating should be greater than 350 mA. Separation between the LM3402HV drivers and the LED arrays mean that heat from the inductor will not threaten the lifetime of the LEDs, but an overheated inductor could still cause the LM3402HV to enter thermal shutdown. The inductance itself of the standard part chosen is ± 20%. With this tolerance the typical, minimum, and maximum inductor current ripples can be calculated: INPUT CAPACITOR Following the calculations from the Input Capacitor section, ∆vIN(MAX) will be 1%P-P = 600 mV. The minimum required capacitance is: CIN(MIN) = (0.35 x 2.7 x 10-6) / 0.6 = 1.6 µF In expectation that more capacitance will be needed to prevent power supply interaction a 2.2 µF ceramic capacitor rated to 100V with X7R dielectric in an 1812 case size will be used. From the Design Considerations section, input rms current is: ∆iF(TYP) = [(60 - 49.2) x 2.7 x 10-6] / 680 x 10-6 = 43 mAP-P ∆iF(MIN) = [(60 - 49.2) x 2.7 x 10-6] / 816 x 10-6 = 36 mAP-P IIN-RMS = 0.35 x Sqrt(0.82 x 0.18) = 134 mA ∆iF(MAX) = [(60 - 49.2) x 2.7 x 10-6] / 544 x 10-6 = 54 mAP-P Ripple current ratings for 1812 size ceramic capacitors are typically higher than 2A, more than enough for this design. RECIRCULATING DIODE The input voltage of 60V ± 5% requires Schottky diodes with a reverse voltage rating greater than 60V. Some manufacturers provide Schottky diodes with ratings of 70, 80 or 90V; however the next highest standard voltage rating is 100V. Selecting a 100V rated diode provides a large safety margin for the ringing of the switch node and also makes crossreferencing of diodes from different vendors easier. The peak LED/inductor current is then estimated: IL(PEAK) = IL + [∆iL(MAX)] / 2 IL(PEAK) = 0.35 + 0.027 = 377 mA In the case of a short circuit across the LED array, the LM3402HV will continue to deliver rated current through the www.national.com 18 PG = (IIN-OP + fSW x QG) x VIN PG = (600 x 10-6 + 3 x 105 x 3 x 10-9) x 60 = 90 mW (Continued) The next parameters to be determined are the forward current rating and case size. In this example the high duty cycle (D = 49.2 / 60 = 82%) places less thermals stress on D1 and more on the internal power MOSFET of the LM3402. The estimated average diode current is: Switching loss, PS, in the internal MOSFET: PS = 0.5 x VIN x IF x (tR + tF) x fSW PS = 0.5 x 60 x 0.361 x 40 x 10-9 x 3 x 105 = 130 mW ID = 0.361 x 0.18 = 65 mA AC rms current loss, PCIN, in the input capacitor: A Schottky with a forward current rating of 0.5A would be adequate, however at 100V the majority of diodes have a minimum forward current rating of 1A. To determine the proper case size, the dissipation and temperature rise in D1 can be calculated as shown in the Design Considerations section. VD for a small case size such as SOD-123F in a 100V, 1A Schottky diode at 350 mA is approximately 0.65V and the θJA is 88˚C/W. Power dissipation and temperature rise can be calculated as: PCIN = IIN(rms)2 x ESR = (0.134)2 x 0.006 = 0.1 mW (negligible) DCR loss, PL, in the inductor PL = IF2 x DCR = 0.352 x 1.1 = 135 mW Recirculating diode loss, PD = 42 mW Current Sense Resistor Loss, PSNS = 69 mW Electrical efficiency, η = PO / (PO + Sum of all loss terms) = 17.76 / (17.76 + 0.62) = 96% PD = 0.065 x 0.65 = 42 mW TRISE = 0.042 x 88 = 4˚C Temperature Rise in the LM3402HV IC is calculated as: CB AND CF The bootstrap capacitor CB should always be a 10 nF ceramic capacitor with X7R dielectric. A 25V rating is appropriate for all application circuits. The linear regulator filter capacitor CF should always be a 100 nF ceramic capacitor, also with X7R dielectric and a 25V rating. TLM3402 = (PC + PG + PS) x θJA = (0.16 + 0.084 + 0.13) x 200 = 74.8˚C Layout Considerations The performance of any switching converter depends as much upon the layout of the PCB as the component selection. The following guidelines will help the user design a circuit with maximum rejection of outside EMI and minimum generation of unwanted EMI. EFFICIENCY To estimate the electrical efficiency of this example the power dissipation in each current carrying element can be calculated and summed. Electrical efficiency, η, should not be confused with the optical efficacy of the circuit, which depends upon the LEDs themselves. Total output power, PO, is calculated as: COMPACT LAYOUT Parasitic inductance can be reduced by keeping the power path components close together and keeping the area of the loops that high currents travel small. Short, thick traces or copper pours (shapes) are best. In particular, the switch node (where L1, D1, and the SW pin connect) should be just large enough to connect all three components without excessive heating from the current it carries. The LM3402/ 02HV operates in two distinct cycles whose high current paths are shown in Figure 6: PO = IF x VO = 0.361 x 49.2 = 17.76W Conduction loss, PC, in the internal MOSFET: PC = (IF2 x RDSON) x D = (0.3612 x 1.5) x 0.82 = 160 mW Gate charging and VCC loss, PG, in the gate drive and linear regulator: 19 www.national.com LM3402/LM3402HV Design Example 2: LM3402HV LM3402/LM3402HV Layout Considerations (Continued) 20192128 FIGURE 6. Buck Converter Current Loops The dark grey, inner loop represents the high current path during the MOSFET on-time. The light grey, outer loop represents the high current path during the off-time. capacitor to connect the component side shapes to the ground plane. A second pulsating current loop that is often ignored is the gate drive loop formed by the SW and BOOT pins and capacitor CB. To minimize this loop at the EMI it generates, keep CB close to the SW and BOOT pins. GROUND PLANE AND SHAPE ROUTING The diagram of Figure 6 is also useful for analyzing the flow of continuous current vs. the flow of pulsating currents. The circuit paths with current flow during both the on-time and off-time are considered to be continuous current, while those that carry current during the on-time or off-time only are pulsating currents. Preference in routing should be given to the pulsating current paths, as these are the portions of the circuit most likely to emit EMI. The ground plane of a PCB is a conductor and return path, and it is susceptible to noise injection just as any other circuit path. The continuous current paths on the ground net can be routed on the system ground plane with less risk of injecting noise into other circuits. The path between the input source and the input capacitor and the path between the recirculating diode and the LEDs/current sense resistor are examples of continuous current paths. In contrast, the path between the recirculating diode and the input capacitor carries a large pulsating current. This path should be routed with a short, thick shape, preferably on the component side of the PCB. Multiple vias in parallel should be used right at the pad of the input www.national.com CURRENT SENSING The CS pin is a high-impedance input, and the loop created by RSNS, RZ (if used), the CS pin and ground should be made as small as possible to maximize noise rejection. RSNS should therefore be placed as close as possible to the CS and GND pins of the IC. REMOTE LED ARRAYS In some applications the LED or LED array can be far away (several inches or more) from the LM3402/02HV, or on a separate PCB connected by a wiring harness. When an output capacitor is used and the LED array is large or separated from the rest of the converter, the output capacitor should be placed close to the LEDs to reduce the effects of parasitic inductance on the AC impedance of the capacitor. The current sense resistor should remain on the same PCB, close to the LM3402/02HV. 20 (Continued) TABLE 1. BOM for Design Example 1 ID Part Number Type Size Parameters Qty Vendor U1 LM3402 LED Driver MSOP-8 40V, 0.5A 1 NSC L1 SLF7045T-330MR82 Inductor 7.0x7.0 x4.5mm 33µH, 0.82A, 96mΩ 1 TDK D1 CMHSH5-4 Schottky Diode SOD-123 40V, 0.5A 1 Central Semi Cf VJ0805Y104KXXAT Capacitor 0805 100nF 10% 1 Vishay Cb VJ0805Y103KXXAT Capacitor 0805 10nF 10% 1 Vishay Cin C3216X7R1H105M Capacitor 1206 1µF 50V 1 TDK Co C2012X7R1A225M Capacitor 0805 2.2 µF 10V 1 TDK Rsns ERJ6BQFR75V Resistor 0805 0.75Ω 1% 1 Panasonic Ron CRCW08055902F Resistor 0805 59.0 kΩ 1% 1 Vishay TABLE 2. BOM for Design Example 2 ID Part Number Type Size Parameters Qty U1 LM3402HV LED Driver MSOP-8 75V, 0.5A 1 NSC L1 DO5022P-684 Inductor 18.5x15.2 x7.1mm 680µH, 1.2A, 1.1Ω 1 Coilcraft D1 CMMSH1-100 Schottky Diode SOD-123F 100V, 1A 1 Central Semi Cf VJ0805Y104KXXAT Capacitor 0805 100nF 10% 1 Vishay Cb VJ0805Y103KXXAT Capacitor 0805 10nF 10% 1 Vishay Cin C4532X7R2A225M Capacitor 1812 2.2µF 100V 1 TDK Rsns ERJ6BQFR56V Resistor 0805 0.56Ω 1% 1 Panasonic Ron CRCW08051214F Resistor 0805 1.21MΩ 1% 1 Vishay 21 Vendor www.national.com LM3402/LM3402HV Layout Considerations LM3402/LM3402HV 0.5A Constant Current Buck Regulator for Driving High Power LEDs Physical Dimensions inches (millimeters) unless otherwise noted 8-Lead MSOP Package NS Package Number MUA08A National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves the right at any time without notice to change said circuitry and specifications. For the most current product information visit us at www.national.com. LIFE SUPPORT POLICY NATIONAL’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT AND GENERAL COUNSEL OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein: 1. Life support devices or systems are devices or systems which, (a) are intended for surgical implant into the body, or (b) support or sustain life, and whose failure to perform when properly used in accordance with instructions for use provided in the labeling, can be reasonably expected to result in a significant injury to the user. 2. A critical component is any component of a life support device or system whose failure to perform can be reasonably expected to cause the failure of the life support device or system, or to affect its safety or effectiveness. BANNED SUBSTANCE COMPLIANCE National Semiconductor follows the provisions of the Product Stewardship Guide for Customers (CSP-9-111C2) and Banned Substances and Materials of Interest Specification (CSP-9-111S2) for regulatory environmental compliance. Details may be found at: www.national.com/quality/green. Lead free products are RoHS compliant. National Semiconductor Americas Customer Support Center Email: [email protected] Tel: 1-800-272-9959 www.national.com National Semiconductor Europe Customer Support Center Fax: +49 (0) 180-530 85 86 Email: [email protected] Deutsch Tel: +49 (0) 69 9508 6208 English Tel: +44 (0) 870 24 0 2171 Français Tel: +33 (0) 1 41 91 8790 National Semiconductor Asia Pacific Customer Support Center Email: [email protected] National Semiconductor Japan Customer Support Center Fax: 81-3-5639-7507 Email: [email protected] Tel: 81-3-5639-7560