AOZ3011PI EZBuck™ 3 A Synchronous Buck Regulator General Description Features The AOZ3011PI is a high efficiency, easy to use, 3 A synchronous buck regulator. The AOZ3011PI operates from 4.5 V to 18 V input voltage range, and provides up to 3 A of continuous output current with an output voltage adjustable down to 0.8 V. 4.5 V to 18 V operating input voltage range The AOZ3011PI comes in an exposed pad SO-8 package and is rated over a -40 °C to +85 °C operating ambient temperature range. Output voltage adjustable to 0.8 V 70 mΩ internal high-side switch and 40 mΩ internal low-side switch (at 12 V) Up to 95 % efficiency External soft start 3 A continuous output current 500 kHz PWM operation Cycle-by-cycle current limit Pre-bias start-up Short-circuit protection Thermal shutdown Exposed pad SO-8 package Applications Point of load DC/DC converters LCD TV Set top boxes DVD and Blu-ray players/recorders Cable modems Typical Application VIN C1 10µF CSS VIN SS L1 4.7µH EN AOZ3011PI R1 COMP RC CC VOUT LX C2, C3 22µF FB AGND PGND R2 Figure 1. 3.3 V, 3 A Synchronous Buck Regulator, Fs = 500 kHz Rev. 0.6 August 2012 www.aosmd.com Page 1 of 14 AOZ3011PI Ordering Information Part Number Ambient Temperature Range Package Environmental AOZ3011PI -40 °C to +85 °C EPAD SO-8 Green Product AOS Green Products use reduced levels of Halogens, and are also RoHS compliant. Please visit www.aosmd.com/media/AOSGreenPolicy.pdf for additional information. Pin Configuration PGND 1 VIN 2 AGND 3 FB 4 PAD (LX) 8 LX 7 SS 6 EN 5 COMP Exposed Pad SO-8 (Top View) Pin Description Pin Number Pin Name Pin Function 1 PGND 2 VIN Supply voltage input. When VIN rises above the UVLO threshold and EN is logic high, the device starts up. 3 AGND Analog ground. AGND is the reference point for controller section. AGND needs to be electrically connected to PGND. 4 FB 5 COMP 6 EN Power ground. PGND needs to be electrically connected to AGND. Feedback input. The FB pin is used to set the output voltage via a resistive voltage divider between the output and AGND. External loop compensation pin. Connect a RC network between COMP and AGND to compensate the control loop. Enable pin. Pull EN to logic high to enable the device. Pull EN to logic low to disable the device. If on/off control in not needed, connect EN to VIN and do not leave it open. 7 SS 8 LX (Sense) Switching node. This pin has to be externally connected to exposed pad LX through PCB. Exposed pad LX Switching node. LX is the drain of the internal PFET. LX is used as the thermal pad of the power stage. Rev. 0.6 August 2012 Soft-start pin. 5 µA current charging current. www.aosmd.com Page 2 of 14 AOZ3011PI Block Diagram VIN UVLO & POR EN Internal +5V 5V LDO Regulator OTP + ISen Reference & Bias Softstart – Q1 ILimit SS SS 5µA 0.8V FB + + EAmp – – PWM Comp PWM Control Logic + Level Shifter + FET Driver LX Q2 COMP Over-Voltage Protection Comparator 500kHz Oscillator + 0.96V – AGND PGND Absolute Maximum Ratings Recommended Operating Conditions Exceeding the Absolute Maximum Ratings may damage the device. The device is not guaranteed to operate beyond the Maximum Recommended Operating Conditions. Parameter Supply Voltage (VIN) LX to AGND LX to AGND (20 ns) EN to AGND FB, SS, COMP to AGND PGND to AGND Rating Parameter 20 V -0.7 V to VIN +0.3 V -5 V to 22 V -0.3 V to VIN +0.3 V -0.3 V to 6.0 V 4.5 V to 18 V Output Voltage Range 0.8 V to 0.85*VIN Ambient Temperature (TA) -40 °C to +85 °C Package Thermal Resistance Exposed Pad SO-8 (JA) 50 °C/W -0.3 V to +0.3 V Junction Temperature (TJ) +150 °C Storage Temperature (TS) -65 °C to +150 °C ESD Rating(1) Supply Voltage (VIN) Rating 2.0 kV Note: 1. Devices are inherently ESD sensitive, handling precautions are required. Human body model rating: 1.5 kΩ in series with 100 pF. Rev. 0.6 August 2012 www.aosmd.com Page 3 of 14 AOZ3011PI Electrical Characteristics TA = 25 °C, VIN = VEN = 12 V, VOUT = 3.3 V unless otherwise specified(2) Symbol VIN VUVLO IIN Parameter Conditions Supply Voltage Input Under-Voltage Lockout Threshold Supply Current (Quiescent) Typ. 4.5 VIN Rising 4.1 VIN Falling 3.7 IOUT = 0, VFB = 1.2 V, VEN > 2 V 1.6 IOFF Shutdown Supply Current VEN = 0 V VFB Feedback Voltage TA = 25 °C 0.788 Max. Units 18 V V 2.5 mA 1 10 µA 0.8 0.812 V Load Regulation 0.5 % Line Regulation 1 % IFB Feedback Voltage Input Current VEN EN Input Threshold 200 Off Threshold On Threshold VHYS Min. 0.6 2 EN Input Hysteresis 200 EN Leakage Current SS Time V mV 1 CSS = 16 nF nA 2 µA ms MODULATOR Frequency 400 DMAX Maximum Duty Cycle 85 TMIN Controllable Minimum On Time fO 500 600 kHz 150 ns % Current Sense Transconductance 8 A/ V Error Amplifier Transconductance 200 µA / V 4.5 A PROTECTION ILIM VOVP Current Limit Over-Voltage Protection Over-Temperature Shutdown Limit 3.5 Off Threshold 960 On Threshold 800 TJ Rising 150 TJ Falling 100 VIN = 12 V 70 VIN = 5 V 110 VIN = 12 V 40 VIN = 5 V 50 mV °C OUTPUT STAGE High-Side Switch On-Resistance Low-Side Switch On-Resistance mΩ mΩ Note: 2. Specification in BOLD indicate an ambient temperature range of -40 °C to +85 °C. These specifications are guaranteed by design. Rev. 0.6 August 2012 www.aosmd.com Page 4 of 14 AOZ3011PI Typical Performance Characteristics Circuit of Figure 1. TA = 25 °C, VIN = VEN = 12 V, VOUT = 3.3 V unless otherwise specified. Full Load Operation Light Load Operation Vin ripple 0.2V/div Vin ripple 0.5V/div Vo ripple 0.2V/div Vo ripple 0.2V/div VLX 10V/div VLX 10V/div IL 2A/div IL 2A/div 2µs/div 2µs/div Start Up to Full Load Short Circuit Protection Vo 2V/div Vin 10V/div Vo 2V/div VLX 10V/div lo 2A/div IL 2A/div 2µs/div 10ms/div 50% to 100% Load Transient Short Circuit Recovery Vo 2V/div Vo ripple 0.2V/div VLX 10V/div Io 2A/div IL 2A/div 100µs/div Rev. 0.6 August 2012 10ms/div www.aosmd.com Page 5 of 14 AOZ3011PI Efficiency Efficiency (VIN = 12V, L = 4.7µH) vs. Load Current 100 95 90 Efficiency (%) 85 80 75 5V OUTPUT 3.3V OUTPUT 1.8V OUTPUT 70 65 1.2V OUTPUT 60 55 50 0 0.5 1.0 1.5 2.0 2.5 3.0 Load Current (A) Detailed Description The AOZ3011PI is a current-mode step down regulator with an integrated high-side PMOS switch and a low-side NMOS switch. The AOZ3011PI operates from a 4.5 V to 18 V input voltage range and supplies up to 3 A of load current. Features include enable control, power-on reset, input under voltage lockout, output over voltage protection, external soft-start and thermal shut down. The AOZ3011PI is available in an exposed pad SO-8 package. Enable and Soft Start The AOZ3011PI has an external soft start feature to limit in-rush current and ensure the output voltage ramps up smoothly to regulation voltage. The soft start process begins when the input voltage rises to 4.1 V and voltage on the EN pin is HIGH. In the soft start process, the FB voltage is ramped to follow the voltage of the soft start pin until it reaches 0.8 V. The voltage of the soft-start pin is charged by an internal 5 µA current. The EN pin of the AOZ3011PI is active high. Connect the EN pin to VIN if the enable function is not used. Pulling EN to ground will disable the AOZ3011PI. Do not leave EN open. The voltage on the EN pin must be above 2 V to enable the AOZ3011PI. When the EN pin voltage falls below 0.6 V, the AOZ3011PI is disabled. Rev. 0.6 August 2012 Steady-State Operation Under heavy load steady-state conditions, the converter operates in fixed frequency and Continuous-Conduction Mode (CCM). The AOZ3011PI integrates an internal P-MOSFET as the high-side switch. Inductor current is sensed by amplifying the voltage drop across the drain to source of the high side power MOSFET. Output voltage is divided down by the external voltage divider at the FB pin. The difference of the FB pin voltage and reference voltage is amplified by the internal transconductance error amplifier. The error voltage, which shows on the COMP pin, is compared against the current signal, which is the sum of inductor current signal and ramp compensation signal, at the PWM comparator input. If the current signal is less than the error voltage, the internal high-side switch is on. The inductor current flows from the input through the inductor to the output. When the current signal exceeds the error voltage, the high-side switch is off. The inductor current is freewheeling through the internal low-side N-MOSFET switch to output. The internal adaptive FET driver guarantees no turn on overlap of both the high-side and the low-side switch. www.aosmd.com Page 6 of 14 AOZ3011PI Compared with regulators using freewheeling Schottky diodes, the AOZ3011PI uses a freewheeling NMOSFET to realize synchronous rectification. This greatly improves the converter efficiency and reduces power loss in the low-side switch. The AOZ3011PI uses a P-Channel MOSFET as the high-side switch. This saves the bootstrap capacitor normally seen in a circuit using an NMOS switch. It also allows 100 % turn-on of the high-side switch to achieve linear regulation mode of operation. The minimum voltage drop from VIN to VO is the load current times DC resistance of the MOSFET plus DC resistance of the buck inductor. It can be calculated by equation below: V O_MAX = V IN – I O R DS ON where; VO_MAX is the maximum output voltage, VIN is the input voltage from 4.5 V to 18 V, IO is the output current from 0 A to 3 A, and RDS(ON) is the on resistance of the internal MOSFET. Output Voltage Programming Output voltage can be set by feeding back the output to the FB pin using a resistor divider network as shown in Figure 1. The resistor divider network includes R1 and R2. Usually, a design is started by picking a fixed R2 value and calculating the required R1 with the equation below: R 1 V O = 0.8 1 + ------- R 2 Since the switch duty cycle can be as high as 100 %, the maximum output voltage can be set as high as the input voltage minus the voltage drop on the upper PMOS and the inductor. Protection Features The AOZ3011PI has multiple protection features to prevent system circuit damage under abnormal conditions. Over Current Protection (OCP) The sensed inductor current signal is also used for over current protection. Since the AOZ3011PI employs peak current mode control, the COMP pin voltage is proportional to the peak inductor current. The COMP pin voltage is limited to be between 0.4 V and 2.5 V internally. The peak inductor current is automatically limited cycle-by-cycle. When the output is shorted to ground under fault conditions, the inductor current slowly decays during a switching cycle because the output voltage is 0 V. To prevent catastrophic failure, a secondary current limit is designed inside the AOZ3011PI. The measured inductor current is compared against a preset voltage which represents the current limit, between 3.5 A and 5.0 A. When the output current is greater than the current limit, the high side switch will be turned off. The converter will initiate a soft start once the over-current condition is resolved. Power-On Reset (POR) Some standard value of R1 and R2 for the most common output voltages are listed in Table 1. VO (V) R1 (kΩ) R2 (kΩ) 0.8 1.0 Open 1.2 4.99 10 1.5 10 11.5 1.8 12.7 10.2 2.5 21.5 10 3.3 31.1 10 5.0 52.3 10 A power-on reset circuit monitors the input voltage. When the input voltage exceeds 4.1 V, the converter starts operation. When input voltage falls below 3.7 V, the converter will be shut down. Thermal Protection An internal temperature sensor monitors the junction temperature. The sensor shuts down the internal control circuit and high side PMOS if the junction temperature exceeds 150 ºC. The regulator will restart automatically under the control of the soft-start circuit when the junction temperature decreases to 100 ºC. Table 1. The combination of R1 and R2 should be large enough to avoid drawing excessive current from the output, which will cause power loss. Rev. 0.6 August 2012 www.aosmd.com Page 7 of 14 AOZ3011PI Application Information The basic AOZ3011PI application circuit is show in Figure 1. Component selection is explained below. Input Capacitor The input capacitor must be connected to the VIN pin and the PGND pin of AOZ3011PI to maintain steady input voltage and filter out the pulsing input current. The voltage rating of input capacitor must be greater than maximum input voltage plus ripple voltage. The input ripple voltage can be approximated by equation below: VO VO IO V IN = ----------------- 1 – --------- --------f C IN V IN V IN For reliable operation and best performance, the input capacitors must have a current rating higher than ICIN_RMS at the worst operating conditions. Ceramic capacitors are preferred for input capacitors because of their low ESR and high current rating. Depending on the application circuits, other low ESR tantalum capacitors may be used. When selecting ceramic capacitors, X5R or X7R type dielectric ceramic capacitors should be used for their better temperature and voltage characteristics. Note that the ripple current rating from capacitor manufactures are based on a certain operating life time. Further de-rating may need to be considered for long term reliability. Inductor Since the input current is discontinuous in a buck converter, the current stress on the input capacitor is another concern when selecting the capacitor. For a buck circuit, the RMS value of input capacitor current can be calculated by: VO VO - 1 – -------- I CIN_RMS = I O -------V IN V IN The inductor is used to supply constant current to output when it is driven by a switching voltage. For a given input and output voltage, inductance and switching frequency together decide the inductor ripple current, which is: VO VO - I L = ----------- 1 – -------fL V IN The peak inductor current is: I L I Lpeak = I O + -------2 if we let m equal the conversion ratio: VO -------- = m V IN The relationship between the input capacitor RMS current and voltage conversion ratio is calculated and shown in Figure 2 below. It can be seen that when VO is half of VIN, CIN is under the worst current stress. The worst current stress on CIN is 0.5 x IO. 0.5 High inductance gives low inductor ripple current but requires larger size inductor to avoid saturation. Low ripple current reduces inductor core losses. It also reduces RMS current through inductor and switches, which results in less conduction loss. Usually, peak to peak ripple current on the inductor is designed to be 20 % to 40 % of output current. When selecting the inductor, confirm it is able to handle the peak current without saturation at the highest operating temperature. 0.4 The inductor takes the highest current in a buck circuit. The conduction loss on the inductor needs to be checked for thermal and efficiency requirements. ICIN_RMS(m) 0.3 IO 0.2 0.1 0 0 0.5 m 1 Surface mount inductors in different shape and styles are available from Coilcraft, Elytone and Murata. Shielded inductors are small and radiate less EMI noise. However, they cost more than unshielded inductors. The choice depends on EMI requirement, price and size. Figure 2. ICIN vs. Voltage Conversion Ratio Rev. 0.6 August 2012 www.aosmd.com Page 8 of 14 AOZ3011PI Output Capacitor The output capacitor is selected based on the DC output voltage rating, output ripple voltage specification and ripple current rating. The selected output capacitor must have a higher rated voltage specification than the maximum desired output voltage including ripple. De-rating needs to be considered for long term reliability. Output ripple voltage specification is another important factor for selecting the output capacitor. In a buck converter circuit, output ripple voltage is determined by inductor value, switching frequency, output capacitor value and ESR. It can be calculated by the equation below: 1 V O = I L ESR CO + ------------------------- 8fC Loop Compensation The AOZ3011PI employs peak current mode control for ease of use and fast transient response. Peak current mode control eliminates the double pole effect of the output L&C filter. It also greatly simplifies the compensation loop design. With peak current mode control, the buck power stage can be simplified to be a one-pole and one-zero system in frequency domain. The pole is dominant pole can be calculated by: 1 f P1 = ----------------------------------2 C O R L O where, CO is output capacitor value, and The zero is a ESR zero due to the output capacitor and its ESR. It is can be calculated by: ESRCO is the equivalent series resistance of the output capacitor. When a low ESR ceramic capacitor is used as the output capacitor, the impedance of the capacitor at the switching frequency dominates. Output ripple is mainly caused by capacitor value and inductor ripple current. The output ripple voltage calculation can be simplified to: 1 V O = I L ------------------------8fC 1 f Z1 = -----------------------------------------------2 C O ESR CO where; CO is the output filter capacitor, RL is load resistor value, and ESRCO is the equivalent series resistance of output capacitor. O If the impedance of ESR at switching frequency dominates, the output ripple voltage is mainly decided by capacitor ESR and inductor ripple current. The output ripple voltage calculation can be further simplified to: V O = I L ESR CO For lower output ripple voltage across the entire operating temperature range, X5R or X7R dielectric type of ceramic, or other low ESR tantalum capacitors are recommended as output capacitors. In a buck converter, output capacitor current is continuous. The RMS current of output capacitor is decided by the peak to peak inductor ripple current. It can be calculated by: I L I CO_RMS = ---------12 Usually, the ripple current rating of the output capacitor is a smaller issue because of the low current stress. When the buck inductor is selected to be very small and inductor ripple current is high, the output capacitor could be overstressed. The compensation design shapes the converter control loop transfer function for the desired gain and phase. Several different types of compensation networks can be used with the AOZ3011PI. For most cases, a series capacitor and resistor network connected to the COMP pin sets the pole-zero and is adequate for a stable high-bandwidth control loop. In the AOZ3011PI, FB and COMP are the inverting input and the output of the internal error amplifier. A series R and C compensation network connected to COMP provides one pole and one zero. The pole is: G EA f P2 = ------------------------------------------2 C C G VEA where; GEA is the error amplifier transconductance, which is 200 x 10-6 A/V, GVEA is the error amplifier voltage gain, which is 500 V/V, and CC is the compensation capacitor in Figure 1. Rev. 0.6 August 2012 www.aosmd.com Page 9 of 14 AOZ3011PI The zero given by the external compensation network, capacitor CC and resistor RC, is located at: 1 f Z2 = ----------------------------------2 C C R C To design the compensation circuit, a target crossover frequency fC to close the loop must be selected. The system crossover frequency is where the control loop has unity gain. The crossover is the also called the converter bandwidth. Generally a higher bandwidth means faster response to load transients. However, the bandwidth should not be too high because of system stability concern. When designing the compensation loop, converter stability under all line and load condition must be considered. Usually, it is recommended to set the bandwidth to be equal or less than 1/10 of the switching frequency. The strategy for choosing RC and CC is to set the cross over frequency with RC and set the compensator zero with CC. Using selected crossover frequency, fC, to calculate RC: VO 2 C C R C = f C ---------- ----------------------------V G G FB EA Thermal Management and Layout Considerations In the AOZ3011PI buck regulator circuit, high pulsing current flows through two circuit loops. The first loop starts from the input capacitors, to the VIN pin, to the LX pad, to the filter inductor, to the output capacitor and load, and then returns to the input capacitor through ground. Current flows in the first loop when the high side switch is on. The second loop starts from the inductor, to the output capacitors and load, to the low side NMOSFET. Current flows in the second loop when the low side NMOSFET is on. In PCB layout, minimizing the area of the two loops will reduce the noise of the circuit and improves efficiency. A ground plane is strongly recommended to connect the input capacitor, the output capacitor, and the PGND pin of the AOZ3011PI. In the AOZ3011PI buck regulator circuit, the major power dissipating components are the AOZ3011PI and the output inductor. The total power dissipation of converter circuit can be measured by input power minus output power: P total_loss = V IN I IN – V O I O CS where; fC is the desired crossover frequency. For best performance, fC is set to be about 1/10 of the switching frequency; VFB is 0.8V, GEA is the error amplifier transconductance, which is 200 x 10-6 A/V, and GCS is the current sense circuit transconductance, which is 8 A/V The compensation capacitor CC and resistor RC together make a zero. This zero is put somewhere close to the dominate pole fp1 but lower than 1/5 of the selected crossover frequency. CC can is selected by: 1.5 C C = ----------------------------------2 R C f P1 The power dissipation of the inductor can be approximately calculated by the output current and DCR value of the inductor: P inductor_loss = IO2 R inductor 1.1 The actual junction temperature can be calculated by the power dissipation in the AOZ3011PI and the thermal impedance from junction to ambient: T junction = P total_loss – P inductor_loss JA The maximum junction temperature of the AOZ3011PI is 150 ºC, which limits the maximum load current capability. Please see the thermal de-rating curves for maximum load current of the AOZ3011PI under different ambient temperature. The thermal performance of the AOZ3011PI is strongly affected by the PCB layout. Care should be taken during the design process to ensure that the IC will operate under the recommended environmental conditions. The above equation can be simplified to: CO RL C C = --------------------RC An easy-to-use application software which helps to design and simulate the compensation loop can be found at www.aosmd.com. Rev. 0.6 August 2012 www.aosmd.com Page 10 of 14 AOZ3011PI Layout Considerations The AOZ3011PI is an exposed pad SO-8 package. Several layout tips are listed for the best electric and thermal performance. 1. The exposed pad (LX) is connected to the internal PFET and NFET drains. Connected a large copper plane to the LX pin to help thermal dissipation. 2. Do not use a thermal relief connection to the VIN pin or the PGND pin. Pour a maximized copper area to the PGND pin and the VIN pin to help thermal dissipation. 3. The input capacitor should be connected as close as possible to the VIN pin and the PGND pin. 4. A ground plane is preferred. If a ground plane is not used, separate PGND from AGND and only connect them at one point to avoid the PGND pin noise coupling to the AGND pin. 5. Make the current trace from the LX pad to L to Co to the PGND as short as possible. 6. Pour copper plane on all unused board area and connect it to stable DC nodes, like VIN, GND or VOUT. 7. Keep sensitive signal trace away from the LX pad. Rev. 0.6 August 2012 www.aosmd.com Page 11 of 14 AOZ3011PI Package Dimensions, SO-8 EP1 Gauge plane 0.2500 D0 C L L1 E2 E1 E3 E L1' D1 Note 5 D θ 7 (4x) A2 e B A A1 Dimensions in millimeters RECOMMENDED LAND PATTERN 3.70 2.20 5.74 2.71 2.87 0.80 1.27 0.635 UNIT: mm Symbols A Min. 1.40 Nom. 1.55 A1 A2 B 0.00 1.40 0.31 0.05 1.50 0.406 C D D0 D1 E e E1 E2 E3 L y θ | L1–L1' | L1 0.17 4.80 3.20 3.10 5.80 — 3.80 2.21 — 4.96 3.40 3.30 6.00 1.27 3.90 2.41 0.40 REF 0.40 0.95 — — 0° — Max. 1.70 0.10 1.60 0.51 0.25 5.00 3.60 3.50 6.20 — 4.00 2.61 1.27 0.10 8° 3° 0.04 0.12 1.04 REF Dimensions in inches Symbols A Min. Nom. Max. 0.055 0.061 0.002 0.059 0.067 0.004 0.063 0.016 — 0.195 0.020 0.010 0.197 A1 A2 B 0.000 0.055 C D D0 D1 E e E1 E2 E3 L y θ | L1–L1' | L1 0.007 0.189 0.012 0.134 0.142 0.130 0.138 0.236 0.244 0.050 — 0.153 0.157 0.095 0.103 0.016 REF 0.016 0.037 0.050 — 0.004 — 0.126 0.122 0.228 — 0.150 0.087 0° — 3° 8° 0.002 0.005 0.041 REF Notes: 1. Package body sizes exclude mold flash and gate burrs. 2. Dimension L is measured in gauge plane. 3. Tolerance 0.10mm unless otherwise specified. 4. Controlling dimension is millimeter, converted inch dimensions are not necessarily exact. 5. Die pad exposure size is according to lead frame design. 6. Followed from JEDEC MS-012 Rev. 0.6 August 2012 www.aosmd.com Page 12 of 14 AOZ3011PI Tape and Reel Dimensions, SO-8 EP1 Carrier Tape P1 D1 P2 T E1 E2 E B0 K0 A0 D0 P0 Feeding Direction UNIT: mm Package SO-8 (12mm) A0 6.40 ±0.10 B0 5.20 ±0.10 K0 2.10 ±0.10 D0 1.60 ±0.10 D1 E 1.50 ±0.10 12.00 ±0.10 Reel E1 1.75 ±0.10 E2 5.50 ±0.10 P0 8.00 ±0.10 P2 2.00 ±0.10 P1 4.00 ±0.10 T 0.25 ±0.10 W1 S G N M K V R H W UNIT: mm N W Tape Size Reel Size M 12mm ø330 ø330.00 ø97.00 13.00 ±0.10 ±0.30 ±0.50 W1 17.40 ±1.00 H K ø13.00 10.60 +0.50/-0.20 S 2.00 ±0.50 G — R — V — Leader/Trailer and Orientation Trailer Tape 300mm min. or 75 empty pockets Rev. 0.6 August 2012 Components Tape Orientation in Pocket www.aosmd.com Leader Tape 500mm min. or 125 empty pockets Page 13 of 14 AOZ3011PI Part Marking Z3011PI FAYWLT Part Number Code Assembly Lot Code Fab & Assembly Location Year & Week Code This data sheet contains preliminary data; supplementary data may be published at a later date. Alpha & Omega Semiconductor reserves the right to make changes at any time without notice. LIFE SUPPORT POLICY ALPHA & OMEGA SEMICONDUCTOR PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS. As used herein: 1. Life support devices or systems are devices or systems which, (a) are intended for surgical implant into the body or (b) support or sustain life, and (c) whose failure to perform when properly used in accordance with instructions for use provided in the labeling, can be reasonably expected to result in a significant injury of the user. Rev. 0.6 August 2012 2. A critical component in any component of a life support, device, or system whose failure to perform can be reasonably expected to cause the failure of the life support device or system, or to affect its safety or effectiveness. www.aosmd.com Page 14 of 14