ETC TPA0211DGNR

TPA0211
2-W MONO AUDIO POWER AMPLIFIER
SLOS275C – JANUARY 2000 – REVISED APRIL 2001
D
D
D
D
D
D
D
D
D
DGN PACKAGE
(TOP VIEW)
Ideal for Wireless Communicators,
Notebook PCs, PDAs, and Other Small
Portable Audio Devices
2 W Into 4 Ω From 5-V Supply
0.6 W Into 4 Ω From 3-V Supply
Wide Power Supply Compatibility
3 V to 5 V
Low Supply Current
– 4 mA Typical at 5 V
– 4 mA Typical at 3 V
Shutdown Control . . . 1 µA Typical
Shutdown Pin Is TTL Compatible
–40°C to 85°C Operating Temperature
Range
Space-Saving, Thermally-Enhanced MSOP
Packaging
IN
SHUTDOWN
VDD
BYPASS
1
8
2
7
3
6
4
5
VO–
GND
SE/BTL
VO+
description
The TPA0211 is a 2-W mono bridge-tied-load (BTL) amplifier designed to drive speakers with as low as 4-Ω
impedance. The device is ideal for use in small wireless communicators, notebook PCs, PDAs, anyplace a
mono speaker and stereo headphones are required. From a 5-V supply, the TPA0211 can deliver 2 W of power
into a 4-Ω speaker.
The gain of the input stage is set by the user-selected input resistor and a 50-kΩ internal feedback resistor
(AV = – RF/RI). The power stage is internally configured with a gain of –1.25 V/V in SE mode, and –2.5 V/V in
BTL mode. Thus, the overall gain of the amplifier is –62.5 kΩ/ RI in SE mode and –125 kΩ/RI in BTL mode. The
input terminals are high-impedance CMOS inputs, and can be used as summing nodes.
The TPA0211 is available in the 8-pin thermally-enhanced MSOP package (DGN) and operates over an ambient
temperature range of –40°C to 85°C.
AVAILABLE OPTIONS
TA
PACKAGED DEVICES
MSOP†
(DGN)
MSOP
SYMBOLIZATION
– 40°C to 85°C
TPA0211DGN
AEG
† The DGN package are available taped and reeled. To order a taped and reeled part, add the
suffix R to the part number (e.g., TPA0211DGNR).
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPAD is a trademark of Texas Instruments.
Copyright  2001, Texas Instruments Incorporated
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
1
TPA0211
2-W MONO AUDIO POWER AMPLIFIER
SLOS275C – JANUARY 2000 – REVISED APRIL 2001
functional block diagram
CB
4
BYPASS
VDD
3
VDD
GND
7
BYPASS
VDD
50 kΩ
1.25*R
Audio
Input
Ci
RI
1
IN
100 kΩ
R
–
CC
–
VO+
+
5
+
BYPASS
100 kΩ
BYPASS
1 kΩ
50 kΩ
SE/BTL
Control
SE/BTL
6
VO–
8
50 kΩ
1.25*R
–
R
–
+
+
BYPASS
BYPASS
From
System Control
2
SHUTDOWN
Shutdown
and Depop Circuitry
Terminal Functions
TERMINAL
NAME
NO.
I/O
DESCRIPTION
I
BYPASS is the tap to the voltage divider for internal mid-supply bias. This terminal should be connected to a
0.1-µF to 1-µF capacitor.
BYPASS
4
GND
7
IN
1
I
IN is the audio input terminal.
SE/BTL
6
I
When SE/BTL is held low, the TPA0211 is in BTL mode. When SE/BTL is held high, the TPA0211 is in SE mode.
SHUTDOWN
2
I
SHUTDOWN places the entire device in shutdown mode when held low. TTL compatible input.
VDD
VO+
3
5
O
VDD is the supply voltage terminal.
VO+ is the positive output for BTL and SE modes.
VO–
8
O
VO– is the negative output in BTL mode and a high-impedance output in SE mode.
2
GND is the ground connection.
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
TPA0211
2-W MONO AUDIO POWER AMPLIFIER
SLOS275C – JANUARY 2000 – REVISED APRIL 2001
absolute maximum ratings over operating free-air temperature range (unless otherwise noted)†
Supply voltage, VDD . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 V
Input voltage, VI . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to VDD +0.3 V
Continuous total power dissipation . . . . . . . . . . . . . . . . . . . . . internally limited (see Dissipation Rating Table)
Operating free-air temperature range, TA . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . – 40°C to 85°C
Operating junction temperature range, TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . – 40°C to 150°C
Storage temperature range, Tstg . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 260°C
†Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE
DGN
TA ≤ 25°C
2.14 W‡
DERATING FACTOR
17.1 mW/°C
TA = 70°C
1.37 W
TA = 85°C
1.11 W
‡ See the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report (literature
number SLMA002), for more information on the PowerPAD package. The thermal data was measured on a
PCB layout based on the information in the section entitled Texas Instruments Recommended Board for
PowerPAD on page 33 of the before mentioned document.
recommended operating conditions
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Supply voltage, VDD
High-level input voltage, VIH
VDD = 3 V
VDD = 5 V
SE/BTL
MAX
2.5
5.5
UNIT
V
2.7
V
4.5
SHUTDOWN
Low-level input voltage, VIL
MIN
2
VDD = 3 V
VDD = 5 V
SE/BTL
1.65
2.75
V
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SHUTDOWN
0.8
Operating free-air temperature, TA
– 40
85
°C
electrical characteristics at specified free-air temperature, VDD = 3 V, TA = 25°C (unless otherwise
noted)
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PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
|VOO|
Output offset voltage (measured differentially)
30
mV
IDD
IDD(SD)
Supply current
4
6
mA
Supply current, shutdown mode
1
10
µA
operating characteristics, VDD = 3 V, TA = 25°C, RL = 4 Ω
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ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
PARAMETER
PO
Output power
THD + N
Total harmonic distortion plus noise
BOM
Maximum output power bandwidth
SNR
Signal-to-noise ratio
Vn
Output noise voltage
TEST CONDITIONS
MIN
TYP
THD = 1%,
BTL mode,
f = 1 kHz
660
THD = 0.1%,
SE mode, f = 1 kHz,
RL = 32 Ω
33
PO = 500 mW,
Gain = 2,
f = 20 Hz to 20 kHz
CB = 0.47 µF,
f = 20 Hz to
20 kHz
mW
20
kHz
88
dB
BTL mode, RL = 8 Ω,
AV= 8 dB
65
SE mode, RL = 32 Ω,
AV= 2 dB
25
• DALLAS, TEXAS 75265
UNIT
0.3%
THD = 2%
POST OFFICE BOX 655303
MAX
µVRMS
3
TPA0211
2-W MONO AUDIO POWER AMPLIFIER
SLOS275C – JANUARY 2000 – REVISED APRIL 2001
electrical characteristics at specified free-air temperature, VDD = 5 V, TA = 25°C (unless otherwise
noted)
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
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PARAMETER
TEST CONDITIONS
|VOO|
Output offset voltage (measured differentially)
IDD
IDD(SD)
Supply current
Supply current, shutdown mode
MIN
TYP
MAX
UNIT
30
mV
4
6
mA
1
10
µA
operating characteristics, VDD = 5 V, TA = 25°C, RL = 4 Ω
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ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
THD = 1%,
BTL mode,
f = 1 kHz
2
W
THD = 0.1%,
SE mode, f = 1 kHz,
RL = 32 Ω
92
mW
Total harmonic distortion plus
noise
PO = 1.5 W,
f = 20 Hz to 20 kHz
BOM
Maximum output power bandwidth
Gain = 2.5,
THD = 2%
SNR
Signal-to-noise ratio
PO
Output power
THD + N
Vn
CB = 0.47 µF,
f = 20 Hz to
20 kHz
Output noise voltage
0.2%
20
kHz
93
dB
BTL mode, RL = 8 Ω,
AV= 8 dB
65
SE mode, RL = 32 Ω,
AV= 2 dB
25
µVRMS
TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
Supply ripple rejection ratio
vs Frequency
IDD
Supply current
vs Supply voltage
3
PO
Output power
vs Supply voltage
4, 5
vs Load resistance
6, 7
THD+N
Total harmonic distortion plus noise
Vn
Output noise voltage
vs Frequency
vs Output power
vs Frequency
Closed loop gain and phase
4
1, 2
8, 9, 10, 11
12, 13, 14, 15,
16, 17
18, 19
20, 21
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
TPA0211
2-W MONO AUDIO POWER AMPLIFIER
SLOS275C – JANUARY 2000 – REVISED APRIL 2001
TYPICAL CHARACTERISTICS
SUPPLY RIPPLE REJECTION RATIO
vs
FREQUENCY
SUPPLY RIPPLE REJECTION RATIO
vs
FREQUENCY
0
0
RL = 8 Ω,
CB = 1 µ,F
Mode = BTL
–20
RL = 32 Ω,
CB = 1 µF,
Mode = SE
–10
Supply Ripple Rejection Ratio – dB
Supply Ripple Rejection Ratio – dB
–10
–30
–40
–50
–60
–70
–80
–90
–20
–30
–40
–50
–60
–70
–80
–90
–100
20
50
100 200
500 1 k 2 k
–100
5 k 10 k 20 k
20
50
f – Frequency – Hz
100 200
Figure 1
OUTPUT POWER
vs
SUPPLY VOLTAGE
4
3
THD+N = 1%,
f = 1 kHz,
Mode = BTL,
AV = 8 dB
3.5
2.5
TA = 125 °C
3
PO – Output Power – W
I DD – Supply Current – mA
5 k 10 k 20 k
Figure 2
SUPPLY CURRENT
vs
SUPPLY VOLTAGE
TA = 25 °C
2.5
2
TA = –40 °C
1.5
SHUTDOWN = High
VDD From Low to High,
Mode = SE,
RL = Open,
Temperature From Hot to Cold
1
0.5
0
2.5
500 1 k 2 k
f – Frequency – Hz
3
3.5
4
4.5
VDD – Supply Voltage – V
5
RL = 4 Ω
2
RL = 8 Ω
1.5
1
0.5
5.5
0
3
3.5
Figure 3
4
4.5
5
VDD – Supply Voltage – V
5.5
Figure 4
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
5
TPA0211
2-W MONO AUDIO POWER AMPLIFIER
SLOS275C – JANUARY 2000 – REVISED APRIL 2001
TYPICAL CHARACTERISTICS
OUTPUT POWER
vs
LOAD RESISTANCE
OUTPUT POWER
vs
SUPPLY VOLTAGE
2.5
500
THD+N = 1%,
f = 1 kHz,
Mode = SE,
AV = 2 dB
2
PO – Output Power – W
PO – Output Power – mW
400
THD+N = 1%,
f = 1 kHz,
Mode = BTL,
AV = 8 dB
RL = 8 Ω
300
200
1.5
VDD = 5 V
1
RL = 32 Ω
100
0.5
0
0
VDD = 3 V
3
3.5
4
4.5
5
VDD – Supply Voltage – V
5.5
0
10
Figure 6
OUTPUT POWER
vs
LOAD RESISTANCE
700
THD+N = 1%,
f = 1 kHz,
Mode = SE,
AV = 2 dB
600
PO – Output Power – mW
30
500
400
300
VDD = 5 V
200
100
VDD = 3 V
0
0
10
20
30
40
50
RL – Load Resistance – Ω
Figure 7
POST OFFICE BOX 655303
40
50
RL – Load Resistance – Ω
Figure 5
6
20
• DALLAS, TEXAS 75265
60
60
TPA0211
2-W MONO AUDIO POWER AMPLIFIER
SLOS275C – JANUARY 2000 – REVISED APRIL 2001
TYPICAL CHARACTERISTICS
10
VDD = 3 V,
PO = 250 mW,
RL = 8 Ω,
Mode = BTL
5
2
1
0.5
0.2
0.1
AV = 20 dB
0.05
0.02
AV = 8 dB
0.01
0.005
0.002
0.001
20
50
100 200
500 1 k 2 k
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
THD+N – Total Harmonic Distortion Plus Noise – %
THD+N – Total Harmonic Distortion Plus Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
5 k 10 k 20 k
10
VDD = 5 V,
PO = 1 W,
RL = 8 Ω,
Mode = BTL
5
2
1
0.5
0.2
0.1
AV = 20 dB
0.05
0.02
0.01
AV = 8 dB
0.005
0.002
0.001
20
50
100 200
f – Frequency – Hz
Figure 8
VDD = 3 V,
PO = 25 mW,
RL = 32 Ω,
Mode = SE
1
0.5
0.2
0.1
AV = 14 dB
0.05
0.02
0.01
AV = 2 dB
0.005
0.002
0.001
20
50
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
THD+N – Total Harmonic Distortion Plus Noise – %
THD+N – Total Harmonic Distortion Plus Noise – %
10
2
100 200
500 1 k 2 k
5 k 10 k 20 k
Figure 9
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
5
500 1 k 2 k
f – Frequency – Hz
5 k 10 k 20 k
10
VDD = 5 V,
PO = 75 mW,
RL = 32 Ω,
Mode = SE
5
2
1
0.5
0.2
0.1
AV = 14 dB
0.05
0.02
0.01
AV = 2 dB
0.005
0.002
0.001
20
50
f – Frequency – Hz
100 200
500 1 k 2 k
5 k 10 k 20 k
f – Frequency – Hz
Figure 10
Figure 11
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• DALLAS, TEXAS 75265
7
TPA0211
2-W MONO AUDIO POWER AMPLIFIER
SLOS275C – JANUARY 2000 – REVISED APRIL 2001
TYPICAL CHARACTERISTICS
10
VDD = 3 V,
RL = 4 Ω,
Mode = BTL,
AV = 8 dB
5
20 kHz
2
1
15 kHz
0.5
1 kHz
0.2
0.1
20 Hz
0.05
0.02
0.01
0.005
0.002
0.001
0.01 0.02 0.05
0.5 1
2
0.1 0.2
PO – Output Power – W
5
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
THD+N – Total Harmonic Distortion Plus Noise – %
THD+N – Total Harmonic Distortion Plus Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
10
10
20 kHz
2
1
0.5
15 kHz
0.2
1 kHz
0.1
0.05
20 Hz
0.02
0.01
0.005
0.002
0.001
0.01 0.02 0.05
Figure 12
VDD = 3 V,
RL = 32 Ω,
Mode = SE,
AV = 2 dB
1
0.5
20 kHz
0.2
0.1
0.05
15 kHz
0.02
20 Hz
0.01
1 kHz
0.005
0.002
0.001
10
10
20
40
PO – Output Power – mW
70
100
10
5
2
1
15 kHz
0.5
0.2
0.1
1 kHz
20 Hz
0.02
0.01
VDD = 5 V,
RL = 4 Ω,
Mode = BTL,
AV = 8 dB
0.005
0.002
0.001
0.01 0.02 0.05
0.5 1
2
0.1 0.2
PO – Output Power – W
Figure 15
POST OFFICE BOX 655303
20 kHz
0.05
Figure 14
8
5
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
THD+N – Total Harmonic Distortion Plus Noise – %
THD+N – Total Harmonic Distortion Plus Noise – %
10
2
0.5 1
2
0.1 0.2
PO – Output Power – W
Figure 13
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
5
VDD = 3 V,
RL = 8 Ω,
Mode = BTL,
AV = 8 dB
5
• DALLAS, TEXAS 75265
5
10
TPA0211
2-W MONO AUDIO POWER AMPLIFIER
SLOS275C – JANUARY 2000 – REVISED APRIL 2001
TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
10
THD+N – Total Harmonic Distortion Plus Noise – %
THD+N – Total Harmonic Distortion Plus Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
VDD = 5 V,
RL = 8 Ω,
Mode = BTL,
AV = 8 dB
5
2
1
20 kHz
0.5
0.2
15 kHz
1 kHz
0.1
0.05
20 Hz
0.02
0.01
0.005
0.002
0.001
0.01 0.02 0.05
0.5 1
2
0.1 0.2
PO – Output Power – W
5
10
10
VDD = 5 V,
RL = 32 Ω,
Mode = SE,
AV = 2 dB
5
2
1
0.5
0.2
20 kHz
0.1
0.05
15 kHz
0.02
20 Hz
0.01
1 kHz
0.005
0.002
0.001
0.01
0.02
0.2
0.05
0.1
PO – Output Power – W
Figure 16
1
Figure 17
OUTPUT NOISE VOLTAGE
vs
FREQUENCY
OUTPUT NOISE VOLTAGE
vs
FREQUENCY
1M
1M
VDD = 5 V,
RL = 8 Ω,
Mode = BTL,
AV = 8 dB
500
200
Vn – Output Noise Voltage – µV RMS
Vn – Output Noise Voltage – µV RMS
0.5
100
50
20
10
5
2
VDD = 5 V,
RL = 32 Ω,
Mode = SE,
AV = 2 dB
500
200
100
50
20
10
5
2
1
20
50
100 200
500 1 k 2 k
f – Frequency – Hz
5 k 10 k 20 k
1
20
50
Figure 18
100 200
500 1 k 2 k
f – Frequency – Hz
5 k 10 k 20 k
Figure 19
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TPA0211
2-W MONO AUDIO POWER AMPLIFIER
SLOS275C – JANUARY 2000 – REVISED APRIL 2001
TYPICAL CHARACTERISTICS
CLOSED LOOP RESPONSE
180°
30
Gain – dB
10
135°
Gain
90°
0
45°
Phase
–10
0°
–20
–45°
–30
–90°
–40
–135°
–50
10
Phase
20
VDD = 5 V,
RL = 4 Ω,
Mode = BTL,
AV = 8 dB
–180°
100
1k
10k
100k
1M
f – Frequency – Hz
Figure 20
CLOSED LOOP RESPONSE
30
Gain – dB
10
135°
Gain
90°
0
45°
Phase
–10
0°
–20
–45°
–30
–90°
–40
–135°
–50
10
100
1k
10k
100k
f – Frequency – Hz
Figure 21
10
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–180°
1M
Phase
20
180°
VDD = 5 V,
RL = 32 Ω,
Mode = SE,
AV = 2 dB
TPA0211
2-W MONO AUDIO POWER AMPLIFIER
SLOS275C – JANUARY 2000 – REVISED APRIL 2001
APPLICATION INFORMATION
gain setting via input resistance
The gain of the input stage is set by the user-selected input resistor and a 50-kΩ internal feedback resistor.
However, the power stage is internally configured with a gain of –1.25 V/V in SE mode, and –2.5 V/V in BTL
mode. Thus, the feedback resistor (RF) is effectively 62.5 kΩ in SE mode and 125 kΩ in BTL mode. Therefore,
the overall gain can be calculated using equations (1) and (2).
A
A
V
+ –125R kW
(BTL)
V
+ –62.5R kW
(SE)
(1)
I
(2)
I
The –3 dB frequency can be calculated using equation 3:
ƒ –3 dB
+ 2p 1R C
(3)
I i
If the filter must be more accurate, the value of the capacitor should be increased while the value of the resistor
to ground should be decreased. In addition, the order of the filter could be increased.
input capacitor, Ci
In the typical application an input capacitor, Ci, is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, Ci and the input resistance of the amplifier, RI, form a
high-pass filter with the corner frequency determined in equation 4.
–3 dB
f c(highpass)
+ 2 p R1 C
(4)
I i
fc
The value of Ci is important to consider as it directly affects the bass (low frequency) performance of the circuit.
Consider the example where RI is 10 kΩ and the specification calls for a flat bass response down to 40 Hz.
Equation 2 is reconfigured as equation 5.
C
i
+ 2 p 1R fc
(5)
I
In this example, CI is 0.4 µF so one would likely choose a value in the range of 0.47 µF to 1 µF. A further
consideration for this capacitor is the leakage path from the input source through the input network (Ci) and the
feedback network to the load. This leakage current creates a dc offset voltage at the input to the amplifier that
reduces useful headroom, especially in high gain applications. For this reason a low-leakage tantalum or
ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor
should face the amplifier input in most applications as the dc level there is held at VDD/2, which is likely higher
than the source dc level. Note that it is important to confirm the capacitor polarity in the application.
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TPA0211
2-W MONO AUDIO POWER AMPLIFIER
SLOS275C – JANUARY 2000 – REVISED APRIL 2001
APPLICATION INFORMATION
power supply decoupling, C(S)
The TPA0211 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling
to ensure the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also
prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is
achieved by using two capacitors of different types that target different types of noise on the power supply leads.
For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance
(ESR) ceramic capacitor, typically 0.1 µF placed as close as possible to the device VDD lead, works best. For
filtering lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 µF or greater placed near
the audio power amplifier is recommended.
midrail bypass capacitor, C(BYP)
The midrail bypass capacitor, C(BYP), is the most critical capacitor and serves several important functions.
During start-up or recovery from shutdown mode, C(BYP) determines the rate at which the amplifier starts up.
The second function is to reduce noise produced by the power supply caused by coupling into the output drive
signal. This noise is from the midrail generation circuit internal to the amplifier, which appears as degraded
PSRR and THD+N.
Bypass capacitor, C(BYP), values of 0.47 µF to 1 µF ceramic or tantalum low-ESR capacitors are recommended
for the best THD and noise performance.
output coupling capacitor, C(C)
In the typical single-supply SE configuration, an output coupling capacitor (C(C)) is required to block the dc bias
at the output of the amplifier thus preventing dc currents in the load. As with the input coupling capacitor, the
output coupling capacitor and impedance of the load form a high-pass filter governed by equation 6.
–3 dB
f c(high)
+ 2 p R1 C
(6)
L (C)
fc
The main disadvantage, from a performance standpoint, is the load impedances are typically small, which drives
the low-frequency corner higher, degrading the bass response. Large values of C(C) are required to pass low
frequencies into the load. Consider the example where a C(C) of 330 µF is chosen and loads vary from 3 Ω,
4 Ω, 8 Ω, 32 Ω, 10 kΩ, to 47 kΩ. Table 1 summarizes the frequency response characteristics of each
configuration.
12
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TPA0211
2-W MONO AUDIO POWER AMPLIFIER
SLOS275C – JANUARY 2000 – REVISED APRIL 2001
APPLICATION INFORMATION
Table 1. Common Load Impedances vs Low Frequency Output Characteristics in SE Mode
RL
C(C)
330 µF
Lowest Frequency
3Ω
4Ω
330 µF
120 Hz
60 Hz
161 Hz
8Ω
330 µF
32 Ω
330 µF
15 Hz
10,000 Ω
330 µF
0.05 Hz
47,000 Ω
330 µF
0.01 Hz
As Table 1 indicates, most of the bass response is attenuated into a 4-Ω load, an 8-Ω load is adequate,
headphone response is good, and drive into line level inputs (a home stereo for example) is exceptional.
Furthermore, the total amount of ripple current that must flow through the capacitor must be considered when
choosing the component. As shown in the application circuit, one coupling capacitor must be in series with the
mono loudspeaker for proper operation of the stereo-mono switching circuit. For a 4-Ω load, this capacitor must
be able to handle about 700 mA of ripple current for a continuous output power of 2 W.
using low-ESR capacitors
Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal)
capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this
resistance the more the real capacitor behaves like an ideal capacitor.
bridged-tied load versus single-ended mode
Figure 22 shows a Class-AB audio power amplifier (APA) in a BTL configuration. The TPA0211 BTL amplifier
consists of two Class-AB amplifiers driving both ends of the load. There are several potential benefits to this
differential drive configuration, but initially consider power to the load. The differential drive to the speaker
means that as one side is slewing up, the other side is slewing down, and vice versa. This in effect doubles the
voltage swing on the load as compared to a ground referenced load. Plugging 2 × VO(PP) into the power
equation, where voltage is squared, yields 4× the output power from the same supply rail and load impedance
(see equation 7).
V
+
(RMS)
V
+
V
Power
O(PP)
Ǹ
2 2
(7)
2
(RMS)
R
L
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TPA0211
2-W MONO AUDIO POWER AMPLIFIER
SLOS275C – JANUARY 2000 – REVISED APRIL 2001
APPLICATION INFORMATION
VDD
VO(PP)
RL
2x VO(PP)
VDD
–VO(PP)
Figure 22. Bridge-Tied Load Configuration
In a typical computer sound channel operating at 5 V, bridging raises the power into an 8-Ω speaker from a
singled-ended (SE, ground reference) limit of 250 mW to 1 W. In sound power, that is a 6-dB improvement—
which is loudness that can be heard. In addition to increased power, there are frequency response concerns.
Consider the single-supply SE configuration shown in Figure 23. A coupling capacitor is required to block the
dc offset voltage from reaching the load. These capacitors can be quite large (approximately 33 µF to 1000 µF)
so they tend to be expensive, heavy, occupy valuable PCB area, and have the additional drawback of limiting
low-frequency performance of the system. This frequency limiting effect is due to the high pass filter network
created with the speaker impedance and the coupling capacitance and is calculated with equation 8.
fc
14
+ 2 p R 1C
(8)
L (C)
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APPLICATION INFORMATION
bridged-tied load versus single-ended mode (continued)
For example, a 68-µF capacitor with an 8-Ω speaker would attenuate low frequencies below 293 Hz. The BTL
configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency
performance is then limited only by the input network and speaker response. Cost and PCB space are also
minimized by eliminating the bulky coupling capacitor.
VDD
–3 dB
VO(PP)
C(C)
RL
VO(PP)
fc
Figure 23. Single-Ended Configuration and Frequency Response
Increasing power to the load does carry a penalty of increased internal power dissipation. The increased
dissipation is understandable considering that the BTL configuration produces 4× the output power of the SE
configuration. Internal dissipation versus output power is discussed further in the crest factor and thermal
considerations section.
single-ended operation
In SE mode (see Figure 22 and Figure 23), the load is driven from one amplifier output (VO+, terminal 5).
The amplifier switches to single-ended operation when the SE/BTL terminal is held high.
BTL amplifier efficiency
Class-AB amplifiers are inefficient. The primary cause of inefficiencies is the voltage drop across the output
stage transistors. There are two components of the internal voltage drop. One is the headroom or dc voltage
drop that varies inversely to output power. The second component is due to the sinewave nature of the output.
The total voltage drop can be calculated by subtracting the RMS value of the output voltage from VDD. The
internal voltage drop multiplied by the RMS value of the supply current, IDDrms, determines the internal power
dissipation of the amplifier.
An easy-to-use equation to calculate efficiency starts out as being equal to the ratio of power from the power
supply to the power delivered to the load. To accurately calculate the RMS and average values of power in the
load and in the amplifier, the current and voltage waveform shapes must first be understood (see Figure 24).
IDD
VO
IDD(avg)
V(LRMS)
Figure 24. Voltage and Current Waveforms for BTL Amplifiers
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TPA0211
2-W MONO AUDIO POWER AMPLIFIER
SLOS275C – JANUARY 2000 – REVISED APRIL 2001
APPLICATION INFORMATION
BTL amplifier efficiency (continued)
Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very
different between SE and BTL configurations. In an SE application the current waveform is a half-wave rectified
shape, whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different.
Keep in mind that for most of the waveform both the push and pull transistors are not on at the same time, which
supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform.
The following equations are the basis for calculating amplifier efficiency.
Efficiency of a BTL amplifier
+ PP L
(9)
SUP
Where:
PL
+ VLRMS
R
L
and P SUP
2
, and V LRMS
+ VDD IDDavg
+ VǸ2P ,
2
therefore, P L
I DDavg
and
+ p1
ŕ
p
0
+ 2VRP
VP
RL
L
sin(t) dt
+ 1p
VP
RL
[cos(t)]
p
2V P
+
0
pR
L
Therefore,
P SUP
+ 2 VpDDR VP
L
substituting PL and PSUP into equation 9,
2
Efficiency of a BTL amplifier
Where:
VP
+ Ǹ2 PL RL
Therefore,
h BTL
+
p
VP
2 RL
+2 V
DD V P
p RL
DD
Ǹ
2 PL RL
4 V DD
(10)
PL = Power devilered to load
PSUP = Power drawn from power supply
VLRMS = RMS voltage on BTL load
RL = Load resistance
16
+ 4p VVP
VP = Peak voltage on BTL load
IDDavg = Average current drawn from the power supply
VDD = Power supply voltage
ηBTL = Efficiency of a BTL amplifier
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TPA0211
2-W MONO AUDIO POWER AMPLIFIER
SLOS275C – JANUARY 2000 – REVISED APRIL 2001
APPLICATION INFORMATION
BTL amplifier efficiency (continued)
Table 2 employs equation 10 to calculate efficiencies for four different output power levels. Note that the
efficiency of the amplifier is quite low for lower power levels and rises sharply as power to the load is increased
resulting in a nearly flat internal power dissipation over the normal operating range. Note that the internal
dissipation at full output power is less than in the half power range. Calculating the efficiency for a specific
system is the key to proper power supply design.
Table 2. Efficiency Vs Output Power in 5-V 8-Ω BTL Systems
Output Power
(W)
Efficiency
(%)
Peak Voltage
(V)
Internal Dissipation
(W)
0.25
31.4
2.00
0.55
0.50
44.4
2.83
0.62
1.00
62.8
0.59
1.25
70.2
4.00
4.47†
0.53
† High peak voltages cause the THD to increase.
A final point to remember about Class-AB amplifiers (either SE or BTL) is how to manipulate the terms in the
efficiency equation to utmost advantage when possible. Note that in equation 10, VDD is in the denominator.
This indicates that as VDD goes down, efficiency goes up.
crest factor and thermal considerations
Class-AB power amplifiers dissipate a significant amount of heat in the package under normal operating
conditions. A typical music CD requires 12 dB to 15 dB of dynamic range, or headroom above the average power
output, to pass the loudest portions of the signal without distortion. In other words, music typically has a crest
factor between 12 dB and 15 dB. When determining the optimal ambient operating temperature, the internal
dissipated power at the average output power level must be used. From the TPA0211 data sheet, one can see
that when the TPA0211 is operating from a 5-V supply into a 4-Ω speaker 4-W peaks are available. Converting
watts to dB:
P dB
+ 10 Log PPW + 10 Log 41 WW + 6 dB
(11)
ref
Subtracting the headroom restriction to obtain the average listening level without distortion yields:
6 dB – 15 dB = –9 dB (15-dB crest factor)
6 dB – 12 dB = –6 dB (12-dB crest factor)
6 dB – 9 dB = –3 dB (9-dB crest factor)
6 dB – 6 dB = 0 dB (6-dB crest factor)
6 dB – 3 dB = 3 dB (3-dB crest factor)
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TPA0211
2-W MONO AUDIO POWER AMPLIFIER
SLOS275C – JANUARY 2000 – REVISED APRIL 2001
APPLICATION INFORMATION
crest factor and thermal considerations (continued)
Converting dB back into watts:
PW
+ 10PdBń10 Pref
+ 63 mW (18-dB crest factor)
+ 125 mW (15-dB crest factor)
+ 250 mW (9-dB crest factor)
+ 500 mW (6-dB crest factor)
+ 1000 mW (3-dB crest factor)
+ 2000 mW (15-dB crest factor)
(12)
This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 2 W of continuous power output with a 3 dB crest
factor, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings for the
system. Using the power dissipation curves for a 5-V, 4-Ω system, the internal dissipation in the TPA0211 and
maximum ambient temperatures is shown in Table 3.
Table 3. TPA0211 Power Rating, 5-V, 4-Ω, Mono
PEAK OUTPUT POWER
(W)
AVERAGE OUTPUT POWER
POWER DISSIPATION
(W)
MAXIMUM AMBIENT
TEMPERATURE
4
2 W (3-dB crest factor)
1.7
– 3°C
4
1000 mW (6-dB crest factor)
1.6
6°C
4
500 mW (9-dB crest factor)
1.4
24°C
4
250 mW (12-dB crest factor)
1.1
51°C
4
125 mW (15-dB crest factor)
0.8
78°C
4
63 mW (18-dB crest factor)
0.6
96°C
As a result, this simple formula for calculating PDmax may be used for an 4-Ω application:
2V 2
P Dmax
+ p2RDD
(13)
L
However, in the case of a 4-Ω load, the PDmax occurs at a point well above the normal operating power level.
The amplifier may therefore be operated at a higher ambient temperature than required by the PDmax formula
for a 4-Ω load.
The maximum ambient temperature depends on the heat sinking ability of the PCB system. The derating factor
for the DGN package is shown in the dissipation rating table. Converting this to ΘJA:
Θ JA
18
1
+ Derating1 Factor + 0.0171
+ 58.48°CńW
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(14)
TPA0211
2-W MONO AUDIO POWER AMPLIFIER
SLOS275C – JANUARY 2000 – REVISED APRIL 2001
APPLICATION INFORMATION
crest factor and thermal considerations (continued)
To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are
per channel so the dissipated power needs to be doubled for two channel operation. Given ΘJA, the maximum
allowable junction temperature, and the total internal dissipation, the maximum ambient temperature can be
calculated with the following equation. The maximum recommended junction temperature for the TPA0211 is
150°C. The internal dissipation figures are taken from the Power Dissipation vs Output Power graphs.
T A Max
+ TJ Max * ΘJA PD + 150 * 58.48 (0.8
2)
+ 56°C (15-dB crest factor)
(15)
NOTE:
Internal dissipation of 0.8 W is estimated for a 2-W system with 15-dB crest factor per channel.
Table 3 shows that for some applications no airflow is required to keep junction temperatures in the specified
range. The TPA0211 is designed with thermal protection that turns the device off when the junction temperature
surpasses 150°C to prevent damage to the IC. Table 3 was calculated for maximum listening volume without
distortion. When the output level is reduced the numbers in the table change significantly. Also, using 8-Ω
speakers dramatically increases the thermal performance by increasing amplifier efficiency.
SE/BTL (stereo/mono) operation
The ability of the TPA0211 to easily switch between mono BTL and stereo SE modes is one of its most important
cost saving features. This feature eliminates the requirement for an additional headphone amplifier in
applications where an internal speaker is driven in BTL mode but an external headphone must be
accommodated. When SE/BTL is held high for SE mode, the VO– output goes into a high impedance state while
the VO+ output operates normally. When SE/BTL is held low, the VO– output operates normally, placing the
amplifier in BTL mode.
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TPA0211
2-W MONO AUDIO POWER AMPLIFIER
SLOS275C – JANUARY 2000 – REVISED APRIL 2001
APPLICATION INFORMATION
ST/BTL operation (continued)
CB
4
BYPASS
VDD
3
VDD
GND
7
BYPASS
VDD
50 kΩ
1.25*R
Audio
Input
Ci
RI
1
IN
100 kΩ
R
–
CC
–
VO+
+
5
+
BYPASS
100 kΩ
BYPASS
1 kΩ
50 kΩ
SE/BTL
Control
SE/BTL
6
VO–
8
50 kΩ
1.25*R
–
R
–
+
+
BYPASS
BYPASS
From
System Control
2
SHUTDOWN
Shutdown
and Depop Circuitry
Figure 25. TPA0211 Resistor Divider Network Circuit
Using a readily available 1/8-in. (3.5 mm) mono headphone jack, the control switch is closed when no plug is
inserted. When closed, the 100-kΩ/1-kΩ divider pulls the SE/BTL input low. When a plug is inserted, the 1-kΩ
resistor is disconnected and the SE/BTL input is pulled high.
20
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TPA0211
2-W MONO AUDIO POWER AMPLIFIER
SLOS275C – JANUARY 2000 – REVISED APRIL 2001
MECHANICAL DATA
DGN (S-PDSO-G8)
PowerPAD PLASTIC SMALL-OUTLINE PACKAGE
0,38
0,25
0,65
8
0,25 M
5
Thermal Pad
(See Note D)
0,15 NOM
3,05
2,95
4,98
4,78
Gage Plane
0,25
1
0°– 6°
4
3,05
2,95
0,69
0,41
Seating Plane
1,07 MAX
0,15
0,05
0,10
4073271/A 01/98
NOTES: A.
B.
C.
D.
All linear dimensions are in millimeters.
This drawing is subject to change without notice.
Body dimensions include mold flash or protrusions.
The package thermal performance may be enhanced by attaching an external heat sink to the thermal pad. This pad is electrically
and thermally connected to the backside of the die and possibly selected leads. The dimension of the thermal pad is 1.40 mm (height
as illustrated) × 1.80 (width as illustrated) mm (maximum). The pad is centered on the bottom of the package.
E. Falls within JEDEC MO-187
PowerPAD is a trademark of Texas Instruments.
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