TI TPA0162

TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B – JUNE 1999 – REVISED MARCH 2000
D
D
D
D
D
D
D
D
D
D
D
Compatible With PC 99 Desktop Line-Out
Into 10-kΩ Load
Compatible With PC 99 Portable Into 8-Ω
Load
Internal Gain Control, Which Eliminates
External Gain-Setting Resistors
Digital Volume Control From 20 dB to
–40 dB
2-W/Ch Output Power Into 3-Ω Load
PC-Beep Input
Depop Circuitry
Stereo Input MUX
Fully Differential Input
Low Supply Current and Shutdown Current
Surface-Mount Power Packaging
24-Pin TSSOP PowerPAD
PWP PACKAGE
(TOP VIEW)
GND
UP
DOWN
LOUT+
LLINEIN
LHPIN
PVDD
RIN
LOUT–
LIN
BYPASS
GND
1
2
3
4
5
6
7
8
9
10
11
12
24
23
22
21
20
19
18
17
16
15
14
13
GND
RLINEIN
SHUTDOWN
ROUT+
RHPIN
VDD
PVDD
CLK
ROUT–
SE/BTL
PC-BEEP
GND
description
The TPA0162 is a stereo audio power amplifier in a 24-pin TSSOP thermally enhanced package capable of
delivering 2 W of continuous RMS power per channel into 3-Ω loads. This device minimizes the number of
external components needed, which simplifies the design and frees up board space for other features. When
driving 1 W into 8-Ω speakers, the TPA0162 has less than 0.22% THD+N across its specified frequency range.
Included within this device is integrated depop circuitry that virtually eliminates transients that cause noise in
the speakers.
The overall gain of the amplifier is controlled digitally by the UP and DOWN terminals. At power up, the gain
is set at the lowest level, –85 dB. It can then be adjusted to any of 31 discrete steps by pulling the voltage down
at the desired pin to logic low. The gain is adjusted in the initial stage of the amplifier as opposed to the power
output stage. As a result, the THD changes very little over all volume levels.
An internal input MUX allows two sets of stereo inputs to the amplifier. In notebook applications, where internal
speakers are driven as BTL and the line outputs (often headphone drive) are required to be SE, the TPA0162
automatically switches into SE mode when the SE/BTL input is activated. This effectively reduces the gain by
6 dB.
The TPA0162 consumes only 20 mA of supply current during normal operation. A miserly shutdown mode is
included that reduces the supply current to less than 150 µA.
The PowerPAD package (PWP) delivers a level of thermal performance that was previously achievable only
in TO-220-type packages. Thermal impedances of approximately 35°C/W are truly realized in multilayer PCB
applications. This allows the TPA0162 to operate at full power into 8-Ω loads at ambient temperatures of 85°C.
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPAD is a trademark of Texas Instruments Incorporated.
Copyright  2000, Texas Instruments Incorporated
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
1
TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B – JUNE 1999 – REVISED MARCH 2000
functional block diagram
RHPIN
RLINEIN
R
MUX
32-Step
Volume
Control
–
ROUT+
+
UP
DOWN
RIN
–
ROUT–
+
PC-BEEP
PCBeep
Depop
Circuitry
SE/BTL
LHPIN
LLINEIN
MUX
Control
L
MUX
Power
Management
PVDD
VDD
BYPASS
SHUTDOWN
GND
32-Step
Volume
Control
–
LOUT+
+
LIN
–
LOUT–
+
2
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B – JUNE 1999 – REVISED MARCH 2000
AVAILABLE OPTIONS
PACKAGED DEVICE
TSSOP†
(PWP)
TA
– 40°C to 85°C
TPA0162PWP
† The PWP package is available taped and reeled. To order a taped and reeled part,
add the suffix R to the part number (e.g., TPA0162PWPR).
Terminal Functions
TERMINAL
NAME
NO.
I/O
DESCRIPTION
BYPASS
11
CLK
17
I
If a 47-nF capacitor is attached, the TPA0162 generates an internal clock. An external clock can override
the internal clock input to this terminal.
DOWN
3
I
A momentary pulse on this terminal decreases the volume level by 2 dB. Holding the terminal low for a period
of time will step the amplifier through the volume levels at a rate determined by the capacitor on the CLK
terminal.
GND
Tap to voltage divider for internal mid-supply bias generator
1, 12
13, 24
Ground connection for circuitry. Connected to thermal pad
LHPIN
6
I
Left-channel headphone input, selected when SE/BTL is held high
LIN
10
I
Common left input for fully differential input. AC ground for single-ended inputs
LLINEIN
5
I
Left-channel line negative input, selected when SE/BTL is held low
LOUT+
4
O
Left-channel positive output in BTL mode and positive in SE mode
LOUT–
9
O
Left-channel negative output in BTL mode and high impedance in SE mode
PC-BEEP
14
I
The input for PC Beep mode. PC-BEEP is enabled when a > 1-V (peak-to-peak) square wave is input to
PC-BEEP or PCB ENABLE is high.
7, 18
I
Power supply for output stage
20
I
Right channel headphone input, selected when SE/BTL is held high
RIN
8
I
Common right input for fully differential input. AC ground for single-ended inputs
RLINEIN
23
I
Right-channel line input, selected when SE/BTL is held low.
ROUT+
21
O
Right-channel positive output in BTL mode and positive in SE mode
ROUT–
16
O
Right-channel negative output in BTL mode and high impedance in SE mode
SE/BTL
15
I
Input MUX control input. When this terminal is held high, the LHPIN or RHPIN and SE output is selected.
When this terminal is held low, the LLINEIN or RLINEIN and BTL output are selected.
SHUTDOWN
22
I
When held low, this terminal places the entire device, except PC-BEEP detect circuitry, in shutdown mode.
UP
2
I
A momentary pulse on this terminal increases the volume level by 2 dB. Holding the terminal low for a period
of time will step the amplifier through the volume levels at a rate determined by the capacitor on the CLK
terminal.
VDD
19
I
Analog VDD input supply. This terminal needs to be isolated from PVDD to achieve highest performance.
PVDD
RHPIN
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
3
TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B – JUNE 1999 – REVISED MARCH 2000
absolute maximum ratings over operating free-air temperature range (unless otherwise noted)†
Supply voltage, VDD . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 V
Input voltage, VI . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to VDD +0.3 V
Continuous total power dissipation . . . . . . . . . . . . . . . . . . . . . internally limited (see Dissipation Rating Table)
Operating free-air temperature range, TA . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . – 40°C to 85°C
Operating junction temperature range, TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . – 40°C to 150°C
Storage temperature range, Tstg . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 260°C
† Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE
PWP
TA ≤ 25°C
2.7 W‡
DERATING FACTOR
21.8 mW/°C
TA = 70°C
1.7 W
TA = 85°C
1.4 W
‡ Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report
(literature number SLMA002), for more information on the PowerPAD package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.
recommended operating conditions
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
Supply voltage, VDD
High level input voltage,
High-level
voltage VIH
MIN
MAX
4.5
5.5
SE/BTL
4
SHUTDOWN
2
SE/BTL
UNIT
V
V
3
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
Low level input voltage,
Low-level
voltage VIL
SHUTDOWN
Operating free-air temperature, TA
4
0.8
– 40
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
85
V
°C
TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B – JUNE 1999 – REVISED MARCH 2000
electrical characteristics at specified free-air temperature, VDD = 5 V, TA = 25°C (unless otherwise
noted)
ÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
Á
ÁÁ
ÁÁÁ
ÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁ
ÁÁÁÁÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁ
ÁÁÁÁÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁ
ÁÁÁÁÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁ
ÁÁÁÁÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁ
ÁÁÁÁÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁ
ÁÁÁÁÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁ
ÁÁÁÁÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁ
ÁÁÁÁÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
ÁÁÁ
PARAMETER
TEST CONDITIONS
|VOO|
Output offset voltage (measured differentially)
PSRR
Power supply rejection ratio
MIN
VI = 0, AV = 2
VDD = 4.9 V to 5.1 V
|IIH|
High-level input current
VDD = 5.5 V,
VI = VDD
|IIL|
Low-level input current
VDD = 5.5 V,
VI = 0 V
IDD
Supply current
IDD(SD)
Supply current, shutdown mode
TYP
MAX
UNIT
25
mV
67
BTL mode
20
SE mode
10
150
dB
900
nA
900
nA
mA
µA
300
operating characteristics, VDD = 5 V, TA = 25°C, RL = 4 Ω, Gain = 2 V/V, BTL mode (unless otherwise
noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
PO
THD + N
Output power
THD = 1%,
f = 1 kHz
Total harmonic distortion plus noise
Maximum output power bandwidth
PO = 1 W,
THD = 5%
f = 20 Hz to 15 kHz
BOM
Supply ripple rejection ratio
f = 1 kHz,,
CB = 0.47 µF
BTL mode
65
SE mode
60
CB = 0.47 µF,
µ
f = 20 Hz to 20 kHz
BTL mode
17
Noise output voltage
SE mode
44
Vn
MAX
UNIT
2
W
0.22%
>15
kHz
dB
µVRMS
TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
vs Output power
THD+N
Vn
SNR
vs Gain
Total harmonic distortion plus noise
vs Frequency
2
3, 5, 7, 9, 11
vs Output voltage
12
Output noise voltage
vs Bandwidth
13
Supply ripple rejection ratio
vs Frequency
14, 15
Crosstalk
vs Frequency
16, 17, 18
Shutdown attenuation
vs Frequency
19
Signal-to-noise ratio
vs Bandwidth
Closed loop respone
PO
1, 4, 6, 8, 10
Output power
PD
Power dissipation
ZI
Input impedance
20
21, 22
POST OFFICE BOX 655303
vs Load resistance
23, 24
vs Output power
25, 26
vs Ambient temperature
27
vs Gain
28
• DALLAS, TEXAS 75265
5
TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B – JUNE 1999 – REVISED MARCH 2000
TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
VOLTAGE GAIN
1%
RL = 4 Ω
1%
RL = 8 Ω
RL = 3 Ω
0.1%
AV = +20 to 4 dB
f = 1 kHz
BTL
0.01%
0.5 0.75
1
1.25 1.5 1.75
2
2.25 2.5 2.75
PO = 1 W for AV≥6dB
VO = 1 VRMS for AV≤4 dB
RL = 8 Ω
BTL
THD+N –Total Harmonic Distortion + Noise
THD+N –Total Harmonic Distortion + Noise
10%
3
0.1%
0.01%
–40
–30
PO – Output Power – W
Figure 1
PO = 1 W
0.1%
PO = 1.75 W
1k
10k 20k
THD+N –Total Harmonic Distortion + Noise
THD+N –Total Harmonic Distortion + Noise
PO = 0.5 W
100
10
20
RL = 3 Ω
AV = +20 to +4 dB
BTL
f = 20 kHz
1%
f = 1 kHz
0.1%
0.01%
0.01
f – Frequency – Hz
Figure 3
6
0
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
10%
RL = 3 Ω
AV = +20 to 0 dB
BTL
1%
0.01%
20
–10
Figure 2
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
10%
–20
A V - Voltage Gain - dB
f = 20 Hz
0.1
1
PO – Output Power – W
Figure 4
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
10
TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B – JUNE 1999 – REVISED MARCH 2000
TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
10%
RL = 4 Ω
AV = +20 to +4 dB
BTL
1%
PO= 0.25 W
0.1%
PO=1.5 W
PO= 1 W
THD+N –Total Harmonic Distortion + Noise
THD+N –Total Harmonic Distortion + Noise
10%
0.01%
20
100
1k
f – Frequency – Hz
RL = 4 Ω
AV = +20 to +4 dB
BTL
1%
f = 20 kHz
f = 1 kHz
0.1%
f = 20 Hz
0.01%
0.01
10k 20k
0.1
1
PO – Output Power – W
Figure 5
Figure 6
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
10%
RL = 8 Ω
AV = +20 to +4 dB
BTL
THD+N –Total Harmonic Distortion + Noise
THD+N –Total Harmonic Distortion + Noise
10%
1%
P0 = 0.25 W
0.1%
PO = 0.5 W
P0 = 1 W
0.01%
20
10
100
1k
10k 20k
RL = 8 Ω
AV = +20 to +4 dB
BTL
1%
f = 20 kHz
0.1%
f = 1 kHz
f = 20 Hz
0.01%
0.01
f – Frequency – Hz
Figure 7
0.1
1
PO – Output Power – W
10
Figure 8
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
7
TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B – JUNE 1999 – REVISED MARCH 2000
TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
10%
RL = 32 Ω
AV = +14 to +4 dB
SE
THD+N –Total Harmonic Distortion + Noise
THD+N –Total Harmonic Distortion + Noise
10%
1%
0.1%
PO = 25 mW
0.01%
PO = 75 mW
PO = 50 mW
0.001%
20
100
1k
f – Frequency – Hz
RL = 32 Ω
AV = +14 to +4 dB
SE
1%
f = 20 kHz
0.1%
f = 1 kHz
0.01%
0.01
10k 20k
f = 20 Hz
0.1
PO – Output Power – W
Figure 9
Figure 10
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT VOLTAGE
10%
RL = 10 kΩ
AV = +14 to 0 dB
SE
THD+N –Total Harmonic Distortion + Noise
THD+N –Total Harmonic Distortion + Noise
10%
1%
0.1%
VO = 1 VRMS
0.01%
0.001%
20
100
1k
10k 20k
1%
f = 20 kHz
0.1%
f = 1 kHz
0.01%
RL = 10 kΩ
AV = +14 to +4 dB
SE
0.001%
0.2 0.4
f – Frequency – Hz
0.6
0.8
f = 20 Hz
1
1.2
Figure 12
POST OFFICE BOX 655303
1.4
1.6
VO – Output Voltage – VRMS
Figure 11
8
1
• DALLAS, TEXAS 75265
1.8
2
TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B – JUNE 1999 – REVISED MARCH 2000
TYPICAL CHARACTERISTICS
OUTPUT NOISE VOLTAGE
vs
BANDWIDTH
SUPPLY RIPPLE REJECTION RATIO
vs
FREQUENCY
0
RL = 8 Ω
CB = 0.47 µF
BTL
VDD = 5 V
RL = 4 Ω
140
Supply Ripple Rejection Ratio – dB
Vn – Output Noise Voltage – µV RMS
160
120
100
80
60
AV = +20 dB
40
AV = +6 dB
20
0
0
100
1k
BW – Bandwidth – Hz
–20
AV = +20 dB
–40
–60
–80
AV = +6 dB
–100
–120
10k 20k
20
100
Figure 13
1k
f – Frequency – Hz
Figure 14
SUPPLY RIPPLE REJECTION RATIO
vs
FREQUENCY
CROSSTALK
vs
FREQUENCY
0
–40
RL = 32 Ω
CB = 0.47 µF
SE
–20
–50
–60
–40
Crosstalk – dB
Supply Ripple Rejection Ratio – dB
10k 20k
AV = 0 dB
–60
PO = 1 W
RL = 8 Ω
AV = +20 dB
BTL
–70
LEFT TO RIGHT
–80
–90
–80
RIGHT TO LEFT
AV = +14 dB
–100
–100
–110
–120
20
100
1k
f – Frequency – Hz
10k 20k
–120
20
100
1k
10k 20k
f – Frequency – Hz
Figure 15
Figure 16
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
9
TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B – JUNE 1999 – REVISED MARCH 2000
TYPICAL CHARACTERISTICS
CROSSTALK
vs
FREQUENCY
CROSSTALK
vs
FREQUENCY
–40
Crosstalk – dB
–60
–60
–70
LEFT TO RIGHT
–80
–90
VO = 1 VRMS
RL = 10 kΩ
AV = +6 dB
SE
–50
Crosstalk – dB
–50
–40
PO = 1 W
RL = 8 Ω
AV = +6 dB
BTL
RIGHT TO LEFT
–70
LEFT TO RIGHT
–80
–90
RIGHT TO LEFT
–100
–100
–110
–110
–120
20
100
1k
–120
20
10k 20k
100
f – Frequency – Hz
Figure 17
Figure 18
SHUTDOWN ATTENUATION
vs
FREQUENCY
120
VI = 1 VRMS
SNR – Signal-To-Noise Ratio – dB
Shutdown Attenuation – dB
PO = 1 W
RL = 8 Ω
BTL
115
–20
RL = 10 kΩ, SE
–40
–60
RL = 32 Ω, SE
–80
–100
RL = 8 Ω, BTL
110
AV = +20 dB
105
100
95
AV = +6 dB
90
85
80
100
1k
10k 20k
0
f – Frequency – Hz
100
1k
BW – Bandwidth – Hz
Figure 19
10
10k 20k
SIGNAL-TO-NOISE RATIO
vs
BANDWIDTH
0
–120
20
1k
f – Frequency – Hz
Figure 20
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
10k 20k
TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B – JUNE 1999 – REVISED MARCH 2000
TYPICAL CHARACTERISTICS
CLOSED LOOP RESPONSE
180°
30
25
RL = 8 Ω
AV = +20 dB
BTL
Gain
90°
15
Phase
0°
10
Phase
Gain – dB
20
5
–90°
0
–5
–10
10
–180°
100
1k
10k
100k
1M
f – Frequency – Hz
Figure 21
CLOSED LOOP RESPONSE
180°
30
25
RL = 8 Ω
AV = +6 dB
BTL
90°
15
Phase
0°
10
Phase
Gain – dB
20
5
Gain
–90°
0
–5
–10
10
–180°
100
1k
10k
100k
1M
f – Frequency – Hz
Figure 22
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
11
TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B – JUNE 1999 – REVISED MARCH 2000
TYPICAL CHARACTERISTICS
OUTPUT POWER
vs
LOAD RESISTANCE
OUTPUT POWER
vs
LOAD RESISTANCE
3.5
3
AV = +14 to 0 dB
SE
1250
PO – Output Power – mW
PO – Output Power – W
1500
AV = +20 to 0 dB
BTL
2.5
2
10% THD+N
1.5
1
1000
750
500
10% THD+N
250
0.5
1% THD+N
1% THD+N
0
0
8
16
24
32
40
48
RL – Load Resistance – Ω
56
0
64
0
8
Figure 23
0.4
3Ω
1.6
0.35
1.4
PD – Power Dissipation – W
PD – Power Dissipation – W
64
POWER DISSIPATION
vs
OUTPUT POWER
1.8
4Ω
1.2
1
0.8
0.6
8Ω
0.4
0.5
1.5
1
PO – Output Power – W
2
4Ω
0.3
0.25
0.2
8Ω
0.15
0.1
32 Ω
f = 1 kHz
BTL
Each Channel
0.2
f = 1 kHz
BTL
Each Channel
0.05
2.5
0
0
0.1
Figure 25
12
56
Figure 24
POWER DISSIPATION
vs
OUTPUT POWER
0
0
24
32
16
40
48
RL – Load Resistance – Ω
0.4
0.5
0.6
0.2
0.3
PO – Output Power – W
Figure 26
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
0.7
0.8
TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B – JUNE 1999 – REVISED MARCH 2000
APPLICATION INFORMATION
POWER DISSIPATION
vs
AMBIENT TEMPERATURE
7
ΘJA1 = 45.9°C/W
ΘJA2 = 45.2°C/W
ΘJA3 = 31.2°C/W
ΘJA4 = 18.6°C/W
ΘJA4
PD – Power Dissipation – W
6
5
4
ΘJA3
3
ΘJA1,2
2
1
0
–40 –20
20 40 60 80 100 120 140 160
0
TA – Ambient Temperature – °C
Figure 27
INPUT IMPEDANCE
vs
GAIN
90
ZI – Input Impedance – kΩ
80
70
60
50
40
30
20
10
–40
–30
–20
–10
0
AV – Gain – dB
10
20
Figure 28
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
13
TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B – JUNE 1999 – REVISED MARCH 2000
APPLICATION INFORMATION
VDD
0.47 µF
UP
100 kΩ
1
100 kΩ
2
L OUT+
3
DOWN
4
0.47 µF
LLINE
0.47 µF
L HP
5
6
7
0.47 µF
8
9
0.47 µF
10
L OUT–
11
0.47 µF
12
GND
GND
UP
RLINEIN
DOWN
SHUTDOWN
LOUT+
ROUT+
LLINEIN
RHPIN
LHPIN
VDD
PVDD
PVDD
RIN
CLK
LOUT–
ROUT–
LIN
SE/BTL
BYPASS
PC-BEEP
GND
GND
R LINE
24
23
22
Shutdown
21
20
R OUT+
0.47 µF
R HP
19
0.1 µF
18
17
47 nF
0.1 µF
VDD
10 µF
16
R OUT–
15
SE/BTL
14
13
GND
PC-BEEP
0.47 µF
Figure 29. Typical TPA0162 Application Circuit
selection of components
Figure 30 and Figure 31 are a schematic diagrams of typical notebook computer application circuits.
14
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B – JUNE 1999 – REVISED MARCH 2000
APPLICATION INFORMATION
Right CIRHP
Head– 0.47 µF
phone
Input
Signal
20
CIRLINE
Right 0.47 µF
23
Line
Input
8
Signal
CRIN
0.47 µF
PC BEEP
14
Input
Signal CPCB
0.47 µF 17
RHPIN
RLINEIN
R
MUX
RIN
PC-BEEP
–
+
ROUT+
21
PCBeep
CLK
Up
CCLK
47 nF
COUTR
330 µF
2
3
100
kΩ
15
UP
–
+
Gain/
MUX
SE/BTL Control
DOWN
ROUT–
16
VDD
100 kΩ
VDD
100
kΩ
Down
Depop
Circuitry
Power
Management
Left CILHP
Head– 0.47 µF
phone
Input
Signal
CILLINE
Left 0.47 µF
Line
Input
Signal
6
LHPIN
5
LLINEIN
10
1 kΩ
PVDD
18
VDD
19
BYPASS
SHUTDOWN
11
GND
L
MUX
See Note A
VDD
CSR
0.1 µF
VDD
CSR
0.1 µF
22
CBYP
0.47 µF
To
System
Control
–
+
LOUT+
4
–
+
LOUT–
9
1 kΩ
1,12,
13,24
COUTR
330 µF
LIN
CLIN
0.47 µF
100 kΩ
NOTE A: A 0.1 µF ceramic capacitor should be placed as close as possible to the IC. For filtering lower-frequency noise signals, a larger
electrolytic capacitor of 10 µF or greater should be placed near the audio power amplifier.
Figure 30. Typical TPA0162 Application Circuit Using Single-Ended Inputs and Input MUX
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
15
TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B – JUNE 1999 – REVISED MARCH 2000
APPLICATION INFORMATION
N/C
Right
Negative
Differential
Input
20
Signal CCRIN–
0.47 µF
23
Right CRIN+
Positive 0.47 µF
Differential
8
Input
Signal
PC BEEP
14
Input
Signal CPCB
0.47 µF 17
RHPIN
RLINEIN
R
MUX
–
+
RIN
PC-BEEP
Up
100
kΩ
15
CLK
COUTR
330 µF
–
+
UP
Gain/
DOWN
MUX
SE/BTL Control
ROUT–
100
kΩ
Down
Depop
Circuitry
Power
Management
6
5
10
LHPIN
LLINEIN
16
VDD
1 kΩ
100 kΩ
VDD
Left CIlHP
Head– 0.47 µF
phone
Input
Signal
CILLINE
Left 0.47 µF
Line
Input
Signal
21
PCBeep
CCLK
47 nF
2
3
ROUT+
PVDD
18
VDD
19
BYPASS
SHUTDOWN
11
GND
L
MUX
See Note A
VDD
CSR
0.1 µF
VDD
CSR
0.1 µF
22
To
System
Control
–
+
LOUT+
4
–
+
LOUT–
9
CBYP
0.47 µF
1 kΩ
1,12,
13,24
COUTR
330 µF
LIN
CLIN
0.47 µF
100 kΩ
NOTE A: A 0.1 µF ceramic capacitor should be placed as close as possible to the IC. For filtering lower-frequency noise signals, a larger
electrolytic capacitor of 10 µF or greater should be placed near the audio power amplifier.
Figure 31. Typical TPA0162 Application Circuit Using Differential Inputs
16
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B – JUNE 1999 – REVISED MARCH 2000
APPLICATION INFORMATION
input resistance
Each gain setting is achieved by varying the input resistance of the amplifier, which can range from its smallest
value to over 6 times that value. As a result, if a single capacitor is used in the input high pass filter, the –3 dB
or cut-off frequency will also change by over 6 times. If an additional resistor is connected from the input pin
of the amplifier to ground, as shown in the figure below, the variation of the cut-off frequency will be much
reduced.
Rf
C
RI
IN
Input Signal
R
The input resistance at each gain setting is given in Figure 28.
The –3 dB frequency can be calculated using equation 1:
f
–3 dB
+
ǒø Ǔ
1
2p C R R
(1)
I
If the filter must be more accurate, the value of the capacitor should be increased while value of the resistor to
ground should be decreased. In addition, the order of the filter could be increased.
input capacitor, CI
In the typical application an input capacitor, CI, is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, CI and the input impedance of the amplifier, ZI, form a
high-pass filter with the corner frequency determined in equation 2.
–3 dB
f
c(highpass)
+ 2 p Z1 C
(2)
I I
fc
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
17
TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B – JUNE 1999 – REVISED MARCH 2000
APPLICATION INFORMATION
input capacitor, CI (continued)
The value of CI is important to consider as it directly affects the bass (low frequency) performance of the circuit.
Consider the example where ZI is 710 kΩ and the specification calls for a flat bass response down to 40 Hz.
Equation 2 is reconfigured as equation 3.
C
I
+ 2 p 1Z fc
(3)
I
In this example, CI is 5.6 nF so one would likely choose a value in the range of 5.6 nF to 1 µF. A further
consideration for this capacitor is the leakage path from the input source through the input network (CI) and the
feedback network to the load. This leakage current creates a dc offset voltage at the input to the amplifier that
reduces useful headroom, especially in high gain applications. For this reason a low-leakage tantalum or
ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor
should face the amplifier input in most applications as the dc level there is held at VDD/2, which is likely higher
that the source dc level. Note that it is important to confirm the capacitor polarity in the application.
power supply decoupling, CS
The TPA0162 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling
to ensure the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also
prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is
achieved by using two capacitors of different types that target different types of noise on the power supply leads.
For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance
(ESR) ceramic capacitor, typically 0.1 µF placed as close as possible to the device VDD lead works best. For
filtering lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 µF or greater placed near
the audio power amplifier is recommended.
midrail bypass capacitor, CBYP
The midrail bypass capacitor, CBYP, is the most critical capacitor and serves several important functions. During
startup or recovery from shutdown mode, CBYP determines the rate at which the amplifier starts up. The second
function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This
noise is from the midrail generation circuit internal to the amplifier, which appears as degraded PSRR and
THD+N.
Bypass capacitor, CBYP, values of 0.47 µF to 1 µF ceramic or tantalum low-ESR capacitors are recommended
for the best THD and noise performance.
output coupling capacitor, CC
In the typical single-supply SE configuration, an output coupling capacitor (CC) is required to block the dc bias
at the output of the amplifier thus preventing dc currents in the load. As with the input coupling capacitor, the
output coupling capacitor and impedance of the load form a high-pass filter governed by equation 4.
–3 dB
f
c(high)
+ 2 p R1 C
(4)
L C
fc
18
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B – JUNE 1999 – REVISED MARCH 2000
APPLICATION INFORMATION
output coupling capacitor, CC (continued)
The main disadvantage, from a performance standpoint, is the load impedances are typically small, which drives
the low-frequency corner higher degrading the bass response. Large values of CC are required to pass low
frequencies into the load. Consider the example where a CC of 330 µF is chosen and loads vary from 3 Ω,
4 Ω, 8 Ω, 32 Ω, 10 kΩ, and 47 kΩ. Table 2 summarizes the frequency response characteristics of each
configuration.
Table 1. Common Load Impedances Vs Low Frequency Output Characteristics in SE Mode
RL
CC
Lowest Frequency
3Ω
330 µF
161 Hz
4Ω
330 µF
120 Hz
8Ω
330 µF
60 Hz
32 Ω
330 µF
15 Hz
10,000 Ω
330 µF
0.05 Hz
47,000 Ω
330 µF
0.01 Hz
As Table 1 indicates, most of the bass response is attenuated into a 4-Ω load, an 8-Ω load is adequate,
headphone response is good, and drive into line level inputs (a home stereo for example) is exceptional.
using low-ESR capacitors
Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal)
capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this
resistance the more the real capacitor behaves like an ideal capacitor.
bridged-tied load versus single-ended mode
Figure 32 shows a class-AB audio power amplifier (APA) in a BTL configuration. The TPA0162 BTL amplifier
consists of two class-AB amplifiers driving both ends of the load. There are several potential benefits to this
differential drive configuration but initially consider power to the load. The differential drive to the speaker means
that as one side is slewing up, the other side is slewing down, and vice versa. This in effect doubles the voltage
swing on the load as compared to a ground referenced load. Plugging 2 × VO(PP) into the power equation, where
voltage is squared, yields 4× the output power from the same supply rail and load impedance (see equation 5).
+ O(PP)
(rms)
2 Ǹ2
V
V
Power
+
V
(5)
2
(rms)
R
L
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
19
TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B – JUNE 1999 – REVISED MARCH 2000
APPLICATION INFORMATION
bridged-tied load versus single-ended mode (continued)
VDD
VO(PP)
2x VO(PP)
RL
VDD
–VO(PP)
Figure 32. Bridge-Tied Load Configuration
In a typical computer sound channel operating at 5 V, bridging raises the power into an 8-Ω speaker from a
singled-ended (SE, ground reference) limit of 250 mW to 1 W. In sound power that is a 6-dB improvement —
which is loudness that can be heard. In addition to increased power there are frequency response concerns.
Consider the single-supply SE configuration shown in Figure 33. A coupling capacitor is required to block the
dc offset voltage from reaching the load. These capacitors can be quite large (approximately 33 µF to 1000 µF)
so they tend to be expensive, heavy, occupy valuable PCB area, and have the additional drawback of limiting
low-frequency performance of the system. This frequency limiting effect is due to the high pass filter network
created with the speaker impedance and the coupling capacitance and is calculated with equation 6.
fc
+ 2 p R1 C
(6)
L C
For example, a 68-µF capacitor with an 8-Ω speaker would attenuate low frequencies below 293 Hz. The BTL
configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency
performance is then limited only by the input network and speaker response. Cost and PCB space are also
minimized by eliminating the bulky coupling capacitor.
VDD
–3 dB
VO(PP)
CC
RL
VO(PP)
fc
Figure 33. Single-Ended Configuration and Frequency Response
20
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B – JUNE 1999 – REVISED MARCH 2000
APPLICATION INFORMATION
bridged-tied load versus single-ended mode (continued)
Increasing power to the load does carry a penalty of increased internal power dissipation. The increased
dissipation is understandable considering that the BTL configuration produces 4× the output power of the SE
configuration. Internal dissipation versus output power is discussed further in the crest factor section.
single-ended operation
In SE mode (see Figure 32 and Figure 33), the load is driven from the primary amplifier output for each channel
(OUT+, terminals 21 and 4).
The amplifier switches single-ended operation when the SE/BTL terminal is held high. This puts the negative
outputs in a high-impedance state, and reduces the amplifier’s gain to 1 V/V.
BTL amplifier efficiency
Class-AB amplifiers are notoriously inefficient. The primary cause of these inefficiencies is voltage drop across
the output stage transistors. There are two components of the internal voltage drop. One is the headroom or
dc voltage drop that varies inversely to output power. The second component is due to the sinewave nature of
the output. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from
VDD. The internal voltage drop multiplied by the RMS value of the supply current, IDDrms, determines the internal
power dissipation of the amplifier.
An easy-to-use equation to calculate efficiency starts out as being equal to the ratio of power from the power
supply to the power delivered to the load. To accurately calculate the RMS and average values of power in the
load and in the amplifier, the current and voltage waveform shapes must first be understood (see Figure 34).
IDD
VO
IDD(avg)
V(LRMS)
Figure 34. Voltage and Current Waveforms for BTL Amplifiers
Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very
different between SE and BTL configurations. In an SE application the current waveform is a half-wave rectified
shape whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different.
Keep in mind that for most of the waveform both the push and pull transistors are not on at the same time, which
supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform.
The following equations are the basis for calculating amplifier efficiency.
Efficiency of a BTL amplifier
+ P PL
(7)
SUP
Where:
P
+
L
and P SUP
V rms 2
L
R
, and V
L
+ Ǹ2P ,
LRMS
+ VDD IDDavg
V
therefore, P
and I DDavg
+ 1p
ŕ
POST OFFICE BOX 655303
V
P
+
L
2R
2
L
p V
P sin(t) dt
R
L
0
• DALLAS, TEXAS 75265
+ p1
P [cos(t)] p
0
R
L
V
+ p2VRP
L
21
TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B – JUNE 1999 – REVISED MARCH 2000
APPLICATION INFORMATION
BTL amplifier efficiency (continued)
Therefore,
P
SUP
+ 2 VpDDR VP
L
substituting PL and PSUP into equation 7,
V
Efficiency of a BTL amplifier
Where:
V
P
+
Ǹ
2 P
Therefore,
h BTL
+
p
L
Ǹ
R
+2 V
L
V
DD P
p RL
+ 4pVVP
DD
L
2 P
4 V
2
P
2R
L
R
L
(8)
DD
PL = Power delivered to load
PSUP = Power drawn from power supply
VLRMS = RMS voltage on BTL load
RL = Load resistance
VP = Peak voltage on BTL load
IDDavg = Average current drawn from the power supply
VDD = Power supply voltage
ηBTL = Efficiency of a BTL amplifier
Table 2 employs equation 4 to calculate efficiencies for four different output power levels. Note that the efficiency
of the amplifier is quite low for lower power levels and rises sharply as power to the load is increased resulting
in a nearly flat internal power dissipation over the normal operating range. Note that the internal dissipation at
full output power is less than in the half power range. Calculating the efficiency for a specific system is the key
to proper power supply design. For a stereo 1-W audio system with 8-Ω loads and a 5-V supply, the maximum
draw on the power supply is almost 3.25 W.
Table 2. Efficiency vs Output Power in 5-V 8-Ω BTL Systems
OUTPUT POWER
(W)
EFFICIENCY
(%)
PEAK VOLTAGE
(V)
INTERNAL DISSIPATION
(W)
0.25
31.4
2.00
0.55
0.50
44.4
2.83
0.62
1.00
62.8
4.00
4.47†
0.59
1.25
70.2
† High peak voltages cause the THD to increase.
0.53
A final point to remember about class-AB amplifiers (either SE or BTL) is how to manipulate the terms in the
efficiency equation to utmost advantage when possible. Note that in equation 8, VDD is in the denominator. This
indicates that as VDD goes down, efficiency goes up.
22
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B – JUNE 1999 – REVISED MARCH 2000
APPLICATION INFORMATION
crest factor and thermal considerations
Class-AB power amplifiers dissipate a significant amount of heat in the package under normal operating
conditions. A typical music CD requires 12 dB to 15 dB of dynamic range, or headroom above the average power
output, to pass the loudest portions of the signal without distortion. In other words, music typically has a crest
factor between 12 dB and 15 dB. When determining the optimal ambient operating temperature the internal
dissipated power at the average output power level must be used. From the TPA0162 data sheet, one can see
that when the TPA0162 is operating from a 5-V supply into a 3-Ω speaker that 4 W peaks are available.
Converting watts to dB:
P
+ 10Log
dB
ǒǓ
P
P
W
ref
ǒ Ǔ
+ 10Log 4W
+ 6 dB
1W
(9)
Subtracting the headroom restriction to obtain the average listening level without distortion yields:
6 dB – 15 dB = –9 dB (15 dB crest factor)
6 dB – 12 dB = –6 dB (12 dB crest factor)
6 dB – 9 dB = –3 dB (9 dB crest factor)
6 dB – 6 dB = 0 dB (6 dB crest factor)
6 dB – 3 dB = 3 dB (3 dB crest factor)
Converting dB back into watts:
P
W
+ 10PdBń10 Pref
+ 63 mW (18 dB crest factor)
+ 125 mW (15 dB crest factor)
+ 250 mW (9 dB crest factor)
+ 500 mW (6 dB crest factor)
+ 1000 mW (3 dB crest factor)
+ 2000 mW (15 dB crest factor)
(10)
This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 2 W of continuous power output with a 3 dB crest
factor, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings for the
system. Using the power dissipation curves for a 5-V, 3-Ω system, the internal dissipation in the TPA0162 and
maximum ambient temperatures is shown in Table 3.
Table 3. TPA0162 Power Rating, 5-V, 3-Ω, Stereo
PEAK OUTPUT POWER
(W)
AVERAGE OUTPUT POWER
POWER DISSIPATION
(W/Channel)
MAXIMUM AMBIENT
TEMPERATURE
4
2 W (3 dB)
1.7
– 3°C
4
1000 mW (6 dB)
1.6
6°C
4
500 mW (9 dB)
1.4
24°C
4
250 mW (12 dB)
1.1
51°C
4
125 mW (15 dB)
0.8
78°C
4
63 mW (18 dB)
0.6
96°C
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
23
TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B – JUNE 1999 – REVISED MARCH 2000
APPLICATION INFORMATION
crest factor and thermal considerations (continued)
Table 4. TPA0162 Power Rating, 5-V, 8-Ω, Stereo
PEAK OUTPUT POWER
AVERAGE OUTPUT POWER
POWER DISSIPATION
(W/Channel)
MAXIMUM AMBIENT
TEMPERATURE
2.5 W
1250 mW (3 dB crest factor)
0.55
100°C
2.5 W
1000 mW (4 dB crest factor)
0.62
94°C
2.5 W
500 mW (7 dB crest factor)
0.59
97°C
2.5 W
250 mW (10 dB crest factor)
0.53
102°C
The maximum dissipated power, PDmax, is reached at a much lower output power level for an 8 Ω load than for
a 3 Ω load. As a result, this simple formula for calculating PDmax may be used for an 8 Ω application:
P
Dmax
+
2V 2
DD
p2R L
(11)
However, in the case of a 3 Ω load, the PDmax occurs at a point well above the normal operating power level.
The amplifier may therefore be operated at a higher ambient temperature than required by the PDmax formula
for a 3 Ω load.
The maximum ambient temperature depends on the heat sinking ability of the PCB system. The derating factor
for the PWP package is shown in the dissipation rating table on page 4. Converting this to ΘJA:
1 + 45°CńW
+ Derating1 Factor + 0.022
Θ JA
(12)
To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are
per channel so the dissipated power needs to be doubled for two channel operation. Given ΘJA, the maximum
allowable junction temperature, and the total internal dissipation, the maximum ambient temperature can be
calculated with the following equation. The maximum recommended junction temperature for the TPA0162 is
150°C. The internal dissipation figures are taken from the Power Dissipation vs Output Power graphs.
T
A
Max
+ TJ Max * ΘJA PD
+ 150 * 45 (0.6 2) + 96°C (15 dB crest factor)
(13)
NOTE:
Internal dissipation of 0.6 W is estimated for a 2-W system with 15 dB crest factor per channel.
Tables 3 and 4 show that for some applications no airflow is required to keep junction temperatures in the
specified range. The TPA0162 is designed with thermal protection that turns the device off when the junction
temperature surpasses 150°C to prevent damage to the IC. Tables 3 and 4 were calculated for maximum
listening volume without distortion. When the output level is reduced the numbers in the table change
significantly. Also, using 8-Ω speakers dramatically increases the thermal performance by increasing amplifier
efficiency.
24
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B – JUNE 1999 – REVISED MARCH 2000
APPLICATION INFORMATION
SE/BTL operation
The ability of the TPA0162 to easily switch between BTL and SE modes is one of its most important cost saving
features. This feature eliminates the requirement for an additional headphone amplifier in applications where
internal stereo speakers are driven in BTL mode but external headphone or speakers must be accommodated.
Internal to the TPA0162, two separate amplifiers drive OUT+ and OUT–. The SE/BTL input (terminal 15)
controls the operation of the follower amplifier that drives LOUT– and ROUT– (terminals 9 and 16). When
SE/BTL is held low, the amplifier is on and the TPA0162 is in the BTL mode. When SE/BTL is held high, the OUT–
amplifiers are in a high output impedance state, which configures the TPA0162 as an SE driver from LOUT+
and ROUT+ (terminals 4 and 21). IDD is reduced by approximately one-half in SE mode. Control of the SE/BTL
input can be from a logic-level CMOS source or, more typically, from a resistor divider network as shown in
Figure 35.
20
23
RHPIN
RLINEIN
R
MUX
–
+
8
ROUT+
21
RIN
COUTR
330 µF
–
+
ROUT–
VDD
16
1 kΩ
100 kΩ
SE/BTL
15
100 kΩ
Figure 35. TPA0162 Resistor Divider Network Circuit
Using a readily available 1/8-in. (3.5 mm) stereo headphone jack, the control switch is closed when no plug is
inserted. When closed the 100-kΩ/1-kΩ divider pulls the SE/BTL input low. When a plug is inserted, the 1-kΩ
resistor is disconnected and the SE/BTL input is pulled high. When the input goes high, the OUT– amplifier is
shut-down causing the speaker to mute (virtually open-circuits the speaker). The OUT+ amplifier then drives
through the output capacitor (CO) into the headphone jack.
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
25
TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B – JUNE 1999 – REVISED MARCH 2000
APPLICATION INFORMATION
PC BEEP operation
The PC BEEP input allows a system beep to be sent directly from a computer through the amplifier to the
speakers with few external components. The input is normally activated automatically, but may be selected
manually by pulling PCB ENABLE high. When the PC BEEP input is active, both of the LINEIN and HPIN inputs
are deselected and both the left and right channels are driven in BTL mode with the signal from PC BEEP. The
gain from the PC BEEP input to the speakers is fixed at 0.3 V/V and is independent of the volume setting. When
the PC BEEP input is deselected, the amplifier will return to the previous operating mode and volume setting.
Furthermore, if the amplifier is in shutdown mode, activating PC BEEP will take the device out of shutdown and
output the PC BEEP signal, then return the amplifier to shutdown mode.
When PCB ENABLE is held low, the amplifier will automatically switch to PC BEEP mode after detecting a valid
signal at the PC BEEP input. The preferred input signal is a square wave or pulse train with an amplitude of 1
Vpp or greater. To be a accurately detected, the signal must have a minimum of 1 Vpp amplitude, rise and fall
times of less than 0.1 µs and a minimum of 8 rising edges. When the signal is no longer detected, the amplifier
will return to its previous operating mode and volume setting.
When PCB ENABLE is held high, PC BEEP is selected and the LINEIN and HPIN inputs are deactivated
regardless of the input signal. PCB ENABLE has an internal 100 kΩ pulldown resistor and will trip at
approximately VDD/2.
If it is desired to ac couple the PC BEEP input, the value of the coupling capacitor should be chosen to satisfy
the following equation:
C
PCB
w 2p f
PCB
1
(100 k )
W
(14)
The PC BEEP input can also be dc coupled to avoid using this coupling capacitor. The pin normally sits at midrail
when no signal is present.
26
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B – JUNE 1999 – REVISED MARCH 2000
APPLICATION INFORMATION
Input MUX operation
Right
Headphone
Input
Signal
CIRHP
0.47 µF
CIRLINE
0.47 µF
20
RHPIN
23
RLINEIN
R
MUX
Right Line
Input
Signal
8
–
+
ROUT+
21
–
+
ROUT–
16
RIN
CRIN
0.47 µF
Figure 36. TPA0162 Example Input MUX Circuit
Another advantage of using the MUX feature is setting the gain of the headphone channel to –1. This provides
the optimum distortion performance into the headphones where clear sound is more important. Refer to the
SE/BTL operation section for a description of the headphone jack control circuit.
shutdown modes
The TPA0162 employs a shutdown mode of operation designed to reduce supply current, IDD, to the absolute
minimum level during periods of nonuse for battery-power conservation. The SHUTDOWN input terminal
should be held high during normal operation when the amplifier is in use. Pulling SHUTDOWN low causes the
outputs to mute and the amplifier to enter a low-current state, IDD = 150 µA. SHUTDOWN should never be left
unconnected because amplifier operation would be unpredictable.
Table 5. Shutdown and Mute Mode Functions
INPUTS†
AMPLIFIER STATE
SE/BTL
SHUTDOWN
INPUT
OUTPUT
Low
High
Line
BTL
X
Low
X
Mute
High
High
HP
SE
† Inputs should never be left unconnected.
X = do not care
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
27
TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B – JUNE 1999 – REVISED MARCH 2000
MECHANICAL DATA
PWP (R-PDSO-G**)
PowerPAD PLASTIC SMALL-OUTLINE
20 PINS SHOWN
0,30
0,19
0,65
20
0,10 M
11
Thermal Pad
(See Note D)
4,50
4,30
0,15 NOM
6,60
6,20
Gage Plane
1
10
0,25
A
0°– 8°
0,75
0,50
Seating Plane
0,15
0,05
1,20 MAX
PINS **
0,10
14
16
20
24
28
A MAX
5,10
5,10
6,60
7,90
9,80
A MIN
4,90
4,90
6,40
7,70
9,60
DIM
4073225/F 10/98
NOTES: A.
B.
C.
D.
All linear dimensions are in millimeters.
This drawing is subject to change without notice.
Body dimensions do not include mold flash or protrusions.
The package thermal performance may be enhanced by bonding the thermal pad to an external thermal plane.
This pad is electrically and thermally connected to the backside of the die and possibly selected leads.
E. Falls within JEDEC MO-153
PowerPAD is a trademark of Texas Instruments Incorporated.
28
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
IMPORTANT NOTICE
Texas Instruments and its subsidiaries (TI) reserve the right to make changes to their products or to discontinue
any product or service without notice, and advise customers to obtain the latest version of relevant information
to verify, before placing orders, that information being relied on is current and complete. All products are sold
subject to the terms and conditions of sale supplied at the time of order acknowledgement, including those
pertaining to warranty, patent infringement, and limitation of liability.
TI warrants performance of its semiconductor products to the specifications applicable at the time of sale in
accordance with TI’s standard warranty. Testing and other quality control techniques are utilized to the extent
TI deems necessary to support this warranty. Specific testing of all parameters of each device is not necessarily
performed, except those mandated by government requirements.
CERTAIN APPLICATIONS USING SEMICONDUCTOR PRODUCTS MAY INVOLVE POTENTIAL RISKS OF
DEATH, PERSONAL INJURY, OR SEVERE PROPERTY OR ENVIRONMENTAL DAMAGE (“CRITICAL
APPLICATIONS”). TI SEMICONDUCTOR PRODUCTS ARE NOT DESIGNED, AUTHORIZED, OR
WARRANTED TO BE SUITABLE FOR USE IN LIFE-SUPPORT DEVICES OR SYSTEMS OR OTHER
CRITICAL APPLICATIONS. INCLUSION OF TI PRODUCTS IN SUCH APPLICATIONS IS UNDERSTOOD TO
BE FULLY AT THE CUSTOMER’S RISK.
In order to minimize risks associated with the customer’s applications, adequate design and operating
safeguards must be provided by the customer to minimize inherent or procedural hazards.
TI assumes no liability for applications assistance or customer product design. TI does not warrant or represent
that any license, either express or implied, is granted under any patent right, copyright, mask work right, or other
intellectual property right of TI covering or relating to any combination, machine, or process in which such
semiconductor products or services might be or are used. TI’s publication of information regarding any third
party’s products or services does not constitute TI’s approval, warranty or endorsement thereof.
Copyright  2000, Texas Instruments Incorporated