NSC LM3211MTX-ADJ

LM3211
Step-up PWM DC/DC Converter Integrated with 4 Buffers
General Description
Features
The LM3211 is a compact bias solution for TFT displays. It
has a current mode PWM step-up DC/DC converter with a
1.4A, 0.17Ω internal switch. Capable of generating 8V at
300mA from a Lithium Ion battery, the LM3211 is ideal for
generating bias voltages for large screen LCD panels. The
LM3211 can be operated at switching frequencies of 600kHz
or 1.25MHz, allowing for easy filtering and low noise. An
external compensation pin gives the user flexibility in setting
frequency compensation, which makes possible the use of
small, low ESR ceramic capacitors at the output. The
LM3211 uses a patented internal circuitry to limit startup
inrush current of the boost switching regulator without the
use of an external softstart capacitor. An external softstart
pin enables the user to tailor the softstart to a specific
application. The LM3211 contains 4 Gamma buffers capable
of supplying 35mA source and sink. The TSSOP-20 package
ensures a low profile overall solution.
n 1.4A, 0.17Ω, internal power switch
n VIN operating range: 2.2V to 7.5V
n 600kHz/1.25MHz selectable frequency step-up DC/DC
converter
n 20 pin TSSOP package
n Inrush current limiting circuitry
n External softstart override
n 4 Gamma buffers
Applications
n
n
n
n
LCD Bias Supplies
Handheld Devices
Portable Applications
Cellular Phones/Digital Cameras
Typical Application Circuit
20062231
© 2004 National Semiconductor Corporation
DS200622
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LM3211 Step-up PWM DC/DC Converter Integrated with 4 Buffers
February 2004
LM3211
Connection Diagram
Top View
20062204
TJMAX
TSSOP 20 package
= 125˚C, θJA = 120˚C/W (Note 1)
Pin Description
Pin
Name
1
VSW
Power switch input.
Function
2
VIN
Switching Regulator Power input.
3
SHDN
Shutdown pin, active low.
4
FSLCT
5
Vs+
Frequency Select pin. FSLCT = VIN for 1.25 MHz, FSLCT = AGND or floating for 600kHz.
6
GMA1-in
Gamma Buffer input.
7
GMA2-in
Gamma Buffer input.
8
GMA3-in
Gamma Buffer input.
9
GMA4-in
Gamma Buffer input.
10
NC
Gamma Buffer input supply.
No Connection, leave open.
11
NC
12
GMA4-out
Gamma Buffer output.
13
GMA3-out
Gamma Buffer output.
14
GMA2-out
Gamma Buffer output.
15
GMA1-out
Gamma Buffer output.
16
SS
Soft start pin.
17
VC
Boost Compensation Network Connection.
18
FB
Output Voltage Feedback input.
19
AGND
20
GND
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No Connection, leave open.
Gamma Buffer ground, Analog ground connection for Regulator.
Switch Power Ground.
2
LM3211
GMA4-out(Pin 12): Gamma Buffer output pin.
GMA3-out(Pin13): Gamma Buffer output pin.
Pin Functions
VSW(Pin 1): This is the drain of the internal NMOS power
switch. Minimize the metal trace area connected to this pin to
minimize EMI.
VIN(Pin 2): Input Supply Pin. Bypass this pin with a capacitor
as close to the device as possible. The capacitor should
connect between VIN and GND.
GMA2-out(Pin 14): Gamma Buffer output pin.
GMA1-out(Pin 15): Gamma Buffer output pin.
SS(Pin 16): Softstart pin. Connect capacitor to SS pin and
AGND to slowly ramp inductor current on startup.
VC(Pin 17): Compensation Network for Boost switching
regulator. Connect resistor/capacitor network between VC
pin and AGND for boost switching regulator AC compensation.
SHDN(Pin 3): Shutdown Pin. The shutdown pin signal is
active low. A voltage of less than 0.3V disables the device. A
voltage greater than 0.85V enables the device.
FSLCT(Pin 4): Frequency Select Pin. Connecting FSLCT to
AGND selects a 600 kHz operating frequency for the switching regulator. Connecting FSLCT to VIN selects a 1.25 MHz
operating frequency. If FSLCT is left floating, the switching
frequency defaults to 600 kHz.
FB(Pin 18): Feedback pin. Set the output voltage by selecting values of R1 and R2 using:
Vs+(Pin 5): Supply pin for the four Gamma buffers. Bypass
this pin with a capacitor as close to the device as possible.
The capacitor should connect between Vs+ and GND.
Connect the ground of the feedback network to the AGND
plane, which can be tied directly to the GND pin.
GMA1-in(Pin 6):Gamma Buffer input pin.
GMA2-in(Pin 7): Gamma Buffer input pin.
AGND(Pin 19): Analog ground pin. Ground connection for
the Gamma buffers and the boost switching regulator. AGND
must be tied directly to GND at the pins.
GND(Pin 20): Power ground pin. Ground connection for the
NMOS power device of the boost switching regulator. GND
must be tied directly to AGND at the pins.
GMA3-in(Pin 8): Gamma Buffer input pin.
GMA4-in(Pin 9): Gamma Buffer input pin.
NC(Pin 10):No Connection.
NC(Pin 11): No Connection.
Ordering Information
Order Number
Package Type
NSC Package Drawing
LM3211MT-ADJ
TSSOP-20
MTC20
73 Units, Rail
LM3211MTX-ADJ
TSSOP-20
MTC20
2500 Units, Tape and Reel
3
Supplied As
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LM3211
Block Diagrams
20062203
20062251
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4
ESD Ratings
(Note 3)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
VIN
-0.3V to 7.5V
VSW Voltage
-0.3V to 18V
FB Voltage
-0.3V to 7V
VC Voltage
0.965V to 1.565V
SHDN Voltage
Human Body Model
-0.3V to 14V
Buffer Input Voltage
Rail-to-Rail
Buffer Output Voltage
Rail-to-Rail
150V
Operating Conditions
Operating Temperature
−40˚C to +125˚C
Storage Temperature
−65˚C to +150˚C
Supply Voltage, VIN
AGND to VIN
Supply Voltage, Vs+
2kV
Machine Model
-0.3V to VIN
FSLCT Voltage
LM3211
Absolute Maximum Ratings (Note 2)
2.2V to 7.5V
VSW Voltage
17V
Supply Gamma Buffer, Vs+
4V to 14V
Electrical Characteristics
Specifications in standard type face are for TJ = 25˚C and those with boldface type apply over the full Operating Temperature Range ( TJ = −40˚C to +125˚C). Unless otherwise specified, VIN =2.2V and Vs+ = 8V.
Switching Regulator
Symbol
IQ
Parameter
Quiescent Current
Conditions
Min
(Note 4)
1.6
2
1.65
2.2
Switching, FSCLT = 0V
2.5
3
Switching, FSCLT = VIN
3.4
4
6
15
µA
1.265
1.291
V
0.03
0.05
%/V
Feedback Voltage
%VFB/∆VIN
Feedback Voltage Line
Regulation
ICL
Switch Current Limit
(Note 6)
VIN = 2.5V, VOUT = 8V
VIN = 2.7V
Switch RDSON (Note 7)
FB Pin Bias Current(Note 8)
Units
Not Switching, FSCLT = VIN
VFB
RDSON
Max
(Note 4)
Not Switching, FSCLT = 0V
Shutdown mode
IB
Typ
(Note 5)
1.239
1.4
VIN
Input Voltage Range
ISS
Soft Start Current
TSS
Internal Soft Start Ramp
Time
gm
Error Amp Transconductance ∆I = 5µA
A
170
30
2.2
5
FSLCT = 0V
60
mA
mΩ
90
nA
7.5
V
11
15
µA
6.7
12
mS
135
250
µmho
AV
Error Amp Voltage Gain
DMAX
Maximum Duty Cycle
fS
Switching Frequency
IL
Switch Leakage Current
VSW = 17V
SHDN
SHDN Threshold
Output High
Output Low
0.6
0.3
V
I SHDN
Shutdown Pin Current
0V ≤ SHDN ≤ VIN
0.5
1
µA
UVP
On Threshold
1.8
1.9
2
V
Off Threshold
1.7
1.8
1.9
V
135
V/V
78
85
FSLCT = 0V
500
600
700
kHz
FSLCT = VIN
0.9
1.25
1.5
MHz
0.185
20
µA
0.85
Hysteresis
0.6
100
5
%
V
mV
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LM3211
Electrical Characteristics
Specifications in standard type face are for TJ = 25˚C and those with boldface type apply over the full Operating Temperature Range ( TJ = −40˚C to +125˚C). Unless otherwise specified, VIN =2.2V and Vs+ = 8V.
BUFFERS
Symbol
Parameter
Conditions
Min
(Note 4)
Typ
(Note 5)
Max
(Note 4)
10
VOS
Input offset voltage
2.5
∆Vos/∆T
Offset Voltage Drift
8
IB
Input Bias Current
170
CMVR
Input Common-mode Voltage
Range
ZIN
Input Impedance
CIN
Input Capacitance
IOUT
Continuous Output Current
VOUT Swing
0.05
mV
µV/˚C
800
nA
Vs+-0.05
V
400
kΩ
1
pF
Vs+=8V, Source
30
39
47
Vs+=8V, Sink
−45
−37
−27
Vs+=12V, Source
35
48
59
Vs+=12V, Sink
−54
−44
RL=10k, Vo min.
Units
mA
−32
0.075
RL=10k, Vo max.
7.88
RL=2k, Vo min.
0.075
RL=2k, Vo max.
V
7.85
0.994
0.9985
AVCL
Voltage Gain
RL =2 kΩ
RL=10 kΩ
NL
Gain Linearity
RL =2 kΩ, Buffer input=0.5 to
(Vs+-0.5V)
Vs+
Supply Voltage
PSRR
Power Supply Rejection
Ratio
Vs+ = 4 to 12V
Is+
Supply Current/Amplifier
SR
BW
φM
0.998
0.9999
V/V
0.01
%
4
12
V
90
316
µV/V
Vo = Vs+/2, No Load
0.75
1.2
mA
Slew Rate
CL =150pf
5.3
V/µs
Bandwidth
-3dB,RL =10 kΩ, CL =100pf
3
MHz
Phase Margin
CL =100pf
78
Deg˚
Note 1: The maximum allowable power dissipation is a function of the maximum junction temperature, TJ(MAX), the junction-to-ambient thermal resistance, θJA,
and the ambient temperature, TA. See the Electrical Characteristics table for the thermal resistance of various layouts. The maximum allowable power dissipation
at any ambient temperature is calculated using: PD (MAX) = (TJ(MAX) − TA)/θJA. Exceeding the maximum allowable power dissipation will cause excessive die
temperature, and the regulator will go into thermal shutdown.
Note 2: Absolute maximum ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions for which the device is intended to
be functional, but device parameter specifications may not be guaranteed. For guaranteed specifications and test conditions, see the Electrical Characteristics.
Note 3: The human body model is a 100 pF capacitor discharged through a 1.5kΩ resistor into each pin. The machine model is a 200pF capacitor discharged
directly into each pin.
Note 4: All limits guaranteed at room temperature (standard typeface) and at temperature extremes (bold typeface). All room temperature limits are 100%
production tested or guaranteed through statistical analysis. All limits at temperature extremes are guaranteed via correlation using standard Statistical Quality
Control (SQC) methods. All limits are used to calculate Average Outgoing Quality Level (AOQL).
Note 5: Typical numbers are at 25˚C and represent the most likely norm.
Note 6: Duty cycle affects current limit due to ramp generator. See Switch Current Limit vs. VIN and Switch Current Limit vs. Temperature graphs in the Typical
Performance Characteristics section.
Note 7: See Typical Performance Characteristics section for Tri-Temperature data for RDSON vs. VIN.
Note 8: Bias current flows into FB pin.
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6
LM3211
Typical Performance Characteristics
Efficiency vs. Load Current
(VOUT = 8V, fS = 1.25 MHz)
Efficiency vs. Load Current
(VOUT = 8V, fS = 600 kHz)
20062226
20062225
Switch Current Limit vs. Temperature
(VOUT = 8V)
Efficiency vs. Load Current
(VOUT = 10V, fS = 1.25 MHz)
20062220
20062260
RDSON vs. VIN
(ISW = 1A)
Switch Current Limit vs. VIN
20062222
20062227
7
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LM3211
Typical Performance Characteristics
(Continued)
IQ vs. VIN
(600 kHz, not switching)
IQ vs. VIN
(600 kHz, switching)
20062221
20062229
IQ vs. VIN
(1.25 MHz, switching)
IQ vs. VIN
(1.25 MHz, not switching)
20062219
20062221
Frequency vs. VIN
(600 kHz)
IQ vs. VIN
(In shutdown)
20062218
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20062223
8
LM3211
Typical Performance Characteristics
(Continued)
Frequency vs. VIN
(1.25 MHz)
Feedback Pin Current vs. Temperature
20062257
20062224
CSS Pin Current vs. VIN
Load Transient Response
20062276
VOUT = 8V, VIN = 3V, F = 1.25MHz
1) Load, 80mA to 260mA to 80mA
2) IL, 500mA/div, DC
20062258
3) VOUT, 100mV/div, AC
T = 100µs/div
Load Transient Response
Load Transient Response
20062275
20062283
VOUT = 8V, VIN = 3V, F = 600kHz
VOUT = 10V, VIN = 5V, F = 1.25MHz
1) Load, 80mA to 260mA to 80mA
1) Load, 195mA to 385mA to 195mA
2) IL, 500mA/div, DC
2) IL, 500mA/div, DC
3) VOUT, 200mV/div, AC
3) VOUT, 500mV/div, AC
T = 100µs/div
T = 100µs/div
9
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LM3211
Typical Performance Characteristics
(Continued)
Internal Soft Start
Internal Soft Start
20062279
20062277
VOUT = 8V, VIN = 3V, RLOAD = 27Ω, CSS = none, F = 600kHz
VOUT = 8V, VIN = 3V, RLOAD = 27Ω, CSS = none, F = 1.25MHz
1) SHDN, 1V/div, DC
1) SHDN, 1V/div, DC
2) IL, 500mA/div, DC
2) IL, 500mA/div, DC
3) VOUT, 5V/div, DC
T = 1ms/div
3) VOUT, 5V/div, DC
T = 1ms/div
Input Offset Voltage vs. Common Mode Voltage
(3 units)
External Soft Start
20062278
VOUT = 8V, VIN = 3V, RLOAD = 27Ω, CSS = 330nF, F = 1.25MHz
1) SHDN, 1V/div, DC
2) IL, 500mA/div, DC
20062261
3) VOUT, 5V/div, DC
T = 4ms/div
Input Offset Voltage vs. Common Mode Voltage
(Over Temperature)
Input Bias Current vs. Common Mode Voltage
20062263
20062262
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10
LM3211
Typical Performance Characteristics
(Continued)
Output Voltage vs. Output Current
(sinking)
Output Voltage vs. Output Current
(sourcing)
20062264
20062265
Supply Current vs. Common Mode Voltage
Large Signal Step Response
20062268
20062266
Positive Slew Rate vs. Capacitive Load
Negative Slew Rate vs. Capacitive Load
20062269
20062270
11
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LM3211
Typical Performance Characteristics
(Continued)
Phase Margin vs. Capacitive Load
Unity Gain Frequency vs. Capacitive Load
20062271
20062272
CMRR vs. Frequency
PSRR vs. Frequency
20062273
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20062274
12
LM3211
Operation
20062202
FIGURE 1. Simplified Boost Converter Diagram
(a) First Cycle of Operation (b) Second Cycle Of Operation
CONTINUOUS CONDUCTION MODE
The LM3211 is a current-mode, PWM boost regulator. A
boost regulator steps the input voltage up to a higher output
voltage. In continuous conduction mode (when the inductor
current never reaches zero at steady state), the boost regulator operates in two cycles.
In the first cycle of operation, shown in Figure 1 (a), the
transistor is closed and the diode is reverse biased. Energy
is collected in the inductor and the load current is supplied by
COUT.
The second cycle is shown in Figure 1 (b). During this cycle,
the transistor is open and the diode is forward biased. The
energy stored in the inductor is transferred to the load and
output capacitor.
The ratio of these two cycles determines the output voltage.
The output voltage is defined approximately as:
SOFT-START CAPACITOR
The LM3211 has patented internal circuitry that is used to
limit the inductor inrush current on start-up. This inrush
current limiting circuitry serves as a soft-start. However,
many applications may require much more soft-start than
what is available with the internal circuitry. The external SS
pin is used to tailor the soft-start for a specific application. A
11µA current charges the external soft-start capacitor, Css.
The soft-start time can be estimated as:
Tss = Css*0.6V/11µA
The minimum soft-start time is set by the internal soft-start
circuitry, typically 7ms for 600kHz operation and approximately half that for 1.25MHz operation. Only longer soft-start
times may be implemented using the SS pin and a capacitor
CSS. If a shorter time is designed for using the above equation, the internal soft-start circuitry will override it.
Due to the unique nature of the dual internal/external softstart, care was taken in the design to ensure temperature
stable operation. As you can see with the Iss data in the
Electrical Characterisitcs table and the graph "Soft-Start
Current vs. VIN" in the Typical Performance Characterisitcs
section, the soft start curent has a temperature coefficient
and would lead one to believe there would be significant
variation with temperature. Though the current has a temperature coefficient the actual programmed external soft
start time does not show this extreme of a temperature
variation. As you can see in the following transient plots:
where D is the duty cycle of the switch, D and D' will be
required for design calculations
SETTING THE OUTPUT VOLTAGE
The output voltage is set using the feedback pin and a
resistor divider connected to the output as shown in the
typical operating circuit. The feedback pin voltage is 1.265V,
so the ratio of the feedback resistors sets the output voltage
according to the following equation:
13
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LM3211
Operation
INTRODUCTION TO COMPENSATION
(Continued)
VOUT = 8V, VIN = 2.5V, RL = 27Ω, CSS = 330nF, T = 4ms/div,
F = 1.25MHz.
Trace:
1) SHDN, 1V/div, DC Coupled
2) IL, 0.5A/div, DC Coupled
3) VOUT, 5V/div, DC Coupled
20062280
TA = −20˚C
20062205
FIGURE 2. (a) Inductor current. (b) Diode current.
The LM3211 is a current mode PWM boost converter. The
signal flow of this control scheme has two feedback loops,
one that senses switch current and one that senses output
voltage.
To keep a current programmed control converter stable
above duty cycles of 50%, the inductor must meet certain
criteria. The inductor, along with input and output voltage,
will determine the slope of the current through the inductor
(see Figure 2 (a)). If the slope of the inductor current is too
great, the circuit will be unstable above duty cycles of 50%.
A 10µH inductor is recommended for most 600 kHz applications, while a 4.7µH inductor may be used for most 1.25 MHz
applications. If the duty cycle is approaching the maximum of
85%, it may be necessary to increase the inductance by as
much as 2X. See Inductor and Diode Selection for more
detailed inductor sizing.
The LM3211 provides a compensation pin (VC) to customize
the voltage loop feedback. It is recommended that a series
combination of RC and CC be used for the compensation
network, as shown in the typical application circuit. For any
given application, there exists a unique combination of RC
and CC that will optimize the performance of the LM3211
circuit in terms of its transient response. The series combination of RC and CC introduces a pole-zero pair according to
the following equations:
20062281
TA = 27˚C
20062282
TA = 85˚C
When programming the softstart time externally, simply use
the equation given in the Soft-Start Capacitor section above.
This equation uses the typical room temperature value of the
soft start current, 11µA, to set the soft start time.
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where RO is the output impedance of the error amplifier,
approximately 1MΩ. For most applications, performance can
be optimized by choosing values within the range 5kΩ ≤ RC
≤ 60kΩ (RC can be up to 200kΩ if CC2 is used, see High
Output Capacitor ESR Compensation) and 680pF ≤ CC ≤
14
tions are possible in the application, the diode current rating
must exceed the switch current limit. Using Schottky diodes
with lower forward voltage drop will decrease power dissipation and increase efficiency.
(Continued)
4.7nF. Refer to the Applications Information section for recommended values for specific circuits and conditions. Refer
to the Compensation section for other design requirement.
DC GAIN AND OPEN-LOOP GAIN
Since the control stage of the converter forms a complete
feedback loop with the power components, it forms a closedloop system that must be stabilized to avoid positive feedback and instability. A value for open-loop DC gain will be
required, from which you can calculate, or place, poles and
zeros to determine the crossover frequency and the phase
margin. A high phase margin (greater than 45˚) is desired for
the best stability and transient response. For the purpose of
stabilizing the LM3211, choosing a crossover point well below where the right half plane zero is located will ensure
sufficient phase margin. A discussion of the right half plane
zero and checking the crossover using the DC gain will
follow.
COMPENSATION FOR BOOST DC/DC
This section will present a general design procedure to help
insure a stable and operational circuit. The designs in this
datasheet are optimized for particular requirements. If different conversions are required, some of the components may
need to be changed to ensure stability. Below is a set of
general guidelines in designing a stable circuit for continuous conduction operation, in most all cases this will provide
for stability during discontinuous operation as well. The
power components and their effects will be determined first,
then the compensation components will be chosen to produce stability.
INDUCTOR AND DIODE SELECTION
Although the inductor sizes mentioned earlier are fine for
most applications, a more exact value can be calculated. To
ensure stability at duty cycles above 50%, the inductor must
have some minimum value determined by the minimum
input voltage and the maximum output voltage. This equation is:
INPUT AND OUTPUT CAPACITOR SELECTION
The switching action of a boost regulator causes a triangular
voltage waveform at the input. A capacitor is required to
reduce the input ripple and noise for proper operation of the
regulator. The size used is dependant on the application and
board layout. If the regulator will be loaded uniformly, with
very little load changes, and at lower current outputs, the
input capacitor size can often be reduced. The size can also
be reduced if the input of the regulator is very close to the
source output. The size will generally need to be larger for
applications where the regulator is supplying nearly the
maximum rated output or if large load steps are expected. A
minimum value of 10µF should be used for the less stressful
conditions while a 22µF to 47µF capacitor may be required
for higher power and dynamic loads. Larger values and/or
lower ESR may be needed if the application requires very
low ripple on the input source voltage.
where fs is the switching frequency, D is the duty cycle, and
RDSON is the ON resistance of the internal switch taken from
the graph "RDSON vs. VIN" in the Typical Performance Characteristics section. This equation is only good for duty cycles
greater than 50% (D > 0.5), for duty cycles less than 50% the
recommended values may be used. The corresponding inductor current ripple as shown in Figure 2 (a) is given by:
The choice of output capacitors is also somewhat arbitrary
and depends on the design requirements for output voltage
ripple. It is recommended that low ESR (Equivalent Series
Resistance, denoted RESR) capacitors be used such as
ceramic, polymer electrolytic, or low ESR tantalum. Higher
ESR capacitors may be used but will require more compensation which will be explained later on in the section. The
ESR is also important because it determines the peak to
peak output voltage ripple according to the approximate
equation:
∆VOUT ) 2∆iLRESR (in Volts)
A minimum value of 10µF is recommended and may be
increased to a larger value. After choosing the output capacitor you can determine a pole-zero pair introduced into the
control loop by the following equations:
The inductor ripple current is important for a few reasons.
One reason is because the peak switch current will be the
average inductor current (input current or ILOAD/D’) plus ∆iL.
As a side note, discontinuous operation occurs when the
inductor current falls to zero during a switching cycle, or ∆iL
is greater than the average inductor current. Therefore, continuous conduction mode occurs when ∆iL is less than the
average inductor current. Care must be taken to make sure
that the switch will not reach its current limit during normal
operation. The inductor must also be sized accordingly. It
should have a saturation current rating higher than the peak
inductor current expected. The output voltage ripple is also
affected by the total ripple current.
The output diode for a boost regulator must be chosen
correctly depending on the output voltage and the output
current. The typical current waveform for the diode in continuous conduction mode is shown in Figure 2 (b). The diode
must be rated for a reverse voltage equal to or greater than
the output voltage used. The average current rating must be
greater than the maximum load current expected, and the
peak current rating must be greater than the peak inductor
current. During short circuit testing, or if short circuit condi-
Where RL is the minimum load resistance corresponding to
the maximum load current. The zero created by the ESR of
the output capacitor is generally very high frequency if the
ESR is small. If low ESR capacitors are used it can be
neglected. If higher ESR capacitors are used see the High
Output Capacitor ESR Compensation section.
15
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LM3211
Operation
LM3211
Operation
HIGH OUTPUT CAPACITOR ESR COMPENSATION
When using an output capacitor with a high ESR value, or
just to improve the overall phase margin of the control loop,
another pole may be introduced to cancel the zero created
by the ESR. This is accomplished by adding another capacitor, CC2, directly from the compensation pin VC to ground, in
parallel with the series combination of RC and CC. The pole
should be placed at the same frequency as fZ1, the ESR
zero. The equation for this pole follows:
(Continued)
RIGHT HALF PLANE ZERO
A current mode control boost regulator has an inherent right
half plane zero (RHP zero). This zero has the effect of a zero
in the gain plot, causing an imposed +20dB/decade on the
rolloff, but has the effect of a pole in the phase, subtracting
another 90˚ in the phase plot. This can cause undesirable
effects if the control loop is influenced by this zero. To ensure
the RHP zero does not cause instability issues, the control
loop should be designed to have a bandwidth of less than 1⁄2
the frequency of the RHP zero. This zero occurs at a frequency of:
To ensure this equation is valid, and that CC2 can be used
without negatively impacting the effects of RC and CC, fPC2
must be greater than 10fZC.
CHECKING THE DESIGN
where ILOAD is the maximum load current.
The final step is to check the design. This is to ensure a
bandwidth of 1⁄2 or less of the frequency of the RHP zero.
This is done by calculating the open-loop DC gain, ADC. After
this value is known, you can calculate the crossover visually
by placing a −20dB/decade slope at each pole, and a +20dB/
decade slope for each zero. The point at which the gain plot
crosses unity gain, or 0dB, is the crossover frequency. If the
crossover frequency is less than 1⁄2 the RHP zero, the phase
margin should be high enough for stability. The phase margin can also be improved by adding CC2 as discussed earlier
in the section. The equation for ADC is given below with
additional equations required for the calculation:
SELECTING THE COMPENSATION COMPONENTS
The first step in selecting the compensation components RC
and CC is to set a dominant low frequency pole in the control
loop. Simply choose values for RC and CC within the ranges
given in the Introduction to Compensation section to set this
pole in the area of 10Hz to 500Hz. The frequency of the pole
created is determined by the equation:
where RO is the output impedance of the error amplifier,
approximately 1MΩ. Since RC is generally much less than
RO, it does not have much effect on the above equation and
can be neglected until a value is chosen to set the zero fZC.
fZC is created to cancel out the pole created by the output
capacitor, fP1. The output capacitor pole will shift with different load currents as shown by the equation, so setting the
zero is not exact. Determine the range of fP1 over the expected loads and then set the zero fZC to a point approximately in the middle. The frequency of this zero is determined by:
mc ) 0.072fs (in V/s)
Now RC can be chosen with the selected value for CC.
Check to make sure that the pole fPC is still in the 10Hz to
500Hz range, change each value slightly if needed to ensure
both component values are in the recommended range. After
checking the design at the end of this section, these values
can be changed a little more to optimize performance if
desired. This is best done in the lab on a bench, checking the
load step response with different values until the ringing and
overshoot on the output voltage at the edge of the load steps
is minimal. This should produce a stable, high performance
circuit. For improved transient response, higher values of RC
should be chosen. This will improve the overall bandwidth
which makes the regulator respond more quickly to transients. If more detail is required, or the most optimal performance is desired, refer to a more in depth discussion of
compensating current mode DC/DC switching regulators.
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where RL is the minimum load resistance, VIN is the maximum input voltage, gm is the error amplifier transconductance found in the Electrical Characteristics table, and RDSON is the value chosen from the graph "RDSON vs. VIN " in
the Typical Performance Characteristics section.
BUFFER COMPENSATION
The Gamma buffers in the LM3211 are internally compensated. They will remain stable with no capacitive load while
sourcing or sinking current. Any capacitance presented by a
load in an application will only provide further stability.
16
input voltage ripple of the IC. For additional input voltage
filtering, a 100nF bypass capacitor can be placed in parallel
with CIN, close to the VIN pin, to shunt any high frequency
noise to ground. The output capacitor, COUT, should also be
placed close to the IC. Any copper trace connections for the
COUT capacitor can increase the series resistance, which
directly effects output voltage ripple and efficiency. The feedback network, resistors R1 and R2, should be kept close to
the FB pin, and away from the inductor, to minimize copper
trace connections that can inject noise into the system.
Trace connections made to the inductor and schottky diode
should be minimized to reduce power dissipation and increase overall efficiency.
The layout of the Gamma buffer traces is important to minimize noise injected into them. Route all input and output
traces for the buffers away from the inductor and the high
power switching traces used for the boost switcher. For best
performance route these traces on the opposite side of the
board from the high current switching traces and use a
ground plane between the two sides for shielding.
(Continued)
LAYOUT CONSIDERATIONS
The LM3211 uses two separate ground connections, GND
for the driver and NMOS power device of the boost regulator
and AGND for the sensitive analog control circuitry of the
boost regulator and the VCOM and Gamma buffers. The
AGND and GND pins should be tied directly together at the
package, see Figure 3 and Figure 4. The feedback, softstart,
and compensation networks should be connected directly to
a dedicated analog ground plane and this ground plane must
connect to the AGND pin, as in Figure 3. If no analog ground
plane is available then the ground connections of the feedback, softstart, and compensation networks must tie directly
to the AGND pin, as show in Figure 4. Connecting these
networks to the GND pin can inject noise into the system and
effect performance. For 600kHz operation the FSLCT pin
should be tied to an analog ground plane or directly to the
AGND pin. For 1.25MHz operation the FSLCT pin should be
tied to the VIN pin.
The input bypass capacitor CIN must be placed close to the
IC. This will reduce copper trace resistance which effects
20062252
FIGURE 3. Multi-Layer Layout
20062253
FIGURE 4. Single Layer Layout
17
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LM3211
Operation
LM3211
Application Information
Some recommended Inductors (others may be used)
Manufacturer
Inductor
Contact Information
Coilcraft
DO3316 series
www.coilcraft.com
Coiltronics
DRQ73 series
www.cooperet.com
Pulse
P0751 series
www.pulseeng.com
Sumida
CDRH8D28 series
www.sumida.com
Some recommended Input and Output Capacitors (others may be used)
Manufacturer
Capacitor
Contact Information
Vishay Sprague
293D, 592D, and 595D series tantalum
www.vishay.com
Taiyo Yuden
High capacitance MLCC ceramic
www.t-yuden.com
Cornell Dubilier
ESRD seriec Polymer Aluminum Electrolytic
SPV and AFK series V-chip series
www.cde.com
Panasonic
High capacitance MLCC ceramic
EEJ-L series tantalum
www.panasonic.com
20062259
FIGURE 5. 600kHz, 8V Application
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18
LM3211
Application Information
(Continued)
20062284
FIGURE 6. 1.25MHz, 5V Application
20062285
FIGURE 7. 1.25MHz, 10V Application
19
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LM3211
Application Information
(Continued)
20062286
FIGURE 8. 1.25MHz, 12V Application
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20
LM3211 Step-up PWM DC/DC Converter Integrated with 4 Buffers
Physical Dimensions
inches (millimeters) unless otherwise noted
TSSOP-20 Pin Package (MTC)
For Ordering, Refer to Ordering Information Table
NS Package Number MTC20
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