LM2575 1.0 A, Adjustable Output Voltage, Step-Down Switching Regulator The LM2575 series of regulators are monolithic integrated circuits ideally suited for easy and convenient design of a step–down switching regulator (buck converter). All circuits of this series are capable of driving a 1.0 A load with excellent line and load regulation. These devices are available in fixed output voltages of 3.3 V, 5.0 V, 12 V, 15 V, and an adjustable output version. These regulators were designed to minimize the number of external components to simplify the power supply design. Standard series of inductors optimized for use with the LM2575 are offered by several different inductor manufacturers. Since the LM2575 converter is a switch–mode power supply, its efficiency is significantly higher in comparison with popular three–terminal linear regulators, especially with higher input voltages. In many cases, the power dissipated by the LM2575 regulator is so low, that no heatsink is required or its size could be reduced dramatically. The LM2575 features include a guaranteed ±4% tolerance on output voltage within specified input voltages and output load conditions, and ±10% on the oscillator frequency (±2% over 0°C to 125°C). External shutdown is included, featuring 80 µA typical standby current. The output switch includes cycle–by–cycle current limiting, as well as thermal shutdown for full protection under fault conditions. http://onsemi.com TO–220 TV SUFFIX CASE 314B 1 5 Heatsink surface connected to Pin 3 TO–220 T SUFFIX CASE 314D 1 5 Pin 1. 2. 3. 4. 5. Vin Output Ground Feedback ON/OFF Features • 3.3 V, 5.0 V, 12 V, 15 V, and Adjustable Output Versions • Adjustable Version Output Voltage Range of 1.23 V to 37 V ±4% • • • • • • • • • Maximum Over Line and Load Conditions Guaranteed 1.0 A Output Current Wide Input Voltage Range: 4.75 V to 40 V Requires Only 4 External Components 52 kHz Fixed Frequency Internal Oscillator TTL Shutdown Capability, Low Power Standby Mode High Efficiency Uses Readily Available Standard Inductors Thermal Shutdown and Current Limit Protection Moisture Sensitivity Level (MSL) Equals 1 1 5 Heatsink surface (shown as terminal 6 in case outline drawing) is connected to Pin 3 ORDERING INFORMATION See detailed ordering and shipping information in the package dimensions section on page 24 of this data sheet. DEVICE MARKING INFORMATION See general marking information in the device marking section on page 24 of this data sheet. Applications • • • • • • D2PAK D2T SUFFIX CASE 936A Simple and High–Efficiency Step–Down (Buck) Regulators Efficient Pre–Regulator for Linear Regulators On–Card Switching Regulators Positive to Negative Converters (Buck–Boost) Negative Step–Up Converters Power Supply for Battery Chargers Semiconductor Components Industries, LLC, 2002 January, 2002 – Rev. 4 1 Publication Order Number: LM2575/D LM2575 Typical Application (Fixed Output Voltage Versions) Feedback 7.0 V - 40 V Unregulated DC Input +Vin Cin 100 µF 4 LM2575 Output 1 3 Gnd 5 2 ON/OFF L1 330 µH D1 1N5819 Cout 330 µF 5.0 V Regulated Output 1.0 A Load Representative Block Diagram and Typical Application Unregulated DC Input Cin +Vin 3.1 V Internal Regulator 1 ON/OFF ON/OFF 5 4 Feedback R2 Fixed Gain Error Amplifier Comparator R1 1.0 k Current Limit Output Voltage Versions R2 (Ω) 3.3 V 5.0 V 12 V 15 V 1.7 k 3.1 k 8.84 k 11.3 k For adjustable version R1 = open, R2 = 0 Ω Driver Latch Freq Shift 18 kHz 1.235 V Band-Gap Reference 52 kHz Oscillator Reset L1 Output Thermal Shutdown 1.0 Amp Switch 2 Gnd Regulated Output Vout Cout D1 3 Load This device contains 162 active transistors. Figure 1. Block Diagram and Typical Application ABSOLUTE MAXIMUM RATINGS (Absolute Maximum Ratings indicate limits beyond which damage to the device may occur.) Rating Symbol Value Unit Maximum Supply Voltage Vin 45 V ON/OFF Pin Input Voltage – –0.3 V ≤ V ≤ +Vin V Output Voltage to Ground (Steady–State) – –1.0 V Power Dissipation Case 314B and 314D (TO–220, 5–Lead) Thermal Resistance, Junction–to–Ambient Thermal Resistance, Junction–to–Case Case 936A (D2PAK) Thermal Resistance, Junction–to–Ambient (Figure 34) Thermal Resistance, Junction–to–Case PD RθJA RθJC PD RθJA RθJC Internally Limited 65 5.0 Internally Limited 70 5.0 W °C/W °C/W W °C/W °C/W Storage Temperature Range Tstg –65 to +150 °C Minimum ESD Rating (Human Body Model: C = 100 pF, R = 1.5 kΩ) – 3.0 kV Lead Temperature (Soldering, 10 s) – 260 °C Maximum Junction Temperature TJ 150 °C http://onsemi.com 2 LM2575 OPERATING RATINGS (Operating Ratings indicate conditions for which the device is intended to be functional, but do not guarantee specific performance limits. For guaranteed specifications and test conditions, see the Electrical Characteristics.) Rating Symbol Value Unit Operating Junction Temperature Range TJ –40 to +125 °C Supply Voltage Vin 40 V SYSTEM PARAMETERS ([Note 1] Test Circuit Figure 14) ELECTRICAL CHARACTERISTICS (Unless otherwise specified, Vin = 12 V for the 3.3 V, 5.0 V, and Adjustable version, Vin = 25 V for the 12 V version, and Vin = 30 V for the 15 V version. ILoad = 200 mA. For typical values TJ = 25°C, for min/max values TJ is the operating junction temperature range that applies [Note 2], unless otherwise noted.) Characteristics Symbol Min Typ Max Unit Output Voltage (Vin = 12 V, ILoad = 0.2 A, TJ = 25°C) Vout 3.234 3.3 3.366 V Output Voltage (4.75 V ≤ Vin ≤ 40 V, 0.2 A ≤ ILoad ≤ 1.0 A) TJ = 25°C TJ = –40 to +125°C Vout 3.168 3.135 3.3 – 3.432 3.465 η – 75 – % Output Voltage (Vin = 12 V, ILoad = 0.2 A, TJ = 25°C) Vout 4.9 5.0 5.1 V Output Voltage (8.0 V ≤ Vin ≤ 40 V, 0.2 A ≤ ILoad ≤ 1.0 A) TJ = 25°C TJ = –40 to +125°C Vout 4.8 4.75 5.0 – 5.2 5.25 η – 77 – % Output Voltage (Vin = 25 V, ILoad = 0.2 A, TJ = 25°C) Vout 11.76 12 12.24 V Output Voltage (15 V ≤ Vin ≤ 40 V, 0.2 A ≤ ILoad ≤ 1.0 A) TJ = 25°C TJ = –40 to +125°C Vout 11.52 11.4 12 – 12.48 12.6 η – 88 – % Output Voltage (Vin = 30 V, ILoad = 0.2 A, TJ = 25°C) Vout 14.7 15 15.3 V Output Voltage (18 V ≤ Vin ≤ 40 V, 0.2 A ≤ ILoad ≤ 1.0 A) TJ = 25°C TJ = –40 to +125°C Vout 14.4 14.25 15 – 15.6 15.75 η – 88 – % Feedback Voltage (Vin = 12 V, ILoad = 0.2 A, Vout = 5.0 V, TJ = 25°C) VFB 1.217 1.23 1.243 V Feedback Voltage (8.0 V ≤ Vin ≤ 40 V, 0.2 A ≤ ILoad ≤ 1.0 A, Vout = 5.0 V) TJ = 25°C TJ = –40 to +125°C VFB 1.193 1.18 1.23 – 1.267 1.28 – 77 – LM2575–3.3 ([Note 1] Test Circuit Figure 14) Efficiency (Vin = 12 V, ILoad = 1.0 A) V LM2575–5 ([Note 1] Test Circuit Figure 14) Efficiency (Vin = 12 V, ILoad = 1.0 A) V LM2575–12 ([Note 1] Test Circuit Figure 14) Efficiency (Vin = 15V, ILoad = 1.0 A) V LM2575–15 ([Note 1] Test Circuit Figure 14) Efficiency (Vin = 18 V, ILoad = 1.0 A) V LM2575 ADJUSTABLE VERSION ([Note 1] Test Circuit Figure 14) η Efficiency (Vin = 12 V, ILoad = 1.0 A, Vout = 5.0 V) V % 1. External components such as the catch diode, inductor, input and output capacitors can affect switching regulator system performance. When the LM2575 is used as shown in the Figure 14 test circuit, system performance will be as shown in system parameters section. 2. Tested junction temperature range for the LM2575: Tlow = –40°C Thigh = +125°C http://onsemi.com 3 LM2575 DEVICE PARAMETERS ELECTRICAL CHARACTERISTICS (Unless otherwise specified, Vin = 12 V for the 3.3 V, 5.0 V, and Adjustable version, Vin = 25 V for the 12 V version, and Vin = 30 V for the 15 V version. ILoad = 200 mA. For typical values TJ = 25°C, for min/max values TJ is the operating junction temperature range that applies [Note 2], unless otherwise noted.) Characteristics Symbol Min Typ Max – – 25 – 100 200 – 47 42 52 – – – 58 63 – – 1.0 – 1.2 1.3 94 98 – 1.7 1.4 2.3 – 3.0 3.2 – – 0.8 6.0 2.0 20 – – 5.0 – 9.0 11 – – 80 – 200 400 Unit ALL OUTPUT VOLTAGE VERSIONS Feedback Bias Current (Vout = 5.0 V [Adjustable Version Only]) TJ = 25°C TJ = –40 to +125°C Ib Oscillator Frequency [Note 3] TJ = 25°C TJ = 0 to +125°C TJ = –40 to +125°C fosc Saturation Voltage (Iout = 1.0 A [Note 4]) TJ = 25°C TJ = –40 to +125°C Vsat Max Duty Cycle (“on”) [Note 5] DC Current Limit (Peak Current [Notes 4 and 3]) TJ = 25°C TJ = –40 to +125°C ICL Output Leakage Current [Notes 6 and 7], TJ = 25°C Output = 0 V Output = –1.0 V IL Quiescent Current [Note 6] TJ = 25°C TJ = –40 to +125°C IQ Standby Quiescent Current (ON/OFF Pin = 5.0 V (“off”)) TJ = 25°C TJ = –40 to +125°C nA kHz V A mA mA µA Istby ON/OFF Pin Logic Input Level (Test Circuit Figure 14) Vout = 0 V TJ = 25°C TJ = –40 to +125°C Vout = Nominal Output Voltage TJ = 25°C TJ = –40 to +125°C % V VIH 2.2 2.4 1.4 – – – – – 1.2 – 1.0 0.8 – – 15 0 30 5.0 VIL µA ON/OFF Pin Input Current (Test Circuit Figure 14) ON/OFF Pin = 5.0 V (“off”), TJ = 25°C ON/OFF Pin = 0 V (“on”), TJ = 25°C IIH IIL 3. The oscillator frequency reduces to approximately 18 kHz in the event of an output short or an overload which causes the regulated output voltage to drop approximately 40% from the nominal output voltage. This self protection feature lowers the average dissipation of the IC by lowering the minimum duty cycle from 5% down to approximately 2%. 4. Output (Pin 2) sourcing current. No diode, inductor or capacitor connected to output pin. 5. Feedback (Pin 4) removed from output and connected to 0 V. 6. Feedback (Pin 4) removed from output and connected to +12 V for the Adjustable, 3.3 V, and 5.0 V versions, and +25 V for the 12 V and 15 V versions, to force the output transistor “off”. 7. Vin = 40 V. http://onsemi.com 4 LM2575 0.6 Vin = 20 V ILoad = 200 mA Normalized at TJ = 25°C 0.4 0.2 0 -0.2 -0.4 -0.6 -50 -25 0 1.0 Vout , OUTPUT VOLTAGE CHANGE (%) Vout , OUTPUT VOLTAGE CHANGE (%) TYPICAL PERFORMANCE CHARACTERISTICS (Circuit of Figure 14) 25 50 75 100 0.6 0.2 5.0 10 15 20 25 30 Vin, INPUT VOLTAGE (V) Figure 2. Normalized Output Voltage Figure 3. Line Regulation 35 40 3.0 IO , OUTPUT CURRENT (A) 1.1 1.0 0.9 -40°C 0.8 25°C 0.7 0.6 125°C 0.5 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 1.8 1.6 0.8 0.9 1.0 1.5 1.0 0.5 Vin = 25 V -25 0 25 50 75 100 TJ, JUNCTION TEMPERATURE (°C) Figure 4. Switch Saturation Voltage Figure 5. Current Limit 20 ∆Vout = 5% Rind = 0.2 Ω ILoad = 1.0 A 1.2 ILoad = 200 mA 0.8 0.6 0.4 -50 2.0 SWITCH CURRENT (A) 1.4 1.0 2.5 0 -50 IQ , QUIESCENT CURRENT (mA) Vsat , SATURATION VOLTAGE (V) 0 TJ, JUNCTION TEMPERATURE (°C) 2.0 INPUT-OUTPUT DIFFERENTIAL (V) 12 V and 15 V 0 1.2 0.4 3.3 V, 5.0 V and Adj 0.4 -0.2 125 ILoad = 200 mA TJ = 25°C 0.8 -25 0 25 50 75 100 Vout = 5.0 V Measured at Ground Pin TJ = 25°C 18 16 14 ILoad = 1.0 A 12 10 ILoad = 200 mA 8.0 6.0 4.0 125 0 5.0 10 15 20 25 30 TJ, JUNCTION TEMPERATURE (°C) Vin, INPUT VOLTAGE (V) Figure 6. Dropout Voltage Figure 7. Quiescent Current http://onsemi.com 5 125 35 40 120 Istby , STANDBY QUIESCENT CURRENT (µA) Istby , STANDBY QUIESCENT CURRENT (µA) LM2575 TJ = 25°C 100 80 60 40 20 0 0 5.0 10 15 20 25 30 35 40 80 60 40 20 0 -50 -25 40 -4.0 -6.0 -8.0 50 75 100 -25 0 25 50 75 100 20 0 -20 -40 -50 125 -25 0 25 50 75 100 TJ, JUNCTION TEMPERATURE (°C) Figure 11. Feedback Pin Current I Load, LOAD CURRENT (A) Vout , OUTPUT VOLTAGE CHANGE (mV) Figure 10. Oscillator Frequency 10 V 0 OUTPUT 1.0 A CURRENT (PIN 2) 0 100 0 -100 1.0 A 0.5 A OUTPUT 20 mV RIPPLE /DIV VOLTAGE 5.0 µs/DIV Figure 12. Switching Waveforms 1.0 0.5 0 100 µs/DIV Figure 13. Load Transient Response http://onsemi.com 6 125 Adjustable Version Only TJ, JUNCTION TEMPERATURE (°C) INDUCTOR CURRENT 25 Figure 9. Standby Quiescent Current -2.0 OUTPUT VOLTAGE (PIN 2) 0 Figure 8. Standby Quiescent Current IFB , FEEDBACK PIN CURRENT (nA) NORMALIZED FREQUENCY (%) 100 TJ, JUNCTION TEMPERATURE (°C) Vin = 12 V Normalized at 25°C -10 -50 Vin = 12 V VON/OFF = 5.0 V Vin, INPUT VOLTAGE (V) 2.0 0 120 125 LM2575 5.0 Output Voltage Versions Feedback 4 Vin + Output Gnd 3 Vin Unregulated DC Input 8.0 V - 40 V Vout Regulated Output L1 330 µH LM2575–5 1 2 ON/OFF 5 Cin 100 µF/50 V D1 1N5819 Cout 330 µF /16 V Load - Adjustable Output Voltage Versions Feedback Vin + LM2575 Adjustable 1 3 Unregulated DC Input 8.0 V - 40 V Gnd 5 4 Vout Regulated Output L1 330 µH Output 2 ON/OFF Cin 100 µF/50 V D1 1N5819 Cout 330 µF /16 V R2 Load R1 V out V R2 R1 1 R2 R1 ref V out V ref 1 Where Vref = 1.23 V, R1 between 1.0 kΩ and 5.0 kΩ Figure 14. Typical Test Circuit PCB LAYOUT GUIDELINES On the other hand, the PCB area connected to the Pin 2 (emitter of the internal switch) of the LM2575 should be kept to a minimum in order to minimize coupling to sensitive circuitry. Another sensitive part of the circuit is the feedback. It is important to keep the sensitive feedback wiring short. To assure this, physically locate the programming resistors near to the regulator, when using the adjustable version of the LM2575 regulator. As in any switching regulator, the layout of the printed circuit board is very important. Rapidly switching currents associated with wiring inductance, stray capacitance and parasitic inductance of the printed circuit board traces can generate voltage transients which can generate electromagnetic interferences (EMI) and affect the desired operation. As indicated in the Figure 14, to minimize inductance and ground loops, the length of the leads indicated by heavy lines should be kept as short as possible. For best results, single–point grounding (as indicated) or ground plane construction should be used. http://onsemi.com 7 LM2575 PIN FUNCTION DESCRIPTION Pin Symbol 1 Vin 2 Output 3 Gnd 4 Feedback This pin senses regulated output voltage to complete the feedback loop. The signal is divided by the internal resistor divider network R2, R1 and applied to the non–inverting input of the internal error amplifier. In the Adjustable version of the LM2575 switching regulator this pin is the direct input of the error amplifier and the resistor network R2, R1 is connected externally to allow programming of the output voltage. 5 ON/OFF It allows the switching regulator circuit to be shut down using logic level signals, thus dropping the total input supply current to approximately 80 µA. The input threshold voltage is typically 1.4 V. Applying a voltage above this value (up to +Vin) shuts the regulator off. If the voltage applied to this pin is lower than 1.4 V or if this pin is connected to ground, the regulator will be in the “on” condition. Description (Refer to Figure 1) This pin is the positive input supply for the LM2575 step–down switching regulator. In order to minimize voltage transients and to supply the switching currents needed by the regulator, a suitable input bypass capacitor must be present (Cin in Figure 1). This is the emitter of the internal switch. The saturation voltage Vsat of this output switch is typically 1.0 V. It should be kept in mind that the PCB area connected to this pin should be kept to a minimum in order to minimize coupling to sensitive circuitry. Circuit ground pin. See the information about the printed circuit board layout. DESIGN PROCEDURE Buck Converter Basics The LM2575 is a “Buck” or Step–Down Converter which is the most elementary forward–mode converter. Its basic schematic can be seen in Figure 15. The operation of this regulator topology has two distinct time periods. The first one occurs when the series switch is on, the input voltage is connected to the input of the inductor. The output of the inductor is the output voltage, and the rectifier (or catch diode) is reverse biased. During this period, since there is a constant voltage source connected across the inductor, the inductor current begins to linearly ramp upwards, as described by the following equation: I L(on) I For the buck converter with ideal components, the duty cycle can also be described as: V d out V in L Figure 16 shows the buck converter idealized waveforms of the catch diode voltage and the inductor current. Vout Cout Power Switch Off RLoad Power Switch On Power Switch Off Power Switch On Time VD(FWD) Inductor Current D1 Von(SW) Diode Voltage L L t d on , where T is the period of switching. T Vin – Vout ton Power Switch Vout – VD toff This period ends when the power switch is once again turned on. Regulation of the converter is accomplished by varying the duty cycle of the power switch. It is possible to describe the duty cycle as follows: During this “on” period, energy is stored within the core material in the form of magnetic flux. If the inductor is properly designed, there is sufficient energy stored to carry the requirements of the load during the “off” period. Vin L(off) Figure 15. Basic Buck Converter The next period is the “off” period of the power switch. When the power switch turns off, the voltage across the inductor reverses its polarity and is clamped at one diode voltage drop below ground by catch dioded. Current now flows through the catch diode thus maintaining the load current loop. This removes the stored energy from the inductor. The inductor current during this time is: Ipk Imin Diode Power Switch Diode ILoad(AV) Power Switch Time Figure 16. Buck Converter Idealized Waveforms http://onsemi.com 8 LM2575 Procedure (Fixed Output Voltage Version) In order to simplify the switching regulator design, a step–by–step design procedure and example is provided. Procedure Example Given Parameters: Vout = Regulated Output Voltage (3.3 V, 5.0 V, 12 V or 15 V) Vin(max) = Maximum DC Input Voltage ILoad(max) = Maximum Load Current Given Parameters: Vout = 5.0 V Vin(max) = 20 V ILoad(max) = 0.8 A 1. Controller IC Selection According to the required input voltage, output voltage and current, select the appropriate type of the controller IC output voltage version. 1. Controller IC Selection According to the required input voltage, output voltage, current polarity and current value, use the LM2575–5 controller IC 2. Input Capacitor Selection (Cin) To prevent large voltage transients from appearing at the input and for stable operation of the converter, an aluminium or tantalum electrolytic bypass capacitor is needed between the input pin +Vin and ground pin Gnd. This capacitor should be located close to the IC using short leads. This capacitor should have a low ESR (Equivalent Series Resistance) value. 2. Input Capacitor Selection (Cin) A 47 µF, 25 V aluminium electrolytic capacitor located near to the input and ground pins provides sufficient bypassing. 3. Catch Diode Selection (D1) A. For this example the current rating of the diode is 1.0 A. 3. Catch Diode Selection (D1) A. Since the diode maximum peak current exceeds the regulator maximum load current the catch diode current rating must be at least 1.2 times greater than the maximum load current. For a robust design the diode should have a current rating equal to the maximum current limit of the LM2575 to be able to withstand a continuous output short B. The reverse voltage rating of the diode should be at least 1.25 times the maximum input voltage. B. Use a 30 V 1N5818 Schottky diode, or any of the suggested fast recovery diodes shown in the Table 4. 4. Inductor Selection (L1) A. According to the required working conditions, select the correct inductor value using the selection guide from Figures 17 to 21. B. From the appropriate inductor selection guide, identify the inductance region intersected by the Maximum Input Voltage line and the Maximum Load Current line. Each region is identified by an inductance value and an inductor code. C. Select an appropriate inductor from the several different manufacturers part numbers listed in Table 1 or Table 2. When using Table 2 for selecting the right inductor the designer must realize that the inductor current rating must be higher than the maximum peak current flowing through the inductor. This maximum peak current can be calculated as follows: Vin–Vout ton I I p(max) Load(max) 2L 4. Inductor Selection (L1) A. Use the inductor selection guide shown in Figures 17 to 21. B. From the selection guide, the inductance area intersected by the 20 V line and 0.8 A line is L330. C. Inductor value required is 330 µH. From the Table 1 or Table 2, choose an inductor from any of the listed manufacturers. where ton is the “on” time of the power switch and V ton out x 1 V fosc in For additional information about the inductor, see the inductor section in the “External Components” section of this data sheet. http://onsemi.com 9 LM2575 Procedure (Fixed Output Voltage Version) (continued)In order to simplify the switching regulator design, a step–by–step design procedure and example is provided. Procedure Example 5. Output Capacitor Selection (Cout) A. Since the LM2575 is a forward–mode switching regulator with voltage mode control, its open loop 2–pole–2–zero frequency characteristic has the dominant pole–pair determined by the output capacitor and inductor values. For stable operation and an acceptable ripple voltage, (approximately 1% of the output voltage) a value between 100 µF and 470 µF is recommended. B. Due to the fact that the higher voltage electrolytic capacitors generally have lower ESR (Equivalent Series Resistance) numbers, the output capacitor’s voltage rating should be at least 1.5 times greater than the output voltage. For a 5.0 V regulator, a rating at least 8V is appropriate, and a 10 V or 16 V rating is recommended. 5. Output Capacitor Selection (Cout) A. Cout = 100 µF to 470 µF standard aluminium electrolytic. B. Capacitor voltage rating = 16 V. Procedure (Adjustable Output Version: LM2575–Adj) Procedure Example Given Parameters: Vout = Regulated Output Voltage Vin(max) = Maximum DC Input Voltage ILoad(max) = Maximum Load Current Given Parameters: Vout = 8.0 V Vin(max) = 12 V ILoad(max) = 1.0 A 1. Programming Output Voltage To select the right programming resistor R1 and R2 value (see Figure 14) use the following formula: 1. Programming Output Voltage (selecting R1 and R2) Select R1 and R2: V out 1.23 1 R2 1 V out V ref R1 where Vref = 1.23 V R2 R1 Resistor R1 can be between 1.0 k and 5.0 kΩ. (For best temperature coefficient and stability with time, use 1% metal film resistors). V out R2 R1 –1 V ref V out V ref R2 R1 Select R1 = 1.8 kΩ 1 1.8 k 8.0 V 1.23 1 V R2 = 9.91 kΩ, choose a 9.88 k metal film resistor. 2. Input Capacitor Selection (Cin) To prevent large voltage transients from appearing at the input and for stable operation of the converter, an aluminium or tantalum electrolytic bypass capacitor is needed between the input pin +Vin and ground pin Gnd This capacitor should be located close to the IC using short leads. This capacitor should have a low ESR (Equivalent Series Resistance) value. 2. Input Capacitor Selection (Cin) A 100 µF aluminium electrolytic capacitor located near the input and ground pin provides sufficient bypassing. For additional information see input capacitor section in the “External Components” section of this data sheet. 3. Catch Diode Selection (D1) A. Since the diode maximum peak current exceeds the regulator maximum load current the catch diode current rating must be at least 1.2 times greater than the maximum load current. For a robust design, the diode should have a current rating equal to the maximum current limit of the LM2575 to be able to withstand a continuous output short. B. The reverse voltage rating of the diode should be at least 1.25 times the maximum input voltage. 3. Catch Diode Selection (D1) A. For this example, a 3.0 A current rating is adequate. B. Use a 20 V 1N5820 or MBR320 Schottky diode or any suggested fast recovery diode in the Table 4. http://onsemi.com 10 LM2575 Procedure (Adjustable Output Version: LM2575–Adj) (continued) Procedure Example 4. Inductor Selection (L1) A. Use the following formula to calculate the inductor Volt x microsecond [V x µs] constant: V out 6 x 10 [V x s] E x T V – V out in F[Hz] V on 4. Inductor Selection (L1) A. Calculate E x T [V x µs] constant: E x T (12 – 8.0) x 8.0 x 1000 51 [V x s] 52 12 B. E x T = 51 [V x µs] B. Match the calculated E x T value with the corresponding number on the vertical axis of the Inductor Value Selection Guide shown in Figure 21. This E x T constant is a measure of the energy handling capability of an inductor and is dependent upon the type of core, the core area, the number of turns, and the duty cycle. C. Next step is to identify the inductance region intersected by the E x T value and the maximum load current value on the horizontal axis shown in Figure 21. D. From the inductor code, identify the inductor value. Then select an appropriate inductor from the Table 1 or Table 2. The inductor chosen must be rated for a switching frequency of 52 kHz and for a current rating of 1.15 x IIoad. The inductor current rating can also be determined by calculating the inductor peak current: I p(max) I Load(max) C. ILoad(max) = 1.0 A Inductance Region = L220 D. Proper inductor value = 220 µH Choose the inductor from the Table 1 or Table 2. Vin – Vout ton 2L where ton is the “on” time of the power switch and t on V out V x 1 f osc in For additional information about the inductor, see the inductor section in the “External Components” section of this data sheet. 5. Output Capacitor Selection (Cout) A. Since the LM2575 is a forward–mode switching regulator with voltage mode control, its open loop 2–pole–2–zero frequency characteristic has the dominant pole–pair determined by the output capacitor and inductor values. 5. Output Capacitor Selection (Cout) A. Cout 7.785 12 53 µF 8.220 To achieve an acceptable ripple voltage, select Cout = 100 µF electrolytic capacitor. For stable operation, the capacitor must satisfy the following requirement: V in(max) [µF] Cout 7.785 V out x L [µH] B. Capacitor values between 10 µF and 2000 µF will satisfy the loop requirements for stable operation. To achieve an acceptable output ripple voltage and transient response, the output capacitor may need to be several times larger than the above formula yields. C. Due to the fact that the higher voltage electrolytic capacitors generally have lower ESR (Equivalent Series Resistance) numbers, the output capacitor’s voltage rating should be at least 1.5 times greater than the output voltage. For a 5.0 V regulator, a rating of at least 8V is appropriate, and a 10 V or 16 V rating is recommended. http://onsemi.com 11 LM2575 INDUCTOR VALUE SELECTION GUIDE 20 15 10 H1000 Vin , MAXIMUM INPUT VOLTAGE (V) Vin , MAXIMUM INPUT VOLTAGE (V) 60 L680 L470 8.0 L330 7.0 L220 L150 6.0 L100 5.0 0.2 0.3 0.4 0.5 0.6 0.8 L680 12 L470 10 L330 9.0 L220 8.0 L150 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 IL, MAXIMUM LOAD CURRENT (A) Figure 17. LM2575–3.3 Figure 18. LM2575–5.0 60 H2200 Vin , MAXIMUM INPUT VOLTAGE (V) Vin , MAXIMUM INPUT VOLTAGE (V) H1000 15 IL, MAXIMUM LOAD CURRENT (A) H1500 H1000 H680 H470 20 18 17 L680 16 L470 15 14 0.2 H1500 7.0 0.2 1.0 60 40 30 25 60 40 25 20 L330 L220 0.3 0.4 0.5 0.6 H2200 40 35 30 H1500 H1000 25 H680 22 20 19 L680 L330 L220 17 0.2 0.7 0.8 0.9 1.0 L470 18 0.3 0.4 0.5 0.6 IL, MAXIMUM LOAD CURRENT (A) IL, MAXIMUM LOAD CURRENT (A) Figure 19. LM2575–12 Figure 20. LM2575–15 ET, VOLTAGE TIME (Vµ s) 200 150 125 H2200 H1500 H1000 100 80 70 60 50 H470 L470 L330 30 20 0.2 H680 L680 40 L220 L150 L100 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 IL, MAXIMUM LOAD CURRENT (A) Figure 21. LM2575–Adj NOTE: This Inductor Value Selection Guide is applicable for continuous mode only. http://onsemi.com 12 H470 0.7 0.8 0.9 1.0 LM2575 Table 1. Inductor Selection Guide Inductor Code Inductor Value Pulse Eng Renco AIE Tech 39 L100 100 µH PE–92108 RL2444 415–0930 77 308 BV L150 150 µH PE–53113 RL1954 415–0953 77 358 BV L220 220 µH PE–52626 RL1953 415–0922 77 408 BV L330 330 µH PE–52627 RL1952 415–0926 77 458 BV L470 470 µH PE–53114 RL1951 415–0927 – L680 680 µH PE–52629 RL1950 415–0928 77 508 BV H150 150 µH PE–53115 RL2445 415–0936 77 368 BV H220 220 µH PE–53116 RL2446 430–0636 77 410 BV H330 330 µH PE–53117 RL2447 430–0635 77 460 BV H470 470 µH PE–53118 RL1961 430–0634 – H680 680 µH PE–53119 RL1960 415–0935 77 510 BV H1000 1000 µH PE–53120 RL1959 415–0934 77 558 BV H1500 1500 µH PE–53121 RL1958 415–0933 – H2200 2200 µH PE–53122 RL2448 415–0945 77 610 BV Table 2. Inductor Selection Guide Inductance Current (µH) (A) THT SMT THT SMT THT SMT SMT 0.32 67143940 67144310 RL–1284–68–43 RL1500–68 PE–53804 PE–53804–S DO1608–68 0.58 67143990 67144360 RL–5470–6 RL1500–68 PE–53812 PE–53812–S DO3308–683 0.99 67144070 67144450 RL–5471–5 RL1500–68 PE–53821 PE–53821–S DO3316–683 1.78 67144140 67144520 RL–5471–5 – PE–53830 PE–53830–S DO5022P–683 0.48 67143980 67144350 RL–5470–5 RL1500–100 PE–53811 PE–53811–S DO3308–104 0.82 67144060 67144440 RL–5471–4 RL1500–100 PE–53820 PE–53820–S DO3316–104 1.47 67144130 67144510 RL–5471–4 – PE–53829 PE–53829–S DO5022P–104 0.39 – 67144340 RL–5470–4 RL1500–150 PE–53810 PE–53810–S DO3308–154 0.66 67144050 67144430 RL–5471–3 RL1500–150 PE–53819 PE–53819–S DO3316–154 1.20 67144120 67144500 RL–5471–3 – PE–53828 PE–53828–S DO5022P–154 0.32 67143960 67144330 RL–5470–3 RL1500–220 PE–53809 PE–53809–S DO3308–224 0.55 67144040 67144420 RL–5471–2 RL1500–220 PE–53818 PE–53818–S DO3316–224 1.00 67144110 67144490 RL–5471–2 – PE–53827 PE–53827–S DO5022P–224 0.42 67144030 67144410 RL–5471–1 RL1500–330 PE–53817 PE–53817–S DO3316–334 0.80 67144100 67144480 RL–5471–1 – PE–53826 PE–53826–S DO5022P–334 68 100 150 220 330 Schott Renco Pulse Engineering Coilcraft NOTE: Table 1 and Table 2 of this Indicator Selection Guide shows some examples of different manufacturer products suitable for design with the LM2575. http://onsemi.com 13 LM2575 Table 3. Example of Several Inductor Manufacturers Phone/Fax Numbers Pulse Engineering Inc. Phone Fax + 1–619–674–8100 + 1–619–674–8262 Pulse Engineering Inc. Europe Phone Fax + 353 93 24 107 + 353 93 24 459 Renco Electronics Inc. Phone Fax + 1–516–645–5828 + 1–516–586–5562 AIE Magnetics Phone Fax + 1–813–347–2181 Coilcraft Inc. Phone Fax + 1–708–322–2645 + 1–708–639–1469 Coilcraft Inc., Europe Phone Fax + 44 1236 730 595 + 44 1236 730 627 Tech 39 Phone Fax + 33 8425 2626 + 33 8425 2610 Schott Corp. Phone Fax + 1–612–475–1173 + 1–612–475–1786 Table 4. Diode Selection Guide gives an overview about both surface–mount and through–hole diodes for an effective design. Device listed in bold are available from ON Semiconductor. Schottky Ultra–Fast Recovery 1.0 A 3.0 A 1.0 A VR SMT THT SMT THT 20 V SK12 1N5817 SR102 SK32 MBRD320 1N5820 MBR320 SR302 30 V MBRS130LT3 SK13 1N5818 SR103 11DQ03 SK33 MBRD330 1N5821 MBR330 SR303 31DQ03 40 V MBRS140T3 SK14 10BQ040 10MQ040 1N5819 SR104 11DQ04 MBRS340T3 MBRD340 30WQ04 SK34 1N5822 MBR340 SR304 31DQ04 50 V MBRS150 10BQ050 MBR150 SR105 11DQ05 MBRD350 SK35 30WQ05 MBR350 SR305 11DQ05 SMT THT MURS120T3 MUR120 11DF1 HER102 SMT THT MURS320T3 http://onsemi.com 14 3.0 A 10BF10 MURD320 31DF1 HER302 MUR320 30WF10 MUR420 LM2575 EXTERNAL COMPONENTS Input Capacitor (Cin) The Input Capacitor Should Have a Low ESR loop, resulting in oscillation at the output. This situation can occur when a tantalum capacitor, that can have a very low ESR, is used as the only output capacitor. For stable operation of the switch mode converter a low ESR (Equivalent Series Resistance) aluminium or solid tantalum bypass capacitor is needed between the input pin and the ground pin to prevent large voltage transients from appearing at the input. It must be located near the regulator and use short leads. With most electrolytic capacitors, the capacitance value decreases and the ESR increases with lower temperatures. For reliable operation in temperatures below –25°C larger values of the input capacitor may be needed. Also paralleling a ceramic or solid tantalum capacitor will increase the regulator stability at cold temperatures. At Low Temperatures, Put in Parallel Aluminium Electrolytic Capacitors with Tantalum Capacitors Electrolytic capacitors are not recommended for temperatures below –25°C. The ESR rises dramatically at cold temperatures and typically rises 3 times at –25°C and as much as 10 times at –40°C. Solid tantalum capacitors have much better ESR spec at cold temperatures and are recommended for temperatures below –25°C. They can be also used in parallel with aluminium electrolytics. The value of the tantalum capacitor should be about 10% or 20% of the total capacitance. The output capacitor should have at least 50% higher RMS ripple current rating at 52 kHz than the peak–to–peak inductor ripple current. RMS Current Rating of Cin The important parameter of the input capacitor is the RMS current rating. Capacitors that are physically large and have large surface area will typically have higher RMS current ratings. For a given capacitor value, a higher voltage electrolytic capacitor will be physically larger than a lower voltage capacitor, and thus be able to dissipate more heat to the surrounding air, and therefore will have a higher RMS current rating. The consequence of operating an electrolytic capacitor above the RMS current rating is a shortened operating life. In order to assure maximum capacitor operating lifetime, the capacitor’s RMS ripple current rating should be: Catch Diode Locate the Catch Diode Close to the LM2575 The LM2575 is a step–down buck converter; it requires a fast diode to provide a return path for the inductor current when the switch turns off. This diode must be located close to the LM2575 using short leads and short printed circuit traces to avoid EMI problems. Use a Schottky or a Soft Switching Ultra–Fast Recovery Diode Since the rectifier diodes are very significant source of losses within switching power supplies, choosing the rectifier that best fits into the converter design is an important process. Schottky diodes provide the best performance because of their fast switching speed and low forward voltage drop. They provide the best efficiency especially in low output voltage applications (5.0 V and lower). Another choice could be Fast–Recovery, or Ultra–Fast Recovery diodes. It has to be noted, that some types of these diodes with an abrupt turnoff characteristic may cause instability or EMI troubles. A fast–recovery diode with soft recovery characteristics can better fulfill a quality, low noise design requirements. Table 4 provides a list of suitable diodes for the LM2575 regulator. Standard 50/60 Hz rectifier diodes such as the 1N4001 series or 1N5400 series are NOT suitable. Irms > 1.2 x d x ILoad where d is the duty cycle, for a buck regulator V t d on out V T in |V out| t on and d for a buckboost regulator. |V out| V T in Output Capacitor (Cout) For low output ripple voltage and good stability, low ESR output capacitors are recommended. An output capacitor has two main functions: it filters the output and provides regulator loop stability. The ESR of the output capacitor and the peak–to–peak value of the inductor ripple current are the main factors contributing to the output ripple voltage value. Standard aluminium electrolytics could be adequate for some applications but for quality design low ESR types are recommended. An aluminium electrolytic capacitor’s ESR value is related to many factors such as the capacitance value, the voltage rating, the physical size and the type of construction. In most cases, the higher voltage electrolytic capacitors have lower ESR value. Often capacitors with much higher voltage ratings may be needed to provide low ESR values that are required for low output ripple voltage. Inductor The magnetic components are the cornerstone of all switching power supply designs. The style of the core and the winding technique used in the magnetic component’s design has a great influence on the reliability of the overall power supply. Using an improper or poorly designed inductor can cause high voltage spikes generated by the rate of transitions in current within the switching power supply, and the possibility of core saturation can arise during an abnormal operational mode. Voltage spikes can cause the semiconductors to enter avalanche breakdown and the part can instantly fail if enough energy is applied. It can also cause significant RFI (Radio Frequency Interference) and EMI (Electro–Magnetic Interference) problems. The Output Capacitor Requires an ESR Value That Has an Upper and Lower Limit As mentioned above, a low ESR value is needed for low output ripple voltage, typically 1% to 2% of the output voltage. But if the selected capacitor’s ESR is extremely low (below 0.05 Ω), there is a possibility of an unstable feedback http://onsemi.com 15 LM2575 Continuous and Discontinuous Mode of Operation amount of EMI (Electro–Magnetic Interference) shielding that the core must provide. The inductor selection guide covers different styles of inductors, such as pot core, E–core, toroid and bobbin core, as well as different core materials such as ferrites and powdered iron from different manufacturers. For high quality design regulators the toroid core seems to be the best choice. Since the magnetic flux is completely contained within the core, it generates less EMI, reducing noise problems in sensitive circuits. The least expensive is the bobbin core type, which consists of wire wound on a ferrite rod core. This type of inductor generates more EMI due to the fact that its core is open, and the magnetic flux is not completely contained within the core. When multiple switching regulators are located on the same printed circuit board, open core magnetics can cause interference between two or more of the regulator circuits, especially at high currents due to mutual coupling. A toroid, pot core or E–core (closed magnetic structure) should be used in such applications. Do Not Operate an Inductor Beyond its Maximum Rated Current Exceeding an inductor’s maximum current rating may cause the inductor to overheat because of the copper wire losses, or the core may saturate. Core saturation occurs when the flux density is too high and consequently the cross sectional area of the core can no longer support additional lines of magnetic flux. This causes the permeability of the core to drop, the inductance value decreases rapidly and the inductor begins to look mainly resistive. It has only the dc resistance of the winding. This can cause the switch current to rise very rapidly and force the LM2575 internal switch into cycle–by–cycle current limit, thus reducing the dc output load current. This can also result in overheating of the inductor and/or the LM2575. Different inductor types have different saturation characteristics, and this should be kept in mind when selecting an inductor. 1.0 0 POWER SWITCH CURRENT (A) INDUCTOR CURRENT (A) POWER SWITCH CURRENT (A) The LM2575 step–down converter can operate in both the continuous and the discontinuous modes of operation. The regulator works in the continuous mode when loads are relatively heavy, the current flows through the inductor continuously and never falls to zero. Under light load conditions, the circuit will be forced to the discontinuous mode when inductor current falls to zero for certain period of time (see Figure 22 and Figure 23). Each mode has distinctively different operating characteristics, which can affect the regulator performance and requirements. In many cases the preferred mode of operation is the continuous mode. It offers greater output power, lower peak currents in the switch, inductor and diode, and can have a lower output ripple voltage. On the other hand it does require larger inductor values to keep the inductor current flowing continuously, especially at low output load currents and/or high input voltages. To simplify the inductor selection process, an inductor selection guide for the LM2575 regulator was added to this data sheet (Figures 17 through 21). This guide assumes that the regulator is operating in the continuous mode, and selects an inductor that will allow a peak–to–peak inductor ripple current to be a certain percentage of the maximum design load current. This percentage is allowed to change as different design load currents are selected. For light loads (less than approximately 200 mA) it may be desirable to operate the regulator in the discontinuous mode, because the inductor value and size can be kept relatively low. Consequently, the percentage of inductor peak–to–peak current increases. This discontinuous mode of operation is perfectly acceptable for this type of switching converter. Any buck regulator will be forced to enter discontinuous mode if the load current is light enough. 1.0 0.1 0 INDUCTOR CURRENT (A) HORIZONTAL TIME BASE: 5.0 µs/DIV 0 0.1 Figure 22. Continuous Mode Switching Current Waveforms 0 HORIZONTAL TIME BASE: 5.0 µs/DIV Selecting the Right Inductor Style Some important considerations when selecting a core type are core material, cost, the output power of the power supply, the physical volume the inductor must fit within, and the Figure 23. Discontinuous Mode Switching Current Waveforms http://onsemi.com 16 LM2575 GENERAL RECOMMENDATIONS Output Voltage Ripple and Transients Source of the Output Ripple Heatsinking and Thermal Considerations The Through–Hole Package TO–220 Since the LM2575 is a switch mode power supply regulator, its output voltage, if left unfiltered, will contain a sawtooth ripple voltage at the switching frequency. The output ripple voltage value ranges from 0.5% to 3% of the output voltage. It is caused mainly by the inductor sawtooth ripple current multiplied by the ESR of the output capacitor. The LM2575 is available in two packages, a 5–pin TO–220(T, TV) and a 5–pin surface mount D2PAK(D2T). There are many applications that require no heatsink to keep the LM2575 junction temperature within the allowed operating range. The TO–220 package can be used without a heatsink for ambient temperatures up to approximately 50°C (depending on the output voltage and load current). Higher ambient temperatures require some heatsinking, either to the printed circuit (PC) board or an external heatsink. Short Voltage Spikes and How to Reduce Them The regulator output voltage may also contain short voltage spikes at the peaks of the sawtooth waveform (see Figure 24). These voltage spikes are present because of the fast switching action of the output switch, and the parasitic inductance of the output filter capacitor. There are some other important factors such as wiring inductance, stray capacitance, as well as the scope probe used to evaluate these transients, all these contribute to the amplitude of these spikes. To minimize these voltage spikes, low inductance capacitors should be used, and their lead lengths must be kept short. The importance of quality printed circuit board layout design should also be highlighted. The Surface Mount Package D 2PAK and its Heatsinking The other type of package, the surface mount D2PAK, is designed to be soldered to the copper on the PC board. The copper and the board are the heatsink for this package and the other heat producing components, such as the catch diode and inductor. The PC board copper area that the package is soldered to should be at least 0.4 in2 (or 100 mm2) and ideally should have 2 or more square inches (1300 mm2) of 0.0028 inch copper. Additional increasing of copper area beyond approximately 3.0 in2 (2000 mm2) will not improve heat dissipation significantly. If further thermal improvements are needed, double sided or multilayer PC boards with large copper areas should be considered. Voltage spikes caused by switching action of the output switch and the parasitic inductance of the output capacitor UNFILTERED OUTPUT VOLTAGE Thermal Analysis and Design The following procedure must be performed to determine whether or not a heatsink will be required. First determine: 1. PD(max) maximum regulator power dissipation in the application. 2. TA(max) maximum ambient temperature in the application. 3. TJ(max) maximum allowed junction temperature (125°C for the LM2575). For a conservative design, the maximum junction temperature should not exceed 110°C to assure safe operation. For every additional 10°C temperature rise that the junction must withstand, the estimated operating lifetime of the component is halved. 4. RθJC package thermal resistance junction–case. 5. RθJA package thermal resistance junction–ambient. (Refer to Absolute Maximum Ratings in this data sheet or RθJC and RθJA values). VERTICAL RESOLUTION: 20 mV/DIV FILTERED OUTPUT VOLTAGE HORIZONTAL TIME BASE: 10 µs/DIV Figure 24. Output Ripple Voltage Waveforms Minimizing the Output Ripple In order to minimize the output ripple voltage it is possible to enlarge the inductance value of the inductor L1 and/or to use a larger value output capacitor. There is also another way to smooth the output by means of an additional LC filter (20 µH, 100 µF), that can be added to the output (see Figure 33) to further reduce the amount of output ripple and transients. With such a filter it is possible to reduce the output ripple voltage transients 10 times or more. Figure 24 shows the difference between filtered and unfiltered output waveforms of the regulator shown in Figure 33. The upper waveform is from the normal unfiltered output of the converter, while the lower waveform shows the output ripple voltage filtered by an additional LC filter. The following formula is to calculate the total power dissipated by the LM2575: PD = (Vin x IQ) + d x ILoad x Vsat where d is the duty cycle and for buck converter http://onsemi.com 17 LM2575 Unregulated DC Input 12 V to 25 V V t d on O , V T in IQ Vin VO ILoad Cin 100 µF /50 V (quiescent current) and Vsat can be found in the LM2575 data sheet, is minimum input voltage applied, is the regulator output voltage, is the load current. Feedback +Vin LM2575–12 1 3 Gnd 5 4 Output 2 ON/OFF D1 1N5819 Cout 1800 µF /16 V Regulated Output -12 V @ 0.35 A The dynamic switching losses during turn–on and turn–off can be neglected if proper type catch diode is used. Figure 25. Inverting Buck–Boost Regulator Using the LM2575–12 Develops –12 V @ 0.35 A Packages Not on a Heatsink (Free–Standing) For a free–standing application when no heatsink is used, the junction temperature can be determined by the following expression: ADDITIONAL APPLICATIONS Inverting Regulator TJ = (RθJA) (PD) + TA An inverting buck–boost regulator using the LM2575–12 is shown in Figure 25. This circuit converts a positive input voltage to a negative output voltage with a common ground by bootstrapping the regulators ground to the negative output voltage. By grounding the feedback pin, the regulator senses the inverted output voltage and regulates it. In this example the LM2575–12 is used to generate a –12 V output. The maximum input voltage in this case cannot exceed +28 V because the maximum voltage appearing across the regulator is the absolute sum of the input and output voltages and this must be limited to a maximum of 40 V. This circuit configuration is able to deliver approximately 0.35 A to the output when the input voltage is 12 V or higher. At lighter loads the minimum input voltage required drops to approximately 4.7 V, because the buck–boost regulator topology can produce an output voltage that, in its absolute value, is either greater or less than the input voltage. Since the switch currents in this buck–boost configuration are higher than in the standard buck converter topology, the available output current is lower. This type of buck–boost inverting regulator can also require a larger amount of startup input current, even for light loads. This may overload an input power source with a current limit less than 1.5 A. Such an amount of input startup current is needed for at least 2.0 ms or more. The actual time depends on the output voltage and size of the output capacitor. Because of the relatively high startup currents required by this inverting regulator topology, the use of a delayed startup or an undervoltage lockout circuit is recommended. where (RθJA)(PD) represents the junction temperature rise caused by the dissipated power and TA is the maximum ambient temperature. Packages on a Heatsink If the actual operating junction temperature is greater than the selected safe operating junction temperature determined in step 3, than a heatsink is required. The junction temperature will be calculated as follows: TJ = PD (RθJA + RθCS + RθSA) + TA where L1 100 µH RθJC is the thermal resistance junction–case, RθCS is the thermal resistance case–heatsink, RθSA is the thermal resistance heatsink–ambient. If the actual operating temperature is greater than the selected safe operating junction temperature, then a larger heatsink is required. Some Aspects That can Influence Thermal Design It should be noted that the package thermal resistance and the junction temperature rise numbers are all approximate, and there are many factors that will affect these numbers, such as PC board size, shape, thickness, physical position, location, board temperature, as well as whether the surrounding air is moving or still. Other factors are trace width, total printed circuit copper area, copper thickness, single– or double–sided, multilayer board, the amount of solder on the board or even color of the traces. The size, quantity and spacing of other components on the board can also influence its effectiveness to dissipate the heat. http://onsemi.com 18 LM2575 Using a delayed startup arrangement, the input capacitor can charge up to a higher voltage before the switch–mode regulator begins to operate. The high input current needed for startup is now partially supplied by the input capacitor Cin. +Vin Shutdown Input 5.0 V The inverting regulator operates in a different manner than the buck converter and so a different design procedure has to be used to select the inductor L1 or the output capacitor Cout. The output capacitor values must be larger than is normally required for buck converter designs. Low input voltages or high output currents require a large value output capacitor (in the range of thousands of µF). The recommended range of inductor values for the inverting converter design is between 68 µH and 220 µH. To select an inductor with an appropriate current rating, the inductor peak current has to be calculated. The following formula is used to obtain the peak inductor current: 0 Off LM2575–12 5 R1 47 k ON/OFF 3 R2 47 k 4 Output Figure 27. Inverting Buck–Boost Regulator Shut Down Circuit Using an Optocoupler With the inverting configuration, the use of the ON/OFF pin requires some level shifting techniques. This is caused by the fact, that the ground pin of the converter IC is no longer at ground. Now, the ON/OFF pin threshold voltage (1.4 V approximately) has to be related to the negative output voltage level. There are many different possible shut down methods, two of them are shown in Figures 27 and 28. +V 0 +Vin Off On Shutdown Input R2 5.6 k +Vin 1 L1 100 µH Q1 2N3906 Cout 1800 µF /16 V D1 1N5819 Gnd R2 47 k LM2575–XX Cin 100 µF 2 Gnd ON/OFF 3 NOTE: This picture does not show the complete circuit. Feedback 1 R3 470 On Under normal continuous inductor current operating conditions, the worst case occurs when Vin is minimal. Note that the voltage appearing across the regulator is the absolute sum of the input and output voltage, and must not exceed 40 V. Cin C1 100 µF /50 V 0.1 µF 5 MOC8101 I (V |V |) O V in x t on Load in 2L 1 V in |V | O where t on x 1 , and f osc 52 kHz. V |V | f osc in O +Vin Cin R1 100 µF 47 k -Vout peak Unregulated DC Input 12 V to 25 V LM2575–XX 1 Design Recommendations: I +Vin 5 ON/OFF 3 R1 12 k Gnd -Vout NOTE: This picture does not show the complete circuit. Regulated Output -12 V @ 0.35 A Figure 28. Inverting Buck–Boost Regulator Shut Down Circuit Using a PNP Transistor Figure 26. Inverting Buck–Boost Regulator with Delayed Startup Negative Boost Regulator This example is a variation of the buck–boost topology and is called a negative boost regulator. This regulator experiences relatively high switch current, especially at low input voltages. The internal switch current limiting results in lower output load current capability. The circuit in Figure 29 shows the negative boost configuration. The input voltage in this application ranges from –5.0 V to –12 V and provides a regulated –12 V output. It has been already mentioned above, that in some situations, the delayed startup or the undervoltage lockout features could be very useful. A delayed startup circuit applied to a buck–boost converter is shown in Figure 26. Figure 32 in the “Undervoltage Lockout” section describes an undervoltage lockout feature for the same converter topology. http://onsemi.com 19 LM2575 cause some problems by coupling the ripple into the ON/OFF pin, the regulator could be switched periodically on and off with the line (or double) frequency. If the input voltage is greater than –12 V, the output will rise above –12 V accordingly, but will not damage the regulator. Cout 1000 µF /16 V 4 +Vin LM2575–12 1 Cin 100 µF /50 V Feedback Output 3 Gnd 5 L1 Unregulated DC Input -Vin = -5.0 V to -12 V 150 µH 2 ON/OFF +Vin +Vin D1 1N5817 LM2575–XX 1 C1 0.1 µF Regulated Output Vout = -12 V Cin 100 µF 5 ON/OFF 3 R1 47 k Load Current from 200 mA for Vin = -5.2 V to 500 mA for Vin = -7.0 V Gnd R2 47 k NOTE: This picture does not show the complete circuit. Figure 30. Delayed Startup Circuitry Figure 29. Negative Boost Regulator Undervoltage Lockout Design Recommendations: Some applications require the regulator to remain off until the input voltage reaches a certain threshold level. Figure 31 shows an undervoltage lockout circuit applied to a buck regulator. A version of this circuit for buck–boost converter is shown in Figure 32. Resistor R3 pulls the ON/OFF pin high and keeps the regulator off until the input voltage reaches a predetermined threshold level, which is determined by the following expression: The same design rules as for the previous inverting buck–boost converter can be applied. The output capacitor Cout must be chosen larger than would be required for a standard buck converter. Low input voltages or high output currents require a large value output capacitor (in the range of thousands of µF). The recommended range of inductor values for the negative boost regulator is the same as for inverting converter design. Another important point is that these negative boost converters cannot provide current limiting load protection in the event of a short in the output so some other means, such as a fuse, may be necessary to provide the load protection. V th V Z1 (Q1) 1 R2 V R1 BE +Vin +Vin Delayed Startup 1 There are some applications, like the inverting regulator already mentioned above, which require a higher amount of startup current. In such cases, if the input power source is limited, this delayed startup feature becomes very useful. To provide a time delay between the time the input voltage is applied and the time when the output voltage comes up, the circuit in Figure 30 can be used. As the input voltage is applied, the capacitor C1 charges up, and the voltage across the resistor R2 falls down. When the voltage on the ON/OFF pin falls below the threshold value 1.4 V, the regulator starts up. Resistor R1 is included to limit the maximum voltage applied to the ON/OFF pin, reduces the power supply noise sensitivity, and also limits the capacitor C1 discharge current, but its use is not mandatory. When a high 50 Hz or 60 Hz (100 Hz or 120 Hz respectively) ripple voltage exists, a long delay time can R2 10 k Cin 100 µF 5 R3 47 k LM2575–5.0 ON/OFF 3 Gnd Z1 1N5242B Q1 2N3904 R1 10 k Vth ≈ 13 V NOTE: This picture does not show the complete circuit. Figure 31. Undervoltage Lockout Circuit for Buck Converter http://onsemi.com 20 LM2575 Adjustable Output, Low–Ripple Power Supply +Vin +Vin LM2575–5.0 1 R2 15 k Cin 100 µF 5 R3 68 k A 1.0 A output current capability power supply that features an adjustable output voltage is shown in Figure 33. This regulator delivers 1.0 A into 1.2 V to 35 V output. The input voltage ranges from roughly 8.0 V to 40 V. In order to achieve a 10 or more times reduction of output ripple, an additional L–C filter is included in this circuit. ON/OFF 3 Gnd Z1 1N5242B Vth ≈ 13 V Q1 2N3904 R1 15 k Vout = -5.0 V NOTE: This picture does not show the complete circuit. Figure 32. Undervoltage Lockout Circuit for Buck–Boost Converter Feedback 4 +Vin L1 150 µH LM2575–Adj 1 Output 3 Gnd 5 L2 20 µH 2 ON/OFF Cin 100 µF /50 V C1 100 µF Cout 2200 µF D1 1N5819 R1 1.1 k Optional Output Ripple Filter Figure 33. Adjustable Power Supply with Low Ripple Voltage JUNCTIONTOAIR (° C/W) R θ JA, THERMAL RESISTANCE 80 3.0 Free Air Mounted Vertically 60 ÎÎÎÎ ÎÎÎÎ ÎÎÎÎ ÎÎÎÎ 2.0 oz. Copper L Minimum Size Pad 50 2.5 2.0 L 40 1.5 RθJA 30 3.5 PD(max) for TA = 50°C 70 0 5.0 10 15 20 25 30 1.0 L, LENGTH OF COPPER (mm) Figure 34. D2PAK Thermal Resistance and Maximum Power Dissipation versus P.C.B. Copper Length http://onsemi.com 21 Regulated Output Voltage 1.2 V to 35 V @1.0 A R2 50 k PD, MAXIMUM POWER DISSIPATION (W) Unregulated DC Input + LM2575 THE LM2575–5.0 STEP–DOWN VOLTAGE REGULATOR WITH 5.0 V @ 1.0 A OUTPUT POWER CAPABILITY. TYPICAL APPLICATION WITH THROUGH–HOLE PC BOARD LAYOUT Feedback 4 +Vin Unregulated DC Input +Vin = +7.0 V to +40 V LM2575–5.0 1 Output 3 C1 100 µF /50 V Gnd L1 330 µH Regulated Output +Vout1 = 5.0 V @ 1.0 A 2 ON/OFF 5 J1 D1 1N5819 Gndin Cout 330 µF /16 V Gndout C1 C2 D1 L1 – – – – 100 µF, 50 V, Aluminium Electrolytic 330 µF, 16 V, Aluminium Electrolytic 1.0 A, 40 V, Schottky Rectifier, 1N5819 330 µH, Tech 39: 77 458 BV, Toroid Core, Through–Hole, Pin 3 = Start, Pin 7 = Finish Figure 35. Schematic Diagram of the LM2575–5.0 Step–Down Converter Gndin C1 L1 Gndout U1 LM2575 D1 J1 C2 DC-DC Converter +Vout1 +Vin NOTE: Not to scale. NOTE: Not to scale. Figure 36. Printed Circuit Board Component Side Figure 37. Printed Circuit Board Copper Side http://onsemi.com 22 LM2575 THE LM2575–ADJ STEP–DOWN VOLTAGE REGULATOR WITH 8.0 V @ 1.0 A OUTPUT POWER CAPABILITY. TYPICAL APPLICATION WITH THROUGH–HOLE PC BOARD LAYOUT Regulated Output Unfiltered Vout1 = 8.0 V @1.0 A 4 Unregulated DC Input +Vin LM2575–Adj 1 +Vin = +10 V to + 40 V Feedback Output 3 Gnd 5 L1 330 µH L2 25 µH Vout2 = 8.0 V @1.0 A 2 ON/OFF C1 100 µF /50 V R2 10 k D1 1N5819 C2 330 µF /16 V – – – – – – – – C3 100 µF /16 V R1 1.8 k V C1 C2 C3 D1 L1 L2 R1 R2 Regulated Output Filtered R2 out V ref 1 R1 Vref = 1.23 V 100 µF, 50 V, Aluminium Electrolytic R1 is between 1.0 k and 5.0 k 330 µF, 16 V, Aluminium Electrolytic 100 µF, 16 V, Aluminium Electrolytic 1.0 A, 40 V, Schottky Rectifier, 1N5819 330 µH, Tech 39: 77 458 BV, Toroid Core, Through–Hole, Pin 3 = Start, Pin 7 = Finish 25 µH, TDK: SFT52501, Toroid Core, Through–Hole 1.8 k 10 k Figure 38. Schematic Diagram of the 8.0 V @ 1.0 V Step–Down Converter Using the LM2575–Adj (An additional LC filter is included to achieve low output ripple voltage) Gndin Gndout U1 LM2575 C3 C1 L1 D1 C2 J1 L2 +Vin +Vout2 +Vout1 R2 R1 NOTE: Not to scale. NOTE: Not to scale. Figure 39. PC Board Component Side Figure 40. PC Board Copper Side References • • • • National Semiconductor LM2575 Data Sheet and Application Note National Semiconductor LM2595 Data Sheet and Application Note Marty Brown “Practical Switching Power Supply Design”, Academic Press, Inc., San Diego 1990 Ray Ridley “High Frequency Magnetics Design”, Ridley Engineering, Inc. 1995 http://onsemi.com 23 LM2575 ORDERING INFORMATION Device Nominal Output Voltage Operating Temperature Range LM2575TV–ADJ TO–220 (Straight Lead) 1 23 V to 37 V 1.23 D2PAK (Surface Mount) LM2575D2T–ADJR4 D2PAK (Surface Mount) LM2575TV–3.3 TO–220 (Vertical Mount) LM2575T–3.3 TO–220 (Straight Lead) 33V 3.3 D2PAK (Surface Mount) LM2575D2T–3.3R4 D2PAK (Surface Mount) LM2575TV–5 TO–220 (Vertical Mount) LM2575D2T–3.3 LM2575T–5 LM2575D2T–5 TJ = –40° 40° to +125°C D2PAK (Surface Mount) D2PAK (Surface Mount) LM2575TV–12 TO–220 (Vertical Mount) LM2575T–12 800 Tape & Reel TO–220 (Straight Lead) 12 V D2PAK (Surface Mount) LM2575D2T–12R4 D2PAK (Surface Mount) LM2575TV–15 TO–220 (Vertical Mount) LM2575T–15 LM2575D2T–15 50 Units/Rail TO–220 (Straight Lead) 50V 5.0 LM2575D2T–5R4 LM2575D2T–12 Shipping TO–220 (Vertical Mount) LM2575T–ADJ LM2575D2T–ADJ Package TO–220 (Straight Lead) 15 V D2PAK (Surface Mount) D2PAK (Surface Mount) LM2575D2T–15R4 MARKING DIAGRAMS TO–220 TV SUFFIX CASE 314B TO–220 T SUFFIX CASE 314D LM 2575T–xxx AWLYWW LM 2575T–xxx AWLYWW D2PAK D2T SUFFIX CASE 936A LM 2575–xxx AWLYWW 1 1 5 1 xxx A WL Y WW 5 = 3.3, 5.0, 12, 15, or ADJ = Assembly Location = Wafer Lot = Year = Work Week http://onsemi.com 24 5 50 Units/Rail LM2575 PACKAGE DIMENSIONS TO–220 TV SUFFIX CASE 314B–05 ISSUE J Q OPTIONAL CHAMFER E A U K NOTES: 1. DIMENSIONING AND TOLERANCING PER ANSI Y14.5M, 1982. 2. CONTROLLING DIMENSION: INCH. 3. DIMENSION D DOES NOT INCLUDE INTERCONNECT BAR (DAMBAR) PROTRUSION. DIMENSION D INCLUDING PROTRUSION SHALL NOT EXCEED 0.043 (1.092) MAXIMUM. C B –P– S L W DIM A B C D E F G H J K L N Q S U V W V F 5X G 5X 0.24 (0.610) M J T H D 0.10 (0.254) M T P N M –T– SEATING PLANE INCHES MIN MAX 0.572 0.613 0.390 0.415 0.170 0.180 0.025 0.038 0.048 0.055 0.850 0.935 0.067 BSC 0.166 BSC 0.015 0.025 0.900 1.100 0.320 0.365 0.320 BSC 0.140 0.153 --0.620 0.468 0.505 --0.735 0.090 0.110 MILLIMETERS MIN MAX 14.529 15.570 9.906 10.541 4.318 4.572 0.635 0.965 1.219 1.397 21.590 23.749 1.702 BSC 4.216 BSC 0.381 0.635 22.860 27.940 8.128 9.271 8.128 BSC 3.556 3.886 --- 15.748 11.888 12.827 --- 18.669 2.286 2.794 TO–220 T SUFFIX CASE 314D–04 ISSUE E –T– –Q– SEATING PLANE C B E A U L J H G D DIM A B C D E G H J K L Q U 1234 5 K 5 PL 0.356 (0.014) M T Q NOTES: 1. DIMENSIONING AND TOLERANCING PER ANSI Y14.5M, 1982. 2. CONTROLLING DIMENSION: INCH. 3. DIMENSION D DOES NOT INCLUDE INTERCONNECT BAR (DAMBAR) PROTRUSION. DIMENSION D INCLUDING PROTRUSION SHALL NOT EXCEED 10.92 (0.043) MAXIMUM. M http://onsemi.com 25 INCHES MIN MAX 0.572 0.613 0.390 0.415 0.170 0.180 0.025 0.038 0.048 0.055 0.067 BSC 0.087 0.112 0.015 0.025 0.990 1.045 0.320 0.365 0.140 0.153 0.105 0.117 MILLIMETERS MIN MAX 14.529 15.570 9.906 10.541 4.318 4.572 0.635 0.965 1.219 1.397 1.702 BSC 2.210 2.845 0.381 0.635 25.146 26.543 8.128 9.271 3.556 3.886 2.667 2.972 LM2575 PACKAGE DIMENSIONS D2PAK D2T SUFFIX CASE 936A–02 ISSUE B –T– OPTIONAL CHAMFER A TERMINAL 6 E U S K B V H NOTES: 1. DIMENSIONING AND TOLERANCING PER ANSI Y14.5M, 1982. 2. CONTROLLING DIMENSION: INCH. 3. TAB CONTOUR OPTIONAL WITHIN DIMENSIONS A AND K. 4. DIMENSIONS U AND V ESTABLISH A MINIMUM MOUNTING SURFACE FOR TERMINAL 6. 5. DIMENSIONS A AND B DO NOT INCLUDE MOLD FLASH OR GATE PROTRUSIONS. MOLD FLASH AND GATE PROTRUSIONS NOT TO EXCEED 0.025 (0.635) MAXIMUM. 1 2 3 4 5 M D 0.010 (0.254) M T L P N G R C http://onsemi.com 26 DIM A B C D E G H K L M N P R S U V INCHES MIN MAX 0.386 0.403 0.356 0.368 0.170 0.180 0.026 0.036 0.045 0.055 0.067 BSC 0.539 0.579 0.050 REF 0.000 0.010 0.088 0.102 0.018 0.026 0.058 0.078 5 REF 0.116 REF 0.200 MIN 0.250 MIN MILLIMETERS MIN MAX 9.804 10.236 9.042 9.347 4.318 4.572 0.660 0.914 1.143 1.397 1.702 BSC 13.691 14.707 1.270 REF 0.000 0.254 2.235 2.591 0.457 0.660 1.473 1.981 5 REF 2.946 REF 5.080 MIN 6.350 MIN LM2575 Notes http://onsemi.com 27 LM2575 ON Semiconductor and are trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages. “Typical” parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. SCILLC does not convey any license under its patent rights nor the rights of others. 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PUBLICATION ORDERING INFORMATION Literature Fulfillment: Literature Distribution Center for ON Semiconductor P.O. Box 5163, Denver, Colorado 80217 USA Phone: 303–675–2175 or 800–344–3860 Toll Free USA/Canada Fax: 303–675–2176 or 800–344–3867 Toll Free USA/Canada Email: [email protected] JAPAN: ON Semiconductor, Japan Customer Focus Center 4–32–1 Nishi–Gotanda, Shinagawa–ku, Tokyo, Japan 141–0031 Phone: 81–3–5740–2700 Email: [email protected] ON Semiconductor Website: http://onsemi.com For additional information, please contact your local Sales Representative. N. American Technical Support: 800–282–9855 Toll Free USA/Canada http://onsemi.com 28 LM2575/D