LM4752 Stereo 11W Audio Power Amplifier General Description The LM4752 is a stereo audio amplifier capable of delivering 11W per channel of continuous average output power to a 4Ω load, or 7W per channel into 8Ω using a single 24V supply at 10% THD+N. The LM4752 is specifically designed for single supply operation and a low external component count. The gain and bias resistors are integrated on chip, resulting in a 11W stereo amplifier in a compact 7 pin TO220 package. High output power levels at both 20V and 24V supplies and low external component count offer high value for compact stereo and TV applications. A simple mute function can be implemented with the addition of a few external components. n PO at 10% THD @ 1 kHz into 8Ω bridged TO-263 pkg. at VCC = 12V 5W (typ) Features n n n n n n n n Drives 4Ω and 8Ω loads Internal gain resistors (AV = 34 dB) Minimum external component requirement Single supply operation Internal current limiting Internal thermal protection Compact 7 lead TO-220 package Low cost-per-watt Applications Key Specifications n Output power at 10% THD+N with 1 kHz into 4Ω at V CC = 24V 11W (typ) n Output power at 10% THD+N with 1 kHz into 8Ω at V CC = 24V 7W (typ) n Closed loop gain 34 dB (typ) n PO at 10% THD @ 1 kHz into 4Ω Single-ended TO-263 pkg. at VCC = 12V 2.5W (typ) Typical Application n n n n Compact stereos Stereo TVs Mini component stereos Multimedia speakers Connection Diagram Plastic Package DS100039-2 Package Description Top View Order Number LM4752T Package Number TA07B DS100039-50 DS100039-1 FIGURE 1. Typical Audio Amplifier Application Circuit © 1999 National Semiconductor Corporation DS100039 Package Description Top View Order Number LM4752TS Package Number TS07B www.national.com LM4752 Stereo 11W Audio Power Amplifier February 1999 Absolute Maximum Ratings (Note 2) Storage Temperature If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. Supply Voltage Operating Ratings Temperature Range TMIN ≤ TA ≤ TMAX 40V ± 0.7V Input Voltage Output Current −40˚C to 150˚C −40˚C ≤ TA ≤ +85˚C Supply Voltage 9V to 32V Internally Limited θJC 2˚C/W 62.5W θJA 79˚C/W Power Dissipation (Note 3) ESD Susceptability (Note 4) 2 kV Junction Temperature 150˚C Soldering Information T Package (10 sec) 250˚C Electrical Characteristics The following specifications apply to each channel with VCC = 24V, TA = 25˚C unless otherwise specified. LM4752 Symbol Itotal Po Parameter Conditions Total Quiescent Power Supply Current VINAC = 0V, Vo = 0V, RL = ∞ Output Power (Continuous f = 1 kHz, THD+N = 10%, RL = 8Ω Average per Channel) f = 1 kHz, THD+N = 10%, RL = 4Ω THD+N Total Harmonic Distortion plus Noise VOSW Output Swing Units (Limits) Typical (Note 5) Limit (Note 6) 10.5 20 mA(max) 7 mA(min) 10 W(min) 7 W VCC = 20V, RL = 8Ω 4 W VCC = 20V, R 7 W f = 1 kHz, THD+N = 10%, RL = 4Ω VS = 12V, TO-263 Pkg. 2.5 W f = 1 kHz, Po = 1 W/ch, RL = 8Ω 0.08 % V L = 4Ω RL = 8Ω, V CC = 20V 15 RL = 4Ω, V CC = 20V 14 V 55 dB 50 dB Xtalk Channel Separation See Figure 1 PSRR Power Supply Rejection Ratio See Figure 1 f = 1 kHz, Vo = 4 Vrms, RL = 8Ω VCC = 22V to 26V, R L = 8Ω VODV Differential DC Output Offset Voltage SR Slew Rate 2 RIN Input Impedance 83 kΩ PBW Power Bandwidth 3 dB BW at Po = 2.5W, RL = 8Ω 65 kHz A VCL Closed Loop Gain (Internally Set) RL = 8Ω 34 ein Noise VINAC = 0V 0.09 IHF-A Weighting Filter, RL = 8Ω 0.4 V(max) V/µs 33 dB(min) 35 dB(max) 0.2 mVrms Output Referred Io Output Short Circuit Current Limit VIN = 0.5V, R L = 2Ω 2 A(min) Note 1: All voltages are measured with respect to the GND pin (4), unless otherwise specified. Note 2: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical specifications under particular test conditions which guarantee specific performance limits. This assumes that the device is within the Operating Ratings. Specifications are not guaranteed for parameters where no limit is given, however, the typical value is a good indication of device performance. Note 3: For operating at case temperatures above 25˚C, the device must be derated based on a 150˚C maximum junction temperature and a thermal resistance of θJC = 2˚C/W (junction to case). Refer to the section Determining the Maximum Power Dissipation in the Application Information section for more information. Note 4: Human body model, 100 pF discharged through a 1.5 kΩ resistor. Note 5: Typicals are measured at 25˚C and represent the parametric norm. Note 6: Limits are guarantees that all parts are tested in production to meet the stated values. Note 7: The TO-263 Package is not recommended for VS > 16V due to impractical heatsinking limitations. www.national.com 2 Test Circuit DS100039-36 FIGURE 2. Test Circuit 3 www.national.com Typical Application with Mute DS100039-3 FIGURE 3. Application with Mute Function www.national.com 4 DS100039-4 Equivalent Schematic Diagram 5 www.national.com System Application Circuit DS100039-5 FIGURE 4. Circuit for External Components Description External Components Description Components Function Description 1, 2 Cs Provides power supply filtering and bypassing. 3, 4 Rsn Works with Csn to stabilize the output stage from high frequency oscillations. 5, 6 Csn Works with Rsn to stabilize the output stage from high frequency oscillations. 7 Cb Provides filtering for the internally generated half-supply bias generator. 8, 9 Ci Input AC coupling capacitor which blocks DC voltage at the amplifier’s input terminals. Also creates a high pass filter with fc=1/(2 • π • Rin • Cin). 10, 11 Co Output AC coupling capacitor which blocks DC voltage at the amplifier’s output terminal. Creates a high pass filter with fc=1/(2 • π • Rout • Cout). 12, 13 Ri Voltage control - limits the voltage level to the amplifier’s input terminals. www.national.com 6 Typical Performance Characteristics THD+N vs Output Power THD+N vs Output Power DS100039-12 THD+N vs Output Power THD+N vs Output Power DS100039-13 THD+N vs Output Power DS100039-6 THD+N vs Output Power THD+N vs Output Power DS100039-7 THD+N vs Output Power DS100039-15 THD+N vs Output Power DS100039-14 DS100039-8 THD+N vs Output Power DS100039-16 THD+N vs Output Power DS100039-9 THD+N vs Output Power DS100039-10 7 DS100039-17 DS100039-11 www.national.com Typical Performance Characteristics (Continued) THD+N vs Output Power THD+N vs Output Power DS100039-38 THD+N vs Output Power DS100039-39 THD+N vs Output Power DS100039-41 THD+N vs Output Power THD+N vs Output Power THD+N vs Output Power THD+N vs Output Power DS100039-40 DS100039-42 DS100039-44 DS100039-43 THD+N vs Output Power DS100039-45 THD+N vs Output Power DS100039-47 www.national.com THD+N vs Output Power THD+N vs Output Power DS100039-48 8 DS100039-46 DS100039-49 Typical Performance Characteristics Output Power vs Supply Voltage (Continued) Output Power vs Supply Voltage DS100039-18 THD+N vs Frequency DS100039-19 THD+N vs Frequency DS100039-21 Channel Separation Frequency Response DS100039-20 Frequency Response DS100039-23 DS100039-22 PSRR vs Frequency DS100039-24 Supply Current vs Supply Voltage DS100039-25 DS100039-26 Power Derating Curve Power Dissipation vs Output Power DS100039-28 DS100039-27 9 Power Dissipation vs Output Power DS100039-29 www.national.com Typical Performance Characteristics Power Dissipation vs Output Power (Continued) Power Dissipation vs Output Power DS100039-51 DS100039-52 differential pair, resulting in an output DC shift towards V SUPPLY. An R-C timing circuit should be used to limit the pull-down time such that output “pops” and signal feedthroughs will be minimized. The pull-down timing is a function of a number of factors, including the external mute circuitry, the voltage used to activate the mute, the bias capacitor, the half-supply voltage, and internal resistances used in the half-supply generator. Table 1 shows a list of recommended values for the external mute circuitry. Application Information CAPACITOR SELECTION AND FREQUENCY RESPONSE With the LM4752, as in all single supply amplifiers, AC coupling capacitors are used to isolate the DC voltage present at the inputs (pins 2,6) and outputs (pins 1,7). As mentioned earlier in the External Components section these capacitors create high-pass filters with their corresponding input/ output impedances. The Typical Application Circuit shown in Figure 1 shows input and output capacitors of 0.1 µF and 1,000 µF respectively. At the input, with an 83 kΩ typical input resistance, the result is a high pass 3 dB point occurring at 19 Hz. There is another high pass filter at 39.8 Hz created with the output load resistance of 4Ω. Careful selection of these components is necessary to ensure that the desired frequency response is obtained. The Frequency Response curves in the Typical Performance Characteristics section show how different output coupling capacitors affect the low frequency rolloff. TABLE 1. Values for Mute Circuit VMUTE R2 C1 R3 CB VCC 5V 10 kΩ 10 kΩ 4.7 µF 360Ω 100 µF 21V–32V VS 20 kΩ 1.2 kΩ 4.7 µF 180Ω 100 µF 15V–32V VS 20 kΩ 910Ω 4.7 µF 180Ω 47 µF 22V–32V OPERATING IN BRIDGE-MODE Though designed for use as a single-ended amplifier, the LM4752 can be used to drive a load differentially (bridgemode). Due to the low pin count of the package, only the non-inverting inputs are available. An inverted signal must be provided to one of the inputs. This can easily be done with the use of an inexpensive op-amp configured as a standard inverting amplifier. An LF353 is a good low-cost choice. Care must be taken, however, for a bridge-mode amplifier must theoretically dissipate four times the power of a single-ended type. The load seen by each amplifier is effectively half that of the actual load being used, thus an amplifier designed to drive a 4Ω load in single-ended mode should drive an 8Ω load when operating in bridge-mode. APPLICATION CIRCUIT WITH MUTE With the addition of a few external components, a simple mute circuit can be implemented, such as the one shown in Figure 3. This circuit works by externally pulling down the half supply bias line (pin 5), effectively shutting down the input stage. When using an external circuit to pull down the bias, care must be taken to ensure that this line is not pulled down too quickly, or output “pops” or signal feedthrough may result. If the bias line is pulled down too quickly, currents induced in the internal bias resistors will cause a momentary DC voltage to appear across the inputs of each amplifier’s internal www.national.com R1 10 Application Information (Continued) DS100039-30 FIGURE 5. Bridge-Mode Application DS100039-31 DS100039-37 FIGURE 6. THD+N vs. POUT for Bridge-Mode Application UNDERVOLTAGE SHUTDOWN PREVENTING OSCILLATIONS With the integration of the feedback and bias resistors onchip, the LM4752 fits into a very compact package. However, due to the close proximity of the non-inverting input pins to the corresponding output pins, the inputs should be AC terminated at all times. If the inputs are left floating, the amplifier will have a positive feedback path through high impedance coupling, resulting in a high frequency oscillation. In most applications, this termination is typically provided by the previous stage’s source impedance. If the application will require an external signal, the inputs should be terminated to ground with a resistance of 50 kΩ or less on the AC side of the input coupling capacitors. If the power supply voltage drops below the minimum operating supply voltage, the internal under-voltage detection circuitry pulls down the half-supply bias line, shutting down the preamp section of the LM4752. Due to the wide operating supply range of the LM4752, the threshold is set to just under 9V. There may be certain applications where a higher threshold voltage is desired. One example is a design requiring a high operating supply voltage, with large supply and bias capacitors, and there is little or no other circuitry connected to the main power supply rail. In this circuit, when the power is disconnected, the supply and bias capacitors will discharge at a slower rate, possibly resulting in audible output distortion as the decaying voltage begins to clip the out11 www.national.com Application Information When determining the proper heatsink, the above equation should be re-written as: (Continued) put signal. An external circuit may be used to sense for the desired threshold, and pull the bias line (pin5) to ground to disable the input preamp. Figure 7 shows an example of such a circuit. When the voltage across the zener diode drops below its threshold, current flow into the base of Q1 is interrupted. Q2 then turns on, discharging the bias capacitor. This discharge rate is governed by several factors, including the bias capacitor value, the bias voltage, and the resistor at the emitter of Q2. An equation for approximating the value of the emitter discharge resistor, R, is given below: θSA ≤ [ (TJ − TA) / PDMAX] − θ JC − θCS TO-263 HEATSINKING Surface mount applications will be limited by the thermal dissipation properties of printed circuit board area. The TO-263 package is not recommended for surface mount applications with VS > 16V due to limited printed circuit board area. There are TO-263 package enhancements, such as clip-on heatsinks and heatsinks with adhesives, that can be used to improve performance. Standard FR-4 single-sided copper clad will have an approximate Thermal resistance (θSA) ranging from: R = (0.7V) / (CB • (V S / 2) / 0.1s) Note that this is only a linearized approximation based on a discharge time of 0.1s. The circuit should be evaluated and adjusted for each application. As mentioned earlier in the Application Circuit with Mute section, when using an external circuit to pull down the bias line, the rate of discharge will have an effect on the turn-off induced distortions. Please refer to the Application Circuit with Mute section for more information. 1.5 x 1.5 in. sq. 20–27˚C/W (TA=28˚C, Sine wave 2 x 2 in. sq. 16–23˚C/W testing, 1 oz. Copper) The above values for θSA vary widely due to dimensional proportions (i.e. variations in width and length will vary θSA). For audio applications, where peak power levels are short in duration, this part will perform satisfactory with less heatsinking/copper clad area. As with any high power design proper bench testing should be undertaken to assure the design can dissipate the required power. Proper bench testing requires attention to worst case ambient temperature and air flow. At high power dissipation levels the part will show a tendency to increase saturation voltages, thus limiting the undistorted power levels. Determining Maximum Power Dissipation For a single-ended class AB power amplifier, the theoretical maximum power dissipation point is a function of the supply voltage, V S, and the load resistance, RL and is given by the following equation: (single channel) PDMAX (W) = [VS 2 / (2 • π2 • RL) ] The above equation is for a single channel class-AB power amplifier. For dual amplifiers such as the LM4752, the equation for calculating the total maximum power dissipated is: (dual channel) PDMAX (W) = 2 • [V S2 / (2 • π2 • RL) ] DS100039-32 FIGURE 7. External Undervoltage Pull-Down THERMAL CONSIDERATIONS Heat Sinking Proper heatsinking is necessary to ensure that the amplifier will function correctly under all operating conditions. A heatsink that is too small will cause the die to heat excessively and will result in a degraded output signal as the internal thermal protection circuitry begins to operate. The choice of a heatsink for a given application is dictated by several factors: the maximum power the IC needs to dissipate, the worst-case ambient temperature of the circuit, the junction-to-case thermal resistance, and the maximum junction temperature of the IC. The heat flow approximation equation used in determining the correct heatsink maximum thermal resistance is given below: TJ–TA = P DMAX • (θJC + θCS + θ or VS2 / (π 2 • RL) (Bridged Outputs) PDMAX (W) = 4[VS2 / (2π2 • RL)] Heatsink Design Example: Determine the system parameters: V = 24V Operating Supply Voltage Minimum load impedance TA = 55˚C Worst case ambient temperature Device parameters from the datasheet: SA) T where: J = 150˚C Maximum junction temperature θJC = 2˚C/W Junction-to-case thermal resistance Calculations: 2 • PDMAX = 2 • [V S2 / (2 • π2 • RL) ] = (24V)2 / (2 • π2 • 4Ω) = 14.6W θSA ≤ [ (TJ − TA) / PDMAX] − θ JC − θCS = [ (150˚C − 55˚C) / 14.6W ] − 2˚C/W − 0.2˚C/W = 4.3˚C/W Conclusion: Choose a heatsink with θSA ≤ 4.3˚C/W. PDMAX = maximum power dissipation of the IC TJ(˚C) = junction temperature of the IC TA(˚C) = ambient temperature θJC(˚C/W) = junction-to-case thermal resistance of the IC θCS(˚C/W) = case-to-heatsink thermal resistance (typically 0.2 to 0.5 ˚C/W) θSA(˚C/W) = thermal resistance of heatsink www.national.com S RL = 4Ω 12 Application Information Layout and Ground Returns Proper PC board layout is essential for good circuit performance. When laying out a PC board for an audio power amplifer, particular attention must be paid to the routing of the output signal ground returns relative to the input signal and bias capacitor grounds. To prevent any ground loops, the ground returns for the output signals should be routed separately and brought together at the supply ground. The input signal grounds and the bias capacitor ground line should also be routed separately. The 0.1 µF high frequency supply bypass capacitor should be placed as close as possible to the IC. (Continued) TO-263 HEATSINK DESIGN EXAMPLES: Example 1: (Stereo Single-Ended Output) Given: TA = 30˚C TJ = 150˚C RL = 4Ω VS = 12V θJC = 2˚C/W PDMAX from PD vs PO Graph: PDMAX ≈ 3.7W Calculating PDMAX: PDMAX = VCC2 / (π2RL) = (12V)2 / π2(4Ω)) = 3.65W Calculating Heatsink Thermal Resistance: θSA < [(T J − TA) / PDMAX] − θJC − θCS θSA < 120˚C / 3.7W − 2.0˚C/W − 0.2˚C/W = 30.2˚C/W Therefore the recommendation is to use 1.5 x 1.5 square inch of single-sided copper clad. Example 2: (Stereo Single-Ended Output) Given: TA = 50˚C TJ = 150˚C RL = 4Ω VS = 12V θJC = 2˚C/W PDMAX from PD vs PO Graph: PDMAX ≈ 3.7W Calculating PDMAX: PDMAX = VCC2 / (π2RL) = (12V)2 / (π2(4Ω)) = 3.65W Calculating Heatsink Thermal Resistance: θSA < [(TJ − TA) / PDMAX] − θJC − θCS θSA < 100˚C / 3.7W − 2.0˚C/W − 0.2˚C/W = 24.8˚C/W Therefore the recommendation is to use 2.0 x 2.0 square inch of single-sided copper clad. Example 3: (Bridged Output) Given: TA = 50˚C TJ = 150˚C RL = 8Ω VS = 12V θJC = 2˚C/W Calculating PDMAX: PDMAX = 4[VCC2 / (2π2RL)] = 4(12V)2 / (2π2(8Ω)) = 3.65W Calculating Heatsink Thermal Resistance: θSA < [(TJ − TA) / PDMAX] − θJC − θCS θSA < 100˚C / 3.7W − 2.0˚C/W − 0.2˚C/W = 24.8˚C/W Therefore the recommendation is to use 2.0 x 2.0 square inch of single-sided copper clad. 13 www.national.com Application Information (Continued) PC BOARD LAYOUT — COMPOSITE DS100039-33 www.national.com 14 Application Information (Continued) PC BOARD LAYOUT — SILK SCREEN DS100039-34 15 www.national.com Application Information (Continued) PC BOARD LAYOUT — SOLDER SIDE DS100039-35 www.national.com 16 Physical Dimensions inches (millimeters) unless otherwise noted Order Number LM4752T NS Package Number TA07B Order Number LM4752TS NS Package Number TS7B 17 www.national.com LM4752 Stereo 11W Audio Power Amplifier LIFE SUPPORT POLICY NATIONAL’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein: 2. A critical component is any component of a life support 1. Life support devices or systems are devices or sysdevice or system whose failure to perform can be reatems which, (a) are intended for surgical implant into sonably expected to cause the failure of the life support the body, or (b) support or sustain life, and whose faildevice or system, or to affect its safety or effectiveness. ure to perform when properly used in accordance with instructions for use provided in the labeling, can be reasonably expected to result in a significant injury to the user. National Semiconductor Corporation Americas Tel: 1-800-272-9959 Fax: 1-800-737-7018 Email: [email protected] www.national.com National Semiconductor Europe Fax: +49 (0) 1 80-530 85 86 Email: [email protected] Deutsch Tel: +49 (0) 1 80-530 85 85 English Tel: +49 (0) 1 80-532 78 32 Français Tel: +49 (0) 1 80-532 93 58 Italiano Tel: +49 (0) 1 80-534 16 80 National Semiconductor Asia Pacific Customer Response Group Tel: 65-2544466 Fax: 65-2504466 Email: [email protected] National Semiconductor Japan Ltd. Tel: 81-3-5639-7560 Fax: 81-3-5639-7507 National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves the right at any time without notice to change said circuitry and specifications.