LT3988 Dual 60V Monolithic 1A Step-Down Switching Regulator Description Features Wide Input Range: Operation from 4.1V to 60V Overvoltage Lockout Protects Circuit Through 80V Transients n Two 1A Output Switching Regulators with Internal Power Switches n Short Circuit Robust n Adjustable 250kHz to 2.5MHz Switching Frequency, Synchronizable Over the Full Range n Integrated Boost Diodes n Integrated Loop Compensation n Anti-Phase Switching Reduces Ripple n Low Shutdown I (<2µA) Q n Uses Small Inductors and Ceramic Capacitors n Thermally Enhanced, 16-Lead MSOP Package The LT®3988 is a dual, current mode, step-down, DC/DC converter that accepts two input voltages up to 60V (80V transient), which may be driven from separate supplies or can be cascaded. High efficiency switches are included on the die along with internal boost diodes and loop compensation. Both converters are capable of generating 1A outputs, are synchronized to a single oscillator programmable up to 2.5MHz, and run with opposite phases, reducing input ripple current. n The switching frequency is set with a single resistor yielding a range of 250kHz to 2.5MHz, or a clock signal can be applied to the SYNC pin. The LT3988’s high switching frequency allows the use of small inductors and capacitors, resulting in a very small dual output supply. The constant switching frequency, combined with low impedance ceramic capacitors, results in low, predictable output ripple. A current mode PWM architecture provides fast transient response with cycle-by-cycle current limiting. Diode current sense and thermal shutdown provide additional protection. Applications Commercial Vehicle Battery Regulation Industrial Supplies n Distributed Supply Regulation n n The LT3988 is available in a 16-lead MSOP package with an exposed pad for low thermal resistance. L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Typical Application Efficiency 4.7µF VIN2 TRACK/SS2 1000pF SYNC LT3988 22µH 10.2k 80 200k 0.22µF 0.22µF 57.6k 47µF 1000pF RT SW1 SW2 DA1 FB1 DA2 FB2 GND VOUT = 5V 85 BD BOOST2 BOOST1 VOUT1 5V, 1A 90 VIN1 EN/UVLO TRACK/SS1 15µH VOUT2 3.3V, 1A EFFICIENCY (%) VIN 7V TO 60V TRANSIENT TO 80V VOUT = 3.3V 75 70 65 60 34k 55 10k 47µF 22pF 3988 TA01 50 VIN = 12V fSW = 500kHz 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 IOUT (A) 1 3988 TA01b 3988f 1 LT3988 Absolute Maximum Ratings Pin Configuration (Note 1) VIN (Notes 7, 8)........................................... –0.3V to 80V BOOST.......................................................................80V EN/UVLO (Note 7)......................................................80V BOOST above SW......................................................30V EN/UVLO above VIN1....................................................6V RT, SYNC.....................................................................6V TRACK/SS, FB.............................................................5V BD..............................................................................20V Operating Junction Temperature Range (Note 2) LT3988E............................................. –40°C to 125°C LT3988I.............................................. –40°C to 125°C LT3988H............................................. –40°C to 150°C Storage Temperature Range................... –65°C to 150°C Lead Temperature (Soldering, 10 sec).................... 300°C TOP VIEW DA1 SW1 BOOST1 BD EN/UVLO BOOST2 SW2 DA2 1 2 3 4 5 6 7 8 16 15 14 13 12 11 10 9 17 GND VIN1 TRACK/SS1 FB1 RT SYNC FB2 TRACK/SS2 VIN2 MSE PACKAGE 16-LEAD PLASTIC MSOP θJA = 40°C/W, θJC = 10°C/W EXPOSED PAD (PIN 17) IS GND, MUST BE SOLDERED TO PCB Order Information LEAD FREE FINISH TAPE AND REEL PART MARKING PACKAGE DESCRIPTION TEMPERATURE RANGE* LT3988EMSE#PBF LT3988EMSE#TRPBF 3988 16-Lead Plastic MSOP –40°C to 125°C LT3988IMSE#PBF LT3988IMSE#TRPBF 3988 16-Lead Plastic MSOP –40°C to 125°C LT3988HMSE#PBF LT3988HMSE#TRPBF 3988 16-Lead Plastic MSOP –40°C to 150°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ Electrical Characteristics The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN1/2 = 12V, unless otherwise noted. (Notes 2, 5, 6) PARAMETER CONDITIONS VIN1 Undervoltage Lockout (Note 3) VIN1 Rising MIN l VIN1 Undervoltage Lockout Hysteresis VIN1 Overvoltage Lockout (Note 3) VIN1 Rising l 60 VIN2 Rising, VIN1 = 4.1V l EN/UVLO Undervoltage Threshold Hysteresis 4.1 64 2 2.6 –0.5 300 VEN/UVLO = Rising l 1.1 66 3.1 V V mV 500 1.2 V V 0.5 120 UNITS mV 135 VEN/UVLO = 1.2V EN/UVLO Enable Threshold EN/UVLO Undervoltage Threshold 3.9 2.1 VIN2 Undervoltage Lockout Hysteresis EN/UVLO Input Current MAX 260 VIN1 Overvoltage Lockout Hysteresis VIN2 Undervoltage Lockout (Note 3) TYP µA mV 1.3 V mV 3988f 2 LT3988 Electrical Characteristics The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN1/VIN2 = 12V, unless otherwise noted. (Notes 2, 5, 6) PARAMETER CONDITIONS TYP MAX VIN1 Quiescent Current VFB1 = 0.9V, VBD = 0V MIN 2.7 4.5 mA VIN1 Quiescent Current VFB1 = 0.9V, VBD = 5V 1.6 3 mA VIN2 Quiescent Current VFB2 = 0.9V, VBD = 5V 250 1000 µA BD Pin Current VBD = 0V –8 –30 µA BD Pin Quiescent Current VBD = 5V 1.1 2.2 mA Shutdown Current (IVIN1+IVIN2) VEN/UVLO ≤ 0.3V 0.1 1 µA 0.75 0.75 0.76 0.765 V V –5 –100 nA FB Voltage l FB Pin Bias Current VFB = 0.75V FB Line Voltage Regulation 5V < VIN < 60V Switching Frequency RT = 40.2k Switching Frequency RT = 200k Switching Frequency RT = 14.7k Switching Phase, SW1 to SW2 RT = 40.2k DA Comparator Current Threshold l 0.01 l ISW = 1A Switch Current Limit (Note 4) Duty Cycle = 35% 0.9 1 %/V 1.1 250 MHz 150 180 210 1.1 1.32 1.58 850 1.4 1.87 MHz kHz 2.15 l Switch VSAT 0.74 0.735 UNITS Deg A mV 2.25 A Switch Leakage Current 0.01 1 µA Minimum Boost Voltage 2 2.5 V 20 50 mA Boost Pin Current ISW = 1A Boost Diode Forward Voltage IBD = 50mA 0.7 0.9 V Boost Diode Leakage Current VR = 5V 0.1 5 µA TRACK/SS Pin Current VTRACK/SS = 1V –1.3 –2.2 µA SYNC Input High Voltage VIH l SYNC Input Low Voltage VIL l SYNC Input Frequency SYNC Pin Input Current –0.8 1.5 V 0.25 VSYNC = 1.5V Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LT3988E is guaranteed to meet performance specifications from 0°C to 125°C junction temperature. Specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LT3988I is guaranteed over the full –40°C to 125°C operating junction temperature range. The LT3988H is guaranteed over the full –40°C to 150°C operating junction temperature range. High junction temperatures degrade operating lifetimes. Operating lifetime is derated at junction temperatures greater than 125°C. Note 3: Undervoltage lockout occurs when VIN is lower than the undervoltage threshold. Overvoltage lockout occurs when VIN exceeds the threshold voltage. See Applications Information. 0.3 0.4 V 2.5 MHz µA Note 4: Current limit is guaranteed by design and/or correlation to static test. Slope compensation reduces current at higher duty cycles. Note 5: Polarity specification for all currents into pins is positive. All voltages are referenced to GND unless otherwise specified. Note 6: This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed the maximum operating junction temperature when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. Note 7: Absolute Maximum Voltage at VIN and EN/UVLO is 80V for nonrepetitive 1 second transients, and 60V for continuous operation. Note 8: If VIN2 is driven above 60V, VIN2 must be connected to VIN1. 3988f 3 LT3988 Typical Performance Characteristics VIN = 12V 75 VIN = 48V 65 60 55 VIN = 24V 75 VIN = 48V 70 65 0 50 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 LOAD CURRENT (A) 70 65 55 f = 500kHz 0 50 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 LOAD CURRENT (A) f = 500kHz 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 LOAD CURRENT (A) 3988 G03 Switch Current Limit vs Temperature 0.755 2.0 0.745 Switch Current Limit vs Duty Cycle 2.5 DUTY CYCLE = 35% 2.0 CURRENT LIMIT (A) 2.5 CURRENT LIMIT (A) 0.760 1.5 1.0 0 VIN = 48V 75 3988 G02 Feedback Voltage vs Temperature 0.740 –50 –25 VIN = 24V 60 3988 G01 0.750 VIN = 12V 80 55 f = 500kHz VBD = 3V Efficiency, VOUT = 5V 85 60 45 FEEDBACK VOLTAGE (V) 90 VIN = 12V 80 50 40 Efficiency, VOUT = 3.3V 85 VIN = 24V 70 EFFICIENCY (%) 90 EFFICIENCY (%) Efficiency, VOUT = 2.5V EFFICIENCY (%) 80 TA = 25°C, unless otherwise noted. 1.5 MINIMUM 1.0 0.5 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) TYPICAL 0 0.5 25 50 75 100 125 150 TEMPERATURE (°C) 3988 G04 0 20 40 60 DUTY CYCLE (%) 3988 G05 Switching Frequency Foldback 1.2 10 2.5 100 3988 G06 Switching Frequency vs Temperature Switching Frequency vs RT 80 2.3 FREQUENCY (MHz) 1.9 1.7 1.5 1.3 1.1 0.9 0.7 1.0 5 f/fNOM FREQUENCY CHANGE (%) 2.1 0 –5 0.8 0.6 0.4 0.5 0.3 0.1 0 20 40 60 80 100 120 140 160 180 200 RT (kΩ) 3988 G07 –10 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 3988 G08 0.2 0 0.1 0.2 0.3 0.4 0.5 FEEDBACK VOLTAGE (V) 0.6 0.7 3988 G09 3988f 4 LT3988 Typical Performance Characteristics No-Load Supply Current vs Input Voltage TA = 25°C, unless otherwise noted. Switch Voltage Drop vs Load Current 3.5 Boost Current vs Load Current 1.0 20 3.4 VOLTAGE DROP (V) IQ (mA) 3.2 3.1 3.0 2.9 2.8 BOOST CURRENT (mA) 0.8 3.3 0.6 0.4 0.2 2.7 15 10 5 2.6 2.5 0 0 5 10 15 20 25 30 35 40 45 50 55 60 VIN (V) 0 0.2 0.4 0.6 LOAD CURRENT (A) 0.8 30 20 40 60 80 BOOST DIODE CURRENT (mA) 100 0.8 200 20 0 1.0 250 10 150 100 50 f = 1MHz 0 0.2 0.4 0.6 LOAD CURRENT (A) 0.8 3988 G13 0 –50 –25 1.0 0 25 50 75 100 125 150 TEMPERATURE (°C) 3988 G14 Minimum Switch Off-Time vs Temperature 3988 G15 Undervoltage Lockout vs Temperature 350 VIN1 RISING 4 300 VIN1 FALLING 250 VIN UVLO (V) 0 0.4 0.6 LOAD CURRENT (A) Minimum Switch On-Time vs Temperature MINIMUM ON-TIME (ns) 0.75 VIN (V) 40 0.25 0.2 3988 G12 Maximum VIN for Full Frequency vs Load Current 1.00 MINIMUM OFF-TIME (ns) BOOST DIODE VOLTAGE (V) Boost Diode Voltage vs Boost Diode Current 0.50 0 3988 G11 3988 G10 0 0 1.0 200 150 100 3 VIN2 RISING VIN2 FALLING 2 1 50 0 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 3988 G16 0 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 3988 G17 3988f 5 LT3988 Typical Performance Characteristics VIN1 Overvoltage Lockout vs Temperature TRACK/SS Pin Current vs Temperature 1.42 TRACK/SS PIN CURRENT (µA) 65 VIN1 OVLO (V) 64 VIN1 RISING 63 62 1.38 1.34 1.30 1.26 VIN1 FALLING 61 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 3988 G18 1.22 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 3988 G19 Pin Functions BD: Internal boost diodes are connected between the BD pin and the BOOST pins. Connect BD to a 3V or higher supply, such as VOUT. BOOST1, BOOST2: The BOOST pins are used to provide drive voltages, higher than the input voltage, to the internal NPN power switches. DA1, DA2: Tie the DA pin to the anode of the external Schottky catch diode. If the DA pin current exceeds 1.32A, which could occur in an overload or short-circuit condition, switching is disabled until the DA pin current falls below 1.32A. EN/UVLO: This pin is used to shut down the LT3988. It can be driven from a logic level, tied directly to the input, or used as an undervoltage lockout by connecting a resistor divider from VIN1. FB1, FB2: The LT3988 regulates each feedback pin to 0.750V. Connect the feedback resistor divider taps to these pins. GND: The exposed pad metal of the package provides both electrical contact to ground and good thermal contact to the printed circuit board. The exposed pad must be soldered to the circuit board for proper operation. RT: The RT pin is used to set the internal oscillator frequency. Tie a resistor from RT to GND for the desired switching frequency. SYNC: To synchronize the part to an external frequency, drive the SYNC pin with a logic-level signal with positive and negative pulse widths of at least 100ns. If the SYNC function is not used, connect the SYNC pin to ground. If using SYNC, minimize coupling to RT and FB2, and add decoupling capacitors as needed up to 22pF. SW1, SW2: The SW pins are the outputs of the internal power switches. Connect these pins to the inductors, catch diodes and boost capacitors. TRACK/SS1, TRACK/SS2: The TRACK/SS pins are used to soft-start the two channels, to allow one channel to track the other output, or to allow both channels to track another output. For tracking, tie a resistor divider to this pin from the tracked output. For soft-start, tie a capacitor to this pin. An internal –1.2μA soft-start current charges the capacitor to create a voltage ramp at the pin. Leave these pins disconnected if unused. VIN1: The VIN1 pin supplies current to the LT3988 internal circuitry and to the internal power switch connected to SW1 and must be locally bypassed. VIN1 must be greater than 3.9V (typ) for channel 1 or channel 2 to operate. VIN2: The VIN2 pin supplies current to the internal power switch connected to SW2 and must be locally bypassed. Connect this pin directly to VIN1 unless power for channel 2 is coming from a different source. VIN2 must be greater than 2.6V (typ) and VIN1 must be greater than 3.9V (typ) for channel 2 to operate. If VIN2 is driven above 60V, VIN2 must be connected to VIN1. 3988f 6 LT3988 Block Diagram RT THERMAL SHUTDOWN SYNC MASTER OSC EN/UVLO INT REG AND REF VIN CIN AV BD CL BOOST SW CTL SLOPE C3 L1 OUT SW CLK SLAVE OSC D1 DA COUT R1 0.675V 1.2µA C2 VC gm TRACK/SS FB R2 TRACK/SS 0.75V ILIMIT CLAMP GND 3988 BD ONE OF TWO SWITCHING REGULATORS SHOWN Figure 1. Block Diagram of the LT3988 with Associated External Components 3988f 7 LT3988 Operation The LT3988 is a dual, constant frequency, current mode regulator with internal power switches. Operation can be best understood by referring to the Block Diagram in Figure 1. If the EN/UVLO pin is pulled low, the LT3988 is shut down and draws minimal current from the input source(s) tied to the VIN pins. If the EN/UVLO pin exceeds 0.5V (typ), the internal bias circuits turn on, including the internal regulator, reference and master oscillator. The switching regulators will only begin to operate when the EN/UVLO pin exceeds 1.2V (typ). The switcher is a current mode regulator. Instead of directly modulating the duty cycle of the power switch, the feedback loop controls the peak current in the switch during each cycle. Compared to voltage mode control, current mode control improves loop dynamics and provides cycle-bycycle current limit. An oscillator enables an RS flip flop, turning on the internal power switch. An amplifier and comparator monitor the current flowing between the VIN and SW pins, turning the switch off when this current reaches a level determined by the voltage at VC. An error amplifier measures the output voltage through an external resistor divider tied to the FB pin and servos the VC voltage. If the error amplifier’s output increases, more current is delivered to the output; if it decreases, less current is delivered. An active clamp on the VC voltage provides a current limit. The switching frequency is set either by the resistance to GND at the RT pin or by the frequency of the logic-level signal driving the SYNC pin. A detection circuit monitors for the presence of a SYNC signal on the pin and switches between the two modes upon detection of a clock applied to the SYNC pin. Use of the SYNC pin as a frequency input requires the use of an RT resistor as well. This requirement is detailed in the Switching Frequency section. Onboard circuitry generates the appropriate slope compensation ramps and generates the 180° out-of-phase clocks for the two channels. Each switcher contains an extra, independent oscillator to perform frequency foldback during overload conditions. This slave oscillator is normally synchronized to the master oscillator. A comparator senses when VFB is less than 50% of its regulated value and switches the regulator from the master oscillator to a slower slave oscillator. VFB is less than 50% of its regulated value during start-up, short-circuit, and overload conditions. Frequency foldback helps limit switch current under these conditions. The TRACK/SS pins override the 0.75V reference of the FB pins when the TRACK/SS pins are below 0.75V. This allows either coincident or ratiometric supply tracking on start-up as well as a soft-start capability. The switch drivers operate either from VIN or from the BOOST pin. An external capacitor and internal Schottky diode are used to generate a voltage at the BOOST pin that is higher than the input supply. This allows the driver to obtain a low VCE across the internal bipolar NPN power switch for efficient operation. The BD pin serves two purposes. The voltage at BD determines the BOOST1 and BOOST2 levels over the VIN1 and VIN2 supply voltages, and allows the internal circuitry to draw its current from a lower voltage supply than VIN1. This reduces power dissipation and increases efficiency. If the voltage at BD falls below 3V, then quiescent current will flow from VIN1. The overvoltage and undervoltage detection shuts down the LT3988 if the input voltage on VIN1 goes above or below thresholds. The overvoltage detector shuts down the regulators when VIN1 exceeds 60V. An undervoltage detector monitoring VIN1 disables both regulators when VIN1 is under 3.7V, an undervoltage detector monitoring VIN2 shuts down channel 2 when VIN2 is under 2.5V. The higher voltage is required on VIN1 to accomodate internal bias circuits. Additionally, tying the EN/UVLO pin to a voltage divider from VIN1 to ground allows a programmable undervoltage threshold. 3988f 8 LT3988 Applications Information Step-Down Considerations FB Resistor Network The output voltage is programmed with a resistor divider (refer to the Block Diagram) between the output and the FB pin. Choose the resistors according to: V R1= R2 OUT – 1 750mV The parallel combination of R1 and R2 should be 20k or less to minimize bias current errors. The maximum error due to VFB bias current is ∆VOUT = IFB(MAX) • R1 Input Voltage Range The minimum operating voltage is determined either by the LT3988’s undervoltage lockout or by its maximum duty cycle. The duty cycle is the fraction of time that the internal switch is on and is determined by the input and output voltages: DC = VOUT + VF VIN – VSW + VF where VF is the forward voltage drop of the catch diode (~0.4V) and VSW is the voltage drop of the internal switch (~0.3V at maximum load). This leads to a minimum input voltage of: VIN(MIN) = VOUT + VF – VF + VSW DCMAX The duty cycle is the fraction of time that the internal switch is on during a clock cycle. The maximum duty cycle is generally given by DCMAX = 1 – tOFF(MIN) • f. However, unlike most fixed frequency regulators, the LT3988 will not switch off at the end of each clock cycle if there is sufficient voltage across the boost capacitor (C3 in Figure 1) to fully saturate the output switch. Forced switch-off for a minimum time will only occur at the end of a clock cycle when the boost capacitor needs to be recharged. This operation has the same effect as lowering the clock frequency for a fixed off time, resulting in a higher duty cycle and lower minimum input voltage. The resultant duty cycle depends on the charging times of the boost capacitor and can be approximated by the following equation: DCMAX = B B+1 where B is the switch pin current divided by the typical boost current from the BOOST pin current vs switch current in the Typical Performance Characteristics section. The maximum operating voltage without pulse-skipping is determined by the minimum duty cycle DCMIN: VIN(PS) = VOUT + VF – VF + VSW DCMIN with DCMIN = tON(MIN) • f. The LT3988 will regulate the output current at input voltages greater than VIN(PS). Exceeding VIN(PS) is safe if the output is in regulation, if the external components have adequate ratings to handle the peak conditions and if the peak inductor current does not exceed 2.3A. A saturating inductor may further reduce performance. For robust operation under fault conditions at input voltages of 40V or greater, use an inductor value of 47µH or larger and a clock rate of 1MHz or lower. Both the maximum and minimum input voltages are a function of the switching frequency and output voltages. Therefore the maximum switching frequency must be set to a value that accommodates all the input and output voltage parameters and must meet both of the following criteria for each channel: fMAX1 = VOUT + VF 1 • VIN(PS) – VSW + VF tON(MIN) VOUT + VF 1 fMAX2 = 1– • VIN(MIN) – VSW + VF tOFF(MIN) The values of tON(MIN) and tOFF(MIN) are functions of ISW and temperature (see chart in the Typical Performance Characteristics section). Worst-case values for switch currents greater than 0.5A are tON(MIN) = 180ns (for TJ > 125°C tON(MIN) = 200ns) and tOFF(MIN) = 240ns. fMAX1 is the frequency at which the minimum duty cycle is exceeded. The regulator will skip ON pulses in order to reduce the overall duty cycle 3988f 9 LT3988 Applications Information at frequencies above fMAX1. It will continue to regulate but with increased inductor current and increased output ripple. fMAX2 is the frequency at which the maximum duty cycle is exceeded. If there is sufficient charge on the BOOST capacitor, the regulator will skip OFF periods to increase the overall duty cycle at frequencies above fMAX2. Note that the restriction on the operating input voltage refers to steady-state limits to keep the output in regulation; the circuit will tolerate input voltage transients up to the absolute maximum rating. Switching Frequency Once the upper and lower bounds for the switching frequency are found from the duty cycle requirements, the frequency may be set within those bounds. Lower frequencies result in lower switching losses, but require larger inductors and capacitors. The user must decide the best trade-off. The switching frequency is set by a resistor connected from the RT pin to ground, or by forcing a clock signal into the SYNC pin. The LT3988 applies a voltage across this resistor and uses the current to set the oscillator speed. The RT resistor value for a given switching frequency is given by: RT = 1.31 46.56 + – 7.322 f f2 250kHz ≤ f ≤ 2.5MHz where f is in MHz and RT is in kΩ. The frequency sync signal will support VIH logic levels from 1.5V to 5V CMOS or TTL. The duty cycle is not important, but it needs a minimum on time of 100ns and a minimum off time of 100ns. RT should be set to provide a frequency within ±25% of the final sync frequency. The slope recovery circuit sets the slope compensation to the appropriate value for the synchronized frequency. Choose the inductor value based on the lowest potential switching frequency. Inductor Selection and Maximum Output Current A good first choice for the inductor value is: L= VOUT + VF 0.6A • f where VF is the voltage drop of the catch diode (~0.4V) and f is in MHz. The inductor’s RMS current rating must be greater than the maximum load current and its saturation current Table 1. Inductors MFG URL PART SERIES INDUCTANCE RANGE (µH) SIZE (mm) (L × W × H) Coilcraft http://www.coilcraft.com XPL7030 XFL4020 XAL50XX 0.13 to 22 1 to 4.7 0.16 to 22 7×7×3 4 × 4 × 2.15 5.28 × 5.48 × 5.1 Cooper http://www.cooperbussmann.com DRA74 DR1040 0.33 to 1000 1.5 to 330 7.6 × 7.6 × 4.35 10.5 × 10.3 × 4 CWS http://www.coilws.com SP-0703 SP-0704 SB-1004 3 to 100 2.2 to 100 10 to 1500 7×7×3 7×7×4 10.1 × 10.1 × 4.5 Murata http://www.murata.com LQH55D LQH6PP LQH88P 0.12 to 10000 1 to 100 1 to 100 5 × 5.7 × 4.7 6 × 6 × 4.3 8 × 8 × 3.8 Sumida http://www.sumida.com CDMC6D28 CDEIR8D38F 0.2 to 4.7 4 to 22 7.25 × 6.7 × 3 8.5 × 8.3 × 4 Toko http://www.toko.co.jp DS84LCB FDV0620 1 to 100 0.2 to 4.7 8.4 × 8.3 × 4 6.7 × 7.4 × 2 Vishay http://www.vishay.com IHLP-2020AB-11 IHLP-2020BZ-11 IHLP-2525CZ-11 0.1 to 4.7 0.1 to 10 1 to 22 5.49 × 5.18 × 1.2 5.49 × 5.18 × 2 6.86 × 6.47 × 3 Würth http://www.we-online.de WE-PD2-S WE-PD-M WE-PD2-XL 1 to 68 1 to 1000 10 to 820 4 × 4.5 × 3.2 7.3 × 7.3 × 4.5 9 × 10 × 5.4 3988f 10 LT3988 Applications Information should be at least 30% higher. For highest efficiency, the series resistance (DCR) should be less than 0.1Ω. Table 1 lists several vendors and types that are suitable. The current in the inductor is a triangle wave with an average value equal to the load current. The peak switch current is equal to the output current plus half the peak-topeak inductor ripple current. The LT3988 limits its switch current in order to protect itself and the system from overload faults. Therefore, the maximum output current that the LT3988 will deliver depends on the switch current limit, the inductor value and the input and output voltages. When the switch is off, the potential across the inductor is the output voltage plus the catch diode drop. This gives the peak-to-peak ripple current in the inductor: ∆IL = (1– DC) • VOUT + VF L•f where f is the switching frequency of the LT3988 and L is the value of the inductor. In continuous mode, the peak inductor and switch current is: ISWPK =ILPK = ∆IL +I 2 L To maintain output regulation, this peak current must be less than the LT3988’s switch current limit, ILIM. For both switches, ILIM is at least 1.5A at low duty cycle and decreases linearly to 1.1A at DC = 90%. (See chart in the Typical Performance Characteristics section). The minimum inductance can now be calculated as: LMIN = 1– DCMIN VOUT + VF • 2•f ILIM – IOUT However, it’s generally better to use an inductor larger than the minimum value. The minimum inductor has large ripple currents which increase core losses and require large output capacitors to keep output voltage ripple low. This analysis is valid for continuous mode operation (IOUT > ∆IL/2). For details of maximum output current in discontinuous mode operation, see Linear Technology’s Application Note AN44. Finally, for duty cycles greater than 50% (VOUT/ VIN > 0.5), a minimum inductance is required to avoid subharmonic oscillations. This minimum inductance is: LMIN = VOUT + VF 1.25A • f with LMIN in μH and f in MHz. For robust operation under fault conditions at input voltages of 40V or greater, use an inductor value of 47µH or larger and a clock rate of 1MHz or lower. Output Capacitor Selection The output capacitor filters the inductor current to generate an output with low voltage ripple. It also stores energy in order to satisfy transient loads and stabilize the LT3988’s control loop. Because the LT3988 operates at a high frequency, minimal output capacitance is necessary. In addition, the control loop operates well with or without the presence of output capacitor series resistance (ESR). Ceramic capacitors, which achieve very low output ripple and small circuit size, are therefore an option. You can estimate output ripple with the following equations: VRIPPLE = ∆IL 8 • f • COUT for ceramic capacitors and V = ∆IL • ESR for electrolytic capacitors RIPPLE (tantalum and aluminum) where ΔIL is the peak-to-peak ripple current in the inductor. The RMS content of this ripple is very low so the RMS current rating of the output capacitor is usually not of concern. It can be estimated with the formula: IC(RMS) = ∆IL 12 Another constraint on the output capacitor is that it must have greater energy storage than the inductor; if the stored energy in the inductor transfers to the output, the resulting voltage step should be small compared to the regulation voltage. For a 5% overshoot, this requirement indicates: I COUT > 10 • L • LIM VOUT 2 3988f 11 LT3988 Applications Information The low ESR and small size of ceramic capacitors make them the preferred type for LT3988 applications. Not all ceramic capacitors are the same, however. Many of the higher value capacitors use poor dielectrics with high temperature and voltage coefficients. In particular, Y5V and Z5U types lose a large fraction of their capacitance with applied voltage and at temperature extremes. Because loop stability and transient response depend on the value of COUT, this loss may be unacceptable. Use X7R and X5R types. Electrolytic capacitors are also an option. The ESRs of most aluminum electrolytic capacitors are too large to deliver low output ripple. Tantalum, as well as newer, lower-ESR organic electrolytic capacitors intended for power supply use are suitable. Choose a capacitor with a low enough ESR for the required output ripple. Because the volume of the capacitor determines its ESR, both the size and the value will be larger than a ceramic capacitor that would give similar ripple performance. One benefit is that the larger capacitance may give better transient response for large changes in load current. Table 2 lists several capacitor vendors. Table 2. Low ESR Surface Mount Capacitors MFG TYPE SERIES AVX Ceramic Tantalum TPS Ceramic X7R, 1812 MLCC Kemet Johansen Tantalum Tantalum Organic Aluminum Organic T491,T494,T498 T520,T521,T528 A700 Panasonic Aluminum Organic SP CAP Sanyo Tantalum Aluminum Organic POSCAP Taiyo-Yuden Ceramic TDK Ceramic Diode Selection The catch diode (D1 from Figure 1) conducts the inductor current during the switch off time. Use a Schottky diode rated for 1A to 2A average current. Peak reverse voltage across the diode is equal to the regulator input voltage. Use a diode with a reverse voltage rating greater than the input voltage. The OVLO function of the LT3988 turns off the switch when VIN > 64V (typ) allowing use of Schottky 12 diodes with a 70V rating for input voltages up to 80V. Table 3 lists several Schottky diodes and their manufacturers. Table 3. Schottky Diodes PART NUMBER VR (V) IAVG (A) VF AT 1A (mV) 40 1 490 VF AT 2A (mV) On Semiconductor NSR10F40NXT5G MBRA160T3 60 1 510 MBRS190T3 90 1 750 MBRS260T3G 60 2 40 1 430 Diodes Inc B140 500 B160 60 1 700 B170 70 1 790 B180 80 1 790 B260 60 2 700 B280 80 2 790 DFLS140L 40 1 550 DFLS160L 60 1 500 DFLS260 60 2 620 Boost Pin Considerations The external capacitor and the internal diode tied to the BOOST pin generate a voltage that is higher than the input voltage. In most cases, a small ceramic capacitor will work well. The capacitor value is a function of the switching frequency, peak current, duty cycle and boost voltage. Figure 2 shows three ways to arrange the boost circuit. The BOOST pin must be more than 2.3V above the SW pin for full efficiency. For outputs of 3.3V and higher, the standard circuit (Figure 2a) is best. For lower output voltages, the BD pin can be tied to the input (Figure 2b). The circuit in Figure 2a is more efficient because the BOOST pin current comes from a lower voltage source. Finally, as shown in Figure 2c, the BD pin can be tied to another source that is at least 3V. For example, if you are generating 3.3V and 1.8V and the 3.3V is on whenever the 1.8V is on, the BD pin can be connected to the 3.3V output. (see Output Voltage Tracking). Be sure that the maximum voltage at the BOOST pin is less than 80V and the voltage difference between the BOOST and SW pins is less than 30V. The minimum operating voltage of an LT3988 application is limited by the internal 4V undervoltage lockout and by the maximum duty cycle. 3988f LT3988 Applications Information VIN3 > 3V BD VIN BOOST VIN BD C3 VOUT SW VIN BOOST VIN GND BD C3 VOUT SW VIN VIN GND VBOOST – VSW ≅ VOUT MAX VBOOST ≅ VIN + VOUT VOUT SW GND VBOOST – VSW ≅ VIN MAX VBOOST ≅ 2VIN (2a) BOOST VBOOST – VSW ≅ VIN3 MAX VBOOST ≅ VIN3 + VIN MIN VALUE FOR VIN3 = 3V (2b) 3988 F02 (2c) Figure 2. Generating the Boost Voltage The boost circuit also limits the minimum input voltage for proper start-up. If the input voltage ramps slowly, or the LT3988 turns on when the output is already in regulation, the boost capacitor may not be fully charged. Because the boost capacitor charges with the energy stored in the inductor, the circuit will rely on some minimum load current to get the boost circuit running properly. This minimum load will depend on input and output voltages, and on the arrangement of the boost circuit. The minimum load current generally goes to zero once the circuit has started. Figure 4 shows a plot of input voltage to start and to run as a function of load current. Even without an output load current, in many cases the discharged output capacitor will present a load to the switcher that will allow it to start. Converter with Backup Output Regulator There is another situation to consider: systems where the output will be held high when the input to the LT3988 is absent. If the VIN pin is grounded while the output is held high, then diodes inside the LT3988 can pull large currents from the output through the SW and VIN pins. A Schottky diode in series with the input to the LT3988, as shown in Figure 3, will protect the LT3988 and the system from a shorted or reversed input. LT3988 D4 SW VIN GND The boost current is generally small but can become significant at high duty cycles. The required boost current is: 3988 F03 Figure 3. Diode D4 Prevents a Shorted Input from Discharging a Backup Battery Tied to the Output V I IBOOST = OUT OUT VIN 40 Minimum Input Voltage, VOUT = 3.3V 5.5 Minimum Input Voltage, VOUT = 5V 7.0 TA = 25°C TA = 25°C 6.6 INPUT VOLTAGE (V) 5.0 INPUT VOLTAGE (V) VOUT TO START 4.5 4.0 TO START 6.2 5.8 TO RUN TO RUN 3.5 0 200 400 600 800 LOAD CURRENT (mA) 1000 3988 F04a 5.4 0 200 400 600 800 LOAD CURRENT (mA) 1000 3988 F04b Figure 4. The Minimum Input Voltage Depends on Output Voltage, Load Current, and Boost Circuit 3988f 13 LT3988 Applications Information Input Capacitor Selection Bypass the input of the LT3988 circuit with a 4.7μF or higher ceramic capacitor of X7R or X5R type. A lower value or a less expensive Y5V type will work if there is additional bypassing provided by bulk electrolytic capacitors, or if the input source impedance is low. The following paragraphs describe the input capacitor considerations in more detail. Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input capacitor is required to reduce the resulting voltage ripple at the LT3988 input and to force this switching current into a tight local loop, minimizing EMI. The input capacitor must have low impedance at the switching frequency to do this effectively and it must have an adequate ripple current rating. With two switchers operating at the same frequency but with different phases and duty cycles, calculating the input capacitor RMS current is not simple; however, a conservative value is the RMS input current for the phase delivering the most power (VOUT • IOUT): IIN(RMS) = IOUT • VOUT ( VIN – VOUT ) VIN I < OUT 2 and is largest when VIN = 2VOUT (50% duty cycle). As the second, lower power channel draws input current, the input capacitor’s RMS current actually decreases as the out-of-phase current cancels the current drawn by the higher power channel. Considering that the maximum load current from a single phase (if SW1 and SW2 are both at maximum current) is ~1A, RMS ripple current will always be less than 0.5A. The high frequency of the LT3988 reduces the energy storage requirements of the input capacitor, so that the capacitance required is often less than 10μF. The combination of small size and low impedance (low equivalent series resistance or ESR) of ceramic capacitors makes them the preferred choice. The low ESR results in very low voltage ripple. Ceramic capacitors can handle larger magnitudes of ripple current than other capacitor types of the same value. An alternative to a high value ceramic capacitor is a lower value along with a larger electrolytic capacitor, for example a 1μF ceramic capacitor in parallel with a low ESR tantalum capacitor. For the electrolytic capacitor, a value larger than 10μF will be required to meet the ESR and ripple current requirements. Because the input capacitor is likely to see high surge currents when the input source is applied, tantalum capacitors should be surge rated. The manufacturer may also recommend operation below the rated voltage of the capacitor. Be sure to place the 1μF ceramic as close as possible to the VIN and GND pins on the IC for optimal noise immunity. A final caution is in order regarding the use of ceramic capacitors at the input. A ceramic input capacitor can combine with stray inductance to form a resonant tank circuit. If power is applied quickly (for example by plugging the circuit into a live power source), this tank can ring, doubling the input voltage and damaging the LT3988. The solution is to either clamp the input voltage or dampen the tank circuit by adding a lossy capacitor in parallel with the ceramic capacitor. For details, see Application Note 88. Frequency Compensation The LT3988 uses current mode control to regulate the output. This simplifies loop compensation. In particular, the LT3988 does not depend on the ESR of the output capacitor for stability, so you are free to use ceramic capacitors to achieve low output ripple and small circuit size. The LT3988 is internally compensated with the RC network tied to the VC node. The internal compensation network is optimized to provide stability over the full frequency range. Figure 5 shows an equivalent circuit for the LT3988 control loop. The error amplifier is a transconductance amplifier with LT3988 CURRENT MODE POWER STAGE OUT gm = 2A/V FB VC RC 300k CC 40pF CPL R1 RESR gm = 40µA/V 7M ERROR AMPLIFIER 0.75V COUT R2 3988 F05 Figure 5. Model For Loop Response 3988f 14 LT3988 Applications Information finite output impedance. The power section, consisting of the modulator, power switch and inductor, is modeled as a transconductance amplifier generating an output current proportional to the voltage at the VC node. Note that the output capacitor integrates this current, and that the capacitor on the VC node (CC) integrates the error amplifier output current, resulting in two poles in the loop. RC provides a zero. With the recommended output capacitor, the loop crossover occurs above the RCCC zero. This simple model works well as long as the value of the inductor is not too high and the loop crossover frequency is much lower than the switching frequency. With a larger ceramic capacitor (very low ESR), crossover may be lower and a phase lead capacitor (CPL) across the feedback divider may improve the phase margin and transient response. Large electrolytic capacitors may have an ESR large enough to create an additional zero, and the phase lead may not be necessary. If the output capacitor is different than the recommended capacitor, stability should be checked across all operating conditions, including load current, input voltage and temperature. The LT1375 data sheet contains a more thorough discussion of loop compensation and describes how to test the stability using a transient load. Shutdown The EN/UVLO pin is used for two purposes, to place the LT3988 in a low current shutdown mode, and to override the internal undervoltage lockout thresholds with a user programmable threshold. When the EN/UVLO pin is pulled to under 0.5V (typ), the LT3988 is in shutdown mode and draws less than 1µA from the input supply. When the EN/UVLO pin is driven above 0.5V (typ) and less than 1.2V (typ), the internal regulator is activated and the oscillators are operating, but the switching operation of both channels remains inhibited. When the EV/UVLO pin is driven above 1.2V (typ), the undervoltage lockout asserted by the EN/UVLO function is released, allowing switching operation of both channels. Internal undervoltage detectors will still prevent switching operation on channel 1 until VIN1 is greater than 3.9V (typ) and on channel 2 until VIN2 is greater than 2.6V (typ). The EN/UVLO undervoltage lockout has 120mV (typ) of hysteresis. The EN/UVLO pin is rated up to 80V and can be connected directly to the input voltage. The EN/UVLO pin may be driven by a voltage divider from VIN1, allowing an externally programmable undervoltage lockout to be set above the internal 3.9V threshold. The undervoltage threshold and hysteresis are given by: R1 V VUVTH = 1.2 1+ ;R1= R2 UVTH – 1 R2 1.2 R1 V VUVHY = 0.12 1+ ;R1= R2 UVHY – 1 R2 0.12 VIN1 R1 – EN/UVLO R2 UVLO 1.2V + 3988 F06 Figure 6. Undervoltage Lockout Circuit Output Voltage Tracking The LT3988 allows the user to program how the output ramps up by means of the TRACK/SS pins. Through these pins, either channel output can be set up to either coincidently or ratiometrically track the other channel output. This example will show the channel 2 output tracking the channel 1 output, as shown in Figure 7. The TRACK/SS2 pin acts as a clamp on channel 2’s reference voltage. VOUT2 is referenced to the TRACK/SS2 voltage when the TRACK/SS2 < 0.8V and to the internal precision reference when TRACK/SS2 > 0.8V. To implement the coincident tracking in Figure 7, connect an extra resistive divider to the output of channel 1 and connect its midpoint to the TRACK/SS2 pin (Figure 8). The ratio of this divider should be selected to be the same as that of channel 2’s feedback divider (R5 = R3 and R6 = R4). In this tracking mode, VOUT1 must be set higher than VOUT2. To implement the ratiometric tracking in Figure 6, change the extra divider ratio to R5 = R1 and R6 = R2 + ΔR. The extra resistance on R6 should be set so that the TRACK/SS2 voltage is ≥1V when VOUT1 is at its final value. The need for this extra resistance is best understood with the help of the equivalent input circuit shown in Figure 9. 3988f 15 LT3988 Applications Information OUTPUT VOLTAGE VOUT1 I I 1.36µA VOUT2 TRACK/SS 0.75V FB D1 + D2 gm – D3 3988 F09 Figure 9. Equivalent Input Circuit of Error Amplifier TIME Coincident Tracking OUTPUT VOLTAGE VOUT1 VOUT2 3988 F07 TIME Ratiometric Tracking Figure 7. Two Different Modes of Output Voltage Tracking VOUT1 TO TRK/SS2 PIN VOUT2 R5 R1 R6 R2 TO FB1 PIN R3 R4 TO FB2 PIN SELECTING VALUES FOR R5 AND R6 COINCIDENT RATIOMETRIC R5 = R3 R1 R6 = R4 R1 VOUT1/1V – 1 3988 F08 R1 VOUT1 R3 VOUT2 = – 1, = –1 R2 0.75 R4 0.75 Figure 8. Setup for Coincident and Ratiometric Tracking At the input stage of the error amplifier, two common anode diodes are used to clamp the equivalent reference voltage and an additional diode is used to match the shifted common mode voltage. The top two current sources are of the same amplitude. In the coincident mode, the TRACK/SS2 voltage is substantially higher than 0.75V at steady state and effectively turns off D1. D2 and D3 will therefore conduct the same current and offer tight matching between VFB2 and the internal precision 0.75V reference. In the ratiometric mode with R6 = R2, TRACK/SS2 equals 0.75V at steady state. D1 will divert part of the bias current and make VFB2 slightly lower than 0.75V. Although this error is minimized by the exponential I-V characteristic of the diodes, it does impose a finite amount of output voltage deviation. Further, when channel 1’s output experiences dynamic excursions (under load transient, for example), channel 2 will be affected as well. Setting R6 to a value that pushes the TRK/SS2 voltage to 1V at steady state will eliminate these problems while providing near ratiometric tracking. The example shows channel 2 tracking channel 1, however either channel may be set up to track the other. Soft-Start If a capacitor is tied from the TRACK/SS pin to ground, then the internal pull-up current will generate a voltage ramp on this pin. This results in a ramp at the output, limiting the inductor current and therefore input current during start-up. A good value for the soft-start capacitor is COUT/10,000, where COUT is the value of the output capacitor. 3988f 16 LT3988 Applications Information the input current at VIN2 when VOUT2 is at maximum load. Figure 10 shows a 12V to 5V, and 1.8V 2-stage converter using this approach. Independent Input Voltages VIN1 and VIN2 are independent and can be powered with different voltages provided VIN1 is present when VIN2 is present. Each supply must be bypassed as close to the VIN pins as possible. For applications requiring large inductors due to high VIN to VOUT ratios, a 2-stage step-down approach may reduce inductor size by allowing an increase in frequency. A dual step-down application steps down the input voltage (VIN1) to the highest output voltage, then uses that voltage to power the other output (VIN2). VOUT1 must be able to provide enough current for its output plus VIN 12V PCB Layout For proper operation and minimum EMI, care must be taken during printed circuit board (PCB) layout. Figure 11 shows the high current paths in the step-down regulator circuit. Note that in the step-down regulators large, switched currents flow in the power switch, the catch diode and the input capacitor. The loop formed by these VOUT1 4.7µF 4.7µF VIN1 VIN2 EN/UVLO TRACK/SS1 TRACK/SS2 2200pF 2200pF RT SYNC LT3988 BOOST1 6.8µH VOUT1 5V, 500mA 10µF BOOST2 0.22µF 57.6k 40.2k BD 0.22µF SW1 SW2 DA1 FB1 DA2 FB2 GND 10.2k 3.3µH VOUT2 1.8V, 500mA 14k 10k 22µF 3988 F10 Figure 10. 1MHz, 2-Stage Step-Down 5V and 1.8V Outputs VIN VIN SW GND SW GND (11a) VIN IC1 (11b) VSW C1 L1 SW D1 GND (11c) C2 3988 F11 Figure 11. Subtracting the Current When the Switch Is ON (11a) From the Current When the Switch Is OFF (11b) Reveals the Path of the High Frequency Switching Current (11c). Keep this Loop Small. The Voltage on the SW and Boost Nodes Will Also Be Switched; Keep These Nodes as Small as Possible. Finally, Make Sure the Circuit Is Shielded with a Local Ground Plane 3988f 17 LT3988 Applications Information components should be as small as possible. Place these components, along with the inductor and output capacitor, on the same side of the circuit board and connect them on that layer. Place a local, unbroken ground plane below these components and tie this ground plane to system ground at one location, ideally at the ground terminal of the output capacitor. Additionally, keep the SW and BOOST nodes as small as possible. Figure 12 shows an example of proper PCB layout. for the H-grade). The die temperature is calculated by multiplying the LT3988 power dissipation by the thermal resistance from junction to ambient. Power dissipation within the LT3988 can be estimated by calculating the total power loss from an efficiency measurement and subtracting the catch diode loss. Thermal resistance depends on the layout of the circuit board, but values from 30°C/W to 60°C/W are typical. Related Linear Technology Publications Thermal Considerations Application Notes 19, 35, 44, 76 and 88 contain more detailed descriptions and design information for buck regulators and other switching regulators. The LT1375 data sheet has a more extensive discussion of output ripple, loop compensation, and stability testing. Design Note 318 shows how to generate a dual polarity output supply using a buck regulator. The die temperature of the LT3988 must be lower than the maximum rating of 125°C (150°C for the H-grade). This is generally not a concern unless the ambient temperature is above 85°C. For higher temperatures, care should be taken in the layout of the circuit to ensure good heat sinking of the LT3988. The maximum load current should be derated as the ambient temperature approaches 125°C (150°C C3 R4 R3 R5 R6 R7 C1 C4 C2 U1 D1 D2 C7 C9 L1 C8 L2 C10 3988 F12 Figure 12. Sample PC Board Layout 3988f 18 LT3988 Typical Applications 400kHz, 5V and 3.3V Outputs VIN 7V TO 40V 80V TRANSIENT C1 4.7µF C2 2200pF VIN1 VIN2 EN/UVLO TRACK/SS1 TRACK/SS2 RT SYNC LT3988 BD VOUT1 5V, 1A L1 22µH R2 57.6k C6 47µF C4 0.22µF BOOST1 BOOST2 SW1 SW2 DA1 FB1 DA2 FB2 D1 R3 10.2k C3 2200pF R1 118k fSW = 400kHz C5 0.22µF L2 15µH D2 GND VOUT2 3.3V, 1A R4 34k R5 10k C7 47µF 3988 TA02 C1 TO C7: X5R OR X7R D1, D2: DIODES, INC. B160 3988f 19 LT3988 Typical Applications 1MHz, Wide Input Range 5V and 1.8V Outputs VIN 7V TO 24V 80V TRANSIENT VOUT1 C1 4.7µF C2 2200pF 4.7µF VIN1 VIN2 EN/UVLO TRACK/SS1 TRACK/SS2 RT SYNC LT3988 BD VOUT1 5V, 0.5A L1 6.8µH R2 57.6k C6 22µF C4 0.22µF BOOST1 BOOST2 SW1 SW2 DA1 FB1 DA2 FB2 D1 R3 10.2k C3 2200pF R1 40.2k fSW = 1MHz C5 0.22µF L2 3.3µH D2 GND VOUT2 1.8V, 0.5A R4 14k R5 10k C7 22µF 3988 TA03 C1 TO C7: X5R OR X7R D1: DIODES, INC. B160 D2: DIODES, INC. B120 3988f 20 LT3988 Typical Applications 700kHz, 24V and 12V Outputs with Coincident Tracking VIN1 26V TO 60V 80V TRANSIENT C1 4.7µF C2 4.7µF VIN2 VIN1 EN/UVLO TRACK/SS1 C3 2200pF RT SYNC R2 10k LT3988 TRACK/SS2 R4 4.7k R3 309k VOUT1 24V, 1A L1 47µH C6 10µF VIN2 14V TO 60V C4 0.22µF BOOST1 BD BOOST2 SW1 SW2 DA1 FB1 DA2 FB2 D1 R5 309k R6 10k R1 61.9k fSW = 700kHz C5 0.22µF D2 GND L2 22µH VOUT2 12V, 1A R7 150k R8 10k C7 10µF 3988 TA04 C1 TO C7: X5R OR X7R D1, D2: DIODES, INC. B160 R4: USE 0.25W RESISTOR DERATE OUTPUT CURRENT AT HIGHER AMBIENT TEMPERATURES AND INPUT VOLTAGES TO MAINTAIN JUNCTION TEMPERATURE BELOW THE ABSOLUTE MAXIMUM. 3988f 21 LT3988 Typical Applications 400kHz, 3.3V and 2.5V Outputs VIN 5.5V TO 32V 80V TRANSIENT C1 4.7µF C2 2200pF VIN1 VIN2 EN/UVLO TRACK/SS1 TRACK/SS2 RT SYNC LT3988 BD VOUT1 3.3V, 1A L1 10µH R2 34k C6 47µF C4 0.22µF BOOST1 BOOST2 SW1 SW2 DA1 FB1 DA2 FB2 D1 R3 10k C3 2200pF R1 118k fSW = 400kHz C5 0.22µF D2 GND VOUT1 L2 10µH VOUT2 2.5V, 1A R4 23.2k R5 10k C7 47µF 3988 TA05 C1 TO C7: X5R OR X7R D1, D2: DIODES, INC. B180 3988f 22 LT3988 Package Description Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings. MSE Package 16-Lead Plastic MSOP, Exposed Die Pad (Reference LTC DWG # 05-08-1667 Rev E) BOTTOM VIEW OF EXPOSED PAD OPTION 2.845 ±0.102 (.112 ±.004) 5.23 (.206) MIN 2.845 ±0.102 (.112 ±.004) 0.889 ±0.127 (.035 ±.005) 8 1 1.651 ±0.102 (.065 ±.004) 1.651 ±0.102 3.20 – 3.45 (.065 ±.004) (.126 – .136) 0.305 ±0.038 (.0120 ±.0015) TYP 16 0.50 (.0197) BSC 4.039 ±0.102 (.159 ±.004) (NOTE 3) RECOMMENDED SOLDER PAD LAYOUT 0.254 (.010) 0.35 REF 0.12 REF DETAIL “B” CORNER TAIL IS PART OF DETAIL “B” THE LEADFRAME FEATURE. FOR REFERENCE ONLY 9 NO MEASUREMENT PURPOSE 0.280 ±0.076 (.011 ±.003) REF 16151413121110 9 DETAIL “A” 0° – 6° TYP 3.00 ±0.102 (.118 ±.004) (NOTE 4) 4.90 ±0.152 (.193 ±.006) GAUGE PLANE 0.53 ±0.152 (.021 ±.006) DETAIL “A” 1.10 (.043) MAX 0.18 (.007) SEATING PLANE 0.17 – 0.27 (.007 – .011) TYP 1234567 8 0.50 (.0197) BSC NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX 6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL NOT EXCEED 0.254mm (.010") PER SIDE. 0.86 (.034) REF 0.1016 ±0.0508 (.004 ±.002) MSOP (MSE16) 0911 REV E 3988f Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 23 LT3988 Typical Application 500kHz External Sync, 5V and 3.3V Outputs with 6V UVLO VIN 7V TO 30V 80V TRANSIENT R6 40.2k R7 10k C1 4.7µF C2 1000pF VIN1 VIN2 EN/UVLO TRACK/SS1 TRACK/SS2 RT LT3988 500kHz CLOCK SYNC C3 1000pF R1 100k BD L1 15µH VOUT1 5V, 1A R2 57.6k C6 22µF C4 0.22µF BOOST1 BOOST2 SW1 SW2 DA1 FB1 DA2 FB2 D1 C5 0.22µF L2 10µH D2 R3 10.2k GND 22pF VOUT2 3.3V, 1A R4 34k R5 10k C7 47µF 3988 TA06 C1 TO C7: X5R OR X7R D1, D2: DIODES, INC. B180 EN/UVLO THRESHOLD = 6.02V Related Parts PART NUMBER DESCRIPTION COMMENTS LT3509 36V with Transient Protection to 60V, Dual 700mA (IOUT), 2.2MHz, High Efficiency Step-Down DC/DC Converter VIN: 3.6V to 36V, VOUT(MIN) = 0.8V, IQ = 1.9mA, ISD = 1µA, 3mm × 4mm DFN-14, MSOP-16E LT3508 36V with Transient Protection to 40V, Dual 1.4A (IOUT), 2.5MHz, High Efficiency Step-Down DC/DC Converter VIN: 3.7V to 36V, VOUT(MIN) = 0.8V, IQ = 4.6mA, ISD = 1µA, 4mm × 4mm QFN-24, TSSOP-16E LT3980 58V with Transient Protection to 80V, 2A (IOUT), 2.4MHz, High Efficiency Step-Down DC/DC Converter with Burst Mode® Operation VIN: 3.6V to 58V, Transient to 80V, VOUT(MIN) = 0.79V, IQ = 75µA, ISD < 1µA, 3mm × 4mm DFN-16, MSOP-16E LT3970 40V, 350mA (IOUT), 2MHz, High Efficiency Step-Down DC/DC Converter with Only 2.5µA of Quiescent Current VIN: 4.2V to 40V, VOUT(MIN) = 1.2V, IQ = 2.5µA, ISD < 1µA, 2mm × 3mm DFN-10, MSOP-10 LT3990 60V, 350mA (IOUT), 2MHz, High Efficiency Step-Down DC/DC Converter with only 2.5µA of Quiescent Current VIN: 4.2V to 60V, VOUT(MIN) = 1.2V, IQ = 2.5µA, ISD < 1µA, 3mm × 3mm DFN-10, MSOP-16E LT3971 38V, 1.2A (IOUT), 2MHz, High Efficiency Step-Down DC/DC Converter with Only 2.8µA of Quiescent Current VIN: 4.2V to 38V, VOUT(MIN) = 1.2V, IQ = 2.8µA, ISD < 1µA, 3mm × 3mm DFN-10, MSOP-10E LT3991 55V, 1.2A (IOUT), 2MHz, High Efficiency Step-Down DC/DC Converter with Only 2.8µA of Quiescent Current VIN: 4.2V to 55V, VOUT(MIN) = 1.2V, IQ = 2.8µA, ISD < 1µA, 3mm × 3mm DFN-10, MSOP-10E LT3507/LT3507A 36V, Triple 2.4A,1.4A, and 1.4A (IOUT), 2.5MHz, High Efficiency Step-Down DC/DC Converter with LDO Controller VIN: 4V to 36V, VOUT(MIN) = 0.8V, IQ = 7mA, ISD = 1µA, 5mm × 7mm QFN-38 LT3680 36V, 3A, 2.4MHz High Efficiency MicroPower Step-Down DC/DC Converter VIN: 3.6V to 36V, VOUT(MIN) = 0.8V, IQ = 75µA, ISD < 1µA, 3mm × 3mm DFN-10, MSOP-10E LT3693 36V, 3A, 2.4MHz High Efficiency Step-Down DC/DC Converter VIN: 3.6V to 36V, VOUT(MIN) = 0.8V, IQ = 1.3mA, ISD < 1µA, 3mm × 3mm DFN-10, MSOP-10E LT3480 36V with Transient Protection to 60V, 2A (IOUT), 2.4MHz, High Efficiency Step-Down DC/DC Converter with Burst Mode Operation VIN: 3.6V to 38V, Transient to 60V, VOUT(MIN) = 0.78V, IQ = 70µA, ISD < 1µA, 3mm × 3mm DFN-10, MSOP-10E 3988f 24 Linear Technology Corporation LT 0412 • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com LINEAR TECHNOLOGY CORPORATION 2012