LT3694/LT3694-1 36V, 2.6A Monolithic Buck Regulator With Dual LDO DESCRIPTION FEATURES n n n n n n n n n n n Wide Input Range: 4V to 36V Overvoltage Shutdown Protects Circuit Through 70V Transients 2.6A Output Switching Regulator with Internal Power Switch Dual, Low Dropout, Linear Regulator Controllers with Programmable Current Limit Tracking/Soft-Start Inputs and Power Good Output Simplify Soft-Start and Supply Sequencing Uses Small Inductors and Ceramic Capacitors VOUT(MIN) = 0.75V (Buck and LDOs) Adjustable 250kHz to 2.5MHz Switching Frequency Accurate Enable Threshold Allows User Programmable Undervoltage Lockout Options for Clock Synchronization (LT3694) or Clock Output to Enable Synchronization to Other Switching Regulators (LT3694-1) Thermally Enhanced 28-Lead 4mm × 5mm QFN and 20-Lead TSSOP Packages The LT®3694/LT3694-1 are monolithic, current mode DC/DC converters with dual, low dropout regulator controllers. The switching converter is a step-down converter capable of generating up to 2.6A at its output. Each regulator has independent track/soft-start circuits simplifying power supply sequencing and interfacing with microcontrollers and DSPs. The switching frequency is set with a single resistor with a range of 250kHz to 2.5MHz. The high switching frequency permits the use of small inductors and ceramic capacitors leading to very small triple output solutions. The constantswitching frequency, combined with low impedance ceramic capacitors, results in low, predictable output ripple. Protection circuitry senses the current in the power switch and external Schottky catch diode to protect the LT3694 against short-circuit conditions. Frequency foldback and thermal shutdown provide additional protection. With its wide input voltage range of 4V to 36V, the LT3694 regulates a broad array of power sources from 4-cell batteries and 5V logic rails to unregulated wall transformers, lead acid batteries and distributed power supplies. The LT3694 can be synchronized to an external clock with the SYNC pin while the LT3694-1 offers a CLKOUT pin allowing other DC/DC converters to synchronize to the LT3694-1 clock. APPLICATIONS n n n n n Automotive Industrial DSL and Cable Modems Distributed Power Regulation Wall Transformer Regulation L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. TYPICAL APPLICATION Efficiency at VOUT = 3.3V VIN 4.5V TO 36V EN/UVLO VIN BIAS BST 0.22μF 1nF TRK/SS1 TRK/SS2 TRK/SS3 0.1Ω LIM2 OUT1 B340A LT3694 DA FB1 VC1 DRV2 OUT2 2.5V 450mA 100 4.7μH SW 41.2k 2.2μF 24.9k FB2 SYNC 10.7k 51.1k 10k OUT1 RT fSW = 800kHz 90 VIN = 12V VIN = 36V 80 70 2.2μF OUT3 1.8V 450mA 60 50 FB3 GND fSW = 800kHz VIN = 4.5V 0.1Ω 15.4k PGOOD 47μF 330pF LIM3 DRV3 34k OUT1 3.3V 1.7A EFFICIENCY (%) 4.7μF 11k 0 1 2 IOUT (A) 3 36941 TA01b 36941 TA01a 36941fa 1 LT3694/LT3694-1 ABSOLUTE MAXIMUM RATINGS (Note 1) VIN, EN/UVLO (Note 6) ............................... –0.3V to 70V BST ...........................................................................55V BST Above SW ..........................................................25V PGOOD......................................................................16V TRK/SS, VC, FB, RT, SYNC Pins ...................................6V BIAS, LIM2, LIM3 Pins ...............................................7V Operating Junction Temperature Range (Notes 2 and 5) LT3694E ............................................. –40°C to 125°C LT3694I .............................................. –40°C to 125°C Storage Temperature Range................... –65°C to 150°C Lead Temperature (Soldering, 10 Sec) (TSSOP Only) ................................................... 300°C PIN CONFIGURATION TOP VIEW SW SW GND GND VIN VIN TOP VIEW VIN 1 20 SW EN/UVLO 2 19 DA SYNC(CLKOUT) 3 18 BST PGOOD 4 17 BIAS 19 VC1 RT 5 18 FB1 TRK/SS1 6 17 TRK/SS3 TRK/SS2 7 14 TRK/SS3 FB2 8 13 FB3 DRV2 9 12 DRV3 LIM2 10 11 LIM3 28 27 26 25 24 23 22 DA EN/UVLO 1 SYNC (CLKOUT) 2 21 BST PGOOD 3 20 BIAS 29 GND RT 4 TRK/SS1 5 TRK/SS2 6 FB2 7 16 FB3 DRV2 8 15 DRV3 LIM3 GND GND GND GND LIM2 9 10 11 12 13 14 UFD PACKAGE 28-LEAD (4mm s 5mm) PLASTIC QFN θJA = 34°C/W EXPOSED PAD (PIN 29) IS GND, MUST BE SOLDERED TO PCB LT3694-1 PINOUT IS SHOWN IN PARENTHESIS 21 GND 16 VC1 15 FB1 FE PACKAGE 20-LEAD PLASTIC TSSOP θJA = 38°C/W EXPOSED PAD (PIN 21) IS GND, MUST BE SOLDERED TO PCB LT3694-1 PINOUT IS SHOWN IN PARENTHESIS ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT3694EUFD#PBF LT3694EUFD#TRPBF 3694 28-Lead (4mm × 5mm) Plastic QFN –40°C to 125°C LT3694IUFD#PBF LT3694IUFD#TRPBF 3694 28-Lead (4mm × 5mm) Plastic QFN –40°C to 125°C LT3694EFE#PBF LT3694EFE#TRPBF LT3694FE 20-Lead Plastic TSSOP –40°C to 125°C LT3694IFE#PBF LT3694IFE#TRPBF LT3694FE 20-Lead Plastic TSSOP –40°C to 125°C LT3694-1EUFD#PBF LT3694-1EUFD#TRPBF 36941 28-Lead (4mm × 5mm) Plastic QFN –40°C to 125°C LT3694-1IUFD#PBF LT3694-1IUFD#TRPBF 36941 28-Lead (4mm × 5mm) Plastic QFN –40°C to 125°C LT3694-1EFE#PBF LT3694-1EFE#TRPBF LT3694FE-1 20-Lead Plastic TSSOP –40°C to 125°C LT3694-1IFE#PBF LT3694-1IFE#TRPBF LT3694FE-1 20-Lead Plastic TSSOP –40°C to 125°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ 36941fa 2 LT3694/LT3694-1 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C, VIN = 12V, VBIAS = 3V, unless otherwise noted. (Notes 2, 9) PARAMETER CONDITIONS MIN TYP VIN Internal Undervoltage Lockout l MAX UNITS 3.5 3.8 4 V Overvoltage Shutdown Threshold l 36 38 40 V Input Quiescent Current Not Switching 1 2 mA Bias Quiescent Current Not Switching 2 3.5 mA Shutdown Current VEN/UVLO = 0.1V 0.1 2 μA EN/UVLO Threshold, Bias On l EN/UVLO Threshold, Switching On Reference Voltage Line Regulation 5V < VIN < 36V Switching Frequency RT = 40.2K 350 500 1.16 1.2 mV 1.23 0.01 l 0.9 VIH, SYNC l 1.5 VIL, SYNC l VOH, CLKOUT ICLKOUT = –50μA l VOL, CLKOUT ICLKOUT = 50μA l PGOOD Output Voltage Low IPGOOD = 250μA PGOOD Leakage VPGOOD = 2V PGOOD Threshold (Relative to VFB) (Note 8) 1.0 V %/V 1.1 MHz V 0.35 V 2.6 V 0.3 V 0.2 0.4 V 10 1000 nA 86 90 94 % 735 750 765 mV –50 –500 nA 1.6 Switching Regulator Feedback Pin Voltage l Feedback Pin Bias Current l Error Amplifier Transconductance 350 Error Amplifier Voltage Gain μS 600 V/V TRK/SS Pull-Up Current –2 –3 –4 μA TRK/SS Threshold to Start Switching 35 50 70 mV VC1 Source Current VC = 0.6V –20 μA VC1 Sink Current VC = 0.6V 28 μA VC1 Clamp Voltage 2 V VC1 Switching Threshold 0.75 V VC1 to Switch Current Gain 3.6 A/V Switch Leakage Current VIN = 36V Minimum Boost Voltage Above Switch (Note 4) l 3.5 0.01 10 μA 1.8 2.5 V 4.9 6 A Switch Current Limit (Note 3) (Note 3) 10% Duty Cycle Switch VCESAT ISW1 = 3A 600 mV BST Operating Current ISW1 = 3A 60 mA VF , BST Diode IBST = 100mA 0.8 V IL BST Diode VBST – VBIAS = 36V 1 μA DA Current Limit l Minimum Switch Off-Time l 2.6 3.6 4.5 A 140 ns 36941fa 3 LT3694/LT3694-1 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C, VIN = 12V, VBIAS = 3V, unless otherwise noted. (Notes 2, 9) PARAMETER CONDITIONS MIN TYP MAX UNITS 735 750 765 mV –50 –500 nA –3 –4 μA 50 70 LDO Regulator Feedback Pin Voltage l Feedback Pin Bias Current l Error Amplifier Voltage Gain 2800 TRK/SS Pull-Up Current –2 TRK/SS Threshold to Shut Down LDO 35 Line Regulation 5V < VIN < 36V 0.025 Load Regulation IDRV From 0.1mA to 10mA mV %/V 0.5 mV/mA Base Drive l 10 15 20 mA Current Limit Threshold l 47 60 70 mV Short-Circuit Current Limit Threshold VFB = 0 26 30 mV Minimum BIAS to DRV Voltage (Note 7) IDRV = 10mA l 0.3 0.9 V Minimum VIN to DRV Voltage IDRV = 10mA l 2.0 2.3 V Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LT3694E is guaranteed to meet performance specifications from 0°C to 125°C junction temperature. Specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LT3694I is guaranteed to meet performance specifications from –40°C to 125°C junction temperature. Note 3: Current limit is guaranteed by design and/or correlation to static test. Slope compensation reduces current limit at higher duty cycles. Note 4: This is the minimum voltage across the boost capacitor needed to guarantee full saturation of the internal power switch. 22 Note 5: This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed the maximum operating range when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. Note 6: Absolute Maximum Voltage at VIN and EN/UVLO pins is 70V for non-repetitive, 1 second transients and 36V for continuous operation. Note 7: The LDO will function if the BIAS to DRV differential is not met, but the base drive current will be drawn from VIN instead of BIAS. Note 8: The PGOOD pin will pull low when the voltage on any of the three FB pins is lower than the PGOOD threshold value. Note 9: Positive currents flow into pins, negative currents flow out of pins. Minimum and maximum values refer to absolute values. 36941fa 4 LT3694/LT3694-1 TYPICAL PERFORMANCE CHARACTERISTICS Efficiency at VOUT = 5V 100 VIN = 12V, TA = 25°C, unless otherwise noted. BST Pin Current vs Switch Current Switch VCESAT vs Switch Current fSW = 800kHz VIN = 6.3V 0.8 70 0.7 60 0.6 SWITCH VCESAT (V) EFFICIENCY (%) VIN = 12V BOOST PIN CURRENT (mA) 90 80 VIN = 36V 70 0.5 0.4 0.3 0.2 60 50 40 30 20 10 0.1 50 2 1 0 IOUT (A) 3 0 0 0 36941 G01 2 1 SWITCH CURRENT (A) 3 1 2 SWITCH CURRENT (A) 0 36941 G02 3 36941 G03 Switch Current Limit vs Duty Cycle Switch Current Limit vs Temperature 5.0 5.0 160 –45°C 25°C 3.5 MINIMUM ON-TIME 120 150°C TIME (ns) SWITCH ILIM (A) 4.0 ISW = 1A 140 4.5 4.5 SWITCH ILIM (A) Switch Minimum On-Time and Off-Time vs Temperature 4.0 100 MINIMUM OFF-TIME 80 60 3.5 40 20 3.0 –50 100 50 TEMPERATURE (°C) 150 0 36941 G04 80 20 40 60 SWITCH DUTY CYCLE (%) 100 VFB vs Temperature 5 2.5 756 FREQUENCY (MHz) VFB (mV) NORMALIZED FREQUENCY SHIFT (%) 758 752 750 748 746 744 2.0 1.5 1.0 0.5 742 0 0 100 50 TEMPERATURE (°C) 150 36941 G07 100 50 TEMPERATURE (°C) 0 50 100 RT (k) 150 36941 G06 Frequency Shift vs Temperature Frequency vs RT 3.0 754 0 36941 G05 760 740 –50 0 –50 3.0 0 150 200 36941 G08 RT = 200k 4 RT = 40.2k 3 2 1 RT = 10.7k 0 –1 –2 –3 –4 –5 –50 0 100 50 TEMPERATURE (°C) 150 36941 G09 36941fa 5 LT3694/LT3694-1 TYPICAL PERFORMANCE CHARACTERISTICS EN/UVLO Thresholds vs Temperature ITRK/SS vs Temperature 4.0 Minimum Input Voltage vs Load Current (VIN to Start) 7.0 1.4 UVLO SWITCHING THRESHOLD 3.5 3.0 2.5 TO START 1.0 0.8 BIAS CURRENT SHUTDOWN THRESHOLD 0.6 0.4 0 0 –50 150 100 50 TEMPERATURE (°C) 5.5 TO RUN 5.0 100 50 TEMPERATURE (°C) 4.0 0.001 150 0.01 0.1 LOAD CURRENT (A) 36941 G11 1 36941 G12 LDO Minimum VIN to DRV Voltage vs DRV Current LDO Current Limit vs VFB (Foldback) 2.5 60 5 +150°C 4 3 2 1 0 –1 –2 –3 50 2.0 –40°C VIN TO DRV VOLTAGE (V) CURRENT LIMIT VOLTAGE (mV) NORMALIZED CURRENT LIMIT (%) 0 36941 G10 LDO Current Limit vs Temperature 40 30 20 10 –4 –5 –50 6.0 4.5 0.2 2.0 –50 VOUT = 5V fSW = 800kHz 6.5 INPUT VOLTAGE (V) EN/UVLO THRESHOLD (V) 1.2 TRK/SS CURRENT (μA) VIN = 12V, TA = 25°C, unless otherwise noted. 0 100 50 TEMPERATURE (°C) 150 0.4 0.6 0.2 FEEDBACK VOLTAGE (V) 0.8 BIAS TO DRV VOLTAGE (V) 0 0 36941 G14 36941 G13 2 4 6 DRV CURRENT (mA) 8 10 36941 G15 LDO Minimum BIAS to DRV Voltage vs DRV Current 0.5 1.0 0.5 0 0 1.5 10Hz to 100kHz LDO Output Noise VOUT = 2.5V IOUT = 0.25A VIN = 5V VBIAS = 4.4V ZXTCM322 PASS XSTR 0.4 0.3 10mV/DIV 0.2 0.1 0 0 2 4 6 DRV CURRENT (mA) 8 10 1ms/DIV 36941 G17 36941 G16 36941fa 6 LT3694/LT3694-1 PIN FUNCTIONS (FE/UFD) VIN (Pin 1/Pins 27, 28): The VIN pin supplies power to the internal switch of the 2.6A regulator and to the LT3694’s internal reference and start-up circuitry. This pin must be locally bypassed. EN/UVLO (Pin 2/Pin 1): The EN/UVLO pin is used to shut down the LT3694. It can be driven from a logic level or used as an undervoltage lockout by connecting a resistor divider from VIN. CLKOUT (Pin 3/Pin 2): Digital Clock Output. The CLKOUT pin allows synchronization of other switching regulators (LT3694-1 only). SYNC (Pin 3/Pin 2): Frequency Synchronization Input. Connect a frequency source to this input if synchronization is desired. Connect SYNC to ground if not used (LT3694 only). PGOOD (Pin 4/Pin 3): Open Collector Output. PGOOD is pulled low when any of the three regulators drops out of regulation (VFB < 90% of nominal value). RT (Pin 5/Pin 4): The RT pin requires a resistor to ground to set the operating frequency of the LT3694. If synchronizing the LT3694 to an external clock, the resistor should be set to program the frequency at least 20% below the synchronization frequency. TRK/SS1, TRK/SS2 , TRK/SS3 (Pins 6, 7, 14/Pins 5, 6, 17): The TRK/SS pins allow a regulator to track the output of another regulator. When the TRK/SS pin is below 0.75V, the FB pin regulates to the TRK/SS voltage. This pin can also be used as a soft-start by connecting a capacitor from TRK/SS to ground. The TRK/SS pins should be left open if neither feature is used. FB1, FB2, FB3 (Pins 15, 8, 13/Pins 18, 7, 16): Negative Inputs of the Error Amplifiers. The LT3694 regulates each feedback pin to the lesser of 0.75V or the corresponding TRK/SS pin voltage. Connect the feedback resistor divider taps to these pins. for the LDO regulators. The DRV pins can provide up to 6V of base drive. LIM2, LIM3 (Pins 10, 11/Pins 9, 14): The LIM pins provide current limiting on the LDO pass transistors by sensing a voltage on an external sense resistor connected to the BIAS pin. These pins should be connected to BIAS if this function is not used. GND (Pins 10, 11, 12, 13, 25, 26) UFD Package Only: Power and Signal Ground. VC1 (Pin 16/Pin 19): Output of the Internal Error Amp. The voltage on this pin controls the peak switch current. This pin is normally used to compensate the control loop. The switching regulator can be shut down by pulling the VC1 pin to ground with an NMOS or NPN transistor. BIAS (Pin 17/Pin 20): The BIAS pin supplies the current to the LT3694’s internal regulator and boost circuits. This must be connected to a voltage source above 3V, usually to VOUT1. The LDO pass transistor base current will also come from the BIAS pin if it is at least 1.8V above the LDO output. BST (Pin 18/Pin 21): The BST pin is used to provide a drive voltage, higher than the input voltage, to the internal bipolar NPN power switch. DA (Pin 19/Pin 22): The DA pin senses the catch diode current to prevent excessive inductor current in output overload or short-circuit conditions. SW (Pin 20/Pins 23, 24): Output of the Internal Power Switch. Connect this pin to the inductor and switching diode. Exposed Pad (Pin 21/Pin 29): Ground. The underside exposed pad metal of the package provides both elec-trical contact to ground and a conductive thermal path to the printed circuit board. The Exposed Pad must be soldered to a grounded pad on the circuit board for proper operation. DRV2, DRV3 (Pins 9, 12/Pins 8, 15): The DRV pins provide the base drive for the external NPN transistors 36941fa 7 LT3694/LT3694-1 BLOCK DIAGRAM RT BIAS SYNC (LT3694) CLKOUT (LT3694-1) OUT1 VIN CLK EN/UVLO + – 0.5V INT REG AND REF – + 1.2V PGOOD OVERVOLTAGE SHUTDOWN MASTER OSC THERMAL SHUTDOWN LDO OUT1 60mV LDO – RLIM3 PG1 + LIM3 + RLIM2 LIM2 OUT1 – 60mV DRV3 DRV2 3μA 3μA TRK/SS2 0.75V FB2 + + SD TRK/SS3 FB3 – – – + 0.68V 0.68V BUCK VIN + VIN CIN BIAS – 0.9V 3 OUT3 0.75V SD + – OUT2 + + + BST R – S SLOPE COMP O C3 L1 SW OUT1 CLK DA + – RC – + + Cf ERROR AMP R1 FB1 – VC1 C1 D1 R2 0.75V 3μA + GND 2V ILIMIT CLAMP PG1 + – CC 0.68V TRK/SS1 36941 F01 Figure 1. LT3694 Block Diagram with Typical External Components 36941fa 8 LT3694/LT3694-1 OPERATION Unless specifically noted, this data sheet refers to both the LT3694 and the LT3694-1 generically as the LT3694. The LT3694 is a constant-frequency, current mode, buck regulator with an internal power switch plus two low dropout linear regulator controllers. The three regulators share common circuitry including input source, voltage reference, undervoltage lockout, and enable, but are otherwise independent. Operation can be best understood by referring to the Block Diagram (Figure 1). the current through the inductor to the output. The internal error amplifier regulates the output voltage by continually adjusting the VC1 pin voltage. The threshold for switching on the VC1 pin is 0.75V and an active clamp of 2V limits the output current. Overcurrent protection is provided by the DA comparator. The DA comparator senses the catch diode current and will delay the switch-on cycle if the diode current is too high at the beginning of a cycle. If the EN/UVLO pin is below 0.35V (min), the LT3694 is shut down and draws <2μA from the input source tied to VIN1. If the EN/UVLO pin is driven above 0.5V (typ), the internal bias circuits turn on, including the internal regulator, reference and master oscillator. The switching regulator will only begin to operate when the EN/UVLO pin reaches >1.20V (typ). The EN/UVLO pin can be driven from a logic gate or can be used as an undervoltage lockout by using a resistor divider to VIN. The switch driver operates either from VIN or from the BST pin. An external capacitor is used to generate a voltage at the BST pin that is higher than the input supply. This allows the driver to saturate the internal bipolar NPN power switch for efficient operation. The switcher is a current mode regulator. Instead of directly modulating the duty cycle of the power switch, the feedback loop controls the peak current in the switch during each cycle. Compared to voltage mode control, current mode control improves loop dynamics and provides cycle-bycycle current limit. The BIAS pin allows the internal circuitry to draw its current from a voltage supply lower than VIN, reducing power dissipation and increasing efficiency. If the voltage on the BIAS pin falls below 2.7V, then its quiescent current will flow from VIN. A pulse from the oscillator sets the RS flip-flop and turns on the internal NPN bipolar power switch. Current in the switch and the external inductor begins to increase. When this current exceeds a level determined by the voltage at VC1, the current comparator resets the RS flip-flop, turning off the switch. The current in the inductor flows through the external, Schottky, catch diode, and begins to decrease. The cycle begins again at the next pulse from the oscillator. In this way, the voltage on the VC1 pin controls The TRK/SS pins override the 0.75V reference for the FB pins when the TRK/SS pins are below 0.75V. This allows either coincident or ratiometric supply tracking on start-up as well as a soft-start capability. The LDO regulator uses an external NPN pass transistor to form a linear regulator. The loop is internally compensated to be stable with a minimum load capacitance of 2.2 μF. The LDO also has a foldback current limiter available to protect the external transistor under overload conditions The overvoltage detection shuts down the LT3694 if the input voltage goes above 38V. This will prevent the switch from turning on under high voltage conditions and allows the LT3694 to survive transient input voltages up to 70V. 36941fa 9 LT3694/LT3694-1 APPLICATIONS INFORMATION STEP DOWN SWITCHING REGULATOR Feedback Resistor Network The output voltage is programmed with a resistor divider (refer to the Block Diagram in Figure 1) between the output and the FB pin. Choose the resistors according to: ⎛ V ⎞ R1= R2 ⎜ OUT − 1⎟ ⎝ 750mV ⎠ The parallel combination of R1 and R2 should be 10k or less to avoid bias current errors. each clock cycle if there is sufficient voltage across the boost capacitor (C3 in Figure 1) to fully saturate the output switch. A forced switch off for a minimum time will only occur at the end of a clock cycle when the boost capacitor needs to be recharged. This operation has the same effect as lowering the clock frequency for a fixed off time, resulting in a higher duty cycle and lower minimum input voltage. The resultant duty cycle depends on the charging times of the boost capacitor and can be approximated by the following equation: DCMAX = B B+1 Input Overvoltage Lockout An important feature of the LT3694 is the ability to survive transient surges on the input voltage of up to 70V. This is accomplished by shutting off the regulators to keep this high voltage off the critical components. The overvoltage lockout trips when the input voltage exceeds 38V. Input Voltage Range The minimum operating voltage is determined either by the LT3694’s internal undervoltage lockout or by its maximum duty cycle. The duty cycle is the fraction of time that the internal switch is on and is determined by the input and output voltage: DC = VOUT + VF VIN − VSW + VF where VF is the forward voltage drop of the catch diode and VSW is the voltage drop of the internal switch (~0.3V at maximum load). This leads to a minimum input voltage of: VIN(MINCF) = VOUT + VF − VF + VSW DCMAX(CF) The duty cycle is the fraction of time that the internal switch is on during a clock cycle. The maximum duty cycle for constant-frequency operation given by DCMAX(CF) = 1 – tOFF(MIN) • fSW . However, unlike most fixed frequency regulators, the LT3694 will not switch off at the end of where B is the output current divided by the typical boost current from the BST Pin Current vs Switch Current curve in the Typical Performance Characteristics section. The maximum voltage, VIN, for constant-frequency operation is determined by the minimum duty cycle DCMIN: VIN(MAXCF) = VOUT + VF − VF + VSW DCMIN with DCMIN = tON(MIN) • fSW Thus, both the maximum and minimum input voltages for constant-frequency operation are a function of the switching frequency and output voltage. Therefore, the maximum switching frequency must be set to a value that accommodates the input and output voltage parameters and must meet both of the following criteria: ⎛ ⎞ VOUT + VF 1 fMAX1 = ⎜ ⎟• ⎝ VIN(MAXCF) − VSW + VF ⎠ tON(MIN) ⎛ ⎞ VOUT + VF 1 fMAX2 = ⎜ 1− ⎟• ⎝ VIN(MINCF) − VSW + VF ⎠ tOFF(MIN) The values of tON(MIN) and tOFF(MIN) are functions of ISW and temperature (see chart in the Typical Performance Characteristics section). Worst-case values for switch currents greater than 0.5A are tON(MIN) = 130ns and 36941fa 10 LT3694/LT3694-1 APPLICATIONS INFORMATION tOFF(MIN) = 140ns. fMAX1 is the frequency at which the minimum duty cycle is exceeded. The regulator will skip ON pulses in order to reduce the overall duty cycle at frequencies above fMAX1. It will continue to regulate but with increased inductor current and greatly increased output ripple. The increased peak inductor current in pulse-skipping will also stress the switch transistor at high voltages and high switching frequency. fMAX2 is the frequency at which the maximum duty cycle is exceeded. If there is sufficient charge on the BST capacitor, the regulator will skip OFF periods to increase the overall duty cycle at frequencies above fMAX2. It will continue to regulate but will not have constant-frequency operation. Note that the restriction on the operating input voltage refers to steady-state limits to keep the output in regulation in constant-frequency mode; the circuit will tolerate input voltage transients up to the absolute maximum rating. Switching Frequency Once the upper limit for the switching frequency is found from the duty cycle requirements, the frequency may be chosen below the upper limit. Lower frequencies result in lower switching losses, but require larger inductors and capacitors. The user must decide the best trade-off. The switching frequency is set by a resistor connected from the RT pin to ground, or by forcing a clock signal into the SYNC pin (LT3694 only). The LT3694 applies a voltage of 0.75V across this resistor and uses the current to set the oscillator speed. The switching frequency is given by the following formula: 49.8 fSW = R T + 8.8 where fSW is in MHz and RT is in kΩ. The formula is accurate within ±2% over the frequency range. Table 1 shows the typical measured value of RT for several common switching frequencies. Table 1: RT for Common Frequencies SWITCHING FREQUENCY (MHz) RT (k) 0.25 193 0.5 90.2 0.75 56.6 1 40.2 1.25 30.5 1.5 23.8 1.75 19.6 2 16.0 2.25 13.5 2.5 11.4 For external clocks applied to the SYNC pin (LT3694 only), the circuit will support VH logic levels from 1.8V to 5V CMOS or TTL. The duty cycle needs a minimum on time of 100ns and a minimum off time of 100ns. When operating in sync mode, RT should be set to provide a frequency at least 20% below the minimum sync frequency. Inductor Selection and Maximum Output Current A good first choice for the inductor value is: L= VOUT + VF 1.25A • f where f is the switching frequency in MHz, L is the inductor value in μH, VOUT is the output voltage and VF is the catch diode voltage drop. The current in the inductor is a triangle wave with an average value equal to the load current. The peak switch current is equal to the output current plus half the peak-topeak inductor ripple current. The LT3694 limits its switch current in order to protect itself and the system from overload faults. Therefore, the maximum output current that the LT3694 will deliver depends on the switch current limit, the inductor value and the input and output voltages. When the switch is off, the potential across the inductor is the output voltage plus the catch diode drop. This gives the peak-to-peak ripple current in the inductor: ΔIL = (1− DC) VOUT + VF L•f 36941fa 11 LT3694/LT3694-1 APPLICATIONS INFORMATION where f is the switching frequency of the LT3694 and L is the value of the inductor. The peak inductor and switch current is: ΔI ISWPK =ILPK =IOUT + L 2 To maintain output regulation, this peak current must be less than the LT3694’s switch current limit, ILIM. ILIM is at least 3.5A at low duty cycles (0.1) and decreases linearly to 2.8A at DC = 0.8. The minimum inductance can now be calculated as: L MIN = 1− DCMIN VOUT + VF • 2•f ILIM −IOUT However, it’s generally better to use an inductor larger than the minimum value. The minimum inductor has large ripple currents which increase core losses and require large output capacitors to keep output voltage ripple low. Select an inductor greater than LMIN that keeps the ripple current below 30% of ILIM. For input voltages greater than 30V, use an inductor with a saturation current of 6A or greater and an inductance value of 3.3μH or greater. The inductor’s RMS current rating must be greater than the maximum load current and its saturation current should be greater than ILPK. For highest efficiency, the series resistance (DCR) should be less than 0.1Ω. Table 2 lists several vendors and types that are suitable. Table 2. Inductors SERIES INDUCTANCE RANGE (μH) CURRENT RANGE (A) WE-HC 1 to 6.5 6 to 15 Würth Elektronik www.we-online.com MSS1048 0.8 to 8 4 to 8 Coilcraft www.coilcraft.com CDRH103R 0.8 to 10 2.8 to 8.3 Sumida www.sumida.com VLF 2.2 to 10 3.8 to 7.7 TDK www.component.tdk. com 1 to 10 2.5 to 9.5 Vishay www.vishay.com IHLP-2525CZ-11 MANUFACTURER This analysis is valid for continuous mode operation (IOUT > ILIM/2). For details of maximum output current in discontinuous mode operation, see the Linear Technology Application Note 44. Finally, for duty cycles greater than 50% (VOUT /VIN > 0.5), a minimum inductance is required to avoid subharmonic oscillations. This minimum inductance is: LMIN = (VOUT + VF ) 2A • fSW with LMIN in μH and fSW in MHz. A detailed discussion of subharmonic oscillations can be found in the Linear Technology Application Note 19. Input Capacitor Selection Bypass the input of the LT3694 circuit with a ceramic capacitor of X7R or X5R type. Y5V types have poor performance over temperature and applied voltage, and should not be used. A 4.7μF to 22μF ceramic capacitor is adequate to bypass the LT3694 and will easily handle the ripple current. Use a 22μF capacitor with fSW between 250kHz and 800kHz. Use a 10μF capacitor with fSW between 800kHz and 1.6MHz. Use a 4.7μF capacitor above 1.6MHz. Always check for sufficient margin by reducing the capacitor value until the dropout increases by >500mV. If the input power source has high impedance, or there is significant inductance due to long wires or cables, additional bulk capacitance may be necessary. This can be provided with a lower performance electrolytic capacitor. Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input capacitor is required to reduce the resulting voltage ripple at the LT3694 and to force this very high frequency switching current into a tight local loop, minimizing EMI. A 10μF capacitor is capable of this task, but only if it is placed close to the LT3694 and the catch diode (see the PCB Layout section). A second precaution regarding the ceramic input capacitor concerns the maximum input voltage rating of the LT3694. A ceramic input capacitor combined with trace or cable inductance forms a high 36941fa 12 LT3694/LT3694-1 APPLICATIONS INFORMATION quality (under damped) tank circuit. If the LT3694 circuit is plugged into a live supply, the input voltage can ring to twice its nominal value, possibly exceeding the LT3694’s maximum input voltage rating. See Linear Technology Application Note 88 for more details. Output Capacitor Selection The output capacitor filters the inductor current to generate an output with low voltage ripple. It also stores energy in order to satisfy transient loads and stabilize the LT3694’s control loop. Because the LT3694 operates at a high frequency, minimal output capacitance is necessary. In addition, the control loop operates well with or without the presence of output capacitor series resistance (ESR). Ceramic capacitors, which achieve very low output ripple and small circuit size, are therefore an option. Output ripple can be estimated with the following equations: ΔIL VRIPPLE = ; Ceramic 8 • f • COUT VRIPPLE = ΔIL • ESR ; Electrolytic where ΔIL is the peak-to-peak ripple current in the inductor. The RMS content of this ripple is very low so the RMS current rating of the output capacitor is usually not of concern. It can be estimated with the formula: IC(RMS) = ΔIL 12 Another constraint on the output capacitor is that it must have greater energy storage than the inductor; if the stored energy in the inductor transfers to the output, the resulting voltage step should be small compared to the regulation voltage. For a 5% overshoot, this requirement indicates: ⎛ I ⎞ COUT > 10 • L • ⎜ LIM ⎟ ⎝ VOUT ⎠ 2 The low ESR and small size of ceramic capacitors make them the preferred type for LT3694 applications. Not all ceramic capacitors are the same, however. Many of the higher value capacitors use poor dielectrics with high temperature and voltage coefficients. In particular, Y5V and Z5U types lose a large fraction of their capacitance with applied voltage and at temperature extremes. Because loop stability and transient response depend on the value of COUT , this loss may be unacceptable. Use X7R and X5R types instead. Electrolytic capacitors are also an option. The ESRs of most aluminum electrolytic capacitors are too large to deliver low output ripple. Surge rated tantalum capacitors or low ESR, organic, electrolytic capacitors intended for power supply use are suitable. Choose a capacitor with a sufficiently low ESR for the required output ripple. Because the volume of the capacitor determines its ESR, both the size and the value will be larger than a ceramic capacitor that would give similar ripple performance. One benefit is that the larger capacitance may give better transient response for large changes in load current. Table 3 lists several capacitor vendors. Table 3. Low ESR Surface Mount Capacitors SERIES TYPE Ceramic TPM, TPS Ceramic, Tantalum MANUFACTURER Taiyo Yuden www.t-yuden.com AVX www.avx.com T494, T495, T510, T520, T525, T530, A700 Ceramic, Tantalum, Tantalum Organic Polymer, Aluminum Organic Polymer Kemet www.kemet.com POSCAP, OS-CON Tantalum Organic Polymer, Aluminum Organic Polymer Sanyo www.sanyo.com SP-CAP Ceramic, Aluminum Organic Polymer Panasonic www.panasonic.com Ceramic TDK www.tdk.com 36941fa 13 LT3694/LT3694-1 APPLICATIONS INFORMATION ID(AVG) =IOUT • VIN − VOUT VIN Consider a diode with a larger current rating than ID(AVG) when the part must survive a shorted output. The DA pin monitors the current in the diode and prevents the switch from turning on at the beginning of a charge cycle if the diode current is above the DA limit. Therefore, under overload conditions, the average diode current will increase to the average of the switch current limit and the DA current limit. Peak reverse voltage is equal to the regulator input voltage, so use a diode with a reverse voltage rating greater than the maximum input voltage. The internal OVLO can protect the diode from excessive reverse voltage by shutting down the regulator if the input voltage exceeds 38V. Table 4 lists several Schottky diodes and their manufacturers. LT3694 CURRENT MODE POWER STAGE gm = 7.5S SW ERROR AMPLIFIER OUTPUT R1 FB + The catch diode (D1 from Figure 1) conducts current only during switch off time. Average forward current in normal operation can be calculated from: VC pin, as shown in Figure 2. Generally a capacitor (CC) and a resistor (RC) in series to ground are used. In addition, there may be lower value capacitor in parallel. This capacitor (CF) is not part of the loop compensation but is used to filter noise at the switching frequency, and is required only if a phase-lead capacitor (CPL) is used or if the output capacitor (C1) has high ESR. – Diode Selection 0.75V gm = 350μS CPL ESR + 3M C1 C1 VC CF POLYMER OR TANTALUM GND RC CERAMIC R2 CC 36941 F02 Figure 2. Model for Loop Response Table 4. Schottky Diodes (40V, 3A) PART NUMBER Vf at 3A (V) OUTLINE MBRS340 MBRD340 0.5 0.6 SMC D-PAK B340 SMB340 0.5 0.5 SMC Powermite 3 CMSH3-40 CSHD3-40 0.5 0.65 SMC D-PAK MANUFACTURER ON Semiconductor www.onsemi.com Diodes, Inc. www.diodes.com Central Semiconductor www.centralsemi.com Frequency Compensation The LT3694 uses current mode control to regulate the output. This simplifies loop compensation. In particular, the LT3694 does not require the ESR of the output capacitor for stability, so the user is free to employ ceramic capacitors to achieve low output ripple and small circuit size. Frequency compensation is provided by the components tied to the Loop compensation determines the stability and transient performance. The best values for the compensation network depend on the application and in particular the type of output capacitor. A practical approach is to start with one of the circuits in this data sheet that is similar to your application and tune the compensation network to optimize the performance. Stability should then be checked across all operating conditions, including load current, input voltage and temperature. The LT1375 data sheet contains a more thorough discussion of loop compensation and describes how to test the stability using a transient load. Figure 2 shows an equivalent circuit for the LT3694 control loop. The error amplifier is a transconductance amplifier with finite output impedance. 36941fa 14 LT3694/LT3694-1 APPLICATIONS INFORMATION The power section, consisting of the modulator, power switch and inductor, is modeled as a transconductance amplifier generating an output current proportional to the voltage at the VC1 pin. Note that the output capacitor integrates this current, and that the capacitor on the VC1 pin (CC) integrates the error amplifier output current, resulting in two poles in the loop. In most cases a zero is required and comes from either the output capacitor ESR or from a resistor RC in series with CC. This simple model works well as long as the value of the inductor is not too high and the loop crossover frequency is much lower than the switching frequency. A phase lead capacitor (CPL) across the feedback divider may improve the transient response. Figure 3 shows the transient response when the load current steps from 1A to 2.6A and back to 1A. capacitor. A 2.5V output presents a special case because it is marginally adequate to support the boosted drive stage while using the internal boost diode. For reliable BST pin operation with 2.5V outputs, use a good external Schottky diode (such as the ON Semi MBR0540), and a 1μF boost capacitor (see Figure 4b). For lower output voltages, the BIAS pin can be tied to the input (Figure 4c), or to another supply greater than 2.8V. Tying BIAS to VIN reduces the maximum input voltage to 7V. The circuit in Figure 4a is more efficient because the BST pin current and BIAS pin quiescent current comes from a lower voltage source. One must also ensure that the maximum voltage ratings of the BST and BIAS pins are not exceeded. The minimum VOUT BIAS VIN VIN GND 4.7μF BST LT3694 C3 SW VOUT 100mV/DIV (4a) For VOUT > 2.8V VOUT D2 IL 1A/DIV BIAS VIN 100μs/DIV 36941 F03 VIN Figure 3. Transient Load Response of the LT3694 Front Page Application as the Load Current Is Stepped from 1A to 2.6A. VOUT = 3.3V LT3694 GND 4.7μF BST C3 SW (4b) For 2.5V < VOUT < 2.8V BST and BIAS Pin Considerations Capacitor C3 and the internal boost Schottky diode (see the Block Diagram in Figure 1) are used to generate a boost voltage that is higher than the input voltage. In most cases a 0.22μF capacitor will work well. Figure 4 shows three ways to arrange the boost circuit. The BST pin must be more than 2.3V above the SW pin for best efficiency. For outputs of 3V and above, the standard circuit (Figure 4a) is best. For outputs between 2.8V and 3V, use a 1μF boost VOUT BIAS VIN 4.7μF VIN BST LT3694 GND C3 SW 36941 FO4 (4c) For VOUT < 2.5V; VIN(MAX) = 7V Figure 4. Three Circuits for Generating the Boost Voltage 36941fa 15 LT3694/LT3694-1 APPLICATIONS INFORMATION At light loads, the inductor current becomes discontinuous and the effective duty cycle can be very high. This reduces the minimum input voltage to approximately 300mV above VOUT. At higher load currents, the inductor current is continuous and the duty cycle is limited by the maximum duty cycle of the LT3694, requiring a higher input voltage to maintain regulation. Internal Undervoltage Lockout The LT3694 features an internal undervoltage lockout that will shut off all three regulators if the input voltage drops too low to maintain regulation of the internal circuitry. This lockout trips when VIN drops below 3.8V (typ). 5.0 VOUT = 3.3V fSW = 800kHz 4.8 TO START INPUT VOLTAGE (V) 4.6 4.4 4.2 4.0 TO RUN 3.8 3.6 3.4 3.2 3.0 0.001 7.0 0.01 0.1 LOAD CURRENT (A) 1 VOUT = 5V fSW = 800kHz 6.5 TO START INPUT VOLTAGE (V) operating voltage of an LT3694 application is limited by the minimum input voltage (4V) and by the maximum duty cycle as outlined in a previous section. For proper start-up, the minimum input voltage is also limited by the boost circuit. If the input voltage is ramped slowly, or the LT3694 is turned on with its EN/UVLO or TRK/SS pin when the output is already in regulation, then the boost capacitor may not be fully charged. Because the boost capacitor is charged with the energy stored in the inductor, the circuit will rely on some minimum load current to get the boost circuit running properly. This minimum load will depend on input and output voltages, and on the arrangement of the boost circuit. The minimum load generally goes to zero once the circuit has started. Figure 5 shows a plot of input voltage to start and to run as a function of load current. In many cases the discharged output capacitor will present a load to the switcher, which will allow it to start. The plots show the worst-case situation in which VIN is ramping very slowly. For lower start-up voltage, the boost diode can be tied to VIN, however, this restricts the input range to one-half of the absolute maximum rating of the BST pin. 6.0 5.5 TO RUN 5.0 4.5 4.0 0.001 0.01 0.1 LOAD CURRENT (A) 1 36941 F05 Figure 5. The Minimum Input Voltage Depends on Output Voltage, Load Current and Boost Circuit Enable and Programmable Undervoltage Lockout The EN/UVLO pin provides both logic enable and programmable undervoltage lockout functions. There are two thresholds on the EN/UVLO pin. The first threshold is at 500mV (typ). When EN/UVLO is below this threshold, the LT3694 is in complete shutdown and the quiescent current drops below 2μA. 36941fa 16 LT3694/LT3694-1 APPLICATIONS INFORMATION Once EN/UVLO climbs above the first threshold, the internal circuitry of the LT3694 is turned on but the switching regulator and LDOs remain shut off. A 2μA current sink on the EN/UVLO pin is activated to provide hysteresis for the programmable undervoltage function. + EN/UVLO INTERNAL CIRCUITRY 0.5V 2μA The second threshold is an accurate 1.2V derived from the internal reference. When EN/UVLO is above the second threshold, the regulators turn on and the 2μA current sink turns off. This allows an accurate programmable UVLO function by placing a resistor divider between VIN, EN/UVLO and ground. Figure 6a shows the EN/UVLO block diagram and Figure 6b shows connections for the programmable UVLO function. The trip level is set by the resistor ratio: – + SHUTDOWN REGULATORS 1.2V – (6a) EN/UVLO Block Diagram VIN UNDERVOLTAGE TRIP LEVEL VIN ⎛ R1+ R2 ⎞ VIN(UVTRIP) = 1.2V ⎜ ⎝ R2 ⎟⎠ R1 1.2V • LT3694 EN/UVLO (R1 + R2) R2 UVLO HYSTERESIS 2μA • R1 R2 36941 FO4 The hysteresis is set by R1: VIN(UVHYS) = 2µA • R1 The EN/UVLO pin may be driven with a logic output if the programmable UVLO is not needed. The requirements for the logic output are a low output voltage less than 0.35V (to insure low current shutdown) and a high output voltage greater than 1.25V. Low Dropout Regulator Each low dropout regulator comprises an error amp, loop compensation and a base drive amp. It uses the same 0.75V reference as the switching regulators. It requires an external NPN pass transistor and 2.2μF of output capacitance for stability. The dropout characteristics will be determined by the pass transistor. The collector-emitter saturation characteristics will limit the dropout voltage. Table 5 lists some suitable NPN transistors with their saturation specifications. (6b) Programmable UVLO Application Figure 6. Programmable UVLO Application The base drive voltage has a maximum voltage of 6V. This will limit the maximum output of the regulator to 6V – VBE(SAT) where VBE(SAT) is the base-emitter saturation voltage of the pass transistor. Table 5. Low VCESAT Transistors PART NUMBER VCESAT at IC = 1A OUTLINE ZXTN25012EZ ZXTN25020DG 0.06 0.075 SOT-89 SOT-223 Zetex www.diodes.com NSS20201JT1G NSS12201LT1G 0.22 0.08 SC-89 SOT-23 ON Semiconductor www.onsemi.com CTLT3410-M621 0.28 MANUFACTURER 1mm × 2mm Central Semiconductor www.central-semi.com TLM621 36941fa 17 LT3694/LT3694-1 APPLICATIONS INFORMATION The LDO may be shut down if it is unused by pulling the FB pin up with a resistor that will source at least 30μA. The FB pin will clamp at about 1.25V and the LDO will shut off reducing power consumption. This pull-up can be sourced from one of the LT3694 outputs provided that channel is always on when the other channels are on. The output stage of the LDO will drive the NPN base from the BIAS voltage if it is at least 1.8V above the LDO DRIVE voltage, otherwise the NPN base current comes from VIN. The base drive current is limited to 15mA. LDO FB Resistor Network The output voltage of the LDO regulator is programmed with a resistor divider (refer to the Block Diagram in Figure 7) between the emitter of the external NPN pass resistor and the feedback pin, FB2 or FB3. Choose the resistors according to: ⎛V ⎞ R1= R2 ⎜ OUT − 1⎟ ⎝ 0.75 ⎠ The parallel combination of R1 and R2 should be 10k or less to avoid bias current errors. OUT1 LT3694 BIAS 60mV RSENSE + LIM2 – DRV2 + OUT2 0.75V R1 – FB2 R2 36941 FO7 Figure 7. LDO with Current Limit LDO Current Limit The LDO has a current limit available to reduce the power consumption of the NPN transistor under overload conditions. The current limit requires the NPN transistor collector to be connected to the BIAS pin through a low resistance sense resistor. The current limit circuit senses the voltage drop across this resistor and reduces the base drive current when the limit voltage exceeds 60mV. This will limit the output current to 60mV/RSENSE. If the overload causes the output voltage to drop, the limit voltage is folded back to reduce power in the NPN transistor. The limit circuit monitors the FB voltage and ramps the limit voltage down once VFB drops to 0.6V. The limit voltage will fold back to 26mV when VFB has dropped to 0V. The current foldback is disabled until the associated TRK/SS pin rises above 0.68V. This insures proper start-up under full load conditions. Figure 7 shows the LDO circuit with current limit. Properly routing the current limit sense resistors is critical to minimize errors in the current limit. The sense connections are the BIAS pin (both channels) on the high side and LIM2 or LIM3 on the bottom side. These sense leads must be routed separately from all current carrying traces. Figure 9 shows a layout that minimizes trace resistance errors. The current limit sense resistors (RLIM2 and RLIM3) are placed close together and the BIAS pin trace is connected to VOUT1 at their junction. The bottom sides of these resistors have a separate via and trace to the LIM2 and LIM3 pins. The foldback can dramatically reduce the power dissipation of the NPN pass transistor under short-circuit conditions. For example, an application that has VOUT1 = 3.3V and VOUT2 = 2.5V will nominally have 0.8V across the pass transistor VCE. Under short-circuit conditions, the pass transistor VCE will increase to 3.3V. Without foldback the power dissipation in the pass transistor will increase by more than 4x, but with foldback the power dissipation only increases by 78%. 36941fa 18 LT3694/LT3694-1 APPLICATIONS INFORMATION If the current feeding the collector of the NPN through the sense resistor comes from a supply that is not connected to BIAS, the current limit cannot be used and the LIM pin must be connected to BIAS to disable the current limit. Tracking and Soft-Start The output of the LT3694 regulates to the lowest voltage present at either the TRK/SS pin or an internal 0.75V reference. A capacitor from the TRK/SS pin to ground is charged by an internal 3μA current source resulting in a linear output ramp from 0V to the regulated output whose duration is given by: t RAMP = CTRKSS • 0.75V 3µA At power-up or at any shutdown event, the TRK/SS pins are internally pulled to ground through 100Ω to insure the soft-start capacitors are discharged. The pins clamp at 1.3V. Ratiometric tracking is achieved by tying the TRK/SS pins tied together and connecting to a single capacitor. The charge current is multiplied by the number of TRK/SS pins connected. Coincident tracking is accomplished by adding an additional resistor divider to the master regulator output and connecting it to the TRK/SS pin of the slave regulator. The resistor divider should be equal to the slave’s feedback divider. Keep in mind that the LDO pass transistor VCE(SAT) will limit how well the LDO output can coincidentally track the switching regulator output. The TRK/SS pin has a low voltage detect that insures the regulator is shut off when TRK/SS is pulled low. The threshold low voltage is nominally 50mV. This allows independent on/off control of the LDOs using the TRK/SS pins. The logic drive should be open collector or have series resistance because the TRK/SS pins are internally pulled to ground during any shutdown event. Shorted and Reversed Input Protection If an inductor is chosen that will not saturate excessively, an LT3694 buck regulator will tolerate a shorted output. There is another situation to consider in systems where the output will be held high when the input to the LT3694 is absent. This may occur in battery charging applications or in battery backup systems where a battery or some other supply is diode ORed with the LT3694’s output. If the VIN pin is allowed to float and the EN/UVLO pin is held high (either by a logic signal or because it is tied to VIN), then the LT3694’s internal circuitry will pull its quiescent current through its SW pin. This is fine if the system can tolerate a few mA in this state. If the EN/UVLO pin is grounded, the SW pin current will drop to essentially zero. However, if the VIN pin is grounded while the output is held high, then parasitic diodes inside the LT3694 can pull large currents from the output through the SW pin and the VIN pin. The circuit in Figure 8 runs only when the input voltage is present—and protects against a shorted or reversed input. D4 VIN VIN BST LT3694 EN/UVLO VOUT SW VC GND FB BACKUP 36941 F08 Figure 8. Diode D4 Prevents a Shorted Input from Discharging a Backup Battery Tied to the Output. It Also Protects the Circuit from a Reversed Input. The LT3694 Runs Only When the Input Is Present 36941fa 19 LT3694/LT3694-1 APPLICATIONS INFORMATION PCB Layout For proper operation and minimum EMI, care must be taken during printed circuit board layout. Figure 9 shows the recommended component placement with trace, ground plane and via locations. Note that large, switched currents flow in the LT3694’s VIN, DA, and SW pins, the catch diode (D1) and the input capacitor (CIN). The loop formed by these components should be as small as possible. These components, along with the inductor and output capacitor, should be placed on the same side of the circuit board, and their connections should be made on that layer. Place a local, unbroken ground plane below these components. The SW and BST nodes should be as small as possible. Finally, keep the FB and VC nodes small so that the ground traces will shield them from the SW and BST nodes. The exposed pad on the bottom of the package must be soldered to ground so that the pad acts as a heat sink. To keep thermal resistance low, extend the top side ground plane as much as possible, and add thermal vias under and near the LT3694 to additional ground planes within the circuit board and on the bottom side. High Temperature Considerations The PCB must provide heat sinking to keep the LT3694 cool. The Exposed Pad on the bottom of the package must be soldered to a ground plane. This ground should be tied to large copper layers below with thermal vias; these layers will spread the heat dissipated by the LT3694. Place additional vias to reduce thermal resistance further. With these steps, the thermal resistance from die (or junction) to ambient can be reduced to θJA = 34°C/W (UFD) or θJA = 38°C/W (FE20). With 100 LFPM airflow, this resistance can fall by another 25%. Further increases in airflow will lead to lower thermal resistance. Because of the large output current capability of the LT3694, it is possible to dissipate enough heat to raise the junction temperature beyond the absolute maximum. When operating at high ambient temperatures, the maximum load current should be derated as the ambient temperature approaches TJ(MAX). Power dissipation within the LT3694 can be estimated by calculating the total power loss from an efficiency measurement and subtracting the catch diode loss and inductor loss. The die temperature is calculated by multiplying the LT3694 power dissipation by the thermal resistance from junction-to-ambient. Keep in mind other heat sources—such as the catch diode, inductor and LDO pass transistors. Other Linear Technology Publications Application Notes 19, 35 and 44 contain more detailed descriptions and design information for buck regulators and other switching regulators. The LT1376 data sheet has a more extensive discussion of output ripple, loop compensation and stability testing. Design Note 318 shows how to generate a bipolar output supply using a buck regulator. 36941fa 20 LT3694/LT3694-1 APPLICATIONS INFORMATION VIN GND L1 VOUT1 COUT1 CIN D1 RLIM2 RLIM3 Q3 Q2 VOUT2 VOUT3 36941 F09 PCB BOTTOM SIDE IS A SOLID GROUND PLANE THERMAL VIAS TO GROUND PLANE VIAS TO LIM2/LIM3 SIGNAL VIAS TO INNER LAYERS VIAS TO Q2 COLLECTOR VIAS TO BIAS Figure 9. A Good PCB Layout Ensures Proper, Low EMI Operation 36941fa 21 LT3694/LT3694-1 TYPICAL APPLICATIONS Automotive Input Range (6V to 16V) to 3.3V, 2.5V, 1.8V VIN 6V TO 16V TRANSIENT TO 70V UVLO 5.8V 4.7μF 100k 26.7k EN/UVLO VIN BIAS BST 0.1μF TRK/SS1 TRK/SS2 1nF FB1 LIM2 OUT2 2.5V 450mA DRV2 34k DA 0.1Ω ZXTN25012EZ OUT1 3.3V 1.7A B340A TRK/SS3 OUT1 1.2μH SW 100pF 34k LT3694 10k VC1 10μF 0.1Ω 2.2μF DRV3 FB2 10.7k OUT1 LIM3 24.9k SYNC ZXTN25012EZ 14k PGOOD OUT3 1.8V 450mA 2.2μF FB3 16k 10k RT GND fSW = 2MHz 36941 TA02 36941fa 22 LT3694/LT3694-1 TYPICAL APPLICATIONS Wide Input Range to (6.3V to 36V) to 5V, 3.3V, 2.5V With Independent On/Off Control of the LDOs VIN 6.3V TO 36V TRANSIENT TO 70V 10μF VIN BIAS EN/UVLO ENABLE 0.22μF 5.4μH TRK/SS1 ENLD02 TRK/SS2 ENLD03 TRK/SS3 1nF 1nF BST B340A 57.6k DA 1nF FB1 10.2k 1000pF 0.1Ω 22μF 20k LIM2 OUT1 OUT2 2.5V 450mA OUT1 5V 1.7A SW ZXTN25020DG LT3694 VC1 DRV2 0.1Ω 2.2μF 24.9k LIM3 OUT1 DRV3 ZXTN25020DG OUT3 3.3V 450mA FB2 SYNC 10.7k PGOOD 34k 2.2μF FB3 66.5k 10k RT SYNC GND CLKIN 36941 TA03 fSW = 800kHz 36941fa 23 LT3694/LT3694-1 TYPICAL APPLICATIONS Wide Input Range (6V to 36V) to 1.8V, 2.5V and 3.3V VIN 6V TO 36V TRANSIENT TO 70V UVLO 5.8V 22μF OUT2 100k 26.7k EN/UVLO VIN BIAS BST 0.22μF TRK/SS1 4.7nF 3.3μH OUT1 1.8V 2.6A SW TRK/SS2 B340A TRK/SS3 14k DA FB1 470pF LT3694 LIM2 ZXTN25020DG 10k 25.5k 47μF VC1 DRV2 LIM3 OUT2 3.3V VIN OUT2 DRV3 2.2μF ZXTN25020DG OUT3 2.5V 34k FB2 10k 90.2k 24.9k PGOOD 2.2μF FB3 10.7k RT SYNC GND fSW = 500kHz 36941 TA04 THE LDO OUTPUT CURRENT CAPABILITY IS LIMITED BY THE POWER DISSIPATION OF THE NPN PASS TRANSISTORS 36941fa 24 LT3694/LT3694-1 PACKAGE DESCRIPTION UFD Package 28-Lead Plastic QFN (4mm × 5mm) (Reference LTC DWG # 05-08-1712 Rev B) 0.70 p0.05 4.50 p 0.05 3.10 p 0.05 2.50 REF 2.65 p 0.05 3.65 p 0.05 PACKAGE OUTLINE 0.25 p0.05 0.50 BSC 3.50 REF 4.10 p 0.05 5.50 p 0.05 RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED 4.00 p 0.10 (2 SIDES) 0.75 p 0.05 R = 0.05 TYP PIN 1 NOTCH R = 0.20 OR 0.35 s 45o CHAMFER 2.50 REF R = 0.115 TYP 27 28 0.40 p 0.10 PIN 1 TOP MARK (NOTE 6) 1 2 5.00 p 0.10 (2 SIDES) 3.50 REF 3.65 p 0.10 2.65 p 0.10 (UFD28) QFN 0506 REV B 0.200 REF 0.00 – 0.05 0.25 p 0.05 0.50 BSC BOTTOM VIEW—EXPOSED PAD NOTE: 1. DRAWING PROPOSED TO BE MADE A JEDEC PACKAGE OUTLINE MO-220 VARIATION (WXXX-X). 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE 36941fa 25 LT3694/LT3694-1 PACKAGE DESCRIPTION FE Package 20-Lead Plastic TSSOP (4.4mm) (Reference LTC DWG # 05-08-1663 Rev H) Exposed Pad Variation CB 6.40 – 6.60* (.252 – .260) 3.86 (.152) 3.86 (.152) 20 1918 17 16 15 14 13 12 11 6.60 ±0.10 2.74 (.108) 4.50 ±0.10 6.40 2.74 (.252) (.108) BSC SEE NOTE 4 0.45 ±0.05 1.05 ±0.10 0.65 BSC 1 2 3 4 5 6 7 8 9 10 RECOMMENDED SOLDER PAD LAYOUT 4.30 – 4.50* (.169 – .177) 0.09 – 0.20 (.0035 – .0079) 0.25 REF 0.50 – 0.75 (.020 – .030) NOTE: 1. CONTROLLING DIMENSION: MILLIMETERS MILLIMETERS 2. DIMENSIONS ARE IN (INCHES) 3. DRAWING NOT TO SCALE 1.20 (.047) MAX 0° – 8° 0.65 (.0256) BSC 0.195 – 0.30 (.0077 – .0118) TYP 0.05 – 0.15 (.002 – .006) FE20 (CB) TSSOP REV H 0910 4. RECOMMENDED MINIMUM PCB METAL SIZE FOR EXPOSED PAD ATTACHMENT *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.150mm (.006") PER SIDE 36941fa 26 LT3694/LT3694-1 REVISION HISTORY REV DATE DESCRIPTION PAGE NUMBER A 01/11 Corrected the Pin Configuration drawing and Package Description for the TSSOP package. 2 36941fa Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 27 LT3694/LT3694-1 TYPICAL APPLICATION 6V to 28V Input Range with Cascaded Step Down — 3.3V, 2.5V and 1.8V Outputs Plus Independently Enabled 1.8V, 1.5V and 1.2V Outputs VIN 6V TO 28V TRANSIENT TO 70V 10μF 200k 52.3k VIN BIAS EN/UVLO BST 0.1μF TRK/SS1 1nF TRK/SS2 B340A TRK/SS3 DA FB1 LT3694-1 30.9k 0.1Ω OUT2 2.5V 450mA 10k ZXTN25012EZ 24.9k FB2 14k CLKOUT 10.7k 10k fSW = 1MHz 40.2k 10μF 10μF GNDA EN4 EN5 EN6 1.5μH 2.2μF PGND VIN PVIN RUN1 PGOOD1 RUN2 PGOOD2 LTC3545 RUN3 SYNC/MODE SW2 SW1 SW3 1.5μH 20pF 226k 226k 1.5μH 20pF 10μF OUT6 1.5V, 800mA 10μF VFB1 511k OUT5 1.2V 800mA VFB2 20pF 2.2μF OUT3 1.8V 200mA FB3 PGOOD RT GND OUT4 1.8V, 800mA 22μF OUT1 LIM3 DRV3 2.2μF 270pF 0.2Ω DRV2 ZXTN25012EZ OUT1 3.3V 500mA 34k VC1 LIM2 OUT1 2.2μH SW GNDA PGND 255k VFB3 200k 301k 36941 TA05 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT3480 36V with Transient Protection to 60V, 2A (IOUT), 2.4MHz, High Efficiency Step-Down DC/DC Converter with Burst Mode® Operation VIN: 3.6V to 38V, VOUT(MIN) = 0.78V, IQ = 70μA, ISD < 1μA, 3mm × 3mm DFN-10 and MSOP-10E Packages LT3500 36V, 40VMAX, 2A, 2.5MHz High Efficiency Step-Down DC/DC Converter and LDO Controller VIN: 3.6V to 36V, VOUT(MIN) = 0.8V, IQ = 2.5mA, ISD < 10μA, 3mm × 3mm DFN-10 Package LT3507 36V, 2.5MHz, Triple (2.4A + 1.5A + 1.5A (IOUT)) with LDO Controller High Efficiency Step-Down DC/DC Converter VIN: 4V to 36V, VOUT(MIN) = 0.8V, IQ = 7mA, ISD < 1μA, 5mm × 7mm QFN-38 Package LT3685 36V with Transient Protection to 60V, 2A (IOUT), 2.4MHz, High Efficiency Step-Down DC/DC Converter VIN: 3.6V to 38V, VOUT(MIN) = 0.78V, IQ = 70μA, ISD < 1μA, 3mm × 3mm DFN-10 and MSOP-10E Packages LT3970 40V, 350mA, 2MHz High Efficiency Micropower Step-Down DC/DC Converter VIN: 4V to 40V, Transient to 60V, VOUT(MIN) = 1.21V, IQ = 2μA, ISD < 1μA, 3mm × 2mm DFN-10 and MSOP-10 Packages LT3980 58V with Transient Protection to 80V, 2A (IOUT), 2.4MHz, High Efficiency Step-Down DC/DC Converter with Burst Mode Operation VIN: 3.6V to 58V, Transient to 80V, VOUT(MIN) = 0.8V, IQ = 85μA, ISD < 1μA, 3mm × 4mm DFN-16 and MSOP-16E Packages 36941fa 28 Linear Technology Corporation LT 0111 REV A • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2010