TRINAMIC TMC603_1

TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
1
TMC603 – DATASHEET
Three phase motor driver with BLDC back EMF
commutation hallFX™ and current sensing
®
TRINAMIC Motion Control GmbH & Co. KG
Sternstraße 67
D – 20357 Hamburg
GERMANY
www.trinamic.com
1 Features
The TMC603 is a three phase motor driver for highly compact and energy efficient drive solutions. It
contains all power and analog circuitry required for a high performance BLDC motor system. The
TMC603 is designed to provide the frontend for a microcontroller doing motor commutation and control
algorithms. It directly drives 6 external N-channel MOSFETs for motor currents up to 30A and up to
50V and integrates shunt less current measurement, by using the MOSFETs channel resistance for
sensing. Integrated hallFX (pat. fil.) allows for sensorless commutation. Protection and diagnostic
features as well as a step down switching regulator further reduce system cost and increase reliability.
Highlights
Up to 30A motor current
9V to 50V operating voltage
3.3V or 5V interface
8mm x 8mm QFN package
Integrated shunt less current measurement using power MOS transistor RDSon
hallFX™ sensorless back EMF commutation emulates hall sensors
Integrated break-before-make logic: No special microcontroller PWM hardware required
EMV optimized current controlled gate drivers – up to 150mA possible
Overcurrent / short to GND and undervoltage protection and diagnostics integrated
Internal QGD protection: Supports latest generation of power MOSFETs
Integrated supply concept: Step down switching regulator up to 500mA / 300kHz
Common rail charge pump allows for 100% PWM duty cycle
Applications
Motor driver for industrial applications
Integrated miniaturized drives
Robotics
High-reliability drives (dual position sensor possible)
Pump and blower applications with sensorless commutation
Motor type
3 phase BLDC, stepper, DC motor
Sine or block commutation
Rotor position feedback: Sensorless, encoder or hall sensor, or any mix
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
Life support policy
TRINAMIC Motion Control GmbH & Co. KG does
not authorize or warrant any of its products for use in
life support systems, without the specific written
consent of TRINAMIC Motion Control GmbH & Co.
KG.
Life support systems are equipment intended to
support or sustain life, and whose failure to perform,
when properly used in accordance with instructions
provided, can be reasonably expected to result in
personal injury or death.
© TRINAMIC Motion Control GmbH & Co. KG 2008
Information given in this data sheet is believed to be
accurate and reliable. However no responsibility is
assumed for the consequences of its use nor for any
infringement of patents or other rights of third parties
which may result from its use.
Specifications subject to change without notice
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
2
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
3
2 Table of contents
1
FEATURES .......................................................................................................................................... 1
2
TABLE OF CONTENTS ........................................................................................................................ 3
3
SYSTEM ARCHITECTURE USING THE TMC603 ................................................................................... 5
4
PINOUT ............................................................................................................................................. 6
4.1
4.2
5
PACKAGE CODES ............................................................................................................................. 6
PACKAGE DIMENSIONS QFN52........................................................................................................... 7
TMC603 FUNCTIONAL BLOCKS .......................................................................................................... 8
5.1
BLOCK DIAGRAM AND PIN DESCRIPTION................................................................................................ 8
5.2
MOSFET DRIVER STAGE ................................................................................................................ 10
5.2.1
Principle of operation ....................................................................................................... 10
5.2.2
Break-before-make logic ................................................................................................... 11
5.2.3
PWM control via microcontroller ...................................................................................... 12
5.2.4
Slope control .................................................................................................................... 13
5.2.5
Reverse capacity (QGD) protection .................................................................................... 14
5.2.6
Considerations for QGD protection ................................................................................... 15
5.2.7
Effects of the MOSFET bulk diode ..................................................................................... 16
5.2.8
Adding Schottky diodes across the MOSFET bulk diodes ................................................. 16
5.2.9
Short to GND detection .................................................................................................... 17
5.2.10
Error logic ......................................................................................................................... 17
5.3
CURRENT MEASUREMENT AMPLIFIERS .................................................................................................. 19
5.3.1
Current measurement timing............................................................................................ 19
5.3.2
Auto zero cycle ................................................................................................................. 20
5.3.3
Measurement depending on chopper cycle ...................................................................... 20
5.3.4
Compensating for offset voltages .................................................................................... 20
5.3.5
Getting a precise current value ........................................................................................ 20
5.4
HALLFX SENSORLESS COMMUTATION .................................................................................................. 21
5.4.1
Adjusting the hallFX spike suppression time ................................................................... 21
5.4.2
Adjusting the hallFX filter frequency ................................................................................ 22
5.4.3
Block commutation chopper scheme for hallFX ............................................................... 22
5.4.4
Start-up sequence for the motor with forced commutation ............................................ 23
5.5
POWER SUPPLY ............................................................................................................................. 25
5.5.1
Switching regulator and charge pump............................................................................. 25
5.5.2
Charge pump .................................................................................................................... 27
5.5.3
Supply voltage filtering .................................................................................................... 27
5.5.4
Reverse polarity protection ............................................................................................... 27
5.5.5
Standby with zero power consumption ........................................................................... 27
5.5.6
Low voltage operation down to 9V ................................................................................. 28
5.6
TEST OUTPUT................................................................................................................................ 29
6
ABSOLUTE MAXIMUM RATINGS ..................................................................................................... 30
7
ELECTRICAL CHARACTERISTICS ...................................................................................................... 30
7.1
7.2
8
OPERATIONAL RANGE ..................................................................................................................... 30
DC CHARACTERISTICS AND TIMING CHARACTERISTICS ........................................................................... 31
DESIGNING THE APPLICATION ...................................................................................................... 38
8.1
CHOOSING THE BEST FITTING POWER MOSFET .................................................................................... 38
8.1.1
Calculating the MOSFET power dissipation ...................................................................... 38
8.2
MOSFET EXAMPLES....................................................................................................................... 40
8.3
DRIVING A DC MOTOR WITH THE TMC603 ......................................................................................... 40
9
TABLE OF FIGURES ......................................................................................................................... 41
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
10
10.1
4
REVISION HISTORY .................................................................................................................... 41
DOCUMENTATION REVISION ......................................................................................................... 41
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
5
3 System architecture using the TMC603
POWER
TMC603
slope
control
BUS / IO
slope HS
12V step
down
regulator
slope LS
1 of 3 shown
5V
linear
regulator
+VM
DRIVER
SECTION
HS
HS-drive
N
S
micro
controller
break
before
make
logic
gate off detection
BLDC motor
LS
LS-drive
position sensor
bridge current
measurement
short to
GND
detection
HallFXTM for
sensorless
commutation
short to GND 1,2,3
NFET power MOS half bridges
error logic
figure 1: application block diagram
The TMC603 is a BLDC driver IC using external power MOS transistors. Its unique feature set allows
equipping inexpensive and small drive systems with a maximum of intelligence, protection and
diagnostic features. Control algorithms previously only found in much more complex servo drives can
now be realized with a minimum of external components. Depending on the desired commutation
scheme and the bus interface requirements, the TMC603 forms a complete motor driver system in
combination with an external 8 bit processor or with a more powerful 32 bit processor. A simple
system can work with three standard PWM outputs even for sine commutation! The complete analog
amplification and filtering frontend as well as the power driver controller are realized in the TMC603.
Its integrated support for sine commutation as well as for back EMF sensing saves cost and allows for
maximum drive efficiency.
The external microcontroller realizes commutation and control algorithms. Based on the position
information from an encoder or hall sensors, the microcontroller can do block commutation or sine
commutation with or without space vector modulation and realizes control algorithms like a PID
regulator for velocity or position or field oriented control based on the current signals from the
TMC603. For sensorless commutation, the microcontroller needs to do a forward controlled motor start
without feedback. This can be realized either using block commutation or sine commutation. A sine
commutated start-up minimizes motor vibrations during start up. As soon as the minimum velocity for
hallFX is reached, it can switch to block commutation and drive the motor based on the hallFX signals.
The TMC603 also supports control of three phase stepper motors as well as two phase stepper
motors using two devices.
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
6
GNDP
HS1
BM1
LS1
VCP
HS2
BM2
LS2
VLS
HS3
BM3
LS3
GNDP
52
51
50
49
48
47
46
45
44
43
42
41
40
4 Pinout
VLS
1
39
VCP
GNDP
2
38
TEST
VM
3
37
SWOUT
GND
4
36
GND
RS2G
5
35
RSLP
H1
6
34
CLR_ERR
H2
7
33
/ERR_OUT
H3
8
32
ENABLE
FILT1
9
31
INV_BL
FILT2
10
30
BBM_EN
FILT3
11
29
SENSE_HI
COSC
12
28
VCC
SCCLK
13
27
SP_SUP
14
15
16
17
18
19
20
21
22
23
24
25
26
BH1
BL1
SAMPLE1
CUR1
BH2
BL2
SAMPLE2
CUR2
BH3
BL3
SAMPLE3
CUR3
5VOUT
TMC 603-LA
QFN52 8mm x 8mm
0.5 pitch
figure 2: pinning / QFN52 package (top view)
4.1
Package codes
Type
TMC603
Package
QFN52 (ROHS)
Temperature range
-40°C ... +125°C
Code/marking
TMC603-LA
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
4.2
Package dimensions QFN52
REF
MIN
NOM
MAX
A
0.80
0.85
0.90
A1
0.00
0.035
0.05
A2
-
0.65
0.67
A3
b
0.203
0.2
0.25
D
8.0
E
8.0
e
0.5
0.3
J
6.1
6.2
6.3
K
6.1
6.2
6.3
L
0.35
0.4
0.45
All dimensions are in mm.
Attention: Drawing not to scale.
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
7
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
8
5 TMC603 functional blocks
5.1
Block diagram and pin description
+VM
220n
220n
100n
(2x)
BAV99 (70V)
BAS40-04W (40V)
4µ7
Tantal 25V
16V
LSW
SS16
TP0610K
or BSS84
(opt. BC857)
12V supply
100µ
100n
(2x)
(150mA with
sel. transistor)
VM+10V
charge pump
LSW: 220µH for 100kHz
VM
COSC
RSLP
ENABLE
BH1
BL1
12V step
down
regulator
slope HS
slope
control
RSLP: 100k -> 100mA
BBM_EN
VLS
VCP
TMC603
COSC: 470p ->100kHz
INV_BL
SWOUT
5V
linear
regulator
5VOUT
VCC
slope LS
100nF
+VM
1 of 3 shown
DRIVER
SECTION
D
D
VCP
HS1
HS-drive
D
break
before
make
logic
D
D
Zener 12V
BZT52B12-V/
BZV55C12
BM1
220R
Gate off detection
BM1
short to
GND
detection
LS1
LS-drive
short to GND 1
GNDP
RS2G: 470k -> 2000ns
SENSE_HI
amplification
4.5x or 18x
D
BM1
RDS
current
sense LS
VCC
1 of 3 power MOS half
bridges
track & hold stage
signed current,
centered at 1/3 VCC
A
SAMPLE1
automatic
sample
point delay
D
D
hall sensor
emulation
BM2
D
D
BM3 switched capacitor filter
A
D
A
A
SENSORLESS
COMMUTATION
test logic
CLR_ERR
CUR1
BRIDGE
CURRENT
MEASUREMENT
BM1
SCCLK
Motor coil
output
opt. for high
QGD FETs :
MSS1P3 /
ZHCS1000
VLS
RS2G
Provide sufficient filtering
capacity near bridge
5V supply transistors (electrolyt
capacitors and ceramic
capacitors)
undervoltage VLS, VCP
short to GND 1,2,3
FILT1
FILT2
FILT3
/ERR_OUT
error logic
set
reset
D
RS2G, RSLP and BMx: Use short
trace and avoid stray capacitance
to switching signals. Place
resistors near pin.
D
H1
H2
H3
TEST
SP_SUP
CSUP: 1n -> 90µs
GND
DIE PAD
GND
figure 3: application diagram
The application diagram shows the basic building blocks of the IC and the connections to the power
bridge transistors, as well as the power supply. The connection of the digital and analog I/O lines to
the microcontroller is highly specific to the microcontroller model used.
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
Pin
Number
Type
Function
VLS
1, 44
Low side driver supply voltage for driving low side gates
GNDP
2, 40, 52
Power GND for MOSFET drivers, connect directly to GND
VM
3
Motor and MOSFET bridge supply voltage
GND
4, 36
Digital and analog low power GND
RS2G
5
AI 5V
Short to GND control resistor. Controls delay time for short to GND
test
Hx
6, 7, 8
DO 5V
hallFX outputs for back EMF based hall sensor emulation
FILTx
9, 10, 11
AO 5V
Output of internal switched capacitor filter
COSC
12
A 5V
Oscillator capacitor for step down regulator
SCCLK
13
DI 5V
Switched capacitor filter clock input for hallFX filters.
BHx
14, 18,
22
DI 5V
High side driver control signal: A positive level switches on the high
side
BLx
15, 19,
23
DI 5V
Low side driver control signal: Polarity can be reversed via INV_BL
SAMPLEx
16, 20,
24
DI 5V
Optional external control for current measurement sample/hold
stage. Set to positive level, if unused
CURx
17, 21,
25
AO 5V
Output of current measurement amplifier
5VOUT
26
SP_SUP
27
VCC
28
Output of internal 5V linear regulator. Provided for VCC supply
A 5V
An external capacitor on this pin controls the commutation spike
suppression time for hallFX.
+5V supply input for digital I/Os and analog circuitry
SENSE_HI 29
DI 5V
Switches current amplifiers to high sensitivity
BBM_EN
30
DI 5V
Enables internal break-before-make circuitry
INV_BL
31
DI 5V
Allows inversion of BLx input active level (low: BLx is active high)
ENABLE
32
DI 5V
Enables the power drivers (low: all MOSFETs become actively
switched off)
/ERR_OUT 33
DO 5V
Error output (open drain). Signals undervoltage or overcurrent. Tie
to ENABLE for direct self protection of the driver
CLR_ERR
34
DI 5V
Reset of error flip-flop (active high). Clears error condition
RSLP
35
AI 5V
Slope control resistor. Sets output current for MOSFET drivers
SWOUT
37
O
Switch regulator transistor output
TEST
38
O 12V
Test multiplexer output (leave open)
VCP
39
LSx
41, 45,
49
O 12V
Low side MOSFET driver output
BMx
42, 46,
50
I (VM)
Sensing input for bridge outputs. Used for MOSFET control and
current measurement.
HSx
43, 47,
51
O
(VCP)
High side MOSFET driver output
Exposed
die pad
-
GND
Connect the exposed die pad to a GND plane. It is used for cooling
of the IC and may either be left open or be connected to GND.
Charge pump supply voltage. Provides high side driver supply
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
9
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
5.2
10
MOSFET Driver Stage
The TMC603 provides three half bridge
drivers, each capable of driving two MOSFET
transistors, one for the high-side and one for
the low-side. In order to provide a low RDSon,
the MOSFET gate driving voltage is about 10V
to 12V.
VLS
VCP
TMC603
HS-DRV
+VM
HS1
Z 12V
BM1
220R
LS-DRV
LS1
GNDP
The TMC603 bridge drivers provide a number
of unique features for simple operation,
explained in the following chapters:
An integrated automatic break-beforemake logic safely switches off one
transistor before its counterpart can be
switched on.
Slope
controlled
operation
allows
adaptation of the driver strength to the
desired slope and to the chosen
transistors.
The drivers protect the bridge actively
against cross conduction (QGD protection)
The bridge is protected against a short to
GND
+VM
HS-DRV
HS2
Z 12V
3 phase
BLDC
motor
BM2
220R
LS-DRV
LS2
+VM
HS-DRV
HS3
Z 12V
BM3
220R
LS-DRV
LS3
figure 4: three phase BLDC driver
5.2.1 Principle of operation
The low side gate driver voltage is supplied by the VLS pins. The low side driver supplies 0V to the
MOSFET gate to close the MOSFET, and VLS to open it.
The TMC603 uses a patented driver principle for driving of the high side:
The high-side MOSFET gate voltage is referenced to its source at the center of the half bridge. Due to
this, the TMC603 references the gate drive to the bridge center (BM) and has to be able to drive it to a
voltage lying above the positive bridge power supply voltage VM. This is realized by a charge pump
voltage generated from the switching regulator via a Villard circuit. When closing the high-side
MOSFET, the high-side driver drives it down to the actual BM potential, since an external induction
current from the motor coil could force the output to stay at high potential. This is accomplished by a
feedback loop and transistor TG1 (see figure). In order to avoid floating of the output BM, a low current
is still fed into the HS output via transistor TG1a. The input BM helps the high side driver to track the
bridge voltage. Since input pins of the TMC603 must not go below -0.7V, the input BM needs to be
protected by an external resistor. The resistor limits the current into BM to a level, the ESD protection
input diodes can accept.
High side driver
VCP
+VM
Ion
HS On
HS
T1
Z 12V
BM
one coil
of motor
220R
HS Off
LS
TG1
TG1a
Ioff
Iholdoff
one NMOS
halfbridge
figure 5: principle of high-side driver (pat.fil.)
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
11
A zener diode at the gate (range 12V to 15V) protects the high-side MOSFET in case of a short to
GND event: Should the bridge be shorted, the gate driver output is forced to stay at a maximum of the
zener voltage above the source of the transistor. Further it prevents the gate voltage from dropping
below source level.
The maximum permissible MOSFET driver current depends on the motor supply voltage:
Parameter
Symbol
Max
Unit
MOSFET driver current with VVM < 30V
IHSX, ILSX
150
mA
150-2.5*(VVM-30V)
mA
100
mA
MOSFET driver current with 30V < VVM < 50V
MOSFET driver current with VVM = 50V
IHSX, ILSX
Pin
Comments
LSx
Low side MOSFET driver output. The driver current is set by resistor RSLP
HSx
High side MOSFET driver output. The driver current is set by resistor RSLP
BMx
Bridge center used for current sensing and for control of the high side driver.
For unused bridges, connect BMx pin to GND and leave the driver outputs
unconnected. Place the external protection resistor near the IC pin.
RSLP
The resistor connected to this pin controls the MOSFET gate driver current. A 40µA
current out of this pin (resistor value of 100k to GND) results in the nominal
maximum current at full supply range. Keep interconnection between IC and resistor
short, to avoid stray capacitance to adjacent signal traces of modulating the set
current.
Resistor range: 60 k
to 500 k
VLS
Low side driver supply voltage for driving low side gates
VCP
Charge pump supply voltage. Provides high side driver supply
GNDP
Power GND for MOSFET drivers, connect directly to GND
BHx
High side driver control signal: A positive level switches on the high side.
For unused bridges, tie to GND.
BLx
Low side driver control signal: Polarity can be reversed via INV_BL
INV_BL
Allows inversion of BLx input active level (low: BLx is active high).
When high, each BLx and BHx can be connected in parallel in order to use only 3
PWM outputs for bridge control. Be sure to switch on internal break-before-make logic
(BBM_EN = Vcc) to avoid bridge short circuits in this case.
5.2.2 Break-before-make logic
Each half-bridge has to be protected against cross conduction during switching events. When
switching off the low-side MOSFET, its gate first needs to be discharged, before the high side
MOSFET is allowed to be switched on. The same goes when switching off the high-side MOSFET and
switching on the low-side MOSFET. The time for charging and discharging of the MOSFET gates
depends on the MOSFET gate charge and the driver current set by RSLP. When the BBM logic is
enabled, the TMC603 measures the gate voltage and automatically delays switching on of the
opposite bridge transistor, until its counterpart is discharged. The BBM logic also prevents
unintentional bridge short circuits, in case both, LSx and HSx, become switched on. The first active
signal has priority.
Alternatively, the required time can be calculated and pre-compensated in the PWM block of the
microcontroller driving the TMC603 (external BBM control).
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
12
Control
signals
Internal BBM control
BLx
0V
BHx
0V
External BBM control
VVLS
Miller plateau
LSx
MOSFET drivers
0V
VVM
BMx
0V
VVCP
VVM
HSx
0V
HSxBMx
Miller plateau
VVCP VVM
0V
tLSON
tLSOFF
tBBMHL
tHSON
tBBMLH
Load pulling BMx down
Load pulling BMx up to +VM
figure 6: bridge driver timing
Pin
Comments
BBM_EN
Enables internal break-before-make circuitry (high = enable)
5.2.3 PWM control via microcontroller
There is a number of different microcontrollers available, which provide specific BLDC commutation
units. However, the TMC603 is designed in a way in order to allow BLDC control via standard
microcontrollers, which have only a limited number of (free) PWM units. The following figure shows
several possibilities to control the BLDC motor with different types of microcontrollers, and shall help to
optimally adapt the TMC603 control interface to the features of your microcontroller. The hall signals
and further signals, like CURx interconnection to an ADC input, are not shown.
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
Microcontroller
with BLDC PWM
unit
PWM1
OUT1
PWM1
OUT2
BH1
TMC603
BL1
PWM1
OUT
BH1
13
Block (Hall or hallFX) or
sine commutated BLDC
motor
TMC603
BL1
Microcontroller with
3 PWM outputs
Sine commutated BLDC
motor
INV_BL
+VCC
BBM_EN
Microcontroller
with 3 PWM
outputs
PWM1
OUT
DIG
OUT
BH1
TMC603
BL1
PWM1
OUT
DIG OUT /
HI-Z
BH1
2k2
Microcontroller with
3 PWM outputs
Block (Hall) commutated
BLDC motor
TMC603
BL1
Block (Hall) or sine
commutated BLDC
motor
INV_BL
+VCC
BBM_EN
Microcontroller
with 1 PWM output
DIG OUT
BH1
DIG OUT
PWM1
OUT
BL1
TMC603
ENABLE
Block (hallFX)
commutated BLDC
motor
2k2
DIG IN
/ERR_OUT
+VCC
BBM_EN
figure 7: examples for microcontroller PWM control
5.2.4 Slope control
The TMC603 driver stage provides a constant current output stage slope control. This allows to adapt
driver strength to the drive requirements of the power MOSFET and to adjust the output slope by
providing for a controlled gate charge and discharge. A slower slope causes less electromagnetic
emission, but at the same time power dissipation of the power transistors rises. The duration of the
complete switching event depends on the total gate charge. The voltage transition of the output takes
place during the so called miller plateau (see figure 6). The miller plateau results from the gate to drain
capacity of the MOSFET charging / discharging during the switching. From the datasheet of the
transistor (see example in figure 8) it can be seen, that the miller plateau typically covers only a part
(e.g. one quarter) of the complete charging event. The gate voltage level, where the miller plateau
starts, depends on the gate threshold voltage of the transistor and on the actual load current.
10
25
VM
8
20
6
15
4
10
2
5
0
0
5
10
15
20
VDS – Drain to source voltage (V)
VGS – Gate to source voltage (V)
MOSFET gate charge vs. switching event
0
25
QMILLER
QG – Total gate charge (nC)
figure 8: MOSFET gate charge as available in device data sheet vs. switching event
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
14
The slope time tSLOPE can be calculated as follows:
Whereas QMILLER is the charge the power transistor needs for the switching event, and IGATE is the
driver current setting of the TMC603.
Taking into account, that a slow switching event means high power dissipation during switching, and,
on the other side a fast switching event can cause EMV problems, the desired slope will be in some
ratio to the switching (chopper) frequency of the system. The chopper frequency is typically slightly
outside the audible range, i.e. 18 kHz to 40 kHz. The lower limit for the slope is dictated by the reverse
recovery time of the MOSFET internal diodes, unless additional Schottky diodes are used in parallel to
the MOSFETs source-drain diode. Thus, for most applications a switching time between 100ns and
750ns is chosen.
The required slope control resistor RSLP can be calculated as follows:
Example:
A circuit using the transistor from the diagram above shall be designed for a slope time of
200ns. The miller charge of the transistor is about 6nC.
The nearest available resistor value is 330 k . It sets the gate driver current to roughly 30 mA.
This is well within the minimum and maximum RSLP resistor limits.
5.2.5 Reverse capacity (QGD) protection
The principle of slope control often is realized by gate series resistors with competitor’s products, but,
as latest MOSFET generations have a fairly high gate-drain charge (QGD), this approach is critical for
safe bridge operation. If the gate is not held in the off state with a low resistance, a sudden raise of the
voltage at the drain (e.g. when switching on the complementary transistor) could cause the gate to be
pulled high via the MOSFETs gate drain capacitance. This would switch on the transistor and lead to a
bridge short circuit.
The TMC603 provides for safe and reliable slope controlled operation by switching on a low resistance
gate protection transistor (see figure).
Vgate
Ion
on
QGD
D
Slope
controlled
G
S
off
QGS
Ioff
full, safe off
TMC603 QGD protected
driver stage
External
MOSFET
figure 9: QGD protected driver stage
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
15
5.2.6 Considerations for QGD protection
This chapter gives the background understanding to ensure a safe operation for MOSFETs with a
gate-drain (Miller) charge QGD substantially larger than the gate-source charge QGS.
In order to guarantee a safe operation of the QGD protection, it is important to spend a few thoughts on
the slope control setting. Please check your transistors’ data sheet for the gate-source charge QGS and
the gate-drain charge QGD (Miller charge). In order to turn on the MOSFET, first the gate-source
charge needs to be charged to the transistor’s gate. Now, the transistor conducts and switching starts.
During the switching event, the additional QGD needs to be charged to the gate in order to complete
the switching event. Wherever QGD is larger than QGS, a switching event of the complementary
MOSFET may force the gate of the switched off MOSFET to a voltage above the gate threshold
voltage. For these MOSFETs the QGD protection ensures a reliable operation, as long as the slopes
are not set too fast.
Calculating the maximum slope setting for high QGD MOSFETs:
Taking into account effects of the MOSFET bulk diode (compare chapter 5.2.7), the maximum slope of
a MOSFET bridge will be around the double slope as calculated from the Miller charge and the gate
current. Based on this, we can estimate the current required to hold the MOSFET safely switched off:
During the bridge switching period, the driver must be able to discharge the difference of Q GD and QGS
while maintaining a gate voltage below the threshold voltage.
Therefore
Thus the minimum value required for IOFFQGD can be calculated:
Where ION is the gate current set via RSLP, and IOFFQGD is the QGD protection gate current.
The low side driver has a lower QGD protection current capability than the high side driver, thus we
need to check the low side. With its RLSOFFQGD of roughly 15 Ohm, the TMC603 can keep the gate
voltage to a level of:
Now we just need to check UGOFF against the MOSFETs output characteristics, to make sure, that no
significant amount of drain current can flow.
Example:
A MOSFET, where QGD is 3 times larger than QGS is driven with 100mA gate current.
The TMC603 thus can keep the gate voltage level to a maximum voltage of
UGOFF = 133mA * 15Ω = 2V
This is sufficient to keep the MOSFET safely off.
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
16
Note:
Do not add gate series resistors to your MOSFETs! This would eliminate the effect of the Q GD
protection. Gate series resistors of a few Ohms only may make sense, when paralleling
multiple MOSFETs in order to avoid parasitic oscillations due to interconnection inductivities.
5.2.7 Effects of the MOSFET bulk diode
Whenever inductive loads are driven, the inductivity will try to sustain current when current becomes
switched off. During bridge switching events, it is important to ensure break-before-make operation,
e.g. one MOSFET becomes switches off, before the opposite MOSFET is switched on. Depending on
the actual direction of the current, this results in a short moment of a few 100 nanoseconds, where the
current flowing through the inductive load forces the bridge output below the lower supply rail or above
the upper supply rail. The respective MOSFET bulk diode in this case takes over the current. The
diode saturates at about -1.2V. But the bulk diode is not an optimum device. It typically has reverse
recovery time of a few ten to several 100ns and a reverse recovery charge in the range of some
100nC or more. Assuming, that the bulk diode of the switching off MOSFET takes over the current, the
complementary MOSFET sees the sum of the coil current and the instantaneous current needed to
recover the bulk diode when trying to switch on. The reverse recovery current may even be higher
than the coil current itself! As a result, a number of very quick oscillations on the output appear,
whenever the bulk diode leaves the reverse recovery area, because up to the half load current
becomes switched off in a short moment. The effect becomes visible as an oscillation due to the
parasitic inductivities of the PCB traces and interconnections. While this is normal, it adds high current
spikes, some amount of dynamic power dissipation and high frequency electromagnetic emission. Due
to its high frequency, the ringing of this current can also be seen on the gate drives and thus can be
easily mistaken as a gate driving problem.
VVM
HS takes over
output current
UBMX
IHS
ILSBULK
0V
-1.2V
IOUT
0A
0A
-IOUT
Phase of switching
event
LS bulk diode
conducting IOUT
HS curr.
rise up to
IOUT
LS bulk
reverse
recovery
overshoot +
ringing
normal slope
switching
complete
HS starts
conducting
figure 10: effect of bulk diode recovery
A further conclusion from this discussion: Do not set the bridge slope time higher than or near to the
reverse recovery time of the MOSFETs, as the parasitic current spikes will multiply the instantaneous
current across the bridge. A plausible time is a factor of three or more for the slope time. If this cannot
be tolerated please see the discussion on adding Schottky diodes.
5.2.8 Adding Schottky diodes across the MOSFET bulk diodes
In order to avoid effects of bulk diode reverse recovery, choose a fast recovery switching MOSFET.
The MOSFET transistors can also be bridged by a Schottky diode, which has a substantially faster
reverse recovery time. This Schottky diode needs to be chosen in a way that it can take over the full
bridge current for a short moment of time only. During this time, the forward voltage needs to be lower
than the MOSFETs forward voltage. A small 5A diode like the SK56 can take over a current of 20A at
a forward voltage of roughly 0.8V.
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
17
+VM
HS1
Z 12V
BM1
Motor
220R
LS1
GNDP
figure 11: parallel Schottky diode avoids current spikes due to bulk diode recovery
5.2.9 Short to GND detection
An overload condition of the high side MOSFET (“short to GND”) is detected by the TMC603, by
monitoring the BM voltage during high side on time. Under normal conditions, the high side power
MOSFET reaches the bridge supply voltage minus a small voltage drop during on time. If the bridge is
overloaded, the voltage cannot rise to the detection level within a limited time, defined by an external
resistor. Upon detection of an error, the error output is activated. By directly tying it to the enable input,
the chip becomes disabled upon detection of a short condition and the error flip flop becomes set.
A variation of the short to GND detection delay allows adaptation to the slope control, and modification
of the sensitivity of the detection during power up.
BHx
0V
VVM
VVM-
BMx
Short
detection
0V
/ERROUT,
ENABLE
Driver off via
ENABLE pin
0V
tS2G
Short to GND
monitor phase
Short to GND
detected
Valid area
VBMS2G
inactive
tS2G
delay
BMx voltage
monitored
inactive
delay
Short detected
figure 12: timing of the short to GND detector
Pin
Comments
RS2G
The resistor connected to this pin controls the delay between switching on the high
side MOSFET and the short to GND check. A 20 µA current out of this pin (resistor
value of 220 k to GND) results in a 500 ns delay, a lower current gives a longer
delay. Disconnecting the pin disables the function. Keep interconnection between IC
and resistor short, to avoid stray capacitance to adjacent signal traces of modulating
the set current.
Resistor range: 47 k
to 1 M
5.2.10 Error logic
The TMC603 has three different sources for signaling an error:
Undervoltage of the low side supply
Undervoltage of the charge pump
Short to GND detector
Upon any of these events the error output is pulled low. After a short to GND detector event, the error
output remains active, until it becomes cleared by the CLR_ERR. By tying the error output to the
enable input, the TMC603 automatically switches off the bridges upon an error. The enable input then
should be driven via an open collector input plus pull-up resistor, or via a resistor.
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
18
Pull-up resistor can be
internal to microcontroller
+VCC
TMC603 error logic
Drive with open drain
output, if feedback is
provided
ENABLE
CLR_ERR
undervoltage VLS
undervoltage VCP
100k
S: priority
D
short to GND 1
short to GND 2
short to GND 3
D
/ERR_OUT
S Q
R Q
D
GND
Feedback connection for automatic self-protection
figure 13: error logic
Pin
Comments
/ERR_OUT
Error output (open drain). Signals undervoltage or overcurrent. Tie to ENABLE for
direct self protection of the driver. The internal error condition generator has a higher
priority than the CLR_ERR input, i.e. the error condition can not be cleared, as long as
it is persistent.
CLR_ERR
Reset of error flip-flop (active high). Clears error condition. The error condition should
at least be cleared once after IC power on.
ENABLE
Enables the power drivers (low: all MOSFETs become actively switched off)
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
5.3
19
Current measurement amplifiers
The TMC603 amplifies the voltage drop in the three lower MOSFET transistors in order to allow
current measurement without the requirement for current sense (shunt) resistors. This saves cost and
board space, as well as the additional power dissipation in the shunt resistors. However, additional
shunt resistors can be added, e.g. source resistors for each lower MOSFET or a common shunt
resistor in the bridge foot point, in order to increase voltage drop and to have a more exact
measurement.
The TMC603 CURx outputs deliver a signal centered to 1/3 of the 5V VCC supply. This allows
measurement of both, negative and positive signals, while staying compatible to a 3.3V
microcontroller. The current amplifier is an inverting type.
+VCC
R
SENSE_HI
D
amplify
4.5x or 18x
R
1/3 VCC
SWC
BMx
R
A
track & hold stage
A
autozero
BLx
SAMPLEx
D
automatic
sample
point delay
CURx
add 1/3 VCC
offset
D
figure 14: schematic of current measurement amplifiers
Pin
Comments
CURx
Output of current measurement amplifier. The output signal is centered to 1/3 VCC.
SENSE_HI
Switches current amplifiers to high sensitivity (high level). Control by processor to get
best sensitivity and resolution for measurement.
SAMPLEx
Optional external control for current measurement sample/hold stage. Set to positive
level, if unused
The voltage drop over the MOSFET is calculated as follows:
whereas x is the ADC output value, x0 is the ADC output value at zero current (e.g. 85 for an 8 bit ADC
with 5V reference voltage), ADCMAX is the range of the ADC (e.g. 256 for an 8 bit ADC), VADCREF is the
reference voltage of the ADC and ACUR is the currently selected amplification (absolute value) of the
TMC603.
With this, the motor current can be calculated using the on resistance RDSON (at 10V) of the MOSFET:
5.3.1 Current measurement timing
Current measurement is self-timed, in order to only provide valid output voltages. Sampling is active
during the low side ON time. The sampling is delayed by an internal time delay, in order to avoid
sampling of instable values during settling time of the bridge current and amplifiers. Thus, a minimum
ON time is required in order to get a current measurement. The output CURx reflects the current
during the measurement. The last value is held in a track and hold circuit as soon as the low side
transistor switches off.
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
20
External
control
Current
sense
out
Bridge
voltage
drop
Control
signals
Internal sample control
SAMPLEx
0V
BLx
0V
BMx
VVM
0.25V
0V
-0.25V
CURx
VVCC/3
0V
tBLHICURX
Phase
tBLHICURX
Hold
(undef.)
CURx tracking -BMx
Hold
CURx tracking
-BMx
Hold
figure 15: timing of the current measurement
The SAMPLEx pins will normally be unused and can be tied to VCC. For advanced applications,
where a precise setting of the current sampling point is desired, e.g. centered to the on-time,
SAMPLEx pins can be deactivated at the desired point of time, enabling the hold stage.
5.3.2 Auto zero cycle
The current measurement amplifiers do an automatic zero cycle during the OFF time of the low side
MOSFETs. The zero offset is stored in internal capacitors. This requires switching off the low side at
least once, before the first measurement is possible, and on a cyclic basis, to avoid drifting away of the
zero reference. This normally is satisfied by the chopper cycle. If commutation becomes stopped, e.g.
due to motor stand still, the respective phase current measurement could drift away. After the first
switching off and on of the low side, the measurement becomes valid again.
5.3.3 Measurement depending on chopper cycle
If the low side on-time on one phase tBLHICURX is too low, a current measurement is not possible. The
TMC603 automatically does not sample the current if the minimum low side-on time is not met. This
condition can arise in normal operation, e.g. due to the commutation angle defined by a sine
commutation chopper scheme. The respective CURx output then does not reflect the phase current.
Thus, the CURx output of a phase should be ignored, if the on-time falls below the minimum low side
on-time for current measurement (please refer to maximum limit). The correct current value can easily
be calculated from the difference of the remaining two current measurements.
5.3.4 Compensating for offset voltages
In order to measure low current values precisely, the “zero” value (x0) of 1/3 VCC can be measured via
the ADC, rather than being hard coded into the measurement software. This is possible by doing a first
current measurement during motor stand-still, with no current flowing in the motor coils, e.g. during a
test phase of the unit. The resulting value can be stored and used as zero reference. However, the
influence of offset voltages can be minimized, by using the high sensitivity setting of the amplifiers for
low currents, and switching to low sensitivity for higher currents.
5.3.5 Getting a precise current value
The ON-resistance of a MOSFET has a temperature co-efficient, which should not be ignored. Thus,
the temperature of the MOSFETs must be measured, e.g. using an NTC resistor, in order to
compensate for the variation. Also, the initial RDSon depends upon fabrication tolerance of the
MOSFETs. If exact measurement is desired, an adjustment should be done during initial testing of
each product. For applications, where an adjustment is not possible, the use of at least one additional
shunt resistor in the common ground line of all three half bridges provides a stable current
measurement base, which allows in-system adjustment of the relative MOSFET resistances. Further,
the TMC603 measurement amplification is slightly different for positive and negative currents.
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
5.4
21
hallFX sensorless commutation
hallFX provides emulated hall sensor signals. The emulated hall sensor signals are available without a
phase shift and there is no error-prone PLL necessary, like with many other systems, nor is the
knowledge of special motor parameters required. Since it is based on the motors’ back-EMF, a
minimum motor velocity is required to get a valid signal. Therefore, the motor needs to be started
without feedback, until the velocity is high enough to generate a reliable hallFX signal.
Position signal
generation (PSG)
Switch Cap filters
Induction pulse
supressor (IPS)
U
Low pass LPU
ULP
BM1
E1
D
H1
V
W
30k
Low pass LPV
VLP
BM2
E1
E2
PSG
IPS
D
H2
E2
30k
E3
Low pass LPW
WLP
BM3
H1
E3
D
H3
H2
H3
30k
A
D
A
A
A
SP_SUP
FILT3
FILT2
FILT1
SC_CLK
CSUP
figure 16: hallFX block diagram and timing
A switched capacitor filter for each coil supplies the measured effective coil voltages. Its filter
frequency can be adapted to the chopper frequency and the desired maximum motor velocity. An
induction pulse suppressor unit gates the commutation spikes which result from the inductive behavior
of the motor coils after switching off the current. The gating time can be adapted by an external
capacitor to fit the motor inductivity and its (maximum) velocity.
Pin
Comments
SP_SUP
A capacitor attached to this pin sets the spike suppression time. This pin charges the
capacitor via an internal current source. If more exact timing is required, an external
47k pull-up resistor to VCC can be used in parallel to the internal current source. The
capacitor becomes discharged upon each valid commutation. The capacitor can
optionally be left away, and the suppression can be done in software.
FILTx
These pins provide the filtered coil voltages. For most applications this will be of no
use, except when an external back-EMF commutation is realized, e.g. using a
microcontroller with ADC inputs. Because of the high output resistance and low
current capability of these pins, it is advised to add an external capacitor of a few
hundred picofarad up to a few nanofarad to GND, if the signals are to be used. This
prevents noise caused by capacitance to adjacent signal traces to disturb the signal.
Hx
Emulated hall sensor output signal of hallFX block.
SCCLK
An external clock controls the corner frequency of the switched capacitor filter. A 1.25
MHz clock gives a filter bandwidth of 3kHz. A lower clock frequency may be better for
lower motor velocities.
5.4.1 Adjusting the hallFX spike suppression time
hallFX needs two minimum motor- and application-specific adjustments: The switched capacitor clock
frequency and the spike suppression time should be adapted. Both can easily be deducted from basic
application parameters and are not very critical. The SCCLK frequency should be matched to the
chopper frequency of the system and the maximum motor velocity. The spike suppression time needs
to be adapted to the desired maximum motor velocity.
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
22
Calculating the commutation frequency fCOM of the motor:
SRPM is the rotation velocity in RPM
nPOLE is the pole count of the actual motor, or the double of the number of pole pairs
The spike suppression time can be chosen as high, as the commutation frequency required for
maximum motor velocity allows. As a thumb rule, we take half of this time to have enough spare.
Example:
Given a 4 pole motor operating at 4000 RPM:
CSUP = 6.25nF. The nearest value is 6.8nF.
5.4.2 Adjusting the hallFX filter frequency
The filter block needs to separate the motors’ back EMF from the chopper pulses. Thus, the target is,
to filter away as much commutation noise as possible, while maintaining as much of the back EMF
signal as possible. Therefore, we need to find a cut-off frequency in between the chopper frequency
and the electrical frequency of the motor. Since we do not want to change the frequency within the
application, we use the nominal or maximum motor velocity to calculate its electrical frequency. The
chopper frequency is given by the system, typically about 20 kHz.
The electrical frequency of the motor is:
Since the filter has a logarithmic behavior, as a thumb rule we can make a logarithmic mean-value as
follows:
With the cut-off frequency being about 1/390 of the switched capacitor clock frequency fSCCLK the
following results as a thumb rule:
The result shall be checked against minimum limit of 250 kHz and maximum limit of 4 MHz, however,
the actual frequency is quite uncritical and can be varied in a wide range.
Example:
Given a 4 pole motor operating at 4000 RPM with a 20 kHz chopper frequency:
fEL = 133 Hz
fCUTOFF = 1.6 kHz
fSCCLK = 0.64 MHz
The result is well within the limits, however, the frequency in a practical application can be
chosen between 300 kHz and 1.5 MHz.
5.4.3 Block commutation chopper scheme for hallFX
hallFX works perfectly with nearly every motor. You can use a standard block commutation scheme,
but the chopper must fulfill the following: The coils must be open for some percentage of the chopper
period, in order to allow the back-EMF of the motor to influence the coil voltages. The motor direction
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
23
is determined by the start-up scheme, since the hallFX signals depend on the direction. Thus, the
same commutation scheme is used for turn right and turn left! Only a single commutation table is
required.
Bridge control signals (high active)
HallFX signals
Motor turning forward
H1
0V
H2
0V
H3
0V
BH1
0V
BL1
0V
BH2
0V
BL2
0V
BH3
0V
BL3
0V
Hall
vector
3
1
5
Chopper on high side
4
Motor turning reverse
6
2
(chopper events not shown)
Chopper on low side
5
1
3
2
6
4
Example: 50% chopper on
high and low side showing
3 chopper events
figure 17: hallFX based commutation
A chopper scheme fulfilling the desired coil open time per chopper period is shown here: Both, the
high side driver and the low side driver are chopped with the same signal. The coil open time
automatically is inverted to the duty cycle. In a practical application, the motor can run with a duty
cycle of 15% to 25% (minimum motor velocity at low load) up to 90% to 95% (maximum motor
velocity). The exact values depend on the actual motor. With a lower duty cycle the motor would not
start, or back EMF would be too small to yield a valid hallFX signal. With a higher duty cycle, the back
EMF would not be visible at the coil voltages, because the coils would be connected to GND or VM
nearly the whole time. The minimum resulting coil open time thus is 5% to 10%. This simple chopper
scheme automatically gives a longer measuring time at low velocities, when back EMF is lower. The
actual borders for the commutation should be checked in the actual application. Provide enough
headroom to compensate for variations due to motor load, mechanics and production stray.
5.4.4 Start-up sequence for the motor with forced commutation
In order to start the motor running with hallFX, it must reach a minimum velocity. The microcontroller
needs to take care of this by starting the motor in a forward control mode, without feedback – just like
a stepper motor. In order to allow a smooth transition to feedback mode, the same chopper scheme
should be used as described above. Alternatively, the chopper scheme can be changed a few
electrical periods before you switch to hallFX. This allows for example to start-up the motor using a
sine commutation, to get a smooth movement also at low motor velocities. In a practical application,
only a few percent up to 10% of the maximum motor velocity are sufficient for hallFX operation.
Turn the motor up to a minimum velocity, where you safely get correct hallFX signals. Since rotation of
the motor can not be measured during this phase, the motor needs to be current controlled, with a
current which in every case is high enough to turn the mechanical load. Current control can be done
by feedback control, or by adapting the duty cycle to the motor characteristics. Further, the minimum
starting speed and acceleration needs to be set matching the application. For sample code, please
see www.trinamic.com. Upon reaching the threshold for hallFX operation, a valid hall signal becomes
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
24
available and allows checking success of the starting phase. The turning direction of the start-up
sequence automatically determines the direction of motor operation with hallFX. You can check
velocity and direction, as soon as valid hallFX signals are available.
When you experience commutation sequence errors during motor operation, probably motor velocity
has dropped below the lower threshold. In this case, the motor could be restarted in forward control
mode, or you could switch to forward control mode on the fly.
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
5.5
25
Power supply
The TMC603 integrates a +12V switching regulator for the gate driver supply and a +5V linear
regulator for supply of the low voltage circuitry. The switching regulator is designed in a way, that it
provides the charge pump voltage by using a Villard voltage doubler circuit. It is able to provide
enough current to supply a number of digital circuits by adding an additional 3.3V or 5V low voltage
linear or switching regulator. If a +5V microcontroller with low current requirement is used, the +5V
regulator is sufficient, to also supply the microcontroller.
+VM
VM+10V
charge pump
BAV99 (70V)
BAS40-04W (40V)
220n
220n
100n
(2x)
4µ7
Tantal 25V
16V
LSW
SS16
TP0610K
or BSS84
(opt. BC857)
12V supply
100µ
(150mA with
sel. transistor)
100n
(2x)
optional supply filter
components when supply
ripple is high due to low
filter capacity for
transistor bridges
SMD
induct.
1µH
or 4R7
1µ
LSW: 220µH for 100kHz
VM
TMC603 voltage
regulators
SWOUT
startup
current
VLS
VCP
5V
linear
regulator
VM-12V /
2mA driver
triangle OSC
COSC
COSC: 470p ->100kHz
14k
+VCC
5VOUT
VCC
5V supply
100nF
+VCC
Q S
Q R
R
R1
5/12 VLS
R
150mV
triangle
1/5R
4/5R
R
R2
10R
dutycycle
limit
GND
figure 18: power supply block
Pin
Comments
COSC
Oscillator capacitor for step down regulator. A 470pF capacity gives 100kHz
operation. Do not leave this pin unconnected. Tie to GND, if oscillator is not used.
SWOUT
Switch regulator transistor output. The output allows driving of a small signal Pchannel MOSFETs as well as PNP small signal transistors
5VOUT
Output of internal 5V linear regulator. Provided for VCC supply
5.5.1 Switching regulator and charge pump
The switching regulator has been designed for high stability. It provides an upper duty cycle limit, in
order to ensure switching operation even at low supply voltage. This allows the combination with a
Villard voltage doubler. The application schematic shows a number of standard values, however, the
coil and oscillator frequency can be altered:
The choice of the external switching regulator transistor depends on the desired load current and the
supply voltage. Especially for high switching frequencies, a low gate charge MOSFET is required. The
following table shows an overview of available transistors and indicative operation limits. For a higher
output current, two transistors can be used in parallel. In this case the switching frequency should be
halved, because of the higher gate charge leading to slower switching slopes.
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
26
transistor
type
manufacturer
gate charge
(typ.)
max. frequency
max. voltage
max. load
current
BC857
div.
- (bipolar)
100 kHz
40V
80 mA
BSS84
Fairchild, NXP
0.9 nC
300 kHz
50V
120 mA
TP0610K
Vishay
1.3 nC
230 kHz
60V
150 mA
NDS0605
Fairchild
1.8 nC
175 kHz
60V
150 mA
TP0202K
Vishay
1 nC
300 kHz
30V
350 mA
For the catching diode, use a Schottky type with sufficient voltage and current rating.
The choice of a high switching frequency allows the use of a smaller and less expensive inductor as
well as a lower capacitance for the Villard circuit and the switching regulator output capacitor.
However, the combination of inductor, transistor and switching frequency should be carefully selected
and should be adapted to the load current, especially if a high load current is desired.
Choice of capacitor for the switching frequency (examples):
COSC
frequency fOSC
inductivity
example
470 pF
100 kHz
220 µH
220 pF
175 kHz
150 µH
100 pF
300 kHz
100 µH
Remark
Not recommended
for VVM < 14V
The switcher inductivity shall be chosen in a way, that it can sustain part of the load current between
each two switching events. If the inductivity is too low, the current will drop to zero and higher
frequency oscillations for the last part of each cycle will result (discontinuous mode). The required
transistor peak current will rise and thus efficiency falls.
For a low load current, operation in discontinuous mode is possible. If a high output current is required,
a good design value for continuous mode is to target a current ripple in the coil of +/-40%.
To give a coarse hint on the required inductor you can use the following formula for calculating the
minimum inductivity required for continuous operation, based on a ripple current which is 100% of the
load current:
VVM is the supply voltage. For low voltage operation (15V or less), the output voltage sinks from 12V to
0.85*VVM. The formula can be adapted accordingly.
IOUT is the current draw at 12V.
For 40% current ripple, you can use roughly the double inductivity.
If ripple is not critical, you can use a much smaller inductivity, e.g. only 5% to 50% of the calculated
value. But at the same time switching losses will rise and efficiency and current capability sink due to
higher losses in the switching transistor. If the TMC603 does not supply additional external circuitry,
current draw is very low, about 20mA in normal operation. This would lead to large inductivity values.
In this case we recommend going for the values given in the table above in order to optimize coil cost.
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
27
Example:
fOSC = 175 kHz, IOUT = 0.2 A, VVM = 48 V:
For continuous operation, a 330µH or 470µH coil would be required. The minimum inductivity
would be around 100µH.
Note:
Use an inductor, which has a sufficient nominal current rating. Keep switching regulator wiring
away from sensitive signals. When using an open core inductor, please pay special care to not
disturbing sensitive signals.
5.5.2 Charge pump
The Villard voltage doubler circuit relies on the switching regulator generating a square wave at the
switching transistor output with a height corresponding to the supply voltage. In order to work properly
the load drawn at +12V needs to be higher than the load drawn at the charge pump voltage. This
normally is satisfied when the IC is supplied by the step down regulator. For low voltage operation, the
charge pump voltage needs to be as high as possible to guarantee a high gate drive voltage, thus, a
dual Schottky diode should be used for the charge pump in low voltage applications.
5.5.3 Supply voltage filtering
As with most integrated circuits, ripple on the supply voltage should be minimized in order to
guarantee a stable operation and to avoid feedback oscillations via the supply voltages. Therefore,
use a ceramic capacitor of 100nF per supply voltage pin (VM to GND, VLS to GND and VCC to GND
and VCP to VM). Please pay attention to also keep voltage ripple on VCC pin low, especially when the
5V output is used to supply additional external circuitry. It also is important to make sure, that the
resistance of the power supply is low when compared to the load circuit. Especially high frequency
voltage ripple >1MHz should be suppressed using filter capacitors near the power bridge or near the
board power supply. The VM terminal is used, to detect short to GND situations, thus, it has to
correspond to the bridge power supply. In high noise applications, it may make sense to filter VCP
supply separately against ripple to GND. A large low ESR electrolytic capacitor across the bridge
supply (VM to GND) should also be used, because it effectively suppresses high frequency ripple. This
cannot be accomplished with ceramic capacitors.
5.5.4 Reverse polarity protection
Some applications need to be protected against a reversed biased power supply, i.e. for automotive
applications. A highly efficient reverse polarity protection based on an N channel MOSFET can simply
be added due to the availability of a charge pump voltage. This type of reverse polarity protection
allows feeding back energy into the supply, and thus is preferable to a pure diode reverse polarity
protection.
+Terminal
-Terminal
BC846
10k
Reverse polarity power
MOS (i.e. same type as
bridge transistors)
10k
+VM protected
(to bridge)
VM
VCP
figure 19: adding a reverse polarity protection
5.5.5 Standby with zero power consumption
In battery powered applications, a standby function often is desired. It allows switching the unit on or
off without the need for a mechanical high power switch. In principle, the bridge driver MOSFETs can
switch off the motor completely, but the TMC603 and its switching regulator still need to be switched
off in order to reduce current consumption to zero. Only a low energy standby power supply will
remain on, in order to wake up the system controller. This standby power supply can be generated by
a low current zener diode plus a resistor to the battery voltage, buffered by a capacitor. The example
in the schematic uses a P channel MOSFET to switch off power for the TMC603 and any additional
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
28
ICs which are directly supplied by the battery. In order to safely switch off the motor bridges during
standby, the gates of the high side MOSFETs become actively held in a discharged state using a low
current signal MOSFET in SOT23 package. Before entering standby mode, the motor shall be stopped
and the TMC603 should be disabled.
+Vbattery
+VM to bridge, only
100K
220n
POWER
SWITCH
FDC5614P
electronic
ON switch
+VM switched,
3A max.
10µ
VM
27k
TMC603
enable
ENABLE
Standby power
supply 5-12V
(only shown for
one high side
MOSFET)
HSx
1M
2N7002
nSTDBY
SAFE
OFF
2N7002
other bridges
figure 20: low power standby
5.5.6 Low voltage operation down to 9V
In low voltage operation, it is important to keep the gate driving voltages as high as possible. The
switching regulator for VLS thus is not needed and can be left out. Tie the pin COSC to GND. VLS
becomes directly tied to +VM, which is possible as long as the supply voltage does not exceed 14V
(16V peak). However, now a source for the Villard voltage doubler is missing. A simple solution is to
use a CMOS 555 timer circuit (e.g. TLC555) oscillating at 250 kHz (square wave) to drive the voltage
doubler.
+9V...14V
VM+10V
charge pump
100n
BAS40-04W
470n
16V
12V supply
VCC
DISCH
100n
(2x)
100n
(2x)
RESET
OUT
TLC555
THRES
CONT
(150mA with
sel. transistor)
optional supply filter
components when supply
ripple is high due to low
filter capacity for
transistor bridges
SMD
induct.
1µH
or 4R7
22k
1µ
TRIG
GND
150p
VM
SWOUT
VLS
VCP
5VOUT
COSC
5V supply
VCC
100nF
TMC603
figure 21: low voltage operation
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
5.6
Test output
The test output is reserved for manufacturing test. Leave open for a normal application.
Pin
Comments
TEST
Output for test voltages. Output resistance 25kOhm +-30%.
Reset: ENABLE(=low); Clock: SCCLK (rising edge).
Test voltage sequence:
0: 0V
1..3 / 4..6 / 7..9: Gate_HS_Off, Gate_LS_On, Gate_LS_Off (driver 1/2/3)
10..14: currently unused
15: 0V (no further counts: Reset for restart)
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
29
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
30
6 Absolute Maximum Ratings
The maximum ratings may not be exceeded under any circumstances. Operating the circuit at or near
more than one maximum rating at a time for extended periods shall be avoided by application design.
Parameter
Symbol
Supply voltage
Supply and bridge voltage max. 20000s
VVM
Min
Max
Unit
-0.5
50
V
55
V
Low side driver supply voltage
VVLS
-0.5
14
V
Low side driver supply voltage max. 20000s
VVLS
-0.5
16
V
-0.5
60
V
65
V
VM-10
VM+16
V
VVCC
-0.5
6.0
V
Logic input voltage
VI
-0.5
VCC+0.5
V
Analog input voltage
VIA
-0.5
VCC+0.5
V
Voltages on driver pins (HSx, LSx, BMx)
VDRVIO
-0.7
0.7
V
Relative high side driver voltage (VHSX – VBMX)
VHSBM
-20
20
V
Charge pump voltage (related to GND)
Charge pump voltage max. 20000s
VVCP
Charge pump voltage during power up / down
Logic supply voltage
Maximum current to / from digital pins
and analog low voltage I/Os
IIO
+/-10
mA
5V regulator continuous output current
I5VOUT
40
mA
5V regulator short time output current
I5VOUT
150
mA
Junction temperature
TJ
-50
150
°C
Storage temperature
TSTG
-55
150
°C
ESD-Protection (Human body model, HBM)
VESD
1
kV
7 Electrical Characteristics
7.1
Operational Range
Parameter
Symbol
Min
Max
Unit
Ambient temperature
TA
-40
125
°C
Junction temperature
TJ
-40
140
°C
10
50
V
9
14
V
Supply voltage (standard circuit)
Supply voltage (low voltage application: VVLS=VVM)
VVM
Low side driver supply voltage
VVLS
9
13
V
Differential charge pump voltage measured to VM (VVCP – VVM)
VCPD
8
12
V
Logic supply voltage
VVCC
4.75
5.25
V
Slope control resistor with VVM <30V
RSLP
60
500
k
Slope control resistor with VVM >30V
RSLP
100
500
k
Short to GND control resistor
RS2G
47
1000
k
Output slope
tSLP
100
1000
ns
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
7.2
31
DC Characteristics and Timing Characteristics
DC characteristics contain the spread of values guaranteed within the specified supply voltage range
unless otherwise specified. Typical values represent the average value of all parts measured at
+25°C. Temperature variation also causes stray to some values. A device with typical values will not
leave Min/Max range within the full temperature range.
NMOS low side driver
note 1)
Parameter
DC-Characteristics
VVCC = 5.0 V, VVLS = 12V
Symbol
Conditions
Min
Typ
Max
Unit
Gate drive current LSx
low side switch ON
ILSON
VLSX = 5V
RSLP = 68k
125
150
180
mA
Gate drive current LSx
low side switch OFF
ILSOFF
VLSX = 5V
RSLP = 68k
-125
-160
-190
mA
Gate drive current LSx
low side switch ON
ILSON
VLSX = 5V
RSLP = 100k
88
105
126
mA
Gate drive current LSx
low side switch OFF
ILSOFF
VLSX = 5V
RSLP = 100k
-90
-115
-138
mA
Gate drive current LSx
low side switch ON
ILSON
VLSX = 5V
RSLP = 220k
50
mA
Gate drive current LSx
low side switch OFF
ILSOFF
VLSX = 5V
RSLP = 220k
-50
mA
Gate Off detector threshold
VGOD
VLSX falling
1
V
QGD protection resistance after
detection of gate off
RLSOFFQGD VLSX = 2V
15
Delay LS driver switch on
BLx to LSx at 50%
tLSON
RSLP = 100k
CLSX = 100pF
35
70
140
ns
Delay LS driver switch off
BLx to LSx at 50%
tLSOFF
RSLP = 100k
CLSX = 100pF
80
160
320
ns
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
NMOS high side driver
note 1)
Parameter
32
DC-Characteristics
VVCC = 5.0 V, VVLS = 12V, VCPD = 10.5V
Symbol
Conditions
Min
Typ
Max
Unit
Gate drive current HSx
high side switch ON
IHSON
VHSX = 5V
RSLP = 68k
112
135
180
mA
Gate drive current HSx
high side switch OFF
IHSOFF
VHSX = VM+5V
RSLP = 68k
-120
-150
-180
mA
Gate drive current HSx
high side switch ON
IHSON
VHSX = 5V
RSLP = 100k
75
90
120
mA
Gate drive current HSx
high side switch OFF
IHSOFF
VHSX = VM+5V
RSLP = 100k
-80
-100
-120
mA
Gate drive current HSx
high side switch ON
IHSON
VHSX = 5V
RSLP = 220k
50
mA
Gate drive current HSx
high side switch OFF
IHSOFF
VHSX = VM+5V
RSLP = 220k
-50
mA
Gate Off detector threshold high
side VHSX-VBMX, BM level high
VGOD
VHSX falling
VBMX > VGOBM
0
V
Gate Off detector threshold high
side VBMX, BM level low
VGOBM
VBMX falling
3.5
V
VBMX = 24V
VHSX = VBMX+2V
300
mA
QGD protection current after
detection of gate off
IHSOFFQGD
Delay HS driver switch on
BHx to HSx at 50%
tHSON
RSLP = 100k
VM = 24V
CHSX = 100pF
75
150
300
ns
Delay HS driver switch off
BHx to HSx at 50%
tHSOFF
RSLP = 100k
VM = 24V
CHSX = 100pF
60
120
240
ns
Min
Typ
Max
Unit
Break-before-make block
note 1)
Parameter
Timing-Characteristics
VVM = 48 V, RSLP = 100K
Symbol
Conditions
Break-before-make delay LSx off
to HSx on
tBBMLH
Measured at 1V
gate-source voltage
160
ns
Break-before-make delay HSx
off to LSx on
tBBMHL
Measured at 1V
gate-source voltage
290
ns
1)
See timing diagram in figure 6: bridge driver timing
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
RSLP input and RS2G input
Parameter
Typical voltage at RSLP and
RS2G input, depending on the
external resistor
33
DC-Characteristics
VVCC = 5.0 V
Symbol
Conditions
VRSLP
VRS2G
RSLP = 100 k
Min
Typ
Max
3.8
Unit
V
RS2G = 100 k
4.0
RSLP = 470 k
RS2G = 470 k
Short to GND detector
Parameter
DC-Characteristics, Timing-Characteristics
VVM = 24 V
Symbol
Short to GND detection level
(VVM – VBM)
VBMS2G
Short to GND detector delay
(HSx going active to short
detector active / ERR_OUT
falling)
tS2G
Supply current
Parameter
Conditions
Min
Typ
Max
Unit
1
1.5
2.3
V
RS2G = 68k
200
320
450
ns
RS2G = 150k
500
750
1000
ns
RS2G = 220k
700
1000
1300
ns
RS2G = 470k
1400
2000
2600
ns
DC-Characteristics
VVCC = 5.0 V, VVLS = 12V, VCPD = 10.5V, RSLP = 100k, VVM = 48V
Symbol
Conditions
Min
Typ
Max
Unit
0.45
0.68
mA
4.6
6.9
mA
VM supply current
IVM
VLS supply current
IVLS
VCP supply current
IVCP
1.6
2.4
mA
VCC supply current
IVCC
2.9
4.4
mA
Min
Typ
Max
Unit
7
7.85
8.5
V
5.8
6.6
Undervoltage detectors
Parameter
DC-Characteristics
VVCC = 5.0 V
Symbol
VLS undervoltage level
VVLSUV
VCP undervoltage level (VVCPVM)
VCPDUV
VCP voltage OK level (VVCP-VM)
not including I5VOUT
Conditions
VVCP falling
VVCP rising
VCP undervoltage detector
Hysteresis
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
7.1
0.5
V
7.8
V
V
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
Switching regulator /
Charge pump
Parameter
34
DC-Characteristics
VVCC = V5VOUT
Symbol
Conditions
Min
Typ
Max
Unit
-1.5
-2.2
-3.0
mA
Switch output drive current (on)
ISWOUT
VSWOUT = VVM
Switch output drive current (off)
ISWOUT
VSWOUT = VVM - 5V
Switch start-up drive current
during VCC undervoltage
ISWOUT
VSWOUT = VVM
VVM = 24V
VVLS < 2V
Switch output drive voltage (on)
VVM - VSWOUT
VSWOUT
Switch regulator output voltage
V12VOUT
10
mA
-0.4
-0.8
mA
ISWOUT = 0
8
12
15
V
VVM > 16V
11
12
13.1
V
VVLSUV < VVM < 16V
0.85 VVM
TJ = 25°C
V
Oscillator output resistance
RCOSC
14.1
Lower oscillator threshold
voltage
VCOSC
1/3 VVCC
V
Upper oscillator threshold
voltage
VCOSC
2/3 VVCC
V
Oscillator threshold voltage for
maximum duty cycle limit
VCOSC
6/15
VVCC
V
k
Maximum duty cycle switch
regulator
DCSWOUT
VVLS = 10V
fCHOP = 100kHz
63
70
77
%
Chopper frequency nominal
fCHOP
COSC = 470pF
70
100
130
kHz
Chopper frequency range
(design reference value only)
fCHOP
300
kHz
Charge pump voltage
(design reference value only)
VCPD
Linear regulator
Parameter
0 (off)
VVLS = 12V
IVCP = 1.6mA
10.6
V
DC-Characteristics
Symbol
Conditions
Min
Typ
Max
Unit
4.75
5.0
5.25
V
Output voltage
V5VOUT
I5VOUT = 10mA
TJ = 25°C
Output resistance
R5VOUT
Static load
Output current capability
I5VOUT
VVLS = 12V
100
mA
VVLS = 8V
60
mA
VVLS = 6.5V
20
mA
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
2
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
Digital logic level
Parameter
35
DC-Characteristics
VVCC = 5.0 V +/-10%
Symbol
Conditions
Min
Typ
Max
Unit
Input voltage low level
VINLO
-0.3
0.8
V
Input voltage high level
VINHI
2.0
VVCC+0.3
V
0.4
V
Output voltage low
(H1, H2, H3, ERR_OUT)
VOUTLO
IOUTLO = 1mA
Output voltage high
(H1, H2, H3)
VOUTHI
IOUTHI = -1mA
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
0.8VVCC
V
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
Current measurement block
36
DC-Characteristics, Timing-Characteristics
VVM = 24 V, VVCC = 5.0 V
Parameter
Symbol
Conditions
Min
Typ
Max
Unit
Amplification of voltage
VBMX to VCURX
for positive voltage (VBMX > 0V)
ACURLO+
SENSE_HI = GND
-4.3
-4.6
-4.9
V/V
ACURHI+
SENSE_HI = VCC
-17
-18.0
-19
V/V
Amplification of voltage
VBMX to VCURX
for negative voltage (VBMX < 0V)
ACURLO-
SENSE_HI = GND
-4.1
-4.4
-4.7
V/V
ACURHI-
SENSE_HI = VCC
-16
-17.1
-18
V/V
Zero current level at CURX
V0CURX
VVCC/3
-50mV
VVCC/3
VVCC/3
+50mV
V
Measurement voltage range at
VBMX
VBMX
SENSE_HI = GND
-250
300
mV
SENSE_HI = VCC
-80
80
mV
VCURX output voltage swing low
VCURX
0.1
V
VCURX output voltage swing high
VCURX
Ripple voltage / hold step noise
at output
note 2)
VCURX
Minimum low side on time for
current measurement
(Delay from BLx going active to
CURx tracking amplified signal)
tBLHICURX
SAMPLEx = VCC
Delay from SAMPLEx going
active to CURx tracking
amplified signal
tSMPHICURX
SAMPLEx = VCC
Delay from BLx or SAMPLEx
going inactive to CURx hold
0.02
VVCC-1.2 VVCC-0.6
V
VBMX = 0V
SENSE_HI = GND
17
26
mV
VBMX = 0V
SENSE_HI = VCC
50
75
mV
5.3
7.2
µs
3.5
tBLHICURX
/2
µs
tBLXLO
0
µs
Sample and hold drop during
hold period
dVCURX
0.001
1.6
V/s
Auto zero drop of current
amplifier during sampling period
(low side on)
dVCURX
0.003
3
V/s
Minimum initial auto zero period
(low side off or SAMPLEx low)
after power on
tBLXLO0
5
µs
Minimum sample period after
tBLHICURX for a 100% current step
tBLXHIADD
1
µs
0.45
mA
Output current limit at CURx
ICURX
Current sourcing
2) Note on first ICs dated before 20/2009:
CURx outputs are sensible to ripple voltage on VCC pin and frequency below 5MHz. Ripple voltage is amplified by 1/3 * Set
amplification, i.e. factor 1.5 with SENSE_HI low and factor 6 with SENSE_HI high. Thus, it is suggested to use 5VOUT only for
VCC supply, if possible, if exact measurements are required.
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
Switched capacitor filter 2
order
nd
Parameter
DC-Characteristics, AC-Characteristics
VVM = 24 V, VVCC = 5.0 V
Symbol
Conditions
Attenuation of voltage
VBMX to VFILTX
AFILTLO
VBMX > 0.9V
Output resistance of FILTX
RFILTX
Output current limit at FILTX
IFILTX
Noise voltage on FILTX
3dB bandwidth
37
Min
fCUTOFF
3.0
fSCCLK = 2.5MHz
Switched capacitor filter clock
frequency for normal operation
hallFX unit
Parameter
fSCCLK
30
Unit
V
40
20
VFILTXNOISE VBMX = 12V
fSCCLK = 1.25MHz
fSCCLK = 1.25MHz
Max
VBMX/15
– 0.9V
21
Current sourcing
Typ
k
µA
20
mV
1/390
fSCCLK
Hz
3.2
3.4
6.4
0.25
kHz
kHz
4
MHz
Typ
Max
Unit
50
150
mV
-300
0
300
mV
DC-Characteristics, AC-Characteristics
VVM = 24 V, VVCC = 5.0 V
Symbol
Conditions
Min
Noise voltage of comparators
VCOMPNOISE VBMX = 12V
including switched capacitor filter
Offset voltage of comparators
VCOMPOFFS VBMX = 12V
including switched capacitor filter
and input attenuation
Spike suppression comparator
threshold
VSP_SUP
VSP_SUP rising
2.0
VVCC/2
2.8
V
Spike suppression capacitor
charging current
ISP_SUP
VSP_SUP = 1V
15
25
35
µA
Spike suppression capacitor
discharging current
ISP_SUP
VSP_SUP = 1V
-0.5
-1
-1.5
mA
Dead time for spike suppression
tSP_SUP
CSP_SUP = 1nF
60
100
180
µs
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
38
8 Designing the application
8.1
Choosing the best fitting power MOSFET
There is a huge choice of power MOSFETs available. MOSFET technology has been improved
dramatically in the last 20 years, and gate drive requirements have shifted from generation to
generation. The first generations of MOSFETs have a comparatively high gate capacity at a moderate
RDSon. Their gate-source capacity is two to five times as high as the capacity of the gate-drain
junction. These MOSFETs have a high gate charge and thus require high current gate drive, but they
are easy to use, because internal feedback is low. In the early 2000s new MOSFETs have emerged,
where RDSon is much lower, and gate-source capacity has been improved by minimizing structural
overlap. Thus, the capacitance ratio has shifted, and feedback has become quite high. These
MOSFETs thus are much more critical, and power drives have to actively force the gate off to prevent
the bridges from cross-conduction due to feedback from the drain to gate. Latest generation
MOSFETs, like the Vishay W-Fet technology, further reduce RDSon, while reducing the capacity
between the channel and the drain. Thus, these MOSFETs have lowest gate charge, and again, are
easier to control than the previous generation of MOSFETs.
When choosing the MOSFET, the following points shall be considered:
Maximum voltage:
Choose at least a few volts above your maximum supply voltage, taking into account that the
motor can feed back energy when slowing down, and thus the supply voltage can rise. On the
other hand, a transistor rated for a higher voltage is more expensive and has a higher gate charge
(see next chapter).
RDSon:
A low RDSon gives low static dissipation, but gate charge and cost increases. Take into account,
that a good part of the power dissipation results from the switching events in a chopped drive
system. Further, to allow a current measurement, the RDSon should be in a range, that the
voltage drop can be used for measurement. A voltage drop of 50mV or higher at nominal motor
current is a good target.
Gate charge and switching speed:
The switching speed should be compared to the required chopper frequency. Choose the chopper
frequency low to reduce dynamic losses. When the application does not require slow, EMV
optimized switching slopes, choose a low gate charge transistor to reduce dynamic losses.
Gate threshold voltage:
Most MOSFETs have a specified on-resistance at a gate drive voltage of 10V. Some MOSFETs
are optimized for direct control from logic ICs with 5 or even 3.3V. They provide a low gate
threshold voltage of 1V to 2V. MOSFETs with higher gate threshold voltage should be preferred,
because they are less sensible to effects of the drain gate capacity and cross conduction.
Package, size and cooling requirements
Cost and availability
8.1.1 Calculating the MOSFET power dissipation
The power dissipation in the MOSFETs has three major components: Static losses (PSTAT) due to
voltage drop, switching losses (PDYN) due to signal rise and fall times, losses due to diode conduction
(PDIODE). The diode power dissipation depends on many factors (back EMF of the motor, inductivity
and motor velocity), and thus is hard to calculate from motor data. Normally, it contributes for a few
percent to some ten percent of overall power dissipation. Other sources for power dissipation are the
reverse recovery time of the transistors and the gate drive energy. Reverse recovery also causes
current spikes on the bridges. If desired, you can add Schottky diodes over the (chopper) transistors to
reduce the diode losses and to eliminate current spikes caused by reverse recovery.
The following sample calculation assumes a three phase BLDC motor operated in block commutation
mode with dual sided chopper. At each time, two coils conduct the full motor current (chopped).
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
39
where
IMOTOR is the motor current, e.g. 10A
RDSON is the on-resistance of the MOSFETs at a gate voltage of about 10V, e.g. 20mΩ
tDUTY is the actual duty cycle of the chopper, e.g. 80% = 0.8
VVM is the motor supply voltage, e.g. 24V or 48V
fCHOP is the chopper frequency, e.g. 20kHz
tSLOPE is the slope (transition) time, e.g. 300ns
Example:
With the example data for a 10A motor at 24V, we get the following power dissipation:
PSTAT = 3.2W
PDYN24 = 2.88W
For comparison: The motor output power is 10A*24V*0.8=192W
The dynamic and static dissipation here are in a good ratio, thus the choice of a 20mΩ
MOSFET is good.
At 48V, the dynamic power dissipation doubles:
PDYN48=5.76W
Here, the dynamic losses are higher than the static losses. Thus, we should reduce the slope time.
Given that the gate capacity would not allow for faster slopes than 300ns, we could go for a 30mΩ
MOSFET, which has a lower gate charge and thus allows faster slopes, e.g. 200ns. With these
modifications we get a static loss of 4.8W and a dynamic loss of 3.84W. This in sum is 8.64W, which is
slightly less than the 8.96W before. At the same time, system cost has decreased due to lower cost
MOSFETs. The loss is still low when compared to a motor power of 384W.
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
8.2
40
MOSFET examples
There is a huge number of MOSFETs on the market, which can be used in combination with the
TMC603. The user choice will depend on the electrical data (voltage, current, RDSon) and on the
package and configuration (single / dual). The following table gives a few examples of SMD MOSFETs
for different motor currents. The MOSFETs explicitly are modern types with a low total gate charge.
With dual configurations, only three MOSFET packages are required to control a BLDC motor, but the
current which can be reached is significantly lower due to thermal restrictions of the packages.
For the actual application, we suggest to calculate static and dynamic power dissipation for a given
MOSFET, as described in the previous chapter. Especially for sine commutation and chopper
frequencies above 20kHz, transistors with a gate charge below 100nC should be preferred.
Transistor
type
manufacturer
unit
RDSon
voltage
mΩ
V
package &
max. motor
configuration current (*)
total gate
charge @10V
A
nC
IBP019N06L3
Infineon
1.9
60
DPACK
30
124
SiE812DF
Vishay
2.6
40
PolarPAK
30
52
IPP032N06N3
Infineon
2.9
60
TO220
30
125
IRFB3306
International
Rectifier
4.2
60
TO220 /
DPACK
30
85
SiE816DF
Vishay
7.4
60
PolarPAK
20
51
SUM75N0609L
Vishay
9.3
60
D2PAK
(TO263)
25
47
FDD10AN06A0 Fairchild
10.5
60
DPAK
(TO252A)
20
28
FDD5353
Fairchild
12.3
60
DPAK
15
46
SI7964DP
Vishay
23
60
PowerPAKSO8 (dual)
9.6
43
SI4946
Vishay
55
60
SO8 (dual)
4.5
19
(*) Remark: The maximum motor current applicable in a given design depends upon PCB size and
layout, since all of these transistors are mainly cooled via the PCB. The data given implies adequate
cooling measures taken by the user, especially for higher current designs.
8.3
Driving a DC motor with the TMC603
The TMC603 can also be used for DC motor applications, using a full bridge or a half bridge for motor
PWM operation with or without reverse direction operation. For single half bridge applications, all
TMC603 gate drivers can be paralleled, taking advantage of the three time increase in gate drive
capability. This way a motor current of up to 100A can be driven. The drive system can use the shunt
less current sensing capability for best efficiency. A Schottky diode across the non-chopped transistor
optimizes slopes and electromagnetic emission characteristics (see chapter 5.2.8).
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG
TMC603 DATA SHEET (V. 1.06 / 26. Mar. 2009)
41
9 Table of figures
FIGURE 1: APPLICATION BLOCK DIAGRAM ..................................................................................................... 5
FIGURE 2: PINNING / QFN52 PACKAGE (TOP VIEW) ....................................................................................... 6
FIGURE 3: APPLICATION DIAGRAM ................................................................................................................ 8
FIGURE 4: THREE PHASE BLDC DRIVER..................................................................................................... 10
FIGURE 5: PRINCIPLE OF HIGH-SIDE DRIVER (PAT.FIL.) ................................................................................ 10
FIGURE 6: BRIDGE DRIVER TIMING ............................................................................................................. 12
FIGURE 7: EXAMPLES FOR MICROCONTROLLER PWM CONTROL .................................................................. 13
FIGURE 8: MOSFET GATE CHARGE AS AVAILABLE IN DEVICE DATA SHEET VS. SWITCHING EVENT ................. 13
FIGURE 9: QGD PROTECTED DRIVER STAGE .............................................................................................. 14
FIGURE 10: EFFECT OF BULK DIODE RECOVERY .......................................................................................... 16
FIGURE 11: PARALLEL SCHOTTKY DIODE AVOIDS CURRENT SPIKES DUE TO BULK DIODE RECOVERY .............. 17
FIGURE 12: TIMING OF THE SHORT TO GND DETECTOR .............................................................................. 17
FIGURE 13: ERROR LOGIC ......................................................................................................................... 18
FIGURE 14: SCHEMATIC OF CURRENT MEASUREMENT AMPLIFIERS ............................................................... 19
FIGURE 15: TIMING OF THE CURRENT MEASUREMENT ................................................................................. 20
FIGURE 16: HALLFX BLOCK DIAGRAM AND TIMING....................................................................................... 21
FIGURE 17: HALLFX BASED COMMUTATION ................................................................................................ 23
FIGURE 18: POWER SUPPLY BLOCK ........................................................................................................... 25
FIGURE 19: ADDING A REVERSE POLARITY PROTECTION.............................................................................. 27
FIGURE 20: LOW POWER STANDBY ............................................................................................................ 28
FIGURE 21: LOW VOLTAGE OPERATION ...................................................................................................... 28
10 Revision History
10.1 Documentation Revision
Version
Author
Description
0.94
BD
TMC603 initial release with preliminary electrical data
0.96
BD
Added package dimensions
0.98
BD
Added microcontroller PWM control examples
0.99
BD
Added reverse polarity protection and MOSFET examples
1.00
BD
Added low power standby and low voltage operation
1.01
BD
Removed “preliminary” indication, modifications in electrical characteristic
tables
1.02
BD
Slightly corrected a few values
1.03
BD
Added transistor examples and temperature information to tables
1.04
BD
Slight beautifications / rewording
1.05
BD
Added mathematical background for QGD protection, discussion on
MOSFET bulk diode and DC motor application
1.06
BD
Added minimum output voltage swing of current amplifiers
Table 1: Documentation Revisions
Copyright © 2008 TRINAMIC Motion Control GmbH & Co. KG