TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 D D D D D D D D D D D Compatible With PC 99 Desktop Line-Out Into 10-kΩ Load Compatible With PC 99 Portable Into 8-Ω Load Internal Gain Control, Which Eliminates External Gain-Setting Resistors 2-W/Ch Output Power Into 3-Ω Load Input MUX Select Terminal PC-Beep Input Depop Circuitry Stereo Input MUX Fully Differential Input Low Supply Current and Shutdown Current Surface-Mount Power Packaging 24-Pin TSSOP PowerPAD PWP PACKAGE (TOP VIEW) GND GAIN0 GAIN1 LOUT+ LLINEIN LHPIN PVDD RIN LOUT– LIN BYPASS GND 1 2 3 4 5 6 7 8 9 10 11 12 24 23 22 21 20 19 18 17 16 15 14 13 GND RLINEIN SHUTDOWN ROUT+ RHPIN VDD PVDD HP/LINE ROUT– SE/BTL PC-BEEP GND description The TPA0222 is a stereo audio power amplifier in a 24-pin TSSOP thermally enhanced package capable of delivering 2 W of continuous RMS power per channel into 3-Ω loads. This device minimizes the number of external components needed, simplifying the design, and freeing up board space for other features. When driving 1 W into 8-Ω speakers, the TPA0222 has less than 0.5% THD+N across its specified frequency range. Included within this device is integrated depop circuitry that virtually eliminates transients that cause noise in the speakers. Amplifier gain is internally configured and controlled by two terminals (GAIN0 and GAIN1). BTL gain settings of 2, 6, 12, and 24 V/V are provided, while SE gain is always configured as 1 V/V for headphone drive. An internal input MUX allows two sets of stereo inputs to the amplifier. The HP/LINE terminal allows the user to select which MUX input is active regardless of whether the amplifier is in SE or BTL mode. In notebook applications, where internal speakers are driven as BTL and the line outputs (often headphone drive) are required to be SE, the TPA0222 automatically switches into SE mode when the SE/BTL input is activated, and reduces the gain to 1 V/V. The TPA0222 consumes only 18 mA of supply current during normal operation. A miserly shutdown mode reduces the supply current to less than 150 µA. The PowerPAD package (PWP) delivers a level of thermal performance that was previously achievable only in TO-220-type packages. Thermal impedances of approximately 35°C/W are truly realized in multilayer PCB applications. This allows the TPA0222 to operate at full power into 8-Ω loads at an ambient temperature of 85°C. Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. PowerPAD is a trademark of Texas Instruments Incorporated. Copyright 1999, Texas Instruments Incorporated PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 1 TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 functional block diagram RHPIN RLINEIN R MUX – ROUT+ + RIN – ROUT– + PC-BEEP GAIN0 GAIN1 SE/BTL PCBeep Gain/ MUX Control Depop Circuitry LLINEIN PVDD VDD BYPASS SHUTDOWN GND HP/LINE LHPIN Power Management L MUX – LOUT+ + LIN – LOUT– + 2 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 AVAILABLE OPTIONS TA PACKAGED DEVICE TSSOP† (PWP) – 40°C to 85°C TPA0222PWP † The PWP package is available taped and reeled. To order a taped and reeled part, add the suffix R to the part number (e.g., TPA0222PWPR). Terminal Functions TERMINAL NAME NO. I/O DESCRIPTION BYPASS 11 GAIN0 2 I Bit 0 of gain control GAIN1 3 I Bit 1 of gain control GND Tap to voltage divider for internal mid-supply bias generator 1, 12, 13, 24 Ground connection for circuitry. Connected to the thermal pad LHPIN 6 I Left channel headphone input, selected when SE/BTL is held high LIN 10 I Common left input for fully differential input. AC ground for single-ended inputs LLINEIN 5 I Left channel line input, selected when SE/BTL is held low LOUT+ 4 O Left channel positive output in BTL mode and positive output in SE mode LOUT– 9 O Left channel negative output in BTL mode and high-impedance in SE mode PC-BEEP 14 I The input for PC Beep mode. PC-BEEP is enabled when a > 1-V (peak-to-peak) square wave is input to PC-BEEP or PCB ENABLE is high. HP/LINE 17 I HP/LINE is the input MUX control input. When the HP/LINE terminal is held high, the headphone inputs (LHPIN or RHPIN [6, 20]) are active. When the HP/LINE terminal is held low, the line BTL inputs (LLINEIN or RLINEIN [5, 23]) are active. 7, 18 I Power supply for output stage PVDD RHPIN 20 I Right channel headphone input, selected when SE/BTL is held high RIN 8 I Common right input for fully differential input. AC ground for single-ended inputs RLINEIN 23 I Right channel line input, selected when SE/BTL is held low ROUT+ 21 O Right channel positive output in BTL mode and positive output in SE mode ROUT– 16 O Right channel negative output in BTL mode and high-impedance in SE mode SHUTDOWN 22 I Places entire IC in shutdown mode when held low, except PC-BEEP remains active SE/BTL 15 I Hold SE/BTL low for BTL mode and hold high for SE mode. VDD 19 I Analog VDD input supply. This terminal needs to be isolated from PVDD to achieve highest performance. POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 3 TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 absolute maximum ratings over operating free-air temperature range (unless otherwise noted)† Supply voltage, VDD . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 V Input voltage, VI . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to VDD +0.3 V Continuous total power dissipation . . . . . . . . . . . . . . . . . . . . . internally limited (see Dissipation Rating Table) Operating free-air temperature range, TA . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . – 40°C to 85°C Operating junction temperature range, TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . – 40°C to 150°C Storage temperature range, Tstg . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –65°C to 150°C Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 260°C † Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. DISSIPATION RATING TABLE TA ≤ 25°C 2.7 W‡ PACKAGE PWP DERATING FACTOR 21.8 mW/°C TA = 70°C 1.7 W TA = 85°C 1.4 W ‡ Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report (literature number SLMA002), for more information on the PowerPAD package. The thermal data was measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended Board for PowerPAD on page 33 of the before mentioned document. recommended operating conditions ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ Supply voltage, VDD High level input voltage, High-level voltage VIH MIN MAX 4.5 5.5 SE/BTL, HP/LINE 4 SHUTDOWN 2 SE/BTL, HP/LINE UNIT V V 3 ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ Low level input voltage, Low-level voltage VIL SHUTDOWN 0.8 Operating free-air temperature, TA – 40 85 V °C electrical characteristics at specified free-air temperature, VDD = 5 V, TA = 25°C (unless otherwise noted) PARAMETER TEST CONDITIONS |VOS| Output offset voltage (measured differentially) PSRR Power supply rejection ratio VI = 0, AV = –2 V/V VDD = 4.9 V to 5.1 V |IIH| High-level input current VDD = 5.5 V, VI = VDD |IIL| Low-level input current VDD = 5.5 V, VI = 0 V IDD Supply current IDD(SD) Supply current, shutdown mode 4 MIN TYP 25 77 BTL mode 18 SE mode 9 150 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 MAX UNIT mV dB 900 nA 900 nA mA 300 µA TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 operating characteristics, VDD = 5 V, TA = 25°C, RL = 8 Ω, Gain = –2 V/V, BTL mode ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ PARAMETER PO Output power THD + N Total harmonic distortion plus noise BOM Maximum output power bandwidth Supply ripple rejection ratio SNR Signal-to-noise ratio Vn Noise output voltage ZI Input impedance TEST CONDITIONS THD = 1%, RL = 4 Ω f = 1 kHz, PO = 1 W, THD = 5% f = 20 Hz to 15 kHz f = 1 kHz, CB = 0.47 µF CB = 0.47 µ µF,, f = 20 Hz to 20 kHz BTL mode MIN TYP MAX UNIT 1.9 W 0.5% >15 kHz 68 dB 105 dB BTL mode 16 SE mode 30 µV RMS See Table 1 TYPICAL CHARACTERISTICS Table of Graphs FIGURE THD+N Total harmonic distortion plus noise vs Output power 1, 4–7, 10–13, 16–19, 21 vs Frequency 2, 3, 8, 9, 14, 15, 20, 22 vs Output voltage Vn SNR Output noise voltage vs Bandwidth 24 Supply ripple rejection ratio vs Frequency 25, 26 Crosstalk vs Frequency 27–29 Shutdown attenuation vs Frequency 30 Signal-to-noise ratio vs Bandwidth 31 Closed loop respone PO PD 23 32–35 Output power Power dissipation vs Load resistance 36, 37 vs Output power 38, 39 vs Ambient temperature POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 40 5 TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 TYPICAL CHARACTERISTICS TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY 10% AV = 2 V/V f = 1 kHz BTL THD+N –Total Harmonic Distortion + Noise THD+N –Total Harmonic Distortion + Noise 10% RL = 4 Ω 1% RL = 8 Ω RL = 3 Ω 0.1% 0.01% 0.5 0.75 1 1.25 1.5 1.75 2 2.25 2.5 2.75 PO = 1.75 W RL = 3 Ω BTL 1% AV = –24 V/V AV = –12 V/V AV = –2 V/V 0.1% AV = –6 V/V 0.01% 20 3 PO – Output Power – W 10% RL = 3 Ω AV = –2 V/V BTL THD+N –Total Harmonic Distortion + Noise THD+N –Total Harmonic Distortion + Noise TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER 1% PO = 1.0 W PO = 0.5 W 0.1% PO = 1.75 W 100 1k 10k 20k f = 15 kHz 1% f = 1 kHz 0.1% f = 20 Hz RL = 3 Ω AV = –2 V/V BTL 0.01% 0.01 f – Frequency – Hz Figure 3 6 10k 20k Figure 2 TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY 0.01% 20 1k f – Frequency – Hz Figure 1 10% 100 0.1 1 PO – Output Power – W Figure 4 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 10 TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 TYPICAL CHARACTERISTICS TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER 10% THD+N –Total Harmonic Distortion + Noise THD+N –Total Harmonic Distortion + Noise 10% f = 15 kHz 1% f = 1 kHz f = 20 Hz 0.1% 0.01% 0.01 RL = 3 Ω AV = –6 V/V BTL 0.1 1 PO – Output Power – W f = 15 kHz 1% f = 1 kHz f = 20 Hz 0.1% 0.01% 0.01 10 RL = 3 Ω AV = –12 V/V BTL 0.1 1 PO – Output Power – W Figure 5 Figure 6 TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY 10% f = 15 kHz 1% THD+N –Total Harmonic Distortion + Noise THD+N –Total Harmonic Distortion + Noise 10% f = 1 kHz f = 20 Hz 0.1% 0.01% 0.01 10 RL = 3 Ω AV = –24 V/V BTL 0.1 1 PO – Output Power – W 10 PO = 1.5 W RL = 4 Ω BTL AV = –24 V/V 1% AV = –12 V/V 0.1% AV = –2 V/V AV = –6 V/V 0.01% 20 100 1k 10k 20k f – Frequency – Hz Figure 7 Figure 8 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 7 TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 TYPICAL CHARACTERISTICS TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY 10% RL = 4 Ω AV = –2 V/V BTL 1% 0.1% THD+N –Total Harmonic Distortion + Noise THD+N –Total Harmonic Distortion + Noise 10% PO = 1.5 W PO = 0.25 W PO = 1.0 W 0.01% 20 TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER 100 1k 10k 20k RL = 4 Ω AV = –2 V/V BTL f = 15 kHz 1% f = 1 kHz 0.1% f = 20 Hz 0.01% 0.01 0.1 1 PO – Output Power – W f – Frequency – Hz Figure 9 Figure 10 TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER 10% THD+N –Total Harmonic Distortion + Noise THD+N –Total Harmonic Distortion + Noise 10% f = 15 kHz 1% f = 1 kHz 0.1% f = 20 Hz RL = 4 Ω AV = –6 V/V BTL 0.01% 0.01 0.1 1 PO – Output Power – W 10 f = 15 kHz 1% f = 1 kHz f = 20 Hz 0.1% RL = 4 Ω AV = –12 V/V BTL 0.01% 0.01 Figure 11 8 10 0.1 1 PO – Output Power – W Figure 12 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 10 TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 TYPICAL CHARACTERISTICS TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY 10% THD+N –Total Harmonic Distortion + Noise THD+N –Total Harmonic Distortion + Noise 10% f = 15 kHz 1% f = 1 kHz f = 20 Hz 0.1% RL = 4 Ω AV = –24 V/V BTL 0.01% 0.01 0.1 1 PO – Output Power – W 10 RL = 8 Ω AV = –2 V/V BTL 1% PO = 0.25 W 0.1% PO = 1.0 W 0.01% PO = 0.5 W 0.001% 20 100 Figure 13 TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER 10% 10% PO = 1 W RL = 8 Ω BTL AV = –24 V/V 1% AV = –12 V/V 0.1% AV = –2 V/V 0.01% AV = –6 V/V 100 1k 10k 20k THD+N –Total Harmonic Distortion + Noise THD+N –Total Harmonic Distortion + Noise 10k 20k Figure 14 TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY 0.001% 20 1k f – Frequency – Hz RL = 8 Ω AV = –2 V/V BTL 1% f = 15 kHz f = 1 kHz 0.1% f = 20 Hz 0.01% 0.001% 0.01 f – Frequency – Hz Figure 15 0.1 1 PO – Output Power – W 10 Figure 16 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 9 TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 TYPICAL CHARACTERISTICS TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER 10% RL = 8 Ω AV = –6 V/V BTL THD+N –Total Harmonic Distortion + Noise THD+N –Total Harmonic Distortion + Noise 10% f = 15 kHz 1% f = 1 kHz 0.1% f = 20 Hz 0.01% 0.01 TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER 0.1 1 PO – Output Power – W f = 15 kHz 1% f = 1 kHz 0.1% f = 20 Hz RL = 8 Ω AV = –12 V/V BTL 0.01% 0.01 10 0.1 1 PO – Output Power – W Figure 17 Figure 18 TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY 10% THD+N –Total Harmonic Distortion + Noise THD+N –Total Harmonic Distortion + Noise 10% f = 15 kHz 1% f = 1 kHz f = 20 Hz 0.1% RL = 8 Ω AV = –24 V/V BTL 0.01% 0.01 0.1 1 PO – Output Power – W 10 RL = 32 Ω AV = –1 V/V SE 1% PO = 25 mW 0.1% PO = 75 mW PO = 50 mW 0.01% 0.001% 20 100 1k f – Frequency – Hz Figure 19 10 10 Figure 20 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 10k 20k TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 TYPICAL CHARACTERISTICS TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY 10% RL = 32 Ω AV = –1 V/V SE THD+N –Total Harmonic Distortion + Noise THD+N –Total Harmonic Distortion + Noise 10% 1% f = 15 kHz 0.1% f = 1 kHz 0.01% f = 20 Hz 0.001% 0.01 0.1 PO – Output Power – W RL = 10 kΩ AV = –1 V/V SE 1% 0.1% VO = 1 VRMS 0.01% 0.001% 20 1 100 Figure 21 OUTPUT NOISE VOLTAGE vs BANDWIDTH 100 RL = 10 kΩ AV = –1 V/V SE 1% 0.1% f = 20 Hz f = 15 kHz 0.01% f = 1 kHz 0.6 0.8 1 1.2 1.4 1.6 80 70 AV = –24 V/V 60 50 AV = –12 V/V 40 30 AV = –6 V/V 20 10 0.001% 0.4 VDD = 5 V RL = 4Ω 90 V n – Output Noise Voltage – µ V THD+N –Total Harmonic Distortion + Noise 10% 0.2 10k 20k Figure 22 TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT VOLTAGE 0 1k f – Frequency – Hz 1.8 2 AV = –2 V/V 0 10 VO – Output Voltage – VRMS 100 1k 10k BW - Bandwidth - Hz Figure 23 Figure 24 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 11 TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 TYPICAL CHARACTERISTICS SUPPLY RIPPLE REJECTION RATIO vs FREQUENCY 0 RL = 8 Ω CB = 0.47 µF BTL –20 Supply Ripple Rejection Ratio – dB Supply Ripple Rejection Ratio – dB 0 –40 –60 AV = –24 V/V –80 –100 AV = –2 V/V –120 20 SUPPLY RIPPLE REJECTION RATIO vs FREQUENCY 100 1k –20 RL = 32 Ω CB = 0.47 µF SE –40 AV = –1 V/V –60 –80 –100 –120 20 10k 20k 100 1k f – Frequency – Hz f – Frequency – Hz Figure 25 Figure 26 CROSSTALK vs FREQUENCY CROSSTALK vs FREQUENCY 0 0 PO = 1 W RL = 8 Ω Av = –2 V/V BTL –20 –40 Crosstalk – dB Crosstalk – dB –20 10k 20k –60 –80 PO = 1 W RL = 8 Ω Av = –24 V/V BTL –40 –60 LEFT TO RIGHT –80 LEFT TO RIGHT –100 –100 RIGHT TO LEFT RIGHT TO LEFT –120 20 100 1k 10k 20k –120 20 f – Frequency – Hz 1k f – Frequency – Hz Figure 27 12 100 Figure 28 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 10k 20k TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 TYPICAL CHARACTERISTICS CROSSTALK vs FREQUENCY SHUTDOWN ATTENUATION vs FREQUENCY 0 VI = 1 VRMS –20 RL = 10 kΩ, SE –60 LEFT TO RIGHT –80 –40 Attenuation – dB –40 –60 RL = 32 Ω, SE –80 –100 RL = 8 Ω, BTL –100 RIGHT TO LEFT –120 20 100 1k –120 20 10k 20k 100 f – Frequency – Hz 1k 10k 20k f – Frequency – Hz Figure 29 Figure 30 SIGNAL-TO-NOISE RATIO vs BANDWIDTH 120 115 SNR – Signal-To-Noise Ratio – dB Crosstalk – dB –20 0 VO = 1 VRMS RL = 10 kΩ Av = –1 V/V SE PO = 1 W RL = 8 Ω BTL AV = –24 V/V AV = –12 V/V 110 105 100 AV = –2 V/V 95 AV = –6 V/V 90 85 80 20 100 1k 10k 20k BW – Bandwidth – Hz Figure 31 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 13 TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 TYPICAL CHARACTERISTICS CLOSED LOOP RESPONSE 180° 10 7.5 Gain Gain – dB 2.5 Phase 0° 0 –2.5 –5 RL = 8 Ω AV = –2 V/V BTL φ m – Phase Margin 90° 5 –90° –7.5 –10 10 100 1k 10k 100k 1M –180° 2M f – Frequency – Hz Figure 32 CLOSED LOOP RESPONSE 180° 30 25 Gain Gain – dB 15 Phase 0° 10 5 0 RL = 8 Ω AV = –6 V/V BTL φ m – Phase Margin 90° 20 –90° –5 –10 10 –180° 100 1k 10k 100k 1M f – Frequency – Hz Figure 33 14 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 2M TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 TYPICAL CHARACTERISTICS CLOSED LOOP RESPONSE 25 180° RL = 8 Ω AV = –12 V/V BTL 90° 20 Gain – dB Gain 15 Phase 0° 10 5 φ m – Phase Margin 30 –90° 0 –5 –10 10 –180° 100 1k 10k 100k 1M 2M f – Frequency – Hz Figure 34 CLOSED LOOP RESPONSE 30 25 180° RL = 8 Ω AV = –24 V/V BTL Gain Gain – dB 15 Phase 0° 10 5 φ m – Phase Margin 90° 20 –90° 0 –5 –10 10 –180° 100 1k 10k 100k 1M 2M f – Frequency – Hz Figure 35 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 15 TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 TYPICAL CHARACTERISTICS OUTPUT POWER vs LOAD RESISTANCE OUTPUT POWER vs LOAD RESISTANCE 3.5 3 AV = –1 V/V SE 1250 PO– Output Power – mW PO – Output Power – W 1500 AV = –2 V/V BTL 2.5 2 10% THD+N 1.5 1 1% THD+N 0.5 1000 750 10% THD+N 500 250 1% THD+N 0 0 8 16 24 32 40 48 RL – Load Resistance – Ω 56 0 64 0 48 16 24 32 40 RL – Load Resistance – Ω 8 Figure 36 POWER DISSIPATION vs OUTPUT POWER 1.8 0.4 3Ω 0.35 1.4 1.2 PD – Power Dissipation – W PD – Power Dissipation – W 1.6 4Ω 1 0.8 0.6 8Ω 0.4 f = 1 kHz BTL Each Channel 0.2 0.5 1.5 1 PO – Output Power – W 2 0.3 4Ω 0.25 0.2 0.15 8Ω 0.1 2.5 0 0 f = 1 kHz SE Each Channel 32 Ω 0.05 0.1 Figure 38 16 64 Figure 37 POWER DISSIPATION vs OUTPUT POWER 0 0 56 0.4 0.5 0.6 0.2 0.3 PO – Output Power – W Figure 39 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 0.7 0.8 TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 APPLICATION INFORMATION POWER DISSIPATION vs AMBIENT TEMPERATURE 7 ΘJA1 = 45.9°C/W ΘJA2 = 45.2°C/W ΘJA3 = 31.2°C/W ΘJA4 = 18.6°C/W ΘJA4 PD – Power Dissipation – W 6 5 4 ΘJA3 3 ΘJA1,2 2 1 0 –40 –20 20 40 60 80 100 120 140 160 0 TA – Ambient Temperature – °C Figure 40 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 17 TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 THERMAL INFORMATION The thermally enhanced PWP package is based on the 24-pin TSSOP, but includes a thermal pad (see Figure 41) to provide an effective thermal contact between the IC and the PWB. Traditionally, surface mount and power have been mutually exclusive terms. A variety of scaled-down TO-220-type packages have leads formed as gull wings to make them applicable for surface-mount applications. These packages, however, have only two shortcomings: they do not address the very low profile requirements (< 2 mm) of many of today’s advanced systems, and they do not offer a terminal-count high enough to accommodate increasing integration. On the other hand, traditional low-power surface-mount packages require power-dissipation derating that severely limits the usable range of many high-performance analog circuits. The PowerPAD package (thermally enhanced TSSOP) combines fine-pitch surface-mount technology with thermal performance comparable to much larger power packages. The PowerPAD package is designed to optimize the heat transfer to the PWB. Because of the very small size and limited mass of a TSSOP package, thermal enhancement is achieved by improving the thermal conduction paths that remove heat from the component. The thermal pad is formed using a patented lead-frame design and manufacturing technique to provide a direct connection to the heat-generating IC. When this pad is soldered or otherwise thermally coupled to an external heat dissipator, high power dissipation in the ultra-thin, fine-pitch, surface-mount package can be reliably achieved. DIE Side View (a) Thermal Pad DIE End View (b) Bottom View (c) Figure 41. Views of Thermally Enhanced PWP Package Figure 42 and Figure 43 are schematic diagrams of typical notebook computer application circuits. 18 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 Right CIRHP Head– 0.47 µF phone Input Signal 20 CIRLINE Right 0.47 µF Line Input Signal 23 RHPIN RLINEIN R MUX – + 8 ROUT+ 21 RIN CRIN 0.47 µF PC BEEP 14 Input Signal CPCB 0.47 µF COUTR 330 µF PC–BEEP – + PC– Beep ROUT– 16 VDD 1 kΩ 100 kΩ 2 3 GAIN0 15 SE/BTL Left CILHP Head– 0.47 µF 17 phone Input Signal 6 CILLINE Left 0.47 µF Line Input Signal 5 10 GAIN1 Gain/ MUX Control Depop Circuitry Power Management HP/LINE LHPIN LLINEIN PVDD 18 VDD 19 BYPASS SHUT– DOWN 11 GND L MUX See Note A VDD CSR 0.1 µF VDD CSR 0.1 µF 22 To System Control – + LOUT+ 4 – + LOUT– 9 CBYP 0.47 µF 1 kΩ 1,12, 13,24 COUTL 330 µF LIN CLIN 0.47 µF 100 kΩ NOTE A: A 0.1 µF ceramic capacitor should be placed as close as possible to the IC. For filtering lower–frequency noise signals, a larger electrolytic capacitor of 10 µF or greater should be placed near the audio power amplifier. Figure 42. Typical TPA0222 Application Circuit Using Single-Ended Inputs and Input MUX POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 19 TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 APPLICATION INFORMATION N/C 20 CIRIN– Right 0.47 µF Negative 23 Differential Input Signal CIRIN+ Right 0.47 µF Positive 8 Differential Input Signal PC BEEP 14 Input Signal C PCB 0.47 µF RHPIN RLINEIN – + 21 COUTR 330 µF PC–BEEP – + PC– Beep ROUT– 16 VDD 1 kΩ 100 kΩ GAIN0 15 SE/BTL GAIN1 Gain/ MUX Control Depop Circuitry Power Management HP/LINE N/C 6 LHPIN 5 LLINEIN CILIN– Left 0.47 µF Negative Differential Input Signal CILIN+ Left 0.47 µF Positive 10 Differential Input Signal ROUT+ RIN 2 3 17 R MUX PVDD 18 VDD 19 BYPASS SHUT– DOWN 11 GND L MUX See Note A VDD CSR 0.1 µF VDD CSR 0.1 µF 22 To System Control – + LOUT+ 4 – + LOUT– 9 CBYP 0.47 µF 1 kΩ 1,12, 13,24 COUTL 330 µF LIN 100 kΩ NOTE A: A 0.1 µF ceramic capacitor should be placed as close as possible to the IC. For filtering lower–frequency noise signals, a larger electrolytic capacitor of 10 µF or greater should be placed near the audio power amplifier. Figure 43. Typical TPA0222 Application Circuit Using Differential Inputs 20 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 APPLICATION INFORMATION gain setting via GAIN0 and GAIN1 inputs The gain of the TPA0222 is set by two input terminals, GAIN0 and GAIN1. Table 1. Gain Settings GAIN0 GAIN1 SE/BTL AV 0 0 0 –2 V/V 0 1 0 –6 V/V 1 0 0 –12 V/V 1 1 0 –24 V/V X X 1 –1 V/V The gains listed in Table 1 are realized by changing the taps on the input resistors inside the amplifier. This causes the input impedance, ZI, to be dependant on the gain setting. The actual gain settings are controlled by ratios of resistors, so the actual gain distribution from part-to-part is quite good. However, the input impedance will shift by 30% due to shifts in the actual resistance of the input impedance. For design purposes, the input network (discussed in the next section) should be designed assuming an input impedance of 10 kΩ, which is the absolute minimum input impedance of the TPA0222. At the higher gain settings, the input impedance could increase as high as 115 kΩ. input resistance Each gain setting is achieved by varying the input resistance of the amplifier, which can range from its smallest value to over 6 times that value. As a result, if a single capacitor is used in the input high pass filter, the –3 dB or cut-off frequency will also change by over 6 times. If an additional resistor is connected from the input pin of the amplifier to ground, as shown in the figure below, the variation of the cut-off frequency will be much reduced. ZF C IN Input Signal ZI R The typical input resistance at each gain setting is given in the table below: Av ZI –24 V/V 14 kΩ –12 V/V 26 kΩ –6 V/V 45.5 kΩ –2 V/V 91 kΩ POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 21 TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 APPLICATION INFORMATION The –3 dB frequency can be calculated using equation 1: ƒ –3 dB + ǒø Ǔ 1 2p C R R (1) I If the filter must be more accurate, the value of the capacitor should be increased while the value of the resistor to ground should be decreased. In addition, the order of the filter could be increased. input capacitor, CI In the typical application an input capacitor, CI, is required to allow the amplifier to bias the input signal to the proper dc level for optimum operation. In this case, CI and the input impedance of the amplifier, ZI, form a high-pass filter with the corner frequency determined in equation 2. –3 dB + 2 p Z1 C f c(highpass) (2) I I fc The value of CI is important to consider as it directly affects the bass (low frequency) performance of the circuit. Consider the example where ZI is 710 kΩ and the specification calls for a flat bass response down to 40 Hz. Equation 2 is reconfigured as equation 3. CI + 2 p Z1 fc (3) I In this example, CI is 5.6 nF so one would likely choose a value in the range of 5.6 nF to 1 µF. A further consideration for this capacitor is the leakage path from the input source through the input network (CI) and the feedback network to the load. This leakage current creates a dc offset voltage at the input to the amplifier that reduces useful headroom, especially in high gain applications. For this reason a low-leakage tantalum or ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor should face the amplifier input in most applications as the dc level there is held at VDD/2, which is likely higher than the source dc level. Note that it is important to confirm the capacitor polarity in the application. 22 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 APPLICATION INFORMATION power supply decoupling, CS The TPA0222 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling to ensure the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is achieved by using two capacitors of different types that target different types of noise on the power supply leads. For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance (ESR) ceramic capacitor, typically 0.1 µF placed as close as possible to the device VDD lead, works best. For filtering lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 µF or greater placed near the audio power amplifier is recommended. midrail bypass capacitor, CBYP The midrail bypass capacitor, CBYP, is the most critical capacitor and serves several important functions. During start-up or recovery from shutdown mode, CBYP determines the rate at which the amplifier starts up. The second function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This noise is from the midrail generation circuit internal to the amplifier, which appears as degraded PSRR and THD+N. Bypass capacitor, CBYP, values of 0.47 µF to 1 µF ceramic or tantalum low-ESR capacitors are recommended for the best THD and noise performance. output coupling capacitor, CC In the typical single-supply SE configuration, an output coupling capacitor (CC) is required to block the dc bias at the output of the amplifier thus preventing dc currents in the load. As with the input coupling capacitor, the output coupling capacitor and impedance of the load form a high-pass filter governed by equation 4. –3 dB f c(high) + 2 p R1 C (4) L C fc The main disadvantage, from a performance standpoint, is the load impedances are typically small, which drives the low-frequency corner higher, degrading the bass response. Large values of CC are required to pass low frequencies into the load. Consider the example where a CC of 330 µF is chosen and loads vary from 3 Ω, 4 Ω, 8 Ω, 32 Ω, 10 kΩ, to 47 kΩ. Table 2 summarizes the frequency response characteristics of each configuration. POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 23 TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 APPLICATION INFORMATION Table 2. Common Load Impedances Vs Low Frequency Output Characteristics in SE Mode RL CC 330 µF Lowest Frequency 3Ω 4Ω 330 µF 120 Hz 60 Hz 161 Hz 8Ω 330 µF 32 Ω 330 µF 15 Hz 10,000 Ω 330 µF 0.05 Hz 47,000 Ω 330 µF 0.01 Hz As Table 2 indicates, most of the bass response is attenuated into a 4-Ω load, an 8-Ω load is adequate, headphone response is good, and drive into line level inputs (a home stereo for example) is exceptional. using low-ESR capacitors Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal) capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this resistance the more the real capacitor behaves like an ideal capacitor. bridged-tied load versus single-ended mode Figure 44 shows a Class-AB audio power amplifier (APA) in a BTL configuration. The TPA0222 BTL amplifier consists of two Class-AB amplifiers driving both ends of the load. There are several potential benefits to this differential drive configuration, but initially consider power to the load. The differential drive to the speaker means that as one side is slewing up, the other side is slewing down, and vice versa. This in effect doubles the voltage swing on the load as compared to a ground referenced load. Plugging 2 × VO(PP) into the power equation, where voltage is squared, yields 4× the output power from the same supply rail and load impedance (see equation 5). V (rms) + O(PP) 2 Ǹ2 Power + V 24 V (rms) (5) 2 RL POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 APPLICATION INFORMATION VDD VO(PP) 2x VO(PP) RL VDD –VO(PP) Figure 44. Bridge-Tied Load Configuration In a typical computer sound channel operating at 5 V, bridging raises the power into an 8-Ω speaker from a singled-ended (SE, ground reference) limit of 250 mW to 1 W. In sound power that is a 6-dB improvement — which is loudness that can be heard. In addition to increased power there are frequency response concerns. Consider the single-supply SE configuration shown in Figure 45. A coupling capacitor is required to block the dc offset voltage from reaching the load. These capacitors can be quite large (approximately 33 µF to 1000 µF) so they tend to be expensive, heavy, occupy valuable PCB area, and have the additional drawback of limiting low-frequency performance of the system. This frequency limiting effect is due to the high pass filter network created with the speaker impedance and the coupling capacitance and is calculated with equation 6. fc + 2 p R1 C (6) L C For example, a 68-µF capacitor with an 8-Ω speaker would attenuate low frequencies below 293 Hz. The BTL configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency performance is then limited only by the input network and speaker response. Cost and PCB space are also minimized by eliminating the bulky coupling capacitor. VDD –3 dB VO(PP) CC RL VO(PP) fc Figure 45. Single-Ended Configuration and Frequency Response POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 25 TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 APPLICATION INFORMATION Increasing power to the load does carry a penalty of increased internal power dissipation. The increased dissipation is understandable considering that the BTL configuration produces 4× the output power of the SE configuration. Internal dissipation versus output power is discussed further in the crest factor and thermal considerations section. single-ended operation In SE mode (see Figure 44 and Figure 45), the load is driven from the primary amplifier output for each channel (OUT+, terminals 21 and 4). The amplifier switches single-ended operation when the SE/BTL terminal is held high. This puts the negative outputs in a high-impedance state, and reduces the amplifier’s gain to 1 V/V. input MUX operation The input MUX allows two separate inputs to be applied to the amplifier. This allows the designer to choose which input is active independent of the state of the SE/BTL terminal. When the HP/LINE terminal is held high, the headphone inputs are active. When the HP/LINE terminal is held low, the line BTL inputs are active. BTL amplifier efficiency Class-AB amplifiers are notoriously inefficient. The primary cause of these inefficiencies is voltage drop across the output stage transistors. There are two components of the internal voltage drop. One is the headroom or dc voltage drop that varies inversely to output power. The second component is due to the sinewave nature of the output. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from VDD. The internal voltage drop multiplied by the RMS value of the supply current, IDDrms, determines the internal power dissipation of the amplifier. An easy-to-use equation to calculate efficiency starts out as being equal to the ratio of power from the power supply to the power delivered to the load. To accurately calculate the RMS and average values of power in the load and in the amplifier, the current and voltage waveform shapes must first be understood (see Figure 46). IDD VO IDD(avg) V(LRMS) Figure 46. Voltage and Current Waveforms for BTL Amplifiers Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very different between SE and BTL configurations. In an SE application the current waveform is a half-wave rectified shape, whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different. Keep in mind that for most of the waveform both the push and pull transistors are not on at the same time, which supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform. The following equations are the basis for calculating amplifier efficiency. 26 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 APPLICATION INFORMATION Efficiency of a BTL amplifier + PP L (7) SUP Where: PL + VLRrms L and P SUP 2 , and V LRMS + VDD IDDavg + VǸ2P , 2 therefore, P L + ŕ 1 I DDavg and p p 0 + 2VRP VP RL L sin(t) dt + 1p VP RL [cos(t)] p 0 + p2VRP L Therefore, P SUP + 2 VpDDR VP L substituting PL and PSUP into equation 7, 2 Efficiency of a BTL amplifier Where: VP +2 V h BTL +p PL = Power devilered to load PSUP = Power drawn from power supply VLRMS = RMS voltage on BTL load RL = Load resistance VP = Peak voltage on BTL load IDDavg = Average current drawn from the power supply VDD = Power supply voltage ηBTL = Efficiency of a BTL amplifier p VP DD V P +4V p RL + Ǹ2 PL RL Therefore, VP 2 RL DD Ǹ 2 PL RL 4 V DD (8) Table 3 employs equation 8 to calculate efficiencies for four different output power levels. Note that the efficiency of the amplifier is quite low for lower power levels and rises sharply as power to the load is increased resulting in a nearly flat internal power dissipation over the normal operating range. Note that the internal dissipation at full output power is less than in the half power range. Calculating the efficiency for a specific system is the key to proper power supply design. For a stereo 1-W audio system with 8-Ω loads and a 5-V supply, the maximum draw on the power supply is almost 3.25 W. Table 3. Efficiency Vs Output Power in 5-V 8-Ω BTL Systems Output Power (W) Efficiency (%) Peak Voltage (V) Internal Dissipation (W) 0.25 31.4 2.00 0.55 0.50 44.4 2.83 0.62 1.00 62.8 4.00 4.47† 0.59 1.25 70.2 † High peak voltages cause the THD to increase. 0.53 A final point to remember about Class-AB amplifiers (either SE or BTL) is how to manipulate the terms in the efficiency equation to utmost advantage when possible. Note that in equation 8, VDD is in the denominator. This indicates that as VDD goes down, efficiency goes up. POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 27 TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 APPLICATION INFORMATION crest factor and thermal considerations Class-AB power amplifiers dissipate a significant amount of heat in the package under normal operating conditions. A typical music CD requires 12 dB to 15 dB of dynamic range, or headroom above the average power output, to pass the loudest portions of the signal without distortion. In other words, music typically has a crest factor between 12 dB and 15 dB. When determining the optimal ambient operating temperature, the internal dissipated power at the average output power level must be used. From the TPA0222 data sheet, one can see that when the TPA0222 is operating from a 5-V supply into a 3-Ω speaker 4-W peaks are available. Converting watts to dB: P dB + 10 Log PPW + 10 Log 41 WW + 6 dB (9) ref Subtracting the headroom restriction to obtain the average listening level without distortion yields: 6 dB – 15 dB = –9 dB (15 dB crest factor) 6 dB – 12 dB = –6 dB (12 dB crest factor) 6 dB – 9 dB = –3 dB (9 dB crest factor) 6 dB – 6 dB = 0 dB (6 dB crest factor) 6 dB – 3 dB = 3 dB (3 dB crest factor) Converting dB back into watts: PW + 10PdBń10 Pref + 63 mW (18 dB crest factor) + 125 mW (15 dB crest factor) + 250 mW (9 dB crest factor) + 500 mW (6 dB crest factor) + 1000 mW (3 dB crest factor) + 2000 mW (15 dB crest factor) (10) This is valuable information to consider when attempting to estimate the heat dissipation requirements for the amplifier system. Comparing the absolute worst case, which is 2 W of continuous power output with a 3 dB crest factor, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings for the system. Using the power dissipation curves for a 5-V, 3-Ω system, the internal dissipation in the TPA0222 and maximum ambient temperatures is shown in Table 4. Table 4. TPA0222 Power Rating, 5-V, 3-Ω, Stereo 28 PEAK OUTPUT POWER (W) AVERAGE OUTPUT POWER POWER DISSIPATION (W/Channel) MAXIMUM AMBIENT TEMPERATURE 4 2 W (3 dB) 1.7 – 3°C 4 1000 mW (6 dB) 1.6 6°C 4 500 mW (9 dB) 1.4 24°C 4 250 mW (12 dB) 1.1 51°C 4 125 mW (15 dB) 0.8 78°C 4 63 mW (18 dB) 0.6 96°C POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 APPLICATION INFORMATION crest factor and thermal considerations (continued) Table 5. TPA0222 Power Rating, 5-V, 8-Ω, Stereo PEAK OUTPUT POWER AVERAGE OUTPUT POWER POWER DISSIPATION (W/Channel) MAXIMUM AMBIENT TEMPERATURE 2.5 W 1250 mW (3 dB crest factor) 0.55 100°C 2.5 W 1000 mW (4 dB crest factor) 0.62 94°C 2.5 W 500 mW (7 dB crest factor) 0.59 97°C 2.5 W 250 mW (10 dB crest factor) 0.53 102°C The maximum dissipated power, PDmax, is reached at a much lower output power level for an 8-Ω load than for a 3-Ω load. As a result, this simple formula for calculating PDmax may be used for an 8-Ω application: 2V 2 P Dmax + p2RDD (11) L However, in the case of a 3-Ω load, the PDmax occurs at a point well above the normal operating power level. The amplifier may therefore be operated at a higher ambient temperature than required by the PDmax formula for a 3 Ω load. The maximum ambient temperature depends on the heat sinking ability of the PCB system. The derating factor for the PWP package is shown in the dissipation rating table (see page 4). Converting this to ΘJA: Θ JA 1 + Derating1 Factor + 0.022 + 45°CńW (12) To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are per channel so the dissipated power needs to be doubled for two channel operation. Given ΘJA, the maximum allowable junction temperature, and the total internal dissipation, the maximum ambient temperature can be calculated with the following equation. The maximum recommended junction temperature for the TPA0222 is 150°C. The internal dissipation figures are taken from the Power Dissipation vs Output Power graphs. T A Max + TJ Max * ΘJA PD + 150 * 45 (0.6 2) + 96°C (15 dB crest factor) (13) NOTE: Internal dissipation of 0.6 W is estimated for a 2-W system with 15 dB crest factor per channel. TableS 4 and 5 show that for some applications no airflow is required to keep junction temperatures in the specified range. The TPA0222 is designed with thermal protection that turns the device off when the junction temperature surpasses 150°C to prevent damage to the IC. Tables 4 and 5 were calculated for maximum listening volume without distortion. When the output level is reduced the numbers in the table change significantly. Also, using 8-Ω speakers dramatically increases the thermal performance by increasing amplifier efficiency. POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 29 TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 APPLICATION INFORMATION SE/BTL operation The ability of the TPA0222 to easily switch between BTL and SE modes is one of its most important cost saving features. This feature eliminates the requirement for an additional headphone amplifier in applications where internal stereo speakers are driven in BTL mode but external headphone or speakers must be accommodated. Internal to the TPA0222, two separate amplifiers drive OUT+ and OUT–. The SE/BTL input (terminal 15) controls the operation of the follower amplifier that drives LOUT– and ROUT– (terminals 9 and 16). When SE/BTL is held low, the amplifier is on and the TPA0222 is in the BTL mode. When SE/BTL is held high, the OUT– amplifiers are in a high output impedance state, which configures the TPA0222 as an SE driver from LOUT+ and ROUT+ (terminals 4 and 21). IDD is reduced by approximately one-half in SE mode. Control of the SE/BTL input can be from a logic-level CMOS source or, more typically, from a resistor divider network as shown in Figure 47. 20 23 RHPIN RLINEIN R MUX – + 8 ROUT+ 21 RIN VDD – + ROUT– 16 100 kΩ SE/BTL 15 COUTR 330 µF 1 kΩ 100 kΩ Figure 47. TPA0222 Resistor Divider Network Circuit Using a readily available 1/8-in. (3.5 mm) stereo headphone jack, the control switch is closed when no plug is inserted. When closed the 100-kΩ/1-kΩ divider pulls the SE/BTL input low. When a plug is inserted, the 1-kΩ resistor is disconnected and the SE/BTL input is pulled high. When the input goes high, the OUT– amplifier is shut down causing the speaker to mute (virtually open-circuits the speaker). The OUT+ amplifier then drives through the output capacitor (CO) into the headphone jack. 30 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 APPLICATION INFORMATION PC BEEP operation The PC BEEP input allows a system beep to be sent directly from a computer through the amplifier to the speakers with few external components. The input is activated automatically. When the PC BEEP input is active, both of the LINEIN and HPIN inputs are deselected and both the left and right channels are driven in BTL mode with the signal from PC BEEP. The gain from the PC BEEP input to the speakers is fixed at 0.3 V/V and is independent of the volume setting. When the PC BEEP input is deselected, the amplifier will return to the previous operating mode and volume setting. Furthermore, if the amplifier is in shutdown mode, activating PC BEEP will take the device out of shutdown and output the PC BEEP signal, then return the amplifier to shutdown mode. The preferred input signal is a square wave or pulse train with an amplitude of 1 Vpp or greater. To be accurately detected, the signal must have a minimum of 1 Vpp amplitude, rise and fall times of less than 0.1 µs and a minimum of 8 rising edges. When the signal is no longer detected, the amplifier will return to its previous operating mode and volume setting. If it is desired to ac-couple the PC BEEP input, the value of the coupling capacitor should be chosen to satisfy equation 14: C PCB w 2p ƒ 1 PCB W (14) (100 k ) The PC BEEP input can also be dc-coupled to avoid using this coupling capacitor. The pin normally sits at midrail when no signal is present. shutdown modes The TPA0222 employs a shutdown mode of operation designed to reduce supply current, IDD, to the absolute minimum level during periods of nonuse for battery-power conservation. The SHUTDOWN input terminal should be held high during normal operation when the amplifier is in use. Pulling SHUTDOWN low causes the outputs to mute and the amplifier to enter a low-current state, IDD = 150 µA. SHUTDOWN should never be left unconnected because amplifier operation would be unpredictable. Table 6. HP/LINE, SE/BTL, and Shutdown Functions INPUTS† HP/LINE SE/BTL AMPLIFIER STATE SHUTDOWN INPUT OUTPUT X X Low X Mute Low Low High Line BTL Low High High Line SE High Low High HP BTL High High High † Inputs should never be left unconnected. X = do not care HP SE POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 31 TPA0222 STEREO 2-W AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL SLOS285 – NOVEMBER 1999 MECHANICAL DATA PWP (R-PDSO-G**) PowerPAD PLASTIC SMALL-OUTLINE PACKAGE 20-PIN SHOWN 0,30 0,19 0,65 20 0,10 M 11 Thermal Pad (See Note D) 4,50 4,30 0,15 NOM 6,60 6,20 Gage Plane 1 10 0,25 A 0°– 8° 0,75 0,50 Seating Plane 0,15 0,05 1,20 MAX PINS ** 0,10 14 16 20 24 28 A MAX 5,10 5,10 6,60 7,90 9,80 A MIN 4,90 4,90 6,40 7,70 9,60 DIM 4073225/E 03/97 NOTES: A. B. C. D. All linear dimensions are in millimeters. This drawing is subject to change without notice. Body dimensions do not include mold flash or protrusions. The package thermal performance may be enhanced by bonding the thermal pad to an external thermal plane. This pad is electrically and thermally connected to the backside of the die and terminals 1, 12, 13, and 24. The dimensions of the thermal pad are 2.40 mm × 4.70 mm (maximum). The pad is centered on the bottom of the package. E. Falls within JEDEC MO-153 PowerPAD is a trademark of Texas Instruments Incorporated. 32 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 IMPORTANT NOTICE Texas Instruments and its subsidiaries (TI) reserve the right to make changes to their products or to discontinue any product or service without notice, and advise customers to obtain the latest version of relevant information to verify, before placing orders, that information being relied on is current and complete. All products are sold subject to the terms and conditions of sale supplied at the time of order acknowledgement, including those pertaining to warranty, patent infringement, and limitation of liability. TI warrants performance of its semiconductor products to the specifications applicable at the time of sale in accordance with TI’s standard warranty. Testing and other quality control techniques are utilized to the extent TI deems necessary to support this warranty. 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