MIC285 512 7 0V 2A Syn nchronous Buck Reg gulator Gen neral Desc cription Featu ures The MIC28512 is i a synchro onous step-d down switching regullator with inte ernal power sw witches capable of providing up to o 2A output current from a wide inpu ut supply range from 4.6V to 70V. The output voltage v is adju ustable down to 0.8V with a gua aranteed accuracy of ±1% %. A consta ant switcching frequen ncy can be prrogrammed from f 200kHz to 680kkHz. The Hy yper Speed Control™ and a HyperLig ght Load d® architecturres of the MIC28512 allo ow for high VIN (low VOUT) opera ation and ultra-fast trans sient response while e reducing the required d output capacitance and providing very goo od light load efficiency. e The MIC28512 offfers a full su uite of protecttion features to ensure protection ns under fault conditions. These include unde er-voltage loc ckout to ensu ure proper operation o und der powe er sag condittions, internal soft-start to o reduce inrush curre ent, fold-back current limit,, “hiccup” mo ode short-circ cuit prote ection, and the ermal shutdow wn. Datasheets and support s docu umentation arre available on Micre el’s web site at: a www.micre el.com. 4.6V V to 70V operrating input vo oltage supply Up tto 2A output ccurrent Integ grated high-sside and low-sside N-channe el MOSFETs Hyp perLight Load (MIC28512-1) and H Hyper Speed d Con ntrol (MIC28512-2) architeccture Ena able input and d power good (PGOOD) ou utput grammable current limit an nd foldback ““hiccup” mode e Prog shorrt-circuit prote ection Builtt-in 5V regula ator for single-supply opera ation Adju ustable 200kH Hz to 680kHz switching fre equency Fixe ed 5ms soft-sttart Interrnal compenssation and the ermal shutdow wn The rmally enhanced 24-pin n 3mm × 4 4mm FCQFN N packkage Juncction tempera ature range off –40°C to +125°C Appllications Indu ustrial power ssupplies Disttributed supply regulation wer supplies Bas e station pow Wal l transformer regulation h-voltage sing gle board systems High Typ pical Application Efficiency y (VIN =12V) vs. Output Currrent MIC28512-1 100 0 5.0V V 3.3V V 90 0 EFFICIENCY (%) 80 0 70 0 60 0 50 0 40 0 30 0 FSW = 300kHz 20 0 10 0 0.01 0.1 1 10 OUTPUT CURRENT (A) Hype er Speed Control is a trademark of o Micrel, Inc. Hype erLight Load is a registered trade emark of Micrel, Inc. I Micrel Inc. • 2180 Fortune Driv ve • San Jose, CA C 95131 • USA • tel +1 (408) 94 44-0800 • fax + 1 (408) 474-1000 0 • http://www.m micrel.com Marcch 25, 2015 Revision 1.2 Micre el, Inc. MIC28512 2 Ord dering Info ormation Arch hitecture Pa ackage(1) Junctiion Temperatu ure Range Lead Finish MIC C28512-1YFL HyperL Light Load 24-Pin 3mm m × 4mm FCQ FN –40°C to +125 5°C Pb-Free MIC C28512-2YFL Hyper Sp peed Control 24-Pin 3mm m × 4mm FCQ FN –40°C to +125 5°C Pb-Free Partt Number Note: 1. FC CQFN is a lead-ffree package. Pb b-free lead finish is Matte Tin. Pin Configuration 24-Pin 3mm m × 4mm FCQF FN (FL) (Top View) Marcch 25, 2015 2 Revision 1.2 2 Micrel, Inc. MIC28512 Pin Description Pin Number Pin Name 1 DL 2 PGND 3 DH 4, 7, 8, 9, 25 (25 is ePad) PVIN Power Input Voltage. The PVIN pins supply power to the internal power switch. Connect all PVIN pins together and locally bypass with ceramic capacitors. The positive terminal of the input capacitor should be placed as close as possible to the PVIN pins; the negative terminal of the input capacitor should be placed as close as possible to the PGND pins 10,11, 22, 23, and 26. 5 LX The LX pin is the return path for the high-side driver circuit. Connect the negative terminal of the bootstrap capacitor directly to this pin. Also connect this pin to the SW pins 12, 21, and 27, with a low impedance path. The controller monitors voltages on this pin and the PGND for zero current detection. 6 BST Bootstrap Pin. This pin provides bootstrap supply for the high-side gate driver circuit. Connect a 0.1µF capacitor and an optional resistor in series from the LX (Pin 5) to the BST. 10, 11, 22, 23, 26 (26 is ePad) PGND Power Ground. These pins are connected to the source of the low-side MOSFET. They are the return path for the step-down regulator power stage and should be tied together. The negative terminal of the input decoupling capacitor should be placed as close as possible to these pins. 12, 21, 27 (27 is ePad) SW 13 AGND 14 FB Feedback Input. The FB pin sets the regulated output voltage relative to the internal reference. This pin is connected to a resistor divider from the regulated output such that the FB pin is at 0.8V when the output is at the desired voltage. 15 PGOOD Power Good. The power good output is an open drain output requiring an external pull-up resistor to external bias. This pin is a high impedance open circuit when the voltage at FB pin is higher than 90% of the feedback reference voltage (typically 0.8V). 16 EN Enable Input. The EN pin enables the regulator. When the pin is pulled below the threshold, the regulator will shut down to an ultra-low current state. A precise threshold voltage allows the pin to operate as an accurate UVLO. Do not tie EN to VDD. 17 VIN Supply voltage for the internal LDO. The VIN operating voltage range is from 4.6V to 70V. A ceramic capacitor from VIN to AGND is required for decoupling. The decoupling capacitor should be placed as close as possible to the supply pin. 18 ILIM Current Limit Setting. Connect a resistor from this pin to the SW pin node to allow for accurate current limit sensing programming of the internal low-side power MOSFET. 19 VDD Internal +5V Linear Regulator. VDD is the internal supply bus for the IC. Connect to an external 1µF bypass capacitor. When VIN is less than 5.5V, this regulator operates in drop-out mode. Connect VDD to VIN. 20 PVDD A 5V supply input for the low-side N-channel MOSFET driver circuit that can be tied to VDD externally. A 1μF ceramic capacitor from PVDD to PGND is recommended for decoupling. 24 FREQ Switching Frequency Adjust pin. Connect this pin to VIN to operate at 680kHz. Place a resistor divider network from VIN to the FREQ pin to program the switching frequency. March 25, 2015 Pin Function Low-side gate drive: Internal low-side power MOSFET gate connection. This pin must be left unconnected or floating. PGND is the return path for the low-side driver circuit. Connect to the source of low-side MOSFET’s (PGND, Pin 10, 11, 22, 23, and 27) through a low impedance path. High-side gate drive: Internal high-side power MOSFET gate connection. This pin must be left unconnected, or floating. Switch Node. The SW pins are the internal power switch outputs. These pins should be tied together and connected to the output inductor. Analog Ground. Signal ground for VDD and the control circuitry. The signal ground return path should be separate from the power ground (PGND) return path. 3 Revision 1.2 Micrel, Inc. MIC28512 Absolute Maximum Ratings(2) Operating Ratings(3) PVIN, VIN to PGND ........................................ 0.3V to 76V VDD, PVDD to PGND ....................................... 0.3V to 6V VBST to VSW, VLX ................................................. 0.3V to 6V VBST to PGND …………………..…………0.3V to (VIN +6V) VSW,VLX to PGND ............................... 0.3V to (VIN +0.3V) VFREQ, VILIM, VEN to AGND .................. 0.3V to (VIN +0.3V) VFB, VPG, to AGND ............................. 0.3V to (VDD+ +0.3V) PGND to AGND ............................................ 0.3V to +0.3V Junction Temperature (TJ) ....................................... +150C Storage Temperature (TS) ......................... 65C to 150C Lead Temperature (soldering, 10s) ............................ 300C ESD HBM Rating(4)...................................................... 1.5kV ESD MM Rating(4) ......................................................... 150V Supply Voltage (PVIN, VIN)............................. +4.6V to +70V Enable Input (VEN) .................................................. 0V to VIN VSW, VFEQ, VILIM, VEN ............................................... 0V to VIN Junction Temperature (TJ) ........................ –40°C to +125°C Junction Thermal Resistance FQFN (JA) ......................................................... 30°C/W Electrical Characteristics(5) VIN = 12V; TA = 25°C, bold values indicate –40°C≤ TA ≤ +125°C, unless noted. Parameter Condition Min. Typ. Max. Units 70 V Power Supply Input Input Voltage Range (PVIN, VIN) Quiescent Supply Current Shutdown Supply Current 4.6 VFB = 1.5V (MIC28512-1) 0.4 0.75 VFB = 1.5V (MIC28512-2) 0.7 1.5 SW = unconnected, VEN = 0V 0.1 10 µA mA VDD Supply VDD Output Voltage VIN =7V to 70V, IVDD = 10mA 4.8 5.2 5.4 V VDD UVLO Threshold VVDD rising 3.8 4.2 4.6 V VDD UVLO Hysteresis 400 Load Regulation @40mA mV 0.6 2 4 % 25ºC (±1.0%) 0.792 0.8 0.808 -40°C ≤ TJ ≤ 125°C (±2%) 0.784 0.8 0.816 5 500 Reference Feedback Reference Voltage FB Bias Current VFB = 0.8V V nA Enable Control EN Logic Level High V 1.8 EN Logic Level Low 0.6 EN Hysteresis EN Bias Current 200 VEN = 12V 23 V mV 40 µA Notes: 2. Exceeding the absolute maximum ratings may damage the device. 3. The device is not guaranteed to function outside its operating ratings. 4. Devices are ESD sensitive. Handling precautions are recommended. Human body model, 1.5kΩ in series with 100pF. 5. Specification for packaged product only. March 25, 2015 4 Revision 1.2 Micrel, Inc. MIC28512 Electrical Characteristics(5) (Continued) VIN = 12V; TA = 25°C, bold values indicate –40°C≤ TA ≤ +125°C, unless noted. Parameter Condition Min. Typ. Max. VFREQ = VIN 450 680 800 Units Oscillator Switching Frequency VFREQ = 50%VIN 340 Maximum Duty Cycle Minimum Duty Cycle VFB>0.8V Minimum Off-time 110 kHz 85 % 0 % 200 270 ns Internal MOSFETs High-Side NMOS On-Resistance 77 m Low-Side NMOS On-Resistance 43 m Short Circuit Protection Current-Limit Threshold VFB = 0.79V -30 -14 0 mV Short-Circuit Threshold VFB = 0V -24 -7 8 mV Current-Limit Source Current VFB = 0.79V 50 70 90 µA Short-Circuit Source Current VFB = 0V 25 36 43 µA 50 µA 95 %VOUT Leakage SW, BST Leakage Current Power Good Power Good Threshold Voltage Sweep VFB from Low to High Power Good Hysteresis Sweep VFB from High to Low 6 %VOUT Power Good Delay Time Sweep VFB from Low to High 100 µs Power Good Low Voltage VFB < 90% x VNOM, IPGOOD = 1mA 70 TJ Rising 160 °C 15 °C 5 ms 85 90 200 mV Thermal Protection Over-Temperature Shutdown Over-Temperature Shutdown Hysteresis Soft Start Soft Start Time March 25, 2015 5 Revision 1.2 Micrel, Inc. MIC28512 Typical Characteristics VIN Shutdown Current vs. Input Voltage VIN Operating Supply Current vs. Input Voltage MIC28512-1 4.0 3.0 2.0 1.0 ENABLE THRESHOLD (V) Rising VOUT = 5V IOUT = 0A FSW = 300kHz SHUTDOWN CURRENT (mA) 30.0 25.0 20.0 15.0 10.0 5.0 5 10 15 20 25 30 35 40 45 50 55 60 65 70 18.0 VIN UVLO Threshold vs. Temperature 6.0 44.0 57.0 CURRENT LIMIT (A) 5.0 4.5 4.3 Falling 3.9 3.7 INPUT VOLTAGE (V) Feedback Voltage vs. Temperature 0.812 4.0 3.0 2.0 0.0 3.3 0 25 50 75 100 -50 125 -25 TEMPERATURE (°C) Output Voltage vs. Input Voltage 5.0 4.9 VOUT = 5V IOUT = 1A FSW = 300kHz 4.8 ENABLE THRESHOLD (V) 5.1 0 25 50 75 100 5 10 15 20 25 30 35 40 45 50 55 60 65 70 INPUT VOLTAGE (V) March 25, 2015 0.800 -50 1.3 1.2 Falling 1.0 0.9 0.8 0.7 VIN = 12V VDD = 5V 0.6 -50 -25 0 25 50 75 TEMPERATURE (°C) 6 25 50 75 100 125 600 Rising 1.1 0 Switching Frequency vs. Output Current 1.5 1.4 -25 TEMPERATURE (°C) 0.5 4.7 0.804 0.792 125 Enable Threshold vs. Temperature 1.6 5.2 0.808 TEMPERATURE (°C) 5.3 VIN = 12V VOUT = 5.0V IOUT = 0A 0.796 1.0 VIN =12V IOUT = 0A -25 Hyst 5 10 15 20 25 30 35 40 45 50 55 60 65 70 70.0 FEEDBACK VOLTAGE (V) 4.7 VIN THRESHOLD (V) 31.0 VIN = 12V VOUT = 5.0V FSW = 300kHz Rising -50 0.30 Output Peak Current Limit vs. Temperature 4.9 3.5 0.60 INPUT VOLTAGE (V) INPUT VOLTAGE (V) 4.1 Falling 0.90 0.00 0.0 0.0 1.20 5.0 SWITCHING FREQUENCY (kHz) SUPPLY CURRENT (mA) 1.50 35.0 5.0 OUTPUT VOLTAGE (V) Enable Threshold vs. Input Voltage 100 125 550 500 450 400 350 300 250 200 VIN = 12V VOUT = 5V 150 100 0.0 0.5 1.0 1.5 2.0 OUTPUT CURRENT (A) Revision 1.2 Micrel, Inc. MIC28512 Typical Characteristics (Continued) Output Voltage vs. Output Current Efficiency (VIN =12V) vs. Output Current MIC28512-1 5.2 100 100 5.0V 3.3V 4.9 70 60 50 40 0.0 0.5 1.0 1.5 10 0.01 OUTPUT CURRENT (A) 0.1 1 20.0 SUPPLY CURRENT (mA) 5.0V 3.3V 70 60 50 40 30 FSW = 300kHz 16.0 14.0 12.0 10.0 8.0 6.0 4.0 VOUT = 5V IOUT = 0A FSW = 300kHz 2.0 1 10 0.804 0.800 0.796 0.792 -50 100 5.2 100 5.0V 3.3V 5.0 70 60 50 40 30 4.9 4.8 March 25, 2015 75 100 125 2.0 10 0.01 Efficiency (VIN =24V) vs. Output Current MIC28512-2 5.0V 3.3V 70 60 50 40 30 FSW = 300kHz 20 VIN = 12V VOUT = 5V OUTPUT CURRENT (A) 50 80 EFFICIENCY (%) EFFICIENCY (%) 5.1 1.5 25 90 80 1.0 0 TEMPERATURE (°C) Efficiency (VIN =12V) vs. Output Current MIC28512-2 90 0.5 -25 INPUT VOLTAGE (V) Output Voltage vs. Output Current MIC28512-2 0.0 10 VIN = 12V VOUT = 5.0V IOUT = 0A 5 10 15 20 25 30 35 40 45 50 55 60 65 70 OUTPUT CURRENT (A) 1 0.808 0.0 0.1 0.1 Feedback Voltage vs. Temperature MIC28512-2 0.812 18.0 10 0.01 FSW = 300kHz OUTPUT CURRENT (A) VIN Operating Supply Current vs. Input Voltage MIC28512-2 90 20 40 OUTPUT CURRENT (A) Efficiency (VIN =48V) vs. Output Current MIC28512-1 80 50 10 0.01 10 FEEDBACK VOLTAGE (V) 100 60 20 FSW = 300kHz 20 2.0 70 30 30 VIN = 12V VOUT = 5V 5.0V 3.3V 80 EFFICIENCY (%) 5.0 4.8 EFFICIENCY (%) 90 80 5.1 EFFICIENCY (%) OUTPUT VOLTAGE (V) 90 OUTPUT VOLTAGE (V) Efficiency (VIN =24V) vs. Output Current MIC28512-1 0.1 1 OUTPUT CURRENT (A) 7 FSW = 300kHz 20 10 10 0.01 0.1 1 10 OUTPUT CURRENT (A) Revision 1.2 Micrel, Inc. MIC28512 Typical Characteristics (Continued) 1.0 90 5.0V 3.3V 70 60 50 40 30 FSW = 300kHz 20 10 0.01 VIN =12V fSW = 300kHz 0.8 0.6 0.4 5.0V 3.3V 0.2 0.0 0.1 1 0.5 1 1.5 5.0V 3.3V 0.6 0.4 0.2 2 0 0.5 OUTPUT CURRENT (A) Vin =48V =24V fSW = 300kHz 5.0V 3.3V 1.2 0.8 1.5 2 24V Input Thermal Derating MIC28512-1 2.5 2.0 5.0V 3.3V 1.5 Vin =24V =12V fSW = 300kHz Tjmax =125°C Θja = 30°C/W 40°C/W 1.0 1 OUTPUT CURRENT (A) 12V Input Thermal Derating 2.5 2.4 1.6 0.8 OUTPUT CURRENT (A) IC Power Dissipation vs. Output Current 2.0 VIN =24V Vin =24V fSW = 300kHz 1.0 0.0 0 10 OUTPUT CURRENT (A) OUTPUT CURRENT (A) EFFICIENCY (%) 80 1.2 IC POWER DISSIPATION (W) IC POWER DISSIPATION (W) 100 IC POWER DISSIPATION (W) IC Power Dissipation vs. Output Current IC Power Dissipation vs. Output Current Efficiency (VIN =48V) vs. Output Current MIC28512-2 0.5 2.0 5.0V 3.3V 1.5 Vin =24V fSW = 300kHz Tjmax =125°C Θja = 30°C/W 1.0 0.5 0.4 0.0 0.0 0.0 0 0.5 1 1.5 25 2 40 55 70 85 100 AMBIENT TEMPERATURE (°C) 25 40 55 70 85 100 AMBIENT TEMPERATURE (°C) OUTPUT CURRENT (A) 48V Input Thermal Derating MIC28512-1 Switching Frequency vs. Input Voltage (MIC2812-1) 600 SWITCHING FREQUENCY (KHz) OUTPUT CURRENT (A) 2.5 2.0 5.0V 3.3V 1.5 Vin =48V fSW = 300kHz Tjmax =125°C Θja = 30°C/W 1.0 0.5 0.0 25 40 55 70 85 AMBIENT TEMPERATURE (°C) 100 590 580 570 560 550 540 530 520 510 500 7.0 16.0 25.0 34.0 43.0 52.0 61.0 70.0 INPUT VOLTAGE (V) March 25, 2015 8 Revision 1.2 Micre el, Inc. MIC28512 2 Fun nctional Characteris stics Marcch 25, 2015 9 Revision 1.2 2 Micre el, Inc. MIC28512 2 Fun nctional Characteris stics (Con ntinued) Marcch 25, 2015 10 Revision 1.2 2 Micre el, Inc. MIC28512 2 Fun nctional Characteris stics (Con ntinued) Marcch 25, 2015 11 Revision 1.2 2 Micre el, Inc. MIC28512 2 Fun nctional Characteris stics (Con ntinued) Marcch 25, 2015 12 Revision 1.2 2 Micre el, Inc. MIC28512 2 Fun nctional Characteris stics (Con ntinued) Marcch 25, 2015 13 Revision 1.2 2 Micre el, Inc. MIC28512 2 Fun nctional Characteris stics (Con ntinued) Marcch 25, 2015 14 Revision 1.2 2 Micre el, Inc. MIC28512 2 Fun nctional Diagram Marcch 25, 2015 15 Revision 1.2 2 Micrel, Inc. MIC28512 It is not recommended to use MIC28512 with an OFFtime close to tOFF(min) during steady-state operation. Functional Description The MIC28512 is an adaptive on-time synchronous buck regulator with integrated high-side and low-side MOSFETs suitable for high-input voltage to low-output voltage conversion applications. It is designed to operate over a wide input voltage range (4.6V to 70V) which is suitable for automotive and industrial applications. The output is adjustable with an external resistive divider. An adaptive on-time control scheme is employed to produce a constant switching frequency in continuous-conduction mode and reduced switching frequency in discontinuousoperation mode, improving light load efficiency. Overcurrent protection is implemented by sensing lowside MOSFET’s RDS(ON). The device features internal softstart, enable, UVLO, and thermal shutdown. The adaptive ON-time control scheme results in a constant switching frequency in the MIC28512. The actual ON-time and resulting switching frequency will vary with the different rising and falling times of the external MOSFETs. Also, the minimum tON results in a lower switching frequency in high VIN to VOUT applications. During load transients, the switching frequency is changed due to the varying OFF-time. Figure 1 shows the allowable range of the output voltage versus the input voltage. The minimum output voltage is 0.8V which is limited by the reference voltage. The maximum output voltage is 24V which is limited by the internal circuitry. Theory of Operation As illustrated in the Functional Diagram, the output voltage of the MIC28512 is sensed by the feedback (FB) pin via the voltage dividers R1 and R2, and compared to a 0.8V reference voltage VREF at the error comparator through a low-gain transconductance (gM) amplifier. If the feedback voltage decreases and the amplifier output is below 0.8V, the error comparator will trigger the control logic and generate an ON-time period. The ON-time period length is predetermined by the “Fixed tON Estimator” circuitry: t ON(ESTIMATED ) VOUT VIN fSW Output Voltage Rrange vs. Input Voltage OUTPUIT VOLTAGE (V) 30 Eq. 1 25 Fsw = 600kHz 20 Fsw = 400kHz Fsw = 200kHz 15 ALLOWABLE RANGE 10 0.8V (MINIMUM) 5 0 5 18 31 44 57 70 INPUT VOLTAGE (V) where VOUT is the output voltage, VIN is the power stage input voltage, and fSW is the switching frequency. Figure 1. Allowable Output Voltage Range vs. Input Voltage At the end of the ON-time period, the internal high-side driver turns off the high-side MOSFET and the low-side driver turns on the low-side MOSFET. The OFF-time period length depends upon the feedback voltage in most cases. When the feedback voltage decreases and the output of the gM amplifier is below 0.8V, the ON-time period is triggered and the OFF-time period ends. If the OFF-time period determined by the feedback voltage is less than the minimum OFF-time tOFF(min), which is about 200ns (typical), the MIC28512 control logic will apply the tOFF(min) instead. The tOFF(min) is required to maintain enough energy in the boost capacitor (CBST) to drive the high-side MOSFET. To illustrate the control loop operation, both the steadystate and load transient scenarios will be analyzed. Figure 2 shows the MIC28512 control loop timing during steady-state operation. During steady-state, the gM amplifier senses the feedback voltage ripple, which is proportional to the output voltage ripple and the inductor current ripple, to trigger the ON-time period. The ON-time is predetermined by the tON estimator. The termination of the OFF-time is controlled by the feedback voltage. At the valley of the feedback voltage ripple, which occurs when VFB falls below VREF, the OFF period ends and the next ON-time period is triggered through the control logic circuitry. The maximum duty cycle is obtained from: DMAX 1 t OFF(MIN) fSW March 25, 2015 Eq. 2 16 Revision 1.2 MIC28512 2 Micre el, Inc. e true current-mode contrrol, the MIC28 8512 uses the e Unlike outpu ut voltage rip pple to trigger an ON-time e period. The e outpu ut voltage riipple is proportional to the inducto or curre nt ripple if th he ESR of the e output capacitor is large e enoug gh. The MIC C28512 contro ol loop has the advantage e of elim minating the n need for slope e compensation. In orrder to meet stability req quirements, th he MIC28512 2 feedb back voltage ripple shou uld be in ph hase with the e inducctor current rip pple and larg ge enough to be sensed by y the gm amplifier and the error comparator. The e recom mmended fee edback voltage ripple is 20mV~100mV. If a low-ESR ou utput capacittor is selectted, then the e feedb back voltage rripple may be e too small to be sensed by y the g m amplifier a and the erro or comparatorr. Also, if the e ESR of the outp put capacitorr is very low w, the outpu ut voltag ge ripple and d the feedba ack voltage rripple are no ot necesssarily in pha ase with the inductor currrent ripple. In n these e cases, ripple injection is required to e ensure prope er opera ation. Please refer to the ““Ripple Injectiion” section in n Appliccation Inform mation for mo ore details ab bout the ripple e injecttion technique e. Figure 2.. MIC28512 Co ontrol Loop Timing Figurre 3 shows th he operation of the MIC28 8512 during a load transient. The T output voltage v drops s due to the ease, which causes the VFB to be les ss sudden load incre than VREF. This will w cause the error comparrator to trigge er an O ON-time perio od. At the end d of the ON-ttime period, a minim mum OFF-tim me tOFF(min) is generated to o charge CBSST since e the feedbac ck voltage is still below VREF. Then, the next ON-time periiod is triggere ed due to the low feedbac ck voltage. Thereforre, the switc ching freque ency change es durin ng the load tra ansient, but returns r to the nominal fixed frequ uency once th he output has s stabilized att the new load curre ent level. With the varying g duty cycle and switching frequ uency, the outtput recovery y time is fast and a the outpu ut voltage deviation is small in MIC28512 conv verter. Disco ontinuous M Mode (MIC285 512-1 Only) ductor curre In co ontinuous mode, the ind ent is always greatter than zero; however, at light loads th he MIC28512 21 is able to forcce the inducctor current tto operate in n disco ontinuous mo ode. Discontin nuous mode is where the e inducctor current fa alls to zero, a as indicated b by trace (IL), is show wn in Figure 4. During thiis period, the e efficiency is s optim mized by shuttting down all the non-esssential circuits and minimizing tthe supply ccurrent. The MIC28512-1 wake es up and turn ns on the hig gh-side MOSF FET when the e feedb back voltage VFB drops bellow 0.8V. The MIC28512-1 has a zero crossing comparator tha at monittors the inducctor current b by sensing the e voltage drop p acrosss the low-sid de MOSFET during its O ON-time. If the e VFB > 0.8V and the e inductor currrent goes slig ghtly negative e, then tthe MIC28512-1 automatically powers down most of o the IC C circuitry and d goes into a low-power m mode. 512-1 goes into discontiinuous mode Once e the MIC285 e, both DH and DL are low, which turns off the high-side e and l ow-side MOS SFETs. The lload current iis supplied by y the o output capacittors and VOUTT drops. If the e drop of VOUT cause es VFB to go o below VREFF, then all th he circuits will wake e up into norrmal continuo ous mode. F First, the bias curre nts of most circuitss reduced during the e disco ontinuous mo ode are resto ored, then a tON pulse is trigge ered before tthe drivers arre turned on to avoid any y possiible glitches. Finally, the high-side drriver is turned d on. Figure 4 sshows the control loo op timing in n disco ontinuous mod de. Figure 3. MIC28512 Load Transient Re esponse Marcch 25, 2015 17 Revision 1.2 2 Micre el, Inc. 2 MIC28512 Curre ent Limit The M MIC28512 usses the RDS(OON) of the inte ernal low-side e powe er MOSFET to o sense overrcurrent condiitions. In each h switch hing cycle, the inducto or current iss sensed by y monittoring the low w-side MOSF FET during itts ON period d. The ssensed voltag ge V(ILIM) is compared w with the powe er groun nd (PGND) after a blanking time off 150ns. The e voltag ge drop of th he resistor RILIM is compared with the e low-sside MOSFET T voltage dro op to set the e over-curren nt trip le evel. The sma all capacitor cconnected fro om ILIM pin to o PGND D can be add ded to filter tthe switching node ringing g, allow wing a betterr short limit measureme ent. The time e consttant created by RILIM and the filter cap pacitor should d uch less than be mu n the minimum m off time. The over currentt limit can b be programm med by using g Equa ation 3. R ILIM M ICLIM 0.5 IL(PP) R DS(ON) VCL ICL Figure 4. MIC28512-1 Control C Loop Mode M (Discontinuous Mode) Eq. 3 Wherre: Durin ng discontinu uous mode, the bias current of mos st circuits are reduc ced. As a res sult, the total power supply curre ent during dis scontinuous mode m is only about 450μA A, allow wing the MIC C28512-1 to achieve high h efficiency in light load applicatiions. ICLIM = Desired currrent limit. RDS(OON) = On-ressistance of llow-side pow wer MOSFET T 40mΩ Ω (typical). VCL = Current-lim mit threshold d 14mV (typ pical absolute e value e). See the Ele ectrical Chara acteristics(5) ta able. VDD Regulator The MIC28512 provides p a 5V regulated VDD to bia as intern nal circuitry VIN ranging fro om 5.5V to 70V. 7 When VIN I is lesss than 5.5V, VDD should d be tied to th he VIN pins to bypa ass the interna al linear regulator. ICL = Current-limit source curre ent 70µA (typ pical). See the e Electr trical Characte eristics(5) table e. ∆IL(PPP) = Inductor ccurrent peak-to-peak (use Equation 4 to o calcu ulate the inducctor ripple currrent). Soft--Start Soft-start reduces s the powerr supply inrush current at a up by controlling the outp put voltage riise time while startu the o output capacittor charges. The p peak-to-peak inductor currrent ripple is: IL(PP) The M MIC28512 im mplements an internal digita al soft-start by ramp ping up the 0.8V 0 referenc ce voltage (VREF) from 0 to 100% % in about 5m ms with 9.7m mV steps. This controls the outpu ut voltage ratte of rise at turn on, minimizing inrush curre ent and eliminating outputt voltage ove ershoot. Once the ssoft-start cycle ends, the related r circuittry is disabled to red duce current consumption. VOU UT VIN(MAX ) VOUT VIN(MAX ) fSW W L Eq. 4 The MOSFET R RDS(ON) va aries 30% tto 40% with h tempe erature; therrefore, it is rrecommended to use the e RDS(OON) at max jun nction temperature with 20% margin to o calcu ulate RILIM in: In ca ase of a harrd short, the current limitt threshold is s folded d down to a allow an indefinite hard short on the e outpu ut without any destructive e effect. It is mandatory to o make e sure that th he inductor ccurrent used to charge the e outpu ut capacitor during soft-sta art is under th he folded shorrt limit; otherwise, th he supply will go into hicccup mode and d may n not finish the soft-start succcessfully. Marcch 25, 2015 18 Revision 1.2 2 Micrel, Inc. MIC28512 Power Good (PG) The power good (PG) pin is an open drain output which indicates logic high when the output is nominally 90% of its steady state voltage. MOSFET Gate Drive The Functional Diagram shows a bootstrap circuit, consisting of DBST, CBST and RBST. This circuit supplies energy to the high-side drive circuit. Capacitor CBST is charged, while the low-side MOSFET is on, and the voltage on the SW pin is approximately 0V. When the high-side MOSFET driver is turned on, energy from CBST is used to turn the MOSFET on. As the high-side MOSFET turns on, the voltage on the SW pin increases to approximately VIN. Diode DBST is reverse-biased and CBST floats high while continuing to bias the high-side gate driver. The bias current of the high-side driver is less than 10mA so a 0.1μF to 1μF is sufficient to hold the gate voltage with minimal droop for the power stroke (highside switching) cycle, i.e. ∆BST = 10mA × 1.25μs/0.1μF = 125mV. When the low-side MOSFET is turned back on, CBST is then recharged through the boost diode. A 30Ω resistor RBST, which is in series with the BST pin, is required to slow down the turn-on time of the high-side Nchannel MOSFET. March 25, 2015 19 Revision 1.2 Micre el, Inc. MIC28512 2 App plication Informatio on VIN N Outp put Voltage Setting S Comp ponents The MIC28512 re equires two resistors r to set s the outpu ut voltage as shown in Figure 5. MIC28512 R3 FR EQ R4 GND Figure 6. S Switching Freq quency Adjus stment g The following formula gives the estimatted switching frequ ency. Figure 5. 5 Voltage Divider Configura ation The o output voltage e is determine ed by Equatio on 5. R1 VOUT VFB 1 R 2 R4 fSW f0 R3 R 4 Eq. 5 Eq. 7 Wherre: f0 = switching frrequency wh hen R4 is o open, 680kHz z typica ally. Where: VFB = 0.8V. Figurre 7 shows th he switch freq quency versu us the resisto or R4 w when R3 = 100 0kΩA. A typ pical value of o R1 used on o the standa ard evaluation board d is 10kΩ. If R1 is too larg ge, it may allo ow noise to be introd duced into th he voltage fe eedback loop p. If R1 is too small, it will decre ease the effic ciency of the power supply y, espe ecially at light loads. Once R1 is selecte ed, R2 can be calcu ulated using Equation E 6. R2 VFB F R1 VOU UT VFB Eq. 6 Setting the Switc ching Freque ency The MIC28512 switching s fre equency can be adjusted between 200kHz and 680kHz z by changing the resisto or divide er network fro om VIN. . Figure 7 7. Switching F Frequency vs. R4 Marcch 25, 2015 20 Revision 1.2 2 Micrel, Inc. MIC28512 The winding resistance must be minimized, although this usually comes at the expense of a larger inductor. The power dissipated in the inductor is equal to the sum of the core and copper losses. At higher output loads, the core losses are usually insignificant and can be ignored. At lower output currents, the core losses can be a significant contributor. Core loss information is usually available from the magnetics vendor. Copper loss in the inductor is calculated by using Equation 11: Inductor Selection Values for inductance, peak, and RMS currents are required to select the output inductor. The input and output voltages and the inductance value determine the peak-to-peak inductor ripple current. Generally, higher inductance values are used with higher input voltages. Larger peak-to-peak ripple currents will increase the power dissipation in the inductor and MOSFETs. Larger output ripple currents will also require more output capacitance to smooth out the larger ripple current. Smaller peak-to-peak ripple currents require a larger inductance value and therefore a larger and more expensive inductor. A good compromise between size, loss and cost is to set the inductor ripple current to be equal to 20% of the maximum output current. The inductance value is calculated by: L VOUT VIN(MAX ) VOUT VIN(MAX ) IL(PP) fSW PL(Cu) = IL(RMS)2 × DCR Eq. 11 The resistance of the copper wire, DCR, increases with the temperature. The value of the winding resistance used should be at the operating temperature. DCR(HT) = DCR20C × (1 + 0.0042 × (TH T20C)) Eq. 8 Eq. 12 Where: Where: TH = temperature of the wire under full load. fSW = switching frequency. T20C = ambient temperature. L(PP) = The peak-to-peak inductor current ripple, Typically 20% of the maximum output current. DCR(20C) = room temperature winding resistance (usually specified by the manufacturer). In the continuous conduction mode, the peak inductor current is equal to the average output current plus one half of the peak-to-peak inductor current ripple. IL(PK ) IOUT 0.5 IL(PP) Output Capacitor Selection The type of the output capacitor is usually determined by its equivalent series resistance (ESR). Voltage and RMS current capability are also important factors in selecting an output capacitor. Recommended capacitor types are ceramic, tantalum, low-ESR aluminum electrolytic, OSCON and POSCAP. For high ESR electrolytic capacitors, ESR is the main cause of the output ripple. The output capacitor ESR also affects the control loop from a stability point of view. For a low ESR ceramic output capacitor, ripple is dominated by the reactive impedance. Eq. 9 The RMS inductor current is used to calculate the I2R losses in the inductor. IL(RMS) I2 OUT(MAX ) I2L(PP) I 2 The maximum value of ESR is calculated with Equation 13. Eq. 10 Maximizing efficiency requires the proper selection of core material and minimizing the winding resistance. The high frequency operation of the MIC28512 requires the use of ferrite materials for all but the most cost sensitive applications. Lower cost iron powder cores may be used, but the increase in core loss will reduce the efficiency of the power supply. This is especially noticeable at low output power. The winding resistance decreases efficiency at the higher output current levels. March 25, 2015 ESR COUT VOUT(PP) IL(PP) Eq. 13 Where: ΔVOUT(pp) = peak-to-peak output voltage ripple. ∆IL(PP) = peak-to-peak inductor current ripple. 21 Revision 1.2 Micrel, Inc. MIC28512 The total output ripple is a combination of the ESR and output capacitance. The total ripple is calculated by Equation 14. 2 IL(PP) IL(PP) ESR COUT VOUT (PP) C OUT f SW 8 2 Input Capacitor Selection The input capacitor for the power stage input VIN should be selected for ripple current rating and voltage rating. Tantalum input capacitors may fail when subjected to high inrush currents, caused by turning the input supply on. A tantalum input capacitor’s voltage rating should be at least two times the maximum input voltage to maximize reliability. Aluminum electrolytic, OS-CON, and multilayer polymer film capacitors can handle the higher inrush currents without voltage de-rating. The input voltage ripple will primarily depend on the input capacitor’s ESR. The peak input current is equal to the peak inductor current, so: Eq. 14 Where: D = duty cycle COUT = output capacitance value fSW = switching frequency. VIN IL(PK ) ESR CIN As described in the “Theory of Operation” section in the Functional Description section, the MIC28512 requires at least 20mV peak-to-peak ripple at the FB pin for the gm amplifier and the error comparator to operate properly. Also, the ripple on FB pin should be in phase with the inductor current. Therefore, the output voltage ripple caused by the output capacitors value should be much smaller than the ripple caused by the output capacitor ESR. If low-ESR capacitors, such as ceramic capacitors, are selected as the output capacitors, a ripple injection method should be applied to provide the enough feedback voltage ripple. Please refer to the “Ripple Injection” section for more details. The input capacitor must be rated for the input current ripple. The RMS value of input capacitor current is determined at the maximum output current. Assuming the peak-to-peak inductor current ripple is low ICIN(RMS) IOUT(MAX ) D 1 D IL(PP ) 12 PDISS(CIN) I2 CIN(RMS) ESR CIN March 25, 2015 Eq. 19 Ripple Injection The VFB ripple required for proper operation of the MIC28512’s gm amplifier and error comparator is 20mV to 100mV. However, the output voltage ripple is generally designed as 1% to 2% of the output voltage. If the feedback voltage ripple is so small that the gm amplifier and error comparator can’t sense it, then the MIC28512 will lose control and the output voltage is not regulated. In order to have some amount of VFB ripple, a ripple injection method is applied for low output voltage ripple applications. Eq. 15 The power dissipated in the output capacitor is calculated using Equation 16. PDISS(COUT ) I2 COUT (RMS) ESR COUT Eq. 18 The power dissipated in the input capacitor is: The voltage rating of the capacitor should be twice the output voltage for a tantalum and 20% greater for aluminum electrolytic or OS-CON. The output capacitor RMS current is calculated by Equation 15. ICOUT(RMS ) Eq. 17 Eq. 16 22 Revision 1.2 Micre el, Inc. MIC28512 2 The applications s are divide ed into three situations according to the amount a of the feedback volltage ripple: 1. E Enough ripple e at the feedback voltage due d to the la arge ESR of the t output capacitors. A As shown in Figure F 8, the converter is stable withou ut a any ripple inje ection. The fee edback voltag ge ripple is: R2 ESR COUT IL(PP C P) R1 R2 VFB(PP) Eq. 20 2 Figu ure 8. Enough Ripple at FB W Where: ∆ ∆IL(pp) is the peak-to-pea ak value of the inducto or ccurrent ripple. 2. Inadequate rip pple at the fe eedback volta age due to the ssmall ESR of the output ca apacitors. T The output voltage v ripple e is fed into o the FB pin through a feed-forward cap pacitor, Cff in this situation n, a as shown in n Figure 9. The typical Cff value is d determined by y: Figure e 9. Inadequatte Ripple at FB B R1 CFF 0 10 fSW W Eq. 21 2 W With the feed--forward capa acitor, the fee edback voltage rripple is very close c to the output o voltage e ripple. VFB(PP) ES SR COUT IL(PP) 2 Eq. 22 3. V Virtually no riipple at the FB pin voltag ge due to the vvery low ESR of the outputt capacitors. Marcch 25, 2015 Figurre 10. Invisible e Ripple at FB B 23 Revision 1.2 2 Micrel, Inc. MIC28512 In this situation, the output voltage ripple is less than 20mV. Therefore, additional ripple is injected into the FB pin from the switching node SW via a resistor Rinj and a capacitor Cinj, as shown in Figure 10. The injected ripple is: ∆VFB(pp) VIN K div D (1- D) K div 1 fSW R1//R2 R inj R1//R2 The process of sizing the ripple injection resistor and capacitors is: Step 1. Select Cff to feed all output ripples into the feedback pin and make sure the large time constant assumption is satisfied. Typical choice of Cff is 1nF to 100nF if R1 and R2 are in kΩ range. Step 2. Select Rinj according to the expected feedback voltage ripple using Equation 26. Eq. 23 K div Eq. 24 ∆VFB(pp) VIN fSW D (1- D) Eq. 26 Then the value of Rinj is obtained using: Where: VIN = power stage input voltage Rinj (R1//R2) ( D = duty cycle fSW = switching frequency τ = (R1//R2//RINJ) × CFF In ∆VFB(pp) VIN K div D (1- D) K div 1) Eq. 27 Step 3. Select CINJ as 100nF, which can be considered a short for a wide range of frequencies. 1 fSW 1 and R1//R2 K div R inj R1//R2 , it is assumed that the time constant associated with CFF must be much greater than the switching period: 1 fSW T 1 Eq. 25 If the voltage divider resistors R1 and R2 are in the kΩ range, a Cff of 1nF to 100nF can easily satisfy the large time constant requirements. Also, a 100nF injection capacitor Cinj is used in order to be considered as short for a wide range of the frequencies. March 25, 2015 24 Revision 1.2 Micrel, Inc. MIC28512 Input Capacitor PCB Layout Guidelines Warning: To minimize EMI and output noise, follow these layout recommendations. PCB Layout is critical to achieve reliable, stable and efficient performance. A ground plane is required to control EMI and minimize the inductance in power, signal and return paths. Figure 11 is optimized from small form factor point of view shows top and bottom layer of a four layer PCB. It is recommended to use mid layer 1 as a continuous ground plane. Place the input capacitors on the same side of the board and as close to the PVIN and PGND pins as possible. Place several vias to the ground plane close to the input capacitor ground terminal. Use either X7R or X5R dielectric input capacitors. Do not use Y5V or Z5U type capacitors. Do not replace the ceramic input capacitor with any other type of capacitor. Any type of capacitor can be placed in parallel with the input capacitor. If a Tantalum input capacitor is placed in parallel with the input capacitor, it must be recommended for switching regulator applications and the operating voltage must be derated by 50%. In “Hot-Plug” applications, a Tantalum or Electrolytic bypass capacitor must be used to limit the overvoltage spike seen on the input supply with power is suddenly applied. SW Node Do not route any digital lines underneath or close to the SW node. Keep the switch node (SW) away from the feedback (FB) pin. Output Capacitor Use a copper island to connect the output capacitor ground terminal to the input capacitor ground terminal. Phase margin will change as the output capacitor value and ESR changes. Contact the factory if the output capacitor is different from what is shown in the BOM. The feedback trace should be separate from the power trace and connected as close as possible to the output capacitor. Sensing a long high-current load trace can degrade the DC load regulation. Figure 11. Top and Bottom Layer of a Four Layer Board The following guidelines should be followed to ensure the proper operation of the MIC28512 converter: IC The analog ground pin (AGND) must be connected directly to the ground planes. Do not route the AGND pin to the PGND pin on the top layer. Place the IC close to the point of load (POL). Use copper planes to route the input and output power lines. Analog and power grounds should be kept separate and connected at only one location. March 25, 2015 25 Revision 1.2 Micrel, Inc. MIC28512 Thermal Measurements Measuring the IC’s case temperature is recommended to insure it is within its operating limits. Although this might seem like a very elementary task, it is easy to get erroneous results. The most common mistake is to use the standard thermal couple that comes with a thermal meter. This thermal couple wire gauge is large, typically 22 gauge, and behaves like a heatsink, resulting in a lower case measurement. Two methods of temperature measurement are using a smaller thermal couple wire or an infrared thermometer. If a thermal couple wire is used, it must be constructed of 36 gauge wire or higher than (smaller wire size) to minimize the wire heat-sinking effect. In addition, the thermal couple tip must be covered in either thermal grease or thermal glue to make sure that the thermal couple junction is making good contact with the case of the IC. Omega brand thermal couple (5SC-TT-K-36-36) is adequate for most applications. Wherever possible, an infrared thermometer is recommended. The measurement spot size of most infrared thermometers is too large for an accurate reading on a small form factor ICs. However, a IR thermometer from Optris has a 1mm spot size, which makes it a good choice for measuring the hottest point on the case. An optional stand makes it easy to hold the beam on the IC for long periods of time. March 25, 2015 26 Revision 1.2 Micre el, Inc. MIC28512 2 MIC C28512 Ev valuation Board B Sch hematic Bill of Materials Item m C1 C2, C3 C4, C7 Part Number UVZ2A3 330MPD 12061Z4 475KAT2A C1608X X7R1A225K080 0AC Man nufacturer Niichicon (6) C9 C10 0, C17 C0603C C104K8RACTU U GRM21B BR72A474KA7 73 08051C4 474KAT2A GRM188 8R72A104KA3 35D 3 33µF/100V 20% % Radial Aluminum Capacito or 1 4 4.7µF/100V, X7 7S, Size 1206 Ceramic Capa acitor 2 (8) 2 2.2µF/10V, X7R R, Size 0603 C Ceramic Capaccitor 2 O OPEN NA 0 0.1µF/10V, X7R R, Size 0603 C Ceramic Capaccitor 2 0 0.47µF/100V, X X7R, Size 0805 5 Ceramic Cap pacitor 1 0 0.1µF/100V, X7 7R, Size 0603 Ceramic Capa acitor 2 AVX TDK (9) Kemet K (10) Murata M AVX Murata C11 C12 2 CGA3E2 2X7R1H102K C14 4, C15 GRM32E ER71A476KE1 15L Qty. (7) C5, C13 C6, C16 D Description O OPEN NA TDK 1 1nF/50V, X7R, Size 0603 Ceramic Capacito or 1 Murata 4 47µF/10V, X7R R, Size 1210 Ceramic Capacitor 2 Notes s: 6. Niichicon: www.nic chicon.co.jp/engliish. 7. AV VX: www.avx.com m. 8. TD DK: www.tdk.com m. 9. Ke emet: www.keme et.com. 10. Murata: www.mura ata.com. Marcch 25, 2015 27 Revision 1.2 2 Micrel, Inc. MIC28512 Bill of Materials (Continued) Item Part Number Manufacturer Description Qty. C18 Open NA C19 Open NA C20 Open NA Open NA C21 D1 BAT46W-TP (11) MCC 100V Small Signal Schottky Diode, SOD123 D3 Open J1, J7, J8, J10, J11, J12, J16, J17, J18 77311-118-02LF L1 XAL7030-682MED R1 CRCW060310K0FKEA FCI (13) Coilcraft(14) Vishay Dale (15) 1 NA CONN HEADER 2POS VERT T/H 9 8.2µH, 10.2A sat current 1 10.0kΩ, Size 0603, 1% Resistor 1 R2 OPEN NA R9 OPEN NA R10 CRCW06033K24FKEA Vishay Dale 3.24kΩ, Size 0603, 1% Resistor 1 R11 CRCW06031K91FKEA Vishay Dale 1.91kΩ, Size 0603, 1% Resistor 1 R14, R15 CRCW06030000FKEA Vishay Dale 0.0 Ω, Size 0603, Resistor Jumper 2 CRCW0603100K0FKEA Vishay Dale 100kΩ, Size 0603, 1% Resistor R26 R16, R19, R17,R3 Open R25 Open NA 4 NA R18 CRCW06031K00JNEA Vishay Dale 1.0kΩ, Size 0603, 5% Resistor 1 R20, R21 CRCW060349R9FKEA Vishay Dale 49.9Ω, Size 0603, 1% Resistor 2 R22 CRCW06031K74FKEA Vishay Dale 2.21kΩ, Size 0603, 1% Resistor 1 R23 CRCW08051R21FKEA Vishay Dale 1.21Ω, Size 0805, 1% Resistor 1 R24 CRCW060340R0FKEA Vishay Dale 40.0Ω, Size 0603, 1% Resistor 1 TP1 TP2 OPEN TP7 TP14 77311-118-02LF FCI CONN HEADER 2POS VERT T/H 1 TP8 TP13 77311-118-02LF FCI CONN HEADER 2POS VERT T/H 1 TP17 TP18 77311-118-02LF FCI CONN HEADER 2POS VERT T/H 1 TP9, TP10, TP11, TP12 1502 Keystone (16) Electronics Testpoint Turret, .090 4 U1 MIC28512-1YFL Micrel. Inc.(17) 70VIN, 2A Synchronous Buck Regulator 1 Notes: 11. MCC: www.mccsemi.com. 12. Diode: www.diodes.com. 13. FCI: www.fciconnect.com. 14. Coilcraft: www.coilcraft.com. 15. Vishay Dale: www.vishay.com. 16. Keystone Electronics: www.keystone.com. 17. Micrel, Inc.: www.micrel.com. March 25, 2015 28 Revision 1.2 Micrel, Inc. MIC28512 MIC28512 Evaluation Board Top Layer Mid Layer 1 March 25, 2015 29 Revision 1.2 Micrel, Inc. MIC28512 MIC28512 Evaluation Board (Continued) Mid Layer 2 Bottom Layer March 25, 2015 30 Revision 1.2 Micre el, Inc. MIC28512 2 Pac ckage Info ormation and a Recom mmended d Land Patttern (18) 24-Pin 3mm m × 4mm FCQF FN (FL) Note: 18. Pa ackage information is correct as of o the publication n date. For updattes and most currrent information, go to www.micrel.com. Marcch 25, 2015 31 Revision 1.2 2 Micrel, Inc. MIC28512 MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com Micrel, Inc. is a leading global manufacturer of IC solutions for the worldwide high-performance linear and power, LAN, and timing & communications markets. The Company’s products include advanced mixed-signal, analog & power semiconductors; high-performance communication, clock management, MEMs-based clock oscillators & crystal-less clock generators, Ethernet switches, and physical layer transceiver ICs. Company customers include leading manufacturers of enterprise, consumer, industrial, mobile, telecommunications, automotive, and computer products. Corporation headquarters and state-of-the-art wafer fabrication facilities are located in San Jose, CA, with regional sales and support offices and advanced technology design centers situated throughout the Americas, Europe, and Asia. Additionally, the Company maintains an extensive network of distributors and reps worldwide. Micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in this datasheet. This information is not intended as a warranty and Micrel does not assume responsibility for its use. Micrel reserves the right to change circuitry, specifications and descriptions at any time without notice. No license, whether express, implied, arising by estoppel or otherwise, to any intellectual property rights is granted by this document. Except as provided in Micrel’s terms and conditions of sale for such products, Micrel assumes no liability whatsoever, and Micrel disclaims any express or implied warranty relating to the sale and/or use of Micrel products including liability or warranties relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright, or other intellectual property right. Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully indemnify Micrel for any damages resulting from such use or sale. © 2014 Micrel, Incorporated. March 25, 2015 32 Revision 1.2