V04N2 - JUNE

LINEAR TECHNOLOGY
JUNE 1994
VOLUME IV NUMBER 2
IN THIS ISSUE...
COVER ARTICLE
LT1251/LT1256 Video Fader
and DC Gain-Controlled
Amplifier ......................... 1
William H. Gross
Editor's Page ................... 2
LT1251/LT1256 Video
Fader and DC GainControlled Amplifier
Rich Markell
Dave Dwelley
LT1302 Micropower DC/DC
Converter Delivers Unprecedented Power Levels in an
Eight-Lead SOIC Package
........................................ 6
Steve Pietkiewicz
The LTC1066-1: Fourteen Bit
DC Accurate Elliptic Lowpass
Filter ............................. 12
Nello Sevastopoulos
Micropower, 12-Bit, SO-8
ADCs Now Available for
Three-Volt Systems ........ 15
William C. Rempfer and Marco Pan
DESIGN IDEAS .......... 17–26
(complete list on page 17)
New Device Cameos ....... 30
LTC in the News ............ 31
Design Tools .................. 32
Sales Offices ................. 32
by William H. Gross
Introduction
The Video Fader
Variable-gain amplifiers are used
extensively to provide effects and level
adjustments in video systems. The
simplest and most common effect is
the fade to black. This is created by
gradually reducing the signal gain to
zero. Another, more specialized kind
of video mixing is the dissolve, where
one signal is reduced while another is
increased. The result is the familiar
fading of one scene into another. The
dissolve is accomplished by a circuit
block called the fader. The fader circuit has two signal inputs, a control
input, and a single output. Of course,
if the second input signal is zero, the
scene fades to black.
The ideal fader would have welldefined gain versus control voltage,
sufficient bandwidth for video, low
distortion, and a DC output level that
is independent of the control signal.
The most difficult design issue in
making a fast, variable-gain amplifier is obtaining good gain accuracy;
the gain must be linear with respect
to the control signal and it must not
change with temperature.
The LT1251/LT1256 is a two-input, one-output current-feedback
amplifier with a linear control that
sets the amount each input contributes to the output. The gain-control
circuitry is trimmed for absolute accuracy at wafer sort and is completely
temperature compensated.
A block diagram of the basic fader
is shown in Figure 1. The control
signal varies the position of the potentiometer such that at one extreme
the output contains only IN1 and at
the other extreme it contains only
IN2. The control is linear; i.e., for the
control signal at 50%, the output is
the sum of one half of IN1 and one
half of IN2. If both inputs are the
same, the output is independent of
the control signal. The design of the
controlled potentiometer is the most
challenging aspect of fader design.
For zero or 100% control input, it is
important that the video fader completely attenuates the appropriate
input signal. Most of the time the
fader is operating with only one input
signal at the output. In this condition, the off signal must not bleed
continued on page 27
IN1
OUT
IN2
IN2
GAIN
DESIGN FEATURES
The LTC1152 Rail-to-Rail
Operational Amplifier ..... 3
IN1
CONTROL
CONTROL
1251_1.eps
Figure 1. Basic fader circuit
EDITOR'S PAGE
Morphing the
Factory Applications Staff
by Rich Markell
Much has been written of late about
the popular visual effect called
“morphing.” An example is when Odo
on Star Trek exquisitely turns himself into a small mouse and scampers
around the space station unnoticed,
then returns as Odo, with the evidence he needs to convict Quork of
running a gambling operation.
Linear Technology is not (yet) in
the business of creating special effects for Hollywood, but as time goes
by and more engineers are added to
the factory applications staff, the personality mix becomes more and more
diverse. Suppose the personalities of
the factory Application Engineers were
all metamorphosed into one. How
would this event be perceived on the
phone or in person? What would it be
like?
You’d probably step out of your
office to meet a nice man who’d have
pens, calculators, and barometric
sensors in one pocket and, perhaps,
gardening tools in the other. He’d
probably be wearing a multicolored
propeller beanie, but the propeller
would be partially broken off and
have CK722’s or nuvistors attached
to it. The vision fits and is only partly
a joke. When you finally sit down to
discuss circuits with our morphed
“all-around engineer,” you’d find
someone with a broad knowledge of
LTC’s product line. He could help you
design the world’s smallest, flattest,
most efficient switcher or a “Betterthan-Bessel” filter. He could evaluate
your HDSL application and suggest
an A/D to fit, or tell you which video
mux to use to switch between HDTV
sumo wrestling and luge. Finally, at
the end of the day, perhaps he’d morph
back into a mouse and crawl into
Williams’ 547 for the night.
Our lead article in this issue features a great new video-product
family, the LT1251/LT1256. These
products incorporate a 30MHz video
fader and a DC gain-controlled amplifier into a single IC. The parts have
greater than 80dB signal-to-noise
ratios with good differential phase
and gain response. These devices are
perfect for new multimedia computer
boards, video products for both the
professional and the consumer, and
a variety of other circuits. We also
introduce a new micropower DC/DC
converter, the LT1302. Designed for
battery-powered applications, no
other part can deliver so much power
from an SO-8 package. The LT1302
operates from two, three, or four cells
and can provide up to 600 milliamps
at 5 volts or up to 1 amp from a 3.3
volt supply.
Also featured in this issue is the
LTC1152. Designed in Singapore, the
LTC1152 is a zero-drift, rail-to-rail
input, rail-to-rail output swing operational amplifier. The part operates
from 2.7V to 14V of total supply voltage. The amplifier has a 1MHz
gain-bandwidth product and can plug
directly into any 8-pin op amp socket.
The LTC1066-1, introduced in these
pages, is the first monolithic filter
that combines RC active techniques
with switched-capacitor technology.
The LTC1066-1 is an eighth-order,
elliptic lowpass filter with only 1.5
millivolts (max.) DC offset and 14 bits
of DC gain linearity.
ADCs are again prominent in this
issue, as we highlight the LTC1285
and LTC1288. These two converters
are small, SO-8 12-bit converters with
serial interfaces for digitizing sensors
or pen screen inputs, and for use in
cellular phones.
We conclude with a palette of Design Ideas in this issue as well as a
variety of New Device Cameos.
FAE Cameo: Jon Dutra
ham radio operator since age 14 and
is a licensed but inactive pilot.
In over five years with LTC he has
had many interesting experiences. “I
love working with customers, helping
them solve their engineering problems in elegant, cost effective ways,”
he says.
Currently, Jon is spending about
50% of his time with regulator issues;
interface products, filters, op amps,
references, and A/Ds consume the
remainder. When Jon is not in the
field, he can often be found in the lab,
building and testing some new cir-
cuit, sometimes just for fun, but usually for a specific customer. One such
breadboard turned into a $1.7 million order and helped him earn the
new award of LTC FAE of the Year.
Jon and his wife Barbara have
been married for eight years. They
have two children, Thomas, 6, and
Brittany, 5, who keep their parents
very busy. He enjoys tennis, gardening, building “things,” and playing
with his children. Jon can be reached
through the LTC Northwest Sales office listed on the back of this magazine.
LTC now has twenty-two Field Application Engineers (FAEs) worldwide
to assist our customers however possible. Jon Dutra is one of two FAEs in
our Silicon Valley Northwest Sales
Office. He now covers Northern Nevada, Idaho, and San Francisco Bay
Area customers with company names
beginning in the letters A–K.
Early in his career, Jon designed
thermocouple measurement systems,
data-acquisition systems, switching
power supplies, and video-speed analog systems. He has been an active
2
Linear Technology Magazine • June 1994
DESIGN FEATURES
The LTC1152 Rail-to-Rail
Operational Amplifier
by Dave Dwelley
Introduction
Over the past few years, the term
“rail-to-rail” has become a common
phrase in op-amp advertising. Generally, this implies an output stage
that can swing to within millivolts of
either power supply. Many CMOS and
BiCMOS parts make this claim; there
are even a few all-bipolar designs that
can come within a saturation voltage
of the rails (approximately 0.7 volts),
close enough to rail-to-rail for most
designers. Some of these parts include inputs whose common-mode
ranges include the negative power
supply rail; this generally earns the
op amp the additional tag line “single
supply.” All this is well and good, but
it obscures an important point: nearly
all the parts boasting “rail-to-rail”
performance don’t include input common mode to the positive rail.
Rail-to-Rail Input CMR
Common-mode range (CMR) is one
of those specs buried deep in the
data sheet that few people look at.
Perhaps, but rail-to-rail input
common-mode range is the distinction that separates true rail-to-rail op
amps from plain vanilla op amps with
fancy output stages. Take one of the
more common op-amp circuits, the
unity-gain follower (Figure 1). Most
designers are familiar with this circuit for a couple of reasons; there are
no pesky resistor-ratio formulas to
figure the gain and you get predictable performance as long as you
remember to pick a unity-gain-stable
op amp. There is, however, a hidden
V+
2
–
7
6
IN
3
+
OUT = IN
4
V–
1152_1.eps
Figure 1. Unity-gain follower
Linear Technology Magazine • June 1994
V + (PIN 7)
CP (PIN 8)
+
(V ) + 2V
+IN (PIN 3)
FRONT
END
CHARGE
PUMP
–IN (PIN 2)
OUT (PIN 6)
V – (PIN 4)
1152_2.eps
Figure 2. LTC1152 functional diagram
trap with this circuit. Operating from
a single 5V supply, most “single supply, rail-to-rail” op amps will follow
the input all the way to ground. However, as we approach the positive
supply, the output stops swinging!
We’re not even close to the rail yet!
What happens in this situation is
that the input exceeds the op amp’s
CMR long before the output gets to
the rail. When op amps ran from ±15V
supplies, input common-mode range
generally ran out a couple of volts
away from either rail, but no one
seemed to mind too much; giving
away 3 or 4 volts out of 30V of possible input swing didn’t seem like too
much of a compromise. That same op
amp running from a single 5V supply
now has only 1V of CMR remaining—
suddenly it’s very significant. Newer
parts that can common mode to V−
get the bottom end of the range back,
but most run out of CMR around 1.5V
below V+. That’s still a large portion of
the total input range; such a part
running from a single 2.7V supply
(3V ±10%) gives away more than half
of its possible input range. To further
aggravate the situation, many op amps
get weird when you exceed their common-mode ranges; the front-end
devices can turn completely off, causing second-order effects that can do
strange things to the output stage
and cause problems with the feedback loop. To make a true rail-to-rail
follower, you need a rail-to-rail input
CMR, rail-to-rail output-swing op amp
that works over a wide range of supply voltages—like the LTC1152.
The Secret
The LTC1152 is a CMOS, zerodrift, rail-to-rail input CMR, rail-to-rail
output-swing operational amplifier
that will work from 2.7V to 14V total
supply voltage. It achieves rail-to-rail
input CMR by using a self-contained
charge pump to generate an internal
voltage regulated to about 2V higher
than V+. This allows it to use a conventional PMOS front-end structure
running from this internal supply
(Figure 2). Input signals at V+ are still
well below this internal supply, allowing the front end to amplify them
without level shifting or extra front
end devices. This same front end
inherently includes V− in its common-mode range, allowing full
rail-to-rail CMR from a traditional
front-end structure. CMR typically
extends about 0.3V beyond either rail
before leakages start to form across
the parasitic clamp diodes and affect
the input impedance.
The charge pump and all of its
support circuitry, including the capacitors, are included on the die; no
3
DESIGN FEATURES
external components are required.
Additionally, the charge pump typically runs at 4.7MHz, well above the
1MHz gain bandwidth of the LTC1152;
this ensures that very little chargepump feedthrough actually reaches
the output pin. Sensitive applications
can further reduce feedthrough by
connecting an external bypass capacitor between the charge pump
output at pin 8 and V+ at pin 7; a
0.1µF cap will pretty much wipe out
any remaining charge-pump noise at
the output. The self-contained charge
pump also allows the LTC1152 to
conform to the industry-standard opamp pinout. It can plug into any
standard 8-pin op-amp socket, either
standard DIP or SO8, provided any
trim circuitry on pins 1, 5, and 8 is
removed. The charge pump is necessarily quite small to fit inside the
package; as a result, it’s not a good
idea to try to drive any external loads
(other than bypass capacitors) from
pin 8. By the same token, don’t connect the bypass capacitor to ground
or V−; the more volts the charge pump
has to put into the capacitor, the
longer it will take for the LTC1152 to
start up.
The Rest of the Secret
In addition to the charge-pump
front end, the LTC1152 has a zerodrift architecture, adapted from LTC’s
family of zero-drift op amps. Like the
other family members, the LTC1152
constantly corrects its own offset and
drift errors for optimum DC performance. The net result is an offset
voltage spec of 10µV maximum,
100nV/°C maximum drift over temperature, minimum PSRR of 105dB
over temperature, and minimum
115dB CMRR over temperature, all
over the entire rail-to-rail input range.
Additionally, the LTC1152 maintains
this performance over a single-supply voltage range of 2.7V to 14V, or a
dual-supply range of ±1.35V to ±7V,
allowing it to run from most standard
digital supplies as well as split analog
supplies. In other words, a follower
built with an LTC1152CS8 will have
an output voltage guaranteed to be
4
within 25µV of the input, in Singapore
or in Lillehamer (over temperature),
all errors accounted for, with almost
any power supply that’s handy, until
the output runs into either powersupply rail. Try that with any other op
amp!
As with all zero-drift op amps, the
LTC1152 pays for its exceptional DC
performance by exhibiting aliasing
behavior at its internal clocking frequency. In the case of the LTC1152,
that clocking frequency is about
2.3kHz (actually, it’s exactly the
charge pump frequency divided by
2048). AC input signals near this
frequency will generate aliasing products, with their magnitude dependent
on the closed-loop gain of the circuit
configuration. As a rule of thumb, the
aliasing products will be about (80dB
– the closed-loop gain) below the input signal. In the case of a unity-gain
follower with the closed-loop gain of
0dB, the sum and difference frequencies will be attenuated by about
80dB—pretty far down. For higher
gain configurations, they may be more
prominent. Because the zeroing clock
is divided down from the charge pump
oscillator, there are no interference
products or “beat frequencies” between the charge pump and the
auto-zero circuit. Many applications,
especially those with relatively low
gain, can use the LTC1152 as a
wideband amplifier all the way up to
its 1MHz gain-bandwidth product,
without ever noticing that it is an
autozeroed amplifier.
Rail-to-Rail Output
The LTC1152’s input stage
wouldn’t be much use without an
equally rail-to-rail output stage. The
output stage is powered off the hard
power supply rails, not the internal
charge pump; although this prevents
it from swinging outside the rails (an
LT1026 charge-pump chip can be
used if you need to do that), it allows
the output to provide much more
current than could be supplied with
an all-internal charge pump. The
LTC1152 output will swing right up
to either rail when unloaded; the
open-loop output impedance is about
190Ω, limiting the output swing with
load to that of the resistor divider
formed between this 190Ω and the
load impedance. Very high value load
resistors allow the output to swing
closer than a millivolt to either rail; a
1k load will swing to about +4.2V with
a 5V supply. The output current is
limited to about ±20mA under shortcircuit conditions.
Single-supply applications with the
load referenced to ground will swing
all the way to ground due to the
pulldown effect of the load; the only
error is the half-wave rectified input
noise amplified by the closed-loop
gain of the circuit. This can be as low
as 1µV for unity gain circuits—that’s
pretty close to ground. Similarly, circuits with the output loaded to V+ will
swing all the way to V+.
2
–
LTC1152
IN
3
+
6
OUT
CLOAD
5
CCOMP
1152_3.eps
Figure 3. Externally compensating the
LTC1152
The output stage is optimized to
drive capacitive loads up to about
10,000pF. Larger capacitive loads can
be driven by externally compensating
the LTC1152 (Figure 3). Connecting a
1000pF capacitor from pin 5 (COMP)
to pin 6 (OUT) ensures unity-gain
stability with loads up to 1µF; 0.1µF
between COMP and OUT allows the
LTC1152 to drive as much capacitance as you can put on it. The
trade-off is speed; large output compensation capacitors work by moving
the compensation pole lower in frequency, directly affecting the
gain-bandwidth product. With a
1000pF compensation capacitor, the
LTC1152 has a gain-bandwidth product of about 20kHz.
Linear Technology Magazine • June 1994
DESIGN FEATURES
Shutdown Mode
The LTC1152 includes a shutdown
feature that disables the part, puts
the output into a high-impedance
state, and drops the supply current
from 2.2mA to about 1µA. All this is
accomplished by pulling pin 1 low,
either with CMOS logic running from
the same supplies as the LTC1152 or
with an open collector/open drain
device. Additionally, the shutdown
pin thresholds are designed so that,
when the LTC1152 is run from dual
supplies, CMOS logic running from
the same positive supply and ground
can interface directly to the shutdown pin. Pin 1 includes an internal
pullup to ensure that the part stays
active if the pin is left floating. This
pullup current increases when the
part is active to ensure that capacitive feedthrough from fast-moving
signals at pin 2 (the amplifier’s negative input) does not inadvertently
couple to the shutdown circuit and
shut the LTC1152 down.
Applications
The combination of features provided by the LTC1152 make it well
suited for instrumentation applications that require high DC precision
and maximum dynamic range. With
a single 5V supply, the LTC1152 can
process signals from its 2µVP-P noise
floor all the way to 5VP-P; almost
128dB, or better than 21 bits at DC.
It is also well suited for use as a
precision unity-gain buffer for transducers operated from single supplies;
the pA-level bias currents and tiny
offset voltage will preserve signals
from even very high impedance devices without degradation, and the
exceptional CMRR over the entire input range allow it to pick out small
signals buried under large commonmode interference. The capacitive load
capabilities of the output stage allow
it to drive long runs of cable, making
it a good buffer amplifier for remotely
located sensors. The wide input common-mode range allows the LTC1152
to be used to sense current in either
power supply rail, while the excellent
DC precision allows extremely small
Linear Technology Magazine • June 1994
value sense resistors to be used, minimizing the effect on the rest of the
circuit.
The high-impedance output state
in shutdown mode can be used to
multiplex several signals into one
by tying the outputs of multiple
LTC1152s together (Figure 4). The
active channel is selected by enabling
the corresponding LTC1152 with its
SD pin and disabling the others. The
low output impedance of the active
channel prevents the feedback networks of the disabled channels from
causing errors in the output. When
all channels are disabled, the mux
output becomes high-impedance if
the individual channels are configured as followers; any other amplifier
configurations will load the output
with the feedback resistor network.
Enabling more than one channel at
once will cause unusual behavior as
two active channels fight with each
other; the LTC1152’s current-limited
output will protect the amplifiers from
damage, but the output may not be
what you expected.
Layout Precautions
As with all zero-drift amplifiers,
circuits using the LTC1152 must be
laid out with some care, or external
parasitics will cause DC errors much
greater than those caused by the
LTC1152 itself. Thermocouples created where the solder meets the copper
PC board traces can create temperature-drift errors as much as ten times
greater than the 10nV/°C due to the
LTC1152. Leakages from leftover solder flux or cheap PC board material
can swamp the input bias current
due to the chip and cause bizarre
low-level behavior at high impedance
inputs (such as the positive input of
a unity gain follower). This topic is
covered in some detail in the LTC1152
data sheet; some of the things that
can cause errors are surprising. Read
the discussion to get that last unaccounted microvolt out of your system.
Although the LTC1152 will work fine
in sloppy layouts, it won’t give all the
DC precision it is capable of. In particular, don’t expect to see microvolt/
SEL 1
–
LTC1152
#1
IN1
MUX
OUTPUT
+
SEL 2
–
LTC1152
#2
IN2
+
SEL 3
–
LTC1152
#3
IN3
+
1152_4.eps
Figure 4. Multiplexing four LTC1152s using
the shutdown pin
picoamp performance if you build
your test circuit on a plug-in protoboard. If you’re a regular reader of
this magazine, you’ve already read
about how using a ground plane will
improve your circuit’s performance
and spice up your life in other ways,
so I won’t go into detail. Just use one.
Conclusions
The LTC1152 comes about as close
as possible to ideal op amp performance at DC and low frequencies:
lots of output current, virtually no
input current, low power-supply current (especially when shut down), and
no DC error over the entire supply
range, input and output. Additionally, it’s simple to use; it uses the
standard op-amp pinout in both DIP
and SO8, and the extra functional
pins will mind their own business if
you chose to ignore them. If you use
the extra features, the LTC1152 can
do things most op amps can only
dream of: sense signals over the entire power supply range, drive huge
capacitive loads, shut down to virtually zero supply current while putting
the output in a high-impedance state,
and create that most elusive of circuits, the true rail-to-rail follower.
5
DESIGN FEATURES
LT1302 Micropower DC/DC
Converter Delivers Unprecedented
Power Levels in an Eight-Lead
SOIC Package
by Steve Pietkiewicz
Introduction
The LT1302 micropower, DC-toDC converter IC achieves new levels
of performance. No other IC converter
on the market today is capable of
efficiently delivering so much power
in such a small package. The internal, low-loss NPN switch can handle
current in excess of 2A with a drop of
just 300mV; the device can deliver 5V
at up to 600mA from a 2V input or up
to 1A from a 3.3V supply. Designed
for battery-powered applications, the
LT1302 can operate from a two-,
is only 200µA and the shutdown pin
can be activated to further reduce
supply current to just 15µA. Inductor
and capacitor size are kept small because the operating frequency is in
the 200kHz–400kHz range, allowing
all components to be surfacemounted. Designers of PDAs, digital
cellular phones, portable transmitting devices, or other systems needing
high efficiency over a broad output
power range will appreciate the performance of the LT1302.
three-, or four-cell input and can
deliver far more output power than
other micropower DC/DC converter
ICs. Finally, designers of battery-operated systems can get the output
power they’ve been looking for
without resorting to complex, spaceconsuming, and expensive discrete
solutions.
The LT1302 maintains high efficiency over a wide range of load
current, thanks to automatic Burst
ModeTM operation. Quiescent current
D1
L1
VIN
C1
+
+
C2
0.1µF
6
7
VIN
36mV
R5
730Ω
A2
–
+
OFF
ENABLE
–
R1
VOS
15mV
R2
220kHz
OSCILLATOR
Q5
SHUTDOWN
Q4
160X
DRIVER
VIN
A1
SHDN
Q3
VIN
–
3
A3
HYSTERETIC
COMPARATOR 2µA
FB
4
C5
100pF
CMP1
SW
R4
2.2Ω
+
1.24V
REFERENCE
VOUT
C3
Q1
BIAS
Q2
+
ERROR
AMPLIFIER
300Ω
1
GND
2
VC
5
ILIM
3.6k
8
PGND
R3
22k
C4
0.01µF
1302_1.eps
Burst Mode is a trademark of Linear Technology Corporation
6
Figure 1. Block Diagram: LT1302
Linear Technology Magazine • June 1994
DESIGN FEATURES
NC
6
7
+
2 CELLS
C3
0.1µF
D1
C1
100µF
ILIM
VIN
SHDN
SW
3
SHUTDOWN
LT1302
8
PGND
GND
+
5
1
C2
100µF
FB
4
VC
2
RC
20k
CC
6800pF
5V
600mA
OUTPUT
C1, C2 = SANYO OS-CON
L1 = COILTRONICS CTX10-3
COILCRAFT DO3316-103
D1 = MOTOROLA MBRS130LT3
R1
100k
1%
200pF
R2
301k
1%
COILTRONICS (407) 241-7876
COILCRAFT (708) 639-2361
1302_2.eps
Figure 2. Two or three cell to 5V converter delivers 600mA, 1A from 3.3V supply
Operation
The LT1302’s operation can best
be understood by examining the block
diagram in Figure 1. The LT1302 operates in one of two modes, depending
on load. With light loads, comparator
CMP1 controls the output; with heavy
loads, control is passed to error amplifier A1. Burst ModeTM operation
consists of monitoring the FB pin
voltage with hysteretic comparator
CMP1. When the FB voltage, related
to the output voltage by external attenuator R1 and R2, falls below the
1.24V reference voltage, the oscillator is enabled. Switch Q4 alternately
turns on, causing current buildup in
inductor L1, then turns off, allowing
the built-up current to flow into output capacitor C3 via D1. As the output
voltage increases, so does the FB voltage; when it exceeds the reference
plus CMP1’s hysteresis (about 5mV)
CMP1 turns the oscillator off. In this
mode, peak switch current is limited
to approximately 800mA by A2, Q2,
and Q3. Q2’s current, set at 34µA,
flows through R5, causing A2’s negative input to be 25mV lower than VIN.
This node must fall more than 36mV
below VIN for A2 to trip and turn off
the oscillator. The remaining 11mV is
generated by Q3’s current flowing
through R4. Emitter-area scaling sets
Q3’s collector current to 0.625% of
switch Q4’s current. When Q4’s current is 800mA, Q3’s current is 5mA,
creating an 11mV drop across R4
Linear Technology Magazine • June 1994
which, added to R5’s 25mV drop, is
enough to trip A2.
When the output load is increased
to the point where the 800mA peak
current cannot support the output
voltage, CMP1 stays on and the peak
switch current is regulated by the
voltage on the VC pin (A1’s output).
VC drives the base of Q1. As the VC
voltage rises, Q2 conducts less current, resulting in less drop across R5.
Q4’s peak current must then increase
in order for A2 to trip. This currentmode control results in good stability
and immunity to input voltage variations. Because this is a linear,
closed-loop system, frequency compensation is required. A series RC
from VC to ground provides the necessary pole-zero combination.
two-cell applications (although this
guideline is ignored in the two-cellto-12V circuit shown later). Lower
values typically have less DC resistance and can handle higher current.
Transient response is better with low
inductance, but more output current
can be had with higher values. Peak
current in Burst ModeTM operation
increases as inductance decreases,
due to the finite response time of the
current sensing comparator in the
LT1302. The Coilcraft DO3316 series
inductors have been found to be excellent in terms of performance, size,
and cost, but their open construction
results in some magnetic flux spray;
try Coiltronics’ OCTAPAC series if
EMI is a problem. Transient response
with a load step of 25mA to 525mA is
detailed in Figure 3. There is no overshoot upon load removal because
switching stops entirely when output
voltage rises above the comparator
threshold. Undershoot at load step is
less than 5%. The circuit’s efficiency
at various input voltages is shown in
Figure 4.
Although efficiency graphs present
useful information, a more “realworld” measure of converter performance comes from battery-lifetime
VOUT
100mV/DIV
AC COUPLED
ILOAD
525mA
25mA
Applications
Two or Three Cell
to 5V Converter
Figure 2 shows a two or three cell
to 5V DC/DC converter that can deliver up to 600mA from a two-cell
input (2V minimum), or up to 900mA
from a three-cell input (2.7V minimum). R1 and R2 set the output
voltage at 5V. The 200pF capacitor
from FB to ground aids stability; without it the FB pin can act as an antenna
and pick up dV/dt from the switch
node, causing some instability in
switch current levels at heavy loads.
L1’s inductance value is not critical;
a minimum of 10µH is suggested in
200ms/DIV
Figure 3. Transient response of DC/DC
converter with 2.5V input. Load step is 25mA
to 525mA.
90
88
VIN = 4V
86
EFFICIENCY (%)
L1
10µH
84
82
VIN = 3V
80
78
76
VIN = 2V
74
72
70
1
10
100
LOAD CURRENT (mA)
1000
1302_4.eps
Figure 4. Efficiency of Figure 2’s circuit.
7
DESIGN FEATURES
5
1 SEC
PEN A
OUTPUT
9 SEC
550mA
4
50mA
chart recordings. Many systems require high power for a short time, for
example, to spin up a hard disk or
transmit a packet of data. Figures 6,
7, and 8 present battery-life data with
a load profile of 50mA for 9 seconds
and 550mA for 1 second, as detailed
in Figure 5. At the chart speeds used,
individual ten-second events are not
discernable and the battery voltage
appears as a very thick line. Figure 6
shows operating life using a two-cell
alkaline (Eveready E91) battery. Battery voltage (pen B) drops 400mV as
the output load changes from 50mA
to 550mA. Battery impedance (330mΩ
when fresh) can be derived from this
data. After 63 minutes, the battery
voltage drops substantially below 2V
when the output load is 550mA, causing the output voltage (pen A) to drop.
The output returns to 5V when the
load drops to 50mA. The LT1302’s
undervoltage lockout prevents the
battery voltage from falling below 1.5V
until the battery is completely discharged (not shown on the chart).
A three-cell alkaline battery has a
significantly longer life, as shown in
Figure 7. Note that the time scale
here is one hour per inch. Usable life
is about 7.3 hours, a sevenfold improvement over the two-cell battery.
Again, battery impedance causes the
battery voltage (pen B) to drop as the
load changes from 50mA to 550mA.
The increasing change between the
loaded and unloaded battery voltage
over time is due to both increased
current demand on the battery as its
voltage decreases and increasing battery impedance as it is discharged.
Replacing the two-cell alkaline with
a two-cell NiCad (AA Gates Millennium) battery results in a surprise,
shown in Figure 8. Although these
AA NiCad cells have one-fourth the
8
3
2
PEN B
BATTERY
1
0
150
90
120
60
30
0
TIME (MINUTES)
Figure 6. Two-cell alkaline battery to 5V converter with load profile of Figure 5 gives 63
minutes operating life. Battery life decreases when 550mA load is applied; impedance is 330
milliohms when fresh. Output voltage drops at 550mA load after 63 minutes, but converter
can still deliver 50mA.
5
PEN A
OUTPUT
4
BATTERY/OUTPUT VOLTAGE (V)
Figure 5. Load profile for battery-life curves
in Figures 6 and 7.
BATTERY/OUTPUT VOLTAGE (V)
1302_5.eps
3
PEN B
BATTERY
2
1
0
9
8
7
6
5
4
3
2
1
0
TIME (HOURS)
Figure 7. Three-cell alkaline battery to 5V converter with pulsed load has 7.3 hours
operating life.
Linear Technology Magazine • June 1994
DESIGN FEATURES
5
PEN A
OUTPUT
4
BATTERY/OUTPUT VOLTAGE (V)
Two-Cell-to-12V Converter
Single-Cell to
5V/150mA Converter
Stand-alone, single-cell converters
can typically provide no more than
40mA–50mA at 5V from a single cell.
2
PEN B
BATTERY
1
0
150
120
90
30
0
1302_8.eps
Figure 8. Two-cell NiCad battery to 5V converter shows dramatically lower ESR of NiCads
compared to alkalines. Battery impedance is 80 milliohms. Although the 600 milliampere hour
NiCad has one fourth the energy of 2.4 Amp/hour alkalines, with 50mA/550mA loads NiCads
outlast alkalines by a factor of 2.8. Low cell impedance is maintained until the battery is
completely discharged.
NC
L1*
3.3µH
7
+
C3
0.1µF
C1
100µF
2 CELLS
D1
6
5
VIN
ILIM
SHDN
SW
+
C2
33µF
+
3
SHUTDOWN
LT1302
8
PGND
FB
GND
4
VC
1
2
RC
20k
C2
33µF
CC
0.02µF
R1
100k
1%
100pF
R2
866k
1%
12V/120mA
OUTPUT
*SEE TEXT C1 = AVX TPSD107M010R0100
C2 = AVX TPSD336M025R0200
D1 = MOTOROLA MBRS130LT3
COILCRAFT (708) 639-2361
L1 = COILCRAFT DO3316-332
FOR 3.3V/5V INPUT USE 22µH (DO3316-223)
1302_9.eps
Figure 9. Two-cell to 12V DC/DC converter delivers 120mA. Changing L1’s value allows
operation from 3.3V/5V supply.
85
300
275
80
VIN = 2.5V
VIN = 3V
75
VIN = 2V
70
65
250
225
200
175
150
125
60
1
10
100
LOAD CURRENT (mA)
Figure 10. Two-cell to 12V converter1302_10.eps
efficiency
Linear Technology Magazine • June 1994
60
TIME (MINUTES)
EFFICIENCY (%)
Portable systems with PCMCIA interfaces often require 12V at currents
of up to 120mA. Figure 9’s circuit can
generate 12V at over 120mA from a
two-cell battery. Operating the converter in continuous mode requires a
higher duty cycle than the LT1302
provides, so a very low inductance
(3.3µH) must be used in order to
provide enough output current in
discontinuous mode. Efficiency for
this circuit is in the 70–80% range, as
Figure 10’s graph shows. Battery life
at this power level would be short
with a continuous load, but the most
common application for this voltage/
current level, flash memory programming, has a rather low duty factor.
Maximum output current versus input voltage is shown in Figure 11. To
operate this circuit from a three-cell
battery, change L1’s value to 6.8µH.
This will result in lower peak
currents, improving efficiency
substantially.
By changing L1’s value to 22µH,
the circuit will operate from a 3.3V or
5V supply. Up to 350mA can be generated from 3.3V; 600mA can be
delivered from a 5V input. Efficiency,
pictured in Figure 12, exceeds 80%
over much of the load range, and
peaks at 89% with a 5V input.
3
LOAD CURRENT (mA)
energy of AA alkaline cells, operating
life is 2.8 times greater with the 50mA/
550mA load profile. Dramatically
lower battery impedance (80mΩ for
the NiCad, versus 330mΩ for the alkaline) is the cause. Battery voltage
(pen B) drops just 100mV as the
output load changes from 50mA to
550mA, compared to 400mV for
alkalines. Additionally, impedance
stays relatively constant over the life
of the battery. This comparison clearly
illustrates the limitations of alkaline
cells in high-power applications.
100
2.00 2.20
2.40 2.60 2.80 3.00 3.20 3.40
INPUT VOLTAGE (V)
Figure 11. Maximum load current of1302_11.eps
two-cell
to 12V converter versus input
9
DESIGN FEATURES
5V/175mA
OUTPUT
L1
3.3µH
D1
220Ω
100k
10Ω
2N3906
1.5V
CELL
90
100k
IL
88
SET
EFFICIENCY (%)
86
VIN = 5V
84
100k
VIN
VIN
SW1
SHDN
FB
SW
LT1073
56.2k
1%
LT1302
82
FB
80
GND
78
VIN = 3.3V
AO
74
VC
IL
SW2
PGND
+
76
R1
301k
1%
(169k FOR 3.3V)
C1
47µF
C2
220µF
GND
100pF
20k
4.99k
1%
+
0.1µF
0.01µF
36.5k
1%
72
70
10
100
LOAD CURRENT (mA)
1000
1302_12.eps
Figure 12. 3.3V/5V to 12V converter
efficiency
When more power is required, the
LT1302 can be used in conjunction
with a single-cell device.1 Figure 13’s
circuit operates from a single cell and
delivers 5V at 150mA output. Although the LT1302 requires a
minimum VIN of 2V, single-cell operation can be achieved by powering the
LT1302 from the 5V output. At startup, VOUT is equal to the cell voltage
minus a diode drop. The LT1073 initially puts the LT1302 in its shutdown
state. The LT1073 switches L1, causing L1’s current to alternately buildup
and dump into C2. When VOUT reaches
approximately 2V, the LT1073’s SET
pin goes above 212mV, causing AO to
go low. This pulls the LT1302’s SHDN
pin low, enabling it. The output, now
booted by the much higher power
LT1302, quickly reaches 2.4V. When
the LT1073’s FB pin reaches 212mV,
its switching action stops. The brief
period when the LT1073 and LT1302
are switching simultaneously has no
detrimental effect. When the output
reaches 5V, the LT1073 has ceased
switching. Circuit efficiency is in the
60–70% range, as shown in Figure 14.
10
L1 = COILCRAFT DO3316-332
D1 = MOTOROLA MBRS130LT3
C1 = AVX TPSD476M016R0150
C2 = AVX TPSE227M010R0100
COILCRAFT (708) 639-2361
1302_13.eps
Figure 13. Single-cell to 5V converter delivers 150mA. Changing R1 to 169K provides
3.3V at 250mA.
Three-Cell to 3.3V/12V
Buck-Boost Converter
Obtaining 3.3V from three cells is
not a straightforward task; a fresh
battery measures over 4.5V and a
fully depleted one 2.7V. Since battery
voltage can be both above and below
the output, common step-up (boost)
or step-down (buck) converters are
inadequate. Figure 15’s circuit provides an efficient solution to the
problem using just one magnetics
component, and also provides an
auxiliary 12V output. When the
LT1302’s switch is on, its SW pin goes
low, causing current buildup in T1D
and T1E (windings are paralleled to
achieve lower DC resistance). D1’s
anode goes to −VIN because of the
phasing of T1C/T1A relative to T1D/
T1E. C1 is charged to VIN. When the
switch opens, SW flies high to a voltage of VIN + VOUT + VDIODE. Energy is
transferred to the output by magnetic coupling from T1D/T1E to T1C/
T1A, and by current flowing through
C1. During this flyback phase, T1A/
T1C has 3.3V plus a diode drop across
the windings. T1B, which has a 3:1
turns ratio, has approximately
10V–11V impressed upon it. T1B
“stands” on the 3.3V output, resulting in about 13–14V at the input of
the LT1121 linear regulator, which
then precisely regulates the 12V output. Since this output is not directly
regulated by the LT1302, it cannot be
loaded without having at least a small
load on the directly regulated 3.3V
output. The LT1121 can be turned off
by pulling its SHDN pin low, isolating
the load from the output. Figure 16
shows the circuit’s efficiency for various input voltages.
72
70
VIN = 1.5V
68
66
EFFICIENCY (%)
1
64
VIN = 1.2V
62
60
58
56
54
52
50
48
1
10
100
LOAD CURRENT (mA)
1000
1302_14.eps
Figure 14. Single-cell to 5V converter
efficiency
Linear Technology Magazine • June 1994
DESIGN FEATURES
Construction Hints
The high-speed, high-current
switching associated with the LT1302
mandates careful attention to layout.
Follow the suggested component
placement in Figure 17 for proper
operation. High-current functions are
separated by the package from sensitive control functions. Feedback
resistors R1 and R2 should be close
to the feedback pin (pin 4). Noise can
easily be coupled into this pin if care
is not taken. If the LT1302 is operated
C3’s ground trace should not carry
switch current. Run a separate
ground trace up under the package
as shown. The battery and load return should go to the power side of
the ground copper. Adherence to these
rules will result in working converters with optimum performance.
off a three cell or higher input, R3 (2Ω)
in series with VIN is recommended.
This isolates the device from noise
spikes on the input voltage. Do not
put in R3 if the device must operate
from a 2V input, as input current will
cause the LT1302’s input voltage to
go below 2V. The 0.1µF ceramic bypass capacitor C3 (use X7R, not Z5U)
should be mounted as close as possible to the package. Grounding
should be segregated as illustrated.
1
Williams, Jim. “200ma Output, 1.5 to 5V Converter.” Linear Technology III:1 (February, 1993)
p. 17
VIN
2.5V-8V
2Ω
SHUTDOWN
SHDN
6
T1D
T1E
4
5
+ C3
D2
SW
100k
1%
2
+ C1
IL
GND
13V
0.1µF
LT1302
VC
100µF
16V
PGND
3
4700pF
25V
8
330k
1%
LT1121
+
ADJ
SHDN
+ C2
3.3µF
GND
330µF
6.3V
T1A
12V
120mA
OUT
+ 22µF
9
1
T1C
IN
T1B
D1
24k
200pF
47µF
16V
VIN
FB
169k
1%
7
150k
1%
10
3.3V OUTPUT
400mA
T1 =
D1, D2 =
C1 =
C2 =
C3 =
DALE LPE-6562-A069
MOTOROLA MBRS130LT3
AVX TPSE107016R0100
AVX TPSE337006R0100
AVX TPSD476016R0150
1:3:1:1:1 TURNS RATIO
DALE (605) 665-9301
1302_15.eps
Figure 15. Three-cell to 3.3V Buck-Boost converter with auxiliary 12V regulated output
80
VIN = 3.5V
75
EFFICIENCY (%)
VIN
R2
VIN = 2.5V
70
C3
R3
2Ω
L1
65
60
+
D1
50
4
3
6
LT1302
C1
VIN = 4.5V
55
5
7
2
8
1
R1
200pF
SHUTDOWN
RC
CC
+
C2
45
VOUT
40
1
10
100
LOAD CURRENT (mA)
1000
GND (BATTERY AND LOAD RETURN)
1302_17.eps
1302_16.eps
Figure 16. 3.3V buck-boost converter
efficiency
Linear Technology Magazine • June 1994
Figure 17. Suggested component placement for LT1302
11
DESIGN FEATURES
The LTC1066-1: Fourteen Bit DC
Accurate Elliptic Lowpass Filter
by Nello Sevastopoulos
Introduction
The LTC1066-1 is the first monolithic filter that combines RC active
techniques with switched-capacitor
technology to achieve outstanding DC
and AC performance. The LTC1066-1
is an eighth-order filter with clocktunable cutoff frequency to 100kHz,
1.5mV maximum DC offset over the
commercial temperature range, and
14 bits of DC gain linearity.
The Technology:
How Is It Done?
Figure 1 shows the block diagram
of the LTC1066-1 (a patent is pending
on the device). A high-speed, precision op amp at the device’s input, A1,
performs the following DC tasks: A1
stores the offset voltage of the eighthorder switched capacitor filter, the
offset voltage of the output unity gain
buffer, A2, and the DC voltage drop
across the feedback resistor, R F ,
across its feedback capacitor, C F. The
DC output offset of the overall filter is
then equal to the DC offset of the
input precision op amp, minus the
voltage drop across R F. The small
voltage drop across RF is caused by
the op amp’s input bias current.
A1 also performs the following AC
tasks. For frequencies above the cutoff frequency (1/2πR FC F of the external
RC), the input op amp serves two
purposes. First, it buffers the input
signal, VIN, and, second, it isolates or
disconnects the IC’s output, VOUT,
from the input terminal of the internal switched-capacitor network as the
input frequency is increased. With
increasing frequency, the AC gain of
the DC correcting loop tends towards
zero and the frequency response characteristics of the LTC1066-1 are
dictated only by the transfer function
of the internal switched-capacitor
network. The transition between DC
and AC is very critical and extreme
care is taken to make it as transparent to the user as possible.
It is important to appreciate the
stringent requirements on the input
op amp: A1 must not only have excellent DC characteristics but must also
be able to handle high-frequency
common-mode signals without introducing distortion.
The output high-frequency op amp,
A2, buffers the switched-capacitor
network and maintains low output
impedance over a wide range of frequencies. The low output impedance
preserves the stop-band characteristics of the internal switched-capacitor
network. The DC offset of the output
op amp is corrected by the loop, as
discussed above. Op amp A2 should
have at least the same AC performance as the input op amp.
LTC1066-1
DC Performance: Let the
Precision Op Amps Show
How Good They Really Are
The LTC1066-1 features both DC
accuracy and low output VOS. Filter
users often confuse DC accuracy with
low DC offset. DC accuracy preserves
the DC information of the input signal. DC accuracy does not imply low
DC offset. If the filter output offset
does not change with the DC value of
the input voltage, the filter is DC
accurate.
Large output DC offsets limit the
filter’s dynamic range. This is especially true with low power-supply
voltages. Adjusting the output offset
over a wide temperature range can be
cumbersome and costly. The DC performance of the LTC1066-1 is
primarily dictated by the DC characteristics of the input precision op
RF
CF
LTC1066-1
2
–
3
+
A1
VIN
V+
5,18
V–
4,10
1
GND 50/100 CLK
15
8
9
14
11
8TH ORDER
SWITCHEDCAPACITOR
NETWORK
6
13
12
–
7
+
A2
16
17
VOUT
PATENT PENDING
1066a_1.eps
Figure 1. LTC1066 functional diagram
12
Linear Technology Magazine • June 1994
DESIGN FEATURES
75
10
VS = ±7.5V
fCLK = 1MHz
fC = 20kHz
50
fCLK= 5MHz
–10
–20
fCLK= 500kHz
–30
0
GAIN (dB)
VIN – VOUT (µV)
25
0
–25
–50
–40
–50
–60
–70
–75
–80
–100
–90
–100
–125
–110
–6 –5 –4 –3 –2 –1 0 1 2 3
INPUT VOLTAGE (VDC)
4
5
6
VS = ±7.5V
fCLK/fC = 50:1
COMPENSATION
= 30k, 15pF
1k
fCLK= 2.5Mz
10k
100k
FREQUENCY (Hz)
1066a_2.eps
1M
1066a_3.eps
Figure 2. DC gain linearity of LTC1066-1 is
better than 14 bits
Figure 3. LTC1066-1 amplitude versus
frequency
amp. DC input voltages in the vicinity
of half the total power supply of the
filter are processed with exactly 0dB
(1V/V) of gain. Using a ±7.5V supply,
a ±5VDC input signal can be processed with better than 14-bit DC
gain linearity. Figure 2 displays the
(VIN(DC) − VOUT(DC)) error for an input
range of +6 to −6 volts. The LTC1066-1
cutoff frequency was set at 20kHz.
The LTC1066-1 output DC offset,
VOS(OUT), is measured with the input
grounded and with dual, symmetrical power supplies. VOS(OUT) is typically
100 microvolts and is optimized for
the combination RF = 20kΩ, CF =
0.1µF (see block diagram). The VOS
temperature drift is 7µV/°C.
VOS can be calculated from the
formula:
Where IB is the input bias current of
A1 (approximately 60nA).
The filter DC output offset voltage
is, for all practical purposes, independent of the clock frequency.
LTC1066-1 AC Performance
Clock Tunability
An external clock tunes the cutoff
frequency of the internal switchedcapacitor network. The device has
been optimized for a clock-to-cutofffrequency ratio of 50:1. The internal
double sampling greatly reduces the
risk of aliasing.
The maximum obtainable cutoff
frequency, fCUTOFF(MAX), depends on
power supply, clock duty cycle, and
temperature; fCUTOFF(MAX) does not
depend on the value of the external
VOS(OUT) = VOS(OA1) − RFIB
100k
CB
0.1µF
33µF
1
2
100k
3
VIN
–7.5V
7.5V
0.1µF
fCLK
OUT B
+IN A
+IN B
V–
V+
8
200Ω
–IN A
5
7
7.5V
V+
4
6
0.1µF
OUT A
9
LTC1066-1
CONNECT 1
FILTEROUT
50/100
CLK
GND
FILTERIN
COMP 2
CONNECT 2
COMP 1
V–
18
7.5V
17
0.1µF
VOUT
16
15
14
13
RC
30k
CC
15pF
12
11
10
0.1µF
resistor/capacitor combination R FCF.
For the commercial temperature range
(0 to +70°C) and ±7.5V supplies, the
maximum obtainable cutoff frequency
is 100kHz (see Figure 3). The R CCC
compensation, as shown in Figure 4,
is needed only for cutoff frequencies
above 60kHz. The data detailed in
Figure 3 reveals the important fact
that for a cutoff frequency of 100kHz,
the stopband attenuation still remains
greater than 70dB for input frequencies up to 1MHz.
The minimum obtainable cutoff
frequency depends on the R FCF time
constant of the servo loop. For a given
RFCF time constant, the minimum
obtainable cutoff frequency of the
LTC1066-1 is:
fCUTOFF(MIN) = 250(1/2πR F CF).
fCUTOFF(MAX) = 100kHz
For instance, if R F = 20kΩ, CF = 1µF,
fCUTOFF(MIN) = 2kHz, and fCLOCK(MIN) =
100kHz.
Under the these conditions, a clock
frequency below 100kHz will “warp”
the passband gain by more than
0.1dB. Please see the LTC1066-1 data
sheet for more details.
Figure 4 shows an application allowing clock tunability from 10Hz to
100kHz. The R CCC frequency compensating components maintain a flat
passband for cutoff frequencies between 50kHz and 100kHz. The input
resistor reduces the output DC offset
caused by the op amp bias current
through the 100kΩ feedback resistor.
The measured DC offset and the gain
nonlinearity are 4mV and ±0.0063%
(84dB), respectively. The 0.1µF bypass capacitor, CB, helps keep the
total harmonic distortion of the filter
from being degraded by the 100kΩ
input resistor. The frequency compensation components (30kΩ, 15pF)
maintain a flat passband for cutoff
frequencies all the way up to 100kHz.
Dynamic Range
–7.5V
The LTC1066-1 wideband noise is
100 microvolts RMS. Figure 5 shows
the noise plus distortion versus RMS
input voltage at 1kHz. With a ±5V
MAXIMUM OUTPUT VOLTAGE OFFSET = 4mV, DC LINEARITY = ±0.0063%, TA = 25°C.
THE PIN 6 TO 12 CONNECTION SHOULD BE UNDER THE IC AND SHIELDED BY AN
ANALOG SYSTEM GROUND PLANE.
RC COMPENSATION BETWEEN PINS 11 AND 13 REQUIRED ONLY FOR fCUTOFF > 50kHz.
THE 33µF CAPACITOR IS A NONPOLARIZED, ALUMINUM ELECTROLYTIC, ±20%, 16V
(NICHICON UUPIC 330MCRIGS OR NIC NACEN 33M16V 6.3 × 5.5 OR EQUIVALENT).
1066a_4.eps
Figure 4. DC-accurate, 10Hz to 100kHz, eighth-order elliptic lowpass filter, fCLK/fC = 50:1
Linear Technology Magazine • June 1994
13
DESIGN FEATURES
(
)
20 log THD + NOISE (dB)
VIN
–45
–50
–55
–60
fIN = 1kHz
fCLK = 1MHz
fCLK /fC = 50:1
VS = ±5V
–65
–70
–75
VS = ±7.5V
–80
–85
–90
0.1
1
INPUT VOLTAGE (VRMS)
5
1066a_5.eps
Figure 5. LTC1066-1 dynamic range
supply, the filter can swing ±2.5V (5V
full scale) with better than 0.01%
distortion plus noise. The maximum
signal-to-noise ratio, in excess of
90dB, is achieved with ±7.5V supplies. Unlike previous monolithic
filters, the data shown in Figure 5 is
taken without using any input or
output op amp buffers. The output
buffer of the LTC1066-1 can drive a
200Ω load without dynamic-range
degradation.
filter in front of the LTC1066-1 is
required. The anti-aliasing filter
should do precisely what it is meant
to do, that is, provide band-limiting.
The anti-aliasing filter should not
degrade the DC or AC performance of
the LTC1066-1.
For fixed-cutoff-frequency filter applications, the anti-aliasing function
is quite trivial. Figure 7 shows the
precision input op amp used to perform both the DC-accurate function
of the LTC1066-1 and as a secondorder, Butterworth anti-aliasing
lowpass filter. The cutoff frequency of
the RC anti-aliasing filter is set three
times higher than the cutoff frequency
of the LTC1066-1. For the example
The LTC1066-1 crams filter
performance usually found in multiple-package, RC active designs into
a single 18-pin SOIC. The filter is an
eighth-order elliptic low pass filter
with fCUTOFF useful to 100kHz. It
boasts true 14-bit gain linearity along
with DC accuracy. The LTC1066-1
will replace larger, more expensive,
and less accurate solutions in instrumentation, data acquisition, and other
types of circuitry.
–80
fCLK – fC fCLK + fC
2fCLK – 2.3fC
2fCLK
2fCLK + 2.3fC
2fCLK – fC 2fCLK + fC
INPUT FREQUENCY
Aliasing and Anti-Aliasing
14
Conclusions
–60
fCLK
All sampled-data systems will alias
if their input signals exceed half the
sampling rate, but aliasing for
high-order, band-limited, switchedcapacitor filters need not be a serious
problem. The LTC1066-1, when
operating with a 50:1 clock-to-cutofffrequency ratio, will have significant
aliasing only for input signals centered around twice the clock frequency
and its even multiples. Figure 6 shows
the input frequencies that will generate aliasing at the filter output. For
instance, if the filter is tuned to a
50kHz cutoff frequency using a
2.5MHz clock, significant aliasing will
occur only for input frequencies of
5MHz ±50kHz. The filter user should
be aware of the spectrum at the input
to the filter. Next, an assessment
should be made as to whether a
simple, continuous-time anti-aliasing
shown in Figure 7, the input antialiasing filter provides a 62dB
attenuation at twice the clock frequency of the switched-capacitor filter.
0
ALIASED OUTPUT (dB)
–40
1066a_6.eps
Figure 6. Aliasing versus frequency fCLK/fC = 50:1 (pin 8 to V+) Clock is a 50% duty-cycle square
wave.
20k
C
10nF
R1
2.37k
1µF
1
2
R2
9k
3
VIN
C1 –7.5V
3.3nF
7.5V
V+
–IN A
OUT B
+IN A
+IN B
4
V–
5
V+
6
7
8
fCLK = 100kHz
OUT A
9
LTC1066-1
CONNECT 1
FILTEROUT
50/100
CLK
GND
FILTERIN
COMP 2
CONNECT 2
COMP 1
V–
18
17
7.5V
VOUT
16
15
14
13
12
11
10
–7.5V
1066a_7.eps
Figure 7. Adding a 2-pole Butterworth input anti-aliasing filter. Set C1 = 0.33C, R2 = 3.8 × R1;
f−3dB (input anti-aliasing) = 0.8993/(2πR1C)
Linear Technology Magazine • June 1994
DESIGN FEATURES
Micropower, 12-Bit, SO-8 ADCs Now
Available for Three-Volt Systems
by William C. Rempfer
and Marco Pan
Portable/Battery Systems:
a Hot Item
One of the fastest growing hightech markets today is that for portable
and battery-operated systems. Manufacturers are scrambling to introduce
new products into this growing area.
Cellular phones, portable computers, personal digital assistants (PDAs),
and portable industrial equipment
are expected to be among the hottest
new products. Even the financial community is aware of this trend, looking
to invest in companies that make
products for these markets. The companies that can capitalize on this
demand can reap huge benefits.
Component suppliers are also
working very hard to develop products that provide value to the system
manufacturers. “Providing value”
simply means helping system designers and their companies succeed
against their competition and succeed in providing the best solutions
to their customers. What these system designers need most from
components is small size, low power,
low cost, and 3V operation.
A/D Converters
in Portable Systems
indirectly, for example to monitor the
system’s health by monitoring voltages and temperatures inside an
instrument. Regardless of the use of
the ADC, system designers have had
a real struggle getting small ADCs at
low enough power levels and at low
cost. To get it all to work on 3V is a
further difficulty.
Linear Technology recently brought
some relief by introducing the world’s
first 12-bit, micropower ADCs in SO8 packages: the LTC1286 and
LTC1298. These two converters provide the micropower, small-size,
low-cost conversion eagerly awaited
by designers of 5V systems. But there
is also a growing need for 3V ADCs.
Now LTC is meeting that need by
releasing a similar pair of ADCs designed for 3V systems: the LTC1285
and LTC1288. This article will discuss some of the system designer’s
challenges and how these converters
meet them.
Meeting the ADC
Needs of Portables
Micropower Operation:
160µA and Auto-Shutdown
Small Size: SO-8
Portable systems use A/D converters. Many use ADCs directly, as when
digitizing the pen-screen input in a
pen-based computer or digitizing the
sensor output in a portable gas meter
or detector. Others use ADCs more
Until recently, it has been impossible to get any 12-bit ADC in an SO-8
sized package. Designers had to suffer with converters in the larger sized
packages such as 16- and 20-pin
SOICs and 8-pin DIPs. The 8-pin deLTC1288
LTC1285
VREF
1
8
VCC
CS/SHDN
1
8
VCC (VREF)
IN+
2
7
CLK
CH0
2
7
CLK
IN –
3
6
DOUT
CH1
3
6
DOUT
GND
4
5
CS/SHDN
GND
4
5
DIN
SO-8
SO-8
1285_1a.eps
1285_1b.eps
Figure 1. These micropower, 3V, 12-bit ADCs come in SO-8 packages and offer one or two
input channels
Linear Technology Magazine • June 1994
vices (such as the LTC1292/LTC1297)
were the most popular because they
were the smallest available at the
time. Their serial I/O made them more
efficient than parallel devices. However, they were still much larger than
the desired SO-8 and were not surface mountable.
The first SO-8, micropower, 12-bit
ADCs on the market were the
LTC1286/LTC1298, introduced by
LTC for 5V systems. The two new
products, the LTC1285 and LTC1288,
bring the same features to 3V systems, again for the first time. As Figure
1 shows, the LTC1285 is a singlechannel, 12-bit device. The LTC1288
has 2 channels. These devices are
already seeing great success in portable applications because of their
size. The total board space of the 12bit ADC (with a single 0.1µF bypass
chip capacitor) is just 0.08 square
inches. The height is also very small,
at 1.7mm.
Micropower, 12-bit A/D conversion is another area that has not been
addressed until recently. In the past,
designers had to cycle the ADCs on
and off to try to achieve micropower
performance. Figure 2 shows an example of this method. A higher power
ADC is switched on and off to reduce
power drain during inactive periods.
This approach has three drawbacks.
First, it requires switching hardware
and a switching signal. Second, while
the ADC power supply is switched off,
any digital or analog inputs to the
ADC need to be disabled or switched
off to prevent forward biasing the
substrate diodes of the ADC (the ADC
15
DESIGN FEATURES
OFF/ON
VDD
ANALOG
INPUT
10µF
SUBSTRATE
DIODE
POWER-HUNGRY
12-BIT ADC
GND
fSAMPLE
ICC = C • V • fSAMPLE
1Hz
10Hz
50µA
500µA
LTC1285 AT 3V
160µA
will load the signals tied to its inputs
and this can cause latchup problems
or excessive current drain). Third,
most power-hungry ADCs require a
large bypass capacitor to be connected
at the supply pin. The charging and
discharging of this capacitor in each
power-down cycle draws a very large
current, ruining the dissipation of the
circuit even at very low sample rates.
For example, a 10µF bypass capacitor switched at a 10Hz rate consumes
500µA.
The first solution to the problem
was provided with auto-shutdown on
low-to-medium power converters (e.g.,
the LTC1297). Here, the ADC is tied
directly to the supply and shuts itself
off whenever it is not converting. The
switching hardware is eliminated, the
need to disable input signals is removed, and the current drain in the
bypass capacitor is eliminated because the cap is never switched. The
power drain is simply the “ON” current of the converter multiplied by
the duty cycle (the percentage of time
it is on). The auto-shutdown is invisible to the user, which is nice.
In many new systems, operating
power becomes more important because the ADC is in use much of the
time. The LTC1286/LTC1298 were
the first devices on the market to offer
the ideal combination: micropower
20µA
2µA
200nA
10Hz
1285_2.eps
Figure 2. High-side switching a power-hungry
ADC takes hardware and wastes power.
Repeatedly switching the required bypass
capacitor consumes 500 microamps even
when taking readings at only 10Hz.
16
operating currents and auto-shutdown. The new LTC1285/LTC1288
bring this same performance to 3V.
Drawing only 160µA at full speed,
they provide the lowest power alternative available. Figure 3 shows the
current drain from a 3V supply versus sample rate. The current drops
linearly with sample rate from 160µA
at 7.5kHz to 2µA at 100Hz. At 10Hz it
draws only 200nA, compared to the
500µA of the switched ADC example
of Figure 1.
SUPPLY CURRENT
5V
3V Operation Guaranteed
100Hz
1kHz
7.5kHz
SAMPLE RATE
Before 3V ADCs became available,
designers had to have a 5V supply on
the board. There was a strong incentive to eliminate the 5V supply entirely.
Now that is possible. LTC has had a
family of 3V ADC products for some
time (as shown in Table 1), but this is
the first time anyone has offered the
combination of 12-bits, micro-power,
SO-8 and 3V all in one device. The
LTC1285/1288 are designed, specified, and tested for 3V.
Low Cost by Design
Cost is a big concern for designers
of portable and battery-operated systems. These systems are typically
high-volume, low-cost products, so
the components they use must be
suitably priced. The LTC1285/
LTC1288 are designed to meet this
need. They use an architecture that
guarantees excellent differential linearity, low drift, and no missing codes,
without expensive calibration, trimming, or processing. They are priced
1285_3.eps
Figure 3. The auto shutdown feature of the
LTC1285/1288 makes the ADC’s supply
current drop as its sample rate is reduced.
The 160 microamp drain at 7.5kHz drops to 2
microamps at 100Hz.
to make them attractive to high volume users.
Conclusion
Designers have had trouble finding 12-bit ADCs which meet any one
of the four criteria we’ve talked about:
small size, micropower operation, 3V
operation, and low cost. But to find
an ADC which meets all four constraints at once has been impossible
until now. There are huge opportunities for system designers in the
portable and battery operated marketplace, if they can meet the
challenges required of them. The
LTC1285 and LTC1288 can help any
designer facing this challenging situation succeed.
Table 1. The 3V ADC family now includes the LTC1285/LTC1288, which combine 12-bits,
micropower operation, 3V, and SO-8 in one device.
3V Device
LTC1096/8
LTC1196
LTC1198
LTC1283
LTC1287
LTC1289
LTC1282
LTC1285/8
Bits
8
8
8
10
12
12
12
12
Speed
20KSPS
383KSPS
287KSPS
15KSPS
30KSPS
25KSPS
140KSPS
7.5KSPS
Channels
1/2
1
2
8
1
8
1
1/2
Package
SO-8
S0-8
SO-8
SO-20
8-DIP
SO-20
SO-24
SO-8
Supply
Current
100µA
2mA
2mA
150µA
1mA
1mA
4mA
120µA
Auto
Shutdown?
Yes
No
Yes
No
No
Software
No
Yes
Linear Technology Magazine • June 1994
DESIGN IDEAS
A Low-Power, Low-Voltage
CCFL Power Supply
Most recently published CCFL
driver circuits require an input supply of 7V to 20V and are optimized for
bulb currents of 5mA or more. This
precludes lower power operation from
two- or three-cell batteries often used
in PDAs, palmtop computers, and the
like. A CCFL power supply that operates from 2 to 6V is shown in Figure
1. This circuit can drive a small
(75mm) CCFL over a 100µA to 2mA
range.
DESIGN IDEAS
A Low-Power, Low-Voltage
CCFL Power Supply ....... 17
Steve Pietkiewicz
LTC1262 Generates 12V for
Programming Flash
Memories without Inductors
...................................... 18
The circuit uses an LT1301 micropower DC-to-DC converter IC in
conjunction with a current-driven,
Royer-class converter comprising T1,
Q1, and Q2. When power is applied
along with intensity-adjust voltage
VA, the LT1301’s ILIM pin is driven
slightly positive, causing maximum
switching current to flow through the
IC’s internal switch pin (SW). L1 conducts current, which flows from T1’s
center tap, through the transistors,
into L1. L1’s current is deposited in
switched fashion to ground by the
regulator’s action.
The Royer converter oscillates at a
frequency set primarily by T1’s characteristics (including its load) and the
0.068µF capacitor. L1 sets the magnitude of the Q1–Q2 tail current, and
hence, T1’s drive level. The 1N5817
diode maintains L1’s current flow
when the LT1301’s switch is off. The
by Steve Pietkiewicz
0.068µF capacitor combines with L1’s
characteristics to produce sine-wave
voltage drive at the Q1 and Q2 collectors. T1 furnishes voltage step-up
and about 1400VP–P appears at its
secondary. Alternating current flows
through the 22pF capacitor into the
lamp. On positive half-cycles the
lamp’s current is steered to ground
via D1. On negative half-cycles the
lamp’s current flows through Q3’s
collector and is filtered by C1. The
LT1301’s ILIM pin acts as a zero summing point, with about 25µA bias
current flowing out of the pin into C1.
The LT1301 regulates L1’s current to
maintain equality of Q3’s average collector current, representing one-half
the lamp current, and R1’s current,
represented by VA/R1. When VA is set
to zero, the I LIM pin’s bias current
forces about 100µA bulb current.
Anthony Ng and Robert Reay
Active-Negation Bus
Terminators .................. 18
9
7
22pF
3kV
TI
Dale Eagar
Extending Op Amp Supplies
to Get More Output Voltage
...................................... 20
1
VIN
2V - 6V
1Ω
5
4
120Ω
1N5817
3
2
0.068µF
Dale Eagar
CCFL
ZTX849
Simple PCMCIA VPP
Socket Switching .......... 23
Doug La Porte
DC-Accurate, Clock-Tunable,
Lowpass Filter............... 24
NC
GND
Philip Karantzalis
+
LT1301
SHDN
10µF
2N3904
ILIM
PGND
+
1µF
Regulated Charge-Pump
Power Supply ................ 26
Tommy Wu
Low-Noise WirelessCommunications Power
Supply ........................... 26
Mitchell Lee and Kevin Vasconcelos
Linear Technology Magazine • June 1994
WIMA
MKP20
ZTX849
SW
SENSE
0.1µF
L1
47µH
SELECT
VIN
7.5K
1%
1N4148
LT1300 TA6
SHUTDOWN
0V - 5V DC IN
INTENSITY ADJUST
100µA TO 2mA BULB CURRENT
T1 = COILTRONICS CTX110654-1
L1 = COILCRAFT D03316-473
COILTRONICS (407) 241-7876
COILCRAFT (708) 639-2361
Figure 1. CCFL power supply
17
DESIGN IDEAS
LTC1262 Generates 12V for
Programming Flash Memories
without Inductors
by Anthony Ng and Robert Reay
Flash memories require a +5V VCC
supply and an additional +12V supply for write or erase cycles. The +12V
supply can be a system supply, or be
generated from the +5V supply using
a DC–DC converter circuit. Singlesupply flash memories (i.e., those with
the +12V converter built-in) are available, but these memories have lower
capacities and slower write/erase performance, and are therefore not as
popular as memories without a builtin 12V supply. Flash memories require
that the +12V supply be regulated to
within 5% and not exceed the permitted maximum voltage (14V
for Intel ETOXTM memories). The
LTC1262 offers a simple and cost
effective +12V programming supply
to meet these requirements.
Figure 1 shows a typical application circuit. The only external
components needed are four surfacemount capacitors. The LTC1262 uses
a triple charge-pump technique to
convert +5V to +12V. It operates
from 4.75V to 5.5V and delivers 30mA
while regulating the 12V output to
within 5%. The TTL-compatible SHDN
1
0.22µF
0.22µF
C1–
SHDN
8
7
GND
LTC1262
6
3
VOUT
C2 –
4
5
+
VCC
C2
2
C1
18
FLASH
MEMORY
VPP
+
4.7µF
+
4.7µF
VCC
VCC
(4.75V TO 5.5V)
dI 1262_1.eps
Figure 1. Typical LTC1262 application circuit
pin can be driven directly by a microprocessor. When the SHDN pin is
taken high (or floated), the LTC1262
enters shutdown mode. In this state,
the supply current of the LTC1262 is
reduced to 0.5 microamps typical and
the +12V output drops to VCC. When
SHDN is taken low, LTC1262 leaves
shutdown mode and the output rises
to 12V without any potentially harmful overshoot (see Figure 2).
The LTC1262 is available in both
8-pin DIP and narrow SO packages.
With small surface-mount capacitors,
the complete +12V supply takes up
very little space on a printed circuit
board. In power sensitive applications, such as PCMCIA flash cards
for portable PCs, the LTC1262
Active-Negation
Bus Terminators
High-speed data buses require
transmission-line techniques, including termination, to preserve signal
integrity. Lost data on a bus can be
attributed to reflections of the signals
from the discontinuities of the bus.
The solution to this problem is proper
termination of the bus.
Early designs of bus terminators
were passive (see Figure 1). Passive
termination works great, but wastes
FROM MICROPROCESSOR
+
lots of precious power, especially when
the bus is not being used.
The ideal solution is a voltage
source capable of both sourcing and
sinking current. Such a voltage
source, with termination resistors, is
shown in Figure 2. This is called
active negation. Active negation uses
minimal quiescent current, essentially providing only the power needed
to properly terminate the bus.
shutdown current is low enough to
preclude the need for external switching devices when the system is
inactive.
A = SHDN
5V/DIV
B = VOUT
2V/DIV
1ms/DIV
Figure 2. LTC1262 taken in and out
of shutdown
by Dale Eagar
Active-Negation Bus
Terminator Using Linear
Voltage Regulation
The active-negation circuit shown
in Figure 3 provides the power to the
output at an efficiency of about 50%;
the rest of the power is dissipated in
either Q1 or U1, depending on the
polarity of the output current.
The circuit will source or sink current. Current is sourced from the
Linear Technology Magazine • June 1994
DESIGN IDEAS
VCC = 5V
VCC
VCC
0.1µF
BUS
LINE
1k
BUS
LINE
Q1
TIP121
1nF
CR1
1N4001
1
COLL
3
V+
dI ACT_1.eps
U1
LT1431
Figure 1. Passive-termination technique
BUS
LINES
2.85V
8
+
•
•
•
+ 100µF
6.3V
–
BUS
LINE
BUS
LINE
2.5V
5k
7
SGND
5
E 2.85V
R1
680Ω
dI ACT_2.eps
dI ACT_3.eps
Figure 2. Active-negation termination
technique
+5V supply through Q1, an NPN
Darlington, to the output. The sink
current flows through CR1 into the
collector (pin 1) of the LT1431, and to
ground. The LT1431 regulates a
scaled version of the output voltage
against the internal 2.5V bandgap
reference, driving the base of Q1 or
drawing current through CR1 to regulate the output voltage. R1 and the
internal 5kΩ resistor of the LT1431
scale the output voltage.
Switching-Power-Supply,
Active-Negation Network
The switching, active-negation terminator shown in Figure 4 is a
synchronous switcher. This solution
further reduces dissipation and therefore achieves higher efficiency. This
type of switcher can both source and
sink current.
The switching power supply operates as follows. The 74AC04 hex
inverters (U1 and U2) form a 1MHz,
variable-duty-factor oscillator. The
duty factor is controlled by the output of the regulator, U3, and is
maintained at the ratio of 2.85V/VIN.
VIN is the +5V supply, which powers
U1, U2, and U3. The output voltage is
the average voltage of the square wave
(VIN × duty factor) from the outputs of
Linear Technology Magazine • June 1994
FGND
6
Figure 3. Linear active-negation voltage source
C1
20pF
U1B
39k
•
•
•
U1F
U1A
CR1
1N5818
U2A
+
•
•
•
U2F
5V
4.7µF
+
14
74AC04
U1
7
C2
100µF
6.3V
L1
40µH
BUS
LINES
VIN = 5V
20k
3.9k
0.1µF
•
•
•
5V
5V
4.7µF
+
3
10k
1
14
74AC04
U2
7
8
U3
LT1431
4
5
6
7
24.9k
0.01µF
5.1k
dI ACT_4.eps
Figure 4. Switching active-negation terminator
U1B–U1F and U2A–U2F. L1 and C2
filter the AC component of the 0-to5V signal, yielding a DC output voltage
of 2.85V.
CR1 is added to prevent latchup of
U1 and U2 during adverse
conditions.
A logic gate could easily be added
to the oscillator to add a disable function to this terminator, further
lowering the quiescent power when
termination is not needed.
19
DESIGN IDEAS
Extending Op Amp Supplies to Get
More Output Voltage
by Dale Eagar
We often hear of applications that
require high-output-voltage, low-output-impedance amplifiers. Here is a
topology that allows you to extend an
op amp’s output voltage swing while
still maintaining its short circuit protection. The trick is to suspend the op
amp between two MOSFET source
followers so that the supply voltages
track the op amp’s output voltage
(see Figure 1). The circuit shown in
Figure 1 will perform very nicely with
any run-of-the-mill ideal op amp. The
problem is in the lead times of ideal
op amps—they just keep getting
pushed out to later dates.
Non-Ideal op amps have realistic
lead times and can be made to work
in the extended-supply mode. They
have bandwidth limitations in both
CMRR and PSRR. The circuit shown
in Figure 2 implements the extended
supply as shown in Figure 1 and has
several additional components: C1 is
added to decouple the supply, improving high frequency PSRR; R3 and
R5 decouple the gates of Q1 and Q2
from AC ground, preventing Q1 and
Q2 from running off together to redirect local air traffic; R1, R2, and C4
form a snubber to de-Q the two pole
system formed by the Miller capacitance of Q1 and Q2 and the high
frequency CMRR of IC1; additionally,
R4, R6, C2, C3, Z1, and Z2 form the
two 15V voltage sources (E1 and E2
in Figure 1); CR1 and CR2 are protecV+
N-CHANNEL
MOSFET
+
E1 15V
–
OP AMP
+
–
+
OUT
E2 15V
–
P-CHANNEL
MOSFET
V–
dale_1.eps
Figure 1. Block diagram of suspended-supply
op amp
20
V+
R6
100k
R5
100Ω
Q1
IRF610
C2
0.1µF
7
+
R2
R1
6
C1
0.1µF
Z1
15V
CR1
C4
IC1
–
4
Z2
15V
CR2
C3
0.1µF
R3
100Ω
Q2
IRF9610
R4
100k
V–
dale_2.eps
Figure 2. Detailed high-voltage op amp
tion diodes that allow the output to be
instantaneously shorted to ground
when the output is at any output
voltage.
The values of R1, R2, and C4 vary
with the MOSFETs’ Miller capacitance
and with the high frequency CMRR of
the op amp used. They are selected to
minimize the overshoot in the step
response of the amplifier.
High-Voltage,
High-Frequency Amplifier
Using the LT1227 current-feedback
amplifier (CFA) in the extended-supply mode as shown in Figure 2, it is
relatively easy to get a 1MHz power
bandwidth at 100VP–P (see Figure 3
for component values). This circuit
has short-circuit protection and is
stable into all capacitive loads.
If One Is Good,
Are Two Better?
Dual and quad op amps can also
be configured with extended supplies,
although the design gets just a wee
bit tricky. When extending supplies
of multiple stages and/or complete
circuits, some design rules need to
change. Op amp circuits generally
require a ground against which to
reference all signals. The problem
encountered when using extendedsupply mode is that “ground” is
swinging through the common-mode
range of the op amp and beyond. This
raises the following question: “If I
cannot reference the signals to
ground, to what can I reference them?”
The answer? “Use the output as the
signal reference.” This works for all
stages except the last stage, where
R7
10k
R8
1k
IN
2
–
LT1227
3
6
OUT
+
R10
1k
R9
9.1k
SUPPLIES SUSPENDED
AS IN FIGURE 2 WITH
R1 = 200Ω , R2 = 1.6K,
AV =
= −11 C4 = INF
R8 R9 – R7 R10
(
R8 R9 + R10
)
dale_3.eps
Figure 3. High-speed suspended op amp
Linear Technology Magazine • June 1994
DESIGN IDEAS
R11
R12
–
–
=
E1
+
E1
R11
+
LOAD
LOAD
R12
A. CONVENTIONAL
B. SUSPENDED
dale_4.eps
Figure 4. Inverting amplifiers (A. conventional/B. suspended)
Ring-Tone Generator
using the output as the reference
would simply discard the signal. In
the last stage, ground is effectively
the output and the feedback resistor
is R12. This is shown in Figures 4a
and 4b. Figure 4a shows a conventional inverting amplifier where the
input and output signals are referred
to ground. Figure 4b shows the equivalent circuit implemented in the
extended-supply mode.
Here are two rules for design in the
extended-supply mode, which will be
demonstrated in the next application:
Rule 1: When designing multiple
stages in the extended-supply mode,
reference the signals of all stages
except the last to the output of the
last stage.
Rule 2: Invert the signal using the
circuit in Figure 4b at the last stage.
Ring-tone generators are sinewave-output, high-voltage inverters
for the specific purpose of ringing
telephone bells. In decades past, the
phone company generated their ring
tones with motor generator sets with
the capacity to ring numerous phones
simultaneously. Often, ring tones are
20Hz at 90V with less than 10mA per
bell output current capability. Since
the power supplied is low, one would
think that the task is minimal. This is
not always so. “It’s simple—no problem,” is often heard in response to
queries about ring-tone generators.
“Just hook a couple of logic-level FETs
to two spare output bits of the microprocessor and hook their drains to
the primary side of a transformer,
with the center tap hooked to 5V, or
12V, or whatever.” At this point everyone is happy until the transformer
comes in. After a few phone calls to
make sure that the transformer maker
shipped the right one, the engineer
(face covered with egg) asks if anyone
needs a rather large paperweight. The
engineer (still wiping egg from his
face) then decides to use switching
power supply technology to solve this
“simple” problem.
Here is a simple ring-tone generator that can be turned on and off with
a logic signal. It has a fully isolated
output, is short-circuit protected, and
can be powered by any input voltage
from 3V to 24V.
How It Works
Suspended along with the dual op
amp in Figure 5 are two voltage references and an oscillator. Keep in mind
when referring to Figure 5 that the
node labeled “A” is the output; this is
the reference common for the references, the oscillator, and the first
lowpass filter (U1a). The two references VR1 and VR2 produce ±2.5V.
The oscillator U2, running on the
±2.5V references, produces a 20Hz
square wave rail-to-rail. U1a is a second-order, Sallen-and-Key lowpass
filter that knocks off the sharp edges,
presenting the somewhat smoothed
signal at point “B.”
Next comes the tricky stuff. U1b is
a second-order, multiple-feedback
(MFB) lowpass filter/amplifier that
0.047µF
110V
1N4001
3
A
+
A
100k
10k
51k
100Ω
0.01µF
IRF610
4
A
15V
0.1µF
0.1µF
8
2k
100V
0.047µF
4
15V
0.1µF
U1 = LT1078
LEFT TRIANGLE
SHOWS POWER
INPUT ONLY
0.15µF
51k
VCC
6
U2
CMOS
555
OUT
VR2
LT1004
–2.5V
B
–
8
R
A
U1
6.8k
VR1
LT1004
–2.5V
1
U1A
2
11k
A
360k
2
6
3
10k
R99
10k 5
VSS
–
U1B
7
+
1
100Ω
100k
360k
0.33µF
IRF9610
C99
0.01µF
100V
LOAD
10k
A
A
1N5817
–110V
dale_5.eps
Figure 5. Ring-tone generator: oscillator, filter, and driver.
Linear Technology Magazine • June 1994
21
500
2.025
460
1.95
420
1.875
380
1.8
340
1.725
300
1.65
260
1.575
220
1.5
180
1.425
140
1.35
100
1.275
60
18 : 112
INPUT
3V TO 15V
MUR120
POWER (W)
INPUT I (mA)
DESIGN IDEAS
1.2
3
6
9
12 15 18 21 24
INPUT VOLTAGE (V)
27
30
dale_6.eps
Figure 6. Input-current and input power
versus input voltage while ringing one bell,
for circuit shown in Figure 5.
performs four functions: first, it subtracts the voltage at point “A” (its own
output voltage) from the voltage at
point “B” (the incoming signal), forming a difference that is the signal;
second, it filters the difference signal
with a two-pole lowpass filter, smoothing out the last wrinkles in the signal;
third, it amplifies the filtered difference signal with a gain of 34; and
fourth, it references the amplified signal to ground, forming the output.
Note that R99 shown in Figure 5 is
there to protect the input of U1b in
the event that the output is shorted
when the output voltage is very high.
This measure is necessary because
the bottom end of C99 is connected to
ground, and C99 could have up to
100V across it. When the output is
shorted to ground from a high voltage, R99 limits the current into the
input of U1b to an acceptable level.
This circuit, when coupled with
the switching power supply shown in
Figure 7, implements a fully isolated
sine-wave ring-tone generator.
The input current and power versus input voltage for the combination
ring-tone generator (Figures 5 and 7)
are shown in Figure 6. The output
waveform (loaded with one bell) is
shown in Figure 8, and the harmonic
distortion is shown in Figure 9.
Although somewhat tricky at first,
extended-supply mode is a valuable
tool to get out of many tight places.
There is also a great deal of satisfaction to be gathered when making it
work, for those of you who love a
technical challenge.
22
+
H = SHUTDOWN
110V
68µF
20V
1
10k
2N3904
VC
0.22µF
160V
5
VIN
LT1070
0.01µF
VSW
2
1N5817
0.22µF
160V
MUR120
4
–110V
MUR120
TRANSFORMER:
EFD25/1319 - 3C8 CORES
15MIL GAP ALL LEGS
FB
510Ω
GND
3
100Ω
1/4W
0.01µF
63V
2
PRI 50µH
18T
SEC
112T
SEC
dale_7a.eps #35
112T
PRI 50µH
18T
Figure 7. High-voltage power supply for ring-tone generator
* #26
#35
2
* #26
dale_7b.eps
Figure 8. Ring-tone generator frequency-spectrum plot
Figure 9. Sine-wave output from ring-tone generator
Linear Technology Magazine • June 1994
DESIGN IDEAS
Simple PCMCIA VPP Socket Switching
for Line-Operated Systems by Doug La Porte
PCMCIA card sockets are not only
for portable systems. Many line-operated systems are using PC cards for
easier system software upgrades or
feature additions. For line-operated
systems requiring a PCMCIA card
socket, there is often a higher voltage
supply (>13V) that can be converted
to the required VPP voltages with a
linear regulator and some additional
logic. Switching supply configurations
could be used, but often EMI considerations and circuit simplicities
outweigh the need for better efficiency.
Figure 1 shows a circuit that uses a
commonly available 24V supply and
the LT1121 adjustable regulator.
The Circuit
The circuit uses a simple linear
voltage regulator to achieve four VPP
voltage states: 0V, 5V, 12V, and a
high-impedance state. The LT1121
has the ability to supply the necessary currents at the VPP pin, and has
current- and thermal-limiting features. The limiting protection can be
very important, because the designer
can never be sure of the condition of
the PC card being plugged into the
socket. To achieve the high-imped-
ance state, the LT1121’s shutdown
feature is used. In this mode, the
circuit consumes only 16µA. The 0V
state is attained by switching Q1 while
the LT1121 is shutdown. The 5V and
12V states are achieved by grounding
R3 and R4, respectively. The capacitor C2 slows the rise and fall times of
the output voltage to minimize overshoot and assures compliance with
flash memory requirements.
The Logic
The logic shown in Figure 1 interfaces directly to PCMCIA logic
controllers. There are many ways to
implement the required logic. The
method shown can be implemented
using some spare logic from a PLD,
gate array, or ASIC. The pulldown
resistors R5 and R6 force the output
to the high-impedance state if the 5V
logic supply fails.
The three-state output HC126
gates must be CMOS. The three-state
condition allows the unselected resistor to float, and a CMOS logic low
level is required to pull the selected
resistor to ground. These two gates
should be located close to the resistors and the LT1121. If spare logic
cannot be close to the LT1121, the
EN0 and EN1 signals can be buffered
remotely to drive two logic level
MOSFET’s close to the LT1121, as
shown in Figure 2.
Thermal Considerations
Connecting the LT1121 directly to
the 24V supply is not recommended.
The maximum voltage rating of the
part is not exceeded in this configuration, but a minimum of 12V will be
dropped across the regulator. At a
current draw of 120mA the part will
have to dissipate 1.44W of power. The
“A” suffix SO-8 part specified here
has a junction-to-ambient thermal
resistance of 60°C/W (that of the “nonA” suffix SO-8 part is 120°C/W) with
a maximum operating junction temperature of 125°C. This would put the
maximum operating ambient temperature at 38°C. This is usually not
acceptable. Two 4.3V, 1W zener diodes lower the voltage drop across
the LT1121A. In this configuration
the input voltage to the LT1121 will
be at about 15V to 16V, depending on
the current requirement, greatly decreasing the power dissipation of the
part.
MLL4731,4.3V ZENER
D1
EN0
EN1
D2
24V
IN
+
10µF
EN0
EN1
HC126 TYPE
R5
100k
OUT
LT1121A
C2
200pF
R1
121k
1%
R5
100k
VPP
+
1µF
R3
SHDN ADJ
GND
0V (100Ω)
5V (30mA)
12V (120mA)
Hi-Z (150k)
ILIM = 250mA
R3
365k, 1%
2N7002
R4
EN
R4
56.2k, 1%
HC126 TYPE
EN
R6
100k
R2
100Ω
2N7002
EN1 EN0 VPP
0
0 0V
0
1 12V
1
0 5V
1
1 Hi-Z
2N7002
Q1
2N7002
REMOTE
LOGIC
DIPCM_1.eps
Figure 1. 24V-input PCMCIA socket VPP switch/regulator
Linear Technology Magazine • June 1994
R2
R6
100k
DIPCM_2.eps
Figure 2. Using additional MOSFETs for
remote logic interface
23
DESIGN IDEAS
DC-Accurate, Clock-Tunable,
Lowpass Filter with Input
Anti-Aliasing Filter
by Philip Karantzalis
LTC1066-1. For the the LTC1066-1 is clock tuned to a
situation when the cutoff frequency of 5kHz (with a clock
LTC1066-1 clock fre- frequency of 250kHz), the 50kHz RC
quency is 500kHz, an filter will provide 20dB attenuation
RC filter with the for aliasing inputs in the range of
–60
–3dB frequency set at 500kHz ±5kHz. Therefore, a first-or–80
50kHz attenuates by der lowpass RC filter will attenuate all
26dB any possible aliasing signals to the LTC1066-1 by
fCLK – fC fCLK + fC
2fCLK – 2.3fC
2fCLK 2fCLK + 2.3fC
aliasing inputs in the a minimum of 20dB for a clock-tunfCLK
2fCLK – fC 2fCLK + fC
range
1MHz ±10kHz. able range of one octave.
INPUT FREQUENCY
For added anti-aliasing bandwidth,
The passband shape
Figure 1. Aliasing versus frequency fCLK/fC = 50:1 (pin 8 to V+);
a
first-order,
lowpass RC filter can be
of
the
50kHz
RC
filter
Clock is a 50% duty-cycle square wave.
does not degrade the tuned by the clock signal of LTC1066In a sampled-data system the sam- flat passband of the LTC1066-1 at 1 to follow the cutoff frequency of the
pling theorem says that if an input 10kHz (the passband attenuation of higher-order filter. The circuit is
signal has any frequency components the 50kHz RC filter for frequencies shown in Figure 2. The circuit operagreater than one half the sampling less than 10kHz is less than 0.2dB). If tion is as follows. The six comparators
frequency, aliasing errors will appear
at the output. In practice, aliasing is
not always a serious problem. HighTable 1. Component calculations for the circuit in Figure 3.
order switched-capacitor lowpass
Definitions: 1. The cutoff frequency of the LTC1066-1 is abbreviated as fC.
filters are band-limited and significant aliasing occurs only for input
2. fC LOW is the lowest cutoff frequency of interest.
signals centered around the clock
3. A range of five octaves is from fC LOW to 32 × fC LOW.
frequency and its multiples.
Figure 1 shows the LTC1066-1
aliasing response when operated with
Component Calculations:
a clock-to-fC ratio of 50:1. With a 50:1
f
1
RIN = RF (if RF can be chosen as 20k
= C LOW
ratio, the LTC1066-1 samples its in2πRFCF
250
RIN and CIN are not needed)
put twice during one clock period and
the effective sampling frequency is
C1 = 1 µF
(fC LOW in Hz); R1 = 1k
fC LOW
twice the clock frequency. Figure 1
shows that the maximum aliased outC2 = C1 ±5%; C3 = 2 × C1 ±5%; C4 = 4 × C1 ±5%; C5 = 8 × C1 ± 5%
put is generated for inputs in the
105 kΩ
CP = 50pF; RP =
range of 2 × fCLK ±fC. (fC is the cutoff
50 fC LOW
frequency of the LTC1066-1.) For in5 × 105 kΩ
stance, if the LTC1066-1 is
CA = 0.047µF; RA =
50
fC LOW
programmed to produce a cutoff frequency of 20kHz with a 1MHz clock,
Example:
for a five-octave range from 1kHz to 32kHz—
maximum aliasing will occur only for
fC LOW = 1kHz
input signals in the narrow range of
Let
CF = 1µF ±20%, then RF = 40.2kΩ ±1%. RIN = RF = 40.2kΩ ±1%, CIN = 0.1µF
2MHz ±20kHz and its multiples.
C1 = 0.001µF ±5%, C2= 0.001µF ±5%, C3 = 0.0022µF ±5%
The simplest anti-aliasing filter is
a passive, first-order lowpass RC filC4 = 0.0039µF ±5%, C5 = 0.0082µF ±5%
ter. The −3dB frequency of the RC
CP = 50pF, RP = 2kΩ ±%, CA = 0.047µF, RA = 10kΩ ±1%
filter should be chosen so that the
passband of the RC filter does not
%,±CA = 0.047µF, RA = 10kΩ ±
CP = 50pF, RP = 2kΩ
influence the passband of the
ALIASED OUTPUT (dB)
0
1066_1.eps
24
Linear Technology Magazine • June 1994
DESIGN IDEAS
inside the LTC1045 detect the clock
frequency. The clock signal of the
LTC1066-1 is converted to a pulse
output whose duty cycle changes with
clock frequency. The average voltage
of the pulse signal is delivered to a
four-window comparator whose outputs drive the four analog switches of
5V
0.1µF
1µF
+
1
12.1k
the LTC202. When the LTC1066-1
clock frequency increases or decreases
by more than one octave (2x or x/2),
a capacitor is switched in or out of
the first order lowpass filter formed
by resistor R1 (1kΩ) and capacitor
C1. The −3dB frequency of the lowpass RC filter is therefore doubled or
halved if the cutoff frequency of the
LTC1066-1 is doubled or halved. Resistor R1 and capacitors C1 through
C5 allow the lowpass RC filter to be
tuned over a range of five octaves,
providing at least 20dB attenuation
to any LTC1066-1 input signals in
the range 2 × fCLK ±fC (the RC filter also
attenuates all aliasing signals near
any multiples of the clock frequency).
The circuit in Figure 2 can be used
for any clock-tunable, five-octave
range for cutoff frequencies from 10Hz
to 80kHz (with ±5 volt supplies for
LTC1066-1) or for cutoff frequency as
high as 100kHz (with ±8 volt supplies
for the LTC1066-1). For cutoff frequencies greater than 50kHz, a 15pF
capacitor in series with a 30kΩ resistor should be connected between pins
11 and 13 of the LTC1066-1 to minimize passband gain peaking. Table 1
provides a design guide for choosing
the component values of R A, R P, R F,
R IN, C F, C1 through C5, CP, and CA.
20
LTC1045
2
+
19
0.1µF
–
2k
0.1µF
3
+
18
–
FIRST ORDER RC
LOWPASS ANTI-ALIASING FILTER
–5V
5V
0.1µF
0.1µF
1k
4
4
+
17
0.1µF
–
500Ω
1
VIN
13
2
CLOCK-TUNABLE,
8TH ORDER LOWPASS FILTER
R1
1k
LTC202
3
C2
14
C3
11
C4
6
C5
C1
20Ω
16
15
RF
9
0.1µF
5
10
+
16
9
1
8
2
7
–
CF
RIN
4
5
500Ω
12
10
3
11
12
5V
CIN
0.1µF
13
5
6
0.1µF
7
8
200Ω
CLOCK INPUT
(TTL OR CMOS)
LTC1045
6
9
LTC1066-1
18
OUT A
V+
17
–IN A
OUT B
16
+IN A
+IN B
15
–
V
GND
14
FIN
V+
13
CON 1
COMP 2
12
F OUT
CON 2
11
50/100
COMP 1
10
CLK
V–
VOUT
0.1µF
–5V
0.1µF
20Ω
0.1µF
+
15
8
–
CP
50pF
7
PULSE
AVERAGE
+
14
RA
RP
CA
0.047µF
–
PULSE
OUTPUT
1066_3.eps
CLOCK FREQUENCY DETECTOR
Figure 2. DC-accurate, clock-tunable lowpass filter with input anti-aliasing.
Linear Technology Magazine • June 1994
25
DESIGN IDEAS
Regulated Charge-Pump
Power Supply
by Tommy Wu
–5
VIN = 5V
–4
1
2
+
1µF
8
LTC1044A
5V INPUT
+
1µF
7
3
6
4
5
2N2219
47kΩ
OUTPUT (V)
NO FEEDBACK
–3
–2
100kΩ
WITH FEEDBACK
–1
–1.7V OUTPUT
–0
10µF
0
5
10
LOAD (mA)
15
20
+
dI 1044_1.eps
dI 1044_2.eps
Figure 1. Regulated charge pump
The circuit shown in Figure 1 uses
an LTC1044A charge pump inverter
to convert a +5 volt input to a −1.7V
potential as required for a certain
LCD panel. Output regulation is provided by a novel feedback scheme,
which uses components Q1, R1, and
R2. Without feedback, the charge
pump would simply develop approximately −5V at its output. With
feedback applied, VOUT charges in the
negative direction until the emitter of
Q1 is biased by the divider compris-
ing R1 and R2. Current flowing in the
collector tends to slow the LTC1044A’s
internal oscillator, reducing the available output current. The output is
thereby maintained at a constant voltage.
In this application less than 5mA
output current is required. As a result, charge-pump capacitor C1 is
reduced to 1 microfarad from the
usual 10 microfarads. Curves of output voltage with and without feedback
are shown in Figure 2. The equivalent
Figure 2. Effect of feedback on output voltage
output impedance of the charge pump
is reduced from approximately 100
ohms to 5 ohms.
A variety of output voltages within
the limits of the curve in Figure 2 can
be set by simply adjusting the VBE
multiplier action of Q1, R1, and R2.
Tighter regulation or a higher tolerance could be obtained by adding a
reference or additional gain, at the
expense of increased complexity
and cost.
Low-Noise Wireless-Communications
Power Supply
by Mitchell Lee and Kevin Vasconcelos
Shown in Figure 1 is a 5V power
supply we designed for a synthesizer
oscillator. The original design used a
three-terminal regulator, but the
regulator’s voltage noise contributed
to excessive phase noise in the oscillator, leading us to this solution. Of
prime importance is noise over the
10Hz-to-10kHz band. Careful measurements show a 40dB improvement
over standard three-terminal regulators.
The regulator is built around a 5V,
buried-zener reference. It is the buried zener’s inherently low noise that
makes the finished supply so quiet.
Measured over a 10Hz-to-10kHz
band, the 5V output contains just
7µVRMS noise at full load. The 10Hz–
26
10kHz noise can be further reduced
to 2.5µVRMS by adding a 100µH,
1000µF output filter. The noise characteristics of the reference are tested
and guaranteed to a maximum of
11µV over the band of interest.
An external boost transistor, the
ZBD949, provides gain to meet a
200mA output current requirement.
Current limiting is achieved by
ballasting the pass transistor and
clamping base drive. Although our
application only requires 200mA, it
is possible to extend the output current to at least 1A by selecting an
appropriate ballast resistor and
addressing attendant thermal considerations in the pass transistor.
9V-12V
INPUT
+
220Ω
4.7Ω
47µF
RED LED**
ZBD949*
10Ω
1/2W
IN
LT1021-5
OUT
5V/200mA
OUTPUT
2.0Ω
GND
+
10µF
TA
* ZETEX INC (516) 864-7630
** GLOWS IN CURRENT LIMIT. DO NOT OMIT.
dI 1021_1.eps
Figure 1. Ultra low noise 5V, 200mA Supply
Output Noise is 7 microvolts RMS over a
10Hz to 10kHz bandwidth. Reference noise is
guaranteed less than 11 microvolts RMS.
Standard, three-terminal regulators have 100
times the noise and no guarantees.
Linear Technology Magazine • June 1994
DESIGN FEATURES
LT1251/LT1256, continued from page 1
through and cause ghosts. The most
common way to ensure that only the
selected input is on is to overdrive the
control input below zero and above
100%. Unfortunately, this means that
the gain is not a simple function of the
control signal. The LT1251 eliminates
this problem with special circuitry
that ensures that, for a control signal
of 2% or 98%, only one input is on and
the other is completely off. The LT1256
does not have this special circuit and
is linear from zero to 100%.
The LT1251/LT1256 has flat response (0.1dB) from DC to 5MHz and
low differential gain and phase (0.1%
and 0.1°) for composite video applications. The signal-to-noise ratio (5MHz
bandwidth) is 80dB referenced to 1V.
The −3dB bandwidth is greater than
30MHz, ideal for computer RGB applications, and the absolute gain
accuracy is better than 3%. The
LT1251/LT1256 operate on a single
+5V supply as well as on dual supplies from ±5V to ±15V. The output is
able to drive all resistive loads, including a doubly terminated cable.
The LT1251/LT1256 can accommodate a variety of control signals
and levels. Zero to one or zero to ten
volts are common, as are bipolar signals around zero. Some systems use
current inputs or voltages connected
to resistors feeding the summing node
of an op amp. Often, in variable-gain
amplifier applications, several control inputs are summed together. In
RG1
order to make the LT1251/LT1256
compatible with as many systems as
possible, the input and output of an
internal op amp are available to the
user.
The LT1251/LT1256 has
flat response from DC to
5MHz and low differential
gain and phase for
composite video
applications...
The —3dB bandwidth is
greater than 30MHz, ideal
for computer RGB
applications
maximum gain for each input is set
by external resistors. The input stages
use current feedback, and the external resistors set both the gain and
bandwidth.
For ideal op amp parameters and
R F1 = R F2, the equation for the gain of
Figure 2 is:
VO = K × IN1 × AV1 + (1 − K) × IN2 × AV2
The closed-loop feedback topology
of the LT1251/LT1256 minimizes control feedthrough and DC shift at the
output. The typical feedthrough is
only 2.5mV peak-to-peak and the DC
shift between inputs is guaranteed to
be less than 4.5mV.
Circuit Description
Figure 2 shows the block diagram
of the LT1251/LT1256 signal path.
The potentiometer of Figure 1 has
been replaced with the mathematical
equivalent blocks: K, 1 − K, and summation. K is a constant determined
by the control circuit and can be any
value between 0 and 1. To make the
LT1251/LT1256 more versatile, the
where A V1=1+R F1 /R G1 and A V2 =
1+R F2/R G2
This shows that as K goes from 0 to
1, the output fades linearly from input 2 times its gain, to input 1 times
its gain. The gains are set by the
external resistors in the same way
that the gain of an op amp is set. For
the inverting case, A V1 and A V2 are
−R F/R G, again just like a standard op
amp.
The complete gain equation with
all the gory details is derived in the
LT1251/LT1256 data sheet.
Figure 3 shows the control-circuit
block diagram. The LT1251/LT1256
consists of two identical voltage-tocurrent converters (V-to-I); each V-to-I
contains an op amp, an NPN transistor and a resistor. The converter on
the right generates a full scale current, IFS; the one on the left generates
a control current, I C. The ratio IC/IFS
is K; K ranges from a minimum of zero
(when IC is zero) to a maximum of one
(when I C is equal to or greater than
IFS). The parameter K determines the
gain from each signal input to the
output.
RF1
V+
2
–
1
+
K
IN1
Σ
14
IN2
8
OUT
VC
3
IC
+
C
+
IFS
+
12
VFS
FS
–
–
1–K
13
RG2
–
IC
RF2
RC
4
5
11
RC
5k
CONTROL V TO I
CONTROL
RFS
5k
IFS
10
RFS
FULL SCALE V TO I
1251_3.eps
SET K FROM 0 TO 1
1251_2.eps
Figure 2. LT1251/1256 signal-path block diagram
Linear Technology Magazine • June 1994
Figure 3. LT1251/1256 Control circuit block diagram
27
DESIGN FEATURES
5
4
1
+
2
14
LT1251/LT1256
–
2
1
+
–
13
CONTROL
0V - 2.5V
CONTROL
IC
3
4
+
–
C
IC
+
FS
–
IFS
5
NULL
V–
6
12
11
2.5V
REF
IFS
10
5k
5k
7
RF2
1.5k
RF1
1.5k
100% down to 0%. The worst-case
error in K (the gain) is ±3%, including
initial accuracy and temperature effects. By using a 2.5V full-scale voltage
and the internal resistors, no additional errors need be accounted for.
In the LT1256, K changes linearly
with IC. To ensure that K is zero, VC
must be −15mV or lower to overcome
the worst-case control-op-amp offset. Similarly, to ensure that K is
100%, VC must be 3% larger than VFS,
based on the guaranteed gain
accuracy.
To eliminate the overdrive requirement, the LT1251 has internal
circuitry that senses when the control current is at 5% and sets K to 0%;
similarly at 95% it sets K to 100%.
The LT1251 guarantees that 2%
(50mV) input ensures K = zero and
98% (2.45V) results in K = 100%.
The operating currents of the
LT1251/LT1256 are derived from IFS
and therefore the quiescent current
is a function of VFS and RFS. An approximate formula for the supply
current is:
10
where VS is the total supply voltage
between pin 9 and 7. Using the internal resistors (5k) with VFS equal to
2.5V results in IFS equal to 500µA and
a supply current of 14.5mA on ±15V
supplies. The supply current can be
reduced by reducing IFS, but the slew
rate and bandwidth will also be reduced. There is no reason to use a
larger value of IFS.
Performance
Figure 4 shows the LT1251/
LT1256 configured as a fader with
unity gain. A full-scale voltage of 2.5V
is applied to pin 12 and the control
input drives pin 3. Figure 5 shows the
typical linearity of LT1256 gain; the
worst-case error, including temperature effects, is ±3%. Figure 6 shows
the frequency response for both the
ON and OFF channels. At 5MHz the
OFF channel is down 60dB to prevent
ghosts from haunting us.
The control path has a 10MHz
bandwidth. Figure 7 shows the
10
6
– 30
INPUT 1 = SMALL SIGNAL
INPUT 2 = GND
– 50
– 70
VC = 0V
100k
VS = ±5V
AV = 1
RF = 1.5kΩ
RL = 100Ω
10M
1M
FREQUENCY (Hz)
100M
1251_6.eps
Figure 6. LT1251/1256 Gain versus frequency
VOLTAGE DRIVE VC
VS = ±5V
6
4
2
0
–2
PIN 4 NOT IN SOCKET
–4
4
2
0
–2
–4
–6
–6
–8
–8
–10
10k
100k
10M
1M
FREQUENCY (Hz)
VOLTAGE DRIVE RC
VC = GND
VS = ±5V
8
VOLTAGE GAIN (dB)
VC = 2.5V
VOLTAGE GAIN (dB)
VOLTAGE GAIN (dB)
IS = 1mA + 24 × IFS + VS/20k
10
8
28
–2
1251_5.eps
Figure 5. LT1256 Gain accuracy versus
control voltage
1251_4.eps
Figure 4. Two-input video fader
– 90
10k
0
–1
V
– 4 GAIN ACCURACY (%) = AVMEAS – C x 100
2.5
–5
0.5
1.5
0
2.0
2.5
1.0
CONTROL VOLTAGE (V)
V+
VOUT
– 10
2
1
–3
9
8
The op amp in each V-to-I converter drives the transistor until the
voltage at the inverting input is the
same as the voltage at the noninverting input. If the open end of the
resistor (pin 5 or 10) is grounded, the
voltage across the resistor is the same
as the voltage at the noninverting
input. The emitter current is therefore equal to the input voltage (VC)
divided by the resistor value (R C). The
collector current is essentially the
same as the emitter current; it is the
ratio of the two collector currents that
sets the gain.
The LT1251/LT1256 is tested with
pins 5 and 10 grounded and a fullscale voltage of 2.5V applied to VFS
(pin 12). This condition sets IFS at
approximately 500µA; the control
voltage (VC) is applied to pin 3. When
the control voltage is negative or zero,
IC is zero and K is zero. When VC is
2.5V or greater, IC equals I FS and K is
one. The gain of channel one ranges
from 0% to 100% as VC increases
from zero to 2.5V. The gain of channel
two moves the opposite way, from
VFS = 2.5V
3
IN2
GAIN ACCURACY (%)
IN1
100M
1251_7.eps
Figure 7. LT1251/1256 Control-path
bandwidth
–10
10k
100k
10M
1M
FREQUENCY (Hz)
100M
1251_8.eps
Figure 8. LT1251/1256 Control-path
bandwidth
Linear Technology Magazine • June 1994
DESIGN FEATURES
50Ω
0.1µF
LT1256
1
1MHz
CARRIER
+
2
–
+
–
13
RF2
1.5k
CONTROL
0.1µF
3
AUDIO
MODULATION
220k
4
+
–
C
+
FS
–
IFS
IC
5
6
220k
VOUT
12
2.5V REF
11
5k
9
NULL
V–
RF1
1.5k
10
5k
7
V+
8
200k
0.1µF
V+
LT1077
+
1251_9.eps
V–
Applications
Grounding IN2 of the LT1256 in
Figure 4 results in a two-quadrant
multiplier. Figure 9 shows the twoquadrant multiplier being used as an
AM modulator. The output will deliver +10dBm into 50 ohms. The
LT1077 op amp senses the LT1256
output DC and drives the null pin,
eliminating any DC at the output.
The null-pin voltage is nominally
100mV above the negative supply and
therefore the op amp output must be
able to swing within a few millivolts of
the negative supply. Without the
LT1077, the worst-case DC output
voltage is 50mV.
14
2
1
–
response while driving pin 3; the peaking is caused by the stray capacitance
from pin 4 to ground. This capacitance is in parallel with the internal
5k resistor between pins 4 and 5.
Grounding pin 3 and driving pin 5
eliminates the effect of the stray capacitance to ground, since pin 4
becomes a virtual ground. Figure 8
shows the true response of the control path. The control path is fast
enough for quick switching between
signals, as when keying on a color or
luminance level. The control path introduces only a small (50mV), short
(50ns) glitch when switched quickly.
A summary of the LT1251/LT1256
performance operating on ±5V supplies in the configuration shown in
Figure 4 is given in Table 1.
Figure 9. AM modulator with DC-output nulling circuit
LT1256
1
14
+
RG1
1.5k
2
MODULATION
–
+
1
2
–
13
RF2
1.5k
CONTROL
3
1MHz
CARRIER
0.1µF
10k*
4
+
–
C
IC
IFS
+
FS
–
5
6
12
10k
VOUT
2.5V REF
11
0.1µF
10
5k
5k
9
*TRIM FOR SYMMETRY
V–
RF1
1.5k
7
V+
8
1251_10.eps
Table 1. LT1251/LT1256 performance
summary
Slew Rate (@ ±2V, RL = 150Ω)
300V/µs
Full-Power Bandwidth (1VRMS)
30MHz
Small-Signal Bandwidth
30MHz
Differential Gain (NTSC, RL = 150Ω)
0.1%
Differential Phase (NTSC, RL = 150Ω) 0.1DEG
Total Harmonic Distortion (1kHz, K = 1) 0.001%
(1kHz, K = 0.5) 0.01%
(1kHz, K = 0.1) 0.4%
Rise Time, Fall Time
11 ns
Overshoot
3%
Propagation Delay
10 ns
65 ns
Settling Time (0.1%, VO = 2V)
Quiescent Supply Current
13.5mA
Linear Technology Magazine • June 1994
Figure 10. Four-quadrant multiplier used as a double-sideband, suppressed-carrier modulator
By operating one input stage in an
inverting configuration and the other
in a noninverting configuration and
driving both inputs, the LT1256 becomes a four-quadrant multiplier.
Figure 10 shows the four-quadrant
multiplier being used as a
double-sideband, suppressed-carrier
modulator. The LT1077 DC-outputnulling circuit could be added if
necessary.
The LT1251/LT1256 can be used
to implement numerous other functions, including voltage-controlled
filters, phase shifters and oscillators.
Squaring and limiting circuits can be
designed by feeding the output or input into the control pins. Gamma
correction and other compression circuits are created in a similar manner.
The applications are limited only by
the designer’s imagination.
29
NEW DEVICE CAMEOS
New Device Cameos
LTC1347 Ultra-Low-Power 5VPowered RS232 Transceiver
transceiver will operate at speeds up and LTC1483 use a unique fabricawith Five Receivers Active
to 50kbaud with the maximum tion process and design that includes
in SHUTDOWN Mode
2500pF, 3kΩ load, or as fast as Shottky diodes in series with the MOS
The LTC1347 is a new three-driver,
five-receiver RS232 interface transceiver with an integral charge-pump
power generator for single 5V supply
operation. The circuit has exceptionally low power consumption: 300µA
in normal operation, and only 80µA
in SHUTDOWN mode. All five receivers remain active during SHUTDOWN
to allow monitoring of all data lines
for incoming data. The LTC1347 is
pin compatible with the LT1137A,
and operates with only four 0.1µF
charge-pump capacitors.
The LTC1347 meets or exceeds all
EIA/TIA-232 specifications. Operation at data rates up to 120k baud is
guaranteed. Slew rate with a 3kΩ
2500pF load is a minimum of 3V/µs.
When powered down or in SHUTDOWN, the driver outputs remain
high impedance for line voltages to
±25V.
Like all Linear Technology RS232
transceivers, the LTC1347 is protected
against ±10kV ESD strikes to the
RS232 inputs and outputs. This integrated ESD resistance saves the
expense and space of external protection devices.
The circuit is available in 28-lead
DIP, SOIC, and SSOP packages.
The LTC1348—
True RS232 from 3.3V
The LTC1348 is a three-driver, fivereceiver RS232 transceiver designed
to work from a single 3.3V supply. It
uses a voltage-tripling charge pump
to generate true RS232 output swings
from a supply as low as 3.0V, while
drawing only 500µA of quiescent current—the lowest in the industry. It
also features low-current shutdown
and receiver keep-alive modes for
additional power savings. The charge
pump requires only five space-saving
0.1µF capacitors to operate. The
30
120kbaud with a 1000pF, 3kΩ load.
It will withstand repeated ESD strikes
of up to ±10kV at the driver outputs
and receiver inputs without damage.
The transceiver operates in one of
four modes: normal, receiver-alive,
receiver-disable, and shutdown. In
normal or receiver-disable modes,
supply current is only 500µA with all
RS-232 outputs unloaded. In receiver-alive mode, all five receivers
are kept alive and the supply current
is reduced to 12µA. Shutdown drops
supply current to less than 1µA.
All RS-232 outputs assume high impedance states in shutdown or
receiver-alive modes, or when the
power is off. The receiver outputs
assume high impedance states in receiver-disable or shutdown modes.
The LTC1348 is available in 28-pin
DIP and SSOP packages.
LTC1481/LTC1483—
Lowest Power RS485
Transceivers Yet
The LTC1481 and LTC1483
are the latest members of Linear
Technology’s growing family of CMOS
RS485 interface devices. Both the
LTC1481 and LTC1483 are pin compatible with the industry standard
75176 pinout. They achieve significant power savings by cutting the
quiescent current when the receiver
is active and by providing a shutdown
mode that reduces the current consumption to below 1µA. The LTC1481
and LTC1483 have a maximum
quiescent current of 120µA in receiver-active mode, more than four
times lower than any other RS485
transceiver. The low power consumption of the LTC1481 and LTC1483
makes them the lowest supply current RS485 transceivers available
today. Similar to the other members
of Linear’s 485 family, the LTC1481
output transistors, allowing the outputs to maintain high impedance
when forced up to ±7V beyond the
supply rails or when the power is off.
The LTC1481 features half-duplex
operation at up to 5M baud, with
receiver input propagation delay of
less than 200ns. The LTC1483 features the same receiver speed as the
LTC1481, but its driver slew rate is
deliberately slowed down to reduce
EMI levels in the transmitted signal.
Both the LTC1481 and LTC1483 will
enter shutdown mode if the driver
and receiver are disabled at the same
time. The LTC1481 and LTC1483 are
offered in 8-pin DIP and SOIC packages, in both commercial and
industrial temperature grades.
Introducing the LT1169
Dual Op Amp—High Input
Impedance with Low Voltage
and Current Noise.
The LT1169 is a low-noise, precision, dual-JFET operational amplifier.
The low voltage noise, (6nV/√Hz), is
better than that of most bipolar op
amps and the low bias current, (2pA),
is better than that of most JFET op
amps. Unlike most monolithic JFET
op amps, the input bias current is
essentially independent of commonmode voltage (delta IB = 2pA over the
input common-mode range of −10
volts to 13 volts). The device has a
very high input resistance (1013Ω)
and the input capacitance is less than
2pF, assuring high gain linearity when
buffering AC signals from high
impedance transducers. The combination of low voltage and current
noise makes the LT1169 the first
choice for amplifying low-level signals from very high impedance
transducers.
The LT1169 is unconditionally
stable for gains of one or more, even
Linear Technology Magazine • June 1994
NEW DEVICE CAMEOS
with 1,000pF capacitive loads. The
low IB of JFET op amps inherently
results in low current noise at the
expense of other parameters, but this
is not the case with the LT1169. Each
individual amplifier is 100% tested
for voltage noise, slew rate (4.1V/
microsecond), and gain bandwidth
product (4.8MHz). Other key features
are 0.65mV VOS and a voltage gain of
three million.
A full set of matching specifications is provided for precision
instrumentation amplifier front ends.
Specifications at ±5 volt supply operation will also be provided.
LTC1159
High Efficiency Synchronous
Regulator Controller
The LTC1159 is the newest member of Linear Technology’s family of
above 90%-efficiency, stepdown DCto-DC converters. The LTC1159
features an extremely wide, 4V-to40V input operating-voltage range
and reduced supply currents. The
quiescent current while regulating
the output is 250µA, and current in
shutdown drops to only 20µA. The
combination of low supply currents
and high-input voltage capability is
ideal for battery-powered applications
that require high-voltage AC wall
adapters.
A unique EXTVCC pin on the
LTC1159 allows the MOSFET drivers
and control circuitry to be powered
from an external source, such as the
output of the regulator itself. Deriving control and driver power from
VOUT improves efficiency at high input voltages because the resulting
current drawn from VIN is scaled
by the duty cycle of the regulator.
During start-up or short-circuit conditions, operating power is supplied
by an internal 4.5V low-dropout
linear regulator. This regulator automatically turns off when the EXTVCC
pin is pulled above 4.5V.
The LTC1159, like other members
of the LTC1148 family, automatically
switches to Burst ModeTM operation
at low output currents to maintain
greater than 90% efficiency over two
Linear Technology Magazine • June 1994
LTC in the News...
Another Record Quarter for
Linear Technology
Thanks again to your outstanding support, Linear Technology
posted record sales and earnings for
the third quarter of fiscal 1994, which
ended April 3, 1994.
Robert H. Swanson, Jr., LTC President and CEO, announced that sales
increased 33% and profits increased
59% over the third quarter of fiscal 1993. Sales were a record
$51,667,000 and net income was
$15,217,000, or 40 cents per share,
compared with sales of $38,806,000
and net income of $9,571,000, or 26
cents per share, for the third quarter
of fiscal 1993.
According to Bob Swanson, “We
have once again reported record financial results. The general business
climate is good. Geographically our
sales were particularly strong overseas, fueled largely by offshore
manufacturing for U.S. companies.
Our business continues to be broadly
based across end markets, with some
additional strength this quarter in a
variety of solutions for desktop and
notebook computers. Our operating
income exceeded 40% of sales for the
first time in our history and we generated an additional $8.8 million in
cash and short-term investments.”
decades of load current range. The
LTC1159 also features a constant offtime architecture and complementary
power MOSFETs drivers. This combination results in a switching regulator
with a dropout voltage lower than
that of most linear low-dropout regulators.
To illustrate the flexibility of the
LTC1159, the data sheet shows applications for high-efficiency 2.5V/5A,
3.3V/2.5A, 5V/10A, and 12V/5A
regulators. It also includes a highefficiency, dual-output application for
obtaining up to 17W combined output power from 3.3V and 5V outputs.
The LTC1159 is available in fixed
5V, fixed 3.3V, and adjustable versions. Package options include
16-lead DIP, 16-lead narrow SOIC,
and 20-lead SSOP.
In its “1994 CEO of the Year”
competition, Financial World magazine presented LTC President & CEO
Bob Swanson a bronze award in the
Electronics and Semiconductor category, ranking him with the CEO of
Intel.
UPSIDE magazine included Linear
Technology at 76th on its list of the
best 200 technology companies in
the U.S.
The Los Angeles Times will include
Linear Technology in its 1994 “Top
100 Companies in California.” The
San Francisco Chronicle included
LTC in its “Top 100 in the Bay Area.”
The Chronicle ranked LTC fourth in
return on sales, 18th in return on
equity, 36th in market value and
40th in growth among the biggest
companies in the Bay Area. Linear
Technology also figured prominently
in the San Jose Mercury list of the
“Silicon Valley 150,” ranking third in
return on sales, 19th in return on
equity, 19th in market capitalization,
24th in profit and 73rd in sales.
The Wall Street Transcript, a magazine published for stock market
broker/analysts, published the results of a poll in which more than 40
analysts selected LTC President Bob
Swanson for a bronze medal among
all CEOs in the semiconductor
industry.
For further information on the
above or any of the other devices
mentioned in this issue of Linear
Technology, use the reader service
card or call the LTC literature service number: 1-800-4-LINEAR. Ask
for the pertinent data sheets and
application notes.
Burst ModeTM is a trademark of Linear
Technology Corporation. LTTM, LT , and LTC 
are trademarks used only to identify products of Linear Technology Corp. Other product
names may be trademarks of the companies
that manufacture the products.
Information furnished by Technology Corporation is believed to be accurate and reliable.
However, Linear Technology makes no representation that the circuits described herein
will not infringe on existing patent rights.
31
DESIGN TOOLS
World Headquarters
Applications on Disk
Linear Technology Corporation
1630 McCarthy Boulevard
Milpitas, CA 95035-7487
Phone: (408) 432-1900
FAX: (408) 434-0507
NOISE DISK
This IBM-PC (or compatible) progam allows the user to
calculate circuit noise using LTC op amps, determine the
best LTC op amp for a low noise application, display the
noise data for LTC op amps, calculate resistor noise, and
calculate noise using specs for any op amp.
Available at no charge.
SPICE MACROMODEL DISK
This IBM-PC (or compatible) high density diskette contains
the library of LTC op amp SPICE macromodels. The
models can be used with any version of SPICE for general
analog circuit simulations. The diskette also contains working circuit examples using the models, and a demonstration
copy of PSPICETM by MicroSim.
Available at no charge.
Technical Books
1990 Linear Databook — This 1440 page collection
of data sheets covers op amps, voltage regulators,
references, comparators, filters, PWMs, data conversion
and interface products (bipolar and CMOS), in both commercial and military grades. The catalog features well over
300 devices.
$10.00
1992 Linear Databook Supplement — This 1248 page
supplement to the 1990 Linear Databook is a collection of
all products introduced since then. The catalog contains full
data sheets for over 140 devices. The 1992 Linear Databook
Supplement is a companion to the 1990 Linear Databook ,
which should not be discarded.
$10.00
Linear Applications Handbook — 928 pages full of
application ideas covered in depth by 40 Application Notes
and 33 Design Notes. This catalog covers a broad range of
“real world” linear circuitry. In addition to detailed, systemsoriented circuits, this handbook contains broad tutorial
content together with liberal use of schematics and scope
photography. A special feature in this edition includes a 22
page section on SPICE macromodels.
$20.00
1993 Linear Applications Handbook Volume II —
Continues the stream of “real world” linear circuitry initiated
by the 1990 Handbook. Similar in scope to the 1990 edition,
the new book covers Application Notes 41 through 54 and
Design Notes 33 through 69. Additionally, references and
articles from non-LTC publications that we have found
useful are also included.
$20.00
Interface Product Handbook — This 200 page handbook
features LTC’s complete line of line driver and receiver
products for RS232, RS485, RS423, RS422 and AppleTalk 
applications. Linear’s particular expertise in this area involves low power consumption, high numbers of drivers
and receivers in one package, 10kV ESD protection of
RS232 devices and surface mount packages.
Available at no charge.
Monolithic Filter Handbook — This 234 page book comes
with a disk which runs on PCs. Together, the book and disk
assist in the selection, design and implementation of the
right switched capacitor filter circuit. The disk contains
standard filter responses as well as a custom mode. The
handbook contains over 20 data sheets, Design Notes and
Application Notes.
$40.00
SwitcherCAD Handbook — This 144 page manual, including disk, guides the user through SwitcherCAD—a
powerful PC software tool which aids in the design and
optimization of switching regulators. The program can cut
days off the design cycle by selecting topologies, calculating operating points and specifying component values and
manufacturer's part numbers.
$20.00
International
Sales Offices
FRANCE
Linear Technology S.A.R.L.
Immeuble “Le Quartz”
58 Chemin de la Justice
92290 Chatenay Malabry
France
Phone: 33-1-41079555
FAX: 33-1-46314613
U.S. Area
Sales Offices
GERMANY
Linear Techonolgy GmbH
Untere Hauptstr. 9
D-85386 Eching
Germany
Phone: 49-89-3197410
FAX: 49-89-3194821
CENTRAL REGION
Linear Technology Corporation
Chesapeake Square
229 Mitchell Court, Suite A-25
Addison, IL 60101
Phone: (708) 620-6910
FAX: (708) 620-6977
JAPAN
Linear Technology KK
5F YZ Bldg.
4-4-12 Iidabashi, Chiyoda-Ku
Tokyo, 102 Japan
Phone: 81-3-3237-7891
FAX: 81-3-3237-8010
NORTHEAST REGION
Linear Technology Corporation
One Oxford Valley
2300 E. Lincoln Hwy.,Suite 306
Langhorne, PA 19047
Phone: (215) 757-8578
FAX: (215) 757-5631
KOREA
Linear Technology Korea Branch
Namsong Building, #505
Itaewon-Dong 260-199
Yongsan-Ku, Seoul
Korea
Phone: 82-2-792-1617
FAX: 82-2-792-1619
Linear Technology Corporation
266 Lowell St., Suite B-8
Wilmington, MA 01887
Phone: (508) 658-3881
FAX: (508) 658-2701
SINGAPORE
Linear Technology Pte. Ltd.
101 Boon Keng Road
#02-15 Kallang Ind. Estates
Singapore 1233
Phone: 65-293-5322
FAX: 65-292-0398
NORTHWEST REGION
Linear Technology Corporation
782 Sycamore Dr.
Milpitas, CA 95035
Phone: (408) 428-2050
FAX: (408) 432-6331
SOUTHEAST REGION
Linear Technology Corporation
17060 Dallas Parkway
Suite 208
Dallas, TX 75248
Phone: (214) 733-3071
FAX: (214) 380-5138
TAIWAN
Linear Technology Corporation
Rm. 801, No. 46, Sec. 2
Chung Shan N. Rd.
Taipei, Taiwan, R.O.C.
Phone: 886-2-521-7575
FAX: 886-2-562-2285
SOUTHWEST REGION
Linear Technology Corporation
22141 Ventura Blvd.
Suite 206
Woodland Hills, CA 91364
Phone: (818) 703-0835
FAX: (818) 703-0517
UNITED KINGDOM
Linear Technology (UK) Ltd.
The Coliseum, Riverside Way
Camberley, Surrey GU15 3YL
United Kingdom
Phone: 44-276-677676
FAX: 44-276-64851
LINEAR TECHNOLOGY CORPORATION
1630 McCarthy Boulevard
Milpitas, CA 95035-7487
(408) 432-1900
Literature Department 1-800-4-LINEAR
AppleTalk is a registered trademark of Apple Computer, Inc.
©32
1994 Linear Technology Corporation/ Printed in U.S.A./27K
Linear Technology Magazine • June 1994