V17N4 - JANUARY

LINEAR TECHNOLOGY
January 2008
IN THIS ISSUE…
COVER ARTICLE
Easy High Density Power: 48A Surface
Mount DC/DC Power Supply Uses Four
Parallel 12A µModule Regulators ........1
alan Chern
Linear in the News… ...........................2
DESIGN FEATURES
Internal 2A, 42V Switch, Adjustable
2.5MHz Operating Frequency and
3mm × 3mm Package Allow Boost
Regulator to Fit Numerous Applications
.....................................................................7
Mathew Wich
36V, 3.5A DC/DC Buck Regulators
for Automotive, Industrial and Wall
Adapter Applications Offer High
Efficiency in a Small Package .........11
Kevin Huang
Monolithic 2A Buck Regulator
Plus Linear Regulator Simplifies
Wide Input Voltage Applications .......13
rich Philpott
Efficient 48V Buck Mode
LED Driver Delivers 50mA .................16
Mohammad J. navabi
Synchronous Boost Converters Provide
High Voltage without the Heat ..........19
Greg Dittmer
Wide Input Voltage Range, Dual
Step-Down Controller Reduces
Power Supply Size and Cost .............22
Wei Gu
Surge Stopper Protects
Sensitive Electronics from
High Voltage Transients ........................... 24
James Herr
DESIGN IDEAS
....................................................27–38
(complete list on page 27)
New Device Cameos ...........................39
Design Tools ......................................43
Sales Offices .....................................44
VOLuME XVII nuMBEr 4
Easy High Density Power:
48A Surface Mount DC/DC
Power Supply Uses Four Parallel
12A µModule Regulators
Introduction
Linear Technology’s µModule DC/DC
regulators simplify power supply
design by offering the black box convenience of traditional power modules
in an IC form factor. For example,
the LTM4601 µModule regulator is
a complete step-down power module
in a 15mm × 15mm × 2.8mm LGa
package.
The LTM4601 accepts 4.5V to
20V inputs and can produce outputs
anywhere from 0.6V to 5V at 12a.
The wide input and output ranges
and excellent thermal performance
of the LTM4601 allow it to be easily
dropped into a variety of applications
with minimal design effort—just set the
output voltage with a single resistor
and determine the requisite bulk input
and output capacitances.
another signiicant advantage of
the LTM4601 over power-moduleor IC-based systems is its ability to
easily scale up as loads increase. If
load requirements are greater than
one µModule regulator can produce,
simply add more modules in parallel.
The design of a parallel system involves
little more than copying and pasting
the layout of each 15mm × 15mm
µModule regulator. Electrical layout
issues are taken care of within the
µModule package—there are no external inductors, switches or other
components to worry about. Even heat
by Alan Chern
The LTM4601 µModule
DC/DC regulator is a high
performance power module
shrunk down to an IC
form factor. The usual
external components are
integrated into the LGA
package—including the
PWM controller, inductor,
input and output capacitors,
ultralow RDS(ON) FETs,
Schottky diodes and
compensation circuitry. Only
external bulk input and
output capacitors and one
resistor are needed to set
the output from 0.6V to 5V.
distribution is improved with parallel regulators, thus enabling surface
mount solutions for high power density
applications.
To demonstrate the simplicity and
performance of a paralleled µModule
regulator design, this article discusses
electrical guidelines, layout considerations, and thermal speciics for
designing a compact 48a, 0.6V–5V
VOuT, 4.5V–20V VIn converter using four LTM4601 µModule DC/DC
regulators.
continued on page L, LT, LTC, LTM, Burst Mode, OPTI-LOOP, Over-The-Top and PolyPhase are registered trademarks of Linear Technology
Corporation. adaptive Power, Bat-Track, BodeCaD, C-Load, DirectSense, Easy Drive, FilterCaD, Hot Swap, LinearView,
µModule, Micropower SwitcherCaD, Multimode Dimming, no Latency ΔΣ, no Latency Delta-Sigma, no rSEnSE, Operational
Filter, PanelProtect, PowerPath, PowerSOT, SmartStart, SoftSpan, Stage Shedding, SwitcherCaD, ThinSOT, True Color PWM,
ultraFast and VLDO are trademarks of Linear Technology Corporation. Other product names may be trademarks of the
companies that manufacture the products.
L LINEAR IN THE NEWS
Linear in the News…
Big Power in Small Packages
Linear has just completed the launch of a new family of
high voltage µModule™ DC/DC converters. These small,
low proile devices are instant power supplies, packing a
range of power system solutions into surface mount packages that can be automatically placed on either side of a PC
board. With the introduction of the LTM802X high voltage
µModule regulators, Linear has expanded its offering to
solutions ideal for 24V industrial, 28V medical, automotive
and avionics applications. (For more, see page 36.)
robert Dobkin, CTO of Linear Technology, stated,
“Manufactured in a bipolar transistor process, the LT3080
expands the easy-to-use linear regulator into modern high
performance systems. With its low voltage operation and
the ability to parallel devices for higher output, it can do
circuit tricks that no other regulator can. This is a new
general purpose and more useful architecture for regulators that will proliferate with time.”
The LT3080 is a 1.1a 3-terminal linear regulator that can
easily be paralleled for heat spreading and is adjustable to
zero with a single resistor. This new architecture regulator uses a current reference and voltage follower to allow
sharing between multiple regulators with a small length
of PC trace as ballast, enabling multiamp linear regulation
in all surface-mount systems without heat sinks.
The LT3080 achieves high performance with wide input
voltage capability from 1.2V to 40V, a dropout voltage of
only 300mV and millivolt regulation. The output voltage
is adjustable, spanning a wide range from 0V to 40V, and
the on-chip trimmed reference achieves high accuracy of
±1%. The LT3080 really shines in generating multirail
systems.
Linear Highlights µModule Regulators
in FPGA Net Seminar
Power Electronics Technology Names
LT3080 Product of the Year
Power Electronics Technology magazine selected Linear
Technology’s LT3080 3-terminal low dropout linear regulator as Product of the year. The award was presented at
the Power Electronics Technology Conference in Dallas
to Linear Technology Vice President Engineering and
Chief Technical Oficer robert Dobkin, who developed the
product. as a historical note, the LT3080 is a signiicant
reinement over the industry-standard 3-terminal linear
regulators irst developed by robert Dobkin over 30 years
ago.
David Morrison, Editor of Power Electronics Technology, stated, “among the hundreds of power components
introduced each year, there are numerous devices with
exciting performance improvements and novel features.
This continuing wave of innovation makes selecting a single
product for special recognition a particularly daunting
challenge. Linear Technology’s LT3080 was selected as this
year’s Product of the year because it offers an intriguing
combination of novelty and usefulness. By redesigning the
low-dropout linear regulator, Linear has given engineers
an extremely lexible building block that should help solve
current and future board-level power challenges.”
2
Linear Technology power module Development Manager
Eddie Beville recently co-presented a web seminar entitled,
“Xilinx Virtex-5 Power Optimization and Power Design
Guidelines.” The online seminar is designed to teach designers how to leverage
the dedicated blocks
in Virtex-5, using the
Xilinx Power Estimator (XPE) to reduce
power consumption,
increase system reliability and simplify
thermal management
and power supply
design for FPGabased systems. It also
demonstrates how
to implement Linear
Technology power
management solutions via real world
design examples for Virtex-5 FPGas. The seminar showed
how to design the power distribution network using Linear
Technology’s µModule DC/DC converters, ultralow noise
VLDOs and other devices for key system functions.
The seminar was conducted on EE Times’ TechOnline
engineering education website. It is currently available for
viewing at www.techonline.com/learning/webinar/. L
Linear Technology Magazine • January 2008
DESIGN FEATURES L
LTM4601, continued from page 1
VOUT
CLOCK SYNC
0° PHASE
VIN
4.5V TO 20V
51.1k
5.9k
LTC6902
SET
V+
MOD
DIV
GND
PH
OUT1 OUT4
OUT2 OUT3
0.1µF
+
51.1k
MPGM
RUN
COMP
INTVCC
DRVCC
CIN*
100µF
25V
10µF
25V
×2
VIN
PGOOD
392k
SGND
5%
MARGIN
4-PHASE
OSCILLATOR
PLLIN TRACK/SS
VOUT
LTM4601
PGND
TRACK/SS CONTROL
VOUT
1.5V
48A
120pF
VFB
MARG0
MARG1
22µF
6.3V
470µF
6.3V
VOUT_LCL
DIFFVOUT
VOSNS+
VOSNS–
fSET
60.4k
+ RSET
N
RSET
N = NUMBER OF PHASES
VOUT = 0.6V
RSET
10k
+
120pF
MARGIN CONTROL
CLOCK SYNC
90° PHASE
TRACK/SS CONTROL
4.5V TO 20V
VIN
PGOOD
PGOOD
MPGM
RUN
COMP
INTVCC
DRVCC
10µF
25V
×2
PLLIN TRACK/SS
VOUT
LTM4601-1
392k
SGND
PGND
22µF
6.3V
VFB
MARG0
MARG1
470µF
6.3V
VOUT_LCL
NC3
NC2
NC1
+
fSET
CLOCK SYNC
180° PHASE
TRACK/SS CONTROL
4.5V TO 20V
VIN
PGOOD
PGOOD
MPGM
RUN
COMP
INTVCC
DRVCC
10µF
25V
×2
PLLIN TRACK/SS
VOUT
LTM4601-1
392k
SGND
PGND
22µF
6.3V
VFB
MARG0
MARG1
470µF
6.3V
VOUT_LCL
NC3
NC2
NC1
+
fSET
CLOCK SYNC
270° PHASE
TRACK/SS CONTROL
4.5V TO 20V
VIN
PGOOD
PGOOD
MPGM
RUN
COMP
INTVCC
DRVCC
10µF
25V
×2
LTM4601-1
392k
SGND
0.1µF
PLLIN TRACK/SS
VOUT
PGND
VFB
MARG0
MARG1
VOUT_LCL
NC3
NC2
NC1
fSET
22µF
6.3V
470µF
6.3V
+
*CIN OPTIONAL TO REDUCE ANY LC RINGING.
NOT NEEDED FOR LOW INDUCTANCE PLANE CONNECTION
Figure 1. Designing a high density power supply for a limited space application could not be easier. Here, four LTM4601 µModule
regulators are paralleled in a simple scheme. Board layout is just as easy, since there are so few external components.
Linear Technology Magazine • January 2008
3
L DESIGN FEATURES
DC/DC µModule Regulator:
A Complete System in an
LGA Package
48A from Four
Parallel µModule Regulators
Figure 1 shows a regulator comprising four parallel LTM4601s, which
can produce a 48a (4 ×12a) output.
The regulators are synchronized but
operate 90° out of phase with respect
to each other, thereby reducing the
amplitude of input and output ripple
currents through cancellation. The attenuated ripple in turn decreases the
external capacitor rMS current rating
and size requirements, further reducing solution cost and board space.
Synchronization and phase shifting is implemented via the LTC6902
oscillator, which provides four clock
outputs, each 90° phase shifted (for 2or 3-phase relationships, the LTC6902
can be adjusted via a resistor.). The
clock signals serve as input to the
PLLIn (phase lock loop in) pins of the
four LTM4601s. The phase-lock loop
of the LTM4601 comprises a phase
detector and a voltage controlled os4
90
80
EFFICIENCY (%)
The LTM4601 µModule DC/DC
regulator is a high performance power
module shrunk down to an IC form
factor. It is a completely integrated
solution—including the PWM controller, inductor, input and output
capacitors, ultralow rDS(On) FETs,
Schottky diodes and compensation
circuitry. Only external bulk input
and output capacitors and one resistor are needed to set the output from
0.6V to 5V. The supply can produce
12a (more if paralleled) from a wide
input range of 4.5V to 20V, making it
extremely versatile. The pin compatible LTM4601HV extends the input
range to 28V.
Output features include output
voltage tracking and margining. The
high switching frequency, typically
850kHz at full load, constant on time,
zero latency controller delivers fast
transient response to line and load
changes while maintaining stability.
Should frequency harmonics be a
concern, an external clock can control
synchronization via an on chip phase
lock loop.
100
70
60
50
40
30
3.3VOUT
2.5VOUT
1.8VOUT
1.5VOUT
1.2VOUT
20
10
0
0
1µs/DIV
Figure 2. Individual LTM4601 switching
waveforms for the circuit in Figure 1 shows
the 90° out-of-phase relationship.
10
20
30
LOAD CURRENT (A)
40
50
Figure 3. Efficiency of the four
parallel LTM4601s remains high
over a wide range of outputs
VIN
CIN
CIN
GND
SIGNAL
GND
COUT
COUT
VOUT
Figure 4. The LTM4601’s pin layout promotes simple power
plane placement and uncomplicated part paralleling
Figure 5. Top layer planes for 4-parallel µModule system
Figure 6. Bottom layer planes for 4-parallel µModule system
Linear Technology Magazine • January 2008
DESIGN FEATURES L
Figure 7. Thermograph of four
parallel LTM4601s without airflow
(20V input to 1.5V output at 40A)
cillator, which combine to lock onto
the rising edge of an external clock
with a frequency range of 850kHz
±30%. The phase lock loop is turned
on when a pulse of at least 400ns
and 2V amplitude at the PLLIn pin is
detected, though it is disabled during
start-up. Figure 2 shows the switching
waveforms of four LTM4601 µModule
regulators in parallel.
Only one resistor is required to set
the output voltage in a parallel setup,
but the value of the resistor depends
on the number of LTM4601s used.
This is because the effective value of
the top (internal) feedback resistor
changes as you parallel LTM4601s.
The LTM4601’s reference voltage is
0.6V and its internal top feedback
resistor value is 60.4kΩ, so the relationship between VOuT, the output
voltage setting resistor (rFB) and the
number of modules (n) placed in
parallel is:
VOUT
60.4k
+ RFB
= 0.6 V n
RFB
Figure 3 illustrates the system’s
high eficiency over the vast output
current range up to 48a. The system
performs impressively with no dipping
in the eficiency curve for a broad range
of output voltages.
Layout
Layout of the parallel µModule regulators is relatively simple, in that there
are few electrical design considerations. nevertheless, if the intent of
a design is to minimize the required
PCB area, thermal considerations
Linear Technology Magazine • January 2008
become paramount, so the important
parameters are spacing, vias, airlow
and planes.
The LTM4601 µModule regulator
has a unique LGa package footprint,
which allows solid attachment to the
PCB while enhancing thermal heat
sinking. The footprint itself simpliies
layout of the power and ground planes,
as shown in Figure 4. Laying out four
parallel µModule regulators is just as
easy, as shown in Figures 5 and 6.
VOUT
GND
VIN
Figure 8. Via placement (cross marks)
under a single µModule regulator
AIRFLOW
DIRECTION
Figure 9. Thermograph of four parallel LTM4601s with 200LFM
bottom-to-top airflow (20V input to 1.5V output at 40A)
AIRFLOW
DIRECTION
Figure 10. Thermograph of four parallel LTM4601s with 400LFM right-to-left
airflow in 50°C ambient chamber (12V input to 1V output at 40A)
AIRFLOW
DIRECTION
Figure 11. Thermograph of four parallel LTM4601s with BGA heat sinks and 400LFM
right-to-left airflow in a 75°C ambient chamber (12V input to 1V output at 40A)
5
L DESIGN FEATURES
If laid out properly, the LGa packaging and the power planes alone can
provide enough heat sinking to keep
the LTM4601 cool.
Figure 7 is a thermal image of the
DC1043a board with readings of the
temperatures at speciic locations.
Cursors 1 to 4 give a rough estimation
of the surface temperature on each
module. Cursors 5 to 7 indicate the
surface temperature of the PCB. notice
the difference in temperature between
the inner two regulators, cursors 1
and 2, and the outside ones, cursors 3
and 4. The LTM4601 µModule regulators placed on the outside have large
planes to the left and right promoting
heat sinking to cool the part down a
few degrees. The inner two only have
small top and bottom planes to draw
heat away, thus becoming slightly
warmer than the outside two.
Further heat dissipation is possible by adding vias underneath the
part. Vias provide a path to the power
planes and into the PCB, which helps
draw heat away. Vias should not be
placed directly under the pads. Figure 8 shows the layout of the vias on
the DC1043a demonstration circuit.
The cross marks indicate the vias in
between the LGa pads.
airlow also has a substantial effect
on the thermal balance of the system.
note the difference in temperature
between Figure 7 and Figure 9. In
Figure 9, a 200LFM airlow travels
evenly from the bottom to the top of
the demo board, causing a 20°C drop
across the board compared to the no
air low case in Figure 7.
The direction of airlow is also
important. In Figure 10 the airlow
travels from right to left, pushing the
heat from one µModule regulator to
the next, creating a stacking effect.
The µModule device on the right, the
closest to the airlow source, is the
coolest. The leftmost µModule regulator has a slightly higher temperature
because of spillover heat from the other
LTM4601 µModule regulators.
Heat transfer to the PCB also
changes with airlow. In Figure 7,
heat transfers evenly to both left and
right sides of the PCB. In Figure 10,
most of the heat moves to the left side.
6
tion of start-up time to VOuT and the
soft-start capacitor (CSS) is:
Layout of the parallel
µModule regulators is
relatively simple, in that
there are few electrical
design considerations.
Nevertheless, if the intent of
a design is to minimize the
required PCB area, thermal
considerations become
paramount. The important
layout parameters are
regulator spacing and usage
of vias, airflow and planes.
VOUT(MARGIN) =
%VOUT
• VOUT
100
t SOFTSTART =
(
)
0.8 • 0.6 V − VOUT(MARGIN) •
Figure 11 shows an extreme case of
heat stacking from one µModule device
to the next. Each of the four µModule
regulators is itted with a BGa heat
sink and entire board is operated in a
chamber with an ambient temperature
of 75°C.
Start-Up, Soft-Start and
Current Sharing
The soft-start feature of the LTM4601
prevents large inrush currents at
start-up by slowly ramping the output
voltage to its nominal value. The rela-
C SS
1.5µA
For example, a 0.1µF soft-start
capacitor yields a nominal 8ms ramp
(see Figure 12) with no margining.
Current sharing among parallel
regulators is well balanced through
start-up to full load. Figure 13 shows
an evenly distributed output current
curve for a 2-parallel LTM4601 system,
as each rises to a nominal 10a each,
20a total.
Conclusion
The LTM4601 µModule regulator is a
self-contained 12a step-down regulator in an IC form factor. It can be
easily paralleled to increase load capability to 48a as shown here. Thermal
performance is equally impressive at
48a of output current with balanced
current sharing and smooth uniform
start-up. The ease and simplicity of
this design minimizes development
time while saving board space. L
12V
VIN
5V/DIV
0V
VOUT
1V/DIV
ILOAD
20A/DIV
VIN = 12V
VOUT = 1.5V
LOAD = 40A
2ms/DIV
Figure 12. Soft-start ramp for four parallel LTM4601s
VIN
5V/DIV
IOUT(IC#1)
5A/DIV
IOUT(IC#2)
5A/DIV
5ms/DIV
Figure 13. Current sharing among parallel regulators is well balanced through start-up
to full load. Two parallel LTM4601s, as each rises to a nominal 10A each, 20A total.
Linear Technology Magazine • January 2008
DESIGN FEATURES L
Internal 2A, 42V Switch, Adjustable
2.5MHz Operating Frequency and
3mm × 3mm Package Allow Boost
Regulator to Fit Numerous Applications
by Mathew Wich
Introduction
The world of switching DC/DC converters is awash with a dizzying array of
product offerings. For a given application, much of the power supply design
effort can be spent simply searching for
the optimum combination of package
size, switching frequency, input and
output voltage range, and desirable
features. In many cases, though, the
LT3580 offers an optimal solution. It
is the right choice for many diverse
applications because of its smart
combination of features, performance
and ease of use.
The LT3580 is a current control
switching regulator available in
The LT3580 supports
a variety of converter
configurations including
boost, inverting, flyback,
and SEPIC. Inputs can be
from 2.5V–32V, and an
integrated 2A, 42V NPN
power switch allows the
LT3580 to provide
efficient power from a
fraction of a watt up to more
than several watts.
L1
4.2µH
VIN
5V
VIN
C2
10µF
SW
SHDN
RT
D1
VOUT
12V
550mA
GND
130k
LT3580
FB
VC
SYNC
C1
2.2µF
75k
SS
10k
0.1µF
1nF
C1: 2.2µF, 25V, X5R, 1206
C2: 10µF, 25V, X5R, 1206
D1: MICROSEMI UPS120
L1: SUMIDA CDR6D23MN-4R2
95
1200
90
1000
80
800
75
600
70
65
400
POWER LOSS (mW)
EFFICIENCY (%)
85
60
200
55
50
0
0
100
200
300
400
LOAD CURRENT (mA)
500
600
Figure 1. This 1.2MHz, 5V to 12V boost converter achieves over 88% efficiency.
Linear Technology Magazine • January 2008
tiny 8-lead packages (MSOP and
3mm × 3mm DFn). Operating from
200kHz–2.5MHz, it supports numerous conigurations including boost,
inverting, lyback and SEPIC. Inputs
can be from 2.5V–32V, and an integrated 2a, 42V nPn power switch
allows the LT3580 to provide eficient
power from a fraction of a watt up to
more than several watts.
Be Picky—Choose the
Ideal Clock Frequency
up to 2.5MHz
Choosing a converter switching frequency is often a compromise between
several performance parameters such
as physical size, output ripple, eficiency and spectral noise issues. While
most converter ICs operate at a single
ixed frequency, the LT3580 operates
at any frequency from 200kHz–2.5MHz
allowing you to choose the ideal frequency for any application.
The high frequency capability (up
to 2.5MHz) of the LT3580 helps to
reduce the overall size of the converter
by permitting the use of smaller inductors and output capacitors. Small
inductors, with correspondingly small
inductances, work best at higher
frequencies because they store and
release less energy in each switching
cycle. This can be seen by looking at
the energy storage relationship for an
inductor,
E=
1 2
LI ,
2
which shows that for a given peak
inductor current (I), the stored energy
is proportional to the inductance
(L). Thus smaller inductances, storing less energy per cycle, switch at
7
L DESIGN FEATURES
RC
VIN
CSS
7
SHDN
–
+
5
1.3V
CC
CIN
2
SS
VC
DISCHARGE
DETECT
L1
275k
UVLO
SR2
R
4
ILIMIT
COMPARATOR
–
Q2
Q
VIN
A3
1.215V
REFERENCE
+
R
S
C1
Q1
Q
RFB
+
+
14.6k
Σ
A1
A4
0.01Ω
–
–
FB
VOUT
SR1
DRIVER
S
3
D1
SW
VC
SOFTSTART
RAMP
GENERATOR
1
+
14.6k
FREQUENCY
FOLDBACK
A2
GND
9
÷N ADJUSTABLE
OSCILLATOR
–
SYNC
BLOCK
SYNC
8
LT3580
RT
6
RT
Figure 2. Block diagram of the LT3580 in a boost converter configuration
V=L
di
ΔI
⇒L
dt
ΔT
and solving for ΔT.
ΔT =
L • ΔI
V
This shows that, for a given inductor voltage (V), a smaller inductor (L)
L1
3.3µH
VIN
5V
D1
VIN
C2
4.7µF
SW
GND
130k
LT3580
FB
VC
SYNC
C1
4.7µF
SS
35.7k
10k
0.1µF
2.2nF
47pF
VOUT
12V
500mA
and uses a smaller inductor and less
output capacitance than the 1.2MHz
solution in Figure 1. The tradeoff is
slightly reduced eficiency due to the
increased switching losses incurred at
the higher switching frequency.
For large voltage gains, the LT3580’s
low frequency capability (down to
200kHz) is very useful. Figure 5 shows
a direct conversion from 5V to 40V
running at 750kHz. Figure 6 shows a
5V to 350V lyback converter running
at 200kHz.
Finally, the LT3580’s wide frequency range makes it easy to avoid
95
1400
90
1200
85
1000
80
75
800
70
600
65
400
60
200
55
C1, C2: 4.7µF, 25V, X5R, 1206
D1: MICROSEMI UPS120
L1: COILCRAFT LPS4018-332ML
POWER LOSS (W)
SHDN
RT
will ramp to its peak current (I) in less
time (T) than a larger inductance, again
leading to higher frequency operation
to make best use of the inductor.
Depending on the load requirements, high frequency operation also
facilitates smaller output capacitors.
Since charge is delivered to the output
in smaller but more frequent packets,
the voltage ripple is reduced for a given
capacitance.
Figure 3 shows an example of
reduced solution size at a higher
switching frequency. The 5V to 12V
boost converter operates at 2.5MHz
EFFICIENCY (%)
higher frequencies to deliver the same
power as larger inductances. also,
smaller inductances reach their peak
current (or energy) faster than large
inductances as seen by rearranging
the relationship
50
0
100
300
200
400
LOAD CURRENT (mA)
500
0
600
Figure 3. The high 2.5MHz switching frequency of this 5V to 12V boost converter allows the use of a tiny 4mm × 4mm × 1.7mm inductor.
8
Linear Technology Magazine • January 2008
DESIGN FEATURES L
RT = 35.7k
2.5
2.3
2.1
1.9
1.7
1.5
Accurate Clocking Options
1.3
The LT3580 provides two options for
generating the clock. First, the integrated oscillator can be accurately set
between 200kHz–2.5MHz by connecting a single resistor from the r T pin
to ground, where
1.1
–50
91.9
R T (kΩ) =
−1
fOSC (MHz)
The boost converter in Figure 3, for
example, uses a 35.7k r T resistor to
set the switching frequency to 2.5MHz.
The internal oscillator’s frequency is
accurate to ±10% with little temperature variation as shown in Figure 4.
The excellent frequency tolerance
maximizes system performance by
reducing necessary design margin.
The switching frequency can also
be synchronized to an external clock
source. The SynC pin overrides the
internal oscillator when toggled at
frequencies greater than 75% of the
internal oscillator’s set frequency.
Simply connect a digital clock signal
to the SynC pin using VIH levels from
1.3V to 5.5V, VIL levels below 0.4V
and any frequency between 200kHz
and 2.5MHz. using an external clock
source is often helpful for several
reasons, including…
q Synchronization of several
switching regulators, often out
of phase, to reduce switching
current spikes
q additional frequency precision
yielding higher performance
q Precisely targeting the frequency
out of sensitive bands for EMI
beneits.
The LTC6908 resistor set oscillator is a nice choice for generating the
SynC clock due to its high precision,
dual phase outputs, spread spectrum
capabilities, small size and simple
operation.
Linear Technology Magazine • January 2008
connecting a single external resistor
from VOuT to the FB pin. The FB pin
automatically servos to the correct
reference voltage for a given topology
(1.215V for positive VOuT and 5mV for
negative VOuT).
Supported conigurations include
boost, SEPIC (Figure 10), and other
topologies such as the lyback (Figure 6) and inverting (Figure 7).
Finally, to improve VOuT accuracy,
the FB pin is factory trimmed to an
accurate current, instead of trimming
the resistance, which is typical of other
parts. This eliminates multiplication of
reference voltage errors to VOuT.
2.7
FREQUENCY (MHz)
sensitive frequency bands that can’t
tolerate spectral noise. For example
radio power supplies may operate
at 2MHz or above to avoid the aM
broadcast band. also, some rF communications products are sensitive to
noise at 455kHz, therefore switching
above 600kHz is desired.
RT = 75k
50
0
TEMPERATURE (°C)
100
Figure 4. Typical internal oscillator
frequency at VIN = 5V
Single-Pin Feedback and
Support for Multiple
Configurations
Soft-Start Feature Limits
Start-Up Current
The novel single-pin feedback of the
LT3580 reduces external component
count and allows it to be used in
many different converter topologies.
The output voltage is set by simply
The LT3580 contains a soft-start
circuit to limit peak switch currents
during start-up. High start-up current
is inherent in switching regulators
since the feedback loop is saturated
L1
47µH
VIN
5V
D1
VIN
C2
2.2µF
SW
SHDN
GND
464k
LT3580
RT
VOUT
40V
150mA
FB
VC
SYNC
C1
2.2µF
SS
10k
121k
0.1µF
47pF
4.7nF
C1, C2: 2.2µF, 25V, X5R, 1206
D1: MICROSEMI UPS140
L1: SUMIDA CDRH105R-470
Figure 5. A 750kHz, 5V to 40V, 150mA boost converter
Danger High Voltage! Operation by High Voltage Trained Personnel Only
7, 8
•
VOUT
350V
4.5mA (VIN = 5V)
2.5mA (VIN = 3.3V)
D1
T1
1:10.4
VIN
3.3V TO 5V
1
4.7µH
•
5, 6
VIN
SW
SHDN
RT
C2
68nF
4
FOR ANY VOUT BETWEEN 50V TO
350V, CHOOSE RFB ACCORDING TO
GND
RFB 4.22M*
LT3580
V
– 1.215
RFB = OUT
83.3µA
FB
VC
SYNC
C1
2.2µF
464k
SS
10k
0.47µF
100pF
10nF
C1: 2.2µF, 25V, X5R, 1206
C2: TDK C3225X7R2J683M
D1: VISHAY GSD2004S DUAL DIODE CONNECTED IN SERIES
T1: TDK LDT565630T-041
FOR 5V INPUT, KEEP MAXIMUM
OUTPUT POWER AT 1.58W
FOR 3.3V INPUT, KEEP MAXIMUM
OUTPUT POWER AT 0.88W
*MAY REQUIRE MULTIPLE SERIES
RESISTORS TO COMPLY WITH
MAXIMUM VOLTAGE RATINGS
Figure 6. This 350V power supply features a tiny
5.8mm × 5.8mm × 3mm transformer switching at 200kHz.
9
L DESIGN FEATURES
due to VOuT being far from its inal
value. The regulator tries to charge
the output capacitors as quickly as
possible, which results in large peak
currents.
The start-up current can be limited
by connecting an external capacitor
(typically 100nF to 1µF) to the SS
pin. This capacitor is slowly charged
to ~2.2V by an internal 275k resistor
once the part is activated. SS voltages
below ~1.1V reduce the internal current limit. Thus, the gradual ramping
of SS also gradually increases the
current limit as the capacitor charges.
This, in turn, allows the VOuT capacitor
to charge gradually toward its inal
value while limiting the start-up current (see Figure 9).
VIN
3.3V TO 12V
VIN
SHDN
10
VOUT
–5V
800mA (VIN = 12V)
C2 620mA (VIN = 5V)
10µF 450mA (VIN = 3.3V)
D1
GND
60.2k
LT3580
RT
FB
VC
SYNC
C1
2.2µF
SS
10k 100pF
35.7k
0.1µF
2.2nF
C1: 2.2µF, 25V, X5R, 1206
C2: 10µF, 25V, X5R, 1206
C3: 1µF, 50V, X5R, 0805
D1: CENTRAL SEMI CMMSH1-40
L1, L2: COILCRAFT LSP4018-472ML
Figure 7. This –5V output inverting converter switches
at 2.5MHz and accepts inputs between 3.3V and 12V
VIN
VIN
Innovative SHDN Pin Resets
Soft-Start and Serves as
Undervoltage Lockout (UVLO)
continued on page 28
L2
4.7µH
SW
ACTIVE/
LOCKOUT
–
1.3V
RUVLO1
The SHDn pin has threshold hysteresis to resist noise and tolerate slowly
varying input voltages. Driving the
SHDn pin to ground shuts down the
LT3580 and reduces input current to
less than 1µa. Driving SHDn above
1.38V enables the part and begins the
soft-start sequence. a built in safety
feature ensures that the SS capacitor
is actively discharged before start-up
begins. This allows for proper soft-start
even in the event of short SHDn pulses
or thermal lockout.
The LT3580 also features an integrated uVLO that shuts down the
chip when the input voltage falls below
~2.3V. However, the SHDn pin can
also be conigured to disable the chip
below even higher voltages as shown
in Figure 8.
Typically, uVLO is needed in
situations where the input supply is
current-limited, has a relatively high
source resistance, or ramps up/down
slowly. a switching regulator draws
constant power from the source, so
source current increases as source
voltage drops. This looks like a negative resistance load to the source and
can cause the source to current-limit
or latch low under low voltage conditions. The conigurable uVLO prevents
the regulator from operating at source
C3
1µF
L1
4.7µH
SHDN
+
11.6µA
AT 1.3V
RUVLO2
(OPTIONAL)
GND
LT3580
Figure 8. Configurable undervoltage lockout
SHDN
2V/DIV
SS
0.5V/DIV
VOUT
5V/DIV
IL
500mA/DIV
VIN = 5V
VOUT = 12V
2ms/DIV
Figure 9. Soft-start of a 5V to 12V boost topology
C3
1µF
L1
4.7µH
VIN
2.6V TO 12V
OPERATING
12V TO 32V
TRANSIENT
VIN
RT
VOUT
5V, 600mA (VIN = 5V OR HIGHER)
500mA (VIN = 4V)
C2
400mA (VIN = 3V)
10µF
300mA (VIN = 2.6V)
L2
4.7µH
SW
SHDN
D1
GND
46.4k
LT3580
FB
VC
SYNC
C1
2.2µF
SS
35.7k
10k
0.1µF
22pF
1nF
C1: 2.2µF, 35V, X5R, 1206
C2: 10µF, 10V, X5R, 1206
C3: 1µF, 50V, X5R, 0805
D1: MICROSEMI UPS140
L1, L2: TDK VLCF4020T-4R7N1R2
Figure 10. Wide input range SEPIC converter with 5V output switches at 2.5MHz
Linear Technology Magazine • January 2008
DESIGN FEATURES L
36V, 3.5A DC/DC Buck Regulators
for Automotive, Industrial and Wall
Adapter Applications Offer High
Efficiency in a Small Package
by Kevin Huang
Introduction
automotive batteries, industrial power
supplies, distributed supplies and
wall transformers are all sources of
wide-ranging, high voltage inputs. The
easiest way to step down the voltage
from these sources is with a high voltage monolithic step-down switching
regulator that can directly accept a
wide input range and produce a wellregulated output. The LT3680 and
LT3693 are new step-down switching
regulators that accept inputs up to 36V
and provide excellent line and load
regulations and dynamic response.
Both regulators offer high eficiency
solutions over wide load range. The
LT3680 adds low ripple Burst Mode®
operation to maximize eficiency at
light load currents.
LT3680 and LT3693 Features
available in either a 10-pin MSOP
or a 3mm × 3mm DFn package, the
LT3680 and LT3693 offer an integrated 5a power switch and external
compensation for design lexibility.
Both regulators employ a constant
frequency, current mode architecture.
The switching frequency can be set be-
VOUT
5V
3.5A
VIN
6.3V TO 36V
VIN
OFF ON
VIN = 12V
VC
10µF
680pF
0.5
3
3.5
EFFICIENCY (%)
Figure 2. Efficiency vs load
current for circuit in Figure 1
Linear Technology Magazine • January 2008
SYNC
536k
GND
FB
47µF
Figure 1. This 600kHz 6.3V–36V input DC/DC converter delivers 3.5A at 5V output.
The easiest way to step
down the voltage from a
wide ranging, high voltage
source is with a monolithic
step-down switching
regulator that can directly
convert the input to a
well-regulated output.
VOUT
10mV/DIV
2
1.5
1
2.5
OUTPUT CURRENT (A)
D
D: ON SEMI MBRA340
L: NEC MPLC0730L4R7
70
0
SW
100k
IL
0.2A/DIV
50
L
4.7µH
PG
63.4k
VSW
5V/DIV
VOUT = 5V
L = 4.7µH
f = 600kHz
LT3680
RT
VIN = 24V
60
BOOST
0.47µF
VIN = 34V
80
RUN/SS
15k
100
90
BD
VIN = 12V
VOUT = 3.3V
ILOAD = 10mA
5µs/DIV
Figure 3. LT3680 Burst Mode
operation at 10mA load
tween 200kHz and 2.4MHz by using a
resistor tied from the rT pin to ground.
This allows a trade off between component size and eficiency. The switching
frequency can also be synchronized
to an external clock for noise sensitive applications. an external resistor
divider programs the output voltage
to any value above the part’s 0.79V
reference.
The LT3680 and LT3693 offer softstart via a resistor and capacitor on the
run/SS pin, thus reducing maximum
inrush currents during start-up. Both
regulators can withstand a shorted
output. a cycle-by-cycle internal
current limit protects the circuit in
overload and limits output power;
when the output voltage is pulled to
ground by a hard short, the LT3680
and LT3693 reduce the operating frequency to limit dissipation and peak
switch current. This lower frequency
allows the inductor current to safely
discharge, thus preventing current
runaway. The high side bootstrapping
boost diode is integrated into the IC to
minimize solution size and cost. When
11
L DESIGN FEATURES
VOUT
1.8V
3.5A
VIN
3.6V TO 27V
VIN
ON OFF
BD
RUN/SS
BOOST
L
3.3µH
0.47µF
VC
4.7µF
LT3693
SW
D
RT
16.9k
PG
78.7k
127k
SYNC
GND
FB
680pF
47µF
100k
f = 500kHz
D: ON SEMI MBRA340
L: NEC MPLC0730L3R3
Figure 4. This 500kHz 3.6V–27V input DC/DC converter delivers 3.5A at 1.8V output.
the output voltage is above 2.5V, the
anode of the boost diode can be connected to output. For output voltages
lower than 2.5V, the boost diode can be
tied to a separate rail or to the input.
For systems that rely on a well-regulated power source, the LT3680 and
LT3693 provide a power good lag that
signals when VOuT reaches 90% of the
programmed output voltage.
Low Ripple Burst Mode
Operation of LT3680
The only difference between LT3680
and LT3693 is that the LT3680 offers low ripple Burst Mode operation,
which can be selected by applying a
logic low to the SynC pin. Low ripple
Burst Mode operation maintains high
eficiency at light load while keeping
the output voltage ripple low. During
Burst Mode operation, the LT3680
delivers single cycle bursts of current
to the output capacitor followed by
sleep periods when the output power is
delivered to the load only by the output
capacitor. Between bursts, all circuitry
associated with controlling the output
switch is shut down, reducing the input supply current and BD quiescent
current to 30µa and 80µa, respectively. as the load current decreases
to a no load condition, the percentage
of time that LT3680 operates in sleep
mode increases and the average input
current is greatly reduced, resulting
in high eficiency. Both LT3680 and
LT3693 have a very low (less than 1µa)
shutdown current which signiicantly
extends battery life in applications that
spend long periods of shutdown mode.
For applications that require constant
frequency operation at no load or light
load, the LT3693 can be used.
6.3V–36V to 5V, 3.5A DC/DC
Converter with All Ceramic
Capacitors
Figure 1 shows the LT3680 producing
5V at 3.5a from an input of 6.3V to
38V with 65V transient. The circuit is
programmed for a 600kHz switching
VIN
6.3V TO 31V
VIN
BD
RUN/SS
BOOST
VC
2.2µF
50V
LT3693
SW
D
RT
FB
15k
PG
100pF
60.4k
1000pF
SYNC
536k
GND
PGOOD
100k
47µF
10V
D: B340LA
L: SUMIDA CDRH8D43-6R8
Figure 5. This negative output DC/DC converter delivers 1.2A at –5V output.
12
3.5V–27V VIN to 1.8V VOUT,
3.5A DC/DC Converter with
All Ceramic Capacitors
For output voltages lower than 2.5V,
the integrated boost diode can be tied to
the input or a separate rail greater than
2.8V. Figure 4 shows a 1.8V output
converter using the LT3680 with the
integrated boost diode tie to input. In
this application, the maximum input
voltage is 27V so that the maximum
voltage rating of Boost pin and BD pin
are not exceeded.
Negative Output from Buck
Regulators
negative output supplies are required
for many applications. The circuit in
Figure 5 can generate a negative voltage of –5V from buck regulators such
as LT3680 or LT3693. The circuit sets
the input ground reference and the
LT3680 ground reference to –5V to
generate negative 5V supply.
Conclusion
L
6.8µH
0.47µF
frequency and requires 100mm2 of
PCB. Figure 2 shows the circuit eficiency at 12V and 24V inputs. at 12V
input, the eficiency peaks above 90%
and remains high across the entire
load range.
The SynC pin is tied to the ground
to enable Burst Mode operation and
achieve high eficiency at light load.
Figure 3 shows the inductor current
and output voltage ripple under single
pulse Burst Mode operation at 10ma
load. The output voltage ripple VP–P
is less than 20mV as a result of low
ripple Burst Mode operation.
an external signal can drive the
run/SS pin through a resistor and
capacitor to program the LT3680’s
soft-start, reducing maximum inrush
current during start-up.
VOUT
–5V
1.2A
The wide input range, small size
and robustness of the LT3680 and
LT3693 make them easy it in automotive, industrial and distributed
power applications. They are highly
eficient over the entire load range.
The unique low ripple Burst Mode
operation of LT3680 helps to save
battery power life while maintaining
low output ripple. L
Linear Technology Magazine • January 2008
DESIGN FEATURES L
Monolithic 2A Buck Regulator
Plus Linear Regulator Simplifies
Wide Input Voltage Applications
by Rich Philpott
Introduction
Wide ranging voltage sources—such
as automotive batteries, unregulated
wall transformers, and industrial
power supplies—require regulation
to provide stable output voltages
during harsh input transient conditions. Simple, robust and relatively
inexpensive linear regulators offer one
solution. They produce low output
ripple and offer excellent power supply ripple rejection, but low eficiency,
high power dissipation and thermal
constraints are problems at high input-to-output ratios.
The typical alternative to the linear
solution is a high voltage monolithic
step-down switching regulator. Switching regulators offer high eficiency,
excellent line and load regulation, and
good dynamic response, but systems
with multiple outputs require multiple
switchers. This can quickly drive up
the power supply cost, space requirements, design effort and noise.
a better solution combines the
advantages of switchers and linear
regulators in a single package. The
LT3500 does just this by integrating
a high frequency switcher and a linear
regulator in a 3mm × 3mm 12-pin DFn
package, thus eliminating the need for
a second switching regulator in a dual
output system.
VIN
6V TO 36V
2.2µF
VIN
BAT54
BST
0.47µF
6.8µH
LT3500
B240A
42.2k
SHDN
SS
0.47µF
53.6k
330pF
40.2k
FB
PG
PG
22µF
8.06k
RT/SYNC LDRV
VC
ZXTCM322
1k
24.9k
GND LFB
8.06k
22µF
VOUT2
3.3V
500mA
Figure 1. Dual step-down converter for 5V at 1A and 3.3V at 1A
Get Two-for-One
and Change…
a common power supply problem
is producing 3.3V and 2.5V power
rails from a high voltage supply. To
solve this problem, the LT3500’s
switcher eficiently converts the high
voltage input to 3.3V, while the linear regulator—plus an external nPn
transistor—generates 2.5V from the
switcher’s 3.3V output. you get two
outputs for the cost of one small
package.
…Or, Just Beat the Heat
In high voltage input, single-output
systems where linear regulation is
preferred because of low output ripple
and power supply rejection, but heat
dissipation is an issue, the LT3500
also offers an elegant solution. For
example, if a linear regulated 3.3V output is needed, the LT3500’s switcher
can eficiently step-down the input
voltage to 3.6V. The integrated linear
regulator (plus an external nPn) can
generate a clean 3.3V from 3.6V with
minimal heat dissipation.
90
80
85
70
VOUT1 =
5V AT 1A
AC COUPLED
2mV/DIV
75
60
PSRR (dB)
80
EFFICIENCY (%)
VOUT1
5V
1A
SW
70
65
60
VIN = 12V
IOUT2 = 0A
FREQUENCY = 800kHz
55
50
0
0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0
LOAD CURRENT (A)
Figure 2. LT3500 switching
regulator efficiency
Linear Technology Magazine • January 2008
VOUT2 =
3.3V AT 1A
AC COUPLED
2mV/DIV
50
40
30
20
10
500ns/DIV
Figure 3. 5V and 3.3V output ripple waveforms
0
100
1k
10k
100k
1M
CENTER FREQUENCY (Hz)
10M
Figure 4. PSRR vs Frequency for VOUT2
for the application shown in Figure 1
13
L DESIGN FEATURES
Features of the LT3500
The LT3500’s switching regulator is
a constant frequency, current mode
PWM step-down DC/DC converter
with an internal 2.3a switch. The wide
3V–36V input range makes the LT3500
ideal for regulating power from a wide
variety of sources, including automotive batteries, 24V industrial supplies
and unregulated wall adapters.
The switching frequency can be
set from 250kHz to 2.2MHz via a
single resistor from the rT/Sync pin
to ground, or synchronized over the
same range by driving the pin with a
square wave. Programmable frequency
range and synchronization capability
enable optimization between eficiency
and external component size. Cycleby-cycle current limit, frequency
foldback and thermal shutdown protect the LT3500 from harmful fault
conditions.
In addition to the switching regulator, the LT3500 contains an internal
nPn transistor capable of delivering
13ma with feedback control, which
can be conigured as a linear regulator
or a linear regulator controller. The
LT3500’s soft-start feature controls
the ramp rate of the output voltages,
eliminating input current surge during start-up, while providing output
tracking between the switcher and
linear outputs. The SHDn pin has
an accurate threshold with current
hysterisis, which enables the user
to program an undervoltage lockout.
The LT3500 provides open collector
power good lags that signal when
the output voltages on both outputs
rise above 90% of their programmed
90
VIN = 12V
FREQUENCY = 800kHz
EFFICIENCY (%)
80
70
60
50
0
0.2
0.4 0.6 0.8 1.0 1.2
LOAD CURRENT (A)
1.4
Figure 7. Efficiency vs load
current for Figure 6 application
14
4.5V TO 36V
VIN
2.2µF
3.3V
LDRV
LT3500
24.9k 8.06k
1µF
BAT54
LFB
SHDN
SS
BST
0.47µF
BAT240A
0.47µF
40.2k
RT/SYNC SW
FB
VC
PG
GND PG
220pF
2.2µH
10k
VOUT1
1.8V
2A
22µF
8.06k
49.9k
Figure 5. 1.8V/2A step-down regulator
BAT54
4.5V TO 36V
VIN
2.2µF
BST
0.47µF 3.3µH
LT3500
SW
B240A
SHDN
SS
0.47µF
22µF
25.5k
8.06k
FB
RT/SYNC LDRV
VC
220pF
40.2k
49.9k
GND
PG
PG
LFB
ZXMN2A03E6
10k
24.9k
VOUT2
3.3V
8.06k
22µF
Figure 6. High efficiency linear regulator
values. The PG pin is high impedance
when the outputs are in regulation
and is typically used for a system reset
function. The PG pin is active when
the outputs are in regulation and is
used as a drive signal for an output
disconnect device. In shutdown mode
the LT3500 draws less than 12µa of
quiescent current.
High Voltage Step-Down
Regulator Plus Low Ripple
Linear Regulator
One of the most common applications
for a high voltage step-down regulator
is as a pre-regulator to other power
supplies. The pre-regulator must be
immune to harsh input transients as
it produces a stable output voltage
for other downstream regulators. In
systems where noise and ripple are
of concern, a linear regulator is often
used to step down the output of the
switcher to the desired voltage.
The LT3500 plus an external nPn
transistor as shown in Figure 1 is a
perfect it in these types of applications. The circuit takes an input from
6V to 36V and generates an interme-
diate 5V output. The LT3500’s linear
regulator is conigured as a controller
for the external nPn with its output
set to 3.3V. note that although the
load current rating for each individual
output is 2a, here the sum of both
outputs must be less than 2a. also,
care must be taken not to violate the
maximum power dissipation of the
external nPn.
The comparison of output ripple
at 1a load current shown in Figure 3
illustrates the beneit of using linear
regulation to reduce switching ripple
and noise. The excellent PSrr versus
frequency of the LT3500’s linear regulator is shown in Figure 4.
High VIN, Low VOUT, and
Boost Pin Problems Solved
Operating the LT3500 at high frequencies allows the use of small low cost
inductors and ceramic capacitors
while maintaining low output ripple.
However, due to minimum on time
restrictions (TOn(MIn) < 140ns) high
VIn-to-VOuT ratios may cause increased
output ripple. The LT3500’s adjustable
frequency allows the user to optimize
Linear Technology Magazine • January 2008
DESIGN FEATURES L
RT/SYNC
VIN
6V TO 20V 2.2µF
499k
100k
BAT54
VIN
BST
LT3500
75k
B240
0.01µF
24.9k
L2
2.2µH
GND
PG
PG
LFB
47µF
VOUT2
1.8V
SW
10k
LT3411
FB
22µF
8.06k
PGOOD
100k
LDRV
VC
PVIN
SVIN
8.06k
FB
330pF
VOUT1
3.3V
100µF
PG
L1
0.47µF 3.3µH
SS
40.2k
40.2k
SW
SHDN
330pF
+
3.3pF
BAT54
SYNC/MODE
4.02k
1k
100k
8.06k
ITH
1000pF
SD/RT
SGND
PGND
16.2k
422k
330pF
ZXMN2B14FH
22µF
VOUT3
1.2V
Figure 8. Triple output application
external component size regardless of
VIn-to-VOuT ratio.
High VIn-to-VOuT ratios also pose a
boost pin problem for most monolithic
step-down regulators. When the desired output voltage is not high enough
to fully turn on the output switch, the
boost voltage must be derived from the
input voltage or another available voltage. Taking the boost voltage from the
input poses a couple of problems. First,
the switcher eficiency suffers due to
the large drop from the boost pin to the
switch pin. Second, the boost pin is exposed to high input transients, which
may violate its ratings. The LT3500
alleviates boost voltage problems by
generating the boost voltage with the
on chip linear regulator as shown in
Figure 5. This circuit generates its
own 3.3V boost rail to regulate 1.8V
from 4.5V to 36V.
High Efficiency
Linear Regulator
In many step-down applications linear
regulators are preferred because of
their excellent PSrr and output ripple,
but are not used due to low eficiency
or thermal constraints. Figure 6 shows
another way to optimally combine the
beneits of a switcher and a linear regulator, resulting in a high eficiency, low
noise regulator. The switcher output
is set to step down the 4.5V to 36V
input voltage range to 3.5V and the
Linear Technology Magazine • January 2008
linear controller is set to generate 3.3V
from the 3.5V output of the switching
regulator. With only 200mV across
the nMOS pass device, the eficiency
of the linear regulator is only 6% less
than a switcher only solution with the
added reduction in output ripple. The
eficiency versus load current for the
application is shown in Figure 7.
NPN or NMOS Pass Transistor
nPn or nMOS pass transistors both
work well when conigured as a linear
controller, but each has its advantages
and disadvantages.
During a shorted linear output
fault, the current through the nPn is
limited to βnPn • ILDrV(MaX), while the
current through an nMOS is essentially unlimited. Since the maximum
nPn current is typically less than the
maximum switcher current, a shorted
output will lag as an error but it will not
LTC3411
SW PIN
2V/DIV
ILOAD = 250mA
affect the switcher output (assuming
the switcher load plus shorted linear
load is less than 2a). a shorted output
on the nMOS will likely cause both
outputs to crash to zero.
The minimum input voltage for the
linear controller to regulate is VOuT2 +
(Vbe or Vgs at max load) + 1.2V. The Vbe
for a nPn is typically 0.7V where as
the nMOS can range from 1.8V to 4.5V
depending on the transistor size. For
example, the minimum input voltage
for a 1.8V output is typically 3.8V for
a nPn pass transistor and 5V for a low
threshold nMOS transistor.
The power loss of the linear regulator is simply the voltage drop across
the device multiplied by the current
through the device. nMOS transistors
can be sized such that the device can
be operated with Vds less than the
saturation voltage of most nPn transistors resulting in lower power loss
(greater eficiency).
Multiple Output Application
LT3500
SW PIN
5V/DIV
ILOAD = 1.25A
VOUT
1.2V AT 1A
10mV/DIV
AC COUPLED
500ns/DIV
Figure 9. Synchronized switch waveforms
for Figure 8 application
The trend in many of today’s systems is
to provide multiple regulated voltages
from a single high voltage source to
optimize performance. When multiple
switching regulators are used, beat
frequencies along with output ripple
can cause problems with some systems. The application circuit in Figure
8 tackles these issues by synchronizcontinued on page 18
15
L DESIGN FEATURES
Efficient 48V Buck Mode
LED Driver Delivers 50mA
by Mohammad J. Navabi
Introduction
LEDs are eficient, compact and durable, and thus they are replacing
other more traditional light sources
in a variety of applications. One such
application is signage. LEDs save
energy, take less space and need less
maintenance than other sign solutions, such as neon, incandescent or
luorescent lighting.
LEDs require proper drivers to perform at their peak. a simple DC/DC
converter is not quite enough. It must
convert an input voltage to the LED
string voltage, but it must do it at constant output current. It must also be
able to dim the LEDs by adjusting the
current applied to the LED string.
Buck Mode
Constant Current LED Driver
The LT3590 is a high voltage current
mode buck mode LED driver capable of
providing a constant current to an LED
string of up to 40V total voltage. It fea-
tures internal compensation, an
internal 55V power switch and an
internal 55V Schottky diode (see
Figure 1). The part can deliver up to
50ma of DC current with eficiencies
as high as 91%. Figure 2 shows a
typical application for the LT3590,
driving a string of ten white LEDs at
50ma current.
The LT3590 uses a constant frequency, current mode architecture
resulting in stable operation over a
wide range of input voltage and output
voltage. The high switching frequency
permits the use of tiny, low proile
inductors and capacitors. The LT3590
is available in 2mm × 2mm DFn and
8-lead SC70 packages
The control scheme is detailed in the
block diagram of Figure 1. at power-up,
the bandgap reference, start-up bias,
and linear regulator are turned on. If
CTrL is pulled higher than 150mV,
the switching converter—including
VIN
48V
VIN
R1
6.81Ω
C1
1µF
+
–
REG
+
VREG
3.3V
1mA
the oscillator, PWM comparator and
error ampliier—is also turned on. The
LT3590 uses a buck mode converter
to regulate the output voltage to the
level needed for the LEDs to run at
the programmed current. It operates
similarly to conventional current
mode buck converters, but uses LED
current rather than output voltage as
the main source of feedback for the
control loop.
The CTrL pin directly controls the
regulated current sense voltage across
the sense resistor (r1 in Figure 1).
as shown in Figure 3, when VCTrL is
less than 100mV, the switcher is in
shutdown mode and the current sense
voltage and LED current are zero.
When VCTrL is greater than 150mV
and less than 1.25V, the current
sense voltage is proportional to VCTrL,
reaching a full scale value of 200mV
±5% when VCTrL is 1.25V. Further
increases in the CTrL input voltage do
EAMP
–
+
+
A = 6.25
–
+
VSENSE
–
LED
C2
1µF
C3
0.1µF
VREF
1.25V
START-UP
CONTROL
SW
VOUT
CURENT MODE
PULSE-WIDTH MODULATOR
CTRL
L1
470µH
GND
CONTROL
Figure 1. Block diagram of the LT3590
16
Linear Technology Magazine • January 2008
DESIGN FEATURES L
C2
1µF
100
90
50mA
4.02Ω
EFFICIENCY (%)
VIN
48V
C1
1µF
VIN
LED
L1
470µH
CTRL
CONTROL
80
70
60
LT3590
50
VREG
C3
0.1µF
SW
C1, C2: GRM21BR71H105KA
C3: GRM188R61E104KA
L1: MURATA LQH43CN471K03
LEDs: LUMILEDS LXCL-PWT1
GND
40
10
0
20
30
LED CURRENT (mA)
40
50
Figure 2. A buck mode converter for ten white LEDs requires very few components
Dimming Control
The LT3590 supports three types of
dimming control. as previously explained, the LED current can be set
by modulating the CTrL pin with a
DC voltage. This method is referred
to as analog dimming. alternatively,
a variable duty cycle PWM signal can
be applied to the CTrL pin through
an rC low-pass ilter. The corner
frequency of the rC network should
be much lower than the frequency of
the PWM signal. The DC value of the
iltered PWM signal seen at the CTrL
pin corresponds to the duty cycle of
the PWM signal and controls the LED
current just as in the analog dimming
scheme.
Direct PWM dimming is also possible and preferred in applications
where the chromaticity of the LEDs
High LED Count
must be maintained over the dimming
range. Dimming the LEDs via a PWM
signal essentially involves turning the
LEDs on and off at the PWM frequency.
With the LT3590, a 200:1 dimming
range is achievable for a 100Hz PWM
frequency.
In most signage and backlighting applications, it is best to place as many
LEDs as possible in the same series
string. This guarantees that all the
LEDs have the same current low and
therefore have uniform brightness
and color. The limiting factor on the
number of LEDs is the forward voltage
drop across the LED string.
The high voltage rating of the
LT3590 allows safe operation with
Onboard 3.3V Regulator
The LT3590 has a 3.3V onboard linear
regulator capable of sourcing up to
1ma of current for use by an external
device. The 3.3V regulator is available
even during shutdown. This feature
could be used to power-up an external controller from the LT3590 which
in turn can control the LED current
by applying a PWM signal directly or
through a lowpass rC ilter to the
CTrL pin. alternatively, the regulator output pin (VrEG) may be directly
connected to the CTrL pin. This way,
at power-up the LED driver is enabled
and will drive the full scale current
programmed by the feedback resistor
through the LED string.
0.25
0.20
VSENSE (V)
not increase the current sense voltage
beyond 200mV. In order to achieve
accurate LED current, 1% precision
resistors should be used.
0.15
0.10
0.05
0
0
1.0
0.5
1.5
2.0
VCTRL (V)
Figure 3. Dimming and shutdown
using the CTRL pin
C2
1µF
100
25mA
R1
4.02Ω
EFFECIENCY (%)
90
25mA
VIN
48V
C1
1µF
CONTROL
>1.5V
VIN
80
70
60
LED
L1
470µH
CTRL
LT3590
50
40
VREG
C3
0.1µF
SW
GND
C1, C2: GRM21BR71H105KA
C3: GRM188R61E104KA
L1: MURATA LQH43CN471K03
LEDs: LUMILEDS LXCL-PWT1
0
5
10
15
LED CURRENT (mA)
20
25
Figure 4. A 48V supply for two strings of ten LEDs, 25mA current
Linear Technology Magazine • January 2008
17
L DESIGN FEATURES
Indicator Light
Single-LED Indicator lights are popular in a wide range of applications from
consumer electronics to automotive.
In applications where a low voltage
supply is available, it is easy to bias
the LED using a simple series resistor.
If the input supply voltage is much
higher than the LED’s forward drop,
using a resistor is ineficient and
could generate excessive heat. also,
C2
2.2µF
80
VIN
12V OR 24V
R1
4.02Ω
50mA
12V
70
C1
1µF
VIN
CONTROL
>1.5V
75
LED
L1
220µH
CTRL
LT3590
VREG
C3
0.1µF
SW
GND
EFFICIENCY (%)
a 48V input power supply. Figure 2
shows the LT3590 driving ten white
LEDs from 48V input supply. Figure 4
shows another high voltage application
for the LT3590. Here, two strings of ten
white LEDs are driven at 25ma. In this
example we rely on the fact that the
voltage drop across each LED string
is a sum of ten average LEDs. Differences in individual LEDs are averaged
across the string. reasonable current
matching is expected in this scheme
with better than 90% eficiency for a
wide range of LED currents.
In larger applications, where
multiple LED strings are used, it is
important to match the string currents
accurately to produce uniform brightness. The LT3590’s accurate current
control makes this possible.
65
24V
60
55
50
45
40
C1: GRM21BR71H105KA
C2: GRM188R61A225KE
C3: GRM188R61E104KA
10
0
L1: MURATA LQH43CN221K03
LEDs: LUMILEDS LXCL-PWT1
20
30
LED CURRENT (mA)
40
50
Figure 5. A 12V or 24V supply for a single LED, 50mA current
in order to handle the power, bulky
power resistors are needed. another
drawback of biasing with a resistor is
that the LED current, and therefore
its brightness, depends on the input
supply voltage.
The LT3590 is the ideal solution
for driving low LED counts from
high voltage supplies. Figure 5
shows the application circuit with
one LED and a 12V or 24V input
supply. The resulting eficiencies for
both input supply voltages are also
shown in Figure 5. at 50ma LED
current, this solution provides 67%
and 61% eficiencies for the 12V and
the 24V input supplies respectively.
In comparison, the resistor-biasing
approach would yield dismal 25%
and 12.5% eficiencies.
LT3500. Operating the LTC3411 in
forced continuous mode generates a
3.3V square wave at its SW pin, which
is used to synchronize the LT3500
to the LTC3411, thus removing any
system beat frequencies. The application switching waveforms are shown
in Figure 9. The LT3500 controls
start-up, and provides power good
information via the SHDn, SS and PG
pins as shown in Figure 10.
The current capability for each output must be determined with the entire
system in mind. The maximum output
current for the LTC3411 is 1.25a,
which must be shared between the
1.8V and 1.2V outputs. The LT3500
powers the LTC3411 so the available
current to the 3.3V rail depends on
whatever power is left. For example,
assuming the 1.2V output maximum
current is 1a, the maximum current
available for the 1.8V output is 250ma.
The maximum output power for the
1.8V output is 2.25W (1.8V • 1.25a).
The load seen by the 3.3V rail due to
the LTC3411 is deined as
Conclusion
The LT3590 offers easy-to-use accurate current drive for LED strings.
Overall solution size is very small
due to its small package size and an
architecture that requires few additional components. Its high eficiency
and wide input voltage range makes it
suitable for a variety of applications,
including driving LED strings with up
to 40V of total LED voltage. L
LT500, continued from page 15
ing the switching regulators and also
providing a low ripple linear output.
The LT3500 in Figure 8 steps
down voltages between 6V and 20V
to 3.3V. The 3.3V output is fed to
the LTC3411, which generates 1.8V
and also provides the drain voltage
for the nMOS pass transistor. The
output of the nMOS provides a low
ripple 1.2V output controlled by the
3.3V
PG
1.8V
1.2V
500µs/DIV
Figure 10. Start-Up waveforms
for Figure 8 application
18
ILOAD(3.3V ) =
POUT(1.8 V )
εLTC3411(1.8 V ) • VIN(LTC3411)
2.25W
0.9 • 3.3V
= 0.75A
=
The current capability of the 3.3V
rail is 1.25a (2a maximum minus
0.75a).
Conclusion
The combination of a wide input range
switcher and a linear regulator makes
the LT3500 a perfect solution to a wide
variety of automotive, industrial and
distributed power problems. L
Linear Technology Magazine • January 2008
DESIGN FEATURES L
Synchronous Boost Converters Provide
High Voltage without the Heat
by Greg Dittmer
Introduction
The LTC3813 and LTC3814-5 reduce
the size of high voltage, high power
boost converters by incorporating
heat-saving features that eliminate
the need for large components and
heat sinks. In particular, two features
signiicantly reduce heat losses over
other high power boost solutions:
q Synchronous control eliminates
the high power loss in the diode
at high output currents
q Strong internal gate drivers
reduce switching losses at high
output voltages.
The LTC3813 can regulate output voltages up to 100V, while the
LTC3814-5 is suitable for applications
up to 60V. They both use a constant
off-time peak current mode control
architecture. Current mode control
provides tight cycle-by-cycle monitoring of inductor current and constant
off-time allows high conversion ratios
such as 7V input to 100V output at
250kHz.
Advantage of Synchronous
Control in High Power
Boost Converters
as load current increases, synchronous boost converters have a
signiicant advantage over non-synchronous boost converters due to the
low power dissipation of the synchronous MOSFET compared to that of
the boost diode in a non-synchronous
converter. For example, an output load
of 5a dissipates 5a • 0.5V = 2.5W in the
diode in a non-synchronous converter.
This high power dissipation requires a
large package (e.g. D2PaK) and a heat
sink, which adds complexity, cost and
area to the power supply. In contrast,
a synchronous converter using a typical 10mΩ MOSFET would dissipate
only (5a)2 • 0.01Ω = 0.25W. Thus the
VIN
VOUT
RNDRV
NDRV
NDRV
INTVCC
INTVCC
+
–
LTC3813
RNDRV
NDRV
10V
a. 6.2V to 14V supply available
10V
LTC3813
EXTVCC
EXTVCC
EXTVCC
D1
INTVCC
LTC3813
6.2V to
14V
VIN < 14.7V
b. INTVCC from VIN, VIN > 14V
c. INTVCC from VOUT
Figure 1. Three ways to generate IC/driver supply
VOUT
ROFF
402k
133k
COFF
100pF
1
20k
BOOST
IOFF
LTC3814-5
2
VOFF
3 V
RNG
4
PGOOD
5
ITH
6
VFB
PGOOD
CSS
1000pF
7
8
TG
SW
PGND
BG
INTVCC
RUN/SS
SGND
EXTVCC
NDRV
CC2
470pF
RFB2
1k
RC
249k
CC1
47pF
SGND
16
15
CMDSH-3
CB
0.1µF
L1
5.9µH
PGND
M1
Si7848DP
14
13
12
VIN
CIN2 5V TO 14V
1µF
20V
CIN1
68µF
20V
CDRVCC
0.1µF
11
M2
Si7848DP
D1
B1100
10
COUT1
330µF
35V × 2
VOUT
24V
4A
COUT2
10µF
50V
9
CVCC
1µF
PGND
RFB1, 29.4k
Figure 2. 5V–14V to 24V, 100W DC/DC converter
Linear Technology Magazine • January 2008
19
L DESIGN FEATURES
100
VIN = 12V
EFFICIENCY (%)
95
VOUT
200mV/
DIV
VOUT
20V/DIV
IOUT
2A/DIV
IL
5A/DIV
VIN = 5V
90
85
80
0
1
2
3
4
500µs/DIV
100µs/DIV
VIN = 12V
0A TO 4A LOAD STEP
VIN = 24V
RSHORT = 1Ω
Figure 4. Load transient performance
of the circuit in Figure 2
Figure 5. Overcurrent performance
of the circuit in Figure 2
LOAD (A)
Figure 3. Efficiency of the circuit in Figure 2
synchronous MOSFET requires only
a small SO8-size package and no heat
sink to carry the same current.
Without heat sinking, the maximum
load current of a non-synchronous
boost converter is limited by the power
dissipation of the boost diode. assuming a thermal resistance of 50°C/W on
the PC board where the boost diode
is mounted, the DC forward current
derating curves of a typical 5a Schottky
diode show that at a 50°C ambient
temperature, the maximum current
the diode can carry is about 3a.
Feature-Rich Controllers
Besides synchronous conversion,
the LTC3813 and LTC3814-5 provide many additional features for a
high performance boost converter.
no rSEnSE™ current sensing utilizes
the voltage drop across the bottom
MOSFET to eliminate the need for a
sense resistor—saving cost and simplifying board layout. For applications
that require more accurate current
limit, the LTC3813 can accommodate
a sense resistor to achieve higher accuracy.
The off-time is programmable with
an external resistor and, with an
additional resistive divider from VIn
to the VOFF, can be compensated for
changes in input voltage to keep the
frequency relatively constant over a
wide supply range. Off-times as low as
100ns can be chosen to provide high
VOuT/VIn step-up ratios. at low duty
cycles, the step up ratio is limited by
the 350ns minimum on time of the
bottom MOSFET.
a high bandwidth error ampliier
provides fast line and load transient
20
response and a precise 0.8V, ±0.5%
reference (0°C to 85°C) provides a very
accurate output voltage. an internal
undervoltage lockout comparator
monitors the driver supply voltage and
shuts down the drivers if the supply
voltage is below a threshold that is safe
for the power MOSFETs (6.2V for the
LTC3813 and 4.2V for the LTC38145). The LTC3813 also provides a pin
for undervoltage lockout on the input
supply that is programmable with a
resistive divider. Finally, the LTC3813
also has a phase-locked loop for external clock synchronization in noise
sensitive applications.
a power good pin, accurate cycleby-cycle inductor current limit, and
overvoltage protection are additional
fault protection features. Programmable soft-start ensures that the
output capacitor ramps up in a controlled manner at start-up with no
overshoot.
The LTC3814-5 provides a simpliied
feature set in a smaller more convenient package (thermally enhanced
16-lead TSSOP). The LTC3814-5 has
a maximum output voltage of 60V and
offers all the features of the LTC3813
except for input supply uVLO and
external clock synchronization.
Strong Gate Drivers
for High Efficiency
Because switching losses are proportional to the square of the output
voltage, these losses can dominate in
high output voltage applications with
inadequate gate drive. The LTC3813
and 3814-5 have strong 1Ω gate drivers that minimize transition losses,
even when multiple MOSFETs are
used for high current applications.
Dual n-channel synchronous drives
combined with strong drivers result in
very high power conversion eficiencies
(see Figures 3 and 7). The LTC3813
uses a high voltage loating driver to
drive the synchronous MOSFET at
output voltages up to 100V (60V for
the LTC3814-5).
The LTC3813 is optimized for
driving 100V MOSFETs, which are
typically rated at a VGS of 6V or higher.
as a result, the LTC3813 has an internal under-voltage lockout that keeps
the drivers off until the driver supply
is greater than 6.2V, with 500mV of
hysterisis. The LTC3814-5 is optimized
for driving logic level MOSFETs, which
are rated at a VGS of 4.5V and this
version has an internal undervoltage
lockout threshold of 4.2V with 500mV
of hysterisis.
IC/Driver Supply Regulator
The LTC3813’s internal control circuitry and top and bottom MOSFET
drivers operate from a supply voltage
in the range of 6.2V to 14V (4.2V to
14V for the LTC3814-5). If the input
supply voltage or another available
supply falls within this voltage range
it can be used to supply IC/driver
power (see Figure 1a). If a supply in
this range is not available, a single low
current external MOSFET and resistor can be added to easily generate a
regulated 10V (5.5V for the LTC38145) IC/driver supply using the internal
linear regulator circuitry (Figure 1b).
using an external pass element has
the advantage of reducing power dissipation on the IC and it also allows
the transistor to be chosen with the
Linear Technology Magazine • January 2008
DESIGN FEATURES L
VOUT
RNDRV
100k
ROFF
806k
143k
COFF
100pF
1
10k
4
IOFF
BOOST
CSS
1000pF
PGND
VOUT
50V
5A
D1
B1100
CDRVCC
0.1µF
CVCC
1µF
SGND
RUV2
10k
CIN2
1µF
50V
M1
Si7850DP
19
BG
18
DRVCC
17
INTVCC
16
EXTVCC
15
NDRV
SS
12
SGND
13
SHDN
14
UVIN
L1
10µH
VIN
12V TO 40V
CB, 0.1µF
25
SENSE+
SENSE– 21
20
BGRTN
11
SHDN
M3
ZXMN10A07F
DB
BAS19
28
27
TG
26
SW
VOFF
5 V
RNG
6
PGOOD
7
SYNC
8
ITH
9
VFB
10
PLL/LPF
PGOOD
RUV1
140k
LTC3813
CIN1
68µF
50V
M2
Si7850
×2
COUT1
220µF
63V
×2
COUT2
10µF
100V × 2
C5
1µF
PGND
CC2
330pF
RFB2
499Ω
RC
300k
CC1
150pF
RFB1
30.9k
Figure 6. 12V–40V to 50V, 250W DC/DC converter
appropriate BVDSS and power rating
for the application—a small SOT23
package will often sufice.
Figure 1c shows a solution for
applications that require the boost
converter to continue operating when
the input voltage has fallen below the
undervoltage threshold of the IC. The
cost is slightly lower eficiency. In this
circuit, the regulator is connected to
the output instead of the input. Diode
D1 supplies power to the IC until the
output voltage is high enough to generate the chip supply from the output.
When the output is in regulation, the
minimum input supply voltage is only
limited by the maximum inductor
current:
VIN(MIN) = IOUT(MAX ) •
LTC3814-5. Synchronous conversion
allows the use of two small Si7848DP
power MOSFETs and results in the
high conversion eficiency shown in
Figure 3. Since the input supply is
within the LTC3814-5’s 4.2V–14V
operating range, it can be connected
directly to the IC supply pin. nDrV
and EXTVCC are shorted to InTVCC
to disable the InTVCC regulator.
a 403kΩ resistor is connected from
VOuT to the IOFF pin to set the frequency
to 250kHz. Connecting the resistor to
the output (as opposed to a constant
supply voltage) has the advantage of
keeping the frequency constant during
output start-up. Connecting the resistive divider from VIn to the VOFF pin
VOUT
100
VIN = 36V
IL(MAX )
VIN = 24V
Since IC/Driver power loss is proportional to the output voltage in this
circuit, it is only practical for output
voltages of ~30V or less.
EFFICIENCY (%)
95
VIN = 12V
90
85
5V–14V to 24V, 100W
DC/DC Converter
The circuit shown in Figure 2 generates a 24V output voltage at 4a from
a 5V–14V input voltage using the
Linear Technology Magazine • January 2008
80
0
1
2
3
4
5
LOAD (A)
Figure 7. Efficiency of the circuit in Figure 6
sets input supply range for constant
frequency operation from 5V to 12V.
The VrnG pin is connected to VIn to
set the max sense voltage to 200mV.
This sets the nominal peak inductor
current limit to 200mV/0.01Ω = 20a
using the Si7848DP MOSFET and,
after accounting for parameter variations and inductor ripple amplitude,
provides a maximum load of 2a at VIn =
5V and 4a at VIn = 12V. Figures 4 and
5 illustrate the outstanding load transient and overcurrent performance of
the power supply.
12V–40V to 50V, 250W
DC/DC Converter
The circuit shown in Figure 6 generates
a 50V output voltage from a 12V–40V
input using the LTC3813. Since the
maximum input voltage is greater
than 14V, the LTC3813 produces a
regulated 10V from the input supply
using a ZXMn10a07F MOSFET in
a SOT23. a resistive divider is connected from VIn to the uVIn pin to set
the undervoltage lockout threshold to
10V on the input supply. This ensures
that the boost converter doesn’t hang
at start-up when the powered by a
current limited source supply when
continued on page 7
21
L DESIGN FEATURES
Wide Input Voltage Range, Dual
Step-Down Controller Reduces
by Wei Gu
Power Supply Size and Cost
Introduction
Internal Step-Up
Bias Converter
The LT3742 integrates a DC/DC
step-up converter to generate the
gate drive voltage for the n-channel
MOSFETs. The gate drive voltage is
regulated to (VIn + 7V), which permits
the use of inexpensive off-the-shelf
5V gate-drive n-channel MOSFETs,
offering higher eficiency than sublogic-level gate-drive MOSFETs. The
gate driver is capable of driving large,
low rDS(On), standard level, n-channel
MOSFETS without the need for a gate
drive buffer.
Integrating the step-up converter
also allows low dropout and 100% duty
22
L3
22µH
VIN
14V-28V
VIN
UVLO
1µF
4.7µF
SWB BIAS
LT3742
VIN
VIN
10µF
VOUT1
5V
4A
M1
G1
L1
4.7µH
RSENSE1
10mΩ
SW1
SENSE1+
SENSE1–
FB1
1.05k
150µF
L2
RSENSE2
6.9µH 10mΩ
SW2
SENSE2+
SENSE2–
FB2
PG1
150µF
1nF
680pF
200Ω
D2
PG1
RUN/SS1
VC1
51k
VOUT2
12V
4A
2.8k
D1
200Ω
10µF
M2
G2
PG2
RUN/SS2
VC2
PG2
1nF
51k
GND
680pF
D1, D2: DIODES INC. PDS540
L1: SUMIDA CDR7D43MN-4R7
L2: COILTRONICS HC8-6R9
M1, M2: RENESAS HAT2168H
Figure 1. Compact, dual-output DC/DC converter: 14V–28V input to 12V at 4A and 5V at 4A
cycle operation. This is in contrast to
the commonly used bootstrap scheme,
which does not allow 100% duty cycle
since a minimum off-time is required
to charge the bootstrap capacitor.
Continuous Inductor
Current Sensing
The LT3742 offers robust short-circuit protection thanks to continuous
inductor current sensing. a wide common-mode input range current sense
ampliier that operates from 0V to 30V
provides continuous inductor current
sensing via an external sense resistor.
a continuous inductor current sensing scheme does not require blanking
intervals or a minimum on-time to
monitor current, limitations that are
common to schemes that sense the
switch current.
The sense ampliier monitors the
inductor current independent of the
switch state, so the gate is held low
until the inductor current is below the
programmed current limit. This turnon decision is performed at the start
of each cycle, and individual switch
cycles are skipped should an over-current condition occur. This eliminates
many of the potential over-current
dangers caused by minimum on-time
requirements, such as those that can
occur during start-up, short-circuit,
or abrupt input transients. Figures 3
and 4 show the switching node voltage waveforms and inductor current
waveforms in normal operation and
in short circuit, respectively.
100
12VOUT
90
EFFICIENCY (%)
The LT3742 is an easy-to-use dual
non-synchronous DC/DC controller
for medium power step-down applications. It offers high eficiency over
a wide input voltage range (4V–30V)
and a wide output voltage range
(0.8V–30V). a 500kHz ixed frequency
current mode architecture provides
fast transient response with simple
loop compensation components and
cycle-by-cycle current limiting. an
internal step-up regulator is used to
generate the gate drive voltage, allowing the gate of the external high side
n-channel MOSFET to be driven to full
enhancement for high eficiency operation. The two channels operate 180°
out of phase to reduce the input ripple
current, minimizing the noise induced
on the input supply and reducing
the input capacitance requirement.
The device also includes individual
shutdown controls and power-good
outputs for each channel. The LT3742
is available in a small 4mm × 4mm
QFn package.
Figure 1 shows the LT3742 in a
compact, dual-output power supply. Figure 2 shows the resulting
eficiency.
5VOUT
80
70
60
50
VIN = 24V
40
0
1
2
3
LOAD CURRENT (A)
4
Figure 2. Efficiency of the converter in Figure 1
Linear Technology Magazine • January 2008
DESIGN FEATURES L
Precision UVLO Voltage
Input supply uVLO for sequencing
or start-up over-current protection is
easily achieved by driving the uVLO
with a resistor divider from the VIn
supply. The resistor divider is set such
that the divider output puts 1.25V
onto the uVLO pin when VIn is at the
desired uVLO rising threshold voltage. The uVLO pin has an adjustable
input hysteresis, which allows the IC
to resist user-deined input supply
droop before disabling the converter.
During a uVLO event, both controllers
and the gate drive boost regulator are
disabled.
2-Phase Operation
When two outputs are derived from
the same input source, any slight
difference in the switching frequencies generates a beat frequency that
is dificult to ilter. To avoid this, the
two output channels must be synchronized. The problem is that if the output
channels are switched in unison, the
input rMS current is maximized as
each channel concurrently calls for
current. This, of course, is counter
to a designers desire to minimize
input current. Minimizing rMS input
current serves to minimize the input
capacitance requirement, reduce
power loss along the input supply path
(batteries, switches, connectors and
protection circuits) and reduce radiated and conducted electromagnetic
interference (EMI).
The LT3742 eliminates the beat
frequency and minimizes the input
rMS current by interleaving the output
channels. The two channels switch at
the same frequency with 180° phase
difference between the rising edges of
G1 and G2. This 2-phase operation
minimizes input rMS current, thus
reducing the solution size, increasing
the overall eficiency and attenuating
EMI.
LT3742 employs a soft-start scheme
that directly controls the DC/DC converter output voltage during start-up.
The rising rate of this voltage is programmed with a capacitor connected
to the SS pin. The capacitor value is
chosen such that the desired ΔV/Δt
of the output results in a 1µa charge
current through the capacitor. Figure
5 shows the output voltage waveforms
during start-up.
If both outputs are always enabled
together, one soft-start capacitor can
be used with the run/SS pins tied
together.
Current Mode Control
The LT3742 uses a current mode
control architecture, enabling a higher
supply bandwidth and thereby improving line and load transient response.
Current mode control also requires
fewer compensation components than
voltage mode control architectures,
making it much easier to compensate
over all operating conditions.
Conclusion
The LT3742 provides a space-saving
and cost-saving solution over a wide
input voltage range. The LT3742 is a
versatile platform on which to build
high voltage DC/DC converter solutions that use few external components
and maintain high eficiencies over
wide load ranges. The integrated
start-up regulator facilitates true
single-supply operation. L
VSW(12V)
20V/DIV
VSW(5V)
20V/DIV
IL(12V)
2A/DIV
IL(5V)
2A/DIV
VIN = 24V
LOAD(12V) = LOAD(5V) = 2A
1µs/DIV
Figure 3. Switching node and inductor current waveforms (normal operation)
VSW(12V)
20V/DIV
VSW(5V)
20V/DIV
IL(12V)
5A/DIV
IL(5V)
5A/DIV
VIN = 24V
2µs/DIV
Figure 4. Switching node and inductor current waveforms (both outputs shorted)
VIN
10V/DIV
Soft-Start
The SS pins are used to enable each
controller independently and to provide a user-programmable soft-start
function that reduces the peak input
current and prevents output voltage overshoot during start-up. The
Linear Technology Magazine • January 2008
VOUT(12V)
10V/DIV
VOUT(5V)
20V/DIV
VIN = 24V
500µs/DIV
Figure 5. Start-up waveforms
23
L DESIGN FEATURES
Surge Stopper Protects Sensitive
Electronics from High Voltage Transients
by James Herr
Introduction
27V CLAMP
VOUT
20V/DIV
12V
100ms/DIV
Figure 2. During overcurrent or overvoltage
conditions, the current amplifier (IA) or the
voltage amplifier (VA) is called into action,
appropriately limiting the output current
or voltage. In the case of an overvoltage
condition, the load circuit continues to
operate, noticing little more than a slight
increase in supply voltage.
24
SNS
50mV
GATE
LT4356-1
OUT
30µA
–
+
IA
FB
VA
SHDN
AOUT
1.25V
1.25V
AMPLIFIER
+
12V
VCC
–
VIN
20V/DIV
OUTPUT
TO LOAD
–
80V INPUT SURGE
SUPPLY
INPUT
+
In automotive and industrial applications, electronics are subjected to high
voltage power supply spikes that can
last from a few microseconds to hundreds of milliseconds. For instance,
microsecond supply spikes result from
load steps transmitted via parasitic
wiring inductance. Longer surges,
such as an automotive load dump,
caused by a break in battery connections, is a voltage surge that stays
at an elevated level for hundreds of
milliseconds. all electronics in these
systems must be protected from high
voltage transients or risk degraded
performance or failure and costly
replacement.
The most common way of protecting electronics from voltage spikes
combines a series iron core inductor
and high value electrolytic bypass
capacitor, augmented by a high power
transient voltage suppressor (TVS) and
fuse. The bulky inductor and capacitor take up valuable board space and
are often the tallest components in
the system. Even with all this protection, supply voltage excursions are
still high enough to warrant the use
of high voltage rated components for
EN
TIMER
FLT
IN+
GND
TMR
Figure 1. Block diagram of the LT4356
The LT4356 surge stopper
eliminates the need for
bulky filtering components
by isolating low voltage
circuitry from damaging
spikes and surges found
in automotive, avionic and
industrial systems. The
LT4356 also guards against
overloads and short circuits,
and withstands input
voltage reversal.
downstream DC/DC converters and
linear regulators.
The LT4356 surge stopper eliminates the need for bulky iltering
components by isolating low voltage
circuitry from damaging spikes and
surges found in automotive, avionic
and industrial systems. The LT4356
also guards against overloads and
short circuits, and withstands input
voltage reversal.
Figure 1 shows a functional block
diagram of the LT4356. under normal
operating conditions, it drives the gate
of an n-channel MOSFET pass device
fully on so that its presence is of no
consequence to the load circuitry.
The MOSFET is called into duty as
a series limiter in case of overvoltage
or overcurrent conditions. If the input
voltage rises above a regulation point
set by the FB divider, the voltage
ampliier Va drives the MOSFET as a
linear regulator, limiting the output
voltage to the prescribed value and
allowing the load circuitry to continue
operating, uninterrupted. To protect
the MOSFET and load from short
circuits, the LT4356 includes current
limiting.
Operation
When power is irst applied, or when
the LT4356 is activated by allowing
SHDn to pull itself high, the MOSFET
is turned on gradually by slowly driving
the gate high. This soft-start minimizes
the effects of dynamic loading on the
input supply. Once the MOSFET is
Linear Technology Magazine • January 2008
DESIGN FEATURES L
80V
10mΩ
VIN
12V
IRLR2908
VOUT
16V
10Ω
12V
t
VCC
383k
SNS
GATE
12V
59k
t
OUT
FB
SHDN
DC-DC
CONVERTER
LT4356-1
IN+
VCC
4.99k
100k
SHDN GND
EN
UNDERVOLTAGE
AOUT
GND
FLT
TMR
FAULT
0.1µF
Figure 3. The spare amplifier is configured to monitor the input
voltage and indicate undervoltage through the AOUT pin.
fully on (VDS < 700mV), the En pin
goes high to activate the load circuitry,
such as a microprocessor.
During overcurrent or overvoltage
conditions, the current ampliier (Ia)
or the voltage ampliier (Va) is called
into action, appropriately limiting the
output current or voltage. In the case
of an overvoltage condition, the load
circuit continues to operate, noticing
little more than a slight increase in
supply voltage as illustrated in Figure 2. The load circuit may continue
operating if, in the case of a current
overload, suficient output voltage is
available. The timer capacitor ramps
up whenever output limiting occurs,
regardless of cause. If the condition
persists long enough for the TMr pin
to reach 1.25V, the FauLT pin goes low
RSNS
10mΩ
VIN
12V
Q3
2N3904
D1
1N4148
C2
0.1µF
R6
10Ω
By using the LT4356’s GATE
output to drive a second,
reverse-connected MOSFET,
the conventional Schottky
blocking diode and its
voltage and power losses
can be eliminated.
MOSFET and waits for a cool-down
interval before attempting to restart.
another feature of the LT4356 is
the spare ampliier (aMP), which may
be used as a power good comparator,
Q2
IRLR2908
D2*
SMAJ58A
6
to give early warning to downstream
circuitry of impending power loss.
at 1.35V the timer shuts down the
Q1
IRLR2908
R4 R5
10Ω 1M
VOUT
12V, 3A
CLAMPED
AT 16V
R3
10Ω
R1
59k
input voltage monitor or low dropout
linear regulator. In shutdown the
supply current is reduced to 5µa, permitting use in applications where the
device is left permanently connected
to a battery supply.
In the circuit of Figure 3, the output
voltage is set to 16V by an external
resistive divider. The spare ampliier is
conigured to monitor the input voltage
and indicate undervoltage through the
aOuT pin. The En pin activates the
downstream load after the MOSFET
is fully on.
Reverse Battery Protection
To protect against reverse inputs,
a Schottky blocking diode is often
included in the power path of an
electronic system. This diode not only
consumes power, it also reduces the
operating voltage range, particularly
with low input voltages such as an
automotive condition known as “cold
crank.” By using the LT4356’s GaTE
output to drive a second, reverse-connected MOSFET, the conventional
Schottky blocking diode and its voltage
and power losses can be eliminated.
Figure 4 shows a reverse protected
circuit with the second MOSFET.
under normal operating conditions
with a positive input, Q2 is enhanced
by the GaTE pin and is fully on, as is
Q1. Q3 is off and plays no role. If the
input connections are reversed and a
VIN
Q2
Si4435
Q1
IRFR2407
VOUT
15V
10k
GATE
R7
10k
5
SNS
4
GATE
VCC
3
OUT
FB
Figure 5. Low loss reverse blocking is also
possible with a P-channel MOSFET
2
R2
4.99k
LT4356-1
7
11
12
SHDN
AOUT
IN+
FLT
GND
10
*DIODES INC.
EN
TMR
1
8
9
CTMR
0.1µF
Want to know more?
visit:
www.linear.com
or call
1-800-4-LINEAR
Figure 4. A reverse protected circuit with the second MOSFET
Linear Technology Magazine • January 2008
25
L DESIGN FEATURES
Q2
2N2905A
negative voltage reaches the LT4356,
Q3 turns on and drags Q2’s gate down
to the negative input, thus isolating
Q1 and points downstream from the
negative voltage. The LT4356’s VCC,
SnS and SHDn pins are protected
from voltages of up to minus 30VDC
without damage.
Low loss reverse blocking is also
possible with a P-channel MOSFET, as
shown in Figure 5. In both cases there
is no need for the blocking MOSFET,
Q2, to be rated at a voltage any higher
than the anticipated negative input.
2.5V, 100mA
RSNS
10mΩ
VIN
12V
6
C2
0.1µF
R6
10Ω
VCC
7
LT4356-1
AOUT
IN+
SHDN
FLT
10
*DIODES INC.
and the controlled gate current set
the slew rate at the GaTE pin. The
slew rate and output capacitor, CL,
set the inrush current at start-up.
The spare ampliier is conigured as
a power good comparator, monitoring
the output voltage. r7 adds hysteresis
to eliminate motorboating.
During an overcurrent event, the
current limit loop regulates the voltage across the VCC and SnS pins
to 50mV and starts the timer. after
timeout, the pass transistor turns off
and remains off until the overcurrent
condition has passed and a cool down
period has elapsed. under conditions
a wide operating range (4V to 80V) and
accurate current limit (10% maximum)
suit the LT4356 for use as a high voltage Hot Swap™ controller, as shown
in Figure 7. The gate capacitor, C1,
Q1
FDB3632
RS
100Ω
CS
0.01µF
R4
140k
R3
10Ω
VOUT
48V, 2.5A
CLAMPED AT 71.5V
R6
27k
CL
300µF
C1
6.8nF
D1
1N4714
BV = 33V
7
6
VCC
5
4
SNS GATE
3
OUT
IN+
SHDN
12
LT4356-1
9
FB
2
EN
GND
TMR
1
AOUT
11
CTMR
0.1µF
Figure 7. High voltage Hot Swap™ controller
26
R1
226k
R2
4.02k
FLT
10
*DIODES INC.
R7
1M
R5
4.02k
R8
47k
8
EN
TMR
1
R4
249k
C3
47nF
12
8
R5
249k
9
CTMR
0.1µF
Figure 6. The LT4356’s internal spare amplifier can
drive an external PNP to provide another supply rail.
Inrush Control
D2*
SMAT70A
2
R2
4.99k
GND
RSNS
15mΩ
R1
59k
3
OUT
FB
11
The internal spare ampliier can drive
an external PnP to provide another
supply rail, as shown in Figure 6. With
2ma available from the aOuT pin, this
PnP based linear regulator can supply 100ma of current as an auxiliary,
regulated output. The spare ampliier
also inds use as an undervoltage
monitor (keeping an eye on the input
voltage as shown in Figure 3), or as
glue for other power system tasks.
The next section shows how the spare
ampliier is conigured as a power good
comparator.
VOUT
12V, 3A
CLAMPED AT 16V
R3
10Ω
4
GATE
5
SNS
Auxiliary Output Voltage
VIN
48V
Q1
IRLR2908
D2*
SMAJ58A
R6
100k
C5
10µF
PWRGD
of overcurrent, MOSFET safe operating
area stress increases as the drainsource voltage drop increases. The
LT4356 monitors VDS and shortens
the timer interval in proportion to increasing VDS. This way a brief, minor
overload may persist for a longer time
interval than a highly stressful output
short circuit condition, ensuring the
MOSFET operates within its safe operating area.
While MOSFET protection is important, the real beneit of current limit
is recognized only after surviving a
short circuit: the upstream fuse also
survives, and need not be replaced.
Conclusion
The electronic content in automotive
and industrial systems is becoming increasingly plentiful and sophisticated,
yet power sources remain riddled with
spikes and surges. as more and more
features are packed into the electronics, less and less space is available
for conventional methods of iltering,
clamping and rejecting the noise. The
LT4356 surge stopper offers a means
for reducing the necessary board
space, while at the same time cutting
the heat dissipation and voltage loss
associated with blocking diodes and
ilter inductors. Higher eficiency and
wider usable voltage range allow more
functionality to be incorporated into
space-constrained products. L
Linear Technology Magazine • January 2008
DESIGN IDEAS L
USB Compatible Li-Ion Battery
Charger and Dual Buck Regulators
in a Single 3mm × 3mm QFN
by Aspiyan Gazder
Introduction
Manufacturers of handheld devices
such as MP3 players are always looking to reduce system size and cost,
even as they increase performance
and functionality. The only way to do
so is to integrate functions at the IC
level. For applications powered from
a single Li-Ion cell, the LTC3559
provides a single chip solution that
charges a Li-Ion cell while eficiently
generating two supply voltage rails to
power the device.
The LTC3559 is a uSB compatible
battery charger and two monolithic
synchronous buck regulators integrated into a low proile 3mm × 3mm
16-lead QFn package. The battery
charger has all the features that a
stand alone battery charger might
offer, such as an nTC input for temperature qualiied charging, internal
timer termination and bad battery
DESIGN IDEAS
USB Compatible Li-Ion Battery Charger
and Dual Buck Regulators in a Single
3mm × 3mm QFN ...............................27
aspiyan Gazder
Entire RGB LED Driver Fits in
Miniscule 3mm × 2mm Package ........29
Zachary Lewko
USB Power Manager with High Voltage
2A Bat-Track Buck Regulator............30
nancy Sun
Complete 3-Rail Power Supply
in a 4mm × 4mm QFN Package ..........32
John Canield
I2C Quad Buck Regulator Packs
Performance, Functionality, Versatility
and Adaptability in a 3mm × 3mm QFN
.........................................................34
Joe Panganiban
µModule Regulators Shrink Power
Supply Size and Design Effort...........36
David ng
Small, High Efficiency Solution Drives
Two Piezo Motors ..............................38
Wei Gu
Linear Technology Magazine • January 2008
ADAPTER
4.5V TO 5.5V
UP TO
950mA
510Ω
1µF
110k
VCC
BAT
+
PVIN
2.2µF
NTC
SINGLE
Li-lon CELL
2.7V TO 4.2V
28.7k
100k
NTC
NTH50603N01
LTC3559
4.7µH
CHRG
887Ω
3.3V AT
400mA
SW1
1.02M
PROG
22pF
10µF
FB1
324k
SUSP
HPWR
DIGITALLY
CONTROLLED
SW2
MODE
1.2V AT
400mA
324k
EN1
22pF
FB2
10µF
649k
EN2
GND
4.7µH
EXPOSED PAD
Figure 1. Full featured USB battery charger and dual buck regulator in one 3mm × 3mm IC
detection. a constant current/constant voltage algorithm is employed
to charge a battery. Only a single
resistor at the PrOG pin is required
to program the charge current up to
950ma. The HPWr input provides
the lexibility to deliver either 100%
or 20% of the programmed charge
current. For applications operating
from a uSB source, charge current
can be programmed to either 100ma
or 500ma per uSB speciications.
The two buck regulators have a
current mode architecture, which provides a quick response to load steps.
To meet the noise and power requirements of a variety of applications, the
buck regulators can be operated in
either Burst Mode operation or pulse
skipping mode. The buck regulators
also have a soft start feature that
prevents large inrush currents at
start up.
at high load currents, the buck
regulator operates as a constant frequency PWM controlled regulator. at
light load currents, pulse skipping is
the normal behavior for a switching
regulator when the inductor current
is not allowed to reverse.
To improve eficiency in light load
conditions, the LTC3559 offers Burst
Mode operation. When in Burst Mode
operation, the buck regulator automatically switches between ixed
frequency PWM control or hysteretic
control, as a function of the load current. at light loads, the regulator has
an output capacitor charging phase
followed by a sleep phase. During the
sleep phase, most of the buck regulators’ circuitry is powered down, saving
battery power. as the load current
increases, the sleep time decreases
to the point where the buck regulator switches to a constant frequency
PWM operating mode—equivalent to
pulse skipping mode at higher output
currents.
Figure 1 shows the LTC3559 with
the nTC input biased using three resistors. a 3-resistor bias provides the user
27
L DESIGN IDEAS
with the lexibility to program both the
upper and lower battery temperature
points that are considered safe for
charging the battery. In this example,
the nTC hot and cold trip points are
set for approximately 55°C and 0°C,
respectively.
One of the buck regulators is
programmed for 3.3V at its output.
When the BaT pin voltage approaches
3.3V, the buck regulator operates in
dropout. an LED at the CHrG pin
gives a visual indication of the battery
charge status.
Figure 2 shows an actual circuit
similar to that shown in Figure 1,
illustrating how little board space
is required to build a full featured
LTC3559 application. Figure 3 shows
how much more eficient Burst Mode
operation is at light loads as compared
to pulse skipping mode.
a basic sequencer function can be
built for the buck regulator outputs
by driving the enable pin on one buck
Figure 4 helps to explain this scenario.
The current being delivered at the BaT
pin is 500ma. Both buck regulators
are enabled. The sum of the average
input currents being drawn by both
buck regulators is 200ma. This makes
the effective battery charging current
only 300ma. If the HPWr pin were tied
low, the BaT pin current would be only
100ma. With the buck regulator conditions unchanged, this would cause the
battery to discharge at 100ma.
Conclusion
Figure 2. A USB battery charger and two buck
regulators small enough to fit in the latest cell
phones, PDAs and MP3 players
regulator with the output of the other
buck regulator. For proper operation,
the BaT and PVIn pins must be tied
together. If a buck regulator is enabled
while the battery is charging, the net
current charging the battery will be lower
than the actual programmed value.
The LTC3559 is ideally suited for
space-constrained applications that
are powered from a single Li-Ion cell
and that need multiple voltage supply
rails. The high switching frequency
allows the use of small low proile
external inductors. The high eficiency
buck regulators and Burst Mode operation combine to maximize battery
life, extending battery operation time
between charge cycles. L
500mA
100
USB (5V)
Burst Mode
OPERATION
90
PROG
EFFICIENCY (%)
70
RPROG
1.62k
60
+
SINGLE Li-lon
CELL 3.6V
200mA
+
2.2µF
LTC3559
PULSE SKIP
MODE
50
300mA
BAT
PVIN
80
SUSP
40
HIGH
30
HIGH
20
VOUT = 1.2V
PVIN = 2.7V
PVIN = 4.2V
10
0
0.1
1
10
ILOAD (mA)
100
HIGH
LOW (PULSE SKIP MODE)
HPWR
SW1
VOUT1
EN1
SW2
VOUT2
EN2
MODE
1000
Figure 4. The net current charging the battery depends
on the operating mode of the buck regulators.
Figure 3. Buck regulator efficiency
LT580, continued from page 10
voltages where these problems might
occur.
The shutdown pin comparator
has voltage hysteresis with typical
thresholds of 1.32V (rising) and 1.29V
(falling). resistor ruVLO2 is optional
but can be included to reduce overall
uVLO voltage variation caused by
variations in SHDn pin current. a good
choice for ruVLO2 is 10k ±1%. after
choosing a value for ruVLO2, ruVLO1
can be determined from either of the
following:
28
VCC
RUVLO1 =
VIN − 1.32V
 1.32V 
R
 + 11.6µA
 UVLO2 
+
or
RUVLO1 =
VIN − − 1.29 V
 1.29 V 
R
 + 11.6µA
 UVLO2 
where VIn+ and VIn- are the VIn
voltages when rising or falling respectively.
Conclusion
The LT3580 is a smart choice for many
DC/DC converter applications. It’s
packed with features without compromising performance or ease of use and
is available in tiny 8-lead packages.
The accurate and adjustable clock,
2a/42V power switch, wide input
voltage range, integrated soft-start
and a conigurable SHDn pin make
the LT3580 an ideal choice for many
DC power supply needs. For additional
information and a complete data sheet
visit www.linear.com. L
Linear Technology Magazine • January 2008
DESIGN IDEAS L
Entire RGB LED Driver Fits in Miniscule
by Zachary Lewko
3mm × 2mm Package
Introduction
The LTC3212 charge pump rGB LED
driver is an ideal solution for highly
space-constrained portable devices
such as cellular phones, PDas, digital cameras and media players. The
LTC3212 features an internal low
noise charge pump utilizing a single
external lying capacitor. This charge
pump operates in 1× mode until one
of the LEDs drops out of regulation,
after which it switches to 2× mode,
automatically maintaining proper LED
current while reducing power loss
and minimizing switching noise. The
LTC3212 is designed with lexibility in
mind and can be used for driving rGB
backlights, keypad back lighting, or a
general purpose LED such as a multicolor status indication LED.
Battery/Supply Voltage
The LTC3212 is designed to operate
from 2.7V to 5.5V inputs, making it an
ideal LED driver for battery powered
and uSB powered devices.
The LTC3212’s charge pump is
enabled when it is necessary to prevent an LED driver from dropping
out of regulation. This reduces losses
and minimizes noise by keeping the
charge pump operating in 1× mode
as long as possible. Once the charge
pump is operating in 2× mode, the
control algorithm ensures switching
noise is reduced by limiting the slew
1µF
CM
VIN
2.7V TO 5.5V
CPO
LTC3212
1µF
LEDR
LEDEN
LEDG
ISETB
ISETR
ISETG
LEDB
Linear Technology Magazine • January 2008
G
R
1µF
B
INDIVIDUAL WHITE
SETTINGS MODE
11.8k
LEDR
LEDG
LEDB
15mA
15mA
15mA
13.5mA
15mA
11.2mA
3212 TA01a
Figure 1. The LTC3212 LED RGB LED driver with minimal external components
rate of the lying capacitor pins and
by reducing the ripple current on the
input supply.
The part has a soft-start circuit
which prevents large inrush currents
on start-up and during a mode switch.
The CPO pin has short circuit protection to protect the part in the event of
a short on the charge pump output.
The CPO output is switched to high
impedance mode when the part enters
shutdown mode.
Compact Solution
With a minimum setup the LTC3212
can be conigured to use only four
external components, three capacitors and one resistor (see Figure 1).
These few external components along
with the small 3mm × 2mm package
make the LTC3212 ideal for space
constrained applications as shown
in Figure 2.
LED Control
Figure 2. A typical LTC3212 RGB LED driver
occupies minimal board real estate.
CP
VIN
The LTC3212 is programmed using a
single wire interface, making it very
easy to integrate into applications
where the controlling device has
limited pins available. The LTC3212
can be programmed to enable any
combination of the red, green and blue
LEDs, resulting in seven colors from
the rGB LED (see Table 1). When all
of the LEDs are enabled the currents
are automatically adjusted to a ratio
that results in white light.
Table 1. LTC3212 Programming Table
Pulses
Red
Blue
Green
0
off
off
off
1
off
off
ON
2
off
ON
off
3
off
ON
ON
4
ON
off
off
5
ON
off
ON
6
ON
ON
off
7+
White Mode
Intensity Setting
The operating currents of the LEDs
can all be the same, two the same, or
they can all be conigured independently—requiring one, two or three
external resistors, respectively. If
independent control of an LED is not
needed, tie its ISET pin to VIn and the
current defaults to the setting of the
ISETG resistor.
Conclusion
The LTC3212 is an rGB LED driver
optimized to be a simple and compact
solution for driving an rGB LED from
a 2.7V to 5.5V supply. The LTC3212 is
well suited for applications requiring
an LED driver with accurate programmable current sources, and compact,
low noise operation. L
29
L DESIGN IDEAS
USB Power Manager with
High Voltage 2A Bat-Track
Buck Regulator
by Nancy Sun
Introduction
Personal navigation devices, HDDbased media players, automotive
accessories, and other handheld products draw on an array of power sources
for recharging their batteries. These
sources include uSB (nominally 5V),
low voltage wall adapters (4.5-5.5V),
high voltage wall adapters (12V–24V),
FireWire (8V–33V) and automotive
batteries (nominally 12V). The large
supply of available sources leads to an
increasing need for handheld devices
that can accept a wide range of multiple
input voltages without the need for
myriad external power adapters.
The LTC4090 is designed to accommodate both uSB and high voltage
sources by integrating a high voltage
2a switching buck regulator, a uSB
input, a PowerPath™ controller and a
linear battery charger into a compact,
thermally enhanced 3mm × 6mm
HIGH
(6V TO 36V)
VOLTAGE
INPUT
package. Figure 1 shows a complete
solution that its into less than 3cm2
with all components on one side of the
PCB (Figure 2).
Complete PowerPath
Controller
The LTC4090 is a complete PowerPath
controller for battery powered applications. It is designed to receive power
from a uSB input (or 5V wall adapter),
a high voltage source, and a single-cell
Li-Ion battery. The PowerPath controller distributes the available power,
with the load on the OuT pin taking
precedence and any remaining current
used to charge the Li-Ion battery. The
high voltage input takes priority over
the uSB input (i.e., if both HVIn and
In are connected to power sources,
load current and charge current are
provided by the HVIn input). Figure 3
HVIN
C1
1µF
50V
1206
BOOST
SW
shows a simpliied block diagram of
the PowerPath operation.
USB Input Current Limit
Power supplies with limited current
capability (such as uSB) should be
connected to the In pin, which has a
programmable current limit. The input
current limit is programmed using a
single external resistor, rCLPrOG, from
the CLPrOG pin to ground. In Figure
1, a 2.1kΩ CLPrOG resistor has been
chosen to program the input current
limit to 476ma in high power mode
(when the HPWr pin is pulled high)
or 95ma in low power mode (when the
HPWr pin is pulled low). This ensures
that the application complies with the
uSB speciication. The sum of battery
charge current and the load current
(which takes priority) will not exceed
the programmed input current limit.
L1
6.8µH
0.47µF
16V
C3
22µF
6.3V
1206
D1
HVEN
HIGH VOLTAGE
INPUT PRESENT
IN
USB
680Ω
59k
1%
HPWR
LTC4090
HVOUT
VC
270pF
SUSP
0.1µF
2.1k
1%
HVPR
Q1
1k
TIMER
LOAD
OUT
4.7µF
6.3V
CLPROG
71.5k
1%
40.2k
1%
GATE
Q2
PROG
BAT
RT
PG
SYNC
+
VNTC
10k
1%
Li-Ion
BATTERY
NTC
T 10k
D: DIODES INC. B360A
L: SUMIDA CDR6D28MN-GR5
Q1, Q2: SILICONIX Si2333DS
680Ω
CHRG
CHARGING
Figure 1. Li-Ion battery charger accepts both USB and high voltage inputs
30
Linear Technology Magazine • January 2008
DESIGN IDEAS L
Ideal Diode from BAT to OUT
an ideal diode function automatically
delivers power to the load via the
ideal diode circuit between the BaT
and OuT pins when the load current
exceeds the programmed input current limit or when the battery is the
only supply available. Powering the
load through the ideal diode instead
of connecting the load directly to the
battery allows a fully charged battery
to remain fully charged until external power is removed. The LTC4090
has a 215mΩ internal ideal diode as
well as a controller for an optional
external ideal diode. In Figure 1, an
external P-channel MOSFET, Q2, is
shown from BaT to OuT and serves
to further increase the conductance
of the ideal diode.
High Voltage Buck Regulator
The LTC4090 has an operating input
voltage range of 6V to 36V and can
withstand voltage transients of up
to 60V. The buck converter output,
HVOuT, maintains approximately
300mV across the battery charger
from OuT to BaT so that the battery
Battery Charger Features
Figure 2. A complete LTC4090-based USB
Power Manager with a 2A high voltage buck
regulator fits into 3cm2.
can be eficiently charged with the
linear charger. The minimum VHVOuT
is 3.6V to ensure the system can operate even if the battery is excessively
discharged. as shown in Figure 1, an
external PFET, Q1, between HVOuT
and OuT is controlled by the HVPr
pin and allows OuT to supply power
to the load and to charge the battery.
The buck converter is capable of up
to 2a of output current.
The LTC4090 battery charger uses a
unique constant-current, constantvoltage, constant-temperature charge
algorithm with programmable charge
current up to 1.5a and a inal loat
voltage of 4.2V ±0.8%. The maximum
charge current is programmed using
a single external resistor, rPrOG, from
the PrOG pin to ground. In Figure 1,
a 71.5k PrOG resistor programs the
maximum charge current to 700ma.
However, in the case where only a uSB
input is present, charge current is reduced to ensure that the programmed
input current limit is not exceeded.
For the circuit in Figure 1, when only
a uSB input is present, the actual
maximum charge current is reduced
to 476ma.
In typical operation, the charge cycle
begins in constant-current mode. a
strong pull-down on the CHrG pin
indicates that the battery is charging.
In constant current mode, the charge
current is set by rPrOG. When the battery approaches the inal loat voltage
of 4.2V, the charge current starts to decontinued on page 42
SW
HVIN
L1
Q1
D1
HIGH VOLTAGE
BUCK REGULATOR
HVOUT
+
4.25V (RISING)
3.15V (FALLING)
C1
–
HVPR
19
+
–
ENABLE
LOAD
75mV (RISING)
25mV (FALLING)
OUT
21
OUT
USB CURRENT LIMIT
CC/CV REGULATOR
CHARGER
+
–
30mV
+
–
IN
+
–
30mV
+
EDA
IDEAL
DIODE
GATE
21
–
BAT
BAT
21
+
Li-Ion
Figure 3. Simplified block diagram of the LTC4090 PowerPath operation
Linear Technology Magazine • January 2008
31
L DESIGN IDEAS
Complete 3-Rail Power Supply in a
4mm × 4mm QFN Package by John Canfield
Introduction
Battery-powered portable electronic
devices such as portable media players, handheld PCs, and GPS receivers
typically require several internal
power supply rails: a 3.0V or 3.3V
supply for audio, motor drivers, and
micro hard disk drives; a 1.2V or 1.5V
rail for a logic core; and often a 1.8V
supply to support Flash memory. For
devices supplied by a Li-Ion battery,
the power system is further complicated by the fact that the 3.0/3.3V
output rail lies within the discharge
voltage range of the battery, thereby
mandating a power supply solution
that can step the input voltage up or
down depending on the battery's state
of charge. In addition, most systems
require speciic power-up sequencing
between the multiple output voltage
rails to ensure consistent and reliable
system initialization.
Figures 1 and 2 show how all of
these requirements can be met with
a single tiny IC and relatively few
additional components. The heart of
this complete power supply system is
the LTC3520, which includes a highVIN
2.2V TO
5.5V
Figure 1. Complete triple-output supply:
Li-Ion to 3.3V, 1.8V, and 1.5V
eficiency, internal 1a buck-boost
converter, a 600ma synchronous buck
converter and an LDO controller, all in
a 4mm × 4mm QFn package.
The LTC3520’s buck-boost converter utilizes an advanced switching
algorithm to precisely regulate the
output voltage with input voltages that
are above, below, or even equal to the
output voltage. Mode transitions occur
seamlessly and high eficiency and low
noise performance are maintained
across all operational modes. The
4.7µH
22µF
4.7µH
249k
10µF
synchronous buck converter operates with current-mode control and is
internally compensated to reduce the
number of external components. If the
input voltage falls below the minimum
buck regulation voltage, the buck
converter automatically transitions
to low dropout mode to extend battery
life. Pin-selectable Burst Mode® operation improves light-load eficiency and
reduces the no-load input current for
both converters to only 70µa.
The extensive array of programmable features on the LTC3520 provide
the lexibility needed to meet the
requirements of a wide range of applications. Both the buck and buck-boost
converters are controlled by a common
oscillator. a single external resistor
sets the switching frequency, making it possible to optimize eficiency
and application size. Both converters
feature voltage mode soft-start with
ramp rates which are independently
set via small external capacitors. The
output voltage of each converter is
programmed via an external resistor
divider. The buck-boost output voltage
PVIN1 PVIN2 PVIN3 SVIN SW1A
SW1B
VOUT1
SW2
27pF
470pF
56pF
VC1
FB2
200k
47µF
1M
15k
10k
FB1
LTC3520
54.9k
SS1
324k
0.047µF
VOUT2
1.8V
600mA
RT
PWM1
BURST PWM
AOUT
CMPT591E
PWM2
SD3
0.047µF
VOUT1
3.3V
500mA
1A FOR
VIN ≥ 3V
100k
SD2
OFF ON
AIN
SS2
RSEQ
1M
33pF
VOUT
1.6V
200mA
4.7µF
115k
SD1
CSEQ
4.7µF
PGND1 SGND PGND2
Figure 2. Sequenced start-up, triple-output converter
32
Linear Technology Magazine • January 2008
DESIGN IDEAS L
can be set as high as 5.25V or as low
as 2.2V. When conigured for a 3.3V
output, the buck-boost can provide
up to 1a load current for input voltages greater than 3V and supports a
500ma load down to an input voltage
of 2.2V. The buck converter delivers
up to 600ma and its output can be
set as low as 0.8V.
Three Output Rails with
Sequenced Start-Up
In many applications, the low voltage
rails that supply the logic core and
memory must be powered and in
regulation before the higher voltage
supply for the peripheral devices is
activated. This provides time for the
processor to initialize and control the
states of its logic outputs to ensure
reliable and consistent initialization
of the system. Figure 2 shows powerup sequencing achieved by using the
buck converter soft-start pin to enable
the buck-boost via the SD1 pin after
a programmable delay created by the
rC ilter composed of resistor rSEQ
and capacitor CSEQ.
Figure 3 shows the output voltages
for this application circuit during
start-up. The buck output voltage
begins its soft-start period soon after
the rising edge of SD2 and the LDO
output rises coincident with the buck
output. approximately 5ms after the
buck reaches regulation, the buckboost soft-start commences. The
length of this delay can be adjusted
via the time constant of the rC ilter,
while the ramp rate of each converter's
soft-start can be independently controlled by the value of the respective
soft-start capacitor. In shutdown, SD2
is held low, which internally forces SS2
low, thereby ensuring the buck-boost
converter remains disabled as well.
Low Battery and
Power-Good Detection
In applications where the third output
rail is not required, the LDO controller can be used instead as a general
purpose comparator. One possibility is
to utilize the uncommitted ampliier as
a low battery indicator with the circuit
shown in Figure 4a. The low battery
Linear Technology Magazine • January 2008
SD2, SD3
5V/DIV
BUCK VOUT
1V/DIV
LDO VOUT
1V/DIV
BUCK-BOOST VOUT
1V/DIV
5ms/DIV
Figure 3. Output voltages during sequenced start-up
output can then be used to provide
the system processor with feedback on
the state of the battery. The uncommitted ampliier is not disabled by the
undervoltage lockout, which allows
the low-battery indicator to remain
functional down to 1.6V typically,
well below the undervoltage lockout
threshold of the LTC3520.
It is also possible to use the uncommitted ampliier as a high accuracy
power-good indicator for either the
buck or buck-boost output rail. The
resultant power-good signal can then
be utilized to enable the opposite channel, providing high accuracy supply
sequencing. For example, the circuit
shown in Figure 4b creates a powergood output for the buck converter
and initiates the buck-boost converter
only after the buck output reaches the
power-good threshold set by resistors
r1 and r2.
USB-Powered
Triple-Output Supply
The uSB specification mandates
that the output voltage provided by a
high power port be maintained in the
range of 4.75 to 5.25V. However, once
resistive drops in the uSB cable and
connectors are taken into account,
along with the potential voltage drop
across an upstream bus-powered
hub, a uSB peripheral must be
LBO
AOUT
able to function with input voltages
over a wider range of 4.25 to 5.25V.
Furthermore, the input voltage seen
by the peripheral can vary dynamically between these limits based on
the particular cable, host, and load
current being drawn. In such applications, the buck-boost converter of
the LTC3520 can provide a restored
5V output rail independent of loading
and cable resistance. additionally, the
buck converter and LDO can be conigured to provide two lower voltage
outputs, such as 3.3V and 1.8V logic
supplies. If both of these additional
voltage outputs are not required, the
uncommitted ampliier can instead
be conigured to monitor the input
uSB voltage to inform the processor
of the presence of a valid uSB input
voltage level.
Conclusion
With its small size, lexible programmability, and high eficiency, the
LTC3520 is well suited to meet the
multiple output power supply needs
of most Li-Ion powered electronic
devices. In addition, the LTC3520 is
ideal for systems powered from uSB
or low voltage wall adapters, which
require an output voltage rail that
lies within the expected input voltage
range due to resistive drops in the
supply path. L
PGOOD
AOUT
SD1
VOUT
VBAT
R1
LTC3520
R1
AIN
R2
LTC3520
AIN
R2
3520 F02
Figure 4. Implementation of low battery and power-good indicators
33
L DESIGN IDEAS
I2C Quad Buck Regulator Packs
Performance, Functionality, Versatility
and Adaptability in a 3mm × 3mm QFN
by Joe Panganiban
I2C Programmable
The LTC3562 is an I2C quadruple erating modes to satisfy the various Output Voltages
Introduction
step-down regulator composed of four
extremely versatile monolithic buck
converters. Two 600ma and two 400ma
highly adjustable step-down regulators
provide a total of 2a of available output
current, all packed inside a 3mm × 3mm
QFn package. all four regulators are
2.25MHz, constant-frequency, current
mode switching buck converters whose
output voltages and operating modes
can be independently adjusted through
I2C control. The 2.7V to 5.5V input
voltage range makes it ideally suited for
single Li-Ion battery-powered applications requiring multiple independent
voltage supply rails.
I2C Programmable
Operating Modes
all four LTC3562 step-down regulators have the unique ability to be
programmed into four distinct op-
noise/power demands of a variety of
applications. These four modes are
pulse skipping mode, Burst Mode operation, forced Burst Mode operation,
and LDO mode.
Pulse skipping mode allows the
regulator to skip pulses at light load
currents, providing very low output
voltage ripple while maintaining high
eficiency. Burst Mode operation and
forced Burst Mode operation deliver
bursts of current to the buck output and regulate the output voltage
through hysteretic control, giving the
highest eficiency at low load currents.
In LDO mode, the bucks are converted
to DC linear regulators and deliver
continuous power from the switch
pins through the inductor, providing
the lowest possible output noise as
well as the lowest no-load quiescent
current.
another unique feature of the LTC3562
is its ability to adjust the output voltage
of each regulator through I2C control.
The chip contains two different lavors
of output adjustable regulators. The
Type a regulators (r600a, r400a)
have programmable feedback servo
voltages, while the Type B regulators (r600B, r400B) have directly
programmable output voltages that
do not need external programming
resistors.
The Type a regulators use external
feedback resistors to set the output
voltage based on a programmable
feedback servo voltage. The feedback
voltage values can be programmed from
800mV (full scale) down to 425mV in
25mV steps. This results in 16 possible
feedback servo voltages, and thus 16
different output voltage settings for the
same external programming resistors.
Table 1. Feature comparison of the LTC3562’s four integrated regulators (two 600mA and two 400mA)
R600A
R400A
R600B
R400B
Type
A
A
B
B
Output Current
600mA
400mA
600mA
400mA
I2C Programmable
Operating Modes
Pulse Skip
Burst
Forced Burst
LDO
Pulse Skip
Burst
Forced Burst
LDO
Pulse Skip
Burst
Forced Burst
LDO
Pulse Skip
Burst
Forced Burst
LDO
Feedback Servo
Voltage
I2C Programmable
425mV–800mV
25mV steps
(16 settings)
I2C Programmable
425mV–800mV
25mV steps
(16 settings)
600mV (Fixed)
600mV (Fixed)
I2C Programmable
600mV–3.775V
25mV steps
(128 settings)
No
34
Output Voltage
Adjustable using
External Resistors
Adjustable using External
Resistors
I2C Programmable
600mV–3.775V
25mV steps
(128 settings)
RUN Pins
Yes
Yes
No
Linear Technology Magazine • January 2008
DESIGN IDEAS L
100k
C5
10µF
Li-Ion BATTERY
3.4V TO 4.2V
SDA
VIN
VOUT 600B
3.3V
600mA
VOUT 400B
1.2V
400mA
L3
3.3µH
C3
10µF
DVCC
R5
100k
LTC3562
SW600B
L1
3.3µH
POR600A
SW600A
OUT600B
R1
634k
FB600A
L4
4.7µH
C4
10µF
SCL
VOUT 600A
1.8V
600mA
C6
10pF
C1
10µF
RUN600A
POR SCL SDA
VCC CORE
VCC I/O
MICROPROCESSOR
SW400B
OUT400B
R2
499k
VOUT 400A
2.5V
400mA
L2
4.7µH
RUN400A
SW400A
R3
1070k
FB400A
PGND AGND
C7
10pF
C2
10µF
R4
499k
L1, L3: TOKO 1098AS-4R7M
L2, L4: TOKO 1098AS-3R3M52
Figure 1. The LTC3562 configured in a quad step-down converter with pushbutton control and power sequencing.
RUN pins and
Default Settings
I2C applications generally have a
microprocessor in charge of the I2C
communications between the various system blocks. a multi-channel
buck converter such as the LTC3562
provides an excellent solution for
eficiently stepping down the microprocessor’s core and I/O supply
voltages from a higher input supply
or battery. at the surface, using an
I2C controllable voltage converter to
generate the microprocessor’s power
supplies seems to pose a bootstrap
problem at system start-up. If the
microprocessor initially has no power
and thus there is no I2C control, what
programs the LTC3562’s output to the
proper voltage for the patiently waiting
microprocessor?
Linear Technology Magazine • January 2008
100
90
80
EFFICIENCY (%)
The Type B regulators (r600B,
r400B) do not require external programming resistors at all because they
are integrated inside the chip. These
internal feedback resistors not only
save valuable board space, they are
also I2C programmable. The values
of the internal feedback resistors can
be adjusted through I2C control to
directly program the regulator output
voltages from 0.6V to 3.775V in 25mV
increments. That is 128 possible output voltage settings for each Type B
regulator.
FORCED
Burst Mode
OPERATION
600mA
BUCKS
70
60
PULSE SKIP
50
40
Burst Mode
OPERATION
30
20
VIN = 3.8V
VOUT = 2.5V
10
0
0.01
0.1
1
10
IOUT (mA)
100
1000
Figure 2. Efficiency of the 2.5V regulator
The LTC3562 gets around this
start-up issue by providing individual
run pins for the two Type a regulators. These run pins bypass the I2C
controls and enable the regulators if
I2C is unavailable. When a run pin
is used, the corresponding Type a
regulator is enabled in a default setting, which is 800mV for the feedback
voltage and pulse skipping mode for
the operating mode. Once I2C becomes
available to the system, these default
settings can always be modiied on the
ly through I2C.
Pushbutton Control and
Power Sequencing
Figure 1 shows an application circuit
that uses the LTC3562 to power the
core and I/O supplies of a system microprocessor. The run pin of r600a
connects to a pushbutton circuit with
a pull-up resistor used to power on the
system. When the button is pushed,
the run pin goes low which enables
r600a to ramp up the power supply
for the microprocessor’s core. The run
pin of r400a is tied to r600a’s poweron-reset output signal (POr600a).
Once r600a reaches regulation,
POr600a goes high after a 230ms
time delay, which would then enable
r400a to power the I/O supply of the
microprocessor.
after both the core and I/O supplies
are up, the microprocessor could then
communicate back to the LTC3562
through I2C to program the part such
that it keeps r600a enabled even after
the pushbutton stimulus is removed.
The microprocessor then can enable
regulators r600B and r400B in any
mode and program the output voltages
to desired levels.
Low Power Adaptability
The ability to change the operating
modes and output voltages at any
time allows the LTC3562 to adapt to
the constantly changing demands of
many high performance systems. an
example of this adaptability would be
during lower power standby operation
in handheld battery-powered systems.
When going into standby mode, the
regulators can be programmed into
Burst Mode operation or forced Burst
continued on page 7
35
L DESIGN IDEAS
µModule Regulators Shrink Power
Supply Size and Design Effort by David Ng
Introduction
When it comes to high density, eficient
power supplies, switching regulators
are a top choice, but what if a project
lacks suficient design resources to
properly layout and test a switching
power supply circuit? Like any other
system, switching power supplies require component selection, derating,
simulation, prototyping, board layout,
analysis and design veriication testing. Design engineers should focus on
the guts of the new whiz-bang gadget,
not the power supply to run it.
The LTM8020, LTM8022 and
LTM8023 are three µModule regulators that require minimal design effort
and only a few inexpensive passive
components to make a complete
power supply. The modules are small,
accept a wide input operating range
and can produce 0.2a, 1a and 2a,
respectively.
VIN
4.5V TO 36V
1µF
VIN
VOUT
LTM8020
SHDN
Figure 1. Generate 3.3V at 200mA with
the LTM8020, two caps and a resistor
The LTM8020, LTM8022 and
LTM8023 are three µModule
regulators that require
minimal design effort and
only a few inexpensive
passive components to make
a complete power supply.
OUT
RUN/SS
AUX
SHARE
BIAS
2.2µF
ADJ
22µF
VOUT
3.3V AT
1A
GND SYNC ADJ
49.9k
154k
Figure 3. Produce 3.3V at 1A with LTM8022 and just four passive components
LTM8022
VIN
14V TO
36V
IN
OUT
RUN/SS
AUX
SHARE
BIAS
2.2µF
PGOOD
RT
49.9k
GND SYNC ADJ
53.6k
Figure 4. The LTM8022 can produce 8V, too
36
VIN
VOUT
LTM8020
SHDN
BIAS
ADJ
GND
2.2µF
–5V
85mA
–5V
165k
1%
10µF
X5R
Figure 2. A simple reconfiguration of the
µModule generates a negative output
Tiny, Self-Contained,
200mA Power Supply
The LTM8020 is small, with a package
measuring only 6.25mm × 6.25mm
× 2.32mm, but it accepts a wide 4V
to 36V input voltage range, and can
produce up to 1W for output voltages
between 1.25V and 5V at 200ma. at
light loads, Burst Mode operation
keeps quiescent current to 50µa at no
load. The current draw is less than 1µa
when shut down. as seen in Figure 1,
a complete LTM8020 power supply requires only an input capacitor, output
capacitor and a single resistor to set
the output voltage.
Negative Power Supply
with Few Components
PGOOD
RT
10µF
X5R
301k
1%
IN
VIN
5V TO 30V
BIAS
GND
LTM8022
VIN
5.5V TO
36V
VOUT
3.3V
200mA
10µF
VOUT
8V AT
1A
Being a self-contained design, the
LTM8020 can be easily conigured to
generate a negative voltage. Figure 2
shows is an example of how to use the
LTM8020 to generate –5V at 85ma
from an input range of 4.5V to 30V.
The part does not operate as a true
buck converter in this coniguration,
so the maximum output current is
less than that achievable in the buck
coniguration.
If You Need More Power…
The LTM8022 comes in a larger
11.25mm × 9mm × 2.82mm package than the LTM8020, but boasts
a wider input range, 3.6V–36V, and
output range, 0.8V–10V, for loads up
to 1a. It also includes more control
features, including a run/SS pin,
Linear Technology Magazine • January 2008
DESIGN IDEAS L
synchronization, user adjustable
switching frequency and a SHarE pin
for paralleling modules. The LTM8022
also employs Burst Mode operation,
drawing only 50µa quiescent current
at no load while maintaining only
30mV of output voltage ripple. Like the
LTM8020, the quiescent current when
shut down is less than 1µa. The schematic is very simple, with examples of
3.3V and 8V output designs shown in
Figures 3 and 4, respectively.
VIN
5.5V TO 36V
VIN
VOUT
RUN/SS
BIAS
AUX
VOUT
3.3V
2A
LTM8023
2.2µF
22µF
SHARE
ADJ
RT
49.9k
GND SYNC
154k
Figure 5. The LTM8023 produces 3.3V at 2A with the same footprint
and components required for the LTM8022 producing 1A.
…Or, Even More Power…
The LTM8023 is the big brother of the
LTM8022, capable of producing up to
2a of output current. The LTM8023
has the same input, output voltage
range, and control features as the
LTM8022. It also features Burst Mode
operation and low quiescent current.
The LTM8022 and LTM8023 share
the same footprint and pin pattern,
so even if you start a design with
the LTM8022 but later ind that you
need more current, you can simply
drop in the LTM8023. In most cases,
the design will use identical passive
components as the LTM8022, as seen
in the 3.3V example in Figure 5.
Conclusion
Table 1. Summary of LTM8000 series µModule regulators
Part Number
VIN Range
Max Load
VOUT Range
Size
LTM8020EV
4V to 36V
200mA
1.25V to 5V
6.25 × 6.25 × 2.32mm
LTM8022EV
3.6V to 36V
1A
0.8V to 10V
11.25 × 9 × 2.82mm
LMT8023EV
3.6V to 36V
2A
0.8V to 10V
11.25 × 9 × 2.82mm
The LTM8020, LTM8022 and LTM8023
µModule regulators make power supply development fast and easy. Their
broad input and output voltage ranges,
load capabilities and small size (see
Table 1) make them readily it into a
wide variety of applications. L
LTC562, continued from page 5
Mode operation to maximize power
eficiency at light loads. under noload conditions, the regulators can
also be programmed into LDO mode,
which provides the lowest quiescent
current (all four regulators in LDO
mode only draw a combined 80µa for
the entire chip).
To save even more power, the
LTC3562 can be programmed to reduce the regulators’ output voltages
in Burst Mode operation or forced
Burst Mode operation during light load
conditions. Since power dissipation
is directly proportional to the supply
voltage multiplied by the load current,
dropping the supply voltage effectively
reduces the circuit’s total power dissipation. If the output load is resistive
in nature, reducing the supply voltages
has an even greater effect, since power
dissipation in the load is proportional
to the supply voltage squared.
Conclusion
The LTC3562 is a highly lexible I2C
quad step-down converter composed
of two 600ma and two 400ma buck
regulators in a 3mm × 3mm QFn
package. The output voltages of the
regulators can be switched on the ly
using servo control or I2C control. Each
regulator can also be switched on the
ly into four possible high eficiency or
low-noise operating modes. This is a
perfect device for high performance
applications that require constant
control of the power supply. It can also
be used to simplify design, build and
test cycles, since output voltages can
easily be changed without changing
components. L
LTC81 and LTC814-5, continued from page 21
the output is drawing full load. Its
eficiency is shown in Figure 7.
Conclusion
The LTC3813 and LTC3814-5’s
synchronous architecture and high
voltage capability make them ideally
suited for high voltage high power
boost converters. They decrease comLinear Technology Magazine • January 2008
plexity by eliminating the requirement
for a large diode package and heat
sink to dissipate its high power loss.
Programmable frequency and current
limit, wide output voltage range, and
ability to drive logic-level or higher
threshold MOSFETs provide maximum lexibility in using them for a
variety of boost applications. Other
features such as such as strong gate
drivers to minimize transition losses,
an accurate voltage reference, accurate
cycle-by-cycle current limit, and an
on-chip bias supply controller make
the LTC3813 and LTC3814-5 the obvious choice for high performance, high
power boost converters. L
37
L DESIGN IDEAS
Small, High Efficiency Solution
Drives Two Piezo Motors
Introduction
Piezoelectric motors are used in
digital cameras for autofocus, zooming and optical image stabilization.
They are relatively small, lightweight
and eficient, but they also require
a complicated driving scheme. Traditionally, this challenge has been
met with the use of separate circuits,
including a step-up converter and an
oversized generic full bridge drive IC.
The resulting high component count
and large board space are especially
problematic in the design of cameras
for ever shrinking cell phones. The
LT3572 solves these problems by
combining a step-up regulator and
a dual full bridge driver in a 4mm ×
4mm QFn package.
A Simple Integrated Solution
to Drive Two Piezo Motors
Figure 1 shows a typical LT3572 Piezo
motor drive circuit. a step-up converter
with a high eficiency internal switch is
used to generate 30V from a low voltage
power source such as a Li-Ion battery
or any input power source within the
part’s wide input voltage range of
2.7V to 10V . The LT3572 uses a peak
current mode control architecture,
which improves line and load transient
response compared to other schemes.
The switching frequency is adjustable
from 500kHz to 2.5MHz, set either by
an external resistor or synchronized
to an external clock source of up to
2.5MHz. This allows selection of the optimum frequency for any given design.
The soft-start feature limits the inrush
current drawn from the supply upon
start-up. a PGOOD pin indicates when
the output of the step-up converter is
in regulation and the Piezo drivers can
start switching. The step-up converter
and both Piezo drivers have their own
shutdown control.
The high output voltage of the stepup converter, adjustable up to 40V, is
available for the drivers at the OuT
pin. The LT3572 is capable of inde38
Wei Gu
10µH CMDSH05-4
VIN
3V TO 5V
100k
4.7µF
42.2k
VOUT
30V
50mA
15pF
VIN
SW
SHDN
SHDNA
SHDNB
PWMA
PWMB LT3572
SYNC
PGOOD
VOUT
RT
OUTB
SS
GND
576k
10µF
FB
OUTA
24.3k
OUTA
OUTB
10nF
Figure 1. A typical LT3572 Piezo motor drive circuit
The LT3572 uses a peak
current mode control
architecture, which improves
line and load transient
response compared to other
schemes. The switching
frequency is adjustable from
500kHz to 2.5MHz, set either
by an external resistor or
synchronized to an external
clock source of up to 2.5MHz.
This allows selection of the
optimum frequency for
any given design.
pendently driving two Piezo motors
with two input PWM signals. The
motors respond accordingly based
on the duty cycle and the frequency
of the PWM signals. The drivers operate in an H-bridge fashion, where the
OuTa and OuTB pins are the same
polarity as the PWMa and PWMB pins
respectively and the OuTa and OuTB
pins are inverted from PWMa and
PWMB respectively. Each H-bridge can
drive a 2.2nF capacitor with rise and
fall times less than 100ns. Figure 2
shows a typical layout. The LT3572
is available in a small 4mm × 4mm
QFn package.
Conclusion
The LT3572 is a complete Piezo motor drive solution with a built-in high
eficiency 40V, 1.2a internal switch
and integrated dual 500ma full bridge
drivers. It includes other features to
minimize the application footprint,
including ixed frequency, soft-start,
and internal compensation. L
Figure 2. Typical layout for
the Figure 1 converter
Linear Technology Magazine • January 2008
NEW DEVICE CAMEOS L
New Device Cameos
I2C ADC Guarantees 16-Bit
Performance in 3mm × 2mm
Package
The LTC2453 is a 16-bit I2C-compatible delta sigma analog-to-digital
converter (aDC) in an ultra-tiny 3mm
× 2mm DFn package. Its tiny size, low
power and guaranteed 16-bit resolution improves performance of portable
instruments and sensors. Operating
from a single 2.7V to 5.5V supply,
the LTC2453 is capable of measuring
a differential input up to ±VCC. This
wide input range is ideal for measuring a wide variety of single-ended or
differential sensors.
The versatile LTC2453 achieves excellent 16-bit DC performance of 2LSB
integral nonlinearity error, 1.4µVrMS
transition noise and 0.01% gain error.
The LTC2453 has an internal oscillator
and allows up to 60 conversions per
second, making it easy to measure
temperature, pressure, voltage or
other low frequency sensor outputs.
The LTC2453 draws 800µa of supply
current at the 60Hz maximum sample
rate. after each conversion, supply
current is reduced to less than 0.2µa,
further preserving battery power. If
the user samples the device once a
second, the LTC2453 dissipates only
40µW from a 3V supply.
The LTC2453 communicates
via a simple I2C-compatible 2-wire
interface, reducing the number of
I/O lines required to read data,
making the LTC2453 ideal for tiny,
space-constrained applications. The
LTC2453 includes continuous internal offset and full-scale calibration of
the input signal, ensuring accuracy
over time and over the full operating temperature range. Linear’s no
Latency Delta-Sigma™ design allows
the aDC to multiplex several inputs
with no delay in reading the output
data. The LTC2453 incorporates a
proprietary sampling network that
reduces the dynamic input current to
less than 50na, making a wide range
of external input protection and ilter
circuits possible.
Linear Technology Magazine • January 2008
SoftSpan 16-/14-/12-Bit
More Choices in Very
Current Output DACs Draw
High Speed ADC Drivers
Less than 1µA Supply Current The LTC6400/LTC6401 is a family
The LTC2751 is a family of extremely
low power, software-programmable
16-/14-/12-bit digital-to-analog converters (DaCs). These current output
DaCs typically draw only 0.7µa of
supply current (2µa max), while generating an output swing up to ±10V. Six
unique output voltage ranges can be
programmed via SoftSpan™ software,
including two unipolar ranges (0V to
5V, 0V to 10V) and four bipolar ranges
(±10V, ±5V, ±2.5V, -2.5V to +7.5V).
Software programmability eliminates
the need for expensive precision resistors, gain stages and manual jumper
switching.
The LTC2751-16 offers accurate DC
speciications, including ±1LSB(max)
InL and DnL over the –40°C to 85°C
industrial temperature range. With its
precision linearity and supply current
less than 1µa, the LTC2751-16 can
be used in DC precision positioning
systems, high-resolution gain and
offset adjustment applications, and
portable instrumentation.
The LTC2751-16 also offers excellent aC speciications, including
full-scale settling time of only 2µs
and low 2nV•s glitch impulse, which
is key for aC applications such as
waveform generation. Low glitch reduces the transient voltages between
code changes in the DaC. Fast settling
and low glitch reduce harmonic distortion, making it possible to produce
higher frequency, lower noise output
waveforms.
The LTC2751 DaCs use a bidirectional input/output parallel interface
that allows readback of any internal
register, as well as the DaC output
span setting. a power-on reset circuit
returns the DaC output to 0V when
power is irst applied and a CLr pin
asynchronously clears the DaC to 0V
in any output range.
The LTC2751 DaCs are available in
pin-compatible 16-bit, 14-bit, and 12bit QFn-38 (5mm × 7mm) packages.
of very high speed differential ampliiers, suitable for driving signals of
up to 300MHz into high performance
pipeline aDCs. Versions of these parts
with gains from 8dB to 26dB are now
available. The “dash” number behind
the part name signiies the voltage gain
in dB. For example, the LTC6400-26
has a voltage gain of 26dB (or 20V/V).
The LTC6401-8 has a voltage gain of
8dB (or 2.5V/V). The LTC6400-20 and
LTC6401-20 (voltage gain of 20dB)
were described in greater detail in an
earlier Design Feature. The difference
between the LTC6400 and LTC6401
part numbers is that the LTC6400
consumes more DC power but has
lower distortion especially at signal frequencies above 140MHz. The LTC6401
consumes less DC power and is recommended for low distortion applications
with signal frequencies up to 140MHz.
Both versions have the same low noise
performance. Inside each IC is a differential op amp with input-referred
noise density of 1nV/√Hz. The gain
is set internally by means of on-chip
resistors. The lower gain versions have
lower output noise (because the op
amp noise is multiplied by less gain)
but the higher gain versions have a
higher gain-bandwidth product (because the bandwidth remains the same
even though the gain is higher).
Typical applications that beneit
from these parts are IF-sampling communications receivers where high
linearity is needed to avoid ‘blockers’
from intermodulating into nearby
bands. For example, at 140MHz the
intermodulation distortion of a 2VP–P
signal is as low as –93dBc. Previously,
the only other way to achieve such
performance was through very power
hungry rF gain blocks with OIP3s of
>50dBm. The LTC6400 saves power,
space and BOM cost compared to
older solutions.
all members of the family are pincompatible and come in a 3mm × 3mm
39
L NEW DEVICE CAMEOS
QFn package. The parts operate from a
3V or 3.3V supply voltage and over the
–40°C to 85°C temperature range.
Tiny Dual Input Li-Ion
Charger Integrates USB
and Wall Adapter Paths
suited for portable applications requiring two different charging input
sources; particularly, if one of those
sources is a uSB port.
2.7GHz, 60dB
Mean-Squared Power
The LTC4097 is a full-featured Li- Detector Responds in 500ns
Ion/Polymer battery charger capable
of charging from either a uSB port or
a wall adapter without the need for
an external multiplexer. Packaged in
a tiny 3mm × 2mm DFn, the LTC4097
includes independently programmable
charge current for both inputs, programmable termination current, an
nTC battery temperature qualiication input, automatic recharge and
more. The LTC4097 is the smallest
IC in a growing line of dual input LiIon chargers including the LTC4075,
LTC4075HVX, LTC4076, LTC4077,
and LTC4096.
Many portable applications—including digital cameras, PDas, mobile
phones, and personal media players—can be charged from a uSB port
while exchanging data with a host
computer, along with the option of
faster charging via a 5V wall adapter.
In such a 2-input system, a singleinput charger requires an external
multiplexer if a different charge current is needed for each type of input.
On the other hand, the LTC4097 Li-ion
charger accomplishes this task with
complete integration, thus avoiding the
cost and board-space requirements of
a multiplexer and related components.
In addition to independent charge
current programming for each input,
the LTC4097 includes a convenient
digital input (HPWr) to switch between
low power (100ma) and high power
(500ma) modes while powered from
a uSB port.
The LTC4097 packs these features
into a tiny package without sacriicing performance. Charge current is
regulated to an accurate 6% and inal
loat voltage is held to a tight ±0.5%.
Furthermore, the termination current
is accurate to within a handful of
milliamps of the programmed value.
This unique combination of small
size, full feature set, and high performance make the LTC4097 ideally
40
a new wide dynamic range meansquared rF detector from Linear
Technology sets a new level of accuracy
and speed performance. The LT5570
provides accurate rMS (root-MeanSquared) power measurement of a
40MHz to 2.7GHz aC signal over 60dB
dynamic range, even with a modulation
crest-factor of up to 12dB. It offers
best-in-class measurement accuracy
of ±0.5dB over its full dynamic range
and over a temperature range of –40°C
to 85°C. Moreover, the device allows
exceptionally fast response with a
full-scale rise time of 500ns.
as nascent next-generation wireless
standards such as mobile WiMaX and
LTE (Long-Term Evolution) adopt
more complex modulation schemes,
combining OFDM (Orthogonal Frequency Division Multiplexing) and
QaM (Quadrature amplitude Modulation) to boost the data rate, it becomes
increasingly dificult to accurately
measure these high crest-factor signals. This problem is not just conined
to wireless infrastructure, as many
other wireless systems are similarly
constrained by limited spectrum bandwidth. as a result, there is an ongoing
need for higher order modulation to
increase data rates. Cable networks,
microwave datalinks, satellite communications, and military radios
have similar needs, and the LT5570
is designed to meet these emerging
challenges.
The LT5570 provides a DC output
proportional to the rMS value of the
input signal power. Even if the input
waveform has high crest-factor content, such as a 4-carrier W-CDMa
modulated waveform, its rMS conformance accuracy is typically within
0.2dB, compared to that of a CW
(continuous waveform) power. The
device offers 61dB dynamic range at
880MHz, and 51dB at 2.14GHz. Its
linear DC output is proportional to the
input power in dBm with a scaling factor of 36.5mV/dB, typical. Minimum
sensitivity is –53dBm at 880MHz, and
–43dBm at 2.14GHz. The device offers
exceptional linearity, deviating less
than ±0.5 dB from the ideal log-linear
straight line, and over the device’s
operating temperature extremes.
The LT5570 operates from a single
5V supply, drawing a quiescent supply current of 26.5ma. a shutdown
feature is provided, reducing supply
current to 0.1µa.
The device comes in a 10-lead
3mm × 3mm DFn surface mount
package.
Single/Dual/Quad/Octal
Precision Voltage Monitors
Guaranteed to 125°C
a family of single, dual, quad, and octal
voltage monitors are now guaranteed
to operate across –40°C to 125°C. The
LTC2910, LTC2912, LTC2913 and
LTC2914 all feature a threshold accuracy of ±1.5% over the automotive
temperature range, allowing them to
accurately monitor single-channel
point-of-load or multichannel applications. These voltage monitors are
all offered in tiny leaded and leadless
packages and draw very little quiescent
current. Set via external resistors, the
entire family includes power supply
glitch iltering that ensures predictable
reset operation without false triggering. Each monitor also includes an
adjustable reset timer and reset output
that signals an undervoltage (uV) or
overvoltage (OV) condition.
The LTC2910 monitors eight low
voltage adjustable uV inputs and the
LTC2914 monitors four adjustable
inputs for OV, uV or negative voltages. Both the LTC2910 and LTC2914
draw just 70µa and are available in
16-lead SSOP and 5mm × 3mm DFn
packages. The LTC2913 monitors
two input channels for OV and uV
conditions, draws only 60µa, and is
offered in 10-lead MSOP and 3mm ×
3mm DFn packages. The LTC2912
monitors a single supply for OV
and uV conditions, with only 40µa
of supply current, and is offered in
8-lead TSOT and 3mm × 2mm DFn
packages.
Linear Technology Magazine • January 2008
NEW DEVICE CAMEOS L
The LTC2910, LTC2912, LTC2913,
and LTC2914 automotive grade voltage
monitors are all available today.
Precision Dual/Quad CMOS
Rail-to-Rail Input/Output
Amplifiers
The LTC6081 and LTC6082 are dual
and quad low offset, low drift, low
noise CMOS operational ampliiers
with rail-to-rail input and output
swings. The 70µV maximum offset,
1pa input bias current, 120dB open
loop gain and 1.3µVP–P 0.1Hz to 10Hz
noise make it perfect for precision
signal conditioning.
The LTC6081 and LTC6082 features
100dB CMrr and 98dB PSrr. Each
ampliier consumes only 330µa of
current on a 3V supply. The 10-lead
DFn has an independent shutdown
function that reduces each ampliier’s
supply current to 1µa. The LTC6081
and LTC6082 are speciied for power
supply voltages of 3V and 5V from
–40°C to 125°C. The dual LTC6081 is
available in 8-lead MSOP and 10-lead
DFn10 packages. The quad LTC6082
is available in 16-lead SSOP and DFn
packages.
0V to 44V Input Range
Precision Current Sense
Amplifier
The LTC6105 is a micropower, precision, current sense ampliier. The
LT6105 monitors unidirectional current via the voltage across an external
sense resistor. any gain between 1V/V
to 100V/V can be conigured with
external resistors. a minimum slew
rate of 2V/µs ensures fast response
to unexpected current changes.
The LT6105 sense inputs have a
voltage range that extends from –0.3V
to 44V and can withstand a differential
voltage of the full supply. This makes
it possible to monitor the voltage
across a MOSFET switch or a fuse of
a nearly depleted battery. The device
can also withstand a reverse-battery
condition on the inputs. CMrr and
PSrr are in excess of 100dB coupled
with low 300µV input offset voltage,
and maximum sense voltage of 1V will
allow a wide dynamic range of current
to be monitored.
Linear Technology Magazine • January 2008
The LT6105 has an independent
power supply, which operates from
2.85V to 36V and draws only 150µa.
When V+ is powered down, the sense
pins are biased off. This prevents
loading of the monitored circuit, irrespective of the sense voltage. The
LT6105 is available in a 6-lead DFn
and 8-lead MSOP packages.
High Power Step-Down
DC/DC Controller Draws Only
30µA in Automotive Systems
The LTC3834/-1 synchronous stepdown DC/DC controller features
ultralow quiescent current. Drawing only 30µa in sleep mode, the
LTC3834/-1 is ideal for preserving
battery energy in “always-on” automotive systems or battery-powered
devices where the system remains
semi-active, or when a car’s engine
is off. When in shutdown mode, the
LTC3834/-1 draws a mere 4µa.
This controller is the latest addition to Linear Technology’s lineup
of over twenty ultralow quiescent
current DC/DC switching regulator
controllers for step-down, step-up,
buck-boost, SEPIC and inverter topologies.
The input supply range of the
LTC3834/-1 at 4V to 36V is wide
enough to protect against high input
voltage transients and it continues to
operate during automotive cold crank.
It can provide an output voltage from
0.8V up to 10V, making it ideal for
the higher voltage supplies typically
required for audio systems, satellite
receivers, analog tuners and CD/DVD
players.
This controller has an onboard LDO
for bias power and a powerful onboard
MOSFET driver to deliver up to 20a
load current at eficiencies as high
as 95%. The LTC3834/-1’s constant
frequency, current mode architecture
provides excellent line and load regulation. The device features a very low
dropout voltage, with up to 99% duty
cycle and smoothly ramps the output
voltage during start-up with its adjustable soft-start and tracking features.
The operating frequency is adjustable
from 250kHz to 530kHz, and can be
synchronized to an external clock from
140kHz to 650kHz using its phasedlocked loop (PLL). In addition, the user
can select from continuous, pulse
skipping or Burst Mode operation at
light loads. Output overvoltage and
overcurrent (short circuit) protection
are integrated and the LTC3834/-1
features ±1% reference voltage accuracy over an operating temperature
range of –40°C to 85°C.
The LTC3834/-1 is available in two
versions. The LTC3834 version has a
power-good output voltage monitor
and an EXTVCC input that allows the
IC to be powered from its output for
maximum eficiency. It is also features
PolyPhase® operation that enables
multiple ICs to be synchronized outof-phase to minimize the required
input and output capacitances. The
LTC3834 is offered in a 20-lead TSSOP
and 4mm × 5mm QFn packages,
whereas the LTC3834-1 is housed in
the smaller 16-pin SSOP and 5mm ×
3mm DFn packages.
100V High Speed
Synchronous N-Channel
3A MOSFET Driver for
High Efficiency Step-Down or
Step-Up DC/DC Converters
The LTC4444 is a high speed, high input supply voltage (100V) synchronous
MOSFET driver designed to drive upper
and lower power n-Channel MOSFETs
in synchronous rectiied converter
topologies. This driver, combined with
power MOSFETs and one of Linear
Technology’s many DC/DC controllers, form a complete high eficiency
synchronous converter.
This powerful driver can source up
to 2.5a with a 1.2Ω pull-down impedance for driving the top MOSFET and
source 3a with a 0.55Ω pull-down
impedance for the bottom MOSFET,
making it ideal for driving high gate
capacitance, high current MOSFETs.
The LTC4444 can also drive multiple
MOSFETs in parallel for higher current applications. The fast 8ns rise
time, 5ns fall time of the top MOSFET,
and 6ns rise time, 3ns fall time of
the bottom MOSFET when driving
a 1000pF load minimize switching
losses. adaptive shoot-through protection is integrated to minimize dead
41
L NEW DEVICE CAMEOS
time while preventing both the upper
and lower MOSFETs from conducting
simultaneously.
The LTC4444 is conigured for two
supply-independent inputs. The high
side input logic signal is internally
level-shifted to the bootstrap supply,
which may function at up to 114V
above ground. Furthermore, this part
drives both upper and lower MOSFET
gates over a range of 7.2V to 13.5V.
The LTC4444 is offered in a thermally enhanced MSOP-8 package.
3.3V 20Mbps 15kV
RS485/RS422 Transceivers
The LTC2850, LTC2851 and LTC2852,
are the latest additions to Linear
Technology’s family of rugged 3.3V
rS485/rS422 transceivers. These
devices offer a variety of advanced
features for industrial, medical and
automotive applications with high
speed operation to 20Mbps.
High receiver input resistance supports up to 256 nodes on a single bus,
while meeting rS485 load requirements. Failsafe operation guarantees
a logic-high receiver output state when
the inputs of the receiver are loating,
shorted or terminated, but not driven.
Current limiting on all driver outputs
and a thermal overload shutdown
feature provide protection from bus
contention and short circuits. Bus
pin protection on all parts exceeds ±
15kV for ESD strikes with no latchup
or damage.
The LTC2850 provides half-duplex operation and the LTC2851 and
LTC2852 are full-duplex. They are
pin-compatible with the 5V LTC485,
LTC490 and LTC491 parts, respectively. Speciied over commercial and
industrial temperature ranges from
–40C to 85C, these parts are available
in SO and MSOP packages as well as
tiny leadless DFn packages.
New Member Added to
the LTC2908 6-Supply
Monitor Family
The LTC2908-C1 is a new addition
to the LTC2908 6-supply monitor
family available in tiny 8-pin TSOT
and DFn packages. The LTC2908-C1,
along with the previously available a1
and B1 versions, provides complete,
precise, space-conscious, micropower
and general purpose voltage monitoring solution for any application. The
inputs can be shorted together for
monitoring systems with fewer than
six supply voltages, and the open drain
rST output of two or more LTC2908
can be wired-Or together for monitoring systems with more than six supply
voltages.
The LTC2908-C1 is designed to
monitor 2.5V and ive positive adjustable voltages. The previously
available LTC2908-a1 is designed to
monitor 5V, 3.3V, 2.5V, 1.8V and two
positive adjustable voltages while the
LTC2908-B1 is designed to monitor
3.3V, 2.5V, 1.8V, 1.5V and two positive
adjustable voltages. The LTC2908 features a low voltage positive adjustable
inputs (+aDJ) with nominal threshold
level at 0.5V, and a low quiescent current on the main supply (the greater
of V1 or V2) of 25µa typical.
The LTC2908 also features ultralow voltage pull downs on the rST pin.
The open drain rST output is guaranteed to be in the correct state as long
as V1 and/or V2 is 0.5V or greater.
The LTC2908 inputs have a tight 1.5%
threshold accuracy over the whole
operating temperature range (–40°C to
85°C) and glitch-immunity to ensure
reliable reset operation without false
triggering. The common rST output
remains low until all six inputs have
been above their respective thresholds
for 200ms. L
LTC4090, continued from page 1
crease as the battery charger switches
to constant-voltage mode. When the
charge current drops to 10% of the
full-scale charge current, commonly
referred to as the C/10 point, the
open-drain charge status pin, CHrG,
assumes a high impedance state. an
external capacitor on the TIMEr pin
sets the total minimum charge time.
In Figure 1, a 0.1µF capacitor on the
TIMEr pin gives a 2.145hr minimum
charge time. When this time elapses,
the charge cycle terminates and the
CHrG pin assumes a high impedance
state, if it has not already done so.
Charge Time is
Automatically Extended
The LTC4090 has a feature that automatically extends charge time if the
charge current in constant current
mode is reduced during the charging
42
cycle. reduction can be due to thermal
regulation or the need to maintain the
programmed input current limit. The
charge time is extended inversely proportional to the actual charge current
delivered to the battery. The decrease
in charge current as the LTC4090 approaches constant-voltage mode is due
to normal charging operation and does
not affect the timer duration.
Trickle Charge and
Defective Battery Detection
at the beginning of a charge cycle, if
the battery voltage is below 2.9V, the
charger goes into trickle charge reducing the charge current to 10% of the
full-scale current. If the low battery
voltage persists for one quarter of the
programmed total charge time, the
battery is assumed to be defective,
the charge cycle is terminated and
the CHrG pin output assumes a high
impedance state. If for any reason the
battery voltage rises above ~2.9V the
charge cycle is restarted.
Conclusion
The LTC4090 combines a high voltage
switching buck regulator, a full-featured Li-Ion battery charger, and a
PowerPath controller in a tiny 3mm
× 6mm DFn package. Its wide input
voltage range, high programmable
charge current, and small footprint
Want to know more?
visit:
www.linear.com
or call
1-800-4-LINEAR
Linear Technology Magazine • January 2008
DESIGN TOOLS L
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LTspice/SwitcherCAD™ III (www.linear.com/swcad)
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Linear Technology Magazine • January 2008
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Linear Technology Magazine • January 2008