90 W, Universal Input, Single Stage, PFC Converter

AND8124/D
90 W, Universal Input,
Single Stage, PFC
Converter
http://onsemi.com
General Description
The NCP1651 demo board uses a quad operational
amplifier on the secondary to perform multiple functions.
One section of the amplifier is used as the error amplifier. A
voltage divider comprised of R23, R24, R25 and R33 senses
the output voltage and divides it down to 2.5 V. This signal
is applied to the negative input of the error amplifier; the
2.5 V reference is applied to the non−inverting input of the
error amplifier.
The output of the error amplifier provides a current sink
that drives the LED of the optocoupler. The primary side
optocoupler circuit sinks current from pin 8. This varies the
voltage into the Voltage−to−Current converter that feeds the
reference multiplier.
The loop operation is as follows: If the output voltage is
less than its nominal value, the voltage at the output of the
voltage divider (inverting input to the error amplifier) will
be less than the reference signal at the non−inverting error
amplifier input. This will cause the output of the error
amplifier to increase. The increase in the output of the error
amplifier will cause the optocoupler LED to conduct less
current, which in turn will reduce the current in the
optocoupler photo−transistor. This will increase the voltage
at pin 8 of the chip, and in turn increase the output of the
reference multiplier, causing an increase in the NCP1651
duty cycle.
The current shaping network is comprised of the ac error
amplifier, buffer and current sense amplifier. This network
will force the average input current to maintain a scaled
replica of the current reference on pin 10. The increase of the
reference voltage will cause the current shaping network to
draw more input current, which translates into an increase in
output current as it passes through the transformer. The
increase in current will increase the output power and
therefore, the output voltage. To calculate the loop stability,
it is recommended that the On Semiconductor spread sheet
be used. This is an easy and convenient way to check the gain
and phase of the control loop.
This application note describes the implementation of a
90 W, universal input Flyback Power−Factor−Correction
(PFC) converter using On Semiconductor’s NCP1651
controller.
The NCP1651 enables a low cost single−stage (with a low
voltage isolated output) PFC converter as demonstrated in
this application circuit, which is designed for 48 Vdc, at
1.9 A of output current. The NCP1651 is designed to operate
in the fixed frequency, continuous mode (CCM), or
discontinuous (DCM) mode of operation, in a Flyback
converter topology. The converter described in this
application note has the following valuable features:
Features
• Wide Input Voltage Range (85 − 265 Vac)
• Galvanic Isolation
• Primary Side Cycle−by−Cycle and Average Current
Limit
• Secondary Side Power Limiting
• High Voltage Start−up Circuit
Detailed Circuit Description
Operational description and design equations are
contained in the NCP1651 Data Sheet. This application note
addresses specific design issues related to this converter
design. Please refer to Figure 2 for component reference
designators.
Voltage Regulation Loop
With a Flyback topology, the output is isolated from the
input by the power transformer. Output voltage regulation
can be accomplished in two ways. The first, and the simplest
method is by sensing the primary side voltage of the
auxiliary winding. This eliminates the feedback isolation
circuitry, at the expense of accuracy of voltage regulation
and current sensing. The second method is to sense the
secondary side voltage which is more complex, but provides
better voltage regulation and transient response.
 Semiconductor Components Industries, LLC, 2003
December, 2003 − Rev. 4
1
Publication Order Number:
AND8124/D
J1
2
Input
C26
1.2 F
R3
180 k
2
R11
1.2 k
NCP1651
16 Start−up
11 AC cmp
5
1st
littr 8
Lavg 7
Ct 3
Ramp 4
GND 2
R35
4.7 k
R8
680
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C3
1
Out
C16
2.2 F
C6
VCC
10 ACref
12
13
Vref
.68 F
9 ACin
8 FB/SD
U1
0
R1 10
1.5kE68CA
D16
1.5kE25CA
D12
R34B
56 k
180 k
R2
D4
1N4006
D3
1N4006
D7
1N4006
Q1
C25 1 nF
R34A
56 k
C11
.001 F
02
2N2222A
100 H
L3
100 H
6
3
4
T3
o
R5
.12
2
D6
MUR160
C13
.1 F
C12
.1 F
470 pF
R7 8.66 k
470 pF
R4 35 k
C10
1 nF
C8
.022 F
1
R9
3.6 k
C9
.022 F
Figure 1. Applications Circuit Schematic
.07
R25
2k
4 U3A
1
+
−
13
12
11
+
−
4
MC3303
U3D
14
R29 2.0 k
C18 .047 F
R22
C17
392 22 F
4
10 + U3C
8
9 −
MC3303
R26
11
3.3 k
11 MC3303
−
MC3303
11
C5
470 pF
4 U5B
5
7
+
6
2
3
C23
1500 F
BAS19LT1
D10
C20
1 F
C19
1 F
0
3
BAS19LT1 2
C4
1000 pF
4
1
D9
R21
2k
R27
7.5 k
R32
R30
300
C22
1500 F
0
D13
AZ23C18
U4
R20
2k
Output
R31 .07
C24
.01 F
R24
174
D11
BAS19LT1
R23
210
R33
40.2 k
D5
14 MUR1620CT
15
11
5
10
D9
BAS19LT1
7
C21
220 F
TP2
GND
R36
12 k
2
D2
1N4006
o
o
L2
J2
TP1
Shutdown
1
F1
C27
o
D1
1N4006
AND8124/D
C28 1 F
1 2
TL431
U2
R28
3.3 k
AND8124/D
Overshoot/Undershoot Circuit
Two sections of the quad amplifier are used as
comparators. One of these monitors the output for
overvoltage condition and the other for undervoltage
condition. The voltage divider requires four resistors (R33,
R23, R24, and R25) in order to make the various ratios
available for the two comparators as well as the error
amplifier.
The undervoltage comparator provides the drive for the
opto−coupler. Its output is normally in the saturated high
state, which allows the flow of current into the opto−coupler
to be determined by the error amplifier or overvoltage
comparator. If an undervoltage condition occurs, the output
of the UV comparator goes low, which reduces the drive
current to the opto−coupler LED. This causes the NCP1651
to go into a high duty cycle state, and will increase the flow
of current into the output until the output voltage is above the
UV limit.
The over−voltage comparator’s output is OR’ed with the
output of the error amplifier. During an overvoltage event
(e.g. a transient load dump), the output of this comparator
will go to ground, and cause the maximum current to flow
in the opto−coupler LED. This will pull pin 8 low and reduce
the duty cycle to zero until the output voltage is below the
OV limit. It should be noted that the purpose of the 680 resistor (R8) in series with the opto−coupler photo transistor,
is there to keep the voltage at pin 8 above the 0.5 V threshold
during such events. This keeps the control chip operational
and will allow immediate operation when the output voltage
is again in its normal operating range. Without this resistor,
the voltage on pin 8 would drop below 0.5 V, causing the
NCP1651 to enter a low power shutdown mode of operation.
in regulation the inverting input voltage is typically 2.5 V).
This causes the error amplifier signal to go low, sinking
more current through the LED in the opto−coupler. This in
turn drives more current in opto−coupler transistor collector,
pulling it low reducing the duty cycle, folding back the
output voltage.
Output Voltage Ripple
The output voltage ripple on the secondary of the
transformer has two components, the traditional high
frequency ripple associated with a flyback converter, and the
low frequency ripple associated with the line frequency
(50 Hz or 60 Hz). In this application our goal was to have the
output ripple 5% of the nominal output voltage, or 2.4 V
pk−pk.
The High Frequency Ripple can be Calculated by:
(eq. 5)
(toff 4T) (Ipk Iped )2))
(eq. 6)
irms ((3.85 10 )) (((13.382 13.38 10.27
10.272) 3) 3.85 10 4)
(13.38 10.27)2) 5.78
(eq. 7)
To meet the capacitors ripple current requirements and
lower the equivalent esr, two 1500 F capacitors were used
in parallel.
Vcap (5.78 3.85 3000 ) 0.00742
Where:
n
Ipk
Iped
CO
esr
T
(eq. 1)
(eq. 2)
(eq. 8)
=Transformer Turns Ratio (3.89)
=Peak Current Secondary (13.38)
=Pedestal Current Secondary (10.27)
=Output Capacitance (1500 each)
=Output Capacitor Equivalent Series Resistance
(0.03 Each)
=Switching Interval
Vesr Ipksec esr
The voltage to the input of the differential amplifier is:
(eq. 9)
Vesr 13.38 Apk 0.015 0.20 V
(eq. 10)
V 0.007422 0.22 0.200
(eq. 11)
The Low Frequency Portion of the Ripple:
The output voltage from the differential amplifier is:
VO 0.33 11 3.63 V
Vcap irms dt CO
irms (toff T) (((Ipk2 (Ipk Iped) Iped2) 3 ))
The fourth section of the amplifier is biased as a
differential amplifier. This section senses the DC output
current, and provides a signal that is diode OR’ed into the
feedback divider.
In the demo board the overload current limit was set to
125% of full load, or 2.375 A. Two resistors are used in
series (to limit their maximum power dissipation) to sense
the output current (R31 and R32). R29 and R30 set−the
current sense amplifier gain.
Where the gain of the amplifier is:
2.375 A 0.14 0.33 V
(eq. 4)
The RMS current at the peak of the sinewave
(phase angle 90°).
Current/Power Limit Circuit
G (R29R30) 1 3000300 1 11
V Vcap2 Vesr2
(eq. 3)
When the output load current increases, the output of the
current sense amplifier will also increase. When the
amplifiers output voltage, minus a diode drop (D11),
increases above the 2.5 V, it pulls up the feedback signal at
the inverting input of the error amplifier ( when the loop is
V Ipk t CO
(eq. 12)
IAVG PO VO
(eq. 13)
Ipk IAVG 0.637
(eq. 14)
Ipk PO VO 0.637
90 (48)(0.637) 2.95
(eq. 15)
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3
AND8124/D
If we divided the output ripple into 10° increments over one
cycle (180°) the sinusoidal ripple voltage with respect to
phase angle is:
V (PO 0.637 VO) sin()
CO 18 fline
th = One Cycle of the Line 16.67ms (60Hz)
Vmax = 48 V
Vmin = 36 V
Pout = 90 W
(eq. 16)
CO (2 90 16.67 ms) (482 362) 3000 F
In Figure 2, the low frequency output voltage ripple are
plotted with respect to phase angle.
(eq. 19)
It is a coincidence that the output capacitor calculated for
voltage ripple and hold−up time are the same value.
1.50
MOSFET Turn−off Snubber
RIPPLE (V)
1.00
The MOSFET in our design has a VDS rating of 800 V, the
peak voltage across the device at turn−off (including the
leakage inductance spike) is:
0.50
VpkTotal Vinmax 1.414 ((VO Vf)n) Vspike
0.00
(eq. 20)
−0.50
Where:
Vinmax
VO
n
Vspike
=265 Vrms
=the Output Voltage (48 V)
=the Transformer Turns Ratio (4)
=Voltage Spike Due to Transformer Leakage
Inductance
To provide a safe operating voltage for the MOSFET we
have selected Vspike to be 130 Vpeak, so when the MOSFET
turns off, the maximum Drain to Source voltage is:
−1.00
−1.50
0
45
90
DEGREES (°)
135
180
Figure 2. Calculated Output Ripple
265 1.414 48(4) 130 697 V
Figure 3. Measured Output Voltage Ripple
It can be seen from the calculations, and the scope
waveform that as long as a capacitor with a low esr is used,
that the output voltage ripple is dominated by the low
frequency (120 Hz) ripple.
E 1 le Ipk2
2
If the user would like to select CO for Hold−Up time
versus, voltage ripple:
E 1 C V2
2
(eq. 17)
Where:
C= Snubber Capacitor
V= the Voltage Across the MOSFET
Rearranging the equation:
CO 2 Pout th V max 2 V min 2
(eq. 22)
Where:
le = Leakage Inductance (9 H Measured)
Ipk = Peak Primary Current
A Second Relationship is:
Hold−Up time
Pout 1 CO V2 f
2
(eq. 21)
To minimize the effect of the leakage inductance spike,
the coupling between the primary and secondary of the
transformer needs to be as tight as possible. This can be
accomplished, if your transformer requires a primary with
multiple layers, by interleaving the primary and secondary
windings. In our 48 Vdc application the transformer primary
has 74 turns, and the secondary has 19 turns. The
manufacture of the transformer, TDK, wound one layer of
the primary with 45 turns, then the 19 turn secondary, and the
remaining 29 turns of the primary. The results were a
leakage inductance of approximately 9 H. If we compare
this to a transformer where the entire 74 turns were wound,
in two layers, then the 19 turn secondary, the leakage
inductance increased to 37 H.
The energy stored in the transformer leakage:
(eq. 18)
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4
(eq. 23)
AND8124/D
Combining Equations:
In Figure 4, the output voltage drops to 40 Vdc, and
recovers in less than 160 ms. In Figure 6 the input voltage
was increased to 230 Vac, and the load was switched from
10% to 100% load. The output voltage now drops only to 44
Vdc, and recovers in approximately 50 ms. The significant
improvement in transient response performance is attributed
to an increase in the DC gain and loop bandwidth at high
line. As the input ac line voltage increases the control loop
DC gain (Refer to www.onsemi.com for a copy of the excel
design spreadsheet for details) increases from 42 dB at
115 Vac to 62 dB at 230 Vac and the control loop bandwidth
increases from 2 Hz to 8 Hz. The result is that at high line,
there is an improvement in transient response, but because
there is less attenuation of the output 120 Hz ripple, it results
in an increase in the input Total Harmonic Distortion (THD).
The system designers will need to trade off their overall
system performance THD, Power Factor, and transient
response to optimize the control loop to meet their
requirements.
C Ipk2 le ((VO Vf)n Vpk Vspike)2
(eq. 24)
((VO Vf)n Vpk)2
Csnubber 3.82 9 H ((192 375 130)2
(eq. 25)
(192 375)2 790 pF
During the MOSFET turn−off, the capacitor C25 is charge
through the Diode D6. Prior to the next ton switching cycle
the capacitor C25 must be fully discharged, so Rsnubber is
selected to be:
Rsnubber ((VO Vf)n Vinmax 1.414 Vspike)
0.63 (Vspike * Csnubber)
(eq. 26)
((192 375 130)0.63(6.5 ) (130 * 790 pF) 28 k
(eq. 27)
The power in the snubber is:
P 1 C V2
2
(0.5)790 pF(1302) 100 kHz 0.68 W
(eq. 28)
After installing the snubber in the NCP1651 Demo Board,
and measuring the voltage spike, the snubber components
where adjusted for maximum performance, C25 was
increased to 1000 pF, and R34 was changed to 30 k. The
difference between the measured and calculated value can
be attributed to the PWB board layout, and other parasitic
components.
Evaluation Board Test Results
The results from the NCP1651 Demo Board show that
using a flyback topology for a PFC converter can provide a
low input Total Harmonic Distortion (THD), a high input
power factor, and excellent steady state output voltage
regulation.
The NCP1651 achieved a THD at 115 Vac input at full
load of 3.12% with a PF of 0.998. The input THD to 6.8%
THD at 230 Vac in, with a PF of 0.971.
The steady state output voltage regulation from 85 Vac to
230 Vac, and no load to full load is less than 0.02%, with an
output voltage ripple meeting our design goal of
2.4 Vpk−pk, measured 2.0 V pk−pk.
Figure 4.
Transient Response
Figures 4 through 7 show the output transient response for
the 90 W converter. The test conditions for each Figure are
listed below:
Table 1. Test Conditions
Vin
IO
Figure 4
115 Vac
0.19 – 1.92 A
Figure 5
115 Vac
1.92 – 0.19 A
Figure 6
230 Vac
0.19 – 1.92 A
Figure 7
230 Vac
1.92 – 0.19 A
Figure 5.
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AND8124/D
Power Dissipation Estimates
The NCP1651 Demo Board power dissipation (measured)
at 115 Vrms, full load, is (106.27 – 47.95 •1.92) = 14.21 W.
Following table provides the calculated and estimated
power loss spread among different power train components.
Components
Pd average
D1−D4
Input Rectifier
1.65 W
Q1
MOSFET
4.1 W
D5
Output rectifier
1.7 W
T3
Flyback transformer
3.5 W
(estimate)
R34
Snubber resistor
0.84 W
D12
Transient suppressor
2.0 W
miscellaneous
0.41 W
Figure 6.
Total
14.20 W
Demo Board Operating Instructions
Connect an Ac source, 85 − 265 Vac, 47 − 64 Hz to the
input terminals J1. Connect a load to the output terminals J2,
the PWB is market +, for the positive output, − for the return.
Turn on the ac source, and the NCP1651 will automatically
start, providing 48 Vdc to the load.
Shutdown Circuit
The shutdown circuit will inhibit the operation of the
power converter and put the NCP1651 into a low power
shutdown mode. To activate this circuit, apply 5 V to the red
test point, with the black jack being “ground”. Be aware that
the black jack is actually hot as it is connected to the output
of the input bridge rectifiers. An isolated 5 V supply should
be used.
If this circuit is not being used, it can be left open as there
is enough resistance built in to the circuit to keep the
transistor (Q2) in it’s off state.
Figure 7.
Table 2. Performance Data Regulation
Line/Load
No Load
45 W
90 W
85 Vrms
47.94
47.95
47.95
115 Vrm
47.94
47.95
47.95
230 Vrms
47.94
47.95
47.95
265 Vrms
47.94
47.94
47.95
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AND8124/D
Table 3. Harmonics & Distortion
115 Vac 90 W
230 Vac 90 W
V harmon
A harm. %
V harm
A harm%
2nd
0.143
0.156
0.08
0.2
3rd
0.203
1.94
0.25
4.74
5th
0.13
0.6
0.12
2.88
7th
0.08
0.28
0.07
0.22
9th
0.04
0.19
0.09
0.76
11th
0.08
0.29
0.08
0.27
13th
0.16
0.32
0.06
0.33
15th
0.28
0.41
0.14
0.68
17th
0.4
0.41
0.28
0.95
19th
0.05
0.29
0.12
0.3
PF
0.998
0.971
THD(A)
3.12
6.8
Ifund
0.918
0.468
Table 4. Efficiency
85 Vrms
115 Vrms
230 Vrms
265 Vrms
1.5
1.52
1.51
1.59
Pin
109.42
106.27
105.35
105.25
Vo
47.95
47.95
47.95
47.95
Io
1.92
1.92
1.92
1.92
Efficiency
0.841
0.866
0.874
0.875
Pin @ No Load
Table 5. Vendor Contact List
Vendor
U. S. Phone / Internet
ON Semiconductor
1−800−282−9855 www.onsemi.com/
TDK
1−847−803−6100 www.component.tdk.com/
Vishay
www.vishay.com/
Bussman (Cooper Ind.)
1−888−414−2645 www.cooperet.com/
Coiltronics (Cooper Ind.)
1−888−414−2645 www.cooperet.com/
Fairchild
www.fairchildsemi.com/
Panasonic
www.eddieray.com/panasonic/
Weidmuller
www.weidmuller.com/
Keystone
1−800−221−5510 www.keyelco.com/
HH Smith
1−888−847−6484 www.hhsmith.com/
Aavid Thermalloy
www.aavid.com/
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AND8124/D
Table 6. NCP1651 Application Circuit Parts List (Specifications:, 90 W, 85 vac to 265 vac Input Range, 48 V Output)
Ref Des
Description
Part Number
Manufacturer
C1
Cap, Ceramic, Chip, 1000 pF, 50 V
VJ0603Y102KXAAT
VISHAY
C3
Cap, Ceramic, Chip, 470 pF, 50 V
VJ0603Y471JXAAT
VISHAY
C5
Cap, Ceramic, Chip, 470 pF, 50 V
VJ0603Y471JXAAT
VISHAY
C6
Cap, Ceramic, Chip, 470 pF, 50 V
VJ0603Y471JXAAT
VISHAY
C8
Cap, Ceramic, Chip, .022 F, 50 V
VJ0603Y223KXXAT
VISHAY
C9
Cap, Ceramic, Chip, 0.022 F, 50 V
VJ0603Y223KXXAT
VISHAY
C10, C11
Cap, Ceramic, chip, 0.001 F, 50 V
VJ0603Y102KXAAT
VISHAY
C12, C13
Cap, Ceramic, Chip, 0.1 F, 50 V
VJ0606Y104KXXAT
VISHAY
C16
2.2 F, alum elect, 450 V (0.394dia x 0.492H)
(.394dia x .492H)
ECA−2WHG2R2
EKA00DC122P00
Panasonic (Digi – P5873)
Vishay Sprague (20)
C17
Cap, Ceramic, Chip, 22 F, 10 V
C3225X5R0J226MT
TDK
C18
Cap, Ceramic, Chip, .047 F, 50 V
VJ0603Y473KXXAT
VISHAY
C19
Cap, Ceramic, Chip, .01 F, 50 V
VJ0603Y103KXAAT
VJ0603Y103KXAAT
C20
Cap, Ceramic, Chip, 1 F, 25 V
C3216X7R1E105KT
TDK
C21
220 F, alum elect, 25 V
ECA1EM331
Panasonic
C22, 23
1800 F, alum elect, 63 V (2.2A rms min)
1500 F, alum elect, 63 V
EEU−FC1J182
EKB00JL415J00
Panasonic (Digi – P11283)
Vishay Sprague (20)
C24
Cap, Ceramic, Chip, .01 F, 50 V
VJ0603Y103KXAAT
VISHAY
C25
Cap,Ceramic, .001 F, 1 KV
ECK−03A102KBP
Panasonic
C26
1.2 F, 275 vac, X cap
F1778−512K2KCT0
VISHAY
C27
Cap, polypropylene, .68 uF, 400 VDC
MKP1841−468−405
Vishey − Sprague
C28
Cap, Ceramic, Chip, 1 F, 25 V
VJ1206V105ZXXAT
VISHAY
D1 – D4
Diode, Rectifier, 800 V, 1 A
1N4006
ON Semiconductor
D5
Diode, Ultrafast, 200 V, 16 A
MUR1620CT
ON Semiconductor
D6
Diode, Ultrafast, 600 V, 1 A
MUR160
ON Semiconductor
D7
Diode, Rectifier, 800 V, 1 A
1N4006
ON Semiconductor
D8 – D11
Diode, Switching, 120 V, 200 mA, SOT−23
BAS19LT1
ON Semiconductor
D12
TVS, 214 V, 5 W
1.5KE250A
ON Semiconductor
D13
Zener Diode, 18 V
AZ23C18
VISHAY
D16
Zener Diode, 68 V
1.5kE68CA
ON Semiconductor
F1
Fuse, 2 A, 250 Vac
1025TD2A
Bussman
L2
2.5 A sat, 100 H inductor, diff mode
TSL1315−101K2R5
TDK
L3
2.5 A sat, 100 H inductor, diff mode
TSL1315−101K2R5
TDK
Q1
FET, 11 a, 800 V, .45 , N−channel
SPA11N80C3
Infineon
Q2
Bipolar, npn, 30 V, SOT−23
MMBT2222ALT1
ON Semiconductor
R1
Resistor, SMT1206, 10
CRCW1206100JRE4
Vishey
R2
Resistor, Axial Lead, 180k, ¼ W
CMF−55−180K00FKRE
Vishey
R3
Resistor, Axial Lead, 180k, ¼ W
CMF−55−180K00FKRE
Vishey
R4
Resistor, SMT1206, 35k
CRCW120635KOJNTA
Vishey
R5
Resistor, SMT, 0.12 , 1 W
WSL2512 .12 1%
Vishey Dale
R7
Resistor, SMT1206, 8.66 k
CRCW12068661F
Vishey
R8
Resistor, SMT1206, 680
CRCW12066800F
Vishey
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AND8124/D
Table 6. NCP1651 Application Circuit Parts List (Specifications:, 90 W, 85 vac to 265 vac Input Range, 48 V Output)
Ref Des
Description
Part Number
Manufacturer
R9
Resistor, axial lead, 3.6k, ¼ W
CMF−55−3K600FKBF
Vishey
R11
Resistor, SMT1206, 1.2k
CRC12061K20JNTA
Vishey
R20
Resistor, SMT1206, 2.0k
CRC12062K00JNTA
Vishey
R21
Resistor, SMT1206, 2.0k
CRC12062K00JNTA
Vishey
R22
Resistor, SMT1206, 392
CRC12052K10JNTA
Vishey
R23
Resistor, SMT1206, 210, 1%
CRCW12062100F
Vishey
R24
Resistor, SMT1206, 174, 1%
CRCW12061740F
Vishey
R25
Resistor, SMT1206, 2.05k, 1%
CRCW12062051F
Vishey
R26
Resistor, SMT1206, 3.3k
CRC12063K30JNTA
Vishey
R27
Resistor, SMT1206, 7.5k
CRC12067K50JNTA
Vishey
R28
Resistor, SMT1206, 3.3k
CRC12063K30JNTA
Vishey
R29
Resistor, SMT1206, 3.01k, 1%
CRCW12063011F
Vishey
R30
Resistor, SMT1206, 301, 1%
CRCW12063010F
Vishey
R31
1w, .07 resistor
WSL251R0700FTB
Vishey
R32
1w, .07 resistor
WSL251R0700FTB
Vishey
R33
Resistor, SMT1206, 40.2k, 1%
CRCW120640022F
Vishey
R34
Resistor, axial lead, 20k, 2W
R35
Resistor, SMT1206, 4.7k
CRCW12064K70NTA
Vishey
R36
Resistor, SMT1206, 12k
CRCW120612K0JNTA
Vishey
R37
Resistor, SMT1206, 100k
CRCR1206100K0JNTA
Vishey
T1
Transformer, Flyback (Lp 1 mH)
SRW42EC−U04H14
TDK
U1
PFC Controller
NCP1651
ON Semiconductor
U2
2.5 V programmable ref, SOIC
TL431ACD
ON Semiconductor
U3
Quad Op A
MC3303D
ON Semiconductor
U4
Optocoupler, 1:1 CTR, 4 pin
SFH615AA−X007
Vishay
Hardware
H1
Printed Circuit Board
H2
Connector
171602
Weidmuller (Digi 281−1435−ND)
H3
Connector
171602
Weidmuller (Digi 281−1435−ND)
H4
Standoff, 4−40, alum, hex, .500 inches
8403
HH Smith (Newark 67F4111)
H5
Standoff, 4−40, alum, hex, .500 inches
8403
HH Smith (Newark 67F4111)
H6
Standoff, 4−40, alum, hex, .500 inches
8403
HH Smith (Newark 67F4111)
H7
Standoff, 4−40, alum, hex, .500 inches
8403
HH Smith (Newark 67F4111)
H8
Heatsink, TO−220
590302B03600
Aavid Thermalloy
H9
Heatsink, TO−220
590302B03600
Aavid Thermalloy
H10
Test point, red
5005
Keystone (Digi 5005K−ND)
H11
Test point, black
5006
Keystone (Digi 5006K−ND)
H12
Shoulder Washer
3049K−ND
Digi−Key
H13
Insulator
4672
Keystone
http://onsemi.com
9
AND8124/D
ON Semiconductor and
are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice
to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability
arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages.
“Typical” parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All
operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. SCILLC does not convey any license under its patent rights
nor the rights of others. SCILLC products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications
intended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation where personal injury or death may occur. Should
Buyer purchase or use SCILLC products for any such unintended or unauthorized application, Buyer shall indemnify and hold SCILLC and its officers, employees, subsidiaries, affiliates,
and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury or death
associated with such unintended or unauthorized use, even if such claim alleges that SCILLC was negligent regarding the design or manufacture of the part. SCILLC is an Equal
Opportunity/Affirmative Action Employer. This literature is subject to all applicable copyright laws and is not for resale in any manner.
PUBLICATION ORDERING INFORMATION
LITERATURE FULFILLMENT:
Literature Distribution Center for ON Semiconductor
P.O. Box 5163, Denver, Colorado 80217 USA
Phone: 303−675−2175 or 800−344−3860 Toll Free USA/Canada
Fax: 303−675−2176 or 800−344−3867 Toll Free USA/Canada
Email: [email protected]
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Order Literature: http://www.onsemi.com/litorder
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Phone: 81−3−5773−3850
http://onsemi.com
10
For additional information, please contact your
local Sales Representative.
AND8124/D
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