A 60 W AC-DC Demonstrator with NCP1256

AND9206/D
A 60 W AC-DC Demonstrator
with NCP1256
Housed in a tiny TSOP6 package, the NCP1256 lends
itself very well to designing moderate to high output power
converters. This application note demonstrates the part
capabilities in a 60-W ac-dc adapter, typical of what is
needed for the high-volume net/notebook market.
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APPLICATION NOTE
The Adapter Schematic
The adopted schematic appears in Figure 1. You can see
the NCP1256 surrounded by components implementing
brown-out sensing, over voltage and over temperature
protection. We start the description by the left side of the
board. The mains is applied on the rectifying diode bridge
through an EMI filter made of a 10-mH common-mode
choke. Its leakage inductance is used together with C11 to
form a differential mode filter. Resistors (R15, R27, R17 and
R20) perform the dual function of brown-out sensing and
also ensure the discharge of the X2-capacitor when the
power cord is un-plugged. These resistors must be carefully
selected to fulfill the IEC−950 safety standards as they can
obviously hamper the no-load standby power. Additional
filtering and protection devices can be necessary (VDR,
spark-gaps) to improve the filter as you start qualifying the
final prototype for safety and surge robustness. On the other
hand, it is also possible to differently design the EMI filter
to adopt an X2 capacitor of 0.1 mF. In this case, discharge
resistors are no longer needed and brown-out sensing
resistors can be slightly increased to the benefit of standby
power.
The controller drives a 5 A/650 V power MOSFET with
a small PNP transistor which helps reduce the turn-off event
for an improved efficiency.
resistors are connected to the bulk capacitor which slowly
discharges at power-off and maintains the voltage on the
VCC capacitor for a long time before reset occurs. In this
circuit, the reset level is typically 250 mV below the UVLO
level, considerably reducing the reset time at power off. If
a slightly faster reset time is necessary, you can
advantageously connect a 1 MW resistor from VCC to
ground: it will help the controller reset faster without
bothering the capacitive start-up circuit. The selected
network ensures a start-up sequence below 3 seconds at the
lowest input line. We have tested this sequence with the
board delivering 3 A while powered from a 85 V rms input
voltage (Figure 2). In this worst case, the time at which the
full output voltage is ready remains below the 3 s limit,
giving some margin for the nominal case at 100 V rms.
Please note that the various leakages to earth brought by the
oscilloscope and other active loads have to be minimized
during this test to avoid altering the start-up time.
The start-up sequence is linked to the Vcc capacitor value.
A small value will bring a short start-up time but can
possibly engender a hiccup at power-on. A sufficient level
of energy must be stored in this capacitor as it is the only
source of energy at power-on before the auxiliary winding
takes over the controller’s supply. In the NCP1256, the
reason why the Under Voltage Lock Out (UVLO) has been
placed high enough is to increase the available CV 2 term at
UVLO high. This helps lower the storage capacitor value
while improving the start-up time. Unfortunately, in these
low-standby power supplies, the recurrence of the switching
pulses in light load conditions can be very long. In this
situation, as the refresh of the VCC capacitor is made by
bunches (the part operates in skip cycle) there are chances
that the VCC level slowly goes down until it touches the
part’s UVLO low level and initiates a new start-up sequence.
If we grow the VCC capacitor, the start-up time will suffer.
A possible solution is that described in Figure 3. It consists
of splitting the capacitors and isolating them via a simple
diode.
Start-Up and Self-Supply
The start-up network benefits from the very low current
consumption of the NCP1256. With a 10 mA maximum
current, the part can be cranked with a weak start-up current
which is good for the standby power. For this demonstration
board, we have adopted a capacitive start-up network
implemented around C16 and C19. The dissipation of this
network is almost inexistent compared to a direct connection
to the bulk capacitor for instance. Furthermore, in case of
a latched event, the VCC on the part will collapse at a faster
pace when the user un-plugs the converter as no additional
current can maintain the VCC capacitor voltage on the
controller. This is different from a situation where start-up
© Semiconductor Components Industries, LLC, 2015
December, 2015 - Rev. 1
1
Publication Order Number:
AND9206/D
+
L1
R27
2Meg
R15
2Meg
2
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250 V
F1
2AT
X2
*
85−265 V rms
R17
2Meg
R20
2Meg
C19
4.7n
R29
10k
C12
100uF
400 V
*
R3
0
Vbulk
D11A
BAW56W
C9
0.1uF
**
*16 V
Jumper
D7
1N966
C8
1nF
SMD
630 V
630 V
C16
4.7n
SMD
R1
0
*
D10A
BAV70
D10B
BAV70
R5
33k
BO
FB
GND
R2
0
3
2
1
U1
NCP1256
CS
Vcc
DRV
*
C15
220p
4
5
6
R28
NTC
NTCLE100
E3104JB0
[email protected]°C
OVP line
25 V
C3
4.7uF
*
*
C10
0.1uF
D6
1N4148
R19
1.5k
C2
10n
x3
*1 W SMD
2
R6a
1
35 V
C17
47uF
Q1
2N2907
R16
.
Naux
6
Np
1
R6b
1
R18
47k
**
1 W SMD
1 W SMD
. 7−8−9
Ns
R6c
1
U3B
C13
2.2nF
Type = Y1
C4
100pF
1 kV
D1
BAV21
25 V 25 V
C6
0.1uF
R4
10k
not
wired
L2
1uH
5A
744772010
Wurth
SOT−23
U2
NCP431
R8
1k
C18
22pF
250 V
U3A
R22
10k
C14
1nF
C5b
C5a
680uF 680uF
10−11−12
**
D5
MBR20200CT
Q2
IPA65R190C7
.
T1
R25
47
R26
47
D4
1N4937
Vertical mount
2W 2W
R11 R13
47k 47k
750314896
Wurth
5
22
R23
910
D3
BAV21
R21
22
D2
BAV21
Film cap.
no disc!
dual common cathode
SOT−23
dual common anode
SOT−23
D11B
BAW56W
* *
C1
0.47uF
R7
1k
*
*
D8
1N4148
*
C11
0.22uF
2 x 10mH
Wurth 744822120
IC4
KBU4K
*
−
*
IN
R24
10k
R9
10k
R10
56k
R12
10k
C7
220uF
25 V
Jumper
D9x
Red LED
SMD
R14
33k
19 V / 3.3 A
AND9206/D
Figure 1. The Adapter Uses All the Features Brought by the NCP1256 to
Implement a High-Performance 60 W Converter
AND9206/D
vCC (t )
Power on
vout (t )
Figure 2. The Start-up Sequence is below 3 s when Powered from a 85 V rms
Input Voltage while Delivering 3 A. Here with a Capacitive Start-up Circuit
R3
R2
D6
D3
VCC
.
C3
C17
aux.
Figure 3. The Split Supply Lets you Power the Controller with a Small VCC Capacitor,
Decoupled from a Larger Value directly Connected to the Auxiliary Winding
Protections
In this case, the start-up time involves C3 only as D6
decouples the discharged capacitor C17 from the charging
circuit. When the auxiliary winding charges C17, the voltage
across its terminals increases until it completely supplies the
controller. In standby, the circuit is decoupled by a capacitor
equal to C3 + C17, enough to maintain the VCC in light- to
no-load operations. In the application board, we have
successfully tested a 4.7 mF value for C3 and a 100 mF
capacitor for C17.
The start-up sequence also involves the internal 4 ms
soft-start. During this time, the peak current setpoint is
linearly increased from a very low value up to the allowable
maximum. This soft-start circuitry is activated upon a fresh
start-up but also every time a restart is attempted, e.g. in an
auto-recovery fault mode.
There are several protections required by ac-dc adapters
for the notebook market. They are listed below:
1. Short Circuit Protection, SCP: the adapter must
sustain a permanent short-circuit on its output
without being destroyed. When the fault has
disappeared, the adapter must recover from the
protection mode and deliver the rated power again.
Auto-recovery OCP comes with NCP1256B (65 or
100 kHz). Some applications require a latched
state when a short circuit is detected. In this case,
extension “A” must be considered.
2. Over Voltage Protection, OVP: in case the loop is
broken, e.g. the optocoupler is destroyed or the
TL431 divider network is affected, the adapter
must be immediately stopped and remain in that
state until the user cycles the input power for reset.
The OVP can be detected via the BO pin or simply
through the VCC pin in case runaway occurs.
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3
AND9206/D
3. Over Temperature Protection, OTP: if the
temperature of the adapter exceeds a certain
ambient value, there is a risk of destruction. To
avoid this from happening, a thermal sensor
permanently monitors the temperature and in case
it exceeds the limit set by the designer, the adapter
shuts down permanently. Again, the adapter is
reset when the user cycles the input power and the
temperature has decreased.
4. Over Power Protection, OPP: for some power
supplies, it is important that the maximum output
current stays in control in worse case conditions,
e.g. when the load is drawing more current that
what it should, without being a real short-circuit.
In our design, the nominal output current is 3.2 A
and must stay below 4.5 A in all input voltage
conditions.
vcc (t )
vDRV (t )
Figure 4. The Auxiliary Winding Collapses in
Presence of a Short-Circuit at the Board Output:
the Converter is well protected but the UVLO Trips
First, not the Timer
Let us know check how each requirement has been
separately addressed.
2. when a short-circuit is applied at the end of the
cable, there are chances that the auxiliary voltage
does not collapse, keeping the controller alive
despite an over current on the secondary side.
The timer can therefore count up to 70 ms (typical)
and make the part enter auto-recovery as before
however with a longer recurrence. This is what
Figure 5 shows. Again, a low duty ratio in burst
mode guarantees a low average input power
(0.07/1.94 = 3.6%).
Short-Circuit Protection
The protection is ensured by monitoring the current sense
signal on pin 4. When this voltage exceeds the maximum
internal current setpoint (0.8 V without OPP at low line or
less when OPP is active), the internal 70 ms timer is started.
The timer is reset if the current sense signal goes back below
the maximum internal current setpoint for 8 consecutive
clock times. This can happen if line ripple is superimposed
on the FB signal, for instance at the lowest input level. If the
timer completes its cycle, meaning the fault has been present
longer than 70 ms without adverse reset, all driving pulses
are immediately stopped and the part reduces its
consumption to around 400 mA. As VCC decreases, it
eventually touches the UVLO low level of 8.9 V where the
part re-enters the start-up mode: consumption goes back to
less than 10 mA, VCC rises up again and when reaching 18 V,
the circuit pulses, attempting to re-start. A kind of
auto-recovery burst mode takes place, ensuring a low
average input power. There are two cases that we can think
of:
1. when the auxiliary and the power winding are well
coupled, a short-circuit on the secondary side, very
close to the board output, can potentially collapse
the auxiliary winding on the primary side. This is
the case during the start-up sequence for instance.
As a result, the internal timer does not have time to
reach completion and the pulses are interrupted by
the UVLO level. This is what Figure 4 shows you.
With the auto-recovery version, the part re-starts in
a double hiccup. With the latched version,
the pre-short capability makes the part latch off at
the first power on sequence when UVLO is
touched first.
vcc (t )
vDRV (t )
Figure 5. In this Case, the Timer Interrupts the
Switching Pulses after 100 ms and the Part Enters
Auto-recovery
Please note that the short circuit test can potentially latch
off the controller as VCC runs away. If the leakage
inductance contribution is too strong, VCC can significantly
vary as IOUT increases. To avoid the problem with this
board, you can a) remove the OVP jumper b) try to calm
down the leakage contribution by inserting a small resistor
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4
AND9206/D
in series with diode D3. A few ohms should be enough to
maintain VCC below the Zener voltage. Please also note that
if VCC approaches the Zener voltage – without tripping the
BO latch – the voltage increase over R7 will potentially
disturb BO thresholds. As a conclusion, the Zener voltage
must be selected so that VCC is always below in worst case
operation. The Zener shall be activated when the loop is
broken only.
In Figure 7, the optocoupler LED has been shorted and the
converter runs open loop. VOUT increases but the auxiliary
VCC also does until Zener diode D7 conducts and lifts the BO
pin for four successive clock cycles. The part latches off and
remains locked until a BO reset or a VCC cycling occurs.
552 V
Over Voltage Protection
vDS (t )
When the optocoupler is broken or when the TL431
divider network is affected by a severe drift (or one of its
resistor is missing or features a wrong value), then the output
voltage can escape from the limits imposed by the
specifications: this is an over voltage condition. In the
majority of cases, this condition is considered hazardous for
the downstream load and the adapter must completely shut
off. The NCP1256 deals with this problem via two possible
options: a) the VCC runs away and touches the upper limit of
26 V (typical) b) a dc voltage lifts the BO pin above 4.5 V
and latches off the part. Both solutions are implemented in
the demonstration board. In a latched state, VCC hiccups up
and down while all pulses are disabled. Reset occurs when
VCC drops below the VCC reset value (8.65 V typical) or
when the BO voltage is cycled (user unplugs the converter).
However, BO cycling can only be detected while the IC
operates. If BO resets while VCC rises up, no reset is done.
One way to accelerate the VCC drop is to add a 1 MW resistor
from VCC to ground. This solution can only be considered
if a) the standby power margin authorizes it b) enough
start−up current exists.
We have captured several oscilloscope shots to illustrate
the behavior of the circuit. Figure 7 displays the VCC
waveform obtained when the optocoupler LED is shorted in
the secondary side at start up. The voltage increases and
when it reaches 26 V, the part latches off.
vout (t )
Figure 7. The Optocoupler LED has been Shorted in
the Secondary Side and Vout Increases. This
Information is Detected in the Primary Side and
Latches off the Part via the BO Pin
Over Temperature Protection
OTP can be implemented by connecting a Negative
Temperature Coefficient resistor (NTC) from the auxiliary
winding to the current sense pin. This is what is shown in
Figure 1 around R28, R19 et D2. In this configuration, when
the temperature increases, the NTC resistance starts
decreasing and lifts up pin 3 voltage during the off time only
(peak current and delivered power are not affected). When
the level reaches 3 V, the part simply latches off and requires
a reset before re-start. Reset occurs when the user cycles the
input voltage.
vcc (t )
vDRV (t )
Figure 6. Here the VCC Runs away at Start up and
Activates the Controller Latch
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AND9206/D
vCS (t )
vCS (t )
Figure 8. OTP is Implemented in a Simple and Efficient Way
datasheet, the current setpoint inside the circuit depends on
pin 2 level divided by 3. In fault conditions, when the loop
is lost, the feedback level can go up to 4 V. To avoid any
current runaway, the maximum voltage setpoint is safely
clamped to 0.8 V. Reference [1] points to an article
describing the over power phenomenon and how to limit the
maximum power the converter can deliver. In the NCP1256,
an offset is created on the CS pin and depends on the BO
level. At the lowest line input, 85 V rms, the offset current
is almost inexistent and full power is authorized. As the
input voltage increases, the offset builds up in relationship
to the series resistor R23 in Figure 1. The below data show
the results obtained at different input voltages.
We tested the demonstration board with the calculated
setup and the test revealed a trip point around 103°C, close
to what was expected.
Over Power Protection
A current-mode power supply works by setting the
inductor peak current according to the output power
demand. The inductor current is transformed into a voltage
by a sense resistor, R6 in our adapter. The peak current
setpoint depends on the error voltage delivered on the
feedback loop pin. In our adapter, this is the current forced
by the TL431 on the secondary side and reflected to the
primary over pin 2 of the NCP1256. As detailed in the
4.6
Output current (A)
4.4
4.2
4
3.8
3.6
3.4
3.2
3
85
135
185
235
Input voltage (V rms)
Figure 9. Maximum Delivered Output Current as a Function on the Input Line
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6
AND9206/D
The Secondary Side Feedback
Efficiency Performance
The feedback is made around a classical TL431 network.
We used the automated spreadsheet to evaluate the
component values based on 1 kHz bandwidth target. A
thorough description of the method is given in [2]. In an
effort to further decrease the no-load standby power, we
have implemented a proprietary technique around the
NCP431. Despite its already low bias current, the data-sheet
specifies a minimum current of 100 mA. Usually, this
minimum bias is ensured by paralleling a resistor with the
optocoupler LED which exhibits a 1 V forward drop. If this
extra current plays a positive role in the converter
performance at high output levels, it is obviously
detrimental to the standby power since it permanently draws
several milliwatts in our 19 V example. The idea is to get rid
of this bias in standby mode, without affecting the transient
response in case the load is suddenly re-applied. Figure 10
shows you the proprietary idea we came up with. The
principle is extremely simple: capacitor C5 delivers a
voltage equal to that of the output at full load, i.e. 19 V. These
19 V are used to bias the NCP431 via R19. As the load is
getting lighter, the controller enters the skip cycle mode.
Given the time constant offered by C5 together with the load
made of R19 and the TL431 bias, the voltage across C5
cannot be maintained: its average value collapses and the
NCP431 bias disappears. In case the load is suddenly
re-applied, the bias is automatically regenerated as the
controller expands the duty ratio and the response is not
affected. With a 19-V output, this technique helps saving
several milliwatts seen from the primary side.
The NCP1256 excels in terms of efficiency and standby
power. We have made a series of tests on the proposed
adapter, carried at both high and low lines. The voltage is
measured at the board output. The results appear below.
Table 1. EFFICIENCY PERFORMANCE
Output Power
Efficiency − Vin =
110 V rms (%)
Efficiency − Vin =
230 V rms (%)
15 W − 25%
89.4
88.8
30 W − 50%
89.5
89.7
45 W − 75%
89.7
89.7
60 W − 100%
89.2
90.1
Average efficiency
89.4
89.6
Table 2. NO-LOAD STANDBY POWER, LED IS OFF
Output Power
Input Power − Vin =
100 V rms (mW)
Input Power − Vin =
230 V rms (mW)
0
30
50
Table 3. NO−LOAD STANDBY POWER, LED IS ON
Output Power
Input Power − Vin =
100 V rms (mW)
Input Power − Vin =
230 V rms (mW)
0
42
60
Table 4. LIGHT LOAD EFFICIENCY
Output Power
Input Power − Vin =
100 V rms (W)
Input Power − Vin =
230 V rms (W)
0.5 W
0.66
0.67
0.6 W
0.79
0.80
0.7 W
0.90
0.92
out
5
.
7
8
5
The performance is linked to the combined action of the
frequency foldback and the skip cycle operation at constant
peak current. Please note that the no-load standby power
includes the 4 MW discharge resistors string placed across
the 0.22 mF X2 capacitor on the input filter. These numbers
are excellent considering a low-voltage controller featuring
a start-up network.
19
5
Figure 10. Simple Peak Rectifier Generates a
Voltage Across C5 whose Amplitude Falls down as
the Controller Starts to Skip Cycle
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7
AND9206/D
Transient Response
transient tests can be performed to check that everything is
ok. In our case, the output has been loaded by a current step
from 0.1 A to 3.5 A with a slew-rate of 1 A/ms. Two input
voltages have been considered, 100 V rms and 230 V rms.
Such a wide loading step is a quite stringent test but as shown
in Figure 11, the response at the board level stays within 2%
of the nominal voltage of 19 V.
The loop small-signal response has been measured and is
the object of a dedicated application note. Please refer to [2]
for more details. Loop stability is an important matter and
must be seriously considered when working on high-volume
projects. No trials and errors in the laboratory while
observing the transient response, please! However, once the
loop has been thoroughly reviewed and analyzed, some
vout (t )
400 mV
vout (t )
100 V rms
390 mV
230 V rms
Figure 11. The Transient Response at Low Line and High Line are almost
Identical and do not Show Signs of Instabilities
The ripple seen in the left side of the figure is due to the
ripple on the bulk capacitor suddenly increasing when the
load current is back.
simple to implement. This makes the part an ideal candidate
where space constraints, performance and cost sensitivity
are key considerations.
Conclusion
References
This application note describes how an ac-dc converter
meeting all new efficiency challenges can be built with the
new NCP1256. Despite a small TSOP−6 package and
a limited amount of pins, the performance of the final board
nicely competes against other more complex circuits by
offering a similar set of options plus protection features
[1] C. Basso, “Switch Mode Power Supplies: SPICE
Simulations and Practical Designs”, McGraw−Hill,
2012
[2] AND8453, “Loop Control Design of an ac−dc
Adapter Using the NCP1250”, www.onsemi.com
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