DATASHEET

DATASHEET
Multiphase PWM Regulator for AMD Fusion™ Desktop
CPUs Using SVI 2.0
ISL95712
Features
The ISL95712 is fully compliant with AMD Fusion™ SVI 2.0 and
provides a complete solution for microprocessor and graphics
processor core power. The ISL95712 controller supports two
Voltage Regulators (VRs) for Core and Northbridge outputs. The
Core VR can be configured for 4-, 3-, 2-, or 1-phase operation while
the Northbridge VR supports 3-, 2- or 1-phase configurations for
maximum flexibility. The two VRs share a serial control bus to
communicate with the AMD CPU and achieve lower cost and
smaller board area compared with two-chip solutions.
• Supports AMD SVI 2.0 serial data bus interface and PMBus
- Serial VID clock frequency range 100kHz to 25MHz
• Dual output controller with 12V integrated core gate drivers
• Precision voltage regulation
- 0.5% system accuracy over-temperature
- 0.5V to 1.55V in 6.25mV steps
- Enhanced load line accuracy
• Supports multiple current sensing methods
- Lossless inductor DCR current sensing
- Precision resistor current sensing
The PWM modulator is based on Intersil’s Robust Ripple
Regulator R3™ Technology. Compared to traditional modulators,
the R3™ modulator can automatically change switching
frequency for faster transient settling time during load transients
and improved light load efficiency.
• Programmable 1-, 2-, 3- or 4-phase for the core output and
1- , 2- or 3-phase for the Northbridge output
The ISL95712 has several other key features. Both outputs
support DCR current sensing with a single NTC thermistor for
DCR temperature compensation or accurate resistor current
sensing. They also utilize remote voltage sense, adjustable
switching frequency, OC protection and power-good indicators.
• Adaptive body diode conduction time reduction
Applications
• High efficiency across entire load range
• Superior noise immunity and transient response
• Output current and voltage telemetry
• Differential remote voltage sensing
• Programmable slew rate
• AMD Fusion CPU/GPU core power
• Programmable VID offset and droop on both outputs
• Desktop computers
• Programmable switching frequency for both outputs
• Excellent dynamic current balance between phases
• Protection: OCP/WOC, OVP, PGOOD and thermal monitor
• Small footprint 52 Ld 6x6 QFN package
- Pb-free (RoHS compliant)
Performance
1.6
100
CORE
90
OUTPUT VOLTAGE (V)
EFFICIENCY (%)
1.5
NORTHBRIDGE
80
70
CORE
(PSI1)
60
50
40
30
20
1.3
NORTHBRIDGE
1.2
1.1
10
0
CORE
1.4
DAC = 1.500V
DAC = 1.500V
0
10
20
30
40
50
60
70
80
LOAD CURRENT (A)
FIGURE 1. EFFICIENCY vs LOAD
November 2, 2015
FN8566.1
1
90
100
110
1.0
0
10
20
30
40
50
60
70
80
90
100
110
LOAD CURRENT (A)
FIGURE 2. VOUT vs LOAD
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Copyright Intersil Americas LLC 2014, 2015. All Rights Reserved
Intersil (and design) and R3 Technology are trademarks owned by Intersil Corporation or one of its subsidiaries.
All other trademarks mentioned are the property of their respective owners.
ISL95712
Table of Contents
Simplified Application Circuit for High Power CPU Core . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
Pin Configuration. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
Pin Descriptions. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
Ordering Information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
Absolute Maximum Ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
Thermal Information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
Recommended Operating Conditions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
Electrical Specifications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
Gate Driver Timing Diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
Theory of Operation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
Multiphase R3™ Modulator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
Diode Emulation and Period Stretching . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Channel Configuration . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Power-On Reset . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Start-Up Timing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
Diode Throttling . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
Voltage Regulation and Load Line Implementation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
Differential Sensing. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Phase Current Balancing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Modes of Operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Dynamic Operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
Adaptive Body Diode Conduction Time Reduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
Resistor Configuration Options. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
VR Offset Programming . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
VID-on-the-Fly Slew Rate Selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
CCM Switching Frequency . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
AMD Serial VID Interface 2.0 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
Pre-PWROK Metal VID. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
SVI Interface Active . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
VID-on-the-Fly Transition . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
SVI Data Communication Protocol . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
SVI Bus Protocol. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
Power States . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
Dynamic Load Line Slope Trim . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
Dynamic Offset Trim . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
Telemetry. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
PMBus Interface . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
Protection Features. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
Overcurrent . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
Current-Balance . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
Undervoltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
Overvoltage. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
Thermal Monitor [NTC, NTC_NB] . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
Fault Recovery . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
Interface Pin Protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
Key Component Selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
Inductor DCR Current-Sensing Network . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
Resistor Current-Sensing Network . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
Overcurrent Protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
Load Line Slope . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
Compensator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
Current Balancing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
Thermal Monitor Component Selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
Layout Guidelines . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
PCB Layout Considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
Revision History. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34
About Intersil . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34
Package Outline Drawing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
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2
FN8566.1
November 2, 2015
ISL95712
NB_PH2
ISEN2_NB
Ri
VNB1
VDDP
ISEN1_NB
VDD
NB_PH1
ENABLE
Simplified Application Circuit for High Power CPU Core
ISUMN_NB
Cn
VNB2
UGATE_NB
PHASE_NB
NTC
NB_PH1
+12V
BOOT_NB
LGATE_NB
NB_PH1
ISUMP_NB
NB_PH2
VNB
PROG
COMP_NB
+12V
FB_NB
*
*OPTIONAL
PWM2_NB
VSEN_NB
VNB_SENSE
ISL6625A
*
VNB1
NB_PH2
IMON_NB
NTC_NB
VNB2
+12V
I2DATA
VR_HOT_L
PWM4
PWROK
SVT
µP
ISL6625A
I2CLK
THERMAL INDICATOR
PH4
SVD
VO4
SVC
VDDIO
+12V
NTC
PWM3
COMP
*
ISL95712
PH3
VO3
FB
*
BOOT2
*OPTIONAL
VCORE_SENSE
VSEN
UGATE2
RTN
PHASE2
PH1
ISEN1
PH2
ISEN2
PH3
ISEN3
PH4
ISEN4
LGATE2
BOOT1
Ri
ISUMP
VCORE
PH2
VO2
+12V
PHASE1
LGATE1
PH1
VO1
PH4
PH3
PH1
PH2
VO4
PGOOD
NTC
GND PAD
Cn
VO3
+12V
UGATE1
ISUMN
PGOOD_NB
VO1
VO2
ISL6625A
IMON
FIGURE 3. TYPICAL APPLICATION CIRCUIT USING INDUCTOR DCR SENSING
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3
FN8566.1
November 2, 2015
ISL95712
Pin Configuration
PWM3_NB
PWM2_NB
PROG
I2DATA
I2CLK
COMP_NB
PGOOD_NB
VSEN_NB
FB_NB
ISUMP_NB
ISUMN_NB
ISEN2_NB
ISEN1_NB
ISL95712
(52 LD QFN)
TOP VIEW
52 51 50 49 48 47 46 45 44 43 42 41 40
ISEN3_NB
1
39 PWM4
NTC_NB
2
38 PWM3
IMON_NB
3
37 BOOT1_NB
SVC
4
36 UGATE1_NB
VR_HOT_L
5
SVD
6
GND
VDDIO
7
(BOTTOM PAD)
SVT
8
32 VDDP
ENABLE
9
31 UGATE2
PWROK 10
30 PHASE2
35 PHASE1_NB
34 LGATE1_NB
33 LGATE2
IMON 11
29 BOOT2
NTC
28 LGATE1
12
ISEN4 13
27 UGATE1
PHASE1
VDD
PGOOD
COMP
BOOT1
RTN
FB
VSEN
ISEN1
ISUMP
ISUMN
ISEN3
IISEN2
14 15 16 17 18 19 20 21 22 23 24 25 26
Pin Descriptions
PIN NUMBER
SYMBOL
1
ISEN3_NB
2
NTC_NB
3
IMON_NB
4
SVC
5
VR_HOT_L
6
SVD
7
VDDIO
VDDIO is the processor memory interface power rail and this pin serves as the reference to the controller
IC for this processor I/O signal level.
8
SVT
Serial VID Telemetry (SVT) data line input to the CPU from the controller IC. Telemetry and VID-on-the-fly
complete signal provided from this pin.
9
ENABLE
Enable input. A high level logic on this pin enables both VRs.
10
PWROK
System power-good input. When this pin is high, the SVI 2 interface is active and the I2C protocol is
running. While this pin is low, the SVC and SVD input states determine the pre-PWROK metal VID. This
pin must be low prior to the ISL95712 PGOOD output going high per the AMD SVI 2.0 Controller
Guidelines.
11
IMON
12
NTC
13
ISEN4
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DESCRIPTION
Individual current sensing for Channel 3 of the Northbridge VR. When ISEN3_NB is pulled to +5V, the
controller will disable Channel 3 and the Northbridge VR will run 2-phase.
Thermistor input to VR_HOT_L circuit to monitor Northbridge VR temperature.
Northbridge output current monitor. A current proportional to the Northbridge VR output current is
sourced from this pin.
Serial VID clock input from the CPU processor master device.
Thermal indicator signal to AMD CPU. Thermal overload open-drain output indicator active LOW.
Serial VID data bidirectional signal from the CPU processor master device to the VR.
Core output current monitor. A current proportional to the Core VR output current is sourced from this pin.
Thermistor input to VR_HOT_L circuit to monitor Core VR temperature.
4
ISEN4 is the individual current sensing for Channel 4 of the Core VR. When ISEN4 is pulled to +5V, the
controller disables Channel 4, and the Core VR runs in three-phase mode.
FN8566.1
November 2, 2015
ISL95712
Pin Descriptions (Continued)
PIN NUMBER
SYMBOL
14
ISEN3
ISEN3 is the individual current sensing for Channel 3 of the Core VR. When ISEN3 is pulled to +5V, the
controller disables Channel 3, and the Core VR runs in two-phase mode.
15
ISEN2
Individual current sensing for Channel 2 of the Core VR. When ISEN2 is pulled to +5V, the controller
disables Channel 2, and the Core VR runs in single-phase mode.
16
ISEN1
Individual current sensing for Channel 1 of the Core VR. If ISEN2 is tied to +5V, this pin cannot be left
open and must be tied to GND with a 10kΩ resistor. If ISEN1 is tied to +5V, the Core portion of the IC is
shut down.
17
ISUMP
Noninverting input of the transconductance amplifier for current monitor and load line of Core output.
18
ISUMN
Inverting input of the transconductance amplifier for current monitor and load line of Core output.
19
VSEN
Output voltage sense pin for the Core controller. Connect to the +sense pin of the microprocessor die.
20
RTN
Output voltage sense return pin for both Core VR and Northbridge VR. Connect to the -sense pin of the
microprocessor die.
21
FB
22
VDD
5V bias power. A resistor [2Ω] and a decoupling capacitor should be used from the +5V supply. A high
quality, X7R dielectric MLCC capacitor is recommended.
23
PGOOD
Open-drain output to indicate the Core output is ready to supply regulated voltage. Pull-up externally to
VDD or 3.3V through a resistor.
24
COMP
Core controller error amplifier output. A resistor from COMP to GND sets the Core VR offset voltage.
25
BOOT1
Connect an MLCC capacitor across the BOOT1 and PHASE1 pins. The boot capacitor is charged, through
an internal boot diode connected from the VDDP pin to the BOOT1 pin, each time the PHASE1 pin drops
below VDDP minus the voltage dropped across the internal boot diode.
26
PHASE1
Current return path for the Phase 1 high-side MOSFET gate driver of VR1. Connect the PHASE1 pin to the
node consisting of the high-side MOSFET source, the low-side MOSFET drain and the output inductor of
Phase 1.
27
UGATE1
Output of the Phase 1 high-side MOSFET gate driver of the Core VR. Connect the UGATE1 pin to the gate
of the Phase 1 high-side MOSFET(s).
28
LGATE1
Output of the Phase 1 low-side MOSFET gate driver of the Core VR. Connect the LGATE1 pin to the gate
of the Phase 1 low-side MOSFET(s).
29
BOOT2
Connect an MLCC capacitor across the BOOT2 and PHASE2 pins. The boot capacitor is charged, through
an internal boot diode connected from the VDDP pin to the BOOT2 pin, each time the PHASE2 pin drops
below VDDP minus the voltage dropped across the internal boot diode.
30
PHASE2
Current return path for the Phase 2 high-side MOSFET gate driver of the Core VR. Connect the PHASE2
pin to the node consisting of the high-side MOSFET source, the low-side MOSFET drain and the output
inductor of Phase 2.
31
UGATE2
Output of the Phase 2 high-side MOSFET gate driver of the Core VR. Connect the UGATE2 pin to the gate
of the Phase 2 high-side MOSFET(s).
32
VDDP
Input voltage bias for the internal gate drivers. Connect +12V to the VDDP pin. Decouple with at least 1µF
of capacitance to GND. A high quality, X7R dielectric MLCC capacitor is recommended.
33
LGATE2
Output of the Phase 2 low-side MOSFET gate driver of the Core VR. Connect the LGATE2 pin to the gate
of the Phase 2 low-side MOSFET(s).
34
LGATE1_NB
Output of Northbridge Phase 1 low-side MOSFET gate driver. Connect the LGATE1_NB pin to the gate of
the Northbridge VR Phase 1 low-side MOSFET(s).
35
PHASE1_NB
Current return path for Northbridge VR Phase 1 high-side MOSFET gate driver. Connect the PHASE1_NB
pin to the node consisting of the high-side MOSFET source, the low-side MOSFET drain and the output
inductor of Northbridge Phase 1.
36
UGATE1_NB
Output of the Phase 1 high-side MOSFET gate driver of the Northbridge VR. Connect the UGATE1_NB pin
to the gate of the Northbridge VR Phase 1 high-side MOSFET(s).
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DESCRIPTION
Output voltage feedback to the inverting input of the Core controller error amplifier.
5
FN8566.1
November 2, 2015
ISL95712
Pin Descriptions (Continued)
PIN NUMBER
SYMBOL
DESCRIPTION
37
BOOT1_NB
Connect an MLCC capacitor across the BOOT1_NB and PHASE1_NB pins. The boot capacitor is charged,
through an internal boot diode connected from the VDDP pin to the BOOT1_NB pin, each time the
PHASE1_NB pin drops below VDDP minus the voltage dropped across the internal boot diode.
38
PWM3
PWM output of Channel 3 of the Core VR. Disabled if ISEN3 is tied to +5V.
39
PWM4
PWM output of Channel 4 of the Core VR. Disabled if ISEN4 is tied to +5V.
40
PWM2_NB
PWM output for Channel 2 of the Northbridge VR. Disabled when ISEN2_NB is tied to +5V.
41
PWM3_NB
PWM output for Channel 3 of the Northbridge VR. Disabled when ISEN3_NB is tied to +5V.
42, 43
I2CLK, I2DATA
44
PROG
45
PGOOD_NB
Open-drain output to indicate the Northbridge output is ready to supply regulated voltage. Pull-up
externally to VDD or 3.3V through a resistor.
46
COMP_NB
Northbridge VR error amplifier output. A resistor from COMP_NB to GND sets the Northbridge VR offset
voltage and is used to set the switching frequency for the Core VR and Northbridge VR.
47
FB_NB
48
VSEN_NB
Output voltage sense pin for the Northbridge controller. Connect to the +sense pin of the microprocessor
die.
49
ISUMN_NB
Inverting input of the transconductance amplifier for current monitor and load line of the Northbridge VR.
50
ISUMP_NB
Noninverting input of the transconductance amplifier for current monitor and load line of the Northbridge
VR.
51
ISEN1_NB
Individual current sensing for Channel 1 of the Northbridge VR. If ISEN1_NB is tied to +5V, this pin cannot
be left open and must be tied to GND with a 10kΩ resistor. If ISEN1_NB is tied to +5V, the Northbridge
portion of the IC is shutdown.
52
ISEN2_NB
Individual current sensing for Channel 2 of the Northbridge VR. When ISEN2_NB is pulled to +5V, the
controller will disable Channels 2 and 3 and the Northbridge VR will run 1-phase.
SMBus/PMBus/I2C interface used for additional communication with the controller outside of the SVI2
pins. Tie to VCC with 4.7kΩ pull-up resistor when not used.
A resistor from the PROG pin to GND programs the switching frequency.
Output voltage feedback to the inverting input of the Northbridge controller error amplifier.
GND (Bottom Pad)
Signal common of the IC. Unless otherwise stated, signals are referenced to the GND pin.
Ordering Information
PART NUMBER
(Notes 1, 2, 3)
PART
MARKING
TEMP.
RANGE (°C)
PACKAGE
(RoHS Compliant)
PKG.
DWG. #
ISL95712HRZ
95712 HRZ
-10 to +100
52 Ld 6x6 QFN
L52.6x6A
ISL95712IRZ
95712 IRZ
-40 to +100
52 Ld 6x6 QFN
L52.6x6A
NOTES:
1. Add “-T” suffix for tape and reel. Please refer to TB347 for details on reel specifications.
2. Intersil Pb-free plus anneal products employ special Pb-free material sets; molding compounds/die attach materials and 100% matte tin plate
termination finish, which are RoHS compliant and compatible with both SnPb and Pb-free soldering operations. Intersil Pb-free products are MSL
classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
3. For Moisture Sensitivity Level (MSL), please see device information page for ISL95712. For more information on MSL please see tech brief TB363.
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6
FN8566.1
November 2, 2015
ISL95712
Absolute Maximum Ratings
Thermal Information
Supply Voltage, VDD . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +7V
Input Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +15V
Gate Driver Supply Voltage, VDDP . . . . . . . . . . . . . . . . . . . . . . -0.3V to + 15V
Boot Voltage (VBOOT) . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to VDDP + 15V
UGATE Voltage (VUGATE). . . . . . . . . . . . . . VPHASE - 0.3VDC to VBOOT + 0.3V
VPHASE - 3.5V (<100ns Pulse Width, 2µJ) to VBOOT + 0.3V
LGATE Voltage (VLGATE) . . . . . . . . . . . . . . . . . GND - 0.3VDC to VDDP + 0.3V
GND - 5V (<100ns Pulse Width, 2µJ) to VDDP + 0.3V
PHASE Voltage (VPHASE) . . . . . . . . . . . . . . . . . . . . . GND - 0.3VDC to 25VDC
GND - 8V (>400ns Pulse Width, 20µ) to 30V (<200ns)
Open-Drain Outputs, PGOOD, PGOOD_NB, VR_HOT_L. . . . . . . -0.3V to +7V
All Other Pins . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-0.3V to VDD + 0.3V
Thermal Resistance (Typical)
JA (°C/W) JC (°C/W)
52 Ld QFN Package (Notes 4, 5) . . . . . . . .
28
2.5
Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . . . . . . . .+150°C
Maximum Storage Temperature Range . . . . . . . . . . . . . .-65°C to +150°C
Pb-Free Reflow Profile . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . see TB493
Recommended Operating Conditions
Supply Voltage, VDD . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +5V ±5%
Input Supply and Gate Drive Voltages, VDDP . . . . . . . . . . . . . . . +12V ±5%
Ambient Temperature
HRZ. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-10°C to +100°C
IRZ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-40°C to +100°C
Junction Temperature
HRZ. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-10°C to +125°C
IRZ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-40°C to +125°C
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product
reliability and result in failures not covered by warranty.
NOTES:
4. JA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See Tech
Brief TB379.
5. For JC, the “case temp” location is the center of the exposed metal pad on the package underside.
Electrical Specifications
Operating Conditions: VDD = 5V, TA = -10°C to +100°C (HRZ), fSW = 300kHz, unless otherwise noted.
Boldface limits apply across the operating temperature range, -40°C to +100°C.
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
(Note 6)
TYP
MAX
(Note 6)
UNIT
12.5
14.0
mA
125
µA
4.50
V
INPUT POWER SUPPLY
+5V Supply Current
IVDD
ENABLE = 1V
ENABLE = 0V
POWER-ON-RESET THRESHOLDS
VDD POR Threshold
VDD_PORr
VDD rising
4.35
VDD_PORf
VDD falling
4.00
HRZ
%Error (VOUT)
No load; closed loop, active mode range,
VID = 0.75V to 1.55V
-0.5
VID = 0.25V to 0.74375V
IRZ
%Error (VOUT)
No load; closed loop, active mode range,
VID = 0.75V to 1.55V
VID = 0.25V to 0.74375V
-12
+12
4.15
V
SYSTEM AND REFERENCES
System Accuracy
+0.5
%
-10
+10
mV
-0.8
+0.8
%
mV
Maximum Output Voltage
VOUT(max)
VID = [00000000]
1.55
V
Minimum Output Voltage
VOUT(min)
VID = [11111111]
0
V
CHANNEL FREQUENCY
Nominal Channel Frequency
280
fSW(nom)
Adjustment Range
300
300
320
kHz
450
kHz
AMPLIFIERS
Current-Sense Amplifier Input Offset
Error Amp DC Gain
HRZ
IFB = 0A
-0.15
+0.15
mV
IRZ
IFB = 0A
-0.20
+0.20
mV
Av0
Error Amp Gain-Bandwidth Product
GBW
CL = 20pF
119
dB
17
MHz
20
nA
ISEN
Input Bias Current
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7
FN8566.1
November 2, 2015
ISL95712
Electrical Specifications
Operating Conditions: VDD = 5V, TA = -10°C to +100°C (HRZ), fSW = 300kHz, unless otherwise noted.
Boldface limits apply across the operating temperature range, -40°C to +100°C. (Continued)
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
(Note 6)
TYP
MAX
(Note 6)
UNIT
0.4
V
POWER-GOOD (PGOOD AND PGOOD_NB) AND PROTECTION MONITORS
PGOOD Low Voltage
VOL
IPGOOD = 4mA
PGOOD Leakage Current
IOH
PGOOD = 3.3V
-1
1
PWROK High Threshold
750
VR_HOT_L Pull-Down
11
µA
mV
Ω
PWROK Leakage Current
1
µA
VR_HOT_L Leakage Current
1
µA
1.5
Ω
GATE DRIVER
UGATE Pull-Up Resistance
RUGPU
200mA source current
1.0
UGATE Source Current
IUGSRC
UGATE - PHASE = 2.5V
2
UGATE Sink Resistance
RUGPD
250mA sink current
UGATE Sink Current
IUGSNK
UGATE - PHASE = 2.5V
2
LGATE Pull-Up Resistance
RLGPU
250mA source current
1.0
LGATE Source Current
ILGSRC
LGATE - VSSP = 2.5V
2
LGATE Sink Resistance
RLGPD
250mA sink current
0.5
LGATE Sink Current
ILGSNK
LGATE - VSSP = 2.5V
4
A
UGATE to LGATE Dead Time
tUGFLGR
UGATE falling to LGATE rising, no load
59
ns
LGATE to UGATE Dead Time
tLGFUGR
LGATE falling to UGATE rising, no load
37
ns
1.0
A
1.5
Ω
A
1.5
Ω
A
0.9
Ω
PROTECTION
Overvoltage Threshold
OVTH
Undervoltage Threshold
UV TH
Current Imbalance Threshold
VSEN rising above setpoint for >1µs
275
325
375
mV
VSEN falls below setpoint for >1µs
275
325
375
mV
One ISEN above another ISEN for >1.2ms
9
mV
15
µA
Way Overcurrent Trip Threshold
[IMONx Current Based Detection]
IMONxWOC
All states, IDROOP = 60µA, RIMON = 135kΩ
Overcurrent Trip Threshold
[IMONx Voltage Based Detection]
VIMONx_OCP
All states, IDROOP = 45µA,
IIMONx = 11.25µA, RIMON = 135kΩ
1.485
1.510
1.535
V
1
V
LOGIC THRESHOLDS
ENABLE Input Low
VIL
ENABLE Input High
ENABLE Leakage Current
VIH
HRZ
1.6
V
VIH
IRZ
1.65
V
IENABLE
ENABLE = 0V
-1
0
ENABLE = 1V
SVT Impedance
1
µA
1
µA
30
%
1
µA
50
SVC, SVD Input Low
VIL
SVC, SVD Input High
VIH
SVC, SVD Leakage
% of VDDIO
% of VDDIO
70
ENABLE = 0V, SVC, SVD = 0V and 1V
-1
ENABLE = 1V, SVC, SVD = 1V
-5
ENABLE = 1V, SVC, SVD = 0V
-35
Ω
%
-20
1
µA
-5
µA
1
V
0.5
µA
PWM
PWM Output Low
V0L
Sinking 5mA
PWM Output High
V0H
Sourcing 5mA
PWM Tri-State Leakage
3.5
V
PWM = 2.5V
THERMAL MONITOR
NTC Source Current
NTC = 0.6V
NTC Thermal Warning Voltage
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8
27
30
33
µA
600
640
680
mV
FN8566.1
November 2, 2015
ISL95712
Electrical Specifications
Operating Conditions: VDD = 5V, TA = -10°C to +100°C (HRZ), fSW = 300kHz, unless otherwise noted.
Boldface limits apply across the operating temperature range, -40°C to +100°C. (Continued)
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
(Note 6)
NTC Thermal Warning Voltage
Hysteresis
TYP
MAX
(Note 6)
20
NTC Thermal Shutdown Voltage
UNIT
mV
530
580
630
mV
Maximum Programmed
16
20
24
mV/µs
Minimum Programmed
8
10
12
mV/µs
SLEW RATE
VID-on-the-Fly Slew Rate
NOTE:
6. Compliance to datasheet limits is assured by one or more methods: production test, characterization and/or design.
Gate Driver Timing Diagram
PWM
tLGFUGR
tFU
tRU
1V
UGATE
1V
LGATE
tRL
tFL
tUGFLGR
FIGURE 4. GATE DRIVER TIMING DIAGRAM
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FN8566.1
November 2, 2015
ISL95712
Theory of Operation
1-phase mode, the master clock signal will be distributed to
Phase 1 only and will be the Clock1 signal.
Multiphase R3™ Modulator
The ISL95712 is a multiphase regulator implementing two voltage
regulators, CORE VR and Northbridge (NB) VR, on one chip
controlled by AMD’s™ SVI2™ protocol. The CORE VR can be
programmed for 1-, 2-, 3- or 4-phase operation. The Northbridge VR
can be configured for 1-, 2-, or 3-phase operation. Both regulators
use the Intersil patented R3™ (Robust Ripple Regulator) modulator.
The R3™ modulator combines the best features of fixed frequency
PWM and hysteretic PWM while eliminating many of their
shortcomings. Figure 5 conceptually shows the multiphase R3™
modulator circuit, and Figure 6 shows the operation principles.
MASTER CLOCK CIRCUIT
MASTER
CLOCK
COMP
PHASE
VCRM
SEQUENCER
GMVO
CLOCK1
CLOCK2
CLOCK3
MASTER
CLOCK
CLOCK1
PWM1
CLOCK3
PWM3
SLAVE CIRCUIT 1
CLOCK1
S
R
Q
PWM1
PHASE1
L1
IL1
VCRS1
COMP
PWM2
CRM
VW
HYSTERETIC
WINDOW
VCRM
CLOCK2
VW
MASTER
CLOCK
VW
VW
VO
CO
GM
VCRS2
VCRS3
VCRS1
CRS1
SLAVE CIRCUIT 2
VW
CLOCK2
S
R
Q
PWM2
PHASE2
L2
IL2
VCRS2
GM
CRS2
SLAVE CIRCUIT 3
VW
CLOCK3
S
R
Q
PWM3
PHASE3
L3
IL3
VCRS3
GM
CRS3
FIGURE 5. R3™ MODULATOR CIRCUIT
Inside the IC, the modulator uses the master clock circuit to
generate the clocks for the slave circuits. The modulator
discharges the ripple capacitor Crm with a current source equal
to gmVo, where gm is a gain factor. Crm voltage VCRM is a
sawtooth waveform traversing between the VW and COMP
voltages. It resets to VW when it hits COMP, and generates a
one-shot master clock signal. A phase sequencer distributes the
master clock signal to the slave circuits. If the Core VR is in
4-phase mode, the master clock signal is distributed to the four
phases, and the Clock 1~4 signals will be 90° out-of-phase. If the
Core VR is in 3-phase mode, the master clock signal is
distributed to the three phases, and the Clock 1~3 signals will be
120° out-of-phase. If the Core VR is in 2-phase mode, the master
clock signal is distributed to Phases 1 and 2, and the Clock1 and
Clock2 signals will be 180° out-of-phase. If the Core VR is in
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10
FIGURE 6. R3™ MODULATOR OPERATION PRINCIPLES IN
STEADY STATE
Each slave circuit has its own ripple capacitor CRS, whose voltage
mimics the inductor ripple current. A gm amplifier converts the
inductor voltage into a current source to charge and discharge
CRS. The slave circuit turns on its PWM pulse upon receiving the
clock signal, and the current source charges CRS. When CRS
voltage VCRS hits VW, the slave circuit turns off the PWM pulse,
and the current source discharges CRS.
Since the controller works with VCRS, which are large amplitude
and noise-free synthesized signals, it achieves lower phase jitter
than conventional hysteretic mode and fixed PWM mode
controllers. Unlike conventional hysteretic mode converters, the
error amplifier allows the ISL95712 to maintain a 0.5% output
voltage accuracy.
Figure 7 shows the operation principles during load insertion
response. The COMP voltage rises during load insertion,
generating the master clock signal more quickly, so the PWM
pulses turn on earlier, increasing the effective switching
frequency. This allows for higher control loop bandwidth than
conventional fixed frequency PWM controllers. The VW voltage
rises as the COMP voltage rises, making the PWM pulses wider.
During load release response, the COMP voltage falls. It takes
the master clock circuit longer to generate the next master clock
signal so the PWM pulse is held off until needed. The VW voltage
falls as the COMP voltage falls, reducing the current PWM pulse
width. This kind of behavior gives the ISL95712 excellent
response speed.
The fact that all the phases share the same VW window voltage
also ensures excellent dynamic current balance among phases.
FN8566.1
November 2, 2015
ISL95712
Figure 9 shows the operation principle in diode emulation mode at
light load. The load gets incrementally lighter in each of the three
cases from top to bottom. The PWM on-time is determined by the
VW window size and therefore is the same, making the inductor
current triangle the same in each of the three cases. The ISL95712
clamps the ripple capacitor voltage VCRS in DE mode to make it
mimic the inductor current. It takes the COMP voltage longer to hit
VCRS, naturally stretching the switching period. The inductor
current triangles move farther apart, such that the inductor current
average value is equal to the load current. The reduced switching
frequency helps increase light-load efficiency.
VW
COMP
V CRM
MASTER
CLOCK
CLOCK1
PWM1
CCM/DCM BOUNDARY
VW
CLOCK2
PWM2
PWM
CLOCK3
V CRS
PWM3
IL
VW
VW
LIGHT DCM
V CRS
VCRS1
VCRS3
VCRS2
IL
FIGURE 7. R3™ MODULATOR OPERATION PRINCIPLES IN LOAD
INSERTION RESPONSE
VW
DEEP DCM
V CRS
Diode Emulation and Period Stretching
The ISL95712 can operate in Diode Emulation (DE) mode to
improve light-load efficiency. In DE mode, the low-side MOSFET
conducts when the current is flowing from source-to-drain and
does not allow reverse current, thus emulating a diode. Figure 8
shows when LGATE is on, the low-side MOSFET carries current,
creating negative voltage on the phase node due to the voltage
drop across the ON-resistance. The ISL95712 monitors the current
by monitoring the phase node voltage. It turns off LGATE when the
phase node voltage reaches zero to prevent the inductor current
from reversing the direction and creating unnecessary power loss.
PHASE
IL
FIGURE 9. PERIOD STRETCHING
Channel Configuration
Individual PWM channels of either VR can be disabled by
connecting the ISENx pin of the channel not required to +5V. For
example, placing the controller in a 3+1 configuration, requires
ISEN4 of the Core VR and ISEN2_NB and ISEN3_NB of the
Northbridge VR to be tied to +5V. This disables Channel 4 of the
Core VR and Channels 2 and 3 of the Northbridge VR. ISEN1_NB
must be tied through a 10kΩ resistor to GND to prevent this pin
from pulling high and disabling the channel. Similarly, if the Core
VR is set to single phase mode, ISEN4, ISEN3 and ISEN2 will be
tied to +5V while ISEN1 is tied to GND through a 10kΩ resistor.
Connecting ISEN1 or ISEN1_NB to +5V will disable the
corresponding VR output. This feature allows debugging of
individual VR outputs.
UG A TE
LG ATE
Power-On Reset
IL
FIGURE 8. DIODE EMULATION
If the load current is light enough, as Figure 8 shows, the inductor
current reaches and stays at zero before the next phase node
pulse, and the regulator is in Discontinuous Conduction Mode
(DCM). If the load current is heavy enough, the inductor current
will never reach 0A, and the regulator is in CCM, although the
controller is in DE mode.
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11
Before the controller has sufficient bias to guarantee proper
operation, the ISL95712 requires a +5V input supply tied to VDD
to exceed the VDD rising Power-On Reset (POR) threshold. Once
this threshold is reached or exceeded, the ISL95712 has enough
bias to check the state of the SVI inputs once ENABLE is taken
high. Hysteresis between the rising and the falling thresholds
assure the ISL95712 does not inadvertently turn off unless the
bias voltage drops substantially (see “Electrical Specifications”
on page 7). Note that VIN must be present for the controller to
drive the output voltage.
FN8566.1
November 2, 2015
ISL95712
1
2
3
4
5
6
7
8
VDD
SVC
SVD
VOTF
SVT
TELEMETRY TELEMETRY
ENABLE
PWROK
METAL_VID
VCORE/ VCORE_NB
V_SVI
PGOOD AND PGOOD_NB
Interval 1 to 2: ISL95712 waits to POR.
Interval 2 to 3: SVC and SVD are externally set to pre-Metal VID code.
Interval 3 to 4: ENABLE locks pre-Metal VID code. Both outputs soft-start to this level.
Interval 4 to 5: PGOOD signal goes HIGH, indicating proper operation.
Interval 5 to 6: PGOOD and PGOOD_NB high is detected and PWROK is taken high. The ISL95712 is prepared for SVI commands.
Interval 6 to 7: SVC and SVD data lines communicate change in VID code.
Interval 7 to 8: ISL95712 responds to VID-ON-THE-FLY code change and issues a VOTF for positive VID changes.
Post 8: Telemetry is clocked out of the ISL95712.
FIGURE 10. SVI INTERFACE TIMING DIAGRAM: TYPICAL PRE-PWROK METAL VID START-UP
Start-Up Timing
With VDD above the POR threshold, the controller start-up
sequence begins when ENABLE exceeds the logic high threshold.
Figure 11 shows the typical soft-start timing of the Core and
Northbridge VRs. Once the controller registers ENABLE as a high,
the controller checks the state of a few programming pins during
the typical 8ms delay prior to beginning soft-starting the Core
and Northbridge outputs. The pre-PWROK Metal VID is read from
the state of the SVC and SVD pins and programs the DAC, the
programming resistors on the COMP, COMP_NB and PROG pins
are read to configure switching frequency, slew rate and output
offsets. These programming resistors are discussed in
subsequent sections. The ISL95712 use a digital soft-start to
ramp up the DAC to the Metal VID level programmed. The
soft-start slew rate is programmed by the PROG resistor, which is
used to set the VID-on-the-fly slew rate as well. See the
“VID-on-the-Fly Slew Rate Selection” on page 17 for more details
on selecting the PROG resistor. PGOOD is asserted high at the
end of the soft-start ramp.
Diode Throttling
During the soft-start ramp-up, the ISL95712 operates in Diode
Throttling mode until the output has exceeded 400mV. In Diode
Throttling mode, the lower MOSFET is kept OFF so that the
MOSFET body diode conducts, similar to a standard buck
regulator.
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12
VDD
SLEW RATE
ENABLE
8ms
MetalVID VID COMMAND
VOLTAGE
DAC
PGOOD
PWROK
VIN
FIGURE 11. TYPICAL SOFT-START WAVEFORMS
Voltage Regulation and Load Line
Implementation
After the soft-start sequence, the ISL95712 regulates the output
voltages to the pre-PWROK metal VID programmed, see Table 6
on page 17. The ISL95712 controls the no-load output voltage to
an accuracy of ±0.5% over the range of 0.75V to 1.55V. A
differential amplifier allows voltage sensing for precise voltage
regulation at the microprocessor die.
FN8566.1
November 2, 2015
ISL95712
amplifier regulates the inverting and noninverting input voltages to
be equal as shown in Equation 4:
Rdroop
+
FB
Vdroop
VR LOCAL VO
“CATCH” RESISTOR
Idroop
+
E/A
COMP
-
VCC SENSE + V
VCC SENSE
-
SVC

+
VDAC
INTERNAL TO IC
DAC
SVD
RTN
X1
-
VSS
FIGURE 12. DIFFERENTIAL SENSING AND LOAD LINE
IMPLEMENTATION
As the load current increases from zero, the output voltage
droops from the VID programmed value by an amount
proportional to the load current, to achieve the load line. The
ISL95712 can sense the inductor current through the intrinsic DC
Resistance (DCR) of the inductors, as shown in Figures 13 and
14, or through resistors in series with the inductors, as shown in
Figure 25 on page 28. In both methods, capacitor Cn voltage
represents the total inductor current. An internal amplifier
converts Cn voltage into an internal current source, Isum, with the
gain set by resistor Ri, see Equation 1.
Rewriting Equation 4 and substituting Equation 3 gives Equation 5
the exact equation required for load line implementation.
Cisen
(EQ. 2)
When using inductor DCR current sensing, a single NTC element
is used to compensate the positive temperature coefficient of the
copper winding, thus sustaining the load line accuracy with
reduced cost.
Idroop flows through resistor Rdroop and creates a voltage drop as
shown in Equation 3.
(EQ. 3)
Vdroop is the droop voltage required to implement load line.
Changing Rdroop or scaling Idroop can change the load line slope.
Since Isum sets the overcurrent protection level, it is
recommended to first scale Isum based on OCP requirement,
then select an appropriate Rdroop value to obtain the desired
load line slope.
Differential Sensing
Figure 12 also shows the differential voltage sensing scheme.
VCCSENSE and VSSSENSE are the remote voltage sensing signals
from the processor die. A unity gain differential amplifier senses
the VSSSENSE voltage and adds it to the DAC output. The error
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13
ISEN1
Rdcr2
L2
PHASE2
Risen
PHASE1
Risen
Rpcb3
IL3
ISEN3
Cisen
Rdcr3
L3
PHASE3
Risen
Cisen
Rpcb4
IL4
ISEN4
ISEN2
Figure 12 shows the load line implementation. The ISL95712
drives a current source (Idroop) out of the FB pin, which is a ratio
of the Isum current, as described by Equation 2.
Rdcr4
L4
PHASE4
Risen
The Isum current is used for load line implementation, current
monitoring on the IMON pins and overcurrent protection.
V droop = R droop  I droop
(EQ. 5)
Phase Current Balancing
(EQ. 1)
5
5 V Cn
I droop = ---  I sum = ---  ----------Ri
4
4
(EQ. 4)
The VCCSENSE and VSSSENSE signals come from the processor die.
The feedback is open circuit in the absence of the processor. As
Figure 12 shows, it is recommended to add a “catch” resistor to
feed the VR local output voltage back to the compensator, and to
add another “catch” resistor to connect the VR local output ground
to the RTN pin. These resistors, typically 10Ω, provide voltage
feedback if the system is powered up without a processor installed.
VSSSENSE
“CATCH” RESISTOR
V Cn
I sum = ----------Ri
= V DAC + VSS SENSE
VCC SENSE – VSS SENSE = V DAC – R droop  I droop
SVID[7:0]
+
droop
Rpcb2
VO
IL2
Rdcr1
L1
Rpcb1
IL1
Cisen
FIGURE 13. CURRENT BALANCING CIRCUIT
The ISL95712 monitors individual phase average current by
monitoring the ISEN1, ISEN2, ISEN3 and ISEN4 voltages.
Figure 13 shows the recommended current balancing circuit for
DCR sensing. Each phase node voltage is averaged by a low-pass
filter consisting of Risen and Cisen, and is presented to the
corresponding ISEN pin. Risen should be routed to the inductor
phase-node pad in order to eliminate the effect of phase node
parasitic PCB DCR. Equations 6 through 9 give the ISEN pin
voltages:
V ISEN1 =  R dcr1 + R pcb1   I L1
(EQ. 6)
V ISEN2 =  R dcr2 + R pcb2   I L2
(EQ. 7)
V ISEN3 =  R dcr3 + R pcb3   I L3
(EQ. 8)
V ISEN4 =  R dcr4 + R pcb4   I L4
(EQ. 9)
Where Rdcr1, Rdcr2, Rdcr3 and Rdcr4 are inductor DCR; Rpcb1,
Rpcb2, Rpcb3 and Rpcb4 are parasitic PCB DCR between the
inductor output side pad and the output voltage rail; and IL1, IL2,
IL3 and IL4 are inductor average currents.
FN8566.1
November 2, 2015
ISL95712
The ISL95712 will adjust the phase pulse-width relative to the
other phases to make VISEN1 = VISEN2 = VISEN3 = VISEN4, thus to
achieve IL1 = IL2 = IL3 = IL4, when Rdcr1 = Rdcr2 = Rdcr3 = Rdcr4
and Rpcb1 = Rpcb2 = Rpcb3 = Rpcb4.
Using the same components for L1, L2, L3 and L4 provides a
good match of Rdcr1, Rdcr2, Rdcr3 and Rdcr4. Board layout
determines Rpcb1, Rpcb2, Rpcb3 and Rpcb4. It is recommended
to have a symmetrical layout for the power delivery path between
each inductor and the output voltage rail, such that
Rpcb1 = Rpcb2 = Rpcb3 = Rpcb4.
V4p
PHASE4
IS E N 4
C is e n
R d c r4
L4
IL 4
V 4n
(EQ. 14)
V 1n + V 2p + V 3n + V
4n
= V 1n + V 2n + V 3p + V 4n
(EQ. 15)
V 1n + V 2n + V 3p + V
4n
= V 1n + V 2n + V 3n + V 4p
(EQ. 16)
Rewriting Equation 14 gives Equation 17:
(EQ. 17)
V 2p – V 2n = V 3p – V 3n
R d c r3
L3
V3p
PHASE3
R is e n
C is e n
IL 3
R is e n
V 3p – V 3n = V 4p – V 4n
V 3n
V 1p – V 1n = V 2p – V 2n = V 3p – V 3n = V 4p – V 4n
R d c r2
L2
V2p
PHASE2
R is e n
C is e n
IL 2
R is e n
R pcb2
Vo
R d c r1
L1
IL 1
R is e n
R pcb1
V 1n
R is e n
R is e n
FIGURE 14. DIFFERENTIAL-SENSING CURRENT BALANCING CIRCUIT
Sometimes, it is difficult to implement symmetrical layout. For
the circuit shown in Figure 13, asymmetric layout causes
different Rpcb1, Rpcb2, Rpcb3 and Rpcb4 values, thus creating a
current imbalance. Figure 14 shows a differential sensing current
balancing circuit recommended for ISL95712. The current
sensing traces should be routed to the inductor pads so they only
pick up the inductor DCR voltage. Each ISEN pin sees the average
voltage of three sources: its own, phase inductor phase-node
pad, and the other two phase inductor output side pads.
Equations 10 through 13 give the ISEN pin voltages:
V ISEN1 = V 1p + V 2n + V 3n + V 4n
(EQ. 10)
V ISEN2 = V 1n + V 2p + V 3n + V 4n
(EQ. 11)
V ISEN3 = V 1n + V 2n + V 3p + V 4n
(EQ. 12)
V ISEN4 = V 1n + V 2n + V 3n + V 4p
(EQ. 13)
14
Therefore:
(EQ. 21)
Current balancing (IL1 = IL2 = IL3 = IL4) is achieved when
Rdcr1 = Rdcr2 = Rdcr3 = Rdcr4. Rpcb1, Rpcb2, Rpcb3 and Rpcb4
do not have any effect.
R is e n
V1p
(EQ. 20)
R dcr1  I L1 = R dcr2  I L2 = R dcr3  I L3 = R dcr4  I L4
V 2n
R is e n
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(EQ. 19)
Combining Equations 17 through 19 give:
R is e n
PHASE1
R is e n
C is e n
(EQ. 18)
Rewriting Equation 16 gives Equation 19:
R pcb3
R is e n
IS E N 1
= V 1n + V 2p + V 3n + V 4n
Rewriting Equation 15 gives Equation 18:
R is e n
IS E N 2
4n
V 1p – V 1n = V 2p – V 2n
R pcb4
R is e n
IN T E R N A L
T O IC
V 1p + V 2n + V 3n + V
R is e n
R is e n
IS E N 3
The ISL95712 will make VISEN1 = VISEN2 = VISEN3 = VISEN4 as
shown in Equations 14 and 16:
Since the slave ripple capacitor voltages mimic the inductor
currents, the R3™ modulator can naturally achieve excellent
current balancing during steady state and dynamic operations.
Figure 15 shows the current balancing performance of a
three-phase evaluation board with load transient of 12A/51A at
different rep rates. The inductor currents follow the load current
dynamic change with the output capacitors supplying the
difference. The inductor currents can track the load current well
at a low repetition rate, but cannot keep up when the repetition
rate gets into the hundred-kHz range, where it is out of the
control loop bandwidth. The controller achieves excellent current
balancing in all cases installed.
FN8566.1
November 2, 2015
ISL95712
REP RATE = 10kHz
Modes of Operation
TABLE 1. CORE VR MODES OF OPERATION
CONFIG.
ISEN4
ISEN3
ISEN2
To Power To Power To Power
4-phase
Stage
Stage
Stage
Core VR
Configuration
REP RATE = 25kHz
Tied to 5V To Power To Power
3-phase
Stage
Stage
Core VR
Configuration
Tied to 5V Tied to 5V To Power
2-phase
Stage
Core VR
Configuration
Tied to 5V Tied to 5V Tied to 5V
1-phase
Core VR
Configuration
PSI0_L
AND
PSI1_L
MODE
11
4-phase CCM
01
2-phase CCM
00
1-phase DE
11
3-phase CCM
01
2-phase CCM
00
1-phase DE
11
2-phase CCM
01
1-phase CCM
00
1-phase DE
11
1-phase CCM
01
1-phase CCM
00
1-phase DE
REP RATE = 50kHz
The Core VR can be configured for 4-, 3-, 2- or 1-phase operation.
Table 1 shows Core VR configurations and operational modes,
programmed by the ISEN4, ISEN3 and ISEN2 pin status and the
PSI0_L and PSI1_L commands via the SVI 2 interface. The SVI 2
interface description of these bits is outlined in Table 9.
The ISENx pins disable the channel which they are related to. For
example, to setup a 3-phase configuration the ISEN4 pin is tied to
5V. This disables Channel 4 of the controller on the Core side.
REP RATE = 100kHz
In a 3-phase configuration, the Core VR operates in 3-phase CCM,
with PSI0_L and PSI_L both high. If PSI0_L is taken low via the
SVI 2 interface, the Core VR sheds Phase 3. The Core VR then
operates 2-phase and remains in CCM. When both PSI0_L and
PSI1_L are taken low, the Core VR sheds Phase 2 and the Core
VR enters 1-phase Diode Emulation (DE) mode.
For 2-phase configurations, the Core VR operates in 2-phase CCM
with PSI0_L and PSI_L both high. If PSI0_L is taken low via the
SVI 2 interface, the Core VR sheds Phase 2 and the Core VR
operates in 1-phase and remains in CCM. When both PSI0_L and
PSI1_L are taken low, the Core VR operates in 1-phase DE mode.
REP RATE = 200kHz
In a 1-phase configuration, the Core VR operates in 1-phase CCM
and remains in this mode when PSI0_L is taken low. When both
PSI0_L and PSI1_L are taken low, the controller enters DE mode.
When the Core VR is taken into PSI1 mode, where both PSI0_L
and PSI1_L are taken low, the ISL95712 will shed any additional
phases in excess of Phase 1. If there is a VID change as well, the
regulator will then slew the output to the new VID level in CCM
mode. Once the output has reached the new VID level, the Core
VR is then placed into DE mode. The Core VR can be disabled
completely by connecting ISEN1 to +5V.
FIGURE 15. CURRENT BALANCING DURING DYNAMIC OPERATION.
CH1: IL1 , CH2: ILOAD, CH3: IL2, CH4: IL3
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15
FN8566.1
November 2, 2015
ISL95712
The ISL95712 Northbridge VR can be configured for 3-, 2-, or 1phase operation. Table 2 shows the Northbridge VR
configurations and operational modes, which are programmed
by the ISEN3_NB and ISEN2_NB pin status and the PSI0_L and
PSI1_L bits of the SVI 2 command.
TABLE 2. NORTHBRIDGE VR MODES OF OPERATION
CONFIG.
ISEN3_NB
To Power
3-phase
Stage
NB VR
Configuration
Tied to 5V
2-phase
NB VR
Configuration
Tied to 5V
1-phase
NB VR
Configuration
ISEN2_NB
To Power
Stage
To Power
Stage
Tied to 5V
PSI0_L AND
PSI1_L
MODE
11
2-phase CCM
01
1-phase CCM
00
1-phase DE
11
2-phase CCM
01
1-phase CCM
00
1-phase DE
11
1-phase CCM
01
1-phase CCM
00
1-phase DE
In a 1-phase configuration, the ISEN2_NB pin is tied to +5V. The
Northbridge VR operates in 1-phase CCM when both PSI0_L and
PSI1_L are high and continues in this mode when PSI0_L is
taken low. The controller enters 1-phase DE mode when both
PSI0_L and PSI1_L are low.
When the Northbridge VR is taken into PSI1 mode, where both
PSI0_L and PSI1_L are taken low, the ISL95712 will shed any
additional phases in excess of Phase 1. If there is a VID change
as well, the regulator will then slew the output to the new VID
level in CCM mode. Once the output has reached the new VID
level, the Northbridge VR is then placed into DE mode.
The Northbridge VR can be disabled completely by tying
ISEN1_NB to 5V.
not reached zero when the low-side MOSFET turns off, it will flow
through the low-side MOSFET body diode, causing the phase
node to have a larger voltage drop until it decays to zero. If the
inductor current has crossed zero and reversed the direction
when the low-side MOSFET turns off, it will flow through the
high-side MOSFET body diode, causing the phase node to have a
spike until it decays to zero. The controller continues monitoring
the phase voltage after turning off the low-side MOSFET. To
minimize the body diode-related loss, the controller also adjusts
the phase comparator threshold voltage accordingly in iterative
steps such that the low-side MOSFET body diode conducts for
approximately 40ns.
Resistor Configuration Options
The ISL95712 uses the COMP, COMP_NB and PROG pins to
configure some functionality within the IC. Resistors from these
pins to GND are read during the first portion of the soft-start
sequence. The following sections outline how to select the
resistor values for each of these pins to correctly program the
output voltage offset of each output, VID-on-the-fly slew rate and
switching frequency used for both VRs.
VR Offset Programming
A positive or negative offset is programmed for the Core VR using
a resistor to ground from the COMP pin and the Northbridge in a
similar manner from the COMP_NB pin. Table 3 provides the
resistor value to select the desired output voltage offset. The 1%
tolerance resistor value shown in Table 3 must be used to program
the corresponding Core or NB output voltage offset. The MIN and
MAX tolerance values provide margin to insure the 1% tolerance
resistor will be read correctly.
TABLE 3. COMP AND COMP_NB OUTPUT VOLTAGE OFFSET SELECTION
RESISTOR VALUE [kΩ]
MIN
1% TOLERANCE
MAX
TOLERANCE
VALUE
TOLERANCE
COMP_NB
COMP
VCORE OFFSET OFFSET
[mV]
[mV]
Dynamic Operation
3.96
4.02
4.07
-43.75
18.75
Core and Northbridge VRs behave the same during dynamic
operation. The controller responds to VID-on-the-fly changes by
slewing to the new voltage at the slew rate programmed, see
Table 4. During negative VID transitions, the output voltage
decays to the lower VID value at the slew rate determined by the
load.
7.76
7.87
7.98
-37.5
31.25
11.33
11.5
11.67
-31.25
43.76
16.65
16.9
17.15
-25
50
19.3
19.6
19.89
-18.75
37.5
24.53
24.9
25.27
-12.5
25
33.49
34.0
34.51
-6.25
12.5
40.58
41.2
41.81
6.25
0
Adaptive Body Diode Conduction Time
Reduction
51.52
52.3
53.08
18.75
18.75
72.10
73.2
74.29
31.25
31.25
In DCM, the controller turns off the low-side MOSFET when the
inductor current approaches zero. During on-time of the low-side
MOSFET, phase voltage is negative and the amount is the
MOSFET rDS(ON) voltage drop, which is proportional to the
inductor current. A phase comparator inside the controller
monitors the phase voltage during on-time of the low-side
MOSFET and compares it with a threshold to determine the zero
crossing point of the inductor current. If the inductor current has
93.87
95.3
96.72
43.76
43.76
119.19
121
112.81
50
50
151.69
154
156.31
37.5
37.5
179.27
182
184.73
25
25
206.85
210
213.15
12.5
12.5
0
0
The R3™ modulator intrinsically has voltage feed-forward. The
output voltage is insensitive to a fast slew rate input voltage
change.
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16
OPEN
FN8566.1
November 2, 2015
ISL95712
TABLE 4. PROG RESISTOR SELECTION
TABLE 5. SWITCHING FREQUENCY SELECTION
RESISTOR VALUE
[kΩ]
SLEW RATE FOR CORE AND
NORTHBRIDGE [mV/µs]
FREQUENCY
[kHz]
COMP_NB RANGE
[kΩ]
PROG RANGE
[kΩ]
4.02
20
300
57.6 to OPEN
7.87
15
19.1 to 41.2
or
154 to OPEN
11.5
12.5
350
4.02 to 41.2
16.9
10
19.1 to 41.2
or
154 to OPEN
19.6
20
400
57.6 to OPEN
24.9
15
5.62 to 16.9
or
57.6 to 121
34.0
12.5
450
4.02 to 41.2
41.2
10
5.62 to 16.9
or
57.6 to 121
52.3
20
73.2
15
95.3
12.5
121
10
154
20
182
15
210
12.5
OPEN
10
VID-on-the-Fly Slew Rate Selection
The PROG resistor is used to select the slew rate for VID changes
commanded by the processor. Once selected, the slew rate is
locked in during soft-start and is not adjustable during operation.
The lowest slew rate that can be selected is 10mV/µs, which is
above the minimum of 7.5mV/µs required by the SVI2
specification. The slew rate selected sets the slew rate for both
Core and Northbridge VRs. The controller does not allow for
independent selection of slew rate.
CCM Switching Frequency
The Core and Northbridge VR switching frequency is set by the
programming resistors on COMP_NB and PROG. When the
ISL95712 is in Continuous Conduction Mode (CCM), the
switching frequency is not absolutely constant due to the nature
of the R3™ modulator. As explained in “Multiphase R3™
Modulator” on page 10, the effective switching frequency
increases during load insertion and decreases during load
release to achieve fast response. Thus, the switching frequency is
relatively constant at steady state. Variation is expected when
the power stage condition, such as input voltage, output voltage,
load, etc. changes. The variation is usually less than 10% and
does not have any significant effect on output voltage ripple
magnitude. Table 5 defines the switching frequency based on the
resistor values used to program the COMP_NB and PROG pins.
Use the previous tables related to COMP_NB and PROG to
determine the correct resistor value in these ranges to program
the desired output offset and slew rate.
The controller monitors SVI commands to determine when to
enter power-saving mode, implement dynamic VID changes and
shut down individual outputs.
AMD Serial VID Interface 2.0
The on-board Serial VID Interface 2.0 (SVI 2) circuitry allows the
AMD processor to directly control the Core and Northbridge
voltage reference levels within the ISL95712. Once the PWROK
signal goes high, the IC begins monitoring the SVC and SVD pins
for instructions. The ISL95712 uses a Digital-to-Analog Converter
(DAC) to generate a reference voltage based on the decoded SVI
value. See Figure 10 on page 12 for a simple SVI interface timing
diagram.
Pre-PWROK Metal VID
Typical motherboard start-up begins with the controller decoding
the SVC and SVD inputs to determine the pre-PWROK Metal VID
setting (see Table 6). Once the ENABLE input exceeds the rising
threshold, the ISL95712 decodes and locks the decoded value
into an on-board hold register.
TABLE 6. PRE-PWROK METAL VID CODES
SVC
SVD
OUTPUT VOLTAGE (V)
0
0
1.1
0
1
1.0
1
0
0.9
1
1
0.8
Once the programming pins are read, the internal DAC circuitry
begins to ramp Core and Northbridge VRs to the decoded
pre-PWROK Metal VID output level. The digital soft-start circuitry
ramps the internal reference to the target gradually at a fixed
rate of approximately 5mV/µs until the output voltage reaches
~250mV and then at the programmed slew rate. The controlled
ramp of all output voltage planes reduces inrush current during
the soft-start interval. At the end of the soft-start interval, the
PGOOD and PGOOD_NB outputs transition high, indicating both
output planes are within regulation limits.
If the ENABLE input falls below the enable falling threshold, the
ISL95712 tri-states both outputs. PGOOD and PGOOD_NB are
pulled low with the loss of ENABLE. The Core and Northbridge VR
output voltages decay, based on output capacitance and load
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FN8566.1
November 2, 2015
ISL95712
leakage resistance. If bias to VDD falls below the POR level, the
The ISL95712 responds in the manner previously described.
Once VDD and ENABLE rise above their respective rising
thresholds, the internal DAC circuitry reacquires a pre-PWROK
metal VID code, and the controller soft-starts.
SVI Interface Active
Once the Core and Northbridge VRs have successfully soft-started
and PGOOD and PGOOD_NB signals transition high, PWROK can
be asserted externally to the ISL95712. Once PWROK is asserted
to the IC, SVI instructions can begin as the controller actively
monitors the SVI interface. Details of the SVI Bus protocol are
provided in the “AMD Serial VID Interface 2.0 (SVI2)
Specification”. See AMD publication #48022.
Once a VID change command is received, the ISL95712 decodes
the information to determine which VR is affected and the VID
target is determined by the byte combinations in Table 7. The
internal DAC circuitry steps the output voltage of the VR
commanded to the new VID level. During this time, one or more
of the VR outputs could be targeted. In the event either VR is
commanded to power-off by serial VID commands, the PGOOD
signal remains asserted.
If the PWROK input is deasserted, then the controller steps both
the Core and the Northbridge VRs back to the stored pre-PWROK
metal VID level in the holding register from initial soft-start. No
attempt is made to read the SVC and SVD inputs during this time.
If PWROK is reasserted, then the ISL95712 SVI interface waits
for instructions.
If ENABLE goes low during normal operation, all external
MOSFETs are tri-stated and both PGOOD and PGOOD_NB are
pulled low. This event clears the pre-PWROK metal VID code and
forces the controller to check SVC and SVD upon restart, storing
the pre-PWROK metal VID code found on restart.
A POR event on VCC during normal operation shuts down both
regulators, and both PGOOD outputs are pulled low. The
pre-PWROK metal VID code is not retained. Loss of VIN during
operation will typically cause the controller to enter a fault
condition on one or both outputs as the output voltage collapses.
The controller will shut down both Core and Northbridge VRs and
latch off. The pre-PWROK metal VID code is not retained during
the process of cycling ENABLE to reset the fault latch and restart
the controller.
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VID-on-the-Fly Transition
Once PWROK is high, the ISL95712 detects this flag and begins
monitoring the SVC and SVD pins for SVI instructions. The
microprocessor follows the protocol outlined in the following
sections to send instructions for VID-on-the-fly transitions. The
ISL95712 decodes the instruction and acknowledges the new
VID code. For VID codes higher than the current VID level, the
ISL95712 begins stepping the commanded VR outputs to the
new VID target at the fixed slew rate of 10mV/µs. Once the DAC
ramps to the new VID code, a VID-on-the-Fly Complete (VOTFC)
request is sent on the SVI lines.
When the VID codes are lower than the current VID level, the
ISL95712 checks the state of power state bits in the SVI
command. If power state bits are not active, the controller begins
stepping the regulator output to the new VID target. If the power
state bits are active, the controller allows the output voltage to
decay and slowly steps the DAC down with the natural decay of
the output. This allows the controller to quickly recover and move
to a high VID code if commanded. The controller issues a VOTFC
request on the SVI lines once the SVI command is decoded and
prior to reaching the final output voltage.
VOTFC requests do not take priority over telemetry per the AMD
SVI 2 specification.
SVI Data Communication Protocol
The SVI WIRE protocol is based on the I2C bus concept. Two wires
[serial clock (SVC) and serial data (SVD)], carry information
between the AMD processor (master) and VR controller (slave) on
the bus. The master initiates and terminates SVI transactions
and drives the clock, SVC, during a transaction. The AMD
processor is always the master and the voltage regulators are the
slaves. The slave receives the SVI transactions and acts
accordingly. Mobile SVI WIRE protocol timing is based on
high-speed mode I2C. See AMD publication #48022 for
additional details.
FN8566.1
November 2, 2015
ISL95712
TABLE 7. SERIAL VID CODES
SVID[7:0]
VOLTAGE (V)
SVID[6:0]
VOLTAGE (V)
SVID[6:0]
VOLTAGE (V)
SVID[6:0]
VOLTAGE (V)
0000_0000
1.55000
0010_0000
1.35000
0100_0000
1.15000
0110_0000
0.95000
0000_0001
1.54375
0010_0001
1.34375
0100_0001
1.14375
0110_0001
0.94375
0000_0010
1.53750
0010_0010
1.33750
0100_0010
1.13750
0110_0010
0.93750
0000_0011
1.53125
0010_0011
1.33125
0100_0011
1.13125
0110_0011
0.93125
0000_0100
1.52500
0010_0100
1.32500
0100_0100
1.12500
0110_0100
0.92500
0000_0101
1.51875
0010_0101
1.31875
0100_0101
1.11875
0110_0101
0.91875
0000_0110
1.51250
0010_0110
1.31250
0100_0110
1.11250
0110_0110
0.91250
0000_0111
1.50625
0010_0111
1.30625
0100_0111
1.10625
0110_0111
0.90625
0000_1000
1.50000
0010_1000
1.30000
0100_1000
1.10000
0110_1000
0.90000
0000_1001
1.49375
0010_1001
1.29375
0100_1001
1.09375
0110_1001
0.89375
0000_1010
1.48750
0010_1010
1.28750
0100_1010
1.08750
0110_1010
0.88750
0000_1011
1.48125
0010_1011
1.28125
0100_1011
1.08125
0110_1011
0.88125
0000_1100
1.47500
0010_1100
1.27500
0100_1100
1.07500
0110_1100
0.87500
0000_1101
1.46875
0010_1101
1.26875
0100_1101
1.06875
0110_1101
0.86875
0000_1110
1.46250
0010_1110
1.26250
0100_1110
1.06250
0110_1110
0.86250
0000_1111
1.45625
0010_1111
1.25625
0100_1111
1.05625
0110_1111
0.85625
0001_0000
1.45000
0011_0000
1.25000
0101_0000
1.05000
0111_0000
0.85000
0001_0001
1.44375
0011_0001
1.24375
0101_0001
1.04375
0111_0001
0.84375
0001_0010
1.43750
0011_0010
1.23750
0101_0010
1.03750
0111_0010
0.83750
0001_0011
1.43125
0011_0011
1.23125
0101_0011
1.03125
0111_0011
0.83125
0001_0100
1.42500
0011_0100
1.22500
0101_0100
1.02500
0111_0100
0.82500
0001_0101
1.41875
0011_0101
1.21875
0101_0101
1.01875
0111_0101
0.81875
0001_0110
1.41250
0011_0110
1.21250
0101_0110
1.01250
0111_0110
0.81250
0001_0111
1.40625
0011_0111
1.20625
0101_0111
1.00625
0111_0111
0.80625
0001_1000
1.40000
0011_1000
1.20000
0101_1000
1.00000
0111_1000
0.80000
0001_1001
1.39375
0011_1001
1.19375
0101_1001
0.99375
0111_1001
0.79375
0001_1010
1.38750
0011_1010
1.18750
0101_1010
0.98750
0111_1010
0.78750
0001_1011
1.38125
0011_1011
1.18125
0101_1011
0.98125
0111_1011
0.78125
0001_1100
1.37500
0011_1100
1.17500
0101_1100
0.97500
0111_1100
0.77500
0001_1101
1.36875
0011_1101
1.16875
0101_1101
0.96875
0111_1101
0.76875
0001_1110
1.36250
0011_1110
1.16250
0101_1110
0.96250
0111_1110
0.76250
0001_1111
1.35625
0011_1111
1.15625
0101_1111
0.95625
0111_1111
0.75625
1000_0000
0.75000
1010_0000
0.55000*
1100_0000
0.35000*
1110_0000
0.15000*
1000_0001
0.74375
1010_0001
0.54375*
1100_0001
0.34375*
1110_0001
0.14375*
1000_0010
0.73750
1010_0010
0.53750*
1100_0010
0.33750*
1110_0010
0.13750*
1000_0011
0.73125
1010_0011
0.53125*
1100_0011
0.33125*
1110_0011
0.13125*
1000_0100
0.72500
1010_0100
0.52500*
1100_0100
0.32500*
1110_0100
0.12500*
1000_0101
0.71875
1010_0101
0.51875*
1100_0101
0.31875*
1110_0101
0.11875*
1000_0110
0.71250
1010_0110
0.51250*
1100_0110
0.31250*
1110_0110
0.11250*
1000_0111
0.70625
1010_0111
0.50625*
1100_0111
0.30625*
1110_0111
0.10625*
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FN8566.1
November 2, 2015
ISL95712
TABLE 7. SERIAL VID CODES (Continued)
SVID[7:0]
VOLTAGE (V)
SVID[6:0]
VOLTAGE (V)
SVID[6:0]
VOLTAGE (V)
SVID[6:0]
VOLTAGE (V)
1000_1000
0.70000
1010_1000
0.50000*
1100_1000
0.30000*
1110_1000
0.10000*
1000_1001
0.69375
1010_1001
0.49375*
1100_1001
0.29375*
1110_1001
0.09375*
1000_1010
0.68750
1010_1010
0.48750*
1100_1010
0.28750*
1110_1010
0.08750*
1000_1011
0.68125
1010_1011
0.48125*
1100_1011
0.28125*
1110_1011
0.08125*
1000_1100
0.67500
1010_1100
0.47500*
1100_1100
0.27500*
1110_1100
0.07500*
1000_1101
0.66875
1010_1101
0.46875*
1100_1101
0.26875*
1110_1101
0.06875*
1000_1110
0.66250
1010_1110
0.46250*
1100_1110
0.26250*
1110_1110
0.06250*
1000_1111
0.65625
1010_1111
0.45625*
1100_1111
0.25625*
1110_1111
0.05625*
1001_0000
0.65000
1011_0000
0.45000*
1101_0000
0.25000*
1111_0000
0.05000*
1001_0001
0.64375
1011_0001
0.44375*
1101_0001
0.24375*
1111_0001
0.04375*
1001_0010
0.63750
1011_0010
0.43750*
1101_0010
0.23750*
1111_0010
0.03750*
1001_0011
0.63125
1011_0011
0.43125*
1101_0011
0.23125*
1111_0011
0.03125*
1001_0100
0.62500
1011_0100
0.42500*
1101_0100
0.22500*
1111_0100
0.02500*
1001_0101
0.61875
1011_0101
0.41875*
1101_0101
0.21875*
1111_0101
0.01875*
1001_0110
0.61250
1011_0110
0.41250*
1101_0110
0.21250*
1111_0110
0.01250*
1001_0111
0.60625
1011_0111
0.40625*
1101_0111*
0.20625*
1111_0111
0.00625*
1001_1000
0.60000*
1011_1000
0.40000*
1101_1000
0.20000*
1111_1000
OFF*
1001_1001
0.59375*
1011_1001
0.39375*
1101_1001
0.19375*
1111_1001
OFF*
1001_1010
0.58750*
1011_1010
0.38750*
1101_1010
0.18750*
1111_1010
OFF*
1001_1011
0.58125*
1011_1011
0.38125*
1101_1011
0.18125*
1111_1011
OFF*
1001_1100
0.57500*
1011_1100
0.37500*
1101_1100
0.17500*
1111_1100
OFF*
1001_1101
0.56875*
1011_1101
0.36875*
1101_1101
0.16875*
1111_1101
OFF*
1001_1110
0.56250*
1011_1110
0.36250*
1101_1110
0.16250*
1111_1110
OFF*
1001_1111
0.55625*
1011_1111
0.35625*
1101_1111
0.15625*
1111_1111
OFF*
NOTE: * Indicates a VID not required for AMD Family 10h processors. Loosened AMD requirements at these levels.
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FN8566.1
November 2, 2015
1
SVC
2
3
4
5
6
7
8
9
10
VID
Bit [0]
VID Bits [7:1]
11
12
13
14
15
16
17
18
19
PSI1_L
PSI0_L
ISL95712
20
21
22
23
24
25
26
27
SVD
START
ACK
ACK
ACK
FIGURE 16. SVD PACKET STRUCTURE
SVI Bus Protocol
Power States
The AMD processor bus protocol is similar to SMBus send byte
protocol for VID transactions. The AMD SVD packet structure is
shown in Figure 16. The description of each bit of the three bytes
that make up the SVI command are shown in Table 8. During a
transaction, the processor sends the start sequence followed by
each of the three bytes, which end with an optional acknowledge
bit. The ISL95712 does not drive the SVD line during the ACK bit.
Finally, the processor sends the stop sequence. After the
ISL95712 has detected the stop, it can then proceed with the
commanded action from the transaction.
SVI2 defines two power state indicator levels, see Tables 1, 2,
and 9. As processor current consumption is reduced, the power
state indicator level changes to improve VR efficiency under low
power conditions.
TABLE 8. SVD DATA PACKET
BITS
1:5
DESCRIPTION
Always 11000b
6
Core domain selector bit, if set then the following data byte
contains VID, power state, telemetry control, load line trim
and offset trim apply to the Core VR.
7
Northbridge domain selector bit, if set then the following data
byte contains VID, power state, telemetry control, load line
trim and offset trim apply to the Northbridge VR.
8
Always 0b
9
Acknowledge bit
10
PSI0_L
11:17
VID code bits [7:1]
18
Acknowledge bit
19
VID code bit [0]
20
PSI1_L
21
TFN (Telemetry Functionality)
22:24
Load line slope trim
25:26
Offset Trim [1:0]
27
Acknowledge bit
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For the Core VR operating in 4-phase mode (when PSI0_L is
asserted) Channels 3 and 4 are tri-stated. The controller
continues to operate in 2-phase CCM. The shedding of phases
improves the efficiency of the VR at the light to moderate load
levels of the CPU in this power state. When PSI1_L is asserted
the Core VR sheds Channel 2. If there is a corresponding VID
change, then the output is moved to the new VID level while in
single phase DE mode. Once the output is at the proper VID level,
Channel 1 enters diode emulation mode to further boost
light-load efficiency in this power state.
For the Northbridge VR operating in 3-phase mode, when PSI0_L
is asserted, Channels 2 and 3 are tri-stated while Channel 1
continues in continuous conduction mode. When PSI1_L is
asserted, the output is moved to the new VID level if one is
commanded and Channel 1 then enters diode emulation mode
to conserve power.
It is possible for the processor to assert or deassert PSI0_L and
PSI1_L out of order. PSI0_L takes priority over PSI1_L. If PSI0_L
is deasserted while PSI1_L is still asserted, the ISL95712 will
return the selected VR back full channel CCM operation. For
example, if the Core VR is configured for 4-Phase operation and
both PSI0_L and PSI1_L are asserted low during a command, the
VR will shed three phases and operate in 1-Phase DE mode. If an
SVI command follows which takes PSI0_L high, but leaves
PSI1_L low, the VR will exit power savings mode and being
operation in 4-Phase CCM mode.
TABLE 9. PSI0_L AND PSI1_L DEFINITION
FUNCTION
BIT
DESCRIPTION
PSI0_L
10
Power State Indicate level 0. When this signal is
asserted (active Low), the processor is in a low
enough power state for the ISL95712 to take action
to boost efficiency by dropping phases.
PSI1_L
20
Power State Indicate level 1. When this signal is
asserted (active Low), the processor is in a low
enough power state for the ISL95712 to take action
to boost efficiency by dropping phases and entering
1-Phase DE.
FN8566.1
November 2, 2015
ISL95712
Dynamic Load Line Slope Trim
Telemetry
The ISL95712 supports the SVI2 ability for the processor to
manipulate the load line slope of the Core and Northbridge VRs
independently using the serial VID interface. The slope
manipulation applies to the initial load line slope. A load line
slope trim will typically coincide with a VOTF change. See
Table 10 for more information about the load line slope trim
feature of the ISL95712. The Disable LL selection is not
recommended unless operation without a LL is required and
considered during the compensation of the VR.
The ISL95712 can provide voltage and current information to the
AMD CPU through the telemetry system outlined by the AMD
SVI2 specification. The telemetry data is transmitted through the
SVC and SVT lines of the SVI2 interface.
TABLE 10. LOAD LINE SLOPE TRIM DEFINITION
LOAD LINE SLOPE TRIM [2:0]
DESCRIPTION
000
Disable LL
001
-40% mΩ Change
010
-20% mΩ Change
011
No Change
100
+20% mΩ Change
101
+40% mΩ Change
110
+60% mΩ Change
111
+80% mΩ Change
Dynamic Offset Trim
The ISL95712 supports the SVI2 ability for the processor to
manipulate the output voltage offset of the Core and Northbridge
VRs. This offset is in addition to any output voltage offset set via
the COMP resistor reader. The dynamic offset trim can disable
the COMP resistor programmed offset of either output when
Disable All Offset is selected.
Current telemetry is based on a voltage generated across a
133kΩ resistor placed from the IMON pin to GND. The current
flowing out of the IMON pin is proportional to the load current in
the VR. The Isum current defined in “Voltage Regulation and Load
Line Implementation” on page 12, provides the base conversion
from the load current to the internal amplifier created Isum
current. The Isum current is then divided down by a factor of 4 to
create the IMON current, which flows out of the IMON pin. The
Isum current will measure 36µA when the load current is at full
load based on a droop current designed for 45µA at the same
load current. The difference between the Isum current and the
droop current is provided in Equation 2. The IMON current will
measure 11.25µA at full load current for the VR and the IMON
voltage will be 1.2V. The load percentage, which is reported by
the IC is based on the this voltage. When the load is 25% of the
full load, the voltage on the IMON pin will be 25% of 1.2V or 0.3V.
The SVI interface allows the selection of no telemetry, voltage
only, or voltage and current telemetry on either or both of the VR
outputs. The TFN bit along with the Core and Northbridge domain
selector bits are used by the processor to change the
functionality of telemetry, see Table 12 for more information.
TABLE 12. TFN TRUTH TABLE
TFN, CORE, NB
BITS [21, 6, 7]
1,0,1
Telemetry is in voltage and current mode. Therefore,
voltage and current are sent for VDD and VDDNB
domains by the controller.
1,0,0
Telemetry is in voltage mode only. Only the voltage of
VDD and VDDNB domains is sent by the controller.
1,1,0
Telemetry is disabled.
1,1,1
Reserved
TABLE 11. OFFSET TRIM DEFINITION
OFFSET TRIM [1:0]
DESCRIPTION
00
Disable All Offset
01
-25mV Change
10
0mV Change
11
+25mV Change
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DESCRIPTION
FN8566.1
November 2, 2015
ISL95712
PMBus Interface
The ISL95712 includes a PMBus interface, which allows for user
programmability of numerous operating parameters and for
monitoring various parameters of the Core and NB regulators.
The PMBus address for the ISL95712 is 1001111.
TABLE 13. PMBus READ AND WRITE REGISTERS
COMMAND
CODE
ACCESS
DEFAULT
COMMAND NAME
9Bh
R
01h
MANUFACTURER REVISION
DESCRIPTION
Silicon revision starts at 01h
D0h
Reserved
D1h
Reserved
D2h
R/W
00h
FAULT_STATUS_2
BIT VALUE
BIT
0
1
5 (Read Only)
ISL95712 Enabled
ISL95712 Fault
Disabled
4
No Fault
Core OV
3
NB OV
2
Core OCP
1
NB OCP
0
CML. Indicates that an
unsupported command
is received or a write
command to a read-only
register or PEC does not
match
D3h
R
xxh
READ_VOUT_CORE
Read the Core Voltage in ADC format. Each LSB is 6.25mV
D4h
R
xxh
READ_IOUT_CORE
Read Core Current in ADC format. FFh = 100% (7.5µA on IMON)
D5h
Reserved
D6h
R
xxh
READ_VOUT_NB
Read the NB voltage in ADC format. Each LSB is 6.25mV
D7h
R
xxh
READ_IOUT_NB
Read NB load current in ADC format. FFh = 100% (7.5µA on IMON)
D8h
Reserved
D9h
Reserved
DAh
Reserved
DBh
Reserved
DCh
Reserved
DDh
Reserved
DEh
R/W
00h
LOCK_SVID
BIT[0] VALUE
FUNCTIONALITY
0
Execute SVI2 Commands. PMBus commands DFh
through E4h are not executed. These registers can
still be read and written to.
1
Execute PMBus commands DFh through E4h
while ignoring SVI2 commands.
DFh
R/W
08h
SET_VID_CORE
Set Core VID, default set to 800mV. Each LSB is 6.25mV. Metal VID
level is determined by SVC/SVD logic levels at power-up.
E0h
R/W
00h
OFFSET_CORE
Set Core offset. The offset range is from -250mV to +200mV. This is a
2’s complement number. Bit[7] is the sign bit.
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November 2, 2015
ISL95712
TABLE 13. PMBus READ AND WRITE REGISTERS (Continued)
COMMAND
CODE
ACCESS
DEFAULT
COMMAND NAME
E1h
R/W
0fh
LOADLINE_PWRSTATE_CORE
DESCRIPTION
BIT
FUNCTIONALITY
[4:2]
Load line slope trim. Refer to Table 10 for proper
usage.
1
Sets PSI0 power state. Refer to Table 9 for proper
usage.
0
Sets PSI1 power state. Refer to Table 9 for proper
usage.
E2h
R/W
08h
SET_VID_NB
Set NB VID, default set to 800mV. Each LSB is 6.25mV. Metal VID level
is determined by SVC/SVD logic levels at power-up.
E3h
R/W
00h
OFFSET_NB
Set NB offset. The offset range is from -250mV to +200mV. This is a
2’s complement number. Bit[7] is the sign bit.
E4h
R/W
0fh
LOADLINE_PWRSTATE_NB
Protection Features
Core VR and Northbridge VR both provide overcurrent,
current-balance, undervoltage and overvoltage fault protections.
The controller also provides over-temperature protection. The
following discussion is based on Core VR and also applies to the
Northbridge VR.
Overcurrent
The IMON voltage provides a means of determining the load
current at any moment in time. The Overcurrent Protection (OCP)
circuitry monitors the IMON voltage to determine when a fault
occurs. Based on the previous description in “Voltage Regulation
and Load Line Implementation” on page 12, the current which
flows out of the IMON pin is proportional to the Isum current. The
Isum current is created from the sensed voltage across Cn, which is
a measure of the load current based upon the sensing element
selected. The IMON current is generated internally and is 1/4 of
the Isum current. The EDC or IDDspike current value for the AMD
CPU load is used to set the maximum current level for droop and
the IMON voltage of 1.2V, which indicates 100% loading for
telemetry. The Isum current level at maximum load, or IDDspike, is
36µA and this translates to an IMON current level of 9µA. The
IMON resistor is 133kΩ and the 9µA flowing through the IMON
resistor results in a 1.2V level at maximum loading of the VR.
The overcurrent threshold is 1.5V on the IMON pin. Based on a
1.2V IMON voltage equating to 100% loading, the additional 0.3V
provided above this level equates to a 25% increase in load current
before an OCP fault is detected. The EDC or IDDspike current is
used to set the 1.2V on IMON for full load current. Thus the OCP
level is 1.25 times the EDC or IDDspike current level. This
additional margin above the EDC or IDDspike current allows the
AMD CPU to enter and exit the IDDspike performance mode
without issue unless the load current is out of line with the
IDDspike expectation, thus the need for overcurrent protection.
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BIT
FUNCTIONALITY
[4:2]
Load line slope trim. Refer to Table 10 for proper
usage.
1
Sets PSI0 power state. Refer to Table 9 for proper
usage.
0
Sets PSI1 power state. Refer to Table 9 for proper
usage.
When the voltage on the IMON pin meets the overcurrent
threshold of 1.5V, this triggers an OCP event. Within 2µs of
detecting an OCP event, the controller asserts VR_HOT_L low to
communicate to the AMD CPU to throttle back. A fault timer
begins counting while IMON is at or above the 1.5V threshold. The
fault timer lasts 7.5µs to 11µs and then the controller takes action
by tri-stating the active channels. This provides the CPU time to
recover and reduce the load current. If the OCP conditions are
relieved, then the fault timer is cleared and VR_HOT_L is taken
high clearing the fault condition. If the load current is not reduced
and the OCP condition is maintained, the output voltage will fall
below the undervoltage threshold due to the lack of switching or a
way-overcurrent fault could occur. Either of these fault conditions
will cause the controller to drop PGOOD of that output. When
PGOOD is taken low, a fault flag from this VR is sent to the other
VR and it is shut down within 10µs and PGOOD of the other output
is taken low.
The ISL95712 also features a Way-Overcurrent [WOC] feature,
which immediately takes the controller into shutdown. This
protection is also referred to as fast overcurrent protection for
short-circuit protection. If the IMON current reaches 15µA, WOC is
triggered. Active channels are tri-stated and the controller is
placed in shutdown and PGOOD is pulled low. There is no fault
timer on the WOC fault, the controller takes immediate action. The
other controller output is also shut down within 10µs.
Current-Balance
The controller monitors the ISENx pin voltages to determine
current-balance protection. If the ISENx pin voltage difference is
greater than 9mV for 1ms, the controller will declare a fault and
latch off.
FN8566.1
November 2, 2015
ISL95712
Undervoltage
If the VSEN voltage falls below the output voltage VID value plus
any programmed offsets by -325mV, the controller declares an
undervoltage fault. The controller deasserts PGOOD and
tri-states the power MOSFETs.
INTERNAL TO
ISL95712
+V
30µA
Overvoltage
If the VSEN voltage exceeds the output voltage VID value plus any
programmed offsets by +325mV, the controller declares an
overvoltage fault. The controller deasserts PGOOD and turns on
the low-side power MOSFETs. The low-side power MOSFETs
remain on until the output voltage is pulled down below the VID
set value. Once the output voltage is below this level, the lower
gate is tri-stated. If the output voltage rises above the overvoltage
threshold again, the protection process is repeated. This behavior
provides the maximum amount of protection against shorted
high-side power MOSFETs while preventing output ringing below
ground.
Thermal Monitor [NTC, NTC_NB]
The ISL95712 features two thermal monitors using an external
resistor network, which includes an NTC thermistor to monitor
motherboard temperature and alert the AMD CPU of a thermal
issue. Figure 17 shows the basic thermal monitor circuit on the
Core VR NTC pin. The Northbridge VR features the same thermal
monitor. The controller drives a 30µA current out of the NTC pin
and monitors the voltage at the pin. The current flowing out of
the NTC pin creates a voltage that is compared to a warning
threshold of 640mV. When the voltage at the NTC pin falls to this
warning threshold or below, the controller asserts VR_HOT_L to
alert the AMD CPU to throttle back load current to stabilize the
motherboard temperature. A thermal fault counter begins
counting toward a minimum shutdown time of 100µs. The
thermal fault counter is an up/down counter, so if the voltage at
the NTC pin rises above the warning threshold, it will count down
and extend the time for a thermal fault to occur. The warning
threshold does have 20mV of hysteresis.
If the voltage at the NTC pin continues to fall down to the
shutdown threshold of 580mV or below, the controller goes into
shutdown and triggers a thermal fault. The PGOOD pin is pulled
low and tri-states the power MOSFETs. A fault on either side will
shutdown both VRs.
Rp
MONITOR
+
VNTC
-
RNTC
WARNING SHUTDOWN
640mV
580mV
Rs
FIGURE 17. CIRCUITRY ASSOCIATED WITH THE THERMAL MONITOR
FEATURE OF THE ISL95712
As the board temperature rises, the NTC thermistor resistance
decreases and the voltage at the NTC pin drops. When the
voltage on the NTC pin drops below the over-temperature trip
threshold, then VR_HOT is pulled low. The VR_HOT signal is used
to change the CPU operation and decrease power consumption.
With the reduction in power consumption by the CPU, the board
temperature decreases and the NTC thermistor voltage rises.
Once the over-temperature threshold is tripped and VR_HOT is
taken low, the over-temperature threshold changes to the reset
level. The addition of hysteresis to the over-temperature
threshold prevents nuisance trips. Once both pin voltages exceed
the over-temperature reset threshold, the pull-down on VR_HOT
is released. The signal changes state and the CPU resumes
normal operation. The over-temperature threshold returns to the
trip level.
Table 14 summarizes the fault protections.
TABLE 14. FAULT PROTECTION SUMMARY
FAULT TYPE
Overcurrent
Phase Current
Unbalance
Undervoltage
-325mV
FAULT DURATION
BEFORE
PROTECTION
NTC Thermal
PROTECTION ACTION
FAULT
RESET
7.5µs to 11.5µs PWM tri-state
1ms
PWM tri-state, PGOOD
latched low
Immediately
Overvoltage
+325mV
25
R
NTC
Way-Overcurrent
(1.5xOC)
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VR_HOT_L
PGOOD latched low.
PWM tri-state.
PGOOD latched low.
Actively pulls the output
voltage to below VID
value, then tri-state.
100µs min
ENABLE
toggle or
VDD
toggle
PGOOD latched low.
PWM tri-state.
FN8566.1
November 2, 2015
ISL95712
Fault Recovery
All of the previously described fault conditions can be reset by
bringing ENABLE low or by bringing VDD below the POR
threshold. When ENABLE and VDD return to their high operating
levels, the controller resets the faults and soft-start occurs.
Interface Pin Protection
The SVC and SVD pins feature protection diodes, which must be
considered when removing power to VDD and VDDIO, but leaving
it applied to these pins. Figure 18 shows the basic protection on
the pins. If SVC and/or SVD are powered but VDD is not, leakage
current will flow from these pins to VDD.
VDD
SVC, SVD
GND
FIGURE 18. PROTECTION DEVICES ON THE SVC AND SVD PINS
Key Component Selection
Inductor DCR Current-Sensing Network
RSUM
RSUM
RSUM
ISUM+
RSUM
L
RNTCS
L
+
RP
DCR
DCR
DCR
CNVCN
-
RNTC
DCR
RO
RI
RO
ISUM-
RO
RO
IO
FIGURE 19. DCR CURRENT-SENSING NETWORK
Figure 19 shows the inductor DCR current-sensing network for a
4-phase solution. An inductor current flows through the DCR and
creates a voltage drop. Each inductor has two resistors in Rsum
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26


R ntcnet

DCR
-----------------------------------------------------V Cn  s  = 

  I  s   A cs  s 
R sum
N  o

 R ntcnet + -------------
N
(EQ. 22)
 R ntcs + R ntc   R p
R ntcnet = ---------------------------------------------------R ntcs + R ntc + R p
(EQ. 23)
s
1 + ------L
A cs  s  = ----------------------s
1 + ------------ sns
(EQ. 24)
DCR
 L = ------------L
(EQ. 25)
1
 sns = -------------------------------------------------------R sum
R ntcnet  --------------N
------------------------------------------  C n
R sum
R ntcnet + --------------N
PHASE1 PHASE2 PHASE3 PHASE4
L
The inductor output side pads are electrically shorted in the
schematic but have some parasitic impedance in actual board
layout, which is why one cannot simply short them together for
the current-sensing summing network. It is recommended to use
1Ω~10ΩRo to create quality signals. Since Ro value is much
smaller than the rest of the current sensing circuit, the following
analysis ignores it.
The summed inductor current information is presented to the
capacitor Cn. Equations 22 through 26 describe the frequency
domain relationship between inductor total current Io(s) and Cn
voltage VCn(s):
INTERNAL TO
ISL95712
L
and Ro connected to the pads to accurately sense the inductor
current by sensing the DCR voltage drop. The Rsum and Ro
resistors are connected in a summing network as shown and
feed the total current information to the NTC network (consisting
of Rntcs, Rntc and Rp) and capacitor Cn. Rntc is a negative
temperature coefficient (NTC) thermistor, used to temperature
compensate the inductor DCR change.
(EQ. 26)
Where N is the number of phases.
Transfer function Acs(s) always has unity gain at DC. The inductor
DCR value increases as the winding temperature increases,
giving higher reading of the inductor DC current. The NTC Rntc
value decrease as its temperature decreases. Proper selection of
Rsum, Rntcs, Rp and Rntc parameters ensures that VCn
represents the inductor total DC current over the temperature
range of interest.
There are many sets of parameters that can properly
temperature-compensate the DCR change. Since the NTC
network and the Rsum resistors form a voltage divider, Vcn is
always a fraction of the inductor DCR voltage. It is recommended
to have a higher ratio of Vcn to the inductor DCR voltage so the
droop circuit has a higher signal level to work with.
A typical set of parameters that provide good temperature
compensation are: Rsum = 3.65kΩ, Rp = 11kΩ, Rntcs = 2.61kΩ
and Rntc = 10kΩ (ERT-J1VR103J). The NTC network parameters
may need to be fine tuned on actual boards. One can apply full
FN8566.1
November 2, 2015
ISL95712
load DC current and record the output voltage reading
immediately; then record the output voltage reading again when
the board has reached the thermal steady state. A good NTC
network can limit the output voltage drift to within 2mV. It is
recommended to follow the Intersil evaluation board layout and
current sensing network parameters to minimize engineering time.
VCn(s) also needs to represent real-time Io(s) for the controller to
achieve good transient response. Transfer function Acs(s) has a
pole sns and a zero wL. One needs to match L and sns so
Acs(s) is unity gain at all frequencies. By forcing L equal to sns
and solving for the solution, Equation 27 gives Cn value.
io
Vo
FIGURE 22. LOAD TRANSIENT RESPONSE WHEN Cn IS TOO LARGE
io
L
C n = --------------------------------------------------------------R sum
R ntcnet  --------------N ---------------------------------------- DCR
R sum
R ntcnet + --------------N
(EQ. 27)
iL
Vo
For example, given N = 4, Rsum = 3.65kΩ, Rp = 11kΩ,
Rntcs = 2.61kΩ, Rntc = 10kΩ, DCR = 0.88mΩ and L = 0.36µH,
Equation 27 gives Cn = 0.518µF.
Assuming the compensator design is correct, Figure 20 shows the
expected load transient response waveforms if Cn is correctly
selected. When the load current Icore has a square change, the
output voltage Vcore also has a square response.
If Cn value is too large or too small, VCn(s) does not accurately
represent real-time Io(s) and worsens the transient response.
Figure 21 shows the load transient response when Cn is too
small. Vcore sags excessively upon load insertion and may create
a system failure. Figure 22 shows the transient response when
Cn is too large. Vcore is sluggish in drooping to its final value.
There is excessive overshoot if load insertion occurs during this
time, which may negatively affect the CPU reliability.
io
RING
BACK
FIGURE 23. OUTPUT VOLTAGE RING-BACK PROBLEM
ISUM+
R ntcs
C n.2
Rp
Rntc
+
Cn.1
Rn
OPTIONAL
Vcn
-
ISUM-
Ri
R ip
C ip
OPTIONAL
Vo
FIGURE 20. DESIRED LOAD TRANSIENT RESPONSE WAVEFORMS
io
Vo
FIGURE 21. LOAD TRANSIENT RESPONSE WHEN Cn IS TOO SMALL
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27
FIGURE 24. OPTIONAL CIRCUITS FOR RING-BACK REDUCTION
Figure 23 shows the output voltage ring-back problem during load
transient response. The load current io has a fast step change, but
the inductor current iL cannot accurately follow. Instead, iL
responds in first-order system fashion due to the nature of the
current loop. The ESR and ESL effect of the output capacitors
makes the output voltage Vo dip quickly upon load current change.
However, the controller regulates Vo according to the droop current
idroop, which is a real-time representation of iL; therefore, it pulls
Vo back to the level dictated by iL, causing the ring-back problem.
This phenomenon is not observed when the output capacitor has
very low ESR and ESL, as is the case with all ceramic capacitors.
Figure 24 shows two optional circuits for reduction of the
ring-back. Cn is the capacitor used to match the inductor time
constant. It usually takes the parallel of two (or more) capacitors
to get the desired value. Figure 24 shows that two capacitors
(Cn.1 and Cn.2) are in parallel. Resistor Rn is an optional
component to reduce the Vo ring-back. At steady state, Cn.1 +
Cn.2 provides the desired Cn capacitance. At the beginning of io
FN8566.1
November 2, 2015
ISL95712
change, the effective capacitance is less because Rn increases
the impedance of the Cn.1 branch. As Figure 21 shows, Vo tends
to dip when Cn is too small, and this effect reduces the Vo
ring-back. This effect is more pronounced when Cn.1 is much
larger than Cn.2. It is also more pronounced when Rn is bigger.
However, the presence of Rn increases the ripple of the Vn signal
if Cn.2 is too small. It is recommended to keep Cn.2 greater than
2200pF. Rn value usually is a few ohms. Cn.1, Cn.2 and Rn values
should be determined through tuning the load transient response
waveforms on an actual board.
Rip and Cip form an R-C branch in parallel with Ri, providing a
lower impedance path than Ri at the beginning of io change. Rip
and Cip do not have any effect at steady state. Through proper
selection of Rip and Cip values, idroop can resemble io rather than
iL, and Vo will not ring back. The recommended value for Rip is
100Ω. Cip should be determined through tuning the load
transient response waveforms on an actual board. The
recommended range for Cip is 100pF~2000pF. However, it
should be noted that the Rip - Cip branch may distort the idroop
waveform. Instead of being triangular as the real inductor
current, idroop may have sharp spikes, which may adversely
affect idroop average value detection and therefore may affect
OCP accuracy. User discretion is advised.
Resistor Current-Sensing Network
PHASE1 PHASE2 PHASE3 PHASE4
L
DCR
L
L
DCR
DCR
RSUM
ISUM+
RSUM
RSEN
RSEN
+
-
VCN
RO
CN
Ri
RO
1
 Rsen = ----------------------------R sum
---------------  C n
N
(EQ. 30)
Transfer function ARsen(s) always has unity gain at DC.
Current-sensing resistor Rsen value does not have significant
variation over-temperature, so there is no need for the NTC
network.
The recommended values are Rsum = 1kΩ and Cn = 5600pF.
Overcurrent Protection
Refer to Equation 2 on page 13 and Figures 19 and 25; resistor
Ri sets the Isum current, which is proportional to droop current
and IMON current. The OCP threshold is 1.5V on the IMON pin,
which equates to an IMON current of 11.25µA using a 133kΩ
IMON resistor. The corresponding Isum is 45µA, which results in
an Idroop of 56.25µA. At full load current, Iomax , the Isum current
is 36µA and the resulting Idroop is 45µA. The ratio of Isum at OCP
relative to full load current is 1.25. Therefore, the OCP current
trip level is 25% higher than the full load current.
(EQ. 31)
R ntcnet
DCR
5 1
I droop = ---  -----  ------------------------------------------  -------------  I o
N
R sum
4 Ri
R ntcnet + --------------N
(EQ. 32)
Therefore:
ISUM-
RO
RO
R ntcnet  DCR  I o
5
R i = ---  ---------------------------------------------------------------------------------R sum
4
N   R ntcnet + ---------------  I droop

N 
(EQ. 33)
Substitution of Equation 23 and application of the OCP condition
in Equation 33 gives Equation 34:
IO
FIGURE 25. RESISTOR CURRENT-SENSING NETWORK
Figure 25 shows the resistor current-sensing network for a
4-phase solution. Each inductor has a series current sensing
resistor, Rsen. Rsum and Ro are connected to the Rsen pads to
accurately capture the inductor current information. The Rsum
and Ro resistors are connected to capacitor Cn. Rsum and Cn
form a filter for noise attenuation. Equations 28 through 30 give
the VCn(s) expression.
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(EQ. 29)
Substitution of Equation 31 into Equation 2 gives Equation 32:
RSUM
RSEN
1
A Rsen  s  = ----------------------s
1 + ------------ sns


R ntcnet

DCR
V Cn =  ------------------------------------------  -------------  I o
R sum
N 

 R ntcnet + -------------
N
RSUM
RSEN
(EQ. 28)
For inductor DCR sensing, Equation 31 gives the DC relationship
of Vcn(s) and Io(s):
L
DCR
R sen
V Cn  s  = -------------  I o  s   A Rsen  s 
N
28
 R ntcs + R ntc   R p
----------------------------------------------------  DCR  I omax
R ntcs + R ntc + R p
5
R i = ---  ----------------------------------------------------------------------------------------------------------------------------4
  R ntcs + R ntc   R p R sum
N   ---------------------------------------------------- + ---------------  I droopmax
N 
 R ntcs + R ntc + R p
(EQ. 34)
Where Iomax is the full load current and Idroopmax is the
corresponding droop current. For example, given N = 4,
Rsum = 3.65kΩ, Rp = 11kΩ, Rntcs = 2.61kΩ, Rntc = 10kΩ,
DCR = 0.88mΩ, Iomax = 100A and Idroopmax = 45μA.
Equation 34 gives Ri = 529Ω.
FN8566.1
November 2, 2015
ISL95712
For resistor sensing, Equation 35 gives the DC relationship of
Vcn(s) and Io(s).
R sen
V Cn = -------------  I o
N
(EQ. 35)
source (= VID) and output impedance Zout(s). If Zout(s) is equal to
the load line slope LL, i.e., a constant output impedance, then in
the entire frequency range, Vo will have a square response when
Io has a square change.
Substitution of Equation 35 into Equation 2 gives Equation 36:
5 1 R sen
I droop = ---  -----  -------------  I o
N
4 Ri
Zout(s) = LL
(EQ. 36)
VR
VID
Therefore:
5 R sen  I o
R i = ---  --------------------------4 N  I droop
LOAD
Vo
(EQ. 37)
Substitution of Equation 37 and application of the OCP condition
in Equation 33 gives Equation 38:
5 R sen  I omax
R i = ---  -------------------------------------4 N  I droopmax
(EQ. 38)
Where Iomax is the full load current and Idroopmax is the
corresponding droop current. For example, given N = 4,
Rsen = 1mΩ, Iomax = 100A and Idroopmax = 45µA, Equation 38
gives Ri = 694Ω.
Load Line Slope
See Figure 12 for load line implementation.
For inductor DCR sensing, substitution of Equation 32 into
Equation 3 gives the load line slope expression:
V droop
R ntcnet
5 R droop
DCR
LL = ------------------- = ---  -------------------  ------------------------------------------  ------------Io
Ri
R sum
4
N
R ntcnet + --------------N
(EQ. 39)
FIGURE 26. VOLTAGE REGULATOR EQUIVALENT CIRCUIT
Intersil provides a Microsoft Excel-based spreadsheet to help
design the compensator and the current sensing network so that
VR achieves constant output impedance as a stable system.
A VR with active droop function is a dual-loop system consisting of
a voltage loop and a droop loop, which is a current loop. However,
neither loop alone is sufficient to describe the entire system. The
spreadsheet shows two loop gain transfer functions, T1(s) and
T2(s), that describe the entire system. Figure 27 conceptually
shows T1(s) measurement set-up, and Figure 28 conceptually
shows T2(s) measurement set-up. The VR senses the inductor
current, multiplies it by a gain of the load line slope, adds it on top
of the sensed output voltage, and then feeds it to the
compensator. T1 is measured after the summing node, and T2 is
measured in the voltage loop before the summing node. The
spreadsheet gives both T1(s) and T2(s) plots. However, only T2(s)
can actually be measured on an ISL95712 regulator.
V droop
5 R sen  R droop
LL = ------------------- = ---  --------------------------------------4
N  Ri
Io
(EQ. 40)
Q1
VIN
GATE
DRIVER
Q2
LOAD LINE SLOPE
20Ω
(EQ. 41)
EA
MOD.
COMP
One can use the full-load condition to calculate Rdroop. For
example, given Iomax = 100A, Idroopmax = 45µA and
LL = 2.1mΩ, Equation 41 gives Rdroop = 4.67kΩ.
It is recommended to start with the Rdroop value calculated by
Equation 41 and fine-tune it on the actual board to get accurate
load line slope. One should record the output voltage readings at
no load and at full load for load line slope calculation. Reading
the output voltage at lighter load instead of full load will increase
the measurement error.
Compensator
Figure 20 shows the desired load transient response waveforms.
Figure 26 shows the equivalent circuit of a Voltage Regulator
(VR) with the droop function. A VR is equivalent to a voltage
29
iO
COUT
Substitution of Equation 33 and rewriting Equation 39, or
substitution of Equation 37 and rewriting Equation 40, gives the
same result as in Equation 41:
Io
R droop = ----------------  LL
I droop
VO
L
For resistor sensing, substitution of Equation 36 into Equation 3
gives the load line slope expression:
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io
+
VID
+
+
ISOLATION
TRANSFORMER
CHANNEL B
LOOP GAIN =
CHANNEL A
CHANNEL A
NETWORK
ANALYZER
CHANNEL B
EXCITATION OUTPUT
FIGURE 27. LOOP GAIN T1(s) MEASUREMENT SET-UP
T1(s) is the total loop gain of the voltage loop and the droop loop.
It always has a higher crossover frequency than T2(s), therefore
has a higher impact on system stability.
T2(s) is the voltage loop gain with closed droop loop, thus having
a higher impact on output voltage response.
FN8566.1
November 2, 2015
ISL95712
Design the compensator to get stable T1(s) and T2(s) with sufficient
phase margin and an output impedance equal to or smaller than
the load line slope.
L
INTERNAL TO
ISL95712
VO
+V
Q1
VIN
30µA
GATE Q2
DRIVER
VR_HOT_L
R
IO
CO
NTC
MONITOR
330kΩ
LOAD LINE SLOPE
EA
MOD.
+
COMP
LOOP GAIN =
+
+
VID
20 
8.45kΩ
ISOLATION
TRANSFORMER
CHANNEL B
CHANNEL A
CHANNEL A
NETWORK
ANALYZER
CHANNEL B
EXCITATION OUTPUT
FIGURE 28. LOOP GAIN T2(s) MEASUREMENT SET-UP
Current Balancing
Refer to Figures 13 through 19 for information on current
balancing. The ISL95712 achieves current balancing through
matching the ISEN pin voltages. Risen and Cisen form filters to
remove the switching ripple of the phase node voltages. It is
recommended to use a rather long RisenCisen time constant,
such that the ISEN voltages have minimal ripple and represent
the DC current flowing through the inductors. Recommended
values are Rs = 10kΩ and Cs = 0.22µF.
Thermal Monitor Component Selection
The ISL95712 features two pins, NTC and NTC_NB, which are
used to monitor motherboard temperature and alert the AMD
CPU if a thermal issues arises. The basic function of this circuitry
is outlined in the “Thermal Monitor [NTC, NTC_NB]” on page 25.
Figure 29 shows the basic configuration of the NTC resistor,
RNTC, and offset resistor, RS, used to generate the warning and
shutdown voltages at the NTC pin.
As the board temperature rises, the NTC thermistor resistance
decreases and the voltage at the NTC pin drops. When the
voltage on the NTC pin drops below the thermal warning
threshold of 0.640V, then VR_HOT_L is pulled low. When the
AMD CPU detects VR_HOT_L has gone low, it will begin throttling
back load current on both outputs to reduce the board
temperature.
If the board temperature continues to rise, the NTC thermistor
resistance will drop further and the voltage at the NTC pin could
drop below the thermal shutdown threshold of 0.580V. Once this
threshold is reached, the ISL95712 shuts down both Core and
Northbridge VRs indicating a thermal fault has occurred prior to
the thermal fault counter triggering a fault.
Submit Document Feedback
30
RNTC
Rs
WARNING SHUTDOWN
580mV
640mV
FIGURE 29. THERMAL MONITOR FEATURE OF THE ISL95712
Selection of the NTC thermistor can vary depending on how the
resistor network is configured. The equivalent resistance at the
typical thermal warning threshold voltage of 0.64V is defined in
Equation 42.
0.64V
---------------- = 21.3k
30A
(EQ. 42)
The equivalent resistance at the typical thermal shutdown
threshold voltage of 0.58V required to shutdown both outputs is
defined in Equation 43.
0.58V
---------------- = 19.3k
30A
(EQ. 43)
The NTC thermistor value correlates to the resistance change
between the warning and shutdown thresholds and the required
temperature change. If the warning level is designed to occur at a
board temperature of +100°C and the thermal shutdown level at
a board temperature of +105°C, then the resistance change of
the thermistor can be calculated. For example, a Panasonic NTC
thermistor with B = 4700 has a resistance ratio of 0.03939 of its
nominal value at +100°C and 0.03308 of its nominal value at
+105°C. Taking the required resistance change between the
thermal warning threshold and the shutdown threshold and
dividing it by the change in resistance ratio of the NTC thermistor
at the two temperatures of interest, the required resistance of
the NTC is defined in Equation 44.
 21.3k – 19.3k 
------------------------------------------------------ = 317k
 0.03939 – 0.03308 
(EQ. 44)
The closest standard thermistor to the value calculated with
B = 4700 is 330kΩ. The NTC thermistor part number is
ERTJ0EV334J. The actual resistance change of this standard
thermistor value between the warning threshold and the
shutdown threshold is calculated in Equation 45.
 330k  0.03939  –  330k  0.03308  = 2.082k
(EQ. 45)
FN8566.1
November 2, 2015
ISL95712
Since the NTC thermistor resistance at +105°C is less than the
required resistance from Equation 43, additional resistance in
series with the thermistor is required to make up the difference.
A standard resistor, 1% tolerance, added in series with the
thermistor will increase the voltage seen at the NTC pin. The
additional resistance required is calculated in Equation 46.
(EQ. 46)
19.3k – 10.916k = 8.384k
The closest, standard 1% tolerance resistor is 8.45kΩ.
The NTC thermistor is placed in a hot spot on the board, typically
near the upper MOSFET of Channel 1 of the respective output.
The standard resistor is placed next to the controller.
Layout Guidelines
PCB Layout Considerations
POWER AND SIGNAL LAYERS PLACEMENT ON THE PCB
As a general rule, power layers should be close together, either
on the top or bottom of the board, with the weak analog or logic
signal layers on the opposite side of the board. The ground-plane
layer should be adjacent to the signal layer to provide shielding.
important to have a symmetrical layout for each power train,
preferably with the controller located equidistant from each
power train. Symmetrical layout allows heat to be dissipated
equally across all power trains. Keeping the distance between
the power train and the control IC short helps keep the gate drive
traces short. These drive signals include the LGATE, UGATE,
PGND, PHASE and BOOT.
VIAS TO
GROUND
PLANE
GND
VOUT
INDUCTOR
PHASE
NODE
HIGH-SIDE
MOSFETs
VIN
OUTPUT
CAPACITORS
SCHOTTKY
DIODE
LOW-SIDE
MOSFETs
INPUT
CAPACITORS
FIGURE 30. TYPICAL POWER COMPONENT PLACEMENT
There are two sets of critical components in a DC/DC converter;
the power components and the small signal components. The
power components are the most critical because they switch
large amounts of energy. The small signal components connect
to sensitive nodes or supply critical bypassing current and signal
coupling.
When placing MOSFETs, try to keep the source of the upper
MOSFETs and the drain of the lower MOSFETs as close as
thermally possible (see Figure 30). Input high-frequency
capacitors should be placed close to the drain of the upper
MOSFETs and the source of the lower MOSFETs. Place the output
inductor and output capacitors between the MOSFETs and the
load. High-frequency output decoupling capacitors (ceramic)
should be placed as close as possible to the decoupling target
(microprocessor), making use of the shortest connection paths to
any internal planes. Place the components in such a way that the
area under the IC has less noise traces with high dV/dt and di/dt,
such as gate signals and phase node signals.
The power components should be placed first and these include
MOSFETs, input and output capacitors, and the inductor. It is
Table 15 shows layout considerations for the ISL95712
controller by pin.
COMPONENT PLACEMENT
TABLE 15. LAYOUT CONSIDERATIONS FOR THE ISL95712 CONTROLLER
PIN NUMBER
SYMBOL
LAYOUT GUIDELINES
BOTTOM PAD
GND
Connect this ground pad to the ground plane through a low impedance path. A minimum of 5 vias are recommended to
connect this pad to the internal ground plane layers of the PCB.
1
ISEN3_NB
Each ISEN pin has a capacitor (Cisen) decoupling it to VSUMN_NB, then through another capacitor (Cvsumn_nb) to GND.
Place Cisen capacitors as close as possible to the controller and keep the following loops small:
1. Any ISENx_NB pin to another ISENx_NB pin
2. Any ISENx_NB pin to GND
2
NTC_NB
3
IMON_NB
4
SVC
5
VR_HOT_L
The NTC thermistor must be placed close to the thermal source that is monitored to determine Northbridge thermal
throttling. Placement at the hottest spot of the Northbridge VR is recommended. Additional standard resistors in the
resistor network on this pin should be placed near the IC.
Place the IMON_NB resistor close to this pin and make a tight GND connection.
Use good signal integrity practices and follow AMD recommendations.
Follow AMD recommendations. Placement of the pull-up resistor near the IC is recommended.
6
SVD
Use good signal integrity practices and follow AMD recommendations.
7
VDDIO
Use good signal integrity practices and follow AMD recommendations.
8
SVT
Use good signal integrity practices and follow AMD recommendations.
9
ENABLE
No special considerations.
10
PWROK
Use good signal integrity practices and follow AMD recommendations.
11
IMON
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Place the IMON resistor close to this pin and make a tight GND connection.
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ISL95712
TABLE 15. LAYOUT CONSIDERATIONS FOR THE ISL95712 CONTROLLER (Continued)
PIN NUMBER
SYMBOL
LAYOUT GUIDELINES
12
NTC
The NTC thermistor must be placed close to the thermal source that is monitored to determine Core thermal throttling.
Placement at the hottest spot of the Core VR is recommended. Additional standard resistors in the resistor network on
this pin should be placed near the IC.
13
ISEN4
Each ISEN pin has a capacitor (Cisen) decoupling it to VSUMN and then through another capacitor (Cvsumn) to GND. Place
Cisen capacitors as close as possible to the controller and keep the following loops small:
1. Any ISEN pin to another ISEN pin
14
ISEN3
15
ISEN2
16
ISEN1
2. Any ISEN pin to GND
The red traces in the following drawing show the loops to be minimized.
Phase1
L3
Ro
R is e n
IS E N 4
C is e n
Phase1
L3
Ro
R is e n
IS E N 3
C is e n
Phase2
V
L2
Ro
R is e n
IS E N 2
C is e n
Phase3
R is e n
IS E N 1
GND
ISUMP
18
ISUMN
Ro
VSUMN
C is e n
17
L1
C vsum n
Place the current sensing circuit in general proximity of the controller.
Place capacitor Cn very close to the controller.
Place the NTC thermistor next to Core VR Channel 1 inductor so it senses the inductor temperature correctly.
Each phase of the power stage sends a pair of VSUMP and VSUMN signals to the controller. Run these two signals traces
in parallel fashion with decent width (>20mil).
IMPORTANT: Sense the inductor current by routing the sensing circuit to the inductor pads. If possible, route the traces on
a different layer from the inductor pad layer and use vias to connect the traces to the center of the pads. If no via is allowed
on the pad, consider routing the traces into the pads from the inside of the inductor. The following drawings show the two
preferred ways of routing current sensing traces.
INDUCTOR
INDUCTOR
VIAS
CURRENT-SENSING TRACES
19
VSEN
20
RTN
21
FB
22
VDD
23
PGOOD
CURRENT-SENSING TRACES
Place the filter on these pins in close proximity to the controller for good coupling.
Place the compensation components in general proximity of the controller.
A high quality, X7R dielectric MLCC capacitor is recommended to decouple this pin to GND. Place the capacitor in close
proximity to the pin with the filter resistor nearby the IC.
No special consideration.
24
COMP
Place the compensation components in general proximity of the controller.
25
BOOT1
Use a wide trace width (>30mil). Avoid routing any sensitive analog signal traces close to or crossing over this trace.
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ISL95712
TABLE 15. LAYOUT CONSIDERATIONS FOR THE ISL95712 CONTROLLER (Continued)
PIN NUMBER
SYMBOL
LAYOUT GUIDELINES
26
PHASE1
27
UGATE1
These two signals should be routed together in parallel. Each trace should have sufficient width (>30mil). Avoid routing
these signals near sensitive analog signal traces or crossing over them. Routing PHASE1 to the Core VR Channel 1 high-side
MOSFET source pin instead of a general connection to PHASE1 copper is recommended for better performance.
28
LGATE1
Use sufficient trace width (>30mil). Avoid routing this signal near any sensitive analog signal traces or crossing over them.
29
BOOT2
Use a wide trace width (>30mil). Avoid routing any sensitive analog signal traces close to or crossing over this trace.
30
PHASE2
31
UGATE2
These two signals should be routed together in parallel. Each trace should have sufficient width (>30mil). Avoid routing
these signals near sensitive analog signal traces or crossing over them. Routing PHASE2 to the Core VR Channel 2 high-side
MOSFET source pin instead of a general connection to PHASE2 copper is recommended for better performance.
32
VDDP
A high quality, X7R dielectric MLCC capacitor is recommended to decouple this pin to GND. Place the capacitor in close
proximity to the pin.
33
LGATE2
Use sufficient trace width (>30mil). Avoid routing this signal near any sensitive analog signal traces or crossing over them.
34
LGATE1_NB Use sufficient trace width (>30mil). Avoid routing this signal near any sensitive analog signal traces or crossing over them.
35
PHASE1_NB These two signals should be routed together in parallel. Each trace should have sufficient width (>30mil). Avoid routing
these signals near sensitive analog signal traces or crossing over them. Routing PHASE1_NB to the high-side MOSFET
UGATE1_NB
source pin instead of a general connection to the PHASE1_NB copper is recommended for better performance.
36
37
BOOT1_NB Use a wide trace width (>30mil). Avoid routing any sensitive analog signal traces close to or crossing over this trace.
38
PWM3
No special considerations.
39
PWM4
No special considerations.
40
PWM2_NB No special considerations.
41
PWM3_NB No special considerations.
42
I2CLK
Use good signal integrity practices
43
I2DATA
Use good signal integrity practices
44
PROG
No special considerations.
45
PGOOD_NB No special consideration.
46
COMP_NB
47
FB_NB
48
49
50
VSEN_NB
Place the compensation components in general proximity of the controller.
Place the filter on this pin in close proximity to the controller for good coupling.
ISUMN_NB Place the current sensing circuit in general proximity of the controller.
Place capacitor Cn very close to the controller.
ISUMP_NB
Place the NTC thermistor next to NB VR Channel 1 inductor so it senses the inductor temperature correctly.
Each phase of the power stage sends a pair of VSUMP and VSUMN signals to the controller. Run these two signals traces
in parallel fashion with decent width (>20mil).
IMPORTANT: Sense the inductor current by routing the sensing circuit to the inductor pads. If possible, route the traces on
a different layer from the inductor pad layer and use vias to connect the traces to the center of the pads. If no via is allowed
on the pad, consider routing the traces into the pads from the inside of the inductor. The following drawings show the two
preferred ways of routing current sensing traces.
INDUCTOR
INDUCTOR
VIAS
CURRENT-SENSING TRACES
51
ISEN1_NB
52
ISEN2_NB
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CURRENT-SENSING TRACES
Each ISEN pin has a capacitor (Cisen) decoupling it to VSUMN_NB, then through another capacitor (Cvsumn_nb) to GND.
Place Cisen capacitors as close as possible to the controller and keep the following loops small:
1. Any ISENx_NB pin to another ISENx_NB pin
2. Any ISENx_NB pin to GND
33
FN8566.1
November 2, 2015
ISL95712
Revision History
The revision history provided is for informational purposes only and is believed to be accurate, but not warranted. Please go to web to make sure you
have the latest revision.
DATE
REVISION
CHANGE
November 2, 2015
FN8566.1
On page 1 under Features, added “Serial VID clock frequency range 100kHz to 25MHz” below “Supports AMD SVI
2.0 serial data bus interface and PMBus”.
Updated Package Outline Drawing L52.6X6A to the latest revision. Changes are as follows:
-Added tolerance ± values.
March 26, 2014
FN8566.0
Initial Release
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FN8566.1
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ISL95712
Package Outline Drawing
L52.6X6A
52 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE CHAMFERED CORNER LEADS
Rev 1, 7/14
4X 4.8
6.00 ± 0.05
48X 0.40
A
B
40
6
PIN 1
INDEX AREA
52
6
PIN #1
INDEX AREA
1
6.00 ± 0.05
39
4.70 ± 0.10
27
(4X)
13
0.15
14
26
4X SEE
0.10
C A B DETAIL "Y"
0.05
C
4 52x0.20
TOP VIEW
52x0.40
BOTTOM VIEW
SEE DETAIL "X"
(5.80 TYP)
0.10 C
C
SEATING PLANE
0.08 C
0.900 ± 0.10
(
4.70)
SIDE VIEW
(48x0.40)
R0.100 TYP.
(52x0.20)
C
0.2 REF
0.165 TYP.
5
(52x0.60)
0.00 MIN.
0.05 MAX.
TYPICAL RECOMMENDED LAND PATTERN
0.165 TYP
DETAIL "X"
DETAIL "Y"
NOTES:
1.
Dimensions are in millimeters.
Dimensions in ( ) for Reference Only.
2.
Dimensioning and tolerancing conform to ASME Y14.5m-1994.
3.
Unless otherwise specified, tolerance: Decimal ± 0.05
4.
Dimension applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
5.
Tiebar shown (if present) is a non-functional feature.
6.
The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 identifier may be
either a mold or mark feature.
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FN8566.1
November 2, 2015
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