bm1q0xx appli-e

Quasi-Resonant type AC/DC converter IC
BM1Q0XX series Quasi-Resonant converter Technical Design
This application note describes the design of Quasi-Resonant converters using ROHM’s AC/DC converter IC BM1Q0xx series
devices. It explains the selection of external components and provides PCB layout guidelines. Please note that all performance
characteristics have to be verified. They are not guaranteed by the PCB layout shown here.
● Description
The BM1Q0xx series of ICs are AC/DC converters for Quasi-Resonant switching, incorporating a built-in starter circuit having
withstanding voltage of 650V. Use of external switching MOSFET and current detection resistors, provides a higher degree of
design freedom. Power efficiency is improved by the built-in starter circuit and the reduction of switching frequency under light
load conditions.
● Key features
Quasi-resonant method
Built-in 650V tolerate start circuit
Low power when load is light (Burst operation)
Maximum frequency control (120kHz)
Frequency reduction function
AC voltage correction function
VCC pin : under voltage protection
VCC pin : overvoltage protection
Over-current protection (cycle-by-cycle)
OUT pin : H voltage 12V clamp
Soft start
ZT trigger mask function
ZT Over voltage protection
FB Over Load protection [Auto-restart]
CS pin open protection [Auto-restart]
● Basic specifications
Operating power supply voltage range(VCC)
: 8.9V to 26.0V
VH voltage range(VH pin)
: up to 600V
Operating current
: Normal mode
0.60 mA (Typ.)
: Burst mode
0.35 mA (Typ.)
Maximum frequency
: 120 KHz (Typ.)
Operating temperature range
: -40°C to +85°C
● BM1Q0xx Series line-up
Product
Package
VCC OVP
ZT OVP
BM1Q001FJ
BM1Q002FJ
SOP-J8
Auto restart
Latch stop
None
Latch stop
● Applications
AC adapters, TVs, household appliances (vacuum cleaners, humidifiers, air filters, air conditioners, refrigerators,
induction heating cookers, rice cookers, etc.)
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1/10
2014.08 - Rev.A
Application Note
BM1Q0XX series Quasi-Resonant converter Technical Design
1.
Design Example of Isolated Type Quasi-Resonant Converter
C3 0
Vin
AC85V
-264V
R3 0
DA 1
Vou t
T1
Filter
C4
3 3 0 0p F
5 0 0V
R6
47k
3W
C3
4 5 0V
1 5 0u F
D3
FRD
8 0 0V 0.5 A
D4
RB 1 6 0L- 6 0
R7
10
R1 0
0.1 2 1W
R9
10 0k
R12
10
D5
RF0 5VA 2 S
C6
5 0V
1 0u F
R1 3
47k
R1 5
2k
PC 1
PC8 1 7
1
R 16
1k
4
FB
3
2
G ND
5
OUT
G ND
6
VC C
CS
8
VH
ZT
IC 1
BM1 Q0 0 1FJ
1
20V 3A
C5
1 0 0p F
1kV
R 11
1k
2
4
R1 8
2 .2 k
R2 0
10k
IC 2
TL4 3 1
R1 4
4. 3k
C7
0.0 1 u F
C 10
0.1 u F
R 17
8 2k
C 20
PC 1
PC 8 1 7
C8
47pF
C1 2
5 0V
6 8 0u F
Q1
R8 0 0 8ANX
R1
10k
R8
150
D6
RFN1 0T2 D C 1 1
50V
6 8 0uF
R 19
1 2k
3
Figure 1-1.Isolated Type Quasi-Resonant Converter Circuit Example
Quasi-Resonant Converters become DCM (Discontinuous Conduction Mode) under light load, and switching frequency
increases with the load increasing. When the load increased further, Quasi-Resonant Converters become BCM (Boundary
Conduction Mode), and switching frequency decreases with the load increasing.
Switc hin g
Fre q u en c y
Boun d a ry
p oin t
DC M
BC M
Ou tp u t
Loa d
Figure 1-2.Switching Frequency – Output Load Characteristics
IC detects Bottom and controls a timing of
switching turn ON.
BCM
DCM
Figure 1-3.Switching waveform (MOSFET Vds,Ids)
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2/10
2014.08 - Rev.A
Application Note
BM1Q0XX series Quasi-Resonant converter Technical Design
1-1. Transformer T1 design
1-1-1. Determination of flyback voltage VOR
Flyback voltage VOR is determined along with turns-ratio Np:Ns
and duty-ratio.
Np ton
=
× VIN
Ns toff
Np VOR
⇒
=
Ns
VO
VOR
⇒ Duty =
VIN + VOR
VOR = VO ×
VOR
VIN→
When VIN=95V(AC85Vx1.4x0.8)、VOR=78V、Vf=1V:
GND→
Np VOR
VOR
78V
=
=
=
= 3.714
Ns
VO Vout + Vf 20V + 1V
VOR
78V
=
= 0.45
Duty(max)=
VIN(min)+ VOR 95V + 78V
Figure 1-4. MOSFET Vds
(*) When duty is 0.5 or above, VOR is adjusted to set it below 0.5 in consideration of MOSFET’s loss, etc.
1-1-2. Determination of Minimum frequency fsw and calculation of primary -side winding inductance Lp
and primary-side maximum current Ippk
When VIN=95V, set minimum frequency to 38kHz.
Other’s parameter is following:
Since of Po=20V x 3A=60W, Po(max)=70W in consideration of over current protection.
Transformer efficiency: η=90%
Resonance capacitor: Cv=100pF
2
⎡
⎤
⎢
⎥
×
VIN
(min)
Duty
(max)
⎥ = 297 uH
Lp = ⎢
⎢ 2 × Po(max) × fsw
⎥
+ VIN(min) × Duty (max) × fsw × π × Cv ⎥
⎢
η
⎣
⎦
Ippk =
2 × Po (max)
= 3.713 A
η× Lp × fsw
1-1-3. Determination of transformer size
Based on Po(max)=70W, the transformer’s core size is EER35.
Table 1-1. Output Voltage and Transformer Core
Core sectional area
Ae (mm2)
~30
EI25/EE25
41
~60
EI28/EE28/EER28
84
~80
EI33/EER35
107
(*) The above are guideline values. For details, check with the transformer manufacturer, etc.
Output power Po(W)
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© 2014 ROHM Co., Ltd. All rights reserved.
Core size
3/10
2014.08 - Rev.A
Application Note
BM1Q0XX series Quasi-Resonant converter Technical Design
1-1-4. Calculation of primary-side turn count Np
Generally, the maximum magnetic flux density B(T) for an ordinary ferrite core is 0.4T @100°C, so Bsat = 0.35T.
Np >
Lp × Ippk
297uH × 3.713A
=
= 29.4 turns
Ae × Bsat
107mm 2 × 0.35T
→
Np is 30 turns or above
Since magnetic saturation does not result from this,
Np is set based on the AL-value-NI characteristics.
When Np=30 turns,
AL − Value =
Lp
297 uH
=
= 330 nH / turns 2
Np 2 30turns 2
NI = Np × Ippk = 30turns× 3.713A = 111.4A・turns
AL-Value=186nH/turns
2
NI=148.5A・turns
In this case, transformer is saturated based on
the AL-value-NI characteristics.
When Np=40 turns,
AL − Value =
Lp
297 uH
=
= 186nH / turns 2
Np 2 40turns 2
Figure 1-5. TDK PC47EER35-Z
AL-Value-NI Limit Characteristics
NI = Np × Ippk = 40turns× 3.713A = 148.5A・turns
In this case, this point is within the tolerance range.
1-1-5. Calculation of secondary-side turn count Ns
Np
= 3.714
Ns
→
40
= 10.8 turns
3.714
Ns =
→
11 turns
1-1-6. Calculation of VCC turn count Nd
When VCC=15V, Vf_vcc=1V,
Nd = Ns ×
VCC + Vf_vcc
15V + 1V
= 11turns ×
= 8.8turns
Vout + Vf
20V + 1V
→
9 turns
As a result, the transformer specifications are as follows.
Table 1-2. Transformer Specifications
Core
Lp
Np
Ns
Nd
TDK PC47EER35-Z or compatible
297 uH
40 turns
11 turns
9 turns
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2014.08 - Rev.A
BM1Q0XX series Quasi-Resonant converter Technical Design
Application Note
1-2. Selection of main components
1-2-1. MOSFET: Q1
Factors to select a MOSFET include the maximum drain-source voltage, peak current, loss due to Ron, and the package’s
maximum allowable loss.
In particular, in the case of world-wide input (AC 85V to AC264 V, etc.), the MOSFET’s ON-period becomes long when the
input voltage is low, and Ron-loss makes more heat generated. Confirm while the MOSFET is assembled in the product,
and, when necessary, use a heat-sink or similar to dissipate the heat.
In this design example, ROHM’s MOSFET R8008ANX (800V, 8A, 0.79Ω) is selected based on worldwide input and
Ippk = 3.713A.
1-2-2. Input capacitor: C3
Use Table 1-3 to select the capacitance of the input capacitor.
Since Pout=20Vx3A=60W, C1=2x60=120 → 150uF.
Table 1-3. Input Capacitor Selection Table
Input voltage (Vac)
Cin(uF)
85-264
2 x Pout(W)
180-264
1 x Pout(W)
(*) The above values are guidelines for full-wave rectification.
When selecting, also consider other specifications such as the retention-time.
The withstanding voltage of the capacitor becomes, Vac (max) × 1.41. Say for AC 264V, it is 264V × 1.41 = 372V, so this should
be 400V or more.
1-2-3.Setting resistor for changing of over current protection point:R13
When input voltage is high, ON time is short, and switching frequency increases. As a result, maximum output power
increases for constant over current limiter. For that, monitoring input voltage, IC switches CS over current voltage level when
ZT input current: Izt=1mA.
Set input voltage to AC150V→DC212V when over current protection point changes.
R13 = VIN(change) ×
Nd 1
9 turns
1
×
= 212V ×
×
= 47.7kΩ
Np Izt
40turns 1mA
→
47kΩ
1-2-4. Setting resistor for ZT terminal voltage:R14
ZT bottom detected voltage is Vzt1=100mV(typ.)(ZT fall), Vzt2=200mV(typ)(ZT rise), and ZT OVP(min) is 4.65V
(BM1Q002FJ), so as a guide, set Vzt to 1V to 3V.
Vzt = (Vout + Vf ) ×
Nd
R14
×
= 1.5V R14 = 4.495kΩ
Ns R13 + R14
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5/10
4.3kΩ
2014.08 - Rev.A
Application Note
BM1Q0XX series Quasi-Resonant converter Technical Design
1-2-5. Current-sensing resistor: R10
The current-sensing resistor limits the current that flows on the primary side to provide protection against output overload.
R10 =
Vcs
0.5V
=
= 0.135Ω
Ippk 3.713A
→ 0.12Ω
Check over current protection point after it was changed.
When IC switches CS over current voltage level, it is changed from 0.5V to 0.35V.
VIN(change) = R13 ×
Ippk' =
40 turns
Np
× Izt = 47kΩ×
× 1mA = 209V
9 turns
Nd
Vcs 0.35V
=
= 2.917A
R10 0.12Ω
tdelay
toff’
ton’
Lp × Ippk'
297uH × 2.917A
ton ' =
=
= 4.145us
VIN(change)
209V
Ispk' =
40turns
Np
× Ippk' =
× 2.917A = 10.61A
11turns
Ns
2
2
⎛ Ns ⎞
⎛ 11turns ⎞
⎟ = 297uH × ⎜
Ls = Lp × ⎜
⎟ = 22.46uH
⎝ 40turns ⎠
⎝ Np ⎠
toff ' =
Ls × Ispk' 22.46uH × 10.61A
=
= 11.35us
20V + 1V
Vout + Vf
Figure 1-6. Switching waveform
tdelay = π × Lp × Cv = 3.14 × 297uH × 100pF = 0.541us
fsw ' =
1
1
=
= 62.36kHz
ton '+ toff '+ tdelay 4.145us + 11.35us + 0.541us
Transformer efficiency: η=90%
1
1
Po' = × Lp × Ippk' 2 ×fsw '×η = × 297uH × 2.917A 2 × 62.36kHz × 0.9 = 70.92W
2
2
When Po’ is under the rated output power, VIN(change), R10, etc. are adjusted to set Po’ above rated output power.
Confirm the overload protection point while the resistor is assembled in the product.
Sensing resistor loss P_R10:
P_R10(peak) = Ippk2 × R10 = 3.713A2 × 0.12Ω = 1.654W
2
2
⎛
⎛
0.45 ⎞
Duty(max)⎞
⎟ × 0.12 = 0.248W
⎟ × R10 = ⎜ 3.713A×
P_R10(rms)= Iprms2 × R10 = ⎜⎜ Ippk ×
⎟
⎜
3 ⎟⎠
3
⎠
⎝
⎝
Set to 1W or above in consideration of pulse resistance.
With regard to pulse resistance, the structure of the resistance may vary even with the same power rating.
Check with the resistor manufacturers for details.
1-2-6. VCC-diode: D5
A high-speed diode is recommended as the VCC-diode.
When D5_Vf=1V, reverse voltage applied to the VCC-diode:
Vdr = VCC(max)+Vf + VINmax×
Nd
Np
When VCC (max) = 29 V,
Vdr = 29V+1V + 372V ×
9turns
= 113.7V
40turns
With a design-margin taken into account, 122.5V / 0.7 = 175V Æ 200V component is selected.
(Example: ROHM’s RF05VA2S 200V, 0.5A)
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Application Note
BM1Q0XX series Quasi-Resonant converter Technical Design
1-2-7. VCC capacitor: C6
VCC Capacitor value ‐Startup time
A VCC capacitor is needed to stabilize the IC’s VCC voltage.
Startup time[ms]
Capacitance of 2.2μF or above is recommended
(example: 50V, 10μF).
Next, determine the startup time of the IC at power-on.
Figure 1-7 illustrates VCC capacitor capacitance and
startup time characteristics.
160
140
120
100
80
60
40
20
0
0
2
4
6
8 10 12 14 16 18
VCC Capacitor value[uF]
20
22
24
Figure 1-7. Startup Time (Reference Values)
1-2-8. VCC winding surge-voltage limiting resistor: R12
Based on the transformer’s leakage inductance (Lleak), a large surge-voltage (spike noise) may occur during the instant
when the MOSFET is switched from ON to OFF. This surge-voltage is induced in the VCC winding, and as the VCC voltage
increases the IC’s VCC overvoltage protection may be triggered.
A limiting resistor R2 (approximately 5Ω to 22Ω) is inserted to reduce the surge-voltage that is induced in the VCC winding.
Confirm the rise in VCC voltage while the resistor is assembled in the product.
1-2-9. Snubber circuits: C4,D3,R6
Based on the transformer’s leakage inductance (Lleak), a large surge-voltage (spike noise) may occur during the instant
when the MOSFET is switched from ON to OFF. This surge-voltage is applied between the MOSFET’s Drain and Source, so
in the worst case damage to MOSFET might occur. RCD snubber circuits are recommended to suppress this surge-voltage.
(1) Determination of clamp voltage (Vclamp) and clamp ripple-voltage (Vripple)
Take a design-margin based on the MOSFET’s withstand voltage for the clamp voltage.
Vclamp = 800V × 0.8 = 640V
The clamp ripple-voltage (Vripple) is about 50V.
(2) Determination of R6
R6 < 2 × Vclamp ×
Vclamp - VOR
Lleak × Ip 2 × fsw(max)
Calculation of Ip, fsw when Lleak = Lp × 10% = 297μH × 10% = 29.7μH, Po=60W and VIN(max)=372V.
1
× Lp × Ip 2 × fsw × η
2
1
Vcs
fsw =
=
Ip =
+
+ tdelay ⎛ Lp
ton
toff
⎞ ⎛
Rcs
× Ip ⎟ + ⎜
⎜
Po =
1
Np
Ls
⎞
×
× Ip ⎟ + π × Lp × Cv
⎠ ⎝ Vo + Vf Ns
⎠
⎝ VIN
⇒ Vcs=0.2657V , Ip=2.214A , fsw=91.6kHz
R6 is derived as:
R6 < 2 × 640V ×
640V - 76.4V
= 54k Ω → 47k Ω
29.7uH × 2.214 2 × 91.6kHz
R6 loss P_R6 is expressed as
P_R6 =
(Vclamp
- VIN
R6
)2
=
(640
- 372
47k Ω
)2
= 1.53W
A 3W component is determined with consideration for design margin.
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2014.08 - Rev.A
BM1Q0XX series Quasi-Resonant converter Technical Design
Application Note
(3) Determination of C4
C4 >
Vclamp
Vripple × fsw(min)
× R4
=
640V
50V × 91.6kHz
× 47k Ω
= 2973pF
→ 3300pF
The voltage applied to C4 is 640V – 264×1.41 = 268V.
Set 400V or above with design margin.
(4) Determination of D3
Choose a fast recovery diode as the diode, with a withstanding voltage that is at or above the MOSFET’s Vds (max) value.
The surge-voltage affects not only the transformer’s leakage inductance but also the PCB substrate’s pattern.
Confirm the Vds voltage while assembled in the product, and adjust the snubber circuit as necessary.
1-2-10. Output rectification diode: D6
Choose a high-speed diode (Schottky barrier diode or fast recovery diode) as the output rectification diode.
When D6_Vf=1V, reverse voltage applied to output diode is
Vdr = Vout(max)+ Vf+VINmax×
Ns
Np
When Vout(max)=20V+5%=21V:
Vdr = 21V+1V + 372V ×
11
= 124.3V
40
A 124.3V/0.7=178V → 200V component is determined with consideration for design margin.
Also, diode loss (approximate value) becomes Pd = Vf × Iout = 1V × 3A = 3W.
(Example: Rohm RFN10T2D: 200V, 10A, TO-220F package)
Use of a voltage margin of 70% or less and current of 50% or less is recommended.
Check temperature rise while assembled in the product. When necessary, reconsider the component and use a heat sink or
similar to dissipate the heat.
1-2-11. Output capacitors: C11,C12
Determine the output capacitors based on the output load‘s allowable peak-to-peak ripple voltage (ΔVpp) and ripple-current.
When the MOSFET is ON, the output diode is OFF. At that time, current is supplied to the load from the output capacitors.
When the MOSFET is OFF, the output diode is ON. At that time, the output capacitors are charged and a load current is also
supplied.
When ΔVpp = 200mV,
ΔVpp
ΔVpp
0.2V
Z_C<
=
=
= 0.0148 Ω
Np
40
Ispk
× 3.713A
× Ippk
11
Ns
at
60kHz (fsw min)
With an ordinary switching power supply electrolytic-capacitor (low-impedance component), impedance is rated at 100kHz,
so it is converted to 100kHz.
60
Z_C < 0.0148 Ω ×
= 0.009 Ω
100
at
100kHz
Ripple-current Is (rms):
Is(rms) = Ispk ×
1 - Duty
40
=
× 3.713A ×
3
11
1 - 0.45
= 5.781A
3
The capacitor’s withstanding voltage should be set to about twice the output voltage.
Vout x 2 = 20V x 2 = 40V → 50V or above
Select an electrolytic capacitor that is suitable for these conditions.
(Example: low impedance type 35V, 1000 μF × 2 parallel for switching power supply )
(*) Use the actual equipment to confirm the actual ripple-voltage and ripple-current.
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2014.08 - Rev.A
BM1Q0XX series Quasi-Resonant converter Technical Design
Application Note
1-2-12. MOSFET gate circuits: R7,R8,D4
The MOSFET’s gate circuits affect the MOSFET’s loss and generate noise. The Switching speed for turn-on is adjusted
using R7+R8, and for turn-off is adjusted using R7, via the drawing diode D4.
(Example: R7: 10Ω/ 0.25W, R8: 150Ω, D4: SBD 60V, 1A)
In the case of Quasi-Resonant converters, switching-loss basically does not occur during turn-on, but occurs predominantly
during turn-off. To reduce switching-loss when turned off, turn-off speed can be increased by reducing R7 value, but sharp
changes in current will occur, which increases the switching-noise. Since there is a trade-off between loss (heat
generation) and noise, measure the MOSFET’s temperature rise and noise while it is assembled in the product, and adjust it
as necessary.
Also, since a pulse current flows to R7, check the pulse resistance of the resistors being used.
1-2-13. FB terminal capacitor: C7
C7 is a capacitor for stability of FB voltage (approximately 1000pF to 0.01uF).
1-2-14. ZT terminal capacitor: C8
C8 is a capacitor for stability of ZT voltage and for timing adjustment of bottom detection.
Check the waveform of ZT terminal and the timing of bottom detection, and adjust it as necessary.
1-2-15. Output voltage setting resistors: R17, R18, R19
When Shunt regulator IC2:Vref=2.495V,
R 17 + R 18 ⎞
82 k Ω + 2 . 2 k Ω ⎞
⎛
⎛
Vo = ⎜ 1 +
⎟ × 2 . 495 V = 20 . 00 V
⎟ × Vref = ⎜ 1 +
R 19
12 k Ω
⎝
⎠
⎝
⎠
1-2-16. Parts for adjustment of control circuit: R15, R16, R20, C10
R20 and C10 are parts for phase compensation. Approximately R20:1k to 30kΩ, C10=0.1uF, and adjust them while they are
assembled in the product.
R15 limits a control circuit current. Approximately R15:300 to 2kΩ, and adjust it while it assembled in the product.
R16 is a resistor for adjustment of minimum operating current of shunt regulator IC2.
In case of IC2: TL431, minimum operating current is 1mA. And when Optocoupler:PC1_Vf is 1V,
R16 = 1V / 1mA = 1kΩ
1-3. EMI countermeasures
Confirm the following with regard to EMI countermeasures.
(*) Constants are reference values. Need to be adjusted based on noise effects.
-
Addition of filter to input block
-
Addition of capacitor between primary-side and secondary-side (approximately C20: Y-Cap 2200pF)
-
Addition of RC snubber to diode (approximately C30: 500V 1000pF, R30: 10Ω, 1W)
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2014.08 - Rev.A
Application Note
BM1Q0XX series Quasi-Resonant converter Technical Design
1-4. Output noise countermeasures
As an output noise countermeasure, add an LC filter
D6
(*) Constants are reference values.
Vout
L
(approximately L:10μH, C: 10μF to 100μF) to the output.
C11
C
C12
20V 3A
Need to be adjusted based on noise effects.
GND
R17
82k
R15
2k
4
1
3
2
R18
2.2k
R16
1k C10
0.1uF
R20
10k
IC2
TL431
R19
12k
Figure 1-8. LC Filter Circuit
1-5. Proposed PCB layout
A proposed layout (example) for these circuits is shown in Figure 1-9.
・ Single-sided board, lead component view
+
・ Components in red are surface-mounted components
DA1
R6
C4
C3
T1
Heat sink
Vin N
D3
Vin L
Q1
D5
D4
Heat sink
C12
C20
IC2
JP
R17
PC1
R20
JP C7
R15
JP
C8
R14
IC1
R16
C6
JP
R19
R18
R8
C10
R11
R13
JP R1
C11
R9
R7
D6
R12
R10
-
50mm
C5
GND
Vout
90mm
Figure 1-9. Proposed PCB Layout (Example)
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10/10
2014.08 - Rev.A
Notice
Notes
1) The information contained herein is subject to change without notice.
2) Before you use our Products, please contact our sales representative and verify the latest specifications :
3) Although ROHM is continuously working to improve product reliability and quality, semiconductors can break down and malfunction due to various factors.
Therefore, in order to prevent personal injury or fire arising from failure, please take safety
measures such as complying with the derating characteristics, implementing redundant and
fire prevention designs, and utilizing backups and fail-safe procedures. ROHM shall have no
responsibility for any damages arising out of the use of our Poducts beyond the rating specified by
ROHM.
4) Examples of application circuits, circuit constants and any other information contained herein are
provided only to illustrate the standard usage and operations of the Products. The peripheral
conditions must be taken into account when designing circuits for mass production.
5) The technical information specified herein is intended only to show the typical functions of and
examples of application circuits for the Products. ROHM does not grant you, explicitly or implicitly,
any license to use or exercise intellectual property or other rights held by ROHM or any other
parties. ROHM shall have no responsibility whatsoever for any dispute arising out of the use of
such technical information.
6) The Products are intended for use in general electronic equipment (i.e. AV/OA devices, communication, consumer systems, gaming/entertainment sets) as well as the applications indicated in
this document.
7) The Products specified in this document are not designed to be radiation tolerant.
8) For use of our Products in applications requiring a high degree of reliability (as exemplified
below), please contact and consult with a ROHM representative : transportation equipment (i.e.
cars, ships, trains), primary communication equipment, traffic lights, fire/crime prevention, safety
equipment, medical systems, servers, solar cells, and power transmission systems.
9) Do not use our Products in applications requiring extremely high reliability, such as aerospace
equipment, nuclear power control systems, and submarine repeaters.
10) ROHM shall have no responsibility for any damages or injury arising from non-compliance with
the recommended usage conditions and specifications contained herein.
11) ROHM has used reasonable care to ensur the accuracy of the information contained in this
document. However, ROHM does not warrants that such information is error-free, and ROHM
shall have no responsibility for any damages arising from any inaccuracy or misprint of such
information.
12) Please use the Products in accordance with any applicable environmental laws and regulations,
such as the RoHS Directive. For more details, including RoHS compatibility, please contact a
ROHM sales office. ROHM shall have no responsibility for any damages or losses resulting
non-compliance with any applicable laws or regulations.
13) When providing our Products and technologies contained in this document to other countries,
you must abide by the procedures and provisions stipulated in all applicable export laws and
regulations, including without limitation the US Export Administration Regulations and the Foreign
Exchange and Foreign Trade Act.
14) This document, in part or in whole, may not be reprinted or reproduced without prior consent of
ROHM.
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More detail product informations and catalogs are available, please contact us.
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http://www.rohm.com/contact/
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© 2014 ROHM Co., Ltd. All rights reserved.
R1102A
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