cd00020010

AN1944
Application note
Developing IGBT applications using an
TD350 advanced IGBT driver
Introduction
The TD350 is an advanced Insulated Gate Bipolar Transistor (IGBT) driver with integrated
control and protection functions. The TD350 is especially adapted for driving 1200V IGBTs
with current ratings from 15 to 75A in Ecopak-like modules.
Main features are:
●
Minimum1.2A sink / 0.75A source peak output current over full temperature range
(-20°C to 125°C)
●
Desaturation protection with adjustable blanking time and fault status signal
●
Active Miller clamp function to reduce the risk of induced turn-on in high dV/dt
conditions without the need of negative gate drive in most cases
●
Optional 2-step turn-off sequence to reduce over-voltage in case of over-current or
short-circuit event to protect IGBT and avoid RBSOA problems
●
Input stage compatible with both optocouplers and pulse transformers
Applications include a three-phase full-bridge inverter used for motor speed control and
UPS systems.
TD350 in 1200V 3-phase inverter application
HV DC
TD350
TD350
TD350
Load
TD350
TD350
TD350
GND
IGBT modules
October 2006
Rev 4
1/21
www.st.com
Contents
AN1944
Contents
1
TD350 application example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
2
Input stage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
3
Output stage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
4
Active Miller clamp . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
5
2-Level turn-off . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
6
Desaturation protection feature . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
7
Application schematics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
8
Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
9
Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
2/21
AN1944
1
TD350 application example
TD350 application example
Figure 1 shows an example of a TD350 application where the device is supplied by a
+16V/-10V isolated voltage source, but a single voltage source can also be used. A pulse
transformer is used for input signal galvanic isolation. Gate resistors at OUTH and OUTL
pins (here 47 Ohms) are to be chosen depending on the IGBT specifications and the
manufacturer recommendations. Sink and source resistor values can be independently
tuned to optimize the turn-on and turn-off behaviors and can help to solve EMI issues.
The pull-down resistor (10kOhms in this example) connected between gate and emitter of
the external IGBT ensures that the external IGBT remains OFF during the TD350 power-up
sequence.
As the driver may be used in a very noisy environment, care should be taken to decouple
the supplies. The use of 100nF ceramic capacitors connected from VH to GND (and from VL
to GND if applicable) is recommended. The capacitors should be located as close as
possible to the TD350 and the ground loops should be reduced as much as possible.
Figure 1.
TD350 application example showing all the features
10K
10K
TD350
10nF
10K
VH
1K
1
IN
2
VREF
3
DESAT 14
VH
13
FAULT
OUTH
12
4
NC
OUTL
11
5
COFF
6
NC
7
LVOFF
10K
16V
VH
470pF
VL
10
CLAMP
9
GND
8
100pF
100nF
47R
VREF
4,7K
1kV
diode
47R
10K
100nF
-10V
10K
11V
3/21
Input stage
2
AN1944
Input stage
The TD350 is compatible with both pulse transformers or optocouplers. The schematic
diagram shown in Figure 2 can be considered as example of use with both solutions.
When using an optocoupler, the IN input must be limited to approximately 5V. The pull-up
resistor to VH must be between 5kOhms and 20kOhms, depending on optocoupler
characteristics. An optional filtering capacitor can be added in the event of a highly noisy
environment, although the TD350 already includes a filtering on input signals and rejects
signals smaller than 100ns (tONMIN specification).
When using a pulse transformer, a 2.5V reference point can be built from the 5V VREF pin
with a resistor divider. The capacitor between the VREF pin and the resistor divider middlepoint provides decoupling of the 2.5V reference, and also ensures a high level on the IN
input pin at power-up to start the TD350 in OFF state.
The waveform from the pulse transformer must comply with the tONMIN and VtON/VtOFF
specifications. To turn ON the TD350 outputs, the input signal must be lower than 0.8V for at
least 220ns. Conversely, the input signal must be higher than 4.2V for at least 200ns to turn
OFF TD350 outputs. A pulse width of about 500ns at these threshold levels is
recommended. In all cases, the input signal at the IN pin must be between 0 and 5V.
Figure 2.
Application schematic (pulse transformer: left / optocoupler: right)
VH
4K7
TD350
10K
1
IN
2
VREF
TD350
1
IN
10nF
10K
47pF
5,1V
10nF
10K
Figure 3.
4/21
Typical input signal waveforms with pulse transformer (left) or
optocoupler (right)
AN1944
Output stage
The output stage is able to sink/source about 2A/1.5A typical at 25°C with a voltage drop
VOL/VOH of 5V (Figure 4). The minimum sink/source currents over the full temperature
range (-20°C/+125°C) are 1.2A sink and 0.75A source. VOL and VOH voltage drops at 0.5A
are guaranteed to 3V and 4V maximum respectively, over the temperature range (Figure 5).
This current capability sets the limit of IGBT driving, and the IGBT gate resistor should not
be lower than approximately 15Ω.
The TD350 uses separate sink and source outputs (OUTL/OUTH) for easy gate driving.
Output current capability can be increased by using an external buffer with two low-cost
bipolar transistors.
Figure 4.
Typical output stage current capability at 25°C (VH = 16V, VL = -10V)
OUT sink current versus voltage (turn-off)
3
3
2,5
2,5
2
2
Iout (A)
Iout (A)
OUT source current versus voltage (turn-on)
1,5
1,5
1
1
0,5
0,5
0
0
-10
-5
0
5
10
15
-10
20
-5
0
Figure 5.
5
10
15
20
Vout (V)
Vout (V)
Typical VOL and VOH voltage variation with temperature
3.0
4.0
2.0
Iosink=500mA
1.0
Iosink=200mA
VH-VOH (V)
3.0
VOL-VL (V)
3
Output stage
Iosource=500mA
2.0
Iosource=200mA
1.0
Iosource=20mA
Iosink=20mA
0.0
0.0
-50
-25
0
25
50
Temp (°C)
75
100
125
-50
-25
0
25
50
75
100
125
Temp (°C)
During the power-on sequence, it is not guaranteed that the Goff signal, which controls the
OUTL-MOS (see TD350 output stage schematic diagram in Figure 6), stays HIGH. In this
case when TD350 goes out from UVLO condition, the OUTL-MOS is turned off and OUTL is
in High-Impedance state until the first IN transition occurs. In these conditions some leakage
effects might slowly charge the external IGBT gate-emitter capacitance.
5/21
Output stage
AN1944
Thus, it is recommended the use of a pull-down resistor of 10 kOhm or less (R3 in Figure 6)
connected between the gate and emitter of the external IGBT.
Figure 6.
6/21
TD350 output stage schematic
AN1944
4
Active Miller clamp
Active Miller clamp
The TD350 offers an alternative solution to the problem of the Miller current in IGBT
switching applications. Instead of driving the IGBT gate to a negative voltage to increase the
safety margin, the TD350 uses a dedicated CLAMP pin to control the Miller current. When
the IGBT is off, a low impedance path is established between IGBT gate and emitter to carry
the Miller current, and the voltage spike on the IGBT gate is greatly reduced (see Figure 7).
The CLAMP switch is opened when the input is activated and is closed when the actual gate
voltage goes close to the ground level. In this way, the CLAMP function doesn’t affect the
turn-off characteristic, but only keeps the gate to the low level throughout the off time. The
main benefit is that negative voltage can be avoided in many cases, allowing a bootstrap
technique for the high side driver supply.
The waveform shown in Figure 8 proves how using the Active Miller clamp provides a
consistent reduction of the voltage spike on IGBT gate.
Figure 7.
Active Miller clamp: principle of operation
TD350
Miller current
Miller current
high dV/dt !
high dV/dt !
active clamp
voltage spike on IGBT gate !
reduced voltage spike
7/21
Active Miller clamp
Figure 8.
AN1944
Reduction of gate voltage spike by active Miller clamp
Miller clamp implemented
in the same conditions,
the Vgs spike is reduced to less than 1V
without Miller clamp
Vgs spike higher than 3V!
For high power applications, a buffer can be used at the CLAMP pin, in the same way as at
the driver output. Figure 9 shows a schematic principle with external buffers for both the
driver output and the CLAMP function.
Figure 9.
Using external buffer to increase the current capability of the driver and
CLAMP outputs
VH
TD350
13
VH
T1
12
11
OUTH
OUTL
T2
10
9
8
VL
CLAMP
T3
GND
For very high-power applications, the Active Clamp function cannot replace the negative
gate drive, due to the effect of the parasitic inductance of the Active Clamp path. In these
cases, the application can benefit from the CLAMP output as an secondary gate discharge
path (see Figure 10).
When the gate voltage goes below 2V (i.e. the IGBT is already driven off), the CLAMP pin is
activated and the gate is rapidly driven to the negative voltage. Again, the benefit is to
improve the time to drive IGBT with large gate capacitance to the low level without affecting
the IGBT turn-off characteristics.
8/21
AN1944
Active Miller clamp
Figure 10. CLAMP used as secondary gate discharge path in large power
applications
VH
TD350
13
VH
T1
12
11
O UTH
O UTL
T2
10
VL
VL
9
8
Caution:
CLA MP
G ND
T3
VL
What to do with the CLAMP pin when not used?
Connect the CLAMP pin to VL.
9/21
2-Level turn-off
5
AN1944
2-Level turn-off
In the event of a short-circuit or over-current in the load, a large voltage overshoot can occur
across the IGBT at turn-off and can exceed the IGBT breakdown voltage. By reducing the
gate voltage before turn-off, the IGBT current is limited and the potential over-voltage is
reduced. This technique is called a 2-level turn-off. Both the level and duration of the
intermediate off-level are adjustable. Duration is set by an external resistor/capacitor in
conjunction with the integrated voltage reference for accurate timing. The level can be easily
set by an external Zener diode, and its value is selected depending on the IGBT
characteristics. This 2-level turn-off sequence takes place at each cycle; it has no effect if
the current does not exceed the normal maximum-rated value, but protects the IGBT in case
of over-current (with a slight increase of conduction losses).
This principle is shown on Figure 11. During the 2-level turn-off time, the OUTL output is
controlled by a comparator between the actual OUTL pin and an external reference voltage.
When the voltage on OUTL goes down as a result of the turn-off and reach the reference
threshold, then the OUTL output is disabled and the IGBT gate is not discharged further.
After the 2-level turn-off delay, the OUTL output is enabled again to end the turn-off
sequence.
To keep the output signal width unchanged relative to the input signal, the turn-on is delayed
by the same value as the 2-level turn-off delay (Figure 12).
Figure 11. Principle schematic for 2-level turn-off feature
VREF
5
COFF
Control
Block
2,5V
Lvoff
off
OUTL
11
VH
7
LVOFF
VL
120µA
10
The duration of the 2-level turn-off is set by the external RC components, and is given by the
formula:
Equation 1
t A [ µs ] = 0.7 • R off [ KΩ] • C off [ nF ]
For example: With Roff=10kΩ and Coff=220pF, tA delay is approximately 1.5 microseconds.
Recommended values are Roff from 10kΩ to 20kΩ, and Coff from 100pF to 330pF, providing
a range of delay from approximately 0.7 to 4.6 microseconds.
10/21
AN1944
2-Level turn-off
Figure 12. Waveforms of the 2-level turn-off function (COFF timing exaggerated for
illustration)
IN input
COFF timing
OUTH/L outputs
Tests with an IGBT module of 1200V and 25A (Eupec FP25R12KE) are shown in Figure 13
for a 150A over-current event.
–
Classical turn-off: OUT voltage is turned-off from VH = 16V to VL = -10V
–
2-level turn-off: OUT voltage is turned-off from VH = 16V to LVOFF = 11V during
1.5µs and ultimately OUT is pulled to VL = -10V
The maximum voltage reached on the IGBT collector and commutation losses are shown in
Table 1 for both nominal rated current at 25°C (40A) and over-current (150A) conditions.
There is no noticeable difference at nominal current, and the over-voltage is greatly reduced
in case of over-current event.
Figure 13. Reduction of IGBT over-voltage stress using 2-level turn-off feature
Without 2-level turn-off
Vce max reaches 1000V!
2-level turn-off implemented
Vce max is reduced to 640V
11/21
2-Level turn-off
AN1944
Table 1.
Comparison between classical turn-off and 2-level turn-off
400V/40A
400V/150A
Turn-off mode
Caution:
Eoff (mJ)
Vce max(V)
Eoff (mJ)
Vce max (V)
Classical turn-off
2.5
620
15
1000
2-level turn-off with LVoff = 11V
2.5
620
23
640
How does one disable the 2-level turn-off feature?
Connect LVOFF to VH, remove Coff capacitor and keep COFF pin connected to Vref by a
4.7kΩ to 10kΩ resistor.
12/21
AN1944
6
Desaturation protection feature
Desaturation protection feature
The desaturation function provides a protection against over-current events. Voltage across
the IGBT is monitored, and the IGBT is turned off if the voltage threshold is reached. A
blanking time is made of an internal 250µA current source and an external capacitor. The
high voltage diode blocks the high voltage during IGBT off state (a standard 1kV or more
diode is usable); the 1kΩ (approx.) resistor filters parasitic spikes and also protects the
DESAT input (see Figure 14).
During operation, the DESAT capacitor is discharged when TD350 output is low (IGBT off).
When the IGBT is turned on, the DESAT capacitor starts charging and desaturation
protection is effective after the blanking time (tB).
Equation 2
Cdesat
t B = 7.2 [ V ] • ---------------------250 [ µA ]
Equation 3
t B [ µs ] = 0.03 • Cdesat [ pF ]
When a desaturation event occurs, the fault output is pulled down and TD350 outputs are
low (IGBT off) until the IN input signal is released (high level), then activated again (low
level).
Figure 15 shows a desaturation fault at 150A on a typical 25A module.
Figure 14. Application schematic for DESAT feature
TD350
VH
250µA
10K
FAULT
1K
DE S A T
1k V
diode
14
3
100pF
Control
7,2V
Vce
B lock
GND
8
Note that during half-bridge commutation, the DESAT pin can experience a voltage peak. It
can depend proportionally to the parasitic capacitante (Cj) of the desaturation diode, to the
voltage value of the DC bus and in inverse proportion to the value of the capacitance placed
on the DESAT pin and to the value of the resistor in series with the desaturation diode. The
voltage peak on the DESAT pin must not exceed the absolute maximum rating indicated in
the TD350 datasheet.
13/21
Desaturation protection feature
AN1944
Figure 15. The collector current ramp-up to 150A triggers the DESAT feature (test on
25A module)
Caution:
What should one do with the DESAT pin when it is not used?
Connect the DESAT pin to GND.
14/21
AN1944
7
Application schematics
Application schematics
The TD350 application designs presented below are based on the Active Miller clamp
concept. With this function, the high-side driver can be supplied with a bootstrap system
instead of using a floating positive/negative supply (see Figure 15). This concept is
applicable to low and medium power systems, up to approximately 10kW. The main benefit
of this is to reduce the global application cost by making the supply system simpler.
Figure 16 shows the half-bridge design concept using the TD350.
It should be highlighted that the Active Miller clamp is fully managed by the TD350 and does
not require any special action from the system controller.
Figure 16. TD350 application concept
high side
5
Rb
+
Cb
15V
Vreg
4.7u
VH
24V
OUT
15k
TD350
IN
VL
CLAMP
VH
15V
OUT
15k
TD350
IN
VL
CLAMP
The TD350 is able to drive 1200V IGBT modules up to 50A or 75A (depending on IGBT
technology and manufacturer). Key parameters to consider are the TD350 peak output
current (0.75A source / 1.2A sink) and the IGBT gate resistor.
The values of gate resistors should be chosen starting with the recommended values from
the IGBT manufacturer. The TD350 allows different values for source and sink. Thanks to
the Active Miller clamp function, the gate resistors can be tuned independently from the
Miller effect that normally put some constraints on the gate resistor. The benefit of this is the
optimization of turn-on and turn-off behavior, especially regarding switching loses and EMI
issues. Table 2 shows the recommended gate resistors values from two major IGBT module
manufacturers, and the peak gate current (with a 15V supply) required for 10A to 100A IGBT
modules. Approximate application power is indicated.
15/21
Application schematics
Table 2.
AN1944
Recommended gate resistors
Application power
1.5
2
3
Eupec: FPxxR12KE3
15
Rgate
Ipeak
4
3
7
11
15
[kW]
25
40
50
75
[A]
75
36
27
18
5
[Ohm]
0.2
0.4
0.55
0.8
3
[A]
Fuji: 6MBIxxS-120
10
15
25
35
50
75
100
[A]
Rgate
120
82
51
33
24
16
12
[Ohm]
Ipeak
0.12
0.2
0.3
0.45
0.6
0.9
1.3
[A]
IGBT modules suitable for the TD350 are indicated in bold. For the FP50R12KE3 and
6MBI75S-120 modules, the source (charging) peak current will be limited to 0.75A in worstcase conditions instead of the theoretical 0.8A or 0.9A peak values, this usually does not
affect the application performance.
An external buffer will be required for higher power applications.
Reference schematics are shown in Figure 17 and Figure 18. Both use the bootstrap
principle for the high-side driver supply. A very simple voltage regulator is used in front of the
TD350 high-side driver. In this way, the bootstrap supply voltage can be made significantly
higher than the target driver supply, and the voltage across the bulk capacitor (CB) can
exhibit large voltage variations during each cycle with no impact on the driver operation.
Gate resistors RgL and RgH depend on the IGBT. It should be noted that the applications
only use two supplies referenced to the ground level.
The application in Figure 17 uses desaturation detection for protection in case of overcurrent. Fault feedback is not used.
The application in Figure 18 uses the two-level turn-off function (level = 11V, duration =
1.5µs) instead of desaturation detection, with the benefit of saving a high voltage diode and
avoiding a connection to the IGBT collector.
It may be useful to use both methods together. In this case, just add the components for
desaturation detection together with the 2-level turn-off schematic diagram.
16/21
AN1944
Application schematics
Figure 17. TD350 application schematic diagram with desaturation protection
high side drivers
5
5
5
2.2k
15k
IN
24V
16V
1k
DESAT
100p
VH
VREF
100n
10n
FAULT
OUTH
RgH
5.1V
TD350
10k
COFF
OUTL
RgL
VL
CLAMP
4.7u
+
GND
LVOFF
100n
15V
15k
IN
1k
DESAT
VREF
100p
VH
100n
10n
FAULT
OUTH
RgH
5.1V
10k
COFF
TD350
OUTL
RgL
VL
CLAMP
LVOFF
GND
low side drivers
17/21
Application schematics
AN1944
Figure 18. TD350 application schematic diagram with 2-level turn-off
high side drivers
5
5
5
2.2k
15k
IN
24V
16V
DESAT
VREF
VH
FAULT
OUTH
100n
10n
RgH
5.1V
10k
TD350
10k
OUTL
RgL
VL
COFF
CLAMP
4.7u
+
LVOFF
100n
GND
220p
11V
15V
15k
IN
DESAT
VREF
VH
FAULT
OUTH
100n
10n
RgH
5.1V
10k
10k
COFF
TD350
OUTL
RgL
VL
CLAMP
LVOFF
GND
220p
11V
low side drivers
18/21
AN1944
8
Conclusion
Conclusion
The TD350 is a versatile device designed for 1200V, 3-phase inverter applications,
especially for motor control and UPS systems. It covers a large range of power applications,
from 0.5kW to more than 100kW.
Thanks to its Active Miller clamp feature and low quiescent current, it can help avoid using
negative gate driving for applications up to 10kW and simplifies the global power supply
system for cost-sensitive applications.
19/21
Revision history
9
AN1944
Revision history
Table 3.
20/21
Revision history
Date
Revision
Changes
09-Sep-2004
1
Initial release
03-May-2006
2
- Quality of drawings improved according to A. Boimond remark.
- AN reviewed according to CCD comments
26-Sept-2006
3
- New template
- Minor editing changes
09-Oct-2006
4
-Figure 2. modified
AN1944
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