FREESCALE MC13176

Freescale Semiconductor, Inc.Order this document by MC13176/D
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The MC13176 is a one chip FM/AM transmitter subsystem designed for
AM/FM communication systems. It include a Colpitts crystal reference
oscillator, UHF oscillator, ÷32 prescaler and phase detector forming a
versatile PLL system. Targeted applications are in the 260 to 470 MHz band
and the 902 to 928 MHz band covered by FCC Title 47; Part 15. Other
applications include local oscillator sources in UHF and 900 MHz receivers,
UHF and 900 MHz video transmitters, RF Local Area Networks (LANs), and
high frequency clock drivers. The MC13176 offers the following features:
• UHF Current Controlled Oscillator
•
•
•
•
•
•
•
•
SEMICONDUCTOR
A
T
TECHNICAL
DA
Uses Easily Available 3rd Overtone or Fundamental Crystals for
Reference
Fewer External Parts Required
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•
UHF FM/AM
TRANSMITTER
Low Operating Supply Voltage (1.8 to 5.0 Vdc)
Low Supply Drain Currents
Power Output Adjustable (Up to 10 dBm)
Differential Output for Loop Antenna or Balun Transformer Networks
16
1
Power Down Feature
ASK Modulated by Switching Output On and Off
fo = 32 x fref
D SUFFIX
PLASTIC PACKAGE
CASE 751B
(SO–16)
ransmitter
ypical
FigureApplication
1. T
as 320 MHz AM T
PIN CONNECTIONS
AM Modulator
Osc
Tank
1
1.3k
16
0.01µ
2
Coilcraft
150–05J08
VEE
3
14
4
13
0.1µ
150p
(1)
150p
RFout
SMA
Z = 50Ω
f/32
RFC1
5
12
6
11
7
10
8
9
1.0k
16 Imod
Osc 1 1
15
0.165µ
(2)
VEE
S2
Out
Gnd
NC 2
15
NC 3
14 Out 2
Osc 4 4
13 Out 1
VEE 5
12 VCC
VCC
27k
S1
VEE
0.1µ
11 Enable
ICont 6
PDout 7
10 Reg.
Gnd
Xtale 8
9 Xtalb
100p
180p
VCC
NOTES: 1.
2.
2.
2.
Crystal
Fundamental
10 MHz
VCC
50 Ω coaxial balun, 1/10 wavelength at 320 MHz equals 1.5 inches.
Pins 5, 10 & 15 are ground and connected to VEE which is the component/DC ground plane
side of PCB. These pins must be decoupled to VCC; decoupling capacitors should be placed
as close as possible to the pins.
TION
ORDERING
INFORMA
Device
Operating
eTmperature Range
Package
MC13176D
TA = – 40 to 85°C
SO–16
 Motorola, Inc. 1998
A
T
OROLA
MOT
RF/IF DEVICE DA
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Rev 0
1
Freescale Semiconductor,
Inc.
MC13176
TINGS
MAXIMUM RA
( TA = 25°C, unless otherwise noted.)
Symbol
Rating
aVlue
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INC. 2005
Power Supply Voltage
V
7.0 (max)
CC
Operating Supply Voltage Range
1.8 to 5.0
Vdc
Junction Temperature
TJ
150
°C
Operating Ambient Temperature
TA
–40 to 85
°C
Tstg
–65 to 150
°C
ELECTRICAL
CHARACTERISTICS
Pin
Symbol
Min
p
Ty
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Max
Unit
Supply Current (Power down: I11 & I16 = 0)
–
IEE1
–0.5
–
–
µA
Supply Current (Enable [Pin 11] to VCC thru 30 k, I16 = 0)
–
IEE2
–18
–14
–
mA
Total Supply Current (Transmit Mode)
(Imod = 2.0 mA; fo = 320 MHz)
–
IEE3
– 39
–34
–
mA
Differential Output Power (fo = 320 MHz; Vref [Pin 9]
= 500 mVp–p; fo = N x fref)
Imod = 2.0 mA (see Figure 7)
Imod = 0 mA
13 & 14
Pout
Hold–in Range (± ∆fref x N) (see Figure 7)
13 & 14
dBm
2.0
–
4.7
–45
–
–
± ∆f H
4.0
8.0
–
MHz
7
lerror
22
27
–
µA
11 & 8
tenable
–
4.0
–
ms
16
BWAM
–
25
–
MHz
Spurious Outputs (Imod = 2.0 mA)
Spurious Outputs (Imod = 0 mA)
13 & 14
13 & 14
Pson
Psoff
–
–
–50
–50
–
–
dBc
Maximum Divider Input Frequency
Maximum Output Frequency
–
13 & 14
fdiv
fo
–
–
950
950
–
–
MHz
Phase Detector Output Error Current
Oscillator Enable Time (see Figure 26)
Amplitude Modulation Bandwidth (see Figure 28)
* For testing purposes, VCC is ground (see Figure 2).
est
Figure
Circuit
2. 320 MHz T
Imod
Osc
Tank
VEE
(1)
Coilcraft
150–03J0
8
16
2
15
0.1µ
0.1µ
14
3
0.098µ
4
f/N
51
0.01µ
51
0.01µ
VCC
RFout 2
(1)
10k
15p
6
11
7
10
8
9
2.2k
RFout 1
13
12
5
0.1µ
27p
1
10k
30k
Ireg. enable
0.1µ
33p
Crystal
Fundamental
VCC
10 MHz
NOTES: 1. VCC is ground; while VEE is negative with respect to ground.
2. Pins 5, 10 and 15 are brought to the circuit side of the PCB via plated through holes.
They are connected together with a trace on the PCB and each Pin is decoupled to VCC (ground).
2
T
OROLA
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(Figure 2; VEE = – 3.0 Vdc, TA = 25°C, unless otherwise noted.)*
Characteristic
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Vdc
VCC
Storage Temperature
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Unit
Freescale Semiconductor,
Inc.
MC13176
PIN FUNCTION DESCRIPTIONS
Internal Equivalent
Pin
Symbol
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BY FREESCALE SEMICONDUCTOR,
INC. 2005
Circuit
Osc 1,
Osc 4
VCC
10k
10k
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1
0sc 1
5
4
Osc 4
VEE
VEE
5
Subcon
VEE
6
ICont
Frequency Control
For VCC = 3.0 Vdc, the voltage at Pin 6 is approximately 1.55
Vdc. The oscillator is current controlled by the error current from
the phase detector. This current is amplified to drive the current
source in the oscillator section which controls the frequency of
the oscillator. Figures 8 and 9 show the ∆fosc versus ICont,
Figure 5 shows the ∆fosc versus ICont at – 40°C, + 25°C and
+ 85°C for 320 MHz. The CCO may be FM modulated as shown
in Figures 17 and 18, MC13176 320 MHz FM Transmitter. A
detailed discussion is found in the Applications Information
section.
VCC
Reg
ICont
PDout
VCC
4.0k
Supply Ground (VEE)
In the PCB layout, the ground pins (also applies to Pins 10 and
15) should be connected directly to chassis ground. Decoupling
capacitors to VCC should be placed directly at
the ground returns.
VEE
6
7
CCO Inputs
The oscillator is a current controlled type. An external oscillator
coil is connected to Pins 1 and 4 which forms a parallel
resonance LC tank circuit with the internal capacitance of the
IC and with parasitic capacitance of the PC board. Three
base–emitter capacitances in series configuration form the
capacitance for the parallel tank. These are the base–emitters
at Pins 1 and 4 and the base–emitter of the differential amplifier.
The equivalent series capacitance in the differential amplifier is
varied by the modulating current from the frequency control
circuit (see Pin 6, internal circuit). A more thorough discussion
is found in the Applications Information section.
4.0k
PDout
7
Phase Detector Output
The phase detector provides ± 30 µA to keep the CCO locked at
the desired carrier frequency. The output impedance of the
phase detector is approximately 53 kΩ. Under closed loop
conditions there is a DC voltage which is dependent upon the
free running oscillator and the reference oscillator frequencies.
The circuitry between Pins 7 and 6 should be selected for
adequate loop filtering necessary to stabilize and filter the loop
response. Low pass filtering between Pin 7 and 6 is needed so
that the corner frequency is well below the sum of the divider
and the reference oscillator frequencies, but high enough to
allow for fast response to keep the loop locked. Refer to the
Applications Information section regarding loop filtering and FM
modulation.
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1&4
Description/External
Circuit Requirements
Freescale Semiconductor,
Inc.
MC13176
PIN FUNCTION DESCRIPTIONS
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SEMICONDUCTOR,
INC. 2005
Internal
Equivalent
Symbol
8
Xtale
Circuit
9
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Xtalb
8.0k
12k
Xtalb
8
10
4.0k
Xtale
Reg. Gnd
Regulator Ground
An additional ground pin is provided to enhance the stability of
the system. Decoupling to the VCC (RF ground) is essential; it
should be done at the ground return for Pin 10.
VCC
Reg
5.0p
11
Enable
11
Enable
Subcon
8.0k
2.4k
10
Reg. Gnd
12
VCC
12
VCC
Out 1 and
Out 2
Differential Output
The output is configured differentially to easily drive a loop
antenna. By using a transformer or balun, as shown in the
application schematic, the device may then drive an unbalanced
low impedance load. Figure 6 shows how much the Output
Power and Free–Running Oscillator Frequency change with
temperature at 3.0 Vdc; Imod = 2.0 mA.
VCC
15
13
Out_Gnd
Out 1
16
Imod
15
Out_Gnd
4
Device Enable
The potential at Pin 11 is approximately 1.25 Vdc. When Pin 11
is open, the transmitter is disabled in a power down mode and
draws less than 1.0 µA ICC if the MOD at Pin 16 is also open
(i.e., it has no current driving it). To enable the transmitter a
current source of 10 µA to 90 µA is provided. Figures 3 and 4
show the relationship between ICC, VCC and Ireg. enable. Note
that ICC is flat at approximately 10 mA for Ireg. enable = 5.0 to
100 µA (Imod = 0).
Supply Voltage (VCC)
The operating supply voltage range is from 1.8 Vdc to 5.0 Vdc.
In the PCB layout, the VCC trace must be kept as wide as
possible to minimize inductive reactances along the trace; it is
best to have it completely fill around the surface mount
components and traces on the circuit side of the PCB.
VCC
13 & 14
Crystal Oscillator Inputs
The internal reference oscillator is configured as a common
emitter Colpitts. It may be operated with either a fundamental
or overtone crystal depending on the carrier frequency and the
internal prescaler. Crystal oscillator circuits and specifications
of crystals
are discussed in detail in the applications
section.
y
pp
Wi h VCC = 3.0 Vd
With
Vdc, the
h voltage
l
at Pi
Pin 8 iis approximately
i
l 1.8
Vdc and at Pin 9 is approximately 2.3 Vdc. 500 to 1000 mVp–p
should be present at Pin 9. The Colpitts is biased at 200 µA;
additional drive may be acquired by increasing the bias to
approximately 500 µA. Use 6.2 k from Pin 8 to ground.
VCC
9
Description/External
Circuit Requirements
14
16
Out 2
Imod
Output Ground
This additional ground pin provides direct access for the output
ground to the circuit board VEE.
AM Modulation/Power Output Level
The DC voltage at this pin is 0.8 Vdc with the current source
active. An external resistor is chosen to provide a source
current of 1.0 to 3.0 mA, depending on the desired output power
level at a given VCC. Figure 27 shows the relationship of Power
Output to Modulation Current, Imod. At VCC = 3.0 Vdc, 3.5 dBm
power output can be acquired with about 35 mA ICC.
For FM modulation, Pin 16 is used to set the desired output
power level as described above.
For AM modulation, the modulation signal must ride on a
positive DC bias offset which sets a static (modulation off)
modulation current. External circuitry for various schemes is
further discussed in the Applications Information section.
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Pin
Freescale Semiconductor,
Inc.
MC13176
Figure 3. Supply Current
versus Supply Voltage
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Figure 4. Supply Current versus
Regulator Enable Current
100
Ireg. enable = 90 µA
Imod = 0
6.0
4.0
2.0
2.0
3.0
VCC, SUPPLY VOLTAGE (Vdc)
4.0
5.0
10
5.0
VCC = 3.0 Vdc
Imod = 2.0 mA
f = 320 MHz (ICont = 0; TA = 25 °C)
Free–Running Oscillator
0
– 40 °C
– 5.0
25 °C
–10
85 °C
60
– 20
0
20
40
ICont, OSCILLATOR CONTROL CURRENT (µA)
f ref , REFERENCE OSCILLATOR FREQUENCY (MHz)
–15
– 40
1.0
10
100
Ireg. enable, REGULATOR ENABLE CURRENT (µA)
1000
Figure 6. Change in Oscillator Frequency and
Output Power versus Ambient Temperature
80
∆ f OSC , OSCILLATOR FREQUENCY (MHz)
Figure 5. Change Oscillator Frequency
versus Oscillator Control Current
∆ f OSC , OSCILLATOR FREQUENCY (MHz)
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1.0
10
1.0
0.1
0
0
VCC = 3.0 Vdc
Imod = 0
4.0
3.0
5.5
∆fosc
PO
5.0
2.0
1.0
4.5
0
4.0
–1.0
– 2.0
– 3.0
– 4.0
– 50
VCC = 3.0 Vdc
Imod = 2.0 mA
f = 320 MHz (ICont = 0; TA = 25°C)
Free–Running Oscillator
0
50
TA, AMBIENT TEMPERATURE (°C)
3.5
PO , OUTPUT POWER (dBm)
8.0
3.0
100
Figure 7. Reference Oscillator
Frequency versus Phase Detector Current
10.3
Closed Loop Response:
VCC = 3.0 Vdc
fo = 32 x fref
Vref = 500 mVp–p
10.2
10.1
Imod = 1.0 mA
ICC = 22 mA
PO = –1.1 dBm
10
Imod = 2.0 mA
ICC = 35.5 mA
PO = 4.7 dBm
9.9
9.8
– 30
– 20
–10
0
10
20
I7, PHASE DETECTOR CURRENT (µA)
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I CC , SUPPLY CURRENT (mA)
I CC , SUPPLY CURRENT (mA)
10
Freescale Semiconductor,
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MC13176
Figure 8. Change in Oscillator Frequency
versus Oscillator Control Current
∆f OSC , OSCILLATOR FREQUENCY (MHz)
10
VCC = 3.0 Vdc
Imod = 2.0 mA
TA = 25 °C
fosc (ICont @ 0) 320 MHz
0
–10
– 20
– 30
– 40
–100
0
400
500
100
200
300
ICont, OSCILLATOR CONTROL CURRENT (µA)
600
10
VCC = 3.0 Vdc
Imod = 2.0 mA
TA = 25 °C
fosc (ICont @ 0) 450 MHz
0
–10
– 20
– 30
– 40
–100
0
100
200
300
400
500
ICont, OSCILLATOR CONTROL CURRENT (µA)
600
APPLICATIONS INFORMATION
Evaluation PC Board
The evaluation PCB, shown in Figures 32 and 33, is very
versatile and is intended to be used across the entire useful
frequency range of this device. The center section of the
board provides an area for attaching all SMT components to
the circuit side and radial leaded components to the
component ground side of the PCB (see Figures 34 and 35).
Additionally, the peripheral area surrounding the RF core
provides pads to add supporting and interface circuitry as a
particular application dictates. This evaluation board will be
discussed and referenced in this section.
Current Controlled Oscillator (Pins 1 to 4)
It is critical to keep the interconnect leads from the CCO
(Pins 1 and 4) to the external inductor symmetrical and equal
in length. With a minimum inductor, the maximum free
running frequency is greater than 1.0 GHz. Since this
inductor will be small, it may be either a microstrip inductor,
an air wound inductor or a tuneable RF coil. An air wound
inductor may be tuned by spreading the windings, whereas
tuneable RF coils are tuned by adjusting the position of an
aluminum core in a threaded coilform. As the aluminum core
coupling to the windings is increased, the inductance is
decreased. The temperature coefficient using an aluminum
core is better than a ferrite core. The UniCoil inductors
made by Coilcraft may be obtained with aluminum cores
(Part No. 51–129–169).
Ground (Pins 5, 10 and 15)
Ground Returns: It is best to take the grounds to a
backside ground plane via plated through holes or eyelets at
the pins. The application PCB layout implements this
technique. Note that the grounds are located at or less than
100 mils from the devices pins.
Decoupling: Decoupling each ground pin to VCC isolates
each section of the device by reducing interaction between
sections and by localizing circulating currents.
Loop Characteristics (Pins 6 and 7)
Figure 10 is the component block diagram of the
MC13176D PLL system where the loop characteristics are
described by the gain constants. Access to individual
components of this PLL system is limited, inasmuch as the
loop is only pinned out at the phase detector output and the
6
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∆ f OSC , OSCILLATOR FREQUENCY (MHz)
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20
20
Figure 9. Change in Oscillator Frequency
versus Oscillator Control Current
frequency control input for the CCO. However, this allows for
characterization of the gain constants of these loop
components. The gain constants Kp, Ko and Kn are well
defined in the MC13176.
Phase Detector (Pin 7)
With the loop in lock, the difference frequency output of the
phase detector is DC voltage that is a function of the phase
difference. The sinusoidal type detector used in this IC has
the following transfer characteristic:
Ie = A Sin θe
The gain factor of the phase detector, Kp (with the loop in lock)
is specified as the ratio of DC output current, le to phase
error, θe:
Kp = Ie/θe (Amps/radians)
Kp = A Sin θe/θe
Sin θe ~ θe for θe ≤ 0.2 radians;
thus, Kp = A (Amps/radians)
Figure 7 shows that the detector DC current is
approximately 30 µA where the loop loses lock
at θ e = + π/2 radians; therefore, K p is 30 µA/radians.
Current Controlled Oscillator, CCO (Pin 6)
Figures 8 and 9 show the non–linear change in frequency
of the oscillator over an extended range of control current for
320 and 450 MHz applications. K o ranges from
approximately 6.3x105 rad/sec/µA or 100 kHz/µA (Figure 8)
to 8.8x105 rad/sec/µA or 140 kHz/µA (Figure 9) over a
relatively linear response of control current (0 to 100 µA). The
oscillator gain factor depends on the operating range of the
control current (i.e., the slope is not constant). Included in the
CCO gain factor is the internal amplifier which can sink and
source at least 30 µA of input current from the phase
detector. The internal circuitry at Pin 6 limits the CCO control
current to 50 µA of source capability while its sink capability
exceeds 200 µA as shown in Figures 8 and 9. Further
information to follow shows how to use the full capabilities of
the CCO by addition of an external loop amplifier and filter
(see Figure 14). This additional circuitry yields at Ko =
0.145 MHz/µA or 9.1x105 rad/sec/µA.
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MC13176
Figure 10. Block Diagram of MC1317XD PLL
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θi(s)
fi = f ref
Pins 9,8
Phase
Detector
θe(s)
Kp = 30 µA/rad
Pin 7
fn = fo/N
Low Pass
Filter
Kf
θn(s) = θo(s)/N
Pin 6
Divider
Kn = 1/N
Where: Kp =
=
Kf =
Kn =
Ko =
Amplifier and
Current Controlled
Oscillator
θo(s)
Phase detector gain constant in
µA/rad; Kp = 30 µA/rad
Filter transfer function
1/N; N = 32
CCO gain constant in rad/sec/µA
Ko = 9.1 x 105 rad/sec/µA
N = 32 : MC13176
Pins 13,14
Loop Filtering
The fundamental loop characteristics, such as capture
range, loop bandwidth, lock–up time and transient response
are controlled externally by loop filtering.
The natural frequency (ωn) and damping factor (∂) are
important in the transient response to a step input of phase or
frequency. For a given ∂ and lock time, ωn can be determined
from the plot shown in Figure 11.
Figure 11. Type 2 Second Order Response
1.9
ζ = 0.1
1.8
For ∂ = 0.707 and lock time = 1.0 ms;
then ωn = 5.0/t = 5.0 krad/sec.
The loop filter may take the form of a simple low pass
filter or a lag–lead filter which creates an additional pole at
origin in the loop transfer function. This additional pole
along with that of the CCO provides two pure integrators
(1/s2). In the lag–lead low pass network shown in Figure
12, the values of the low pass filtering parameters R1, R2
and C determine the loop constants ωn and ∂. The
equations t1 = R1C and t2 = R2C are related in the loop filter
transfer functions F(s) = 1 + t2s/1 + (t1 + t2)s.
Figure 12. Lag–Lead Low Pass Filter
1.7
1.6
0.2
Vin
1.5
R1
The closed loop transfer function takes the form of a 2nd
order low pass filter given by,
0.4
1.3
1.2
0.5
1.1
H(s) = KvF(s)/s + KvF(s)
From control theory, if the loop filter characteristic has F(0) =
1, the DC gain of the closed loop, Kv is defined as,
0.6
1.0
0.7
0.9
0.8
Kv = KpKoKn
and the transfer function has a natural frequency,
1.0
0.8
ωn = (Kv/t1 + t2)1/2
1.5
0.7
and a damping factor,
2.0
0.6
VO
R2
C
0.3
1.4
∂ = (ωn/2) (t2 + 1/Kv)
0.5
Rewriting the above equations and solving for the MC13176
with ∂ = 0.707 and ωn = 5.0 k rad/sec:
0.4
0.3
Kv = KpKoKn = (30) (0.91
106) (1/32) = 0.853
106
6
t1 + t2 = Kv/ωn2 = 0.853
10 /(25
106) = 34.1 ms
t2 = 2∂/ωn = (2) (0.707)/(5
103) = 0.283 ms
t1 = (Kv/ωn2) – t2= (34.1 – 0.283) = 33.8 ms
0.2
0.1
0
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fo = nfi
θ o (t), NORMALIZED OUTPUT RESPONSE
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Ko = 0.91Mrad/sec/µA
0
1.0 2.0 3.0
4.0 5.0 6.0 7.0 8.0 9.0
ωnt
10
11
12 13
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VCC
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For C = 0.47 µ;
measurement of the hold–in range (i.e. ∆fref
N = ±∆fH
2π). Since sin θe cannot exceed ±1.0, as θe approaches ±π/2
then, R1 = t1/C = 33.8
10–3/0.47
10–6 = 72 k
the2005
hold–in range is equal to the DC loop gain, Kv
N.
dthus, R2 = t2/C
= 0.283
10–3/0.47
10–6 = 0.60 k
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In the above example, the following standard value
±∆ωH = ± Kv
N
components are used,
where, Kv = KpKoKn.
C = 0.47 µ; R2 = 620 and R′1 = 72 k – 53 k ~ 18 k
In the above example,
(R′1 is defined as R1 – 53 k, the output impedance of the
±∆ωH = ± 27.3 Mrad/sec
phase detector.)
±∆fH = ± 4.35 MHz
Since the output of the phase detector is high impedance
(~50 k) and serves as a current source, and the input to the
Extended Hold–in Range
frequency control, Pin 6 is low impedance (impedance of the
The hold–in range of about 3.4% could cause problems
two diode to ground is approximately 500 Ω), it is imperative
over temperature in cases where the free–running oscillator
that the second order low pass filter design above be
drifts more than 2 to 3% because of relatively high
modified. In order to minimize loading of the R2C shunt
temperature coefficients of the ferrite tuned CCO inductor.
network, a higher impedance must be established to Pin 6. A
This problem might worsen for lower frequency applications
simple solution is achieved by adding a low pass network
where the external tuning coil is large compared to internal
between the passive second order network and the input to
capacitance at Pins 1 and 4. To improve hold–in range
Pin 6. This helps to minimize the loading effects on the
performance, it is apparent that the gain factors involved
second order low pass while further suppressing the
must be carefully considered.
sideband spurs of the crystal oscillator. A low pass filter with
R3 = 1.0 k and C2 = 1500 p has a corner frequency (fc) of
Kn = is 1/32 in the MC13176.
106 kHz; the reference sideband spurs are down greater
Kp = is fixed internally and cannot be altered.
than – 60 dBc.
Ko = Figures 8 and 9 suggest that there is capability
Ko = of greater control range with more current swing.
Figure 13. Modified Low Pass Loop Filter
Ko = However, this swing must be symmetrical about
Ko = the center of the dynamic response. The
1.0k
Pin 7 18k
Pin 6
Ko = suggested zero current operating point for
R′1
Ko = ±100 µA swing of the CCO is at about + 70 µA
R3
620 R2
Ko = offset point.
C3
1500p
Ka
= External loop amplification will be necessary
C
0.47
Ka = since the phase detector only supplies ± 30 µA.
In the design example in Figure 14, an external resistor
(R5) of 15 k to VCC (3.0 Vdc) provides approximately 100 µA
of current boost to supplement the existing 50 µA internal
source current. R4 (1.0 k) is selected for approximately
0.1 Vdc across it with 100 µA. R1, R2 and R3 are selected to
set the potential at Pin 7 and the base of 2N4402 at
approximately 0.9 Vdc and the emitter at 1.55 Vdc when error
current to Pin 6 is approximately zero µA. C1 is chosen to
reduce the level of the crystal sidebands.
Hold–In Range
The hold–in range, also called the lock range, tracking
range and synchronization range, is the ability of the CCO
frequency, fo to track the input reference signal, fref • N as it
gradually shifted away from the free running frequency, ff.
Assuming that the CCO is capable of sufficient frequency
deviation and that the internal loop amplifier and filter are not
overdriven, the CCO will track until the phase error, θe
approaches ±π/2 radians. Figures 5 through 7 are a direct
Figure 14. External Loop Amplifier
VCC = 3.0Vdc
12
30µA
Phase
Detector
Output
R3
1000p
R1
68k
R2
33k
4.7k
R5
R4
1.0k
2N4402
7
30µA
8
C1
50µA
15k
1.6V
6
Oscillator
Control
Circuitry
5, 10, 15
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MC13176
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f ref , REFERENCE OSCILLATOR FREQUENCY (MHz)
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fc = 0.159/RC;
Figure 15 shows the improved hold–in range of the loop.
The ∆fref is moved 950 kHz with over 200 µA swing of control
For R = 1.0 k + R7 (R7 = 53 k) and C = 390 pF
current for anBY
improved
hold–in range
of ±15.2 MHz or INC. 2005
fc = 7.55 kHz or ωc = 47 krad/sec
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The application example in Figure 17 of a 320 MHz FM
transmitter demonstrates the FM capabilities of the IC. A high
Figure 15. MC13176 Reference Oscillator
value series resistor (100 k) to Pin 6 sets up the current
Frequency versus Oscillator Control Current
source to drive the modulation section of the chip. Its value is
dependent on the peak to peak level of the encoding data
10.6
and the maximum desired frequency deviation. The data
Closed Loop Response:
input is AC coupled with a large coupling capacitor which is
f
=
32
x
f
o
ref
10.4
VCC = 3.0 Vdc
selected for the modulating frequency. The component
ICC = 38 mA
placements on the circuit side and ground side of the PC
10.2
Pout = 4.8 dB
board are shown in Figures 34 and 35, respectively.
Imod = 2.0 mA
Figure 19 illustrates the input data of a 10 kHz modulating
V
=
500
mV
ref
p–p
10
signal at 1.6 Vp–p. Figures 20 and 21 depict the deviation
and resulting modulation spectrum showing the carrier null at
9.8
– 40 dBc. Figure 22 shows the unmodulated carrier power
output at 3.5 dBm for VCC = 3.0 Vdc.
9.6
For voice applications using a dynamic or an electret
microphone,
an op amp is used to amplify the microphone’s
9.4
low level output. The microphone amplifier circuit is shown in
–100
– 50
0
50
100
–150
Figure 16. Figure 18 shows an application example for NBFM
I6, OSCILLATOR CONTROL CURRENT (µA)
audio or direct FSK in which the reference crystal oscillator is
Lock–in Range/Capture Range
modulated.
If a signal is applied to the loop not equal to free running
frequency, ff , then the loop will capture or lock–in the
Figure 16. Microphone Amplifier
signal by making f s = f o (i.e. if the initial frequency
difference is not too great). The lock–in range can be
VCC
Data
expressed as ∆ω L ~ ± 2∂ω n
100k 120k
Input
FM Modulation
Noise external to the loop (phase detector input) is
minimized by narrowing the bandwidth. This noise is minimal
in a PLL system since the reference frequency is usually
derived from a crystal oscillator. FM can be achieved by
applying a modulation current superimposed on the control
current of the CCO. The loop bandwidth must be narrow
enough to prevent the loop from responding to the
modulation frequency components, thus, allowing the CCO
to deviate in frequency. The loop bandwidth is related to the
natural frequency ωn. In the lag–lead design example where
the natural frequency, ωn = 5.0 krad/sec and a damping
factor, ∂ = 0.707, the loop bandwidth = 1.64 kHz.
Characterization data of the closed loop responses at 320
MHz (Figure 7) show satisfactory performance using only a
simple low–pass loop filter network. The loop filter response
is strongly influenced by the high output impedance of the
push–pull current output of the phase detector.
3.3k
Voice
Input
1.0k
3.9k
10k
10k
Electret
Microphone
VCC
1.0
MC33171
Data or
Audio
Output
Local Oscillator Application
To reduce internal loop noise, a relatively wide loop
bandwidth is needed so that the loop tracks out or cancels
the noise. This is emphasized to reduce inherent CCO and
divider noise or noise produced by mechanical shock and
environmental vibrations. In a local oscillator application the
CCO and divider noise should be reduced by proper
selection of the natural frequency of the loop. Additional low
pass filtering of the output will likely be necessary to reduce
the crystal sideband spurs to a minimal level.
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Freescale Semiconductor,
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MC13176
Figure 17. 320 MHz MC13176D FM Transmitter
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RF Level Adjust
1.1k
5.0k
16
1
0.047µ
2
15
3
14
CW
Coilcraft
146–04J08
(1)
SMA
0.146µ
0.47µ
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130k
510p
13
f/32
0.1µ
9.1k
RFC1 (3)
5
12
6
11
VCC
(2)
VEE
15k
18k
2N4402
7
10
8
9
0.47µ
100k
33k
VCC
27k
1.0k
620
RF Output
to Antenna
50Ω
4
VCC = 3.8 to
3.3 Vdc
VCC
VCC
51p
Data Input
(1.6 Vp–p)
220p
51p
6.8 (4)
Crystal
Fundamental
10 MHz
(5)
NOTES: 1. 50 Ω coaxial balun, 2 inches long.
2. Pins 5, 10 and 15 are grounds and connnected to VEE which is the component’s side ground plane.
These pins must be decoupled to VCC; decoupling capacitors should be placed as close as possible to the pins.
3. RFC1 is 180 nH Coilcraft surface mount inductor or 190 nH Coilcraft 146–05J08.
4. Recommended source is a Coilcraft “slot seven” 7.0 mm tuneable inductor, part #7M3–682.
5. The crystal is a parallel resonant, fundamental mode calibrated with 32 pF load capacitance.
Figure 18. 320 MHz NBFM Transmitter
RF Level Adjust
1.0k
Osc
Tank
5.0k
16
1
0.047µ
2
15
3
14
Coilcraft
146–04J08
SMA
130k
6.2k
0.1µ
9.1k
f/32
470p
13
RFC1 (3)
5
12
6
11
VCC (3.6 Vdc – Lithium Battery)
(2)
VEE
15k
VCC
27k
1.0k
15k
2N4402
7
10
8
9
0.47µ
33k
VCC
10p
External
Loop Amp
100p
180p
(6)
Crystal
Fundamental
10MHz
RFC2
(4)
VCC
1.0k
10µ
RFC3
(5) MMBV432L
NOTES: 1. 50 Ω coaxial balun, 2 inches long.
2. Pins 5, 10 and 15 are grounds and connnected to VEE which is the component’s side ground plane. These
pins must be decoupled to VCC; decoupling capacitors should be placed as close as possible to the pins.
3. RFC1 is 180 nH Coilcraft surface mount inductor.
4. RFC2 and RFC3 are high impedance crystal frequency of 10 MHz; 8.2 µH molded inductor gives XL > 1000 Ω..
5. A single varactor like the MV2105 may be used whereby RFC2 is not needed.
6. The crystal is a parallel resonant, fundamental mode calibrated with 32 pF load capacitance.
10
RF Output
to Antenna
UT–034
4
4700p
CW
(1)
0.146µ
VCC
VCC
+
0.01µ
Audio or
Data Input
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Osc
Tank
Freescale Semiconductor,
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MC13176
Figure 20. Frequency Deviation
Figure 19. Input Data Waveform
Figure 21. Modulation Spectrum
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Figure 22. Unmodulated Carrier
–10
– 20
– 30
– 40
(dBc)
(dBc)
Reference Crystal Oscillator (Pins 8 and 9)
Selection of Proper Crystal: A crystal can operate in a
number of mechanical modes. The lowest resonant
frequency mode is its fundamental while higher order modes
are called overtones. At each mechanical resonance, a
crystal behaves like a RLC series–tuned circuit having a
large inductor and a high Q. The inductor Ls is series
resonance with a dynamic capacitor, Cs determined by the
elasticity of the crystal lattice and a series resistance Rs,
which accounts for the power dissipated in heating the
crystal. This series RLC circuit is in parallel with a static
capacitance, Cp which is created by the crystal block and by
the metal plates and leads that make contact with it.
Figure 23 is the equivalent circuit for a crystal in a single
resonant mode. It is assumed that other modes of resonance
are so far off frequency that their effects are negligible.
Series resonant frequency, fs is given by;
fs = 1/2π(LsCs)1/2
and parallel resonant frequency, fp is given by;
fp = fs(1 + Cs/Cp)1/2
Figure 23. Crystal Equivalent Circuit
L3
Cp
R3
C3
the frequency separation at resonance is given by;
∆f = fp–fs = fs[1 – (1 + Cs/Cp)1/2]
Usually fp is less than 1% higher than fs, and a crystal exhibits
an extremely wide variation of the reactance with frequency
between fp and fs. A crystal oscillator circuit is very stable
with frequency. This high rate of change of impedance with
frequency stabilizes the oscillator, because any significant
change in oscillator frequency will cause a large phase shift
in the feedback loop keeping the oscillator on frequency.
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Enable (Pin 11)
The enabling resistor at Pin 11 is calculated by:
Reg. enable = VCC – 1.0 Vdc/Ireg. enable
From Figure 4, Ireg. enable is chosen to be 75 µA. So, for a
VCC = 3.0 Vdc Rreg. enable = 26.6 kΩ, a standard value
27 kΩ resistor is adequate.
Layout Considerations
Supply (Pin 12): In the PCB layout, the VCC trace must be
kept as wide as possible to minimize inductive reactance
12
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ARCHIVE INFORMATION
along the trace; it is best that VCC (RF ground) completely fills
Manufacturers specify crystal for either series or parallel
around the surface mounted components and interconnect
resonant operation. The frequency for the parallel mode is
traces
on the circuit side of the board. This technique is
calibrated withBY
a specified
shunt capacitance
called a “load INC.
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2005
demonstrated in the evaluation PC board.
capacitance.” The most common value is 30 to 32 pF. If the
Battery/Selection/Lithium Types
load capacitance is placed in series with the crystal, the
equivalent circuit will be series resonance at the specified
The device may be operated from a 3.0 V lithium battery.
parallel–resonant frequency. Frequencies up to 20 MHz use
Selection of a suitable battery is important. Because one of
parallel resonant crystal operating in the fundamental mode,
the major problems for long life battery powered equipment is
while above 20 MHz to about 60 MHz, a series resonant
oxidation of the battery terminals, a battery mounted in a
crystal specified and calibrated for operation in the overtone
clip–in socket is not advised. The battery leads or contact
mode is used.
post should be isolated from the air to eliminate oxide
build–up. The battery should have PC board mounting tabs
Application Examples
which can be soldered to the PCB. Consideration should be
Two types of crystal oscillator circuits are used in the
given for the peak current capability of the battery. Lithium
applications circuits: 1) fundamental mode common emitter
batteries have current handling capabilities based on the
Colpitts (Figures 1, 17, 18, and 24), and 2) third overtone
composition of the lithium compound, construction and the
impedance inversion Colpitts (also Figures 1 and 24).
battery size. A 1300 mA/hr rating can be achieved in the
The fundamental mode common emitter Colpitts uses a
c y l i n d r i c a l c e l l b a t t e r y. T h e R a y o v a c C R 2 / 3 A
parallel resonant crystal calibrated with a 32 pf load
lithium–manganese dioxide battery is a crimp sealed, spiral
capacitance. The capacitance values are chosen to provide
wound 3.0 Vdc, 1300 mA/hr cylindrical cell with PC board
excellent frequency stability and output power
mounting tabs. It is an excellent choice based on capacity
of > 500 mVp–p at Pin 9. In Figures 1 and 24, the
and size (1.358″ long by 0.665″ in diameter).
fundamental mode reference oscillator is fixed tuned relying
on the repeatability of the crystal and passive network to
Differential Output (Pins 13, 14)
maintain the frequency, while in the circuit shown in Figures
The availability of micro–coaxial cable and small baluns in
17 and 18, the oscillator frequency can be adjusted with the
surface
mount and radial–leaded components allows for
variable inductor for the precise operating frequency.
simple
interface
to the output ports. A loop antenna may be
The reference oscillator can be operated as high as
directly connected with bias via RFC or 50 Ω resistors.
60 MHz with a third overtone crystal. Therefore, it is
Antenna configuration will vary depending on the space
possible to use the MC13176 up to 950 MHz (based on the
available and the frequency of operation.
maximum capability of the divider network).
AM Modulation (Pin 16)
Amplitude Shift Key: The MC13176 is designed to
accommodate Amplitude Shift Keying (ASK). ASK
modulation is a form of digital modulation corresponding to
AM. The amplitude of the carrier is switched between two or
more values in response to the PCM code. For the binary
case, the usual choice is On–Off Keying (often abbreviated
OOK). The resultant amplitude modulated waveform
consists of RF pulses called marks, representing binary 1
and spaces representing binary 0.
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Freescale Semiconductor,
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MC13176
Figure 24. ASK 320 MHz Application Circuit
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Rmod
3.3k
1
16
2
15
3
14
0.01µ
Coilcraft
150–05J08
VEE
(1)
0.165µ
0.1µ
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150p
SM
A
Z = 50
4
(2)
VEE
(3)
On–Off Keyed Input
TTL Level 10 kHz
f/32
150p
13
RFOut
RFC1
5
12
6
11
1.0k
7
10
8
9
VCC
27k S1
VEE
(4)
0.1µ
180p
100p
Crystal
Fundamental
10 MHz
VCC
NOTES: 1. 50 Ω coaxial balun, 1/10 wavelength line (1.5″) provides the best
match to a 50 Ω load.
2. Pins 5, 10 and 15 are ground and connnected to VEE which is
the component/DC ground plane side of PCB. These pins must
be decoupled to VCC; decoupling capacitors should be placed
as close as possible to the pins.
Figure 24 shows a typical application in which the output
power has been reduced for linearity and current drain. The
current draw on the device is 16 mA ICC (average) and
– 22.5 dBm (average power output) using a 10 kHz
modulating rate for the on–off keying. This equates to 20 mA
and – 2.3 dBm “On”, 13 mA and – 41 dBm “Off”. In Figure 25,
the device’s modulating waveform and encoded carrier are
VCC
3. The On–Off keyed signal turns the output of the transmitter off and on with
TTL level pulses through Rmod at Pin 16. The “On” power and ICC is set
by the resistor which sets Imod = VTTL – 0.8 / Rmod. (see Figure 27).
4. S1 simulates an enable gate pulse from a microprocessor which will
enable the transmitter. (see Figure 4 to determine precise value of the
enabling resistor based on the potential of the gate pulse and the
desired enable.)
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Osc
Tank
displayed. The crystal oscillator enable time is needed to set
the acquisition timing. It takes typically 4.0 msec to reach full
magnitude of the oscillator waveform (see Figure 26,
Oscillator Waveform, at Pin 8). A square waveform of 3.0 V
peak with a period that is greater than the oscillator enable
time is applied to the Enable (Pin 11).
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Freescale Semiconductor,
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Figure 25. ASK Input Waveform and Modulated Carrier
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Pin 16
OOK Input Modulation
10 kHz TTL Waveform
Pin 8
Oscillator Waveform
Figure 27. Power Output versus Modulation Current
10
5.0
0
– 5.0
VCC = 3.0 Vdc
f = 320 MHz
–10
–15
– 20
– 25
0.1
14
1.0
Imod, MODULATION CURRENT (mA)
10
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Figure 26. Oscillator Enable Time, Tenable
PO, POWER OUTPUT (dBm)
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On–Off Keying Encoded
Carrier Envelope
Analog AM
In analog AM applications, the output amplifier’s linearity
must be carefully considered. Figure 27 is a plot of Power
Output versus Modulation Current at 320 MHz, 3.0 Vdc. In
order to achieve a linear encoding of the modulating
sinusoidal waveform on the carrier, the modulating signal
must amplitude modulate the carrier in the linear portion of its
power output response. When using a sinewave modulating
signal, the signal rides on a positive DC offset called Vmod
which sets a static (modulation off) modulation current, Imod.
Imod controls the power output of the IC. As the modulating
signal moves around this static bias point the modulating
current varies causing power output to vary or to be AM
modulated. When the IC is operated at modulation current
levels greater than 2.0 mAdc the differential output stage
starts to saturate.
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MC13176
In the design example, shown in Figure 28, the operating
point is selected as a tradeoff between average power output
and quality of the
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FREESCALE SEMICONDUCTOR, INC. 2005
For VCC = 3.0 Vdc; lCC = 18.5 mA and Imod = 0.5 mAdc and
a static DC offset of 1.04 Vdc, the circuit shown in Figure 28
completes the design. Figures 29, 30 and 31 show the results
of – 6.9 dBm output power and 100% modulation by the 10
kHz and 1.0 MHz modulating sinewave signals. The
amplitude of the input signals is approximately 800 mVp–p.
Where Rmod = (VCC – 1.04 Vdc)/0.5 mA = 3.92 k, use a
standard value resistor of 3.9 k.
Figure 28. Analog AM Transmitter
3.9k 1.04Vdc 560
VCC
16
R
3.0Vdc mod
0.8Vdc
Data
Input
800mVp–p
+
6.8µ
Figure 30. Input Signal and AM Modulated
Carrier for fmod = 10 kHz
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Figure 29. Power Output of Unmodulated Carrier
Figure 31. Input Signal and AM Modulated
Carrier for fmod = 1.0 MHz
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Freescale Semiconductor,
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Figure 32. Circuit Side View of MC13176D
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4″
4″
Figure 33. Ground Side View
4″
4″
16
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MC13176
Figure 34. Surface Mounted Components Placement
(on Circuit Side)
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Figure 35. Radial Leaded Components Placement
(on Ground Side)
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Freescale Semiconductor,
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OUTLINE DIMENSIONS
ARCHIVED BY FREESCALE SEMICONDUCTOR,
INC. 2005
D SUFFIX
PLASTIC PACKAGE
CASE 751B–05
(SO–16)
ISSUE J
16
9
–B–
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1
Freescale Semiconductor, Inc...
NOTES:
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2. CONTROLLING DIMENSION: MILLIMETER.
3. DIMENSIONS A AND B DO NOT INCLUDE
MOLD PROTRUSION.
4. MAXIMUM MOLD PROTRUSION 0.15 (0.006)
PER SIDE.
5. DIMENSION D DOES NOT INCLUDE DAMBAR
PROTRUSION. ALLOWABLE DAMBAR
PROTRUSION SHALL BE 0.127 (0.005) TOTAL
IN EXCESS OF THE D DIMENSION AT
MAXIMUM MATERIAL CONDITION.
P
8 PL
0.25 (0.010)
8
M
B
S
G
R
K
F
X 45 _
C
–T–
SEATING
PLANE
M
D
16 PL
0.25 (0.010)
T B
M
S
A
S
J
DIM
A
B
C
D
F
G
J
K
M
P
R
MILLIMETERS
MIN
MAX
9.80
10.00
3.80
4.00
1.35
1.75
0.35
0.49
0.40
1.25
1.27 BSC
0.19
0.25
0.10
0.25
0_
7_
5.80
6.20
0.25
0.50
INCHES
MIN
MAX
0.386
0.393
0.150
0.157
0.054
0.068
0.014
0.019
0.016
0.049
0.050 BSC
0.008
0.009
0.004
0.009
0_
7_
0.229
0.244
0.010
0.019
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the suitability of its products for any particular purpose, nor does Motorola assume any liability arising out of the application or use of any product or circuit, and
specifically disclaims any and all liability, including without limitation consequential or incidental damages. “Typical” parameters which may be provided in Motorola
data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All operating parameters, including “Typicals”
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arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that
Motorola was negligent regarding the design or manufacture of the part. Motorola and
are registered trademarks of Motorola, Inc. Motorola, Inc. is an Equal
Opportunity/Affirmative Action Employer.
ARCHIVE INFORMATION
–A–
Mfax is a trademark of Motorola, Inc.
How to reach us:
USA / EUROPE / Locations Not Listed: Motorola Literature Distribution;
P.O. Box 5405, Denver, Colorado 80217. 1–303–675–2140 or 1–800–441–2447
JAPAN: Motorola Japan Ltd.; SPD, Strategic Planning Office, 141,
4–32–1 Nishi–Gotanda, Shinagawa–ku, Tokyo, Japan. 81–3–5487–8488
Customer Focus Center: 1–800–521–6274
Mfax: [email protected] – TOUCHTONE 1–602–244–6609
ASIA/PACIFIC: Motorola Semiconductors H.K. Ltd.; 8B Tai Ping Industrial Park,
Motorola Fax Back System
– US & Canada ONLY 1–800–774–1848 51 Ting Kok Road, Tai Po, N.T., Hong Kong. 852–26629298
– http://sps.motorola.com/mfax/
HOME PAGE: http://motorola.com/sps/
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For More Information On This Product, MOTOROLA RF/IF DEVICE DATA
Go to: www.freescale.com