LINER LTC3586-2

LTC3586-2/LTC3586-3
High Efficiency USB Power
Manager with Boost,
Buck-Boost and Dual Bucks
DESCRIPTION
FEATURES
Power Manager
n High Efficiency Switching PowerPath™ Controller
with Bat-Track™ Adaptive Output Control and
Instant-On Operation
n Programmable USB or Wall Current Limit
(100mA/500mA/1A)
n Full Featured Li-Ion/Polymer Battery Charger with
Float Voltage of 4.2V (LTC3586-2) or 4.1V
(LTC3586-3) with 1.5A Maximum Charge Current
n Internal 180mΩ Ideal Diode Plus External Ideal Diode
Controller Powers Load in Battery Mode
n <30µA No-Load Quiescent Current when Powered
from BAT
DC/DCs
n Dual High Efficiency Buck DC/DCs (400mA I
OUT)
n High Efficiency Buck-Boost DC/DC (1A I
)
OUT
n High Efficiency Boost DC/DC (800mA I
OUT)
nDC/DC FAULT Output
n Compact (4mm × 6mm) 38-Pin QFN Package
APPLICATIONS
The LTC®3586-2/LTC3586-3 are highly integrated power
management and battery charger ICs for Li-Ion/Polymer
battery applications. They include a high efﬁciency current limited switching PowerPath manager with automatic
load prioritization, battery charger, ideal diode, and four
synchronous switching regulators (two bucks, one buckboost and one boost). Designed speciﬁcally for USB applications, the LTC3586-2/LTC3586-3’s switching power
manager automatically limits input current to a maximum
of either 100mA or 500mA for USB applications or 1A for
adapter-powered applications.
Unlike linear chargers, the LTC3586-2/LTC3586-3 switching
architecture transmits nearly all of the power available from
the USB port to the load with minimal loss and heat which
eases thermal constraints in small places. The two buck
regulators can provide up to 400mA each, the buck-boost
can deliver 1A, and the boost delivers at least 800mA.
The LTC3586-2/LTC3586-3 are available in a low profile
(0.75mm) 38-pin 4mm × 6mm QFN package.
L, LT, LTC, LTM, Burst Mode, Linear Technology and the Linear logo are registered trademarks
and PowerPath and Bat-Track are trademarks of Linear Technology Corporation. All other
trademarks are the property of their respective owners. Protected by U.S. Patents including
6522118, 6404251.
Portable Medical/Industrial Devices
Other USB-Based Handheld Products
n
n
TYPICAL APPLICATION
High Efficiency PowerPath Manager, Dual Buck, Buck-Boost and Boost
CURRENT
CONTROL
700
CC/CV
BATTERY
CHARGER
OPTIONAL
0V
CHARGE
T
LTC3586-2/LTC3586-3
+
3.3V/20mA
4
HIGH EFFICIENCY
BUCK-BOOST
MODE
ILIM
DUAL HIGH EFFICIENCY
BUCKS
1
2
3
2.5V to 3.3V/1A
2
HIGH EFFICIENCY
BOOST
RTC/LOW
POWER LOGIC
0.8V TO 3.6V/400mA
0.8V TO 3.6V/400mA
5V/800mA
4
FAULT
BATTERY CHARGE CURRENT
600
Li-Ion
ALWAYS ON LDO
EN
Battery Charge Current vs
Battery Voltage (LTC3586-2)
TO OTHER
LOADS
USB COMPLIANT
STEP-DOWN
REGULATOR
MEMORY/
CORE µP
I/O
SYSTEM
CHARGE CURRENT (mA)
USB/WALL
4.5V TO 5.5V
EXTRA CURRENT
FOR FASTER CHARGING
500
500mA USB CURRENT LIMIT
400
300
200
100
0
VBUS = 5V
5x MODE
BATTERY CHARGER PROGRAMMED FOR 1A
2.8
3
3.2 3.4 3.6
3.8
BATTERY VOLTAGE (V)
4
4.2
358623 TA01b
AUDIO/
MOTOR
358623 TA01
358623f
1
LTC3586-2/LTC3586-3
PIN CONFIGURATION
VBUS (Transient) t < 1ms,
Duty Cycle < 1%...........................................– 0.3V to 7V
VIN1, VIN2, VIN3, VIN4, VBUS (Static),
BAT, NTC, CHRG, FAULT, ILIM0 , ILIM1,
EN3, EN4, MODE, FB4, VOUT4 ......................– 0.3V to 6V
FB1.................... – 0.3V to Lesser of 6V and (VIN1 + 0.3V)
FB2.................... – 0.3V to Lesser of 6V and (VIN2 + 0.3V)
FB3, VC3............ – 0.3V to Lesser of 6V and (VIN3 + 0.3V)
EN1, EN2.................................– 0.3V to Lesser of 6V and
Max (VBUS, VOUT, BAT) + 0.3V
ICLPROG.....................................................................3mA
IFAULT, ICHRG............................................................50mA
IPROG.........................................................................2mA
ILDO3V3....................................................................30mA
ISW1, ISW2.............................................................600mA
ISW, IBAT, IVOUT.............................................................2A
ISWAB3 , ISWCD3 , ISW4, IVOUT3....................................2.5A
Operating Temperature Range (Note 2)....–40°C to 85°C
Junction Temperature (Note 3).............................. 125°C
Storage Temperature Range....................–65°C to 125°C
BAT
EN4
VOUT
VBUS
VBUS
SW
FAULT
TOP VIEW
38 37 36 35 34 33 32
ILIM0 1
31 GATE
ILIM1 2
30 CHRG
LDO3V3 3
29 PROG
CLPROG 4
28 FB1
NTC 5
27 VIN1
VOUT4 6
26 SW1
39
GND
VOUT4 7
25 SW2
SW4 8
24 VIN2
MODE 9
23 FB2
FB4 10
22 VIN4
FB3 11
21 EN1
20 EN2
VC3 12
EN3
SWCD3
VOUT3
VOUT3
SWAB3
13 14 15 16 17 18 19
VIN3
(Notes 1, 5)
VIN3
ABSOLUTE MAXIMUM RATINGS
UFE PACKAGE
38-LEAD (4mm × 6mm) PLASTIC QFN
TJMAX = 125°C, θJA = 38.7°C/W
EXPOSED PAD (PIN 39) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3586EUFE-2#PBF
LTC3586EUFE-2#TRPBF
35862
38-Lead (4mm × 6mm) Plastic QFN
– 40°C to 85°C
LTC3586EUFE-3#PBF
LTC3586EUFE-3#TRPBF
35863
38-Lead (4mm × 6mm) Plastic QFN
– 40°C to 85°C
Consult LTC Marketing for parts specified with wider operating temperature ranges.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
LTC3586 PRODUCT OPTIONS
BOOST OVERVOLTAGE
THRESHOLD (VOV4)
BOOST OVERVOLTAGE
HYSTERESIS (∆VOV4)
Bi-Directional with Latch
5.3V
300mV
4.1V
Bi-Directional with Latch
5.3V
300mV
4.2V
Output Only, No Latch
5.5V
100mV
4.1V
Output Only, No Latch
5.5V
100mV
OPTIONS
FLOAT VOLTAGE (VFLOAT)
FAULT PIN FUNCTIONALITY
LTC3586
4.2V
LTC3586-1
LTC3586-2
LTC3586-3
358623f
2
LTC3586-2/LTC3586-3
ELECTRICAL
CHARACTERISTICS
The
l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VBUS = 5V, BAT = 3.8V, VIN1 = VIN2 = VIN3 = VIN4 = VOUT3 = 3.8V,
VOUT4 = 5V, RPROG = 1k, RCLPROG = 3.01k, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
95
460
860
0.38
100
500
1000
0.50
UNITS
PowerPath Switching Regulator
VBUS
Input Supply Voltage
IBUSLIM
Total Input Current
1x Mode, VOUT = BAT
5x Mode, VOUT = BAT
10x Mode, VOUT = BAT
Suspend Mode, VOUT = BAT
IVBUSQ
VBUS Quiescent Current
1x Mode, IVOUT = 0mA
5x Mode, IVOUT = 0mA
10x Mode, IVOUT = 0mA
Suspend Mode, IVOUT = 0mA
7
15
15
0.044
mA
mA
mA
mA
hCLPROG (Note 4)
Ratio of Measured VBUS Current to
CLPROG Program Current
1x Mode
5x Mode
10x Mode
Suspend Mode
224
1133
2140
9.3
mA/mA
mA/mA
mA/mA
mA/mA
IOUT(POWERPATH)
VOUT Current Available Before Loading
BAT
1x Mode, BAT = 3.3V
5x Mode, BAT = 3.3V
10x Mode, BAT = 3.3V
Suspend Mode
135
672
1251
0.32
mA
mA
mA
mA
VCLPROG
CLPROG Servo Voltage in Current Limit 1x, 5x, 10x Modes
Suspend Mode
1.188
100
V
mV
VUVLO_VBUS
VBUS Undervoltage Lockout
Rising Threshold
Falling Threshold
VUVLO_VBUS-BAT
VBUS to BAT Differential Undervoltage
Lockout
Rising Threshold
Falling Threshold
VOUT
VOUT Voltage
1x, 5x, 10x Modes, 0V < BAT < 4.2V,
IVOUT = 0mA, Battery Charger Off
3.5
BAT + 0.3
4.7
V
USB Suspend Mode, IVOUT = 250µA
4.5
4.6
4.7
V
1.8
2.25
2.7
MHz
fOSC
4.35
l
l
l
l
87
436
800
0.31
3.95
5.5
4.30
4.00
4.35
200
50
Switching Frequency
V
mA
mA
mA
mA
V
V
mV
mV
RPMOS_POWERPATH PMOS On-Resistance
0.18
Ω
RNMOS_POWERPATH NMOS On-Resistance
0.30
Ω
2
3
A
A
IPEAK_POWERPATH
Peak Switch Current Limit (Note 5)
1x, 5x Modes
10x Mode
BAT Regulated Output Voltage
LTC3586-2
LTC3586-2
LTC3586-3
LTC3586-3
Battery Charger
VFLOAT
l
l
4.179
4.165
4.079
4.065
4.200
4.200
4.100
4.100
4.221
4.235
4.121
4.135
V
V
V
V
980
185
1022
204
1065
223
mA
mA
2
3.5
29
5
41
µA
µA
ICHG
Constant-Current Mode Charge Current RPROG = 1k
RPROG = 5k
IBAT
Battery Drain Current
VPROG
PROG Pin Servo Voltage
VPROG_TRKL
PROG Pin Servo Voltage in Trickle
Charge
VC/10
C/10 Threshold Voltage at PROG
100
mV
hPROG
Ratio of IBAT to PROG Pin Current
1022
mA/mA
ITRKL
Trickle Charge Current
100
mA
VBUS > VUVLO , IVOUT = 0µA
VBUS = 0V, IVOUT = 0µA (Ideal Diode Mode)
BAT < VTRKL
BAT < VTRKL
1.000
V
0.100
V
358623f
3
LTC3586-2/LTC3586-3
ELECTRICAL
CHARACTERISTICS
The
l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VBUS = 5V, BAT = 3.8V, VIN1 = VIN2 = VIN3 = VIN4 = VOUT3 = 3.8V,
VOUT4 = 5V, RPROG = 1k, RCLPROG = 3.01k, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
VTRKL
Trickle Charge Threshold Voltage
BAT Rising
2.7
2.85
3.0
–75
–100
∆VTRKL
Trickle Charge Hysteresis Voltage
VRECHRG
Recharge Battery Threshold Voltage
Threshold Voltage Relative to VFLOAT
130
tTERM
Safety Timer Termination
Timer Starts When BAT = VFLOAT
3.3
tBADBAT
Bad Battery Termination Time
BAT < VTRKL
0.42
hC/10
End-of-Charge Indication Current Ratio (Note 6)
0.088
UNITS
V
mV
–125
mV
4
5
Hour
0.5
0.63
Hour
0.1
0.112
mA/mA
65
100
mV
1
µA
VCHRG
CHRG Pin Output Low Voltage
ICHRG = 5mA
ICHRG
CHRG Pin Leakage Current
VCHRG = 5V
RON_CHG
Battery Charger Power FET
On-Resistance (Between VOUT and BAT)
0.18
Ω
TLIM
Junction Temperature in Constant
Temperature Mode
110
°C
NTC
VCOLD
Cold Temperature Fault Threshold
Voltage
Rising Threshold
Hysteresis
75.0
76.5
1.5
78.0
%VBUS
%VBUS
VHOT
Hot Temperature Fault Threshold
Voltage
Falling Threshold
Hysteresis
33.4
34.9
1.73
36.4
%VBUS
%VBUS
VDIS
NTC Disable Threshold Voltage
Falling Threshold
Hysteresis
0.7
1.7
50
2.7
%VBUS
mV
INTC
NTC Leakage Current
VNTC = VBUS = 5V
–50
50
nA
VFWD
Forward Voltage
VBUS = 0V, IVOUT = 10mA
IVOUT = 10mA
RDROPOUT
Internal Diode On-Resistance, Dropout
VBUS = 0V
IMAX_DIODE
Internal Diode Current Limit
Ideal Diode
2
15
mV
mV
0.18
Ω
1.6
A
Always On 3.3V Supply
VLDO3V3
Regulated Output Voltage
0mA < ILDO3V3 < 20mA
3.1
3.3
3.5
V
RCL_LDO3V3
Closed-Loop Output Resistance
4
Ω
ROL_LDO3V3
Dropout Output Resistance
23
Ω
Logic Input (EN1, EN2, EN3, EN4, MODE, ILIM0, ILIM1)
VIL
Logic Low Input Voltage
VIH
Logic High Input Voltage
IPD
Pull-Down Current
0.4
1.2
V
V
1
µA
FAULT Output
VFAULT
FAULT Pin Output Low Voltage
FAULT Delay
FBx Voltage Threshold
for FAULT (x = 1, 2, 3, 4)
IFAULT = 5mA
65
14
0.736
100
mV
ms
V
358623f
4
LTC3586-2/LTC3586-3
ELECTRICAL
CHARACTERISTICS
The
l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VBUS = 5V, BAT = 3.8V, VIN1 = VIN2 = VIN3 = VIN4 = VOUT3 = 3.8V,
VOUT4 = 5V, RPROG = 1k, RCLPROG = 3.01k, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
5.5
V
VIN1,2,3,4 Connected to VOUT Through
Low Impedance. Switching Regulators
are Disabled in UVLO
2.5
2.6
2.8
2.9
V
V
1.8
2.25
2.7
MHz
50
nA
0.80
0.82
V
60
1
µA
µA
µA
1100
mA
Switching Regulators 1, 2, 3 and 4
VIN1,2,3,4
Input Supply Voltage
VOUTUVLO
VOUT UVLO—VOUT Falling
VOUT UVLO—VOUT Rising
fOSC
Oscillator Frequency
IFB1,2,3,4
FBx Input Current
VFB1,2,3,4
VFBx Servo Voltage
2.7
VFB1,2,3,4 = 0.85V
–50
l
0.78
Switching Regulators 1 and 2 (Buck)
IVIN1,2
Pulse-Skipping Mode Input Current
Burst Mode® Input Current
Shutdown Input Current
IVOUT1,2 = 0µA, (Note 7)
IVOUT1,2 = 0µA, (Note 7)
IVOUT1,2 = 0µA, (Note 7)
ILIM1,2
PMOS Switch Current Limit
Pulse-Skipping/Burst Mode Operation (Note 5)
225
35
600
800
RP1,2
PMOS RDS(ON)
0.6
Ω
RN1,2
NMOS RDS(ON)
0.7
Ω
D1,2
Maximum Duty Cycle
RSW1,2
SW1,2 Pull-Down in Shutdown
100
%
10
kΩ
Switching Regulator 3 (Buck-Boost)
IVIN3
Input Current
PWM Mode, IVOUT3 = 0µA
Burst Mode Operation, IVOUT3 = 0µA
Shutdown
220
13
0
400
20
1
µA
µA
µA
VOUT3(LOW)
Minimum Regulated Output Voltage
For Burst Mode Operation or PWM Mode
2.65
2.75
V
VOUT3(HIGH)
Maximum Regulated Output Voltage
ILIMF3
Forward Current Limit (Switch A)
l
2
2.5
3
IPEAK3(BURST)
Forward Burst Current Limit (Switch A) Burst Mode Operation
l
200
275
350
mA
IZERO3(BURST)
Reverse Burst Current Limit (Switch D) Burst Mode Operation
l
–30
0
30
mA
IMAX3(BURST)
Maximum Deliverable Output Current in 2.7V ≤ VIN3 ≤ 5.5V, 2.75V ≤ VOUT3 ≤ 5.5V
Burst Mode Operation
(Note 8)
RDS(ON)P
PMOS RDS(ON)
Switches A, D
0.22
Ω
RDS(ON)N
NMOS RDS(ON)
Switches B, C
0.17
Ω
ILEAK(P)
PMOS Switch Leakage
Switches A, D
–1
1
µA
ILEAK(N)
NMOS Switch Leakage
Switches B, C
–1
1
µA
RVOUT3
VOUT3 Pull-Down in Shutdown
DBUCK(MAX)
Maximum Buck Duty Cycle
PWM Mode
DBOOST(MAX)
Maximum Boost Duty Cycle
PWM Mode
tSS3
Soft-Start Time
5.5
PWM Mode (Note 5)
5.6
V
50
mA
10
l
A
100
kΩ
%
75
%
0.5
ms
358623f
5
LTC3586-2/LTC3586-3
ELECTRICAL
CHARACTERISTICS
The
l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VBUS = 5V, BAT = 3.8V, VIN1 = VIN2 = VIN3 = VIN4 = VOUT3 = 3.8V,
VOUT4 = 5V, RPROG = 1k, RCLPROG = 3.01k, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Switching Regulator 4 (Boost)
IVIN4
Input Current
FB4 > 0.8V, IVOUT4 = 0µA
Shutdown, VOUT4 = 0V
180
IVOUT4
Q-Current Drawn from Boost Output
FB4 = 0V
ILIMF4
NMOS Switch Current Limit
(Note 5)
VOUT4
Output Voltage Adjust Range
VOV4
Overvoltage Shutdown
∆VOV4
Overvoltage Shutdown Hysteresis
RDS(ON)P4
PMOS RDS(ON)
RDS(ON)N4
NMOS RDS(ON)
ILEAK(P)4
PMOS Switch Leakage
Synchronous Switch
–1
ILEAK(N)4
NMOS Switch Leakage
Main Switch
–1
RVOUT4
VOUT4 Pull-Down in Shutdown
10
DBOOST(MAX)
Maximum Boost Duty Cycle
91
tSS4
Soft-Start Time
1
µA
µA
7.5
mA
2000
2800
mA
5.3
5.5
5
V
5.7
V
0.1
V
Synchronous Switch
0.25
Ω
Main Switch
0.17
Ω
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC3586E-2/LTC3586E-3 are guaranteed to meet performance
speciﬁcations from 0°C to 85°C. Speciﬁcations over the –40°C to 85°C
operating temperature range are assured by design, characterization and
correlation with statistical process controls.
Note 3: The LTC3586E-2/LTC3586E-3 include overtemperature protection
that is intended to protect the device during momentary overload
conditions. Junction temperature will exceed 125°C when overtemperature
protection is active. Continuous operation above the speciﬁed maximum
operating junction temperature may impair device reliability.
1
1
0.375
µA
µA
kΩ
94
%
ms
Note 4: Total input current is the sum of quiescent current, IVBUSQ, and
measured current given by:
VCLPROG/RCLPROG • (hCLPROG +1)
Note 5: The current limit features of this part are intended to protect the
IC from short term or intermittent fault conditions. Continuous operation
above the maximum specified pin current rating may result in device
degradation or failure.
Note 6: hC/10 is expressed as a fraction of measured full charge current
with indicated PROG resistor.
Note 7: FBx above regulation such that regulator is in sleep. Specification
does not include resistive divider current reflected back to VINX.
Note 8: Guaranteed by design.
358623f
6
LTC3586-2/LTC3586-3
TYPICAL PERFORMANCE CHARACTERISTICS
0.20
INTERNAL IDEAL
DIODE ONLY
INTERNAL IDEAL DIODE
0.15
0.10
INTERNAL IDEAL DIODE
WITH SUPPLEMENTAL
EXTERNAL VISHAY
Si2333 PMOS
0.05
0.2
VBUS = 0V
VBUS = 5V
0
0.04
0.12
0.16
0.08
FORWARD VOLTAGE (V)
0
2.7
0.20
3.0
3.6
3.9
3.3
BATTERY VOLTAGE (V)
USB Limited Battery Charge
Current vs Battery Voltage
600
25
CHARGE CURRENT (mA)
125
300
LTC3586-3
200
100
5x USB SETTING,
BATTERY CHARGER SET FOR 1A
0
3.0
3.3
3.6
3.9
2.7
BATTERY VOLTAGE (V)
4.2
100
75
LTC3586-3
50
VBUS = 5V
RPROG = 1k
25 RCLPROG = 3k
1x USB SETTING,
BATTERY CHARGER SET FOR 1A
0
2.7
3.0
3.3
3.6
3.9
BATTERY VOLTAGE (V)
358623 G04
10
5
VBUS = 5V
(SUSPEND MODE, RCLPROG = 3.01k)
3.0
3.6
3.9
3.3
BATTERY VOLTAGE (V)
4.2
358623 G06
Battery Charging Efficiency vs
Battery Voltage with No External
Load (PBAT/PBUS)
1x MODE
5x, 10x MODE
90
80
EFFICIENCY (%)
EFFICIENCY (%)
15
0
2.7
4.2
VBUS = 0V
20
100
BAT = 3.8V
90
1000
IVOUT = 0µA
358623 G05
PowerPath Switching Regulator
Efficiency vs Output Current
100
600
800
400
OUTPUT CURRENT (mA)
Battery Drain Current
vs Battery Voltage
LTC3586-2
VBUS = 5V
RPROG = 1k
RCLPROG = 3k
200
0
358623 G03
150
LTC3586-2
CHARGE CURRENT (mA)
3.25
4.2
USB Limited Battery Charge
Current vs Battery Voltage
700
400
BAT = 3.4V
3.75
358623 G02
358623 G01
500
4.00
3.50
BATTERY DRAIN CURRENT (µA)
0
VBUS = 5V
5x MODE
4.25
OUTPUT VOLTAGE (V)
0.6
0.4
4.50
BAT = 4V
RESISTANCE (Ω)
CURRENT (A)
0.25
INTERNAL IDEAL DIODE
WITH SUPPLEMENTAL
EXTERNAL VISHAY
Si2333 PMOS
0.8
Output Voltage vs Output Current
(Battery Charger Disabled)
Ideal Diode Resistance
vs Battery Voltage
Ideal Diode V-I Characteristics
1.0
(TA = 25°C unless otherwise noted)
70
60
1x CHARGING EFFICIENCY
80
5x CHARGING EFFICIENCY
70
RCLPROG = 3.01k
RPROG = 1k
IVOUT = 0mA
50
40
0.01
0.1
OUTPUT CURRENT (A)
1
358623 G07
60
2.7
3
3.5
3.9
3.3
BATTERY VOLTAGE (V)
4.2
358623 G08
358623f
7
LTC3586-2/LTC3586-3
TYPICAL PERFORMANCE CHARACTERISTICS
(TA = 25°C unless otherwise noted)
Output Voltage vs Output Current
in Suspend
VBUS Current vs VBUS Voltage
(Suspend)
45
VBUS Current vs Output Current in
Suspend
5.0
0.5
4.5
0.4
VBUS = 5V
BAT = 3.3V
RCLPROG = 3.01k
30
25
20
15
10
VBUS CURRENT (mA)
35
OUTPUT VOLTAGE (V)
4.0
3.5
3.0
0
1
0
3
VBUS VOLTAGE (V)
4
2
2.5
5
0.1
0
0.3
0.4
0.2
OUTPUT CURRENT (mA)
358623 G09
BAT = 3.4V
BAT = 3.6V
BAT = 3V
BAT = 3.1V
BAT = 3.2V
BAT = 3.3V
0
5
4.21
4.20
THERMAL REGULATION
300
200
25
RPROG = 2k
10x MODE
0
–40 –20
0
20 40 60 80
TEMPERATURE (°C)
4.17
–40
–15
35
10
TEMPERATURE (°C)
60
85
358623 G14
Oscillator Frequency
vs Temperature
2.6
BAT = 2.7V
IVOUT = 100mA
5x MODE
3.66
2.4
FREQUENCY (MHz)
OUTPUT VOLTAGE (V)
100 120
358623 G13
Low Battery (Instant On) Output
Voltage vs Temperature
3.64
3.62
3.60
–40
4.19
4.18
358623 G12
3.68
0.5
Battery Charger Float Voltage
vs Temperature
400
100
15
20
10
OUTPUT CURRENT (mA)
0.3
0.4
0.2
OUTPUT CURRENT (mA)
500
3.0
2.6
0.1
0
358623 G11
600
BAT = 3.5V
3.2
2.8
0
Battery Charge Current
vs Temperature
CHARGE CURRENT (mA)
OUTPUT VOLTAGE (V)
BAT = 3.9V, 4.2V
0.5
358623 G10
3.3V LDO Output Voltage
vs Output Current, VBUS = 0V
3.4
0.2
0.1
VBUS = 5V
BAT = 3.3V
RCLPROG = 3.01k
5
0.3
FLOAT VOLTAGE (V)
VBUS QUIESCENT CURRENT (µA)
40
VBUS = 5V
BAT = 3.6V
VBUS = 0V
2.2
BAT = 3V
VBUS = 0V
2.0
BAT = 2.7V
VBUS = 0V
–15
35
10
TEMPERATURE (°C)
60
85
358623 G15
1.8
–40
–15
35
10
TEMPERATURE (°C)
60
85
358623 G16
358623f
8
LTC3586-2/LTC3586-3
TYPICAL PERFORMANCE CHARACTERISTICS
VBUS Quiescent Current in
Suspend vs Temperature
VBUS Quiescent Current
vs Temperature
70
5x MODE
12
9
1x MODE
6
3
–40
–15
35
10
TEMPERATURE (°C)
60
100
IVOUT = 0µA
60
50
40
30
–40
85
CHRG Pin Current vs Voltage
(Pull-Down State)
–15
35
10
TEMPERATURE (°C)
60
40
20
0
85
1
0
3
4
2
CHRG PIN VOLTAGE (V)
Battery Drain Current
vs Temperature
ILDO3V3
5mA/DIV
VLDO3V3
20mV/DIV
ACCOUPLED
358623 G20
40
Switching Regulators 1, 2 PulseSkipping Mode Quiescent Currents
325
BAT = 3.8V
VBUS = 0V
ALL REGULATORS OFF
30
20
10
0
–40
–15
35
10
TEMPERATURE (°C)
60
85
1.95
VIN1, 2 = 3.8V
300
1.90
VOUT1, 2 = 2.5V
(CONSTANT FREQUENCY)
275
250
1.80
225
200
–40
–15
35
10
TEMPERATURE (°C)
100
100
80
VOUT1, 2 = 1.2V
EFFICIENCY (%)
EFFICIENCY (%)
90
VOUT1, 2 = 2.5V
70
VOUT1, 2 = 1.8V
60
50
40
30
85
1.70
358623 G22
70
VOUT1, 2 = 2.5V
VOUT1, 2 = 1.2V
VOUT1, 2 = 1.8V
60
50
40
30
20
20
10
0
60
Switching Regulators 1, 2
Burst Mode Efficiency
Switching Regulators 1, 2
Pulse-Skipping Mode Efficiency
80
1.75
VOUT1, 2 = 1.25V
(PULSE SKIPPING)
358623 G21
90
1.85
INPUT CURRENT (mA)
0mA
BATTERY DRAIN CURRENT (µA)
50
5
358623 G19
QUIESCENT CURRENT (µA)
3.3V LDO Step Response
(5mA to 15mA)
20µs/DIV
60
358623 G18
358623 G17
BAT = 3.8V
VBUS = 5V
BAT = 3.8V
80
CHRG PIN CURRENT (mA)
VBUS = 5V
IVOUT = 0µA
VBUS QUIESCENT CURRENT (µA)
VBUS QUIESCENT CURRENT (mA)
15
(TA = 25°C unless otherwise noted)
VIN1, 2 = 3.8V
1
10
100
OUTPUT CURRENT (mA)
1000
358623 G23
10
0
0.1
VIN1, 2 = 3.8V
1
10
100
OUTPUT CURRENT (mA)
1000
358623 G24
358623f
9
LTC3586-2/LTC3586-3
TYPICAL PERFORMANCE CHARACTERISTICS
1.845
VBUS = 3.8V
Burst Mode
OPERATION
1.200
PULSE-SKIPPING
MODE
1.185
1.170
0.1
1
10
100
OUTPUT CURRENT (mA)
1000
1.823
2.56
VBUS = 3.8V
Burst Mode OPERATION
PULSE-SKIPPING MODE
1.800
1.778
1.755
0.1
1
10
100
OUTPUT CURRENT (mA)
90
PWM MODE
CURVES
PMOS RDS(ON) (Ω)
EFFICIENCY (%)
50
VIN3 = 3V
VIN3 = 3.6V
VIN3 = 4.5V
40
30
20
10
0
0.1
VOUT3 = 3.3V
TYPE 3 COMPENSATION
10
ILOAD (mA)
1
100
1000
0.40
2600
0.35
2550
0.20
0.30
2500
NMOS VIN3 = 3V
NMOS VIN3 = 3.6V
NMOS VIN3 = 4.5V
0.15
0.25
0.10
0.20
0.05
0.15
0
–55 –35 –15
0.10
5 25 45 65 85 105 125
TEMPERATURE (°C)
2400
2350
2300
–55 –35 –15
5 25 45 65 85 105 125
TEMPERATURE (°C)
358623 G30
Reduction in Current
Deliverability at Low VIN3
VIN1 = 3V
VIN3 = 3.6V
11.5
5 25 45 65 85 105 125
TEMPERATURE (°C)
358623 G31
REDUCTION BELOW 1A (mA)
IQ (µA)
VIN3 = 4.5V
2450
300
12.0
11.0
–55 –35 –15
VIN3 = 3V
358623 G29
VIN3 = 4.5V
1000
VIN3 = 3.6V
14.0
12.5
1
10
100
OUTPUT CURRENT (mA)
Buck-Boost Regulator Forward
Current Limit
PMOS VIN3 = 3V
PMOS VIN3 = 3.6V
0.25
PMOS VIN3 = 4.5V
Buck-Boost Regulator Burst Mode
Operation Quiescent Current
13.0
2.47
0.30
358623 G28
13.5
2.50
358623 G27
NMOS RDS(ON) (Ω)
60
PULSE-SKIPPING MODE
RDS(ON) For Buck-Boost Regulator
100
70
2.53 Burst Mode OPERATION
358623 G26
Buck-Boost Regulator Efficiency
vs ILOAD
Burst Mode
OPERATION
CURVES
VIN3 = 3V
VIN3 = 3.6V
VIN3 = 4.5V
VBUS = 3.8V
2.44
0.1
1000
358623 G25
80
Switching Regulators 1, 2 Load
Regulation at VOUT1, 2 = 2.5V
ILIMF (mA)
1.215
OUTPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
1.230
Switching Regulators 1, 2 Load
Regulation at VOUT1, 2 = 1.8V
OUTPUT VOLTAGE (V)
Switching Regulators 1, 2 Load
Regulation at VOUT1, 2 = 1.2V
(TA = 25°C unless otherwise noted.)
STEADY-STATE ILOAD
START-UP WITH A
RESISTIVE LOAD
START-UP WITH A
CURRENT SOURCE LOAD
250
200
150
100
50
0
VOUT3 = 3.3V
TYPE 3 COMPENSATION
2.7
3.1
3.5
3.9
VIN3 (V)
4.3
4.7
358623 G32
358623f
10
LTC3586-2/LTC3586-3
TYPICAL PERFORMANCE CHARACTERISTICS
Boost Efficiency vs VIN4
Boost Efficiency (VIN4 = 3.8V)
100
0.7
VOUT4 = 5V
90
EFFICIENCY (%)
300mA
IVOUT3
200mA/DIV
0
70
0.5
EFFICIENCY
60
0.4
50
POWER LOSS
40
0.3
30
0.2
20
358623 G33
100µs/DIV
VIN3 = 3.8V
VOUT3 = 3.3V
80
70
60
10
1
100
IVOUT4 (mA)
1000
SYNCH
PMOS
OFF
50
40
30
20
0.1
10
0
90
0.6
80
POWER LOSS (W)
VOUT3
100mV/DIV
ACCOUPLED
100
EFFICIENCY (%)
Buck-Boost Step Response
(TA = 25°C unless otherwise noted.)
IVOUT4 = 300mA
VOUT4 = 5V
10
0
2.6
0
3
3.4 3.8 4.2 4.6
INPUT VOLTAGE VIN4 (V)
358623 G23
2200
4.995
2000
VIN4 = 2.7V
VOUT4 (V)
4.980
OUTPUT CURRENT IVOUT4 (mA)
5.000
4.985
VIN4 = 4.5V
VIN4 = 3.8V
4.975
5.4
358623 G35
Maximum Deliverable Boost
Output Current
Boost Output Voltage
vs Temperature
4.990
5
4.970
4.965
4.960
4.955
L = 2.2µH
VOUT4 = 4.9V (SET FOR 5V)
1800
1600
T = –45°C
1400
T = 90°C
1200
T = 25°C
1000
800
600
400
200
4.950
–45 –30 –15
0 15 30 45 60
TEMPERATURE (°C)
75
90
0
2.7
3
3.3
3.6
3.9
VIN4 (V)
358623 G36
4.2
4.5
358623 G37
Maximum Boost Duty Cycle
vs VIN4
Boost Step Response
(50mA to 300mA)
MAXIMUM DUTY CYCLE (%)
100
VOUT4
100mV/DIV
ACCOUPLED
95
T = 90°C
T = 25°C
90
T = –45°C
300mA
IVOUT4
125mA/DIV
50mA
85
80
2.7
3
3.3
3.6
3.9
VIN4 (V)
4.2
4.5
VIN4 = 3.8V
VOUT4 = 5V
L = 2.2µH
C = 10µF
50µs/DIV
358623 G39
358623 G38
358623f
11
LTC3586-2/LTC3586-3
PIN FUNCTIONS
ILIM0, ILIM1 (Pins 1, 2): Logic Inputs. ILIM0 and ILIM1
control the current limit of the PowerPath switching
regulator. See Table 1.
SW4 (Pin 8): Switch Node for the (Boost) Switching
Regulator 4. An external inductor connects between this
pin and VIN4.
(ILIM1)
(ILIM0)
0
0
1x Mode (USB 100mA Limit)
MODE (Pin 9): Digital Input. The MODE pin controls different modes of operation for the switching regulators
according to Table 2.
0
1
10x Mode (Wall 1A Limit)
Table 2. Switching Regulators Mode
1
0
Suspend
1
1
5x Mode (USB 500mA Limit)
Table 1. USB Current Limit Settings
USB SETTING
LDO3V3 (Pin 3): 3.3V LDO Output Pin. This pin provides
a regulated always-on 3.3V supply voltage. LDO3V3
gets its power from VOUT. It may be used for light loads
such as a watch dog microprocessor or real time clock.
A 1µF capacitor is required from LDO3V3 to ground. If
the LDO3V3 output is not used it should be disabled by
connecting it to VOUT.
CLPROG (Pin 4): USB Current Limit Program and Monitor Pin. A resistor from CLPROG to ground determines
the upper limit of the current drawn from the VBUS pin.
A fraction of the VBUS current is sent to the CLPROG pin
when the synchronous switch of the PowerPath switching
regulator is on. The switching regulator delivers power until
the CLPROG pin reaches 1.188V. Several VBUS current limit
settings are available via user input which will typically
correspond to the 500mA and 100mA USB speciﬁcations.
A multilayer ceramic averaging capacitor is required at
CLPROG for ﬁltering.
NTC (Pin 5): Input to the Thermistor Monitoring Circuits.
The NTC pin connects to a battery’s thermistor to determine if the battery is too hot or too cold to charge. If the
battery’s temperature is out of range, charging is paused
until it re-enters the valid range. A low drift bias resistor
is required from VBUS to NTC and a thermistor is required
from NTC to ground. If the NTC function is not desired,
the NTC pin should be grounded.
VOUT4 (Pins 6, 7): Power Output for the (Boost) Switching
Regulator 4. A 10µF MLCC capacitor should be placed as
close to the pins as possible.
REGULATION MODE
Mode
Buck
Buck-Boost
Boost
0
Pulse Skipping
PWM
Pulse Skipping
1
Burst
Burst
Pulse Skipping
FB4 (Pin 10): Feedback Input for the (Boost) Switching
Regulator 4. When the control loop is complete, the voltage on this pin servos to 0.8V.
FB3 (Pin 11): Feedback Input for (Buck-Boost) Switching
Regulator 3. When regulator 3’s control loop is complete,
this pin servos to 0.8V.
VC3 (Pin 12): Output of the Error Amplifier and Voltage Compensation Node for (Buck-Boost) Switching Regulator 3.
External Type I or Type III compensation (to FB3) connects
to this pin. See the Applications Information section for
selecting buck-boost compensation components.
SWAB3 (Pin 13): Switch Node for (Buck-Boost) Switching Regulator 3. Connected to Internal Power Switches A
and B. An external inductor connects between this node
and SWCD3.
VIN3 (Pins 14, 15): Power Input for (Buck-Boost) Switching
Regulator 3. These pins will generally be connected to VOUT.
A 1µF MLCC capacitor is recommended on these pins.
VOUT3 (Pins 16, 17): Output Voltage for (Buck-Boost)
Switching Regulator 3.
EN3 (Pin 18): Digital Input. This input enables the
buck-boost switching regulator 3.
SWCD3 (Pin 19): Switch Node for (Buck-Boost) Switching Regulator 3 Connected to Internal Power Switches C
and D. An external inductor connects between this node
and SWAB3.
358623f
12
LTC3586-2/LTC3586-3
PIN FUNCTIONS
EN2 (Pin 20): Digital Input. This input enables the buck
switching regulator 2.
EN1 (Pin 21): Digital Input. This input enables the buck
switching regulator 1.
VIN4 (Pin 22): Power Input for Switching Regulator 4
(Boost). This pin will generally be connected to VOUT.
A 1µF MLCC capacitor is recommended on this pin.
FB2 (Pin 23): Feedback Input for (Buck) Switching Regulator 2. When regulator 2’s control loop is complete, this
pin servos to 0.8V.
VIN2 (Pin 24): Power Input for (Buck) Switching Regulator 2. This pin will generally be connected to VOUT.
A 1µF MLCC capacitor is recommended on this pin.
SW2 (Pin 25): Power Transmission Pin for (Buck) Switching Regulator 2.
SW1 (Pin 26): Power Transmission Pin for (Buck) Switching Regulator 1.
VIN1 (Pin 27): Power Input for (Buck) Switching Regulator 1. This pin will generally be connected to VOUT. A 1µF
MLCC capacitor is recommended on this pin.
FB1 (Pin 28): Feedback Input for (Buck) Switching Regulator 1. When regulator 1’s control loop is complete, this
pin servos to 0.8V.
PROG (Pin 29): Charge Current Program and Charge
Current Monitor Pin. Connecting a resistor from PROG
to ground programs the charge current. If sufﬁcient input power is available in constant-current mode, this pin
servos to 1V. The voltage on this pin always represents
the actual charge current.
CHRG (Pin 30): Open-Drain Charge Status Output. The
CHRG pin indicates the status of the battery charger. Four
possible states are represented by CHRG: charging, not
charging, unresponsive battery and battery temperature
out of range. CHRG is modulated at 35kHz and switches
between a low and a high duty cycle for easy recognition by either humans or microprocessors. See Table 3.
CHRG requires a pull-up resistor and/or LED to provide
indication.
GATE (Pin 31): Analog Output. This pin controls the gate
of an optional external P-channel MOSFET transistor used
to supplement the ideal diode between VOUT and BAT. The
external ideal diode operates in parallel with the internal
ideal diode. The source of the P-channel MOSFET should
be connected to VOUT and the drain should be connected
to BAT. If the external ideal diode FET is not used, GATE
should be left ﬂoating.
BAT (Pin 32): Single Cell Li-Ion Battery Pin. Depending on
available VBUS power, a Li-Ion battery on BAT will either
deliver power to VOUT through the ideal diode or be charged
from VOUT via the battery charger.
EN4 (Pin 33): Digital Input. This input enables the boost
switching regulator 4.
VOUT (Pin 34): Output Voltage of the Switching PowerPath
Controller and Input Voltage of the Battery Charger. The
majority of the portable product should be powered from
VOUT. The LTC3586-2/LTC3586-3 will partition the available
power between the external load on VOUT and the internal
battery charger. Priority is given to the external load and
any extra power is used to charge the battery. An ideal
diode from BAT to VOUT ensures that VOUT is powered even
if the load exceeds the allotted power from VBUS or if the
VBUS power source is removed. VOUT should be bypassed
with a low impedance ceramic capacitor.
VBUS (Pins 35, 36): Primary Input Power Pin. These
pins deliver power to VOUT via the SW pin by drawing
controlled current from a DC source such as a USB port
or wall adapter.
SW (Pin 37): Power Transmission Pin for the USB
Power Path. The SW pin delivers power from VBUS to VOUT
via the buck switching regulator. A 3.3µH inductor should
be connected from SW to VOUT.
FAULT (Pin 38): Open-Drain Status Output. Used to indicate fault condition in any of the four general purpose
voltage regulators.
GND (Exposed Pad Pin 39): Ground. The exposed pad
should be connected to a continuous ground plane on the
second layer of the printed circuit board by several vias
directly under the LTC3586-2/LTC3586-3.
358623f
13
LTC3586-2/LTC3586-3
BLOCK DIAGRAM
VBUS
35, 36
2.25MHz
PowerPath
SWITCHING
REGULATOR
37 SW
3 LDO3V3
3.3V LDO
SUSPEND
LDO
500µA
34 VOUT
CLPROG 4
BATTERY
TEMPERATURE
MONITOR
NTC 5
1.188V
+
IDEAL
CC/CV
CHARGER
+
+
–
–
+
–
+
0.3V
+–
31 GATE
–
15mV
32 BAT
3.6V
29 PROG
27 VIN1
EN1
CHRG 30
CHARGE
STATUS
FAULT 38
26 SW1
400mA
2.25MHz
(BUCK)
SWITCHING
REGULATOR 1
FAULT
LOGIC
EN1 21
28 FB1
24 VIN2
EN2
400mA
2.25MHz
(BUCK)
SWITCHING
REGULATOR 2
EN2 20
EN3 18
EN4 33
MASTER LOGIC
25 SW2
23 FB2
MODE 9
ILIM0 1
14, 15
ILIM1 2
EN3
13 SWAB3
VIN4 22
VOUT4
SW4 8
VIN3
A
6, 7
800mA
2.25MHz
(BOOST)
SWITCHING
REGULATOR 4
EN4
1A
2.25MHz
(BUCK-BOOST)
SWITCHING
REGULATOR 3
B
16, 17
VOUT3
D
19 SWCD3
C
11 FB3
FB4 10
12 VC3
39
GND
358623 BD
358623f
14
LTC3586-2/LTC3586-3
OPERATION
Introduction
The LTC3586-2/LTC3586-3 are highly integrated power
management ICs which include a high efficiency switch
mode PowerPath controller, a battery charger, an ideal
diode, an always-on LDO, two 400mA buck switching
regulators, a 1A buck-boost switching regulator, and an
800mA boost switching regulator. All of the regulators can
be independently controlled via ENABLE pins.
Designed specifically for USB applications, the PowerPath
controller incorporates a precision average input current
buck switching regulator to make maximum use of the
allowable USB power. Because power is conserved, the
LTC3586-2/LTC3586-3 allow the load current on VOUT to
exceed the current drawn by the USB port without exceeding the USB load specifications.
The PowerPath switching regulator and battery charger
communicate to ensure that the input current never violates
the USB speciﬁcations.
The ideal diode from BAT to VOUT guarantees that ample
power is always available to VOUT even if there is insufﬁcient
or absent power at VBUS.
An always-on LDO provides a regulated 3.3V from available power at VOUT. Drawing very little quiescent current,
this LDO will be on at all times and can be used to supply
up to 20mA.
Along with constant frequency PWM mode, the buck and
the buck-boost switching regulators have a low power
burst mode setting for signiﬁcantly reduced quiescent
current under light load conditions.
High Efﬁciency Switching PowerPath Controller
Whenever VBUS is available and the PowerPath switching regulator is enabled, power is delivered from VBUS to
VOUT via SW. VOUT drives the combination of the external
load (including switching regulators 1, 2, 3 and 4) and
the battery charger.
If the combined load does not exceed the PowerPath
switching regulator’s programmed input current limit, VOUT
will track 0.3V above the battery (Bat-Track). By keeping
the voltage across the battery charger low, efﬁciency is
optimized because power lost to the linear battery charger is minimized. Power available to the external load is
therefore optimized.
If the combined load at VOUT is large enough to cause the
switching PowerPath supply to reach the programmed
input current limit, the battery charger will reduce its
charge current by that amount necessary to enable the
external load to be satisﬁed. Even if the battery charge
current is set to exceed the allowable USB current, the USB
speciﬁcation will not be violated. The PowerPath switching regulator will limit the average input current so that
the USB speciﬁcation is never violated. Furthermore, load
current at VOUT will always be prioritized and only excess
available power will be used to charge the battery.
If the voltage at BAT is below 3.3V, or the battery is not
present, and the load requirement does not cause the
PowerPath switching regulator to exceed the USB
speciﬁcation, VOUT will regulate at 3.6V, as shown in
Figure 1. This “instant-on” feature will allow a portable
product to run immediately when power is applied without
waiting for the battery to charge. If the load exceeds the
current limit at VBUS, VOUT will range between the no-load
voltage and slightly below the battery voltage, indicated
by the shaded region of Figure 1.
For very low-battery voltages, the battery charger acts like
a load and, due to limited input power, its current will tend
to pull VOUT below the 3.6V “instant-on” voltage. To prevent
VOUT from falling below this level, an undervoltage circuit
automatically detects that VOUT is falling and reduces the
battery charge as needed. This reduction ensures that load
current and output voltages are always priortized while
allowing as much battery charge current as possible. See
Over-Programming the Battery Charger in Applications
Information Section.
The power delivered from VBUS to VOUT is controlled by a
2.25MHz constant-frequency buck switching regulator. To
meet the USB maximum load speciﬁcation, the switching
regulator includes a control loop which ensures that the
average input current is below the level programmed at
CLPROG.
358623f
15
LTC3586-2/LTC3586-3
OPERATION
The current at CLPROG is a fraction (hCLPROG–1) of the
VBUS current. When a programming resistor and an averaging capacitor are connected from CLPROG to GND,
the voltage on CLPROG represents the average input
current of the PowerPath switching regulator. When the
input current approaches the programmed limit, CLPROG
reaches VCLPROG , 1.188V and power out is held constant.
The input current limit is programmed by the ILIM0 and
ILIM1 pins to limit average input current to one of several
possible settings as well as be deactivated (USB Suspend).
The input current limit will be set by the VCLPROG servo
voltage and the resistor on CLPROG according to the following expression:
IVBUS = IVBUSQ +
VCLPROG
• (hCLPROG + 1)
RCLPROG
Figure 1 shows the range of possible voltages at VOUT as
a function of battery voltage.
Ideal Diode from BAT to VOUT
The LTC3586-2/LTC3586-3 have an internal ideal diode as
well as a controller for an optional external ideal diode.
The ideal diode controller is always on and will respond
quickly whenever VOUT drops below BAT.
If the load current increases beyond the power allowed
from the switching regulator, additional power will be
pulled from the battery via the ideal diode. Furthermore,
if power to VBUS (USB or wall power) is removed, then
all of the application power will be provided by the battery via the ideal diode. The transition from input power
to battery power at VOUT will be quick enough to allow
only the10µF capacitor to keep VOUT from drooping. The
ideal diode consists of a precision ampliﬁer that enables
a large on-chip P-channel MOSFET transistor whenever
the voltage at VOUT is approximately 15mV (VFWD) below
the voltage at BAT. The resistance of the internal ideal
diode is approximately 180mΩ. If this is sufﬁcient for the
application, then no external components are necessary.
However, if more conductance is needed, an external
P-channel MOSFET transistor can be added from BAT to
VOUT. See Figure 2.
When an external P-channel MOSFET transistor is present, the GATE pin of the LTC3586-2/LTC3586-3 drive its
gate for automatic ideal diode control. The source of the
external P‑channel MOSFET should be connected to VOUT
and the drain should be connected to BAT. Capable of
driving a 1nF load, the GATE pin can control an external
P-channel MOSFET transistor having an on-resistance of
40mΩ or lower.
2200
4.5
1800
1600
CURRENT (mA)
VOUT (V)
3.9
3.6
NO LOAD
300mV
3.3
3.0
LTC3586-2/
LTC3586-3
IDEAL DIODE
1400
1200
1000
800
600
ON
SEMICONDUCTOR
MBRM120LT3
400
2.7
2.4
2.4
VISHAY Si2333
OPTIONAL EXTERNAL
IDEAL DIODE
2000
4.2
200
2.7
3.0
3.6
3.3
BAT (V)
3.9
4.2
358623 F01
Figure 1. VOUT vs BAT
0
0
60 120 180 240 300 360 420 480
FORWARD VOLTAGE (mV) (BAT – VOUT)
358623 F02
Figure 2. Ideal Diode Operation
358623f
16
LTC3586-2/LTC3586-3
OPERATION
Suspend LDO
VBUS Undervoltage Lockout (UVLO)
If the LTC3586-2/LTC3586-3 are conﬁgured for USB suspend mode, the switching regulator is disabled and the
suspend LDO provides power to the VOUT pin (presuming
there is power available to VBUS). This LDO will prevent the
battery from running down when the portable product has
access to a suspended USB port. Regulating at 4.6V, this
LDO only becomes active when the switching converter
is disabled (Suspended). To remain compliant with the
USB speciﬁcation, the input to the LDO is current limited
so that it will not exceed the 500µA low power suspend
speciﬁcation. If the load on VOUT exceeds the suspend
current limit, the additional current will come from the
battery via the ideal diode.
An internal undervoltage lockout circuit monitors VBUS and
keeps the PowerPath switching regulator off until VBUS
rises above 4.30V and is about 200mV above the battery
voltage. Hysteresis on the UVLO turns off the regulator if
VBUS drops below 4.00V or to within 50mV of BAT. When
this happens, system power at VOUT will be drawn from
the battery via the ideal diode.
3.3V Always-On Supply
The LTC3586-2/LTC3586-3 include a low quiescent current
low dropout regulator that is always powered. This LDO
can be used to provide power to a system pushbutton
controller, standby microcontroller or real-time clock. Designed to deliver up to 20mA, the always-on LDO requires
at least a 1µF low impedance ceramic bypass capacitor
for compensation. The LDO is powered from VOUT, and
therefore will enter dropout at loads less than 20mA as
VOUT falls near 3.3V. If the LDO3V3 output is not used, it
should be disabled by connecting it to VOUT.
TO USB
OR WALL
ADAPTER
Battery Charger
The LTC3586-2/LTC3586-3 include a constant-current/
constant-voltage battery charger with automatic recharge,
automatic termination by safety timer, low voltage trickle
charging, bad cell detection and thermistor sensor input
for out-of-temperature charge pausing.
Battery Preconditioning
When a battery charge cycle begins, the battery charger
ﬁrst determines if the battery is deeply discharged. If the
battery voltage is below VTRKL, typically 2.85V, an automatic
trickle charge feature sets the battery charge current to
10% of the programmed value. If the low voltage persists
for more than 1/2 hour, the battery charger automatically
terminates and indicates via the CHRG pin that the battery
was unresponsive.
VBUS
SW
35, 36
VOUT
PWM AND
GATE DRIVE
IDEAL
DIODE
ISWITCH/
hCLPROG
CONSTANT-CURRENT
CONSTANT-VOLTAGE
BATTERY CHARGER
15mV
CLPROG
1.188V
–
+
AVERAGE INPUT
CURRENT LIMIT
CONTROLLER
+
+
–
4
0.3V
3.6V
+–
–
+
+
–
GATE
BAT
3.5V TO
(BAT + 0.3V)
TO SYSTEM
LOAD
37
34
OPTIONAL
EXTERNAL
IDEAL DIODE
PMOS
31
32
AVERAGE OUTPUT
VOLTAGE LIMIT
CONTROLLER
+
SINGLE CELL
Li-Ion
358623 F03
Figure 3. PowerPath Block Diagram
358623f
17
LTC3586-2/LTC3586-3
OPERATION
Once the battery voltage is above 2.85V, the battery charger
begins charging in full power constant-current mode. The
current delivered to the battery will try to reach 1022V/
RPROG . Depending on available input power and external
load conditions, the battery charger may or may not be
able to charge at the full programmed rate. The external
load will always be prioritized over the battery charge
current. The USB current limit programming will always
be observed and only additional power will be available to
charge the battery. When system loads are light, battery
charge current will be maximized.
Charge Termination
The battery charger has a built-in safety timer. When the
voltage on the battery reaches the pre-programmed ﬂoat
voltage, the battery charger will regulate the battery voltage and the charge current will decrease naturally. Once
the battery charger detects that the battery has reached
the float voltage, the four hour safety timer is started.
After the safety timer expires, charging of the battery will
discontinue and no more current will be delivered.
Automatic Recharge
After the battery charger terminates, it will remain off
drawing only microamperes of current from the battery.
If the portable product remains in this state long enough,
the battery will eventually self discharge. To ensure that
the battery is always topped off, a charge cycle will automatically begin when the battery voltage falls below the
recharge threshold which is typically 100mV less than
the charger’s float voltage. In the event that the safety
timer is running when the battery voltage falls below the
recharge threshold, it will reset back to zero. To prevent
brief excursions below the recharge threshold from resetting the safety timer, the battery voltage must be below
the recharge threshold for more than 1.3ms. The charge
cycle and safety timer will also restart if the VBUS UVLO
cycles low and then high (e.g., VBUS is removed and then
replaced).
Charge Current
The charge current is programmed using a single resistor from PROG to ground. 1/1022th of the battery charge
current is sent to PROG which will attempt to servo to
1.000V. Thus, the battery charge current will try to reach
1022 times the current in the PROG pin. The program
resistor and the charge current are calculated using the
following equations:
RPROG =
1022V
1022V
, ICHG =
ICHG
RPROG
In either the constant-current or constant-voltage charging
modes, the voltage at the PROG pin will be proportional to
the actual charge current delivered to the battery. Therefore, the actual charge current can be determined at any
time by monitoring the PROG pin voltage and using the
following equation:
IBAT =
VPROG
• 1022
RPROG
In many cases, the actual battery charge current, IBAT, will
be lower than ICHG due to limited input power available and
prioritization with the system load drawn from VOUT.
Charge Status Indication
The CHRG pin indicates the status of the battery charger.
Four possible states are represented by CHRG which include charging, not charging, unresponsive battery, and
battery temperature out of range.
The signal at the CHRG pin can be easily recognized as
one of the above four states by either a human or a microprocessor. An open-drain output, the CHRG pin can
drive an indicator LED through a current limiting resistor
for human interfacing or simply a pull-up resistor for
microprocessor interfacing.
358623f
18
LTC3586-2/LTC3586-3
OPERATION
To make the CHRG pin easily recognized by both humans
and microprocessors, the pin is either LOW for charging,
HIGH for not charging, or it is switched at high frequency
(35kHz) to indicate the two possible faults, unresponsive
battery and battery temperature out of range.
When charging begins, CHRG is pulled low and remains
low for the duration of a normal charge cycle. When
charging is complete, i.e., the BAT pin reaches the float
voltage and the charge current has dropped to one tenth
of the programmed value, the CHRG pin is released (Hi‑Z).
If a fault occurs, the pin is switched at 35kHz. While
switching, its duty cycle is modulated between a high
and low value at a very low frequency. The low and high
duty cycles are disparate enough to make an LED appear
to be on or off thus giving the appearance of “blinking”.
Each of the two faults has its own unique “blink” rate for
human recognition as well as two unique duty cycles for
machine recognition.
The CHRG pin does not respond to the C/10 threshold if
the LTC3586-2/LTC3586-3 are in VBUS current limit. This
prevents false end-of-charge indications due to insufﬁcient
power available to the battery charger.
Table 3 illustrates the four possible states of the CHRG
pin when the battery charger is active.
Table 3. CHRG Signal
STATUS
Charging
Not Charging
NTC Fault
Bad Battery
FREQUENCY
0Hz
0Hz
35kHz
35kHz
MODULATION
(BLINK) FREQUENCY
DUTY CYCLES
0Hz (Lo-Z)
100%
0Hz (Hi-Z)
0%
1.5Hz at 50%
6.25% to 93.75%
6.1Hz at 50%
12.5% to 87.5%
An NTC fault is represented by a 35kHz pulse train whose
duty cycle varies between 6.25% and 93.75% at a 1.5Hz
rate. A human will easily recognize the 1.5Hz rate as a
“slow” blinking which indicates the out-of-range battery
temperature while a microprocessor will be able to decode
either the 6.25% or 93.75% duty cycles as an NTC fault.
If a battery is found to be unresponsive to charging (i.e.,
its voltage remains below 2.85V for 1/2 hour), the CHRG
pin gives the battery fault indication. For this fault, a human
would easily recognize the frantic 6.1Hz “fast” blink of the
LED while a microprocessor would be able to decode either
the 12.5% or 87.5% duty cycles as a bad battery fault.
Note that the LTC3586-2/LTC3586-3 are 3-terminal
PowerPath products where system load is always prioritized over battery charging. Due to excessive system
load, there may not be sufﬁcient power to charge the
battery beyond the trickle charge threshold voltage
within the bad battery timeout period. In this case, the
battery charger will falsely indicate a bad battery. System
software may then reduce the load and reset the battery
charger to try again.
Although very improbable, it is possible that a duty cycle
reading could be taken at the bright-dim transition (low
duty cycle to high duty cycle). When this happens the
duty cycle reading will be precisely 50%. If the duty cycle
reading is 50%, system software should disqualify it and
take a new duty cycle reading.
NTC Thermistor
The battery temperature is measured by placing a negative temperature coefﬁcient (NTC) thermistor close to the
battery pack.
To use this feature, connect the NTC thermistor, RNTC ,
between the NTC pin and ground and a resistor, RNOM ,
from VBUS to the NTC pin. RNOM should be a 1% resistor with a value equal to the value of the chosen NTC
thermistor at 25°C (R25). A 100k thermistor is recommended since thermistor current is not measured by the
LTC3586-2/LTC3586-3 and will have to be budgeted for
USB compliance.
The LTC3586-2/LTC3586-3 will pause charging when the
resistance of the NTC thermistor drops to 0.54 times the
value of R25 or approximately 54k. For Vishay “Curve 1”
thermistor, this corresponds to approximately 40°C. If the
battery charger is in constant voltage (ﬂoat) mode, the
safety timer also pauses until the thermistor indicates a
return to a valid temperature. As the temperature drops,
the resistance of the NTC thermistor rises. The LTC3586‑2/
LTC3586-3 are also designed to pause charging when the
value of the NTC thermistor increases to 3.25 times the
value of R25. For Vishay “Curve 1” this resistance, 325k,
corresponds to approximately 0°C. The hot and cold
comparators each have approximately 3°C of hysteresis
to prevent oscillation about the trip point. Grounding the
NTC pin disables the NTC charge pausing function.
358623f
19
LTC3586-2/LTC3586-3
OPERATION
Thermal Regulation
To optimize charging time, an internal thermal feedback
loop may automatically decrease the programmed charge
current. This will occur if the die temperature rises to
approximately 110°C. Thermal regulation protects the
LTC3586-2/LTC3586-3 from excessive temperature due to
high power operation or high ambient thermal conditions
and allows the user to push the limits of the power handling capability with a given circuit board design without
risk of damaging the LTC3586-2/LTC3586-3 or external
components. The beneﬁt of the LTC3586-2/LTC3586-3
thermal regulation loop is that charge current can be set
according to actual conditions rather than worst-case
conditions with the assurance that the battery charger will
automatically reduce the current in worst-case conditions.
A flow chart of battery charger operation can be seen in
Figure 4.
Low Supply Operation
The LTC3586-2/LTC3586-3 incorporate an undervoltage
lockout circuit on VOUT which shuts down all four general
purpose switching regulators when VOUT drops below
VOUTUVLO. This UVLO prevents unstable operation.
FAULT Pin
FAULT is an open-drain output used to indicate a fault
condition on any of the general purpose regulators. If
the FB pin voltage of any of the enabled regulators stays
below 92% of the internal reference voltage (0.8V) for
more than 14ms, a fault condition will be reported by
FAULT going low. Since FAULT is an open-drain output,
it requires a pull-up resistor to the input voltage of the
monitoring microprocessor or another appropriate power
source such as LD03V3.
General Purpose Buck Switching Regulators
The LTC3586-2/LTC3586-3 contain two 2.25MHz constantfrequency current mode buck switching regulators. Each
buck regulator can provide up to 400mA of output current.
Both buck regulators can be programmed for a minimum
output voltage of 0.8V and can be used to power a microcontroller core, microcontroller I/O, memory, disk drive or
other logic circuitry. Both buck converters support 100%
duty cycle operation (low dropout mode) when their input
voltage drops very close to their output voltage. To suit a
variety of applications, selectable mode functions can be
used to trade-off noise for efﬁciency. Two modes are available to control the operation of the LTC3586-2/LTC3586-3’s
buck regulators. At moderate to heavy loads, the pulseskipping mode provides the least noise switching solution.
At lighter loads, Burst Mode operation may be selected.
The buck regulators include soft-start to limit inrush current when powering on, short-circuit current protection
and switch node slew limiting circuitry to reduce radiated
EMI. No external compensation components are required.
The operating mode of the buck regulators can be set by
the MODE pin. The buck converters can be individually
enabled by the EN1 and EN2 pins. Both buck regulators
have a fixed feedback servo voltage of 800mV. The buck
regulator input supplies VIN1 and VIN2 will generally be
connected to the system load pin VOUT.
Buck Regulator Output Voltage Programming
Both buck regulators can be programmed for output voltages greater than 0.8V. The output voltage for each buck
regulator is programmed using a resistor divider from the
buck regulator output connected to the feedback pins (FB1
and FB2) such that:
 R1 
VOUTX = VFBX  +1
 R2 
where VFB is ﬁxed at 0.8V and X = 1, 2. See Figure 5.
Typical values for R1 are in the range of 40k to 1M. The
capacitor CFB cancels the pole created by feedback resistors
and the input capacitance of the FBx pin and also helps
to improve transient response for output voltages much
greater than 0.8V. A variety of capacitor sizes can be used
for CFB but a value of 10pF is recommended for most applications. Experimentation with capacitor sizes between
2pF and 22pF may yield improved transient response.
358623f
20
LTC3586-2/LTC3586-3
OPERATION
POWER ON
CLEAR EVENT TIMER
ASSERT CHRG LOW
NTC OUT OF RANGE
YES
INHIBIT CHARGER
NO
BAT < 2.85V
BATTERY STATE
BAT > 4.15V
CHRG CURRENTLY
HIGH-Z
2.85V < BAT < 4.15V
NO
NO
CHARGE AT
100V/RPROG (C/10 RATE)
CHARGE AT
1022V/RPROG RATE
CHARGE WITH
FIXED VOLTAGE
(4.200V)
RUN EVENT TIMER
PAUSE EVENT TIMER
RUN EVENT TIMER
TIMER > 30 MINUTES
TIMER > 4 HOURS
YES
YES
INDICATE
NTC FAULT
AT CHRG
NO
YES
INHIBIT CHARGING
IBAT < C/10
STOP CHARGING
NO
YES
INDICATE BATTERY
FAULT AT CHRG
BAT RISING
THROUGH 4.1V
YES
RELEASE CHRG
HIGH-Z
RELEASE CHRG
HIGH-Z
NO
BAT > 2.85V
YES
NO
BAT FALLING
THROUGH 4.1V
NO
YES
BAT < 4.1V
NO
YES
358623 F04
Figure 4. Flow Chart for Battery Charger Operation (LTC3586-2)
358623f
21
LTC3586-2/LTC3586-3
OPERATION
VINx
L
SWx
LTC3586-2/
LTC3586-3
FBx
VOUTx
CFB
R1
COUT
R2
X = 1, 2
GND
358623 F05
Figure 5. Buck Converter Application Circuit
Buck Regulator Operating Modes
The LTC3586-2/LTC3586-3’s buck regulators include two
possible operating modes to meet the noise/ power needs
of a variety of applications.
In pulse-skipping mode, an internal latch is set at the
start of every cycle which turns on the main P-channel
MOSFET switch. During each cycle, a current comparator compares the peak inductor current to the output of
an error ampliﬁer. The output of the current comparator
resets the internal latch which causes the main P-channel
MOSFET switch to turn off and the N-channel MOSFET
synchronous rectiﬁer to turn on. The N-channel MOSFET
synchronous rectiﬁer turns off at the end of the 2.25MHz
cycle or if the current through the N-channel MOSFET
synchronous rectiﬁer drops to zero. Using this method
of operation, the error ampliﬁer adjusts the peak inductor
current to deliver the required output power. All necessary compensation is internal to the switching regulator
requiring only a single ceramic output capacitor for stability. At light loads, the inductor current may reach zero
on each pulse which will turn off the N-channel MOSFET
synchronous rectiﬁer. In this case, the switch node (SW1,
SW2) goes high impedance and the switch node voltage
will “ring”. This is discontinuous mode operation, and is
normal behavior for a switching regulator. At very light
loads, the buck regulators will automatically skip pulses
as needed to maintain output regulation.
At high duty cycles (VOUTx > VINx /2) it is possible for the
inductor current to reverse, causing the buck regulator
to operate continuously at light loads. This is normal and
regulation is maintained, but the supply current will increase
to several milliamperes due to continuous switching.
In Burst Mode operation, the buck regulator automatically switches between fixed frequency PWM operation
and hysteretic control as a function of the load current.
At light loads, the buck regulators operate in hysteretic
mode in which the output capacitor is charged to a voltage slightly higher than the regulation point. The buck
converter then goes into sleep mode, during which the
output capacitor provides the load current. In sleep mode,
most of the regulator’s circuitry is powered down, helping
conserve battery power. When the output voltage drops
below a predetermined value, the buck regulator circuitry
is powered on and the normal PWM operation resumes.
The duration for which the buck regulator operates in
sleep mode depends on the load current. The sleep time
decreases as the load current increases. Beyond a certain
load current point (about 1/4 rated output load current)
the step-down switching regulators will switch to a low
noise constant frequency PWM mode of operation, much
the same as pulse-skipping operation at high loads. For
applications that can tolerate some output ripple at low
output currents, Burst Mode operation provides better
efficiency than pulse skip at light loads while still providing the full specified output current of the buck regulator.
The buck regulators allow mode transition on the ﬂy,
providing seamless transition between modes even under
load. This allows the user to switch back and forth between
modes to reduce output ripple or increase low current
efﬁciency as needed.
Buck Regulator in Shutdown
The buck regulators are in shutdown when not enabled for
operation. In shutdown, all circuitry in the buck regulator
is disconnected from the buck regulator input supply
leaving only a few nanoamps of leakage current. The
buck regulator outputs are individually pulled to ground
through a 10k resistor on the switch pins (SW1 and SW2)
when in shutdown.
Buck Regulator Dropout Operation
It is possible for a buck regulator’s input voltage, VINx , to
approach its programmed output voltage (e.g., a battery
voltage of 3.4V with a programmed output voltage of 3.3V).
358623f
22
LTC3586-2/LTC3586-3
OPERATION
When this happens, the PMOS switch duty cycle increases
until it is turned on continuously at 100%. In this dropout
condition, the respective output voltage equals the buck
regulator’s input voltage minus the voltage drops across
the internal P-channel MOSFET and the inductor.
Buck Regulator Soft-Start Operation
Soft-start is accomplished by gradually increasing the
peak inductor current for each buck regulator over a 500µs
period. This allows each output to rise slowly, helping
minimize the battery in-rush current. A soft-start cycle
occurs whenever a given buck regulator is enabled, or
after a fault condition has occurred (thermal shutdown
or UVLO). A soft-start cycle is not triggered by changing
operating modes. This allows seamless output operation
when transitioning between modes.
Buck Regulator Switching Slew Rate Control
The buck regulators contain new patent pending circuitry
to limit the slew rate of the switch node (SW1 and SW2).
This new circuitry is designed to transition the switch node
over a period of a couple of nanoseconds, signiﬁcantly
reducing radiated EMI and conducted supply noise.
BUCK-BOOST DC/DC SWITCHING REGULATOR
The LTC3586-2/LTC3586-3 contain a 2.25MHz constantfrequency voltage-mode buck-boost switching regulator.
The regulator provides up to 1A of output load current.
The buck-boost can be programmed to a minimum output
voltage of 2.5V and can be used to power a microcontroller core, microcontroller I/O, memory, disk drive, or
other logic circuitry. The converter is enabled by pulling
EN3 high. To suit a variety of applications, a selectable
mode function allows the user to trade-off noise for efficiency. Two modes are available to control the operation
of the LTC3586-2/LTC3586-3’s buck-boost regulator. At
moderate to heavy loads, the constant frequency PWM
mode provides the least noise switching solution. At
lighter loads Burst Mode operation may be selected. The
output voltage is programmed by a user-supplied resistive
divider returned to FB3. An error amplifier compares the
divided output voltage with a reference and adjusts the
compensation voltage accordingly until the FB3 pin has
stabilized to the reference voltage (0.8V). The buck-boost
regulator includes a soft-start to limit inrush current and
voltage overshoot when powering on, short-circuit current protection, and switch node slew limiting circuitry
for reduced radiated EMI.
Input Current Limit
The input current limit comparator will shut the input
PMOS switch off once current exceeds 2.5A (typical). The
2.5A input current limit also protects against a grounded
VOUT3 node.
Output Overvoltage Protection
If the FB3 node were inadvertently shorted to ground, then
the output would increase indefinitely with the maximum
current that could be sourced from VIN3 . The LTC3586-2/
LTC3586-3 protect against this by shutting off the input
PMOS if the output voltage exceeds 5.6V (typical).
Low Output Voltage Operation
When the output voltage is below 2.65V (typical) during
start-up, Burst Mode operation is disabled and switch D
is turned off (allowing forward current through the well
diode and limiting reverse current to 0mA).
Buck-Boost Regulator PWM Operating Mode
In PWM mode the voltage seen at FB3 is compared to the
reference voltage (0.8V). From the FB3 voltage an error
amplifier generates an error signal seen at VC3 . This error
signal commands PWM waveforms that modulate switches
A, B, C, and D. Switches A and B operate synchronously
as do switches C and D. If VIN3 is significantly greater
than the programmed VOUT3 , then the converter will operate in buck mode. In this case switches A and B will be
modulated, with switch D always on (and switch C always
off), to step-down the input voltage to the programmed
output. If VIN3 is significantly less than the programmed
VOUT3 , then the converter will operate in boost mode. In
this case switches C and D are modulated, with switch A
358623f
23
LTC3586-2/LTC3586-3
OPERATION
always on (and switch B always off), to step-up the input
voltage to the programmed output. If VIN3 is close to the
programmed VOUT3 , then the converter will operate in
4‑switch mode. In this case the switches sequence through
the pattern of AD, AC, BD to either step the input voltage
up or down to the programmed output.
Buck-Boost Regulator Burst-Mode Operation
In Burst Mode operation, the buck-boost regulator uses
a hysteretic FB3 voltage algorithm to control the output
voltage. By limiting FET switching and using a hysteretic
control loop, switching losses are greatly reduced. In this
mode output current is limited to 50mA typical. While
operating in Burst Mode operation, the output capacitor
is charged to a voltage slightly higher than the regulation
point. The buck-boost converter then goes into a sleep
state, during which the output capacitor provides the load
current. The output capacitor is charged by charging the
inductor until the input current reaches 250mA typical
and then discharging the inductor until the reverse current
reaches 0mA typical. This process is repeated until the
feedback voltage has charged to 6mV above the regulation
point. In the sleep state, most of the regulator’s circuitry
is powered down, helping to conserve battery power.
When the feedback voltage drops 6mV below the regulation point, the switching regulator circuitry is powered on
and another burst cycle begins. The duration for which
the regulator sleeps depends on the load current and
output capacitor value. The sleep time decreases as the
load current increases. The buck-boost regulator will not
go to sleep if the current is greater than 50mA, and if the
load current increases beyond this point while in Burst
Mode operation the output will lose regulation. Burst
Mode operation provides a significant improvement in
efficiency at light loads at the expense of higher output ripple
when compared to PWM mode. For many noise-sensitive
systems, Burst Mode operation might be undesirable at
certain times (i.e., during a transmit or receive cycle of a
wireless device), but highly desirable at others (i.e., when
the device is in low power standby mode). The MODE pin
is used to enable or disable Burst Mode operation at any
time, offering both low noise and low power operation
when they are needed.
Buck-Boost Regulator Soft-Start Operation
Soft-start is accomplished by gradually increasing the
maximum VC3 voltage over a 0.5ms (typical) period.
Ramping the VC3 voltage limits the duty cycle and thus
the VOUT3 voltage minimizing output overshoot during
startup. A soft-start cycle occurs whenever the buck-boost
is enabled, or after a fault condition has occurred (thermal
shutdown or UVLO). A soft-start cycle is not triggered by
changing operating modes. This allows seamless output
operation when transitioning between Burst Mode operation and PWM mode.
SYNCHRONOUS BOOST DC/DC SWITCHING
REGULATOR
The LTC3586-2/LTC3586-3 contain a 2.25MHz constantfrequency current mode synchronous boost switching
regulator with true output disconnect feature. The regulator
provides at least 800mA of output load current and the
output voltage can be programmed up to a maximum of
5V. The converter is enabled by pulling EN4 high. The
boost regulator also includes soft-start to limit inrush
current and voltage overshoot when powering on, short
circuit current protection and switch node slew limiting
circuitry for reduced radiated EMI.
Error Amp
The boost output voltage is programmed by a user-supplied resistive divider returned to the FB4 pin. An internally
compensated error amplifier compares the divided output
voltage with an internal 0.8V reference and adjusts the
voltage accordingly until FB4 servos to 0.8V.
Current Limit
Lossless current sensing converts the NMOS switch current signal to a voltage to be summed with the internal
slope compensation signal. The summed signal is then
compared to the error amplifier output to provide a peak
current control command for the peak comparator. Peak
switch current is limited to 2.8A independent of output
voltage.
358623f
24
LTC3586-2/LTC3586-3
OPERATION
Zero Current Comparator
The zero current comparator monitors the inductor current
to the output and shuts off the synchronous rectifier once
the current drops to approximately 65mA. This prevents
the inductor current from reversing in polarity thereby
improving efficiency at light loads.
Antiringing Control
The antiringing control circuitry prevents high frequency
ringing of the SW pin as the inductor current goes to zero
in discontinuous mode. The damping of the resonant
circuit formed by L and CSW (capacitance of the SW4
pin) is achieved internally by switching a 150Ω resistor
across the inductor.
PMOS Synchronous Rectifier
To prevent the inductor current from running away,
the PMOS synchronous rectifier is only enabled when
VOUT > (VIN + 130mV).
Output Disconnect and Inrush Limiting
The LTC3586-2/LTC3586-3 boost converter is designed to
allow true output disconnect by eliminating body diode
conduction of the internal PMOS rectifier. This allows VOUT
to go to zero volts during shutdown, drawing zero current
from the input source. It also allows for inrush current
limiting at start-up, minimizing surge currents seen by the
input supply. Note that to obtain the advantage of output
disconnect, there must not be an external Schottky diode
connected between the SW4 and VOUT4 pin.
Short-Circuit Protection
Unlike most boost converters, the LTC3586-2/LTC3586‑3
boost converter allows its output to be short-circuited
due to the output disconnect feature. It incorporates
internal features such as current limit foldback and thermal
shutdown for protection from an excessive overload or
short circuit.
VIN > VOUT Operation
The LTC3586-2/LTC3586-3 boost converter will maintain
voltage regulation even if the input voltage is above
the output voltage. This is achieved by terminating the
switching of the synchronous PMOS and applying VIN4
statically on its gate. This ensures that the slope of the
inductor current will reverse during the time when current is flowing to the output. Since the PMOS no longer
acts as a low impedance switch in this mode, there will
be more power dissipation within the IC. This will cause
a sharp drop in the efficiency (see Typical Performance
Characteristics, Boost Efficiency vs VIN4). The maximum
output current should be limited in order to maintain an
acceptable junction temperature.
Boost Soft-Start
The LTC3586-2/LTC3586-3 boost converter provides softstart by slowly ramping the peak inductor current from
zero to a maximum of 2.8A in about 500µs. Ramping the
peak inductor current limits transient inrush currents
during start-up. A soft-start cycle occurs whenever the
boost is enabled, or after a fault condition has occurred
(thermal shutdown or UVLO).
Boost Overvoltage Protection
If the FB4 node were inadvertently shorted to ground, then
the boost converter output would increase indefinitely with
the maximum current that could be sourced from VIN4 . The
LTC3586-2/LTC3586-3 protects against this by shutting
off the main switch if the output voltage exceeds 5.5V.
358623f
25
LTC3586-2/LTC3586-3
APPLICATIONS INFORMATION
PowerPath CONTROLLER APPLICATIONS SECTION
VBUS and VOUT Bypass Capacitors
CLPROG Resistor and Capacitor
The style and value of capacitors used with the LTC3586‑2/
LTC3586-3 determine several important parameters
such as regulator control-loop stability and input voltage
ripple. Because the LTC3586-2/LTC3586-3 use a buck
switching power supply from VBUS to VOUT, its input
current waveform contains high frequency components.
It is strongly recommended that a low equivalent series
resistance (ESR) multilayer ceramic capacitor be used to
bypass VBUS . Tantalum and aluminum capacitors are not
recommended because of their high ESR. The value of the
capacitor on VBUS directly controls the amount of input
ripple for a given load current. Increasing the size of this
capacitor will reduce the input ripple.
As described in the High Efﬁciency Switching PowerPath
Controller section, the resistor on the CLPROG pin determines the average input current limit when the switching
regulator is set to either the 1x mode (USB 100mA), the
5x mode (USB 500mA) or the 10x mode. The input current will be comprised of two components, the current
that is used to drive VOUT and the quiescent current of the
switching regulator. To ensure that the USB speciﬁcation is
strictly met, both components of input current should be
considered. The Electrical Characteristics table gives the
worst-case values for quiescent currents in either setting
as well as current limit programming accuracy. To get as
close to the 500mA or 100mA speciﬁcations as possible,
a 1% resistor should be used. Recall that IVBUS = IVBUSQ
+ VCLPROG/RCLPPROG • (hCLPROG +1).
An averaging capacitor is required in parallel with the
CLPROG resistor so that the switching regulator can
determine the average input current. This network also
provides the dominant pole for the feedback loop when
current limit is reached. To ensure stability, the capacitor
on CLPROG should be 0.1µF.
Choosing the PowerPath Inductor
Because the input voltage range and output voltage range
of the power path switching regulator are both fairly narrow, the LTC3586-2/LTC3586-3 are designed for a specific
inductance value of 3.3µH. Some inductors which may be
suitable for this application are listed in Table 4.
Table 4. Recommended Inductors for PowerPath Controller
INDUCTOR L
TYPE
(µH)
MAX
IDC
(A)
MAX
DCR
(Ω)
0.08
SIZE IN mm
(L × W × H)
MANUFACTURER
LPS4018
3.3
2.2
3.9 × 3.9 × 1.7 Coilcraft
www.coilcraft.com
D53LC
DB318C
3.3
3.3
2.26 0.034
Toko
5×5×3
1.55 0.070 3.8 × 3.8 × 1.8 www.toko.com
WE-TPC
Type M1
3.3
1.95 0.065 4.8 × 4.8 × 1.8 Wurth Elektronik
www.we-online.com
CDRH6D12
CDRH6D38
3.3
3.3
2.2 0.0625 6.7 × 6.7 × 1.5 Sumida
3.5 0.020
www.sumida.com
7×7×4
To prevent large VOUT voltage steps during transient load
conditions, it is also recommended that a ceramic capacitor be used to bypass VOUT. The output capacitor is used
in the compensation of the switching regulator. At least
4µF of actual capacitance with low ESR are required on
VOUT. Additional capacitance will improve load transient
performance and stability.
Multilayer ceramic chip capacitors typically have exceptional ESR performance. MLCCs combined with a tight
board layout and an unbroken ground plane will yield very
good performance and low EMI emissions.
There are several types of ceramic capacitors available
each having considerably different characteristics. For
example, X7R ceramic capacitors have the best voltage and
temperature stability. X5R ceramic capacitors have apparently higher packing density but poorer performance over
their rated voltage and temperature ranges. Y5V ceramic
capacitors have the highest packing density, but must be
used with caution, because of their extreme non-linear
characteristic of capacitance verse voltage. The actual
in-circuit capacitance of a ceramic capacitor should be
measured with a small AC signal as is expected in-circuit.
Many vendors specify the capacitance verse voltage with
a 1V RMS AC test signal and as a result overstate the capacitance that the capacitor will present in the application.
Using similar operating conditions as the application, the
user must measure or request from the vendor the actual
capacitance to determine if the selected capacitor meets
the minimum capacitance that the application requires.
358623f
26
LTC3586-2/LTC3586-3
APPLICATIONS INFORMATION
Over-Programming the Battery Charger
The USB high power speciﬁcation allows for up to 2.5W to
be drawn from the USB port (5V • 500mA). The PowerPath
switching regulator transforms the voltage at VBUS to just
above the voltage at BAT with high efﬁciency, while limiting
power to less than the amount programmed at CLPROG.
In some cases the battery charger may be programmed
(with the PROG pin) to deliver the maximum safe charging
current without regard to the USB speciﬁcations. If there
is insufﬁcient current available to charge the battery at the
programmed rate, the PowerPath regulator will reduce
charge current until the system load on VOUT is satisﬁed
and the VBUS current limit is satisﬁed. Programming the
battery charger for more current than is available will
not cause the average input current limit to be violated.
It will merely allow the battery charger to make use of
all available power to charge the battery as quickly as
possible, and with minimal power dissipation within the
battery charger.
Alternate NTC Thermistors and Biasing
The LTC3586-2/LTC3586-3 provide temperature qualiﬁed
charging if a grounded thermistor and a bias resistor
are connected to NTC. By using a bias resistor whose
value is equal to the room temperature resistance of the
thermistor (R25) the upper and lower temperatures are
pre-programmed to approximately 40°C and 0°C, respectively (assuming a Vishay “Curve 1” thermistor).
The upper and lower temperature thresholds can be adjusted by either a modiﬁcation of the bias resistor value
or by adding a second adjustment resistor to the circuit.
If only the bias resistor is adjusted, then either the upper
or the lower threshold can be modiﬁed but not both. The
other trip point will be determined by the characteristics
of the thermistor. Using the bias resistor in addition to an
adjustment resistor, both the upper and the lower temperature trip points can be independently programmed with
the constraint that the difference between the upper and
lower temperature thresholds cannot decrease. Examples
of each technique are given below.
NTC thermistors have temperature characteristics which
are indicated on resistance-temperature conversion tables.
The Vishay-Dale thermistor NTHS0603N011-N1003F, used
in the following examples, has a nominal value of 100k
and follows the Vishay “Curve 1” resistance-temperature
characteristic.
In the explanation below, the following notation is used.
R25 = Value of the Thermistor at 25°C
RNTC|COLD = Value of thermistor at the cold trip point
RNTC|HOT = Value of the thermistor at the hot trip point
rCOLD = Ratio of RNTC|COLD to R25
rHOT = Ratio of RNTC|COLD to R25
RNOM = Primary thermistor bias resistor
(see Figure 6a)
R1 = Optional temperature range adjustment resistor
(see Figure 6b)
The trip points for the LTC3586-2/LTC3586-3’s temperature
qualiﬁcation are internally programmed at 0.349 • VBUS for
the hot threshold and 0.765 • VBUS for the cold threshold.
Therefore, the hot trip point is set when:
RNTC|HOT
RNOM + RNTC|HOT
• VBUS = 0.349 • VBUS
and the cold trip point is set when:
RNTC|COLD
RNOM + RNTC|COLD
• VBUS = 0.765 • VBUS
Solving these equations for RNTC|COLD and RNTC|HOT results
in the following:
RNTC|HOT = 0.536 • RNOM
and
RNTC|COLD = 3.25 • RNOM
By setting RNOM equal to R25, the above equations result
in rHOT = 0.536 and rCOLD = 3.25. Referencing these ratios
to the Vishay Resistance-Temperature Curve 1 chart gives
a hot trip point of about 40°C and a cold trip point of about
0°C. The difference between the hot and cold trip points
is approximately 40°C.
358623f
27
LTC3586-2/LTC3586-3
APPLICATIONS INFORMATION
By using a bias resistor, RNOM , different in value from
R25, the hot and cold trip points can be moved in either
direction. The temperature span will change somewhat due
to the non-linear behavior of the thermistor. The following
equations can be used to easily calculate a new value for
the bias resistor:
RNOM =
rHOT
• R25
0.536
RNOM =
rCOLD
• R25
3.25
From the Vishay Curve 1 R-T characteristics, rHOT is
0.2488 at 60°C. Using the above equation, RNOM should
be set to 46.4k. With this value of RNOM , the cold trip point
is about 16°C. Notice that the span is now 44°C rather
than the previous 40°C. This is due to the decrease in
VBUS
RNOM
100k
NTC
0.765 • VBUS
T
LTC3586-2/LTC3586-3
NTC BLOCK
–
+
RNTC
100k
–
0.349 • VBUS
For example, to set the trip points to 0°C and 45°C with
a Vishay Curve 1 thermistor choose:
RNOM =
3.266 – 0.4368
• 100k = 104.2k
2.714
the nearest 1% value is 105k:
R1 = 0.536 • 105k – 0.4368 • 100k = 12.6k
the nearest 1% value is 12.7k. The ﬁnal circuit is shown
in Figure 6b and results in an upper trip point of 45°C and
a lower trip point of 0°C.
VBUS
RNOM
105k
NTC
0.765 • VBUS
LTC3586-2/LTC3586-3
NTC BLOCK
–
TOO_COLD
5
+
R1
12.7k
–
TOO_HOT
+
rCOLD – rHOT
• R25
2.714
R1= 0.536 • RNOM – rHOT • R25
VBUS
TOO_COLD
5
The upper and lower temperature trip points can be independently programmed by using an additional bias resistor
as shown in Figure 6b. The following formulas can be used
to compute the values of RNOM and R1:
RNOM =
where rHOT and rCOLD are the resistance ratios at the desired hot and cold trip points. Note that these equations
are linked. Therefore, only one of the two trip points can
be chosen, the other is determined by the default ratios
designed in the IC. Consider an example where a 60°C
hot trip point is desired.
VBUS
“temperature gain” of the thermistor as absolute temperature increases.
T
RNTC
100k
0.349 • VBUS
TOO_HOT
+
+
+
NTC_ENABLE
0.017 • VBUS
–
NTC_ENABLE
0.017 • VBUS
–
358623 F06b
358623 F06a
(6a)
(6b)
Figure 6. NTC Circuits
358623f
28
LTC3586-2/LTC3586-3
APPLICATIONS INFORMATION
USB Inrush Limiting
When a USB cable is plugged into a portable product,
the inductance of the cable and the high-Q ceramic input
capacitor form an L-C resonant circuit. If the cable does
not have adequate mutual coupling or if there is not much
impedance in the cable, it is possible for the voltage at
the input of the product to reach as high as twice the
USB voltage (~10V) before it settles out. In fact, due to
the high voltage coefﬁcient of many ceramic capacitors, a
nonlinearity, the voltage may even exceed twice the USB
voltage. To prevent excessive voltage from damaging the
LTC3586-2/LTC3586-3 during a hot insertion, it is best to
have a low voltage coefﬁcient capacitor at the VBUS pin to
the LTC3586-2/LTC3586-3. This is achievable by selecting
an MLCC capacitor that has a higher voltage rating than
that required for the application. For example, a 16V, X5R,
10µF capacitor in a 1206 case would be a better choice
than a 6.3V, X5R, 10µF capacitor in a smaller 0805 case.
Alternatively, the soft connect circuit (Figure 7) can be
employed. In this circuit, capacitor C1 holds MP1 off
when the cable is ﬁrst connected. Eventually C1 begins
to charge up to the USB input voltage applying increasing
gate support to MP1. The long time constant of R1 and
C1 prevent the current from building up in the cable too
fast thus dampening out any resonant overshoot.
disconnected, a 4.7µF capacitor in series with a 0.2Ω to
1Ω resistor from BAT to GND is required to keep ripple
voltage low.
High value, low ESR multilayer ceramic chip capacitors
reduce the constant-voltage loop phase margin, possibly
resulting in instability. Ceramic capacitors up to 22µF may
be used in parallel with a battery, but larger ceramics should
be decoupled with 0.2Ω to 1Ω of series resistance.
In constant-current mode, the PROG pin is in the feedback loop rather than the battery voltage. Because of the
additional pole created by any PROG pin capacitance,
capacitance on this pin must be kept to a minimum. With
no additional capacitance on the PROG pin, the battery
charger is stable with program resistor values as high
as 25k. However, additional capacitance on this node
reduces the maximum allowed program resistor. The pole
frequency at the PROG pin should be kept above 100kHz.
Therefore, if the PROG pin has a parasitic capacitance,
CPROG, the following equation should be used to calculate
the maximum resistance value for RPROG:
RPROG ≤
1
2π • 100kHz • CPROG
BUCK REGULATOR APPLICATIONS SECTION
Battery Charger Stability Considerations
Buck Regulator Inductor Selection
The LTC3586-2/LTC3586-3’s battery charger contains both
a constant-voltage and a constant-current control loop.
The constant-voltage loop is stable without any compensation when a battery is connected with low impedance
leads. Excessive lead length, however, may add enough
series inductance to require a bypass capacitor of at least
1µF from BAT to GND. Furthermore, when the battery is
Many different sizes and shapes of inductors are available from numerous manufacturers. Choosing the right
inductor from such a large selection of devices can be
overwhelming, but following a few basic guidelines will
make the selection process much simpler.
MP1
Si2333
5V USB
INPUT
VBUS
C1
100nF
USB CABLE
R1
40k
C2
10µF
LTC3586-2/
LTC3586-3
GND
358623 F07
Figure 7. USB Soft Connect Circuit
The buck converters are designed to work with inductors
in the range of 2.2µH to 10µH. For most applications a
4.7µH inductor is suggested for both buck regulators.
Larger value inductors reduce ripple current which improves output ripple voltage. Lower value inductors result
in higher ripple current and improved transient response
time. To maximize efﬁciency, choose an inductor with a
low DC resistance. For a 1.2V output, efﬁciency is reduced
about 2% for 100mΩ series resistance at 400mA load current, and about 2% for 300mΩ series resistance at 100mA
358623f
29
LTC3586-2/LTC3586-3
APPLICATIONS INFORMATION
load current. Choose an inductor with a DC current rating
at least 1.5 times larger than the maximum load current to
ensure that the inductor does not saturate during normal
operation. If output short circuit is a possible condition,
the inductor should be rated to handle the maximum peak
current speciﬁed for the buck converters.
Different core materials and shapes will change the size/
current and price/current relationship of an inductor. Toroid
or shielded pot cores in ferrite or Permalloy materials are
small and don’t radiate much energy, but generally cost
more than powdered iron core inductors with similar
electrical characteristics. Inductors that are very thin or
have a very small volume typically have much higher core
and DCR losses, and will not give the best efﬁciency. The
choice of which style inductor to use often depends more
on the price vs size, performance and any radiated EMI
requirements than on what the LTC3586-2/LTC3586-3
require to operate.
The inductor value also has an effect on Burst Mode
operations. Lower inductor values will cause the Burst
Mode switching frequencies to increase.
Buck Regulator Input/Output Capacitor Selection
Low ESR (equivalent series resistance) MLCC capacitors
should be used at both buck regulator outputs as well as at
each buck regulator input supply (VIN1 and VIN2). Only X5R
or X7R ceramic capacitors should be used because they
retain their capacitance over wider voltage and temperature
ranges than other ceramic types. A 10µF output capacitor is sufﬁcient for most applications. For good transient
response and stability the output capacitor should retain
at least 4µF of capacitance over operating temperature
and bias voltage. Each buck regulator input supply should
be bypassed with a 1µF capacitor. Consult with capacitor
manufacturers for detailed information on their selection
and speciﬁcations of ceramic capacitors. Many manufacturers now offer very thin (<1mm tall) ceramic capacitors
ideal for use in height-restricted designs. Table 6 shows a
list of several ceramic capacitor manufacturers.
Table 6. Recommended Ceramic Capacitor Manufacturers
AVX
www/avxcorp.com
Murata
www.murata.com
Taiyo Yuden
www.t-yuden.com
Table 5 shows several inductors that work well with the
LTC3586-2/LTC3586-3’s buck regulators. These inductors
offer a good compromise in current rating, DCR and physical size. Consult each manufacturer for detailed information
on their entire selection of inductors.
Vishay Siliconix
www.vishay.com
TDK
www.tdk.com
Table 5. Recommended Inductors for Buck Regulators
Inductor selection criteria for the buck-boost are similar
to those given for the buck switching regulator. The buckboost converter is designed to work with inductors in the
range of 1µH to 5µH. For most applications a 2.2µH inductor
will suffice. Choose an inductor with a DC current rating
at least 2 times larger than the maximum load current to
ensure that the inductor does not saturate during normal
operation. If output short circuit is a possible condition,
the inductor should be rated to handle the maximum peak
current specified for the buck-boost converter.
MAX
INDUCTOR L
IDC
TYPE
(µH) (A)
DE2818C
SIZE IN mm
(L × W × H)
MANUFACTURER
DE2812C
4.7
1.25 0.072* 3.0 × 2.8 × 1.8 Toko
www.toko.com
1.15 0.13* 3.0 × 2.8 × 1.2
CDRH3D16
4.7
0.9
0.11
SD3118
4.7
1.3
SD3112
4.7
0.8
0.162 3.1 × 3.1 × 1.8 Cooper
www.cooperet.com
0.246 3.1 × 3.1 × 1.2
LPS3015
4.7
1.1
*Typical DCR
4.7
MAX
DCR
(Ω)
0.2
4 × 4 × 1.8
Sumida
www.sumida.com
3.0 × 3.0 × 1.5 Coilcraft
www.coilcraft.com
BUCK-BOOST REGULATOR APPLICATIONS SECTION
Buck-Boost Regulator Inductor Selection
Table 7 shows several inductors that work well with the
LTC3586-2/LTC3586-3’s buck-boost regulator. These inductors offer a good compromise in current rating, DCR
and physical size. Consult each manufacturer for detailed
information on their entire selection of inductors.
358623f
30
LTC3586-2/LTC3586-3
APPLICATIONS INFORMATION
Table 7. Recommended Inductors for Buck-Boost Regulator
INDUCTOR L
TYPE
(µH)
MAX
IDC
(A)
MAX
DCR
(Ω)
SIZE IN mm
(L × W × H)
MANUFACTURER
LPS4018
3.3
2.2
2.2
2.5
0.08
0.07
3.9 × 3.9 × 1.7 Coilcraft
3.9 × 3.9 × 1.7 www.coilcraft.com
D53LC
2.0
3.25
0.02
5.0 × 5.0 × 3.0 Toko
www.toko.com
7440430022
2.2
2.5
0.028 4.8 × 4.8 × 2.8 Würth-Elektronik
www.we-online.com
CDRH4D22/
HP
2.2
2.4
0.044 4.7 × 4.7 × 2.4 Sumida
www.sumida.com
SD14
2.0
2.56 0.045
5.2 × 5.2 ×
1.45
Cooper
www.cooperet.com
Buck-Boost Regulator Input/Output Capacitor
Selection
Low ESR ceramic capacitors should be used at both the
buck-boost regulator output (VOUT3) as well as the buckboost regulator input supply (VIN3). Again, only X5R or
X7R ceramic capacitors should be used because they
retain their capacitance over wider voltage and temperature
ranges than other ceramic types. A 22µF output capacitor is
sufficient for most applications. The buck-boost regulator
input supply should be bypassed with a 2.2µF capacitor.
Refer to Table 6 for recommended ceramic capacitor
manufacturers.
Buck-Boost Regulator Output Voltage Programming
The buck-boost regulator can be programmed for output
voltages greater than 2.75V and less than 5.5V. The full
scale output voltage is programmed using a resistor divider
from the VOUT3 pin connected to the FB3 pin such that:
 R1 
VOUT3 = VFB3  + 1
 R2 
where VFB3 is 0.8V. See Figure 8 or 9.
Closing the Feedback Loop
The LTC3586-2/LTC3586-3 incorporate voltage mode PWM
control. The control to output gain varies with operation
region (buck, boost, buck-boost), but is usually no greater
than 20. The output filter exhibits a double pole response
given by:
fFILTER _ POLE =
1
Hz
2 • π • L • COUT
where COUT is the output filter capacitor.
The output filter zero is given by:
fFILTER _ ZERO =
1
2 • π • RESR • COUT
Hz
where RESR is the capacitor equivalent series resistance.
A troublesome feature in boost mode is the right-half plane
zero (RHP), and is given by:
fRHPZ =
VIN2
Hz
2 • π •IOUT • L • VOUT
The loop gain is typically rolled off before the RHP zero
frequency.
A simple Type I compensation network (as shown in
Figure 8) can be incorporated to stabilize the loop but
at the cost of reduced bandwidth and slower transient
response. To ensure proper phase margin, the loop must
cross unity-gain decade before the LC double pole.
The unity-gain frequency of the error amplifier with the
Type I compensation is given by:
fUG =
1
Hz
2 • π • R1• CP1
Most applications demand an improved transient response
to allow a smaller output filter capacitor. To achieve a higher
bandwidth, Type III compensation is required. Two zeros
are required to compensate for the double-pole response.
Type III compensation also reduces any VOUT3 overshoot
seen during a start-up condition.
358623f
31
LTC3586-2/LTC3586-3
APPLICATIONS INFORMATION
The compensation network depicted in Figure 9 yields the
transfer function:
R1+ R3
VC3
=
VOUT3 R1• R3 • C1
•

1
1  

 s +
 •  s +
R2 • C2 
(R1+ R3) • C3 
C1+ C2  
1 

s•s+
• s+
 R2 • C1• C2   R3 • C3 
A Type III compensation network attempts to introduce
a phase bump at a higher frequency than the LC double
pole. This allows the system to cross unity gain after the
LC double pole, and achieve a higher bandwidth. While
attempting to crossover after the LC double pole, the
system must still crossover before the boost right-half
plane zero. If unity gain is not reached sufficiently before
the right-half plane zero, then the –180° of phase from
the LC double pole combined with the –90° of phase from
the right-half plane zero will negate the phase bump of
the compensator.
The compensator zeros should be placed either before
or only slightly after the LC double pole such that their
positive phase contributions of the compensation network
offset the –180° that occurs at the filter double pole. If they
are placed at too low of a frequency, however, they will
introduce too much gain to the system and the crossover
frequency will be too high. The two high frequency poles
should be placed such that the system crosses unity gain
during the phase bump introduced by the zeros yet before
the boost right-half plane zero and such that the compensator bandwidth is less than the bandwidth of the error
amp (typically 900kHz). If the gain of the compensation
network is ever greater than the gain of the error amplifier,
then the error amplifier no longer acts as an ideal op amp,
another pole will be introduced where the gain crossover
occurs, and the total compensation gain will not exceed
that of the amplifier.
Recommended Type III Compensation Components for
a 3.3V output:
R1: 324k
RFB: 105k
C1: 10pF
R2: 15k
C2: 330pF
R3: 121k
C3: 33pF
COUT: 22µF
LOUT: 2.2µH
BOOST REGULATOR APPLICATIONS SECTION
Boost Regulator Inductor Selection
The boost converter is designed to work with inductors in
the range of 1µH to 5µH. For most applications a 2.2µH
inductor will suffice. Larger value inductors will allow
greater output current capability by reducing the inductor
ripple current. However, using too large an inductor may
push the right-half-plane zero too far inside and cause loop
instability. Lower value inductors result in higher ripple
current and improved transient response time. Refer to
Table 7 for recommended inductors.
Boost Regulator Input/Output Capacitor Selection
Low ESR (equivalent series resistance) ceramic capacitors
should be used at both the boost regulator output (VOUT4)
as well as the boost regulator input supply (VIN4). Only
X5R or X7R ceramic capacitors should be used because
they retain their capacitance over wider voltage and temperature ranges than other ceramic types. At least 10µF of
output capacitance at the rated output voltage is required to
ensure stability of the boost converter output voltage over
the entire temperature and load range. Refer to Table 6 for
recommended ceramic capacitor manufacturers.
358623f
32
LTC3586-2/LTC3586-3
APPLICATIONS INFORMATION
+
ERROR
AMP
Boost Regulator Output Voltage Programming
VOUT3
0.8V
The boost regulator can be programmed for output voltages up to 5V. The output voltage is programmed using a
resistor divider from the VOUT4 pin connected to the FB4
pin such that:
R1
FB3
–
CP1
VC3
R2
358623 F08
Figure 8. Error Amplifier with Type I Compensation
where VFB4 is 0.8V. See Figure 10.
VOUT3
+
ERROR
AMP
0.8V
R1
FB3
R3
C3
–
VC3
R2
C2
RFB
C1
358623 F09
Figure 9. Error Amplifier with Type III Compensation
L
Typical values for R1 are in the range of 40k to 1M. Too
small a resistor will result in a large quiescent current
in the feedback network and may hurt efficiency at low
current. Too large a resistor coupled with the FB4 pin capacitance will create an additional pole which may result
in loop instability. If large values are chosen for R1 and
R2, a phase-lead capacitor, CPL, across resistor R1 can
improve the transient response. Recommended values
for CPL are in the range of 2pF to 10pF.
Printed Circuit Board Layout Considerations
VIN4
SW4
LTC3586-2/
LTC3586-3
VOUT4
CPL
 R1 
VOUT4 = VFB4  +1
 R2 
R1
COUT
FB4
R2
In order to be able to deliver maximum current under all
conditions, it is critical that the exposed pad on the backside
of the LTC3586-2/LTC3586-3 packages be soldered to the
PC board ground. Failure to make thermal contact between
the exposed pad on the backside of the package and the
copper board will result in higher thermal resistances.
358623 F10
Figure 10. Boost Converter Application Circuit
358623f
33
LTC3586-2/LTC3586-3
APPLICATIONS INFORMATION
Furthermore, due to its high frequency switching circuitry,
it is imperative that the input capacitors, inductors and
output capacitors be as close to the LTC3586-2/LTC3586-3
as possible and that there be an unbroken ground plane
under the LTC3586-2/LTC3586-3 and all of its external
high frequency components. High frequency currents,
such as the VBUS, VIN1, VIN2 , VIN3 , VOUT3 , and VOUT4
currents on the LTC3586-2/LTC3586-3, tend to ﬁnd their
way along the ground plane in a myriad of paths ranging
from directly back to a mirror path beneath the incident
path on the top of the board. If there are slits or cuts
in the ground plane due to other traces on that layer,
the current will be forced to go around the slits. If high
frequency currents are not allowed to ﬂow back through
their natural least-area path, excessive voltage will build
up and radiated emissions will occur. There should be a
group of vias under the grounded backside of the package leading directly down to an internal ground plane. To
minimize parasitic inductance, the ground plane should
be on the second layer of the PC board.
The GATE pin for the external ideal diode controller has
extremely limited drive current. Care must be taken to
minimize leakage to adjacent PC board traces. 100nA of
leakage from this pin will introduce an offset to the 15mV
ideal diode of approximately 10mV. To minimize leakage,
the trace can be guarded on the PC board by surrounding
it with VOUT connected metal, which should generally be
less that one volt higher than GATE.
358623 F11
Figure 11. Higher Frequency Ground Currents Follow Their
Incident Path. Slices in the Ground Plane Cause High Voltage
and Increased Emmisions
358623f
34
LTC3586-2/LTC3586-3
PACKAGE DESCRIPTION
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
UFE Package
38-Lead Plastic QFN (4mm × 6mm)
(Reference LTC DWG # 05-08-1750 Rev B)
0.70 ±0.05
4.50 ± 0.05
3.10 ± 0.05
2.40 REF
2.65 ± 0.05
4.65 ± 0.05
PACKAGE OUTLINE
0.20 ±0.05
0.40 BSC
4.40 REF
5.10 ± 0.05
6.50 ± 0.05
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
4.00 ± 0.10
0.75 ± 0.05
R = 0.10
TYP
PIN 1 NOTCH
R = 0.30 OR
0.35 × 45°
CHAMFER
2.40 REF
37
38
0.40 ± 0.10
PIN 1
TOP MARK
(NOTE 6)
1
2
4.65 ± 0.10
6.00 ± 0.10
4.40 REF
2.65 ± 0.10
(UFE38) QFN 0708 REV B
0.200 REF
0.00 – 0.05
R = 0.115
TYP
0.20 ± 0.05
0.40 BSC
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING IS NOT A JEDEC PACKAGE OUTLINE
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
358623f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
35
LTC3586-2/LTC3586-3
TYPICAL APPLICATION
Watchdog Microcontroller Operation
USB/WALL
4.5V TO 5.5V
35, 36
C1
22µF
100k
5
29
T
4
2k
0.1µF
SW
VBUS
VOUT
GATE
NTC
BAT
PROG
CLPROG
GND
2.94k
CHRG
VOUT3
LTC3586-2
LTC3586-3
3.3V, 20mA
3
LDO3V3
1µF
10k
38
HOUSEKEEPING
MICROCONTROLLER
4
AUDIO/
MOTOR DRIVE
2
1, 2
9
ILIM
MODE
SWAB3
VIN3
SW2
6, 7
88.7k
10pF
10
TO OTHER
LOADS
34
31
MP1
32
+
39
RED
3.3V
1A
10pF
12
15k
121k
VIN2
324k
22µF
11
L2
2.2µH
13
2.2µF
105k
14, 15
25
1.8V
400mA
L3 4.7µH
1.02M
23
26
10µF
1µF
28
MICROPROCESSOR
1.6V
400mA
L4 4.7µH
806k
806k
I/O/MEMORY
10pF
24
FB4
FB1
SYSTEM RAIL/
I/O
33pF
330pF
806k
SW1
16.9k
Li-Ion
30
EN
VOUT4
510Ω
C2
22µF
16, 17
FB3
19
SWCD3
FB2
18, 20, 21, 33
5V
800mA
22µF
FAULT
VC3
L1
3.3µH
37
CORE
10pF
10µF
1µF
27
VIN1
VIN4 22
C1, C2: TDK C2012X5R0J226M
L1: COILCRAFT LPS4018-332LM
L2, L5: TOKO 1098AS-2R2M
L3, L4: TOKO 1098AS-4R7M
MP1: SILICONIX Si2333
L5
2.2µH
SW4
10µF
8
358623 TA02
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC3555
I2C Controlled High Efficiency USB
Maximizes Available Power from USB Port, Bat-Track, “Instant On” Operation, 1.5A Max
Charge Current, 3.3V/25mA Always-On LDO, Three Synchronous Buck Regulators, One 1A
Buck-Boost Regulator, 4mm × 5mm QFN28 Package
Power Manager Plus Triple Step-Down
DC/DC
LTC3556
High Efficiency USB Power Manager Plus Maximizes Available Power from USB Port, Bat-Track, “Instant On” Operation, 1.5A Max
Dual Buck Plus Buck-Boost DC/DC
Charge Current, 3.3V/25mA Always-On LDO, Two 400mA Synchronous Buck Regulators,
One 1A Buck-Boost Regulator, 4mm × 5mm QFN28 Package
LTC3566
Switching USB Power Manager with
Li-Ion/Polymer Charger, 1A Buck-Boost
Converter Plus LDO
LTC3586/
LTC3586-1
Switching USB Power Manager with
Complete Multifunction PMIC: Switching Power Manager, 1A Buck-Boost + 2 Bucks +
Li-Ion/Polymer Charger, 1A Buck-Boost + Boost + LDO, Synchronous Buck/Buck-Boost Converter Efficiency: >95%; Charge Current
Dual Sync Buck Converter + Boost + LDO 1.5A; LTC3586-1 version has 4.1V VFLOAT; 4mm × 6mm QFN-38 Package
Multifunction PMIC: Switchmode Power Manager and 1A Buck-Boost Regulator + LDO,
Charge Current Programmable Up to 1.5A from Wall Adapter Input, Thermal Regulation
Synchronous Buck-Boost Converters Efficiency: >95%, 4mm × 4mm QFN24 Package
358623f
36 Linear Technology Corporation
LT 0312 • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
 LINEAR TECHNOLOGY CORPORATION 2012

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