MICREL MIC2165YMME

MIC2165
Adaptive On-Time DC-DC Controller
Featuring Hyper Light LoadTM
Hyper Speed Control™ Family
General Description
Features
The Micrel MIC2165 is a synchronous adaptive on-time
buck controller targeting high performance, cost sensitive
applications such as set-top boxes, gateways, routers,
computing
peripherals,
and
telecom/networking
equipment.
The MIC2165 operates over a supply range of 4.5V to
28V. It has an internal linear regulator which provides a
regulated 5V supply to power the internal control circuitry.
MIC2165 operates at a constant 600kHz switching
frequency and can be used to drive up to 25A of output
current. The output voltage is adjustable down to 0.8V.
A unique Hyper Speed Control™ architecture enables
ultra-fast transient response while reducing the output
capacitance and also makes High VIN/Low VOUT operation
possible.
A UVLO feature is provided to ensure proper operation
under power-sag conditions to prevent the external power
MOSFET from over heating. Also, a soft start feature is
provided to reduce the inrush current. Short current
sensing on the bottom MOSFET with hiccup mode
operation ensures protection in case of an output short
circuit. Further, the MIC2165 includes an EN pin to shut
down the converter and a PGOOD pin to allow simple
sequencing.
The MIC2165 is available in a 10-pin MSOP ePad
package with a junction operating temperature range from
–40 ºC to +125 ºC. All support documentation can be
found on Micrel’s web site at: www.micrel.com.
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Hyper Speed Control™ architecture enables
- High VIN/VOUT operation (VIN=28V and VOUT=0.8V)
- Smallest output capacitance
Hyper Light Load™ Efficiency
Built-in 5V regulator for single-supply operation
TM
Any Capacitor stable
- Zero ESR to high ESR
Power-Good output
Input voltage range: 4.5V to 28V
5μA typical shutdown current
25A output current drive capability
Output down to +0.8V with ±1% FB Accuracy
600kHz switching frequency
Internal 5ms digital Soft Start
Thermal shutdown and hiccup current limit protection
No external current-sense resistor required
Pre-bias output safe
10-pin MSOP ePad package
–40°C to +125°C junction temperature range
Applications
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Set-top box, gateways, routers and DSL modems
Printers, scanners, graphic and video cards
Servers, PCs and processor core supply
Low-Voltage Distributed Power
Typical Application
12V to 3.3V Efficiency
100
90
EFFICIENCY (%)
80
70
60
50
40
30
20
10
MIC2165 Adjustable Output 600kHz Buck Converter
0
0.01
0.10
1.00
10.00
OUTPUT CURRENT (A)
MLF and MicroLeadFrame are registered trademarks of Amkor Technology, Inc.
Hyper Light Load and Any Capacitor are trademarks of Micrel, Inc.
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
June 2010
M9999-060810-D
Micrel, Inc.
MIC2165
Ordering Information
Part Number
Voltage
Switching Frequency
Junction Temp. Range
Package
Lead Finish
MIC2165YMME
Adj.
600kHz
–40° to +125°C
10-pin ePad MSOP
Pb-Free
Pin Configuration
10-Pin ePad MSOP (MME)
Pin Description
Pin
Number
Pin Name
Pin Function
1
FB
2
PGOOD
3
EN
Enable (input): A logic level control of the output. The EN pin is CMOS-compatible. Logic high = enable,
logic low = shutdown. In the off state, supply current of the device is greatly reduced (typically 5µA). The
EN pin should not be left open. Connect to VIN if sequencing not required.
4
VIN
Supply Voltage: Input voltage for the internal +5V linear regulator. The VIN operating voltage range is
from 4.5V to 28V. A 0.1µF capacitor between VIN and the ground is required.
5
VDD
5V Internal Linear Regulator (Output): VDD is the external MOSFET gate drive supply voltage and an
internal supply bus for the IC. VDD is created by internal LDO from VIN. When VIN <+5.5V, VDD Should
be tied to VIN. A 2.2µF (minimum) ceramic capacitor from VDD to GND is recommended for clean
operation.
6
DL
Low-Side Gate Drive (output): High-current driver output for external low-side MOSFET. The DL driving
voltage swings from ground to VDD.
7
PGND
Power Ground. PGND is the ground path for the MIC2165 buck converter power stage. The PGND pin
connects to the sources of low-side N-Channel MOSFETs, the negative terminals of input capacitors,
and the negative terminals of output capacitors. The loop for the power ground should be as small as
possible and separate from the Signal ground (GND) loop.
8
DH
High-Side Gate Drive (output): High-current driver output for external high-side MOSFET. The DH
driving voltage is floating on the switch node voltage (SW). It swings from ground-to-VDD minus the
diode drop.
9
SW
Switch Node (input): High current output driver return. The SW pin connects directly to the switch node.
Due to the high speed switching on this pin, the SW pin should be routed away from sensitive nodes.
Feedback (input): Input to the transconductance amplifier of the control loop. The FB pin is regulated to
0.8V. A resistor divider connecting the output to FB is used to set the desired output voltage.
Power Good (Output): Open Drain Output. The PGOOD pin is externally tied with a resistor to VDD.
High output when VOUT>90% nominal.
Current Sense input (input): SW pin also senses the current by monitoring the voltage across the lowside MOSFET during OFF-time. The current sensing is necessary for short circuit protection and zero
current cross comparator. In order to sense the current accurately, connect the low-side MOSFET drain
to SW using a Kelvin connection.
10
BST
EP
GND
June 2010
Boost (output): Bootstrapped voltage to the high-side N-channel MOSFET driver. A Schottky diode is
connected between the VDD pin and the BST pin. A boost capacitor of 0.1μF is connected between the
BST pin and the SW pin. Adding a small resistor at BST pin can slow down the turn-on time of high-side
N-Channel MOSFETs.
Thermal Pad and Signal ground. GND is the ground path for VDD and the control circuitry. The loop for
the signal ground should be separate from the power ground (PGND) loop.
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M9999-060810-D
Micrel, Inc.
MIC2165
Absolute Maximum Ratings(1)
Operating Ratings(2)
VIN to GND ...................................................... -0.3V to +29V
VDD, VFB, VPGOOD to GND .................................. -0.3V to +6V
VBST to VSW ....................................................... -0.3V to +6V
VBST to GND ................................................... -0.3V to +35V
VEN to GND.............................................-0.3V to (VIN+0.3V)
VDH to VSW ..........................................-0.3V to (VBST + 0.3V)
VDL to GND ..........................................-0.3V to (VDD + 0.3V)
PGND to GND ............................................... -0.3V to +0.3V
Junction Temperature .............................................. +150°C
Storage Temperature (TS)..........................-65°C to +150°C
Lead Temperature (soldering, 10sec) ........................ 260°C
Supply Voltage (VIN) .......................................... 4.5V to 28V
Enable Input Voltage (VEN)..................................... 0V to VIN
Junction Temperature (TJ) .........................-40°C to +125°C
(3)
Package Thermal Resistance
MSOP-10L ePad (θJA) ......................................77°C/W
MSOP-10L ePad (θJC) ......................................10°C/W
Electrical Characteristics
VIN = VEN= 12V; VBST – VSW = 5V; TJ = 25°C, unless noted. Bold values indicate -40°C ≤ TJ ≤ 125°C.
Parameter
Condition
Min
Typ
Max
Units
28
V
450
750
µA
5
10
µA
Power Input Supply
Input Voltage Range (VIN)
4.5
Quiescent Supply Current
VFB = 1.5V (non-switching)
Shutdown Current
VEN = 0V
VDD Supply
VDD Output Voltage
VIN = 7V to 28V, IDD = 40mA
4.8
5.2
5.4
V
VDD UVLO Threshold
VDD rising
3.7
4.2
4.5
V
VDD UVLO Hysteresis
Dropout Voltage (VIN - VDD)
400
IDD = 25mA
380
mV
600
mV
DC-DC Controller
Output-Voltage Adjust Range
(VOUT)
V
0.8
Reference
Feedback Regulation Voltage
Load Regulation
TJ = 25°C
0.792
0.808
V
0°C ≤ TJ ≤ 85°C
0.788
0.812
V
-40°C ≤ TJ ≤ 125°C
0.784
0.816
V
IOUT = 2A to 10A (Continuous Mode)
0.8
0.25
%
0.25
%
Depends on external components
Line Regulation
VIN = 4.5V to 28V
Depends on external components
FB Bias Current
Enable Control
VFB = 0.8V
50
nA
(5)
Enable Logic Level High
V
1.6
Enable Logic Level Low
0.6
Enable Hysteresis
Enable Bias Current
June 2010
500
100
VEN = 12V
6
3
V
mV
30
µA
M9999-060810-D
Micrel, Inc.
MIC2165
On Timer
Switching Frequency
Minimum Off-Time
450
600
750
kHz
200
300
400
ns
Maximum Duty Cycle
Results from Switching Frequency and Minimum
Off-Time
82
%
Minimum Duty Cycle
VFB = 1.0V
0
%
Bottom FET Active
Zero Crossing Comparator Offset
-16
0.5
16
mV
Short Current Protection
Current Limit Threshold
VFB = 0.79V
98
133
182
mV
Short Circuit Current
VFB = 0V
24
48
72
mV
0.1
V
FET Drives
DH, DL Output Low Voltage
ISINK = 10mA
DH, DL Output High Voltage
ISOURCE = 10mA
DH On-Resistance
Pull Up, ISOURCE = 20mA
Pull Down, ISINK = 20mA
2
3
Ω
1.5
3
Ω
Pull Up, ISOURCE = 20mA
2
3
Ω
Pull Down, ISINK = 20mA
1
2
Ω
30
µA
DL On-Resistance
SW, BST Leakage Current
V
VDD-0.1V
or
VBST-0.1V
VSW = VBST = 0
Power Good
Power Good Threshold Voltage
Sweep VFB from Low to High
Power Good Hysteresis
Power Good Delay Time
Sweep VFB from High to Low
Sweep VFB from Low to High
6.0
100
Power Good Low Voltage
VFB<0.9 × VNOM, IPGOOD = 1mA
70
TJ Rising
160
°C
15
°C
85
90
95
%VOUT
%VOUT
µs
200
mV
Thermal Protection
Over-temperature Shutdown
Over-temperature Shutdown
Hysteresis
Notes:
1. Exceeding the absolute maximum rating may damage the device.
2. The device is not guaranteed to function outside its operating rating.
3. The maximum allowable power dissipation of any TA (ambient temperature) is PD(max) = (TJ(max) – TA) / θJA. Exceeding the maximum allowable power
dissipation will result in excessive die temperature, and the regulator will go into thermal shutdown.
4. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5k in series with 100pF.
5. Enable pin should not be left open.
June 2010
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Micrel, Inc.
MIC2165
Typical Characteristics
12V to 3.3V Efficiency
12V to 1.2V Efficiency
Output Voltage vs. Load
90
90
1.28
80
80
1.26
70
70
60
50
40
30
OUTPUT VOLTAGE (V)
1.30
EFFICIENCY (%)
100
EFFICIENCY (%)
100
60
50
40
30
1.22
1.20
1.18
1.16
20
20
1.14
10
10
1.12
0
0.01
0.10
1.00
10.00
0.10
1.00
Feedback Voltage
vs. Temperature
0.812
1.24
IOUT = 100mA
1.20
IOUT = 10A
1.16
1.14
0.808
0.804
0.800
0.796
0.792
1.12
1.10
12
16
20
24
28
-40
0
40
80
Switching Frequency
vs. Temperature
Switching Frequency
vs. Input Voltage
750
SWITCHING FREQUENCY (kHz)
660
630
600
570
540
510
VIN = 12V
VOUT = 2.5V
480
690
660
630
600
570
540
510
IOUT = 5A
0
40
80
TEMPERATURE (°C)
June 2010
120
600
570
540
510
VIN = 12V
VOUT = 2.5V
480
0
2
8
12
16
20
INPUT VOLTAGE (V)
5
6
8
10
Current Limit Threshold vs.
Feedback Voltage Percentage
140
120
100
80
60
40
20
0
4
4
0
450
-40
630
160
720
480
450
660
OUTPUT CURRENT (A)
TEMPERATURE (ºC)
690
10
690
120
INPUT VOLTAGE (V)
720
8
720
CURRENT LIMIT THRESHOLD
(mV)
750
8
6
450
0.788
4
4
Switching Frequency vs. Load
(Continuous Mode)
750
SWITCHING FREQUENCY (kHz)
FEEDBACK VOLTAGE (V)
OUTPUT VOLTAGE (V)
1.26
1.18
2
OUTPUT CURRENT (A)
1.28
1.22
0
10.00
OUTPUT CURRENT (A)
Output Voltage
vs. Input Voltage
1.30
VIN = 12V
VOUT = 1.2V
1.10
0
0.01
OUTPUT CURRENT (A)
SWITCHING FREQUENCY (kHz)
1.24
24
28
20
40
60
80
100
FEEDBACK VOLTAGE PERCENTAGE (%)
M9999-060810-D
Micrel, Inc.
MIC2165
Typical Characteristics (Continued)
Current Limit Threshold
vs. Temperature
140
5.5
550
5.4
VFB = 0.79V
500
100
80
VFB = 0V
60
40
VDD DROPOUT (mV)
5.3
120
VDD VOLTAGE (V)
CURRENT LIMIT THRESHOLD
(mV)
160
5.2
5.1
5.0
4.9
4.8
4.7
20
4.6
0
4.5
-40
0
40
80
0
40
80
120
-40
Enable Threshold
vs. Input Voltage
1.20
1.15
1.10
1.05
1.00
1.2
0.9
0.6
0.3
0.95
0.0
0.90
8
12
16
20
24
0
500
96
95
94
93
92
91
90
40
80
120
-40
0
TEMPERATURE (°C)
INPUT VOLTAGE (V)
Quiescent Current
vs. Input Voltage
420
400
380
360
340
320
120
VDD Dropout vs. VDD Load
600
80
VDD DROPOUT (mV)
SHUTDOWN CURRENT (µA)
440
80
700
90
460
40
TEMPERATURE (°C)
Shutdown Current
vs. Input Voltage
100
480
120
89
-40
28
80
Power Good Threshold
vs. Temperature
97
POWER GOOD THRESHOLD (%)
1.25
40
TEMPERATURE (°C)
Enable Threshold
vs. Temperature
1.5
1.30
4
0
TEMPERATURE (°C)
ENABLE THRESHOLD (V)
ENABLE THRESHOLD (V)
350
IDD = 25mA
1.35
QUIESCENT CURRENT (µA)
400
250
-40
120
450
300
TEMPERATURE (°C)
1.40
VDD Dropout vs.
Temperature
VDD vs. Temperature
70
60
50
40
30
500
400
300
200
20
100
10
300
0
0
4
8
12
16
20
INPUT VOLTAGE (V)
June 2010
24
28
4
8
12
16
20
INPUT VOLTAGE (V)
6
24
28
0
5
10
15
20
25
30
35
40
VDD OUTPUT CURRENT (mA)
M9999-060810-D
Micrel, Inc.
MIC2165
Functional Characteristics
June 2010
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Micrel, Inc.
MIC2165
Functional Characteristics (Continued)
June 2010
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Micrel, Inc.
MIC2165
Functional Characteristics (Continued)
June 2010
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Micrel, Inc.
MIC2165
Functional Diagram
Figure 1. MIC2165 Block Diagram
June 2010
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Micrel, Inc.
MIC2165
Functional Description
DMAX =
The MIC2165 is an adaptive ON-time buck controller
built for low cost and high performance. Featuring an
internal 5V linear regulator and PGOOD output, it is
designed for a wide input voltage range from 4.5V to
28V, high output power buck converters. An estimated
ON-time method is used in the MIC2165 to obtain a
constant switching frequency and to simplify the control
compensation. Over-current protection is implemented
without the use of an external sense resistor. It includes
an internal soft-start function which reduces the power
supply input surge current at start-up by controlling the
output voltage rise time.
Continuous Mode
In the continuous mode, the output voltage variation will
be sensed by the MIC2165 feedback pin FB via the
voltage divider. The FB voltage VFB is compared to a
0.8V reference voltage VREF at the error comparator
through a low gain transconductance (gm) amplifier at
switching frequency. This gm amplifier improves the
MIC2165 converter output voltage regulation. If the FB
voltage VFB decreases and the output of the gm amplifier
is below 0.8V, The error comparator will trigger the
control logic and generate an ON-time period, in which
DH pin is logic high and DL pin is logic low. The ON-time
period length is predetermined by the “Fixed TON
Estimator” circuitry:
VOUT
VIN × f sw
= 1−
300ns
TS
(1)
where VOUT is the output voltage, VIN is the power stage
input voltage, and fSW is the switching frequency
(600kHz for MIC2165).
After an ON-time period, the MIC2165 goes into the
OFF-time period, in which DH pin is logic low and DL pin
is logic high. The OFF-time period length depends on
VFB in most cases. When VFB decreases and the output
of the gm amplifier is below 0.8V, the ON-time period is
triggered and the OFF-time period ends. If the OFF-time
period determined by VFB is less than the minimum OFF
time TOFF(min), which is about 300ns typical, then the
MIC2165 control logic will apply the TOFF(min) instead.
TOFF(min) is required to maintain enough energy in the
boost capacitor (CBST) to drive the high-side MOSFET.
The maximum duty cycle is obtained from the 300ns
TOFF(min):
June 2010
TS
where TS = 1/fSW. It is not recommended to use MIC2165
with an OFF-time close to TOFF(min) during steady state
operation.
The estimated ON-time method results in a constant
switching frequency in the MIC2165. The actual ON-time
varies slightly with the different rising and falling times of
the external MOSFETs. Therefore, the type of the
external MOSFETs and the output load current will
modify the actual ON-time and the switching frequency.
Also, the minimum TON results in a lower switching
frequency in high VIN and low VOUT applications, such as
24V to 1.0V. The minimum TON measured on the
MIC2165 evaluation board is about 100ns. During the
load transient, the switching frequency is changed due to
the varying OFF-time.
To illustrate the control loop, the steady-state scenario
and the load transient scenario are analyzed. For easy
analysis, the gain of the gm amplifier is assumed to be 1.
With this assumption, the inverting input of the error
comparator is the same as VFB. Figure 2 shows the
MIC2165 control loop timing during steady-state
operation in continuous mode. During steady-state, the
gm amplifier senses VFB ripple, which is proportional to
the output voltage (VOUT) ripple and the inductor current
ripple, to trigger the ON-time period. The ON-time is
predetermined by the estimation. The ending of OFFtime is controlled by VFB. At the valley of VFB ripple,
which occurs when VFB falls below VREF, the OFF period
ends and the next ON-time period is triggered through
the control logic circuitry.
Theory of Operation
The MIC2165 is an adaptive on-time buck controller.
Figure 1 illustrates the block diagram for the control loop.
MIC2165 is able to operate in two modes: continuous
mode and discontinuous mode. The operation mode of
MIC2165 is determined by the output of Zero Cross
Comparator (ZC), as shown in Figure 1.
TON(estimated) =
TS − TOFF(min)
Figure 2. MIC2165 Control Loop Timing
(Continuous Mode)
Figure 3 shows the load transient operation of the
MIC2165 converter. The output voltage drops due to the
sudden load increase, which causes VFB to be less than
VREF. This will cause the error comparator to trigger an
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Micrel, Inc.
MIC2165
ON-time period. At the end of the ON-time period, a
minimum OFF-time TOFF(min) is generated to charge CBST
since VFB is still below VREF. Then, the next ON-time
period is triggered due to the low VFB. Therefore, the
switching frequency changes during the load transient.
With the varying duty-cycle and switching frequency, the
output recovery time is fast and the output voltage
deviation is small in MIC2165 converter.
The MIC2165 has a zero crossing comparator that
monitors the inductor current by sensing the voltage
drop across the low-side MOSFET during its ON-time. If
the VFB > 0.8V and the inductor current goes slightly
negative, then the MIC2165 automatically powers down
most of the IC circuitry and goes into a low-power mode.
Once the MIC2165 goes into discontinuous mode, both
DL and DH are low, which turns off the high-side and
low-side MOSFETs. The load current is supplied by the
output capacitors and VOUT drops. If the drop of VOUT
causes VFB to go below VREF, then all the circuits will
wake up into normal continuous mode. First, the bias
currents of most circuits reduced during the
discontinuous mode are restored, then a TON pulse is
triggered before the drivers are turned on to avoid any
possible glitches. Finally, the high-side driver is turned
on. Figure 4 shows the control loop timing in
discontinuous mode.
Figure 3. MIC2165 Load-Transient Response
Unlike in current-mode control, the MIC2165 uses the
output voltage ripple, which is proportional to the
inductor current ripple if the ESR of the output capacitor
is large enough, to trigger an ON-time period. The
predetermined ON-time makes MIC2165 control loop
have the advantage of constant on-time mode control
and eliminates the need for slope compensation.
The MIC2165 has its own stability concern: VFB ripple
should be in phase with the inductor current ripple and
large enough to be sensed by the gM amplifier and the
error comparator. The recommended VFB ripple is
20mV~100mV. If a low ESR output capacitor is selected,
the VFB ripple may be too small to be sensed by the gm
amplifier and the error comparator. Also, the VOUT ripple
and the VFB ripple are not in phase with the inductor
current ripple if the ESR of the output capacitor is very
low. Therefore, ripple injection is required for a low ESR
output capacitor. Please refer to “Ripple Injection”
subsection in “Application Information” for more details.
Figure 4. MIC2165 Control Loop Timing
(Discontinuous Mode)
An external Schottky diode D2 is recommended in
parallel with the low-side MOSFET for high efficiency
performance as shown in the typical application
schematic. Please refer to “External Schottky Diode”
subsection in “Application Information” for more details.
During discontinuous mode, the zero crossing
comparator and the current limit comparator are turned
off. The bias current of most circuits are reduced. As a
result, the total power supply current during
discontinuous mode is only about 450μA, allowing the
MIC2165 to achieve high efficiency in light load
applications.
Discontinuous Mode
In continuous mode, the inductor current is always
greater than zero; however, at light loads the MIC2165 is
able to force the inductor current to operate in
discontinuous mode. Discontinuous mode is where the
inductor current falls to zero, as indicated by trace (IL)
shown in Figure 4. During this period, the efficiency is
optimized by shutting down all the non-essential circuits
and minimizing the supply current. The MIC2165 wakes
up and turns on the high-side MOSFET when the
feedback voltage VFB drops below 0.8V.
June 2010
Soft-Start
Soft-start reduces the power supply input surge current
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MIC2165
at startup by controlling the output voltage rise time. The
input surge appears while the output capacitor is
charged up. A slower output rise time will draw a lower
input surge current.
The MIC2165 implements an internal digital soft-start by
making the 0.8V reference voltage VREF ramp from 0 to
100% in about 5ms. Therefore, the output voltage is
controlled to increase slowly by a stair-case VREF ramp.
Once the soft-start cycle ends, the related circuitry is
disabled to reduce current consumption. During soft-start,
the discontinuous mode is disabled in MIC2165.
Current Limit
The MIC2165 uses the RDS(ON) of the low-side power
MOSFET to sense over-current conditions. This method
will avoid adding cost, board space and power losses
taken by discrete current sense resistors. The low-side
MOSFET is used because it displays much lower
parasitic oscillations during switching than the high-side
MOSFET.
In each switching cycle of the MIC2165 converter, the
inductor current is sensed by monitoring the low-side
MOSFET in the OFF period. The sensed voltage is
compared with a current-limit threshold voltage VCL after
a blanking time of 150ns. If the sensed voltage is over
VCL, which is 133mV typical at 0.8V VFB, then the
MIC2165 turns off the high-side and low-side MOSFETs
and a soft-start sequence is triggered. This mode of
operation is called “hiccup mode” and its purpose is to
protect the downstream load in case of a hard short. The
current limit threshold VCL has a foldback characteristic
related to the FB voltage. Please refer to the “Typical
Characteristics” for the curve of current limit threshold vs.
FB voltage percentage. The circuit in Figure 5 illustrates
the MIC2165 current limiting circuit.
operation, calculating the current limit ICL should take
into account that one is sensing the peak inductor
current and that there is a blanking delay of
approximately 150ns.
ICL =
VOUT × (1 − D)
f SW ×L
(3)
where:
VOUT = The output voltage
tDLY = Current limit blanking time, 150ns typical
ΔIL(PP) = Inductor current ripple peak-to-peak value
D = Duty Cycle
fSW = Switching frequency
The MOSFET RDS(ON) varies between 30% to 40% with
temperature; therefore, it is recommended to add 50%
margin to ICL in the above equation to avoid false current
limiting due to increased MOSFET junction temperature
rise. It is also recommended to connect SW pin directly
to the drain of the low-side MOSFET to accurately sense
the MOSFETs RDS(ON).
MOSFET Gate Drive
The MIC2165 high-side drive circuit is designed to
switch an N-Channel MOSFET. The typical application
schematic shows a bootstrap circuit, consisting of D1 (a
Schottky diode is recommended) and CBST. This circuit
supplies energy to the high-side drive circuit. Capacitor
CBST is charged while the low-side MOSFET is on and
the voltage on the SW pin is approximately 0V. When
the high-side MOSFET driver is turned on, energy from
CBST is used to turn the MOSFET on. As the high-side
MOSFET turns on, the voltage on the SW pin increases
to approximately VIN. Diode D1 is reversed biased and
CBST floats high while continuing to keep the high-side
MOSFET on. The bias current of the high-side driver is
less than 10mA so a 0.1μF to 1μF is sufficient to hold
the gate voltage with minimal droop for the power stroke
(high-side switching) cycle, i.e., ΔBST = 10mA x
1.67μs/0.1μF = 167mV. When the low-side MOSFET is
turned back on, CBST is recharged through D1. A small
resistor RG at BST pin can be used to slow down the
turn-on time of the high-side N-channel MOSFET.
The drive voltage is derived from the internal linear
regulator VDD. The nominal low-side gate drive voltage is
VDD and the nominal high-side gate drive voltage is
approximately VDD – VDIODE, where VDIODE is the voltage
drop across D1. A dead time of approximate 30ns delay
between the high-side and low-side driver transitions is
used to prevent current from simultaneously flowing
unimpeded through both MOSFETs.
Using the typical VCL value of 133mV, the current limit
value is roughly estimated as:
133mV
RDS(ON)
For designs where the current ripple is significant
compared to the load current IOUT, or for low duty-cycle
June 2010
(2)
ΔIL(PP) =
Figure 5. MIC2165 Current Limiting Circuit
ICL ≈
ΔIL(PP)
×t
V
133mV
+ OUT DLY −
R DS(ON)
L
2
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Micrel, Inc.
MIC2165
For the Low-Side (LS) MOSFET:
Application Information
IG[LS] (avg) = C ISS × VGS × f SW
MOSFET Selection
The MIC2165 controller works from an input voltage of
4.5V to 28V and has an internal 5V VDD to provide power
to turn the external N-Channel power MOSFETs for the
high-side and low-side switches. For applications where
VIN < 5.5V, it is recommended to connect VDD-to-VIN to
bypass the internal linear regulator. The external power
MOSFETs should be logic-level MOSFETs, whose
operation is specified at VGS = 4.5V.
There are different criteria for choosing the high-side and
low-side MOSFETs. These differences are more
significant at lower duty cycles such as 24V to 1.2V
conversion. In such an application, the high-side
MOSFET is required to switch as quickly as possible to
minimize transition losses, whereas the low-side
MOSFET can switch slower, but must handle larger
RMS currents. When the duty cycle approaches 50%,
the current carrying capability of the high-side MOSFET
starts to become critical.
It is important to note that the on-resistance of a
MOSFET increases with increasing temperature. For a
MOSFET with a 0.4%/°C thermal coefficient a 75°C rise
in junction temperature will increase the channel
resistance of the MOSFET by 30% resistance specified
at 25°C. This change in resistance must be accounted
for when calculating MOSFET power dissipation and the
value of current limit. Total gate charge is the charge
required to turn the MOSFET on and off under specified
operating conditions (VDS and VGS). The gate charge is
supplied by the MIC2165 gate-drive circuit. At 600kHz
switching frequency, the gate charge can be a significant
source of power dissipation in the MIC2165. At light
output load, this power dissipation is noticeable as a
reduction in efficiency. The average current required to
drive the high-side MOSFET is:
IG[HS] (avg) = Q G × f SW
(5)
Since the current from the gate drive comes from the
VDD, which is the output of the internal linear regulator
power by VIN, the power dissipated in the MIC2165 due
to gate drive is:
PGATEDRIVE = VIN × (IG[high-side] (avg) + IG[low -side] (avg)) (6)
A convenient figure of merit for switching MOSFETs is
the on resistance times the total gate charge (RDS(ON) ×
QG). Lower numbers translate into higher efficiency. Low
gate-charge logic-level MOSFETs are a good choice for
use with the MIC2165. Also, the RDS(ON) of the low-side
MOSFET will determine the current limit value. Please
refer to “Current Limit” subsection is “Functional
Description” for more details.
Parameters that are important to MOSFET switch
selection are:
•
Voltage rating
•
On-resistance
• Total gate charge
The voltage ratings for the high-side and low-side
MOSFETs are essentially equal to the power stage input
voltage VIN. A safety factor of 20% should be added to
the VDS(max) of the MOSFETs to account for voltage
spikes due to circuit parasitic elements.
The power dissipated in the MOSFETs is the sum of the
conduction losses during the on-time (PCONDUCTION) and
the switching losses during the period of time when the
MOSFETs turn on and off (PAC).
PSW = PCONDUCTION + PAC
(7)
PCONDUCTION = ISW(RMS) 2 × R DS(ON)
PAC = PAC(off ) + PAC(on)
(4)
(8)
(9)
where:
RDS(ON) = on-resistance of the MOSFET switch
D = Duty Cycle = VOUT / VIN
Making the assumption that the turn-on and turn-off
transition times are equal; the transition times can be
approximated by:
where:
IG[HS](avg) = Average High-Side (HS) MOSFET gate
current
QG = Total gate charge for the high-side MOSFET taken
from the manufacturer’s data sheet for VGS = VDD.
fSW = Switching Frequency
The low-side MOSFET is turned on and off at VDS = 0V
because an internal body diode or external freewheeling
diode is conducting during this time. The switching loss
for the low-side MOSFET is usually negligible. Also, the
gate-drive current for the low-side MOSFET is more
accurately calculated using CISS at VDS = 0 instead of
gate charge.
tT =
C ISS × VDD + C OSS × VIN
IG
(10)
where:
CISS and COSS are measured at VDS = 0
IG = gate-drive current
The total high-side MOSFET switching loss is:
PAC = (VIN + VD ) × IPK × t T × f SW
(11)
where:
June 2010
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M9999-060810-D
Micrel, Inc.
MIC2165
Lower cost iron powder cores may be used but the
increase in core loss will reduce the efficiency of the
power supply. This is especially noticeable at low output
power. The winding resistance decreases efficiency at
the higher output current levels. The winding resistance
must be minimized although this usually comes at the
expense of a larger inductor. The power dissipated in the
inductor is equal to the sum of the core and copper
losses. At higher output loads, the core losses are
usually insignificant and can be ignored. At lower output
currents, the core losses can be a significant contributor.
Core loss information is usually available from the
magnetics vendor. Copper loss in the inductor is
calculated by the equation below:
2
PINDUCTORCu=IL(RMS) × RWINDING
(16)
The resistance of the copper wire, RWINDING, increases
with the temperature. The value of the winding
resistance used should be at the operating temperature.
RWINDING = RWINDING(20°C) × (1+ 0.0042 × (TH – T20°C)) (17)
where:
TH = temperature of wire under full load
T20°C = ambient temperature
RWINDING(20°C) = room temperature winding resistance
(usually specified by the manufacturer)
tT = Switching transition time
VD = Diode drop
fSW = Switching Frequency
The high-side MOSFET switching losses increase with
the switching frequency and the input voltage VIN. The
low-side MOSFET switching losses are negligible and
can be ignored for these calculations.
Inductor Selection
Values for inductance, peak, and RMS currents are
required to select the output inductor. The input and
output voltages and the inductance value determine the
peak-to-peak inductor ripple current. Generally, higher
inductance values are used with higher input voltages.
Larger peak-to-peak ripple currents will increase the
power dissipation in the inductor and MOSFETs. Larger
output ripple currents will also require more output
capacitance to smooth out the larger ripple current.
Smaller peak-to-peak ripple currents require a larger
inductance value and therefore a larger and more
expensive inductor. A good compromise between size,
loss and cost is to set the inductor ripple current to be
equal to 20% of the maximum output current. The
inductance value is calculated by the equation below.
L=
(
VOUT × VΙΝ(max) − VOUT
)
VΙΝ(max) × f SW × 20% × IOUT(max)
(12)
Output Capacitor Selection
The type of the output capacitor is usually determined by
its ESR (equivalent series resistance). Voltage and RMS
current capability are two other important factors for
selecting the output capacitor. Recommended capacitors
are tantalum, low-ESR aluminum electrolytic, OS-CON,
POSCAPS, and ceramic. The output capacitor’s ESR is
usually the main cause of the output ripple. The output
capacitor ESR also affects the control loop from a
stability point of view. The maximum value of ESR is
calculated:
where:
fSW = switching frequency
20% = ratio of AC ripple current to DC output current
VIN(max) = maximum power stage input voltage
The peak-to-peak inductor current ripple is:
ΔIL(PP ) =
VOUT × ( VIN(max) − VOUT )
VIN(max) × f SW × L
(13)
The peak inductor current is equal to the average output
current plus one half of the peak-to-peak inductor current
ripple.
IL(PK) = IOUT(max) + 0.5 × ΔIL(PP)
ESR COUT ≤
(14)
The RMS inductor current is used to calculate the I R
losses in the inductor.
ΔIL(PP)2
12
(15)
2
(
ΔIL(PP)
⎞
⎛
⎟ + ΔIL(PP) × ESR C
ΔVOUT(PP) = ⎜⎜
⎟
OUT
⎝ COUT × fSW × 8 ⎠
Maximizing efficiency requires the proper selection of
core material and minimizing the winding resistance. The
high frequency operation of the MIC2165 requires the
use of ferrite materials for all but the most cost sensitive
applications.
June 2010
(18)
ΔIL(PP)
where:
ΔVOUT(PP) = peak-to-peak output voltage ripple
ΔIL(PP) = peak-to-peak inductor current ripple
The total output ripple is a combination of the ESR and
output capacitance. The total ripple is calculated below:
2
IL(RMS) = IOUT(max)2 +
ΔVOUT(pp)
)2
(19)
where:
C = output capacitance value
fSW = switching frequency
OUT
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M9999-060810-D
Micrel, Inc.
MIC2165
As described in the “Theory of Operation” subsection in
“Functional Description”, MIC2165 requires at least
20mV peak-to-peak ripple at the FB pin to make the gm
amplifier and the error comparator to behavior properly.
Also, the output voltage ripple should be in phase with
the inductor current. Therefore, the output voltage ripple
caused by the output capacitor COUT should be much
smaller than the ripple caused by the output capacitor
ESR. If low ESR capacitors, such as ceramic capacitors,
are selected as the output capacitors, then a ripple
injection method should be applied to provide the
enough FB voltage ripples. Please refer to the “Ripple
Injection” subsection for more details.
The voltage rating of the capacitor should be twice the
output voltage for a tantalum and 20% greater for
aluminum electrolytic or OS-CON. The output capacitor
RMS current is calculated below:
ICOUT (RMS) =
ΔIL(PP)
12
Figure 6. Voltage-Divider Configuration
The output voltage is determined by the equation:
R1
)
(25)
R2
where VREF = 0.8V. If R1 is too large, it may allow noise
to be introduced into the voltage feedback loop. If R1 is
too small in value, it will decrease the efficiency of the
power supply, especially at light loads. The total voltage
divider resistance R1+R2 is recommended to be 7.5kΩ.
Once R1 is selected, R2 can be calculated using:
VOUT = VREF × (1 +
(20)
The power dissipated in the output capacitor is:
PDISS(COUT ) = ICOUT (RMS) 2 × ESR COUT
(21)
R2 =
Input Capacitor Selection
The input capacitor for the power stage input VIN should
be selected for ripple current rating and voltage rating.
Tantalum input capacitors may fail when subjected to
high inrush currents, caused by turning on a “hotplugging”. A tantalum input capacitor’s voltage rating
should be at least two times the maximum input voltage
to maximize reliability. Aluminum electrolytic, OS-CON,
and multilayer polymer film capacitors can handle the
higher inrush currents without voltage de-rating. The
input voltage ripple will primarily depend upon the input
capacitor’s ESR. The peak input current is equal to the
peak inductor current, so:
ΔVIN = IL(PK ) × ESR CIN
(22)
ID(avg)CM = IOUT × 2 × 30ns × f SW
ID(avg)DM ≈ (1 − D) × (
(23)
ΔIL(PP )
VZC
−
) (28)
Rds(on)
2
where VZC is the zero cross comparator offset.
The reverse voltage requirement of the diode is:
VDIODE(rrm) = VIN
(24)
The power dissipated by the Schottky diode is:
Voltage Setting Components
The MIC2165 requires two resistors to set the output
voltage, as shown in Figure 6.
June 2010
(27)
In the discontinuous mode, the average current through
the diode is large.
The power dissipated in the input capacitor is:
PDISS(CIN ) = ICIN (RMS) 2 × ESR CIN
(26)
External Schottky Diode
An external freewheeling diode, which is recommended
to improve the efficiency in discontinuous mode, can be
used to keep the inductor current flow continuous while
both MOSFETs are turned off.
In continuous mode, the diode conducts current during
the dead-time. The dead-time prevents current from
flowing unimpeded through both MOSFETs and is
typically 30ns. The diode conducts twice during each
switching cycle. Although the average current through
this diode is small, the diode must be able to handle the
peak current.
The input capacitor must be rated for the input current
ripple. The RMS value of input capacitor current is
determined at the maximum output current. Assuming
the peak-to-peak inductor current ripple is low:
ICIN (RMS) ≈ IOUT(max) × D × (1 − D)
VREF × R1
VOUT − VREF
PDIODE = ID(avg) × VF
(29)
where, VF = forward voltage at the peak diode current.
An external Schottky diode is recommended, even
though the low-side MOSFET contains a parasitic body
diode since the Schottky diode has much less forward
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Micrel, Inc.
MIC2165
parasitic capacitance of the MOSFET COSS. A capacitor
that is too small will have high impedance and prevent
the resistor from damping the ringing. A capacitor that is
too large causes unnecessary power dissipation in the
resistor, which lowers efficiency.
voltage than the body diode. The external diode will
improve efficiency and reduce the high frequency noise.
If the MOSFET body diode is used, it must be rated to
handle the peak and average current. The body diode
has a relatively slow reverse recovery time and a
relatively high forward voltage drop. The power lost in
the diode is proportional to the forward voltage drop of
the diode. As the high-side MOSFET starts to turn on,
the body diode becomes a short circuit for the reverse
recovery period, dissipating additional power. The diode
recovery and the circuit inductance will cause ringing
during the high-side MOSFET turn-on.
An external Schottky diode conducts at a lower forward
voltage preventing the body diode in the MOSFET from
turning on. The lower forward voltage drop dissipates
less power than the body diode. The lack of a reverse
recovery mechanism in a Schottky diode causes less
ringing and less power loss.
LSTRAY1
LSTRAY2
RDS
LSTRAY3
RS
COSS2
CS
LSTRAY4
Snubber Design
A snubber is used to damp out high frequency ringing
caused by parasitic inductance and capacitance in the
buck converter circuit. Figure 7 shows a simplified
schematic of the buck converter. Stray capacitance
consists mostly of the two MOSFETs’ output
capacitance (COSS). The stray inductance consists
mostly package inductance and trace inductance. The
arrows show the resonant current path when the high
side MOSFET turns on. This ringing causes stress on
the semiconductors in the circuit as well as increased
EMI.
Figure 8. Snubber Circuit
The snubber components should be placed as close as
possible to the low-side MOSFET and/or external
schottky diode since it contributes to most of the stray
capacitance. Placing the snubber too far from the
MOSFET or using trace that is too long or thin will add
inductance to the snubber and diminishes its
effectiveness.
A proper snubber design requires that the parasitic
inductance and capacitance be known. A method of
determining these values and calculating the damping
resistor value is outlined below.
1. Measure the ringing frequency at the switch node
which is determined by parasitic LP and CP. Define this
frequency as f1.
2. Add a capacitor CS (such as 2 times as big as the
COSS of the FET) from the switch node to ground and
measure the new ringing frequency. Define this new
(lower) frequency as f2. LP and CP can now be solved
using the values of f1, f2 and CS.
3. Add a resistor RS in series with CS to generate critical
damping.
Step 1: First measure the ringing frequency on the
switch node voltage when the high-side MOSFET turns
on. This ringing is characterized by the equation:
COSS1
+
LSTRAY1
LSTRAY2
L
Q1
CIN
VDC
LSTRAY3
Sync_buck
Controller
Q2
COSS2
COUT
LSTRAY4
–
Figure 7. Output Parasitics
One method of reducing the ringing is to use a resistor
and capacitor to lower the Q of the resonant circuit, as
shown in Figure 8. Capacitor CS is used to block DC and
minimize the power dissipation in the resistor. This
capacitor value should be between 2 and 10 times the
June 2010
f1 =
1
(30)
2π L P × C P
where CP and LP are the parasitic capacitance and
inductance.
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Micrel, Inc.
MIC2165
Step 2: Add a capacitor, CS, in parallel with the
synchronous MOSFET, Q2. The capacitor value should
be approximately 2 times the COSS of Q2. Measure the
frequency of the switch node ringing, f2:
1
f2 =
2π Lp × (Cs + Cp)
1) Enough ripple at VOUT due to the large ESR of the
output capacitors.
As shown in Figure 9a, the converter is stable without
any ripple injection. The VFB ripple is:
(31)
ΔVFB(pp) =
(37)
where ΔVOUT = ESR COUT ⋅ ΔIL (PP) , ΔIL(PP) is the peak-
Define f’ as:
f' =
f1
f2
to-peak value of the inductor current ripple.
2) Inadequate ripple at VOUT due to the small ESR of the
output capacitors.
The output voltage ripple is fed into the FB pin through a
feedforward capacitor Cff in this situation, as shown in
Figure 9b. The typical Cff value is between 1nF and
100nF. With the feedforward capacitor, VFB ripple is very
close to the output voltage ripple:
Combining the equations for f1, f2 and f’ to derive CP, the
parasitic capacitance:
CP =
CS
' 2
(32)
(f ) − 1
LP is solved by re-arranging the equation for f1:
LP =
1
(2π)2 × CP × ( f1 ) 2
ΔVFB(pp) ≈ ΔVOUT
(33)
Q = RS ×
CP
=1
LP
(34)
Solving for RS
RS =
LP
Cp
ΔVFB(PP) = VIN × K div × D × (1 - D) ×
(35)
K div =
Figure 8 shows the snubber in the circuit and the
damped switch node waveform. The snubber capacitor,
CS, is charged and discharged each switching cycle. The
energy stored in CS is dissipated by the snubber resistor,
RS, two times per switching period. This power is
calculated in the equation below:
PSNUBBER = fSW × CS × VIN2
(38)
3) Virtually no ripple at VOUT due to the very low ESR of
the output capacitors.
In this situation, the output voltage ripple is less than
20mV. Therefore, additional ripple is injected into the FB
pin from the switching node SW via a resistor Rinj and a
capacitor Cinj, as shown in Figure 9c. The injected ripple
is:
Step 3: Calculate the damping resistor.
Critical damping occurs at Q = 1:
1
f SW × τ
R1//R2
R inj + R1//R2
(39)
(40)
where:
VIN = Power stage input voltage at VIN pin
D = Duty Cycle
fSW = switching frequency
τ = (R1 // R2 // R inj ) ⋅ C ff
(36)
In equations (39) and (40), it is assumed that the time
constant associated with Cff must be much greater than
the switching period:
Ripple Injection
The VFB ripple required for proper operation of the
MIC2165 gm amplifier and error comparator is 20mV to
100mV. However, the output voltage ripple is generally
designed as 1% to 2% of the voltage. For a low output
voltage, such as a 1V output, the output voltage ripple is
only 10mV to 20mV, and the VFB ripple is less than
20mV. If the VFB ripple is so small that the gm amplifier
and error comparator cannot sense it, the MIC2165 will
lose control and the output voltage is not regulated. In
order to have some amount of VFB ripple, a ripple
injection method is applied for low output voltage ripple
applications.
The applications are divided into three situations
according to the amount of the VFB ripple:
June 2010
R2
× ΔVOUT
R1 + R2
1
f SW × τ
=
T
<< 1
τ
If the voltage divider resistors R1 and R2 are in the kΩ
range, a Cff of 1nF to 100nF can easily satisfy the large
time constant consumption. Also, a 100nF injection
capacitor Cinj is used in order to be considered as short
for a wide range of the frequencies.
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Micrel, Inc.
MIC2165
The process of sizing the ripple injection resistor and
capacitors is:
Step 1. Select Cff to feed all output ripples into the
feedback pin and make sure the large time constant
assumption is satisfied. Typical choice of Cff is 1nF to
100nF if R1 and R2 are in kΩ range.
Step 2. Select Rinj according to the expected feedback
voltage ripple. According to the equation (39),
Figure 9a.
R2
× ΔVOUT > 20mV
R1 + R2
K div =
ΔVFB(PP )
VIN
×
f SW × τ
D × (1 − D)
(41)
Then the value of Rinj is obtained as:
R inj = (R1 // R2) × (
1
K div
− 1)
(42)
Step 3. Select Cinj as 100nF, which could be considered
as short for a wide range of the frequencies.
Figure 9b.
R2
× ΔVOUT < 20mV and ΔVOUT > 20mV
R1 + R2
Figure 9c. ΔVOUT < 20mV
June 2010
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Micrel, Inc.
MIC2165
Inductor
PCB Layout Guidelines
Warning!!! To minimize EMI and output noise, follow
these layout recommendations.
PCB Layout is critical to achieve reliable, stable and
efficient performance. A ground plane is required to
control EMI and minimize the inductance in power,
signal and return paths.
The following guidelines should be followed to insure
proper operation of the MIC2165 converter.
IC
•
Place the IC and MOSFETs close to the point of
load (POL).
•
Use fat traces to route the input and output power
lines.
•
Signal and power grounds should be kept separate
and connected at only one location.
•
The exposed pad (ePad) on the bottom of the IC
must be connected to the ground through several
vias.
•
The feedback resistors should be placed close to the
FB pin. The top feedback resistor should connect
directly to the output node. Run this trace away from
the switch node (SW).
•
Keep the inductor connection to the switch node
(SW) short.
•
Do not route any digital lines underneath or close to
the inductor.
•
Keep the switch node (SW) away from the feedback
(FB) pin.
•
The SW pin should be connected directly to the
drain of the low-side MOSFET to accurate sense the
voltage across the low-side MOSFET.
•
To minimize noise, place a ground plane underneath
the inductor.
Output Capacitor
•
Use a wide trace to connect the output capacitor
ground terminal to the input capacitor ground
terminal.
•
Phase margin will change as the output capacitor
value and ESR changes. Contact the factory if the
output capacitor is different from what is shown in
the BOM.
•
The feedback trace should be separate from the
power trace and connected as close as possible to
the output capacitor. Sensing a long high current
load trace can degrade the DC load regulation.
Schottky Diode
Input Capacitor
•
Place the VIN input capacitor next.
•
•
Place the VIN input capacitors on the same side of
the board and as close to the MOSFETs as
possible.
Place the Schottky diode on the same side of the
board as the MOSFETs and VIN input capacitor.
•
•
Keep both the VIN and PGND connections short.
The connection from the Schottky diode’s Anode to
the input capacitors ground terminal must be as
short as possible.
•
Place several vias to the ground plane close to the
VIN input capacitor ground terminal.
•
Use either X7R or X5R dielectric input capacitors.
Do not use Y5V or Z5U type capacitors.
•
Do not replace the ceramic input capacitor with any
other type of capacitor. Any type of capacitor can be
placed in parallel with the input capacitor.
•
•
•
•
The diode’s Cathode connection to the switch node
(SW) must be keep as short as possible.
RC Snubber
•
Place the RC snubber on the same side of the board
and as close to the MOSFETs as possible.
MOSFETs
If a Tantalum input capacitor is placed in parallel
with the input capacitor, it must be recommended for
switching regulator applications and the operating
voltage must be derated by 50%.
In “Hot-Plug” applications, a Tantalum or Electrolytic
bypass capacitor must be used to limit the overvoltage spike seen on the input supply with power is
suddenly applied.
The 2.2µF (minumum) capacitors, which connect to
the VDD terminal, must be located right at the IC. The
VDD terminal is very noise sensitive and placement
of the capacitor is very critical. Connections must be
made with wide trace.
June 2010
20
•
Low-side MOSFET gate drive trace (DL pin to
MOSFET gate pin) must be short and routed over a
ground plane. The ground plane should be the
connection between the MOSFET source and PGND.
•
Chose a low-side MOSFET with a high CGS/CGD ratio
and a low internal gate resistance to minimize the
effect of dv/dt inducted turn-on.
•
Do not put a resistor between the LSD output and
the gate.
•
Use a 4.5V VGS rated MOSFET. Its higher gate
threshold voltage is more immune to glitches than a
2.5V or 3.3V rated MOSFET. MOSFETs that are
rated for operation at less than 4.5V VGS should not
be used.
M9999-060810-D
Micrel, Inc.
MIC2165
Evaluation Board Schematic
Figure 10. Schematic of MIC2165 8-24 VIN to 1.2 VOUT/10A Evaluation Board
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MIC2165
Bill of Materials
Item
C1
C2,C3
Part Name
B41125A7227M
C7
12105C475KAZ2A
AVX
06035C104KAT2A
TDK
C1608X7R1H102K
06035C472KAT2A
GRM188R71H472KA01D
C13
D1
AVX
AVX
TDK
AVX
Murata
AVX
SD103BWS
1nF Ceramic Capacitor, X7R, Size 0603,
50V
Murata
12106D107MAT2A
SD103BWS-7
10µF Ceramic Capacitor, X5R, Size 0805,
10V
Murata
TDK
6SEPC560MX
0.1µF Ceramic Capacitor, X7R, Size
0603, 50V
(5)
C1608X7R1H472K
GRM32ER60J107ME20L
C15
AVX
C1608X7R1H104K
06035C102KAT2A
4.7µF Ceramic Capacitor, X7R, Size
1210, 50V
(4)
Murata
GRM188R71H102KA01D
C12
Murata
Qty
220µF Aluminum Capacitor, SMD, 35V
(3)
GRM188R71H104KA93D
0805ZD106KAT2A
Description
(1)
(2)
Vishay
GRM21BR61A106KE19L
C11
EPCOS
222215095001E3
GRM32ER71H475KA88L
C6,C8,C10
Manufacturer
Murata
(6)
SANYO
Diodes Inc
Q1
FDS6298
Fairchild
(9)
Q2
FDS8672S
Fairchild
(9)
R1,R14
CRCW06032R21FKEA
R2
R3
(8)
1
1
100µF Ceramic Capacitor, X5R, Size
1210, 6.3V
1
560µF OSCON Capacitor, 6.3V
1
1
1.0µH Inductor, 24A Saturation Current
1
30V 13A N-Channel MOSFET 12mΩ
Rds(on) @ 4.5V
1
30V 18A N-Channel MOSFET 7mΩ
Rds(on) @ 4.5V
1
Vishay/Dale
2.21Ω Resistor, Size 0603, 1%
2
CRCW08051R21FKEA
Vishay/Dale
1.21Ω Resistor, Size 0805, 1%
1
CRCW060319K6FKEA
Vishay/Dale
19.6kΩ Resistor, Size 0603, 1%
1
R4
CRCW06032K49FKEA
Vishay/Dale
2.49kΩ Resistor, Size 0603, 1%
1
R5
CRCW06034K99FKEA
Vishay/Dale
4.99kΩ Resistor, Size 0603, 1%
1
R13
CRCW060320R0FKEA
Vishay/Dale
20Ω Resistor, Size 0603, 1%
1
R15,R16
CRCW060310K0FKEA
Vishay/Dale
10kΩ Resistor, Size 0603, 1%
2
R17
CRCW060349R9FKEA
Vishay/Dale
49.9Ω Resistor, Size 0603, 1%
1
U1
MIC2165YMME
600kHz Buck Controller
1
June 2010
Cooper Bussmann
3
1
Small Signal Schottky Diode
Vishay
HCF1305-1R0-R
2
4.7nF Ceramic Capacitor, X7R, Size
0603, 50V
(7)
L1
1
(10)
Micrel Inc.
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MIC2165
Notes:
1.
EPCOS: www.epcos.com
2.
Vishay: www.vishay.com
3.
AVX: www.avx.com
4.
MuRata: www.murata.com
5.
TDK: www.tdk.com
6.
Sanyo: www.sanyo.com
7.
Diode Inc.: www.diodes.com
8.
Cooper Bussmann: www.cooperbussmann.com
9.
Fairchild: www.fairchildsemi.com
10. Micrel, Inc: www.micrel.com
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MIC2165
PCB Layout
Figure 11. MIC2165 10A Evaluation Board Top Layer
Figure 12. MIC2165 10A Evaluation Board Bottom Layer
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MIC2165
Figure 13. MIC2165 10A Evaluation Board Mid-Layer 1 (GND Plane)
Figure 14. MIC2165 10A Evaluation Board Mid-Layer 2
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MIC2165
Package Information
10-Pin ePad MSOP (MME)
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MIC2165
Recommended Landing Pattern
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its
use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product
can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant
into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A
Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully
indemnify Micrel for any damages resulting from such use or sale.
© 2010 Micrel, Incorporated.
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