LTC1147-3.3 LTC1147-5/LTC1147L High Efficiency Step-Down Switching Regulator Controllers U DESCRIPTIO FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ Very High Efficiency: Over 95% Possible Wide VIN Range: 3.5V* to 16V Current Mode Operation for Excellent Line and Load Transient Response High Efficiency Maintained Over Three Decades of Output Current Low 160µA Standby Current at Light Loads Logic Controlled Micropower Shutdown: IQ < 20µA Short-Circuit Protection Very Low Dropout Operation: 100% Duty Cycle High Efficiency in a Small Amount of Board Space Output Can Be Externally Held High in Shutdown Available in 8-Pin SO Package U APPLICATIO S ■ ■ ■ ■ ■ ■ The LTC®1147 series are step-down switching regulator controllers featuring automatic Burst ModeTM operation to maintain high efficiencies at low output currents. These devices drive an external P-channel power MOSFET at switching frequencies exceeding 400kHz using a constant off-time current mode architecture providing constant ripple current in the inductor. The operating current level is user-programmable via an external current sense resistor. Wide input supply range allows operation from 3.5V* to 14V (16V maximum). Constant off-time architecture provides low dropout regulation limited by only the RDS(ON) of the external MOSFET and resistance of the inductor and current sense resistor. The LTC1147 series incorporates automatic power saving Burst Mode operation to reduce switching losses when load currents drop below the level required for continuous operation. Standby power is reduced to only 2mW at VIN = 10V (at IOUT = 0). Load currents in Burst Mode operation are typically 0mA to 300mA. Notebook and Palmtop Computers Portable Instruments Battery-Operated Digital Devices Cellular Telephones DC Power Distribution Systems GPS Systems For applications where even higher efficiency is required, refer to the LTC1148 data sheet and Application Note 54. , LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode is a trademark of Linear Technology Corporation. *LTC1147L and LTC1147L-3.3 only. UO TYPICAL APPLICATI LTC1147-5 Efficiency VIN (5.2V TO 14V) + 1µF VIN PDRIVE LTC1147-5 SHDN SENSE + ITH SENSE – CT 0V = NORMAL >1.5V = SHUTDOWN RC 1k CC 3300pF CT 470pF GND P-CHANNEL Si4431DY L* 50µH CIN 100µF RSENSE** 0.05Ω 95 VOUT 5V/2A 1000pF + D1 MBRD330 COUT 390µF EFFICIENCY (%) + 100 VIN = 6V 90 VIN = 10V 85 80 75 LT1147 • F01 *COILTRONICS CTX50-2-MP **KRL SL-1-C1-0R050J 70 0.001 0.01 0.1 1 LOAD CURRENT (A) LT1147 • TA01 Figure 1. High Efficiency Step-Down Converter sn1147 1147fds 1 LTC1147-3.3 LTC1147-5/LTC1147L U W W W ABSOLUTE AXI U RATI GS Input Supply Voltage (Pin 1) .................... 16V to – 0.3V Continuous Output Current (Pin 8) ...................... 50mA Sense Voltages (Pins 4, 5) VIN ≥ 12.7V ...........................................13V to – 0.3V VIN < 12.7V ............................... (VIN + 0.3V) to – 0.3V Operating Ambient Temperature Range LTC1147C ............................................... 0°C to 70°C LTC1147I ............................................. –40°C to 85°C Extended Commercial Temperature Range (Note 4) ................. – 40°C to 85°C Junction Temperature (Note 1) ............................ 125°C Storage Temperature Range ................ – 65°C to 150°C Lead Temperature (Soldering, 10 sec)................. 300°C U W U PACKAGE/ORDER I FOR ATIO ORDER PART NUMBER TOP VIEW VIN 1 8 PDRIVE CT 2 7 ITH 3 6 SENSE – 4 5 GND SHDN (VFB*) SENSE + N8 PACKAGE S8 PACKAGE 8-LEAD PLASTIC DIP 8-LEAD PLASTIC SO * ADJUSTABLE OUTPUT VERSION LTC1147CN8-3.3 LTC1147CN8-5 LTC1147CS8-3.3 LTC1147CS8-5 LTC1147IS8-3.3 S8 PART MARKING LTC1147IS8-5 LTC1147LCS8 LTC1147LCS8-3.3 LTC1147LIS8 11473 11475 1147I3 1147I5 1147L 1147L3 1147LI TJMAX = 125°C, θJA = 110°C/W (N) TJMAX = 125°C, θJA = 150°C/W (S) Consult factory for Military grade parts. ELECTRICAL CHARACTERISTICS TA = 25°C, VIN = 10V, VSHDN = 0V, unless otherwise noted. SYMBOL PARAMETER CONDITIONS V6 Feedback Voltage (LTC1147L) VIN = 9V I6 Feedback Current (LTC1147L) VOUT Regulated Output Voltage LTC1147-3.3, LTC1147L-3.3 LTC1147-5 VIN = 9V ILOAD = 700mA ILOAD = 700mA Output Voltage Line Regulation VIN = 7V to 12V, ILOAD = 50mA Output Voltage Load Regulation LTC1147-3.3, LTC1147L-3.3 LTC1147-5 5mA < ILOAD < 2A 5mA < ILOAD < 2A ∆VOUT IQ ● MIN TYP MAX UNITS 1.21 1.25 1.29 V 0.2 1 µA 3.23 4.90 3.33 5.05 3.43 5.20 V V – 40 0 40 mV 40 60 65 100 mV mV ● Burst Mode Output Ripple ILOAD = 0A Input DC Supply Current (Note 2) LTC1147 Series Normal Mode Sleep Mode Sleep Mode (LTC1147-5) Shutdown LTC1147L Series Normal Mode Sleep Mode Shutdown (LTC1147L-3.3) (Note 5) ● ● ● ● 50 mVP-P 4V < VIN < 12V 4V < VIN < 12V 5V < VIN < 12V VSHDN = 2.1V, 4V < VIN < 12V 1.6 160 160 10 2.1 230 230 20 mA µA µA µA 3.5V < VIN < 12V 3.5V < VIN < 12V VSHDN = 2.1V, 3.5V < VIN < 12V 1.6 160 10 2.1 230 20 mA µA µA sn1147 1147fds 2 LTC1147-3.3 LTC1147-5/LTC1147L ELECTRICAL CHARACTERISTICS SYMBOL PARAMETER V5 – V 4 Current Sense Threshold Voltage (Note 6) LTC1147-3.3, LTC1147L-3.3 LTC1147–5 LTC1147L V6 SHDN Pin Threshold LTC1147-3.3/LTC1147-5/LTC1147L-3.3 I6 SHDN Pin Input Current LTC1147-3.3/LTC1147-5/LTC1147L-3.3 TA = 25°C, VIN = 10V, VSHDN = 0V, unless otherwise noted. CONDITIONS MIN VSENSE– = VOUT + 100mV (Forced) VSENSE– = VOUT – 100mV (Forced) VSENSE– = VOUT + 100mV (Forced) VSENSE– = VOUT – 100mV (Forced) VSENSE– = 5V, V6 = VOUT/4 + 25mV (Forced) VSENSE– = 5V, V6 = VOUT/4 – 25mV (Forced) I2 CT Pin Discharge Current VOUT in Regulation, VSENSE VOUT = 0V tOFF Off-Time (Note 3) CT = 390pF, ILOAD = 700mA tr, tf Driver Output Transition Times CL = 3000pF (Pin 8), VIN = 6V UNITS 130 0.5 0.8 2 V 1.2 5 µA 70 2 90 10 µA µA 130 50 OUT MAX 25 150 25 150 25 150 130 0V < VSHDN < 8V, VIN = 16V –=V TYP 4 170 170 170 mV mV mV mV mV mV 5 6 µs 100 200 ns – 40°C ≤ TA ≤ 85°C (Note 4), VIN = 10V, unless otherwise noted. SYMBOL PARAMETER V6 Feedback Voltage (LTC1147L) VIN = 9V VOUT Regulated Output Voltage LTC1147-3.3/LTC1147L-3.3 LTC1147-5 VIN = 9V ILOAD = 700mA ILOAD = 700mA Input DC Supply Current (Note 2) LTC1147 Series Normal Mode Sleep Mode Sleep Mode (LTC1147-5) Shutdown LTC1147L Series Normal Mode Sleep Mode Shutdown (LTC1147L-3.3) (Note 5) IQ V 5 – V4 Current Sense Threshold Voltage (Note 6) LTC1147-3.3 LTC1147-5 LTC1147L V6 tOFF CONDITIONS MIN TYP MAX UNITS ● 1.20 1.25 1.30 V ● ● 3.17 4.85 3.33 5.05 3.43 5.20 V V 4V < VIN < 12V 4V < VIN < 12V 5V < VIN < 12V VSHDN = 2.1V, 4V < VIN < 12V 1.6 160 160 10 2.4 260 260 22 mA µA µA µA 3.5V < VIN < 12V 3.5V < VIN < 12V VSHDN = 2.1V, 3.5V < VIN < 12V 1.6 160 10 2.4 260 22 mA µA µA VSENSE – = VOUT + 100mV (Forced) VSENSE – = VOUT – 100mV (Forced) VSENSE – = VOUT + 100mV (Forced) VSENSE – = VOUT – 100mV (Forced) VSENSE– = 5V, 6V = VOUT/4 + 25mV (Forced) VSENSE– = 5V, 6V = VOUT/4 – 25mV (Forced) 125 25 150 25 150 25 150 ● 125 ● 125 185 185 185 mV mV mV mV mV mV SHDN Pin Threshold LTC1147-3.3/LTC1147-5/LTC1147L-3.3 0V < VSHDN < 8V, VIN = 16V 0.5 0.8 2 V Off-Time (Note 3) CT = 390pF, ILOAD = 700mA 3.8 5 6.5 µs The ● denotes specifications which apply over the full specified temperature range. Note 1: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formulas: LTC1147CN8-3.3/LTC1147CN8-5: TJ = TA + (PD)(110°C/W) LTC1147LIS/LTC1147IS8/LTC1147LCS/ LTC1147CS8-3.3/LTC1147CS8-5: TJ = TA + (PD)(150°C/W) Note 2: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. See Applications Information. Note 3: In applications where RSENSE is placed at ground potential, the offtime increases approximately 40%. Note 4: The LTC1147C is guaranteed to meet specified performance from 0°C to 70°C and is designed, characterized and expected to meet these extended temperature limits, but is not tested at – 40°C and 85°C. The LTC1147I is guaranteed to meet the extended temperature limits. Note 5: The LTC1147L/LTC1147L-3.3 allow operation to VIN = 3.5V. Note 6: The LTC1147L is tested with external feedback resistors resulting in a nominal output voltage of 2.5V. sn1147 1147fds 3 LTC1147-3.3 LTC1147-5/LTC1147L U W TYPICAL PERFOR A CE CHARACTERISTICS Efficiency vs Input Voltage Line Regulation 100 FIGURE 1 CIRCUIT 98 Load Regulation 40 20 FIGURE 1 CIRCUIT ILOAD = 1A 30 20 ILOAD = 1A 92 90 88 ILOAD = 100mA 86 –20 10 ∆VOUT (mV) 94 ∆VOUT (mV) EFFICIENCY (%) 96 0 –10 –40 VIN = 12V –60 –80 –30 82 80 –40 4 0 16 12 8 INPUT VOLTAGE (V) –100 0 8 4 12 20 1.6 VSHUTDOWN = 2V (NOT AVAILABLE ON LTC1147L) 18 VOUT = 5V 1.4 1.2 0.9 0.6 NORMALIZED FREQUENCY SUPPLY CURRENT (µA) ACTIVE MODE 14 12 10 8 6 4 0.3 SLEEP MODE 0 0 2 4 8 10 12 14 6 INPUT VOLTAGE (V) 16 0 18 2 4 6 8 10 12 14 INPUT VOLTAGE (V) Gate Charge Supply Current 10 60 4 16 2 260 80 200 140 OPERATING FREQUENCY (kHz) LTC1147 • G07 2 4 10 8 6 (VIN – VOUT) VOLTAGE (V) Current Sense Threshold Voltage MAXIMUM THRESHOLD 150 50 40 30 0 1 2 125 100 75 50 MINIMUM THRESHOLD 25 LTC1147-5 0 12 175 LTC1147-3.3 20 0 LTC1148 • G06 VSENSE – = VOUT 10 0 0.4 0 20 QP = 29nC 0.6 18 SENSE VOLTAGE (mV) 12 70 OFF-TIME (µs) 80 6 25°C 0.8 Off-Time vs VOUT 14 QP = 50nC 70°C 1.0 LTC1147 • G05 LTC1147 • G04 8 0°C 1.2 0.2 2 0 2.5 2.0 Operating Frequency vs (VIN – VOUT) 16 1.5 1.0 1.5 LOAD CURRENT (A) LTC1147 • G03 Supply Current in Shutdown NOT INCLUDING GATE CHARGE CURRENT 1.8 0.5 LTC1147 • G02 DC Supply Current 2.1 0 16 INPUT VOLTAGE (V) LTC1147 • G01 SUPPLY CURRENT (mA) VIN = 6V –20 84 GATE CHARGE CURRENT (mA) FIGURE 1 CIRCUIT RSENSE = 0.05Ω 0 0 3 4 5 OUTPUT VOLTAGE (V) LTC1147 • G08 0 20 60 40 TEMPERATURE (°C) 80 100 LTC1147 • G09 sn1147 1147fds 4 LTC1147-3.3 LTC1147-5/LTC1147L U U U PI FU CTIO S VIN (Pin 1): Main Supply Pin. Must be closely decoupled to ground Pin 7. CT (Pin 2): External capacitor CT from Pin 2 to ground sets the operating frequency. The actual frequency is also dependent upon the input voltage. ITH (Pin 3): Gain Amplifier Decoupling Point. The current comparator threshold increases with the Pin 3 voltage. SENSE – (Pin 4): Connects to internal resistive divider which sets the output voltage. Pin 4 is also the (–) input for the current comparator. SENSE + (Pin 5): The (+) input to the current comparator. A built-in offset between Pins 4 and 5 in conjunction with RSENSE sets the current trip threshold. SHDN/VFB (Pin 6): When grounded, the fixed output versions of the LTC1147 family operate normally. Pulling Pin 6 high holds the P-channel MOSFET off and puts the LTC1147 in micropower shutdown mode. Requires CMOS logic signal with tr, tf < 1µs. Do not leave this pin floating. On the LTC1147L this pin serves as the feedback pin from an external resistive divider used to set the output voltage. GND (Pin 7): Two independent ground lines must be routed separately to: 1) the (–) terminal of COUT, and 2) the cathode of the Schottky diode and (–) terminal of CIN. PDRIVE (Pin 8): High current drive for the P-channel MOSFET. Voltage swing at this pin is from VIN to ground. U U W FU CTIO AL DIAGRA Pin 6 Connection Shown For LTC1147-3.3 and LTC1147-5; Changes Create LTC1147L. 1 VIN SENSE+ SENSE – 5 4 VFB 8 PDRIVE 6 7 GND – V + SLEEP – R S + + VTH2 VTH1 + 5pF VOS – + 13k T ITH 3 G + – – – S 25mV TO 150mV C Q 1.25V 100k 2 CT OFF-TIME CONTROL VIN SENSE – SHDN 6 REFERENCE LTC1147 • FD sn1147 1147fds 5 LTC1147-3.3 LTC1147-5/LTC1147L U OPERATIO (Refer to Functional Diagram) The LTC1147 series uses a current mode, constant offtime architecture to switch an external P-channel power MOSFET. Operating frequency is set by an external capacitor at CT (Pin 2). The output voltage is sensed by an internal voltage divider connected to SENSE– (Pin 4). A voltage comparator V, and a gain block G, compare the divided output voltage with a reference voltage of 1.25V. To optimize efficiency, the LTC1147 series automatically switchs between two modes of operation, burst and continuous. The voltage comparator is the primary control element when the device is in Burst Mode operation, while the gain block controls the output voltage in continuous mode. During the switch “on” cycle in continuous mode, current comparator C monitors the voltage between Pins 4 and 5 connected across an external shunt in series with the inductor. When the voltage across the shunt reaches its threshold value, the PDRIVE output is switched to VIN, turning off the P-channel MOSFET. The timing capacitor connected to Pin 2 is now allowed to discharge at a rate determined by the off-time controller. The discharge current is made proportional to the output voltage (measured by Pin 4) to model the inductor current, which decays at a rate which is also proportional to the output voltage. When the voltage on the timing capacitor has discharged past VTH1, comparator T trips, setting the flip-flop. This causes the PDRIVE output to go low turning the P-channel MOSFET back on. The cycle then repeats. As the load current increases, the output voltage decreases slightly. This causes the output of the gain stage (Pin 3) to increase the current comparator threshold, thus tracking the load current. The sequence of events for Burst Mode operation is very similar to continuous operation with the cycle interrupted by the voltage comparator. When the output voltage is at or above the desired regulated value, the P-channel MOSFET is held off by comparator V and the timing capacitor continues to discharge below VTH1. When the timing capacitor discharges past VTH2, voltage comparator S trips, causing the internal sleep line to go low. The circuit now enters sleep mode with the power MOSFET turned off. In sleep mode, a majority of the circuitry is turned off, dropping the quiescent current from 1.6mA to 160µA. The load current is now being supplied from the output capacitor. When the output voltage has dropped by the amount of hysteresis in comparator V, the P-channel MOSFET is again turned on and this process repeats. To avoid the operation of the current loop interfering with Burst Mode operation, a built-in offset VOS is incorporated in the gain stage. This prevents the current comparator threshold from increasing until the output voltage has dropped below a minimum threshold. Using constant off-time architecture, the operating frequency is a function of the input voltage. To minimize the frequency variation as dropout is approached, the off-time controller increases the discharge current as VIN drops below VOUT + 1.5V. In dropout the P-channel MOSFET is turned on continuously (100% duty cycle), providing low dropout operation with VOUT ≈ VIN. U W U U APPLICATIO S I FOR ATIO LTC1147L Adjustable Applications When an output voltage other than 3.3V or 5V is required, the LTC1147L adjustable version is used with an external resistive divider from VOUT to VFB (Pin 6) (see Figure 7). The regulated voltage is determined by: ) VOUT = 1.25 1 + R2 R1 ) To prevent stray pickup a 100pF capacitor is suggested across R1 located close to the LTC1147L. For Figure 1 applications with VOUT below 2V, or when RSENSE is moved to ground, the current sense comparator inputs operate near ground. When the current comparator is operated at less than 2V common mode, the off-time increases approximately 40%, requiring the use of a smaller timing capacitor CT. sn1147 1147fds 6 LTC1147-3.3 LTC1147-5/LTC1147L U W U U APPLICATIO S I FOR ATIO The basic LTC1147 application circuit is shown in Figure 1. External component selection is driven by the load requirement and begins with the selection of RSENSE. Once RSENSE is known, CT and L can be chosen. Next, the power MOSFET and D1 are selected. Finally, CIN and COUT are selected and the loop is compensated. The circuit shown in Figure 1 can be configured for operation up to an input voltage of 16V. If the application requires higher input voltage, then the synchronous switched LTC1149 should be used. Consult factory for lower minimum input voltage version. 0.20 RSENSE (Ω) 0.15 0.05 0 Since efficiency generally increases with ripple current, the maximum allowable ripple current is assumed, i.e., IRIPPLE(P-P) = 25mV/RSENSE (see CT and L Selection for Operating Frequency). Solving for RSENSE and allowing a margin for variations in the LTC1147 series and external component values yields: RSENSE = 100mV IMAX A graph for selecting RSENSE versus maximum output current is given in Figure 2. The load current below in which Burst Mode operation commences, IBURST and the peak short-circuit current ISC(PK), both track IMAX. Once RSENSE has been chosen, IBURST and ISC(PK) can be predicted from the following: IBURST ≈ 15mV RSENSE ISC(PK) = 150mV RSENSE 0 1 3 4 2 MAXIMUM OUTPUT CURRENT (A) 5 LTC1147 • F02 RSENSE Selection for Output Current RSENSE is chosen based on the required output current. The LTC1147 series current comparator has a threshold range which extends from a minimum of 25mV/ RSENSE to a maximum of 150mV/RSENSE. The current comparator threshold sets the peak of the inductor ripple current, yielding a maximum output current IMAX equal to the peak value less half the peak-to-peak ripple current. For proper Burst Mode operation, IRIPPLE(P-P) must be less than or equal to the minimum current comparator threshold. 0.10 Figure 2. Selecting RSENSE The LTC1147 series automatically extend tOFF during a short circuit to allow sufficient time for the inductor current to decay between switch cycles. The resulting ripple current causes the average short-circuit current ISC(AVG) to be reduced to approximately IMAX. L and CT Selection for Operating Frequency The LTC1147 series use a constant off-time architecture with tOFF determined by an external timing capacitor CT. Each time the P-channel MOSFET switch turns on, the voltage on CT is reset to approximately 3.3V. During the off-time, CT is discharged by a current which is proportional to VOUT. The voltage on CT is analogous to the current in inductor L, which likewise decays at a rate proportional to VOUT. Thus the inductor value must track the timing capacitor value. The value of CT is calculated from the desired continuous mode operating frequency: CT = ) ) VIN – VOUT 1 (1.3)(104)(f) VIN + VD Where VD is the drop across the Schottky diode. A graph for selecting CT versus frequency including the effects of input voltage is given in Figure 3. As the operating frequency is increased the gate charge losses will reduce efficiency (see Efficiency Considerations). The complete expression for operating frequency sn1147 1147fds 7 LTC1147-3.3 LTC1147-5/LTC1147L U U W U APPLICATIO S I FOR ATIO 1000 VSENSE – = VOUT = 5V CAPACITANCE (pF) 800 Inductor Core Selection 600 400 VIN = 12V 200 VIN = 7V VIN = 10V 0 0 200 100 FREQUENCY (kHz) 300 LTC1147 • F03 Figure 3. Timing Capacitor Value is given by: f≈ 1 tOFF series will delay entering Burst Mode operation and efficiency will be degraded at low currents. ) 1– VOUT VIN ) where: ) ) V tOFF = (1.3)(104)(CT) REG VOUT VREG is the desired output voltage (i.e., 5V, 3.3V). VOUT is the measured output voltage. Thus VREG/VOUT = 1 in regulation. Note that as VIN decreases, the frequency decreases. When the input to output voltage differential drops below 1.5V, the LTC1147 reduces tOFF by increasing the discharge current in CT. This prevents audible operation prior to dropout. Once the frequency has been set by CT, the inductor L must be chosen to provide no more than 25mV/RSENSE of peak-to-peak inductor ripple current. This results in a minimum required inductor value of: LMIN = (5.1)(105)(RSENSE)(CT)(VREG) As the inductor value is increased from the minimum value, the ESR requirements for the output capacitor are eased at the expense of efficiency. If too small an inductor is used, the inductor current will become discontinuous before the LTC1147 series enters Burst Mode operation. A consequence of this is that the LTC1147 Once the minimum value for L is known, the type of inductor must be selected. Highest efficiency will be obtained using ferrite, Kool Mµ® (from Magnetics, Inc.) or molypermalloy (MPP) cores. Lower cost powdered iron cores provide suitable performance but cut efficiency by 3% to 5%. Actual core loss is independent of core size for a fixed inductor value, but it is very dependent on inductance selected. As inductance increases, core losses go down. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core loss, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates “hard,” which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequent output voltage ripple which can cause Burst Mode operation to be falsely triggered in the LTC1147. Do not allow the core to saturate! Kool Mµ is a very good, low loss core material for toroids with a “soft” saturation characteristic. Molypermalloy is slightly more efficient at high (>200kHz) switching frequencies but quite a bit more expensive. Toroids are very space efficient, especially when you can use several layers of wire. Because they generally lack a bobbin, mounting is more difficult. However, new designs for surface mount are available from Coiltronics, Sumida and Beckman Industrial Corp. which do not increase the height significantly. Power MOSFET Selection An external P-channel power MOSFET must be selected for use with the LTC1147 series. The main selection criteria for the power MOSFET are the threshold voltage VGS(TH) and “on” resistance RDS(ON). The minimum input voltage determines whether a standard threshold or logic-level threshold MOSFET must be Kool Mµ is a registered trademark of Magnetics, Inc. sn1147 1147fds 8 LTC1147-3.3 LTC1147-5/LTC1147L U W U U APPLICATIO S I FOR ATIO used. For VIN > 8V, a standard threshold MOSFET (VGS(TH) < 4V) may be used. If VIN is expected to drop below 8V, a logic-level threshold MOSFET (VGS(TH) < 2.5V) is strongly recommended. When a logic-level MOSFET is used, the LTC1147 supply voltage must be less than the absolute maximum VGS ratings for the MOSFET. The maximum output current IMAX determines the RDS(ON) requirement for the power MOSFET. When the LTC1147 series is operating in continuous mode, the simplifying assumption can be made that either the MOSFET or Schottky diode is always conducting the average load current. The duty cycles for the MOSFET and diode are given by: V P-Ch Duty Cycle = OUT VIN (V – VOUT + VD) Schottky Diode Duty Cycle = IN VIN From the duty cycle the required RDS(ON) for the MOSFET can be derived: P-Ch RDS(ON) = (VIN)(PP) (VOUT)(IMAX2)(1 + δP) where PP is the allowable power dissipation and δP is the temperature dependency of RDS(ON). PP will be determined by efficiency and/or thermal requirements (see Efficiency Considerations). (1 + δ) is generally given for a MOSFET in the form of a normalized RDS(ON) vs temperature curve, but δ = 0.007/°C can be used as an approximation for low voltage MOSFETs. Output Diode Selection (D1) The Schottky diode D1 shown in Figure 1 only conducts during the off-time. It is important to adequately specify the diode peak current and average power dissipation so as not to exceed the diode ratings. The most stressful condition for the output diode is under short circuit (VOUT = 0V). Under this condition the diode must safely handle ISC(PK) at close to 100% duty cycle. Under normal load conditions the average current conducted by the diode is: ID1 = (VIN – VOUT + VD) (ILOAD) VIN Remember to keep lead lengths short and observe proper grounding (see Board Layout Checklist) to avoid ringing and increased dissipation. The forward voltage drop allowable in the diode is calculated from the maximum short-circuit current as: VF ≈ PD ISC(PK) where PD is the allowable power dissipation and will be determined by efficiency and/or thermal requirements (see Efficiency Considerations). CIN and COUT Selection In continuous mode, the source current of the P-channel MOSFET is a square wave of duty cycle VOUT/VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: CIN Required IRMS ≈ IMAX [VOUT(VIN – VOUT)]1/2 VIN This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that capacitor manufacturer’s ripple current ratings are often based on only 2000 hours of life. This makes it advisable to further derate the capacitor, or to choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet size or height requirements in the design. Always consult the manufacturer if there is any question. An additional 0.1µF to 1µF ceramic decoupling capacitor is also required on VIN (Pin 1) for high frequency decoupling. The selection of COUT is driven by the required effective series resistance (ESR). The ESR of COUT must be less than twice the value of RSENSE for proper operation of the LTC1147: COUT Required ESR < 2RSENSE sn1147 1147fds 9 LTC1147-3.3 LTC1147-5/LTC1147L U W U U APPLICATIO S I FOR ATIO Optimum efficiency is obtained by making the ESR equal to RSENSE. As the ESR is increased up to 2RSENSE, the efficiency degrades by less than 1%. If the ESR is greater than 2RSENSE, the voltage ripple on the output capacitor will prematurely trigger Burst Mode operation, resulting in disruption of continuous mode and an efficiency hit which can be several percent. Manufacturers such as Nichicon and United Chemicon should be considered for high performance capacitors. The OS-CON semiconductor dielectric capacitor available from Sanyo has the lowest ESR/size ratio of any aluminum electrolytic at a somewhat higher price. Once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. In surface mount applications multiple capacitors may have to be paralleled to meet the capacitance, ESR or RMS current handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available in surface mount configurations. In the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalums, available in case heights ranging from 2mm to 4mm. For example, if 200µF/10V is called for in an application requiring 3mm height, two AVX 100µF/10V (P/N TPSD 107K010) could be used. Consult the manufacturer for other specific recommendations. At low supply voltages, a minimum capacitance at COUT is needed to prevent an abnormal low frequency operating 1000 L = 50µH RSENSE = 0.02Ω COUT (µF) 800 L = 25µH RSENSE = 0.02Ω 600 200 L = 50µH RSENSE = 0.05Ω 0 1 3 4 2 (VIN – VOUT) VOLTAGE (V) Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in DC (resistive) load current. When a load step occurs, VOUT shifts by an amount equal to ∆ILOAD(ESR), where ESR is the effective series resistance of COUT. ∆ILOAD also begins to charge or discharge COUT until the regulator loop adapts to the current change and returns VOUT to its steady state value. During this recovery time VOUT can be monitored for overshoot or ringing which would indicate a stability problem. The external components shown in the Figure 1 circuit will prove adequate compensation for most applications. A second, more severe transient is caused by switching in loads with large (>1µF) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem if the load switch resistance is low and it is driven quickly. The only solution is to limit the rise time of the switch drive so that the load rise time is limited to approximately (25)CLOAD. Thus a 10µF capacitor would require a 250µs rise time, limiting the charging current to about 200mA. Efficiency Considerations 400 0 mode (see Figure 4). When COUT is made too small, the output ripple at low frequencies will be large enough to trip the voltage comparator. This causes Burst Mode operation to be activated when the LTC1147 series would normally be in continuous operation. The effect is most pronounced with low values of RSENSE and can be improved by operating at higher frequencies with lower values of L. The output remains in regulation at all times. 5 LTC1147 • F04 The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Percent efficiency can be expressed as: %Efficiency = 100% – (L1 + L2 + L3 + ...) Figure 4. Minimum Value of COUT sn1147 1147fds 10 LTC1147-3.3 LTC1147-5/LTC1147L U W U U APPLICATIO S I FOR ATIO Although all dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in LTC1147 circuits: 1) LTC1147 DC bias current, 2) MOSFET gate charge current, 3) I2R losses, and 4) voltage drop of the Schottky diode. 1. The DC supply current is the current which flows into VIN (Pin 1) less the gate charge current. For VIN = 10V the LTC1147 series DC supply current is 160µA for no load, and increases proportionally with load up to a constant 1.6mA after the LTC1147 series has entered continuous mode. Because the DC bias current is drawn from VIN, the resulting loss increases with input voltage. For VIN = 10V the DC bias losses are generally less than 1% for load currents over 30mA. However, at very low load currents the DC bias current accounts for nearly all of the loss. 2. MOSFET gate charge current results from switching the gate capacitance of the power MOSFET. Each time a MOSFET gate is switched from low to high to low again, a packet of charge dQ moves from VIN to ground. The resulting dQ/dt is a current out of VIN which is typically much larger than the DC supply current. In continuous mode, IGATECHG = f(QP). The typical gate charge for a 0.135Ω P-channel power MOSFET is 40nC. This results in IGATECHG = 4mA in 100kHz continuous operation for a 2% to 3% typical midcurrent loss with VIN = 10V. Note that the gate charge loss increases directly with both input voltage and operating frequency. This is the principal reason why the highest efficiency circuits operate at moderate frequencies. Furthermore, it argues against using a larger MOSFET than necessary to control I2R losses, since overkill can cost efficiency as well as money! 3. I2R losses are easily predicted from the DC resistances of the MOSFET, inductor and current shunt. In continuous mode the average output current flows through L and RSENSE, but is “chopped” between the P-channel and Schottky diode. The MOSFET RDS(ON) multiplied by the P-channel duty cycle can be summed with the resistances of L and RSENSE to obtain I2R losses. For example, if RDS(ON) = 0.1Ω, RL = 0.15Ω, and RSENSE = 0.05Ω, then the total resistance is 0.3Ω at VIN ≈ 2VOUT. This results in losses ranging from 3% to 10% as the output current increases from 0.5A to 2A. I2R losses cause the efficiency to roll off at high output currents. 4. The Schottky diode is a major source of power loss at high currents and gets worse at high input voltages. The diode loss is calculated by multiplying the forward voltage drop times the Schottky diode duty cycle multiplied by the load current. For example, assuming a duty cycle of 50% with a Schottky diode forward voltage drop of 0.4V, the loss increases from 0.5% to 8% as the load current increases from 0.5A to 2A. Figure 5 shows how the efficiency losses in a typical LTC1147 series regulator end up being apportioned. The gate charge loss is responsible for the majority of the efficiency lost in the midcurrent region. If Burst Mode operation was not employed at low currents, the gate charge loss alone would cause efficiency to drop to unacceptable levels. With Burst Mode operation, the DC supply current represents the lone (and unavoidable) loss component which continues to become a higher percentage as output current is reduced. As expected, the I2R losses and Schottky diode loss dominate at high load currents. 100 I2R GATE CHARGE EFFICIENCY/LOSS (%) where L1, L2, etc., are the individual losses as a percentage of input power. (For high efficiency circuits only small errors are incurred by expressing losses as a percentage of output power.) 95 LTC1147 IQ SCHOTTKY DIODE 90 85 80 0.01 0.03 0.3 1 0.1 OUTPUT CURRENT (A) 3 LTC1147 • F05 Figure 5. Efficiency Loss sn1147 1147fds 11 LTC1147-3.3 LTC1147-5/LTC1147L U W U U APPLICATIO S I FOR ATIO Other losses including CIN and COUT ESR dissipative losses, MOSFET switching losses, and inductor core losses, generally account for less than 2% total additional loss. fMIN = Design Example PP = As a design example, assume VIN = 5V (nominal), VOUT = 3.3V, IMAX = 1A, and f = 130kHz; RSENSE, CT and L can immediately be calculated: RSENSE = 100mV/1A = 0.1Ω tOFF = (1/130kHz)[1 – (3.3/5)] = 2.61µs CT = 2.61µs/(1.3)(104) = 220pF L = (5.1)(105)(0.1Ω)(220pF)(3.3V) = 33µH Assume that the MOSFET dissipation is to be limited to PP = 250mW. If TA = 50°C and the thermal resistance of the MOSFET is 50°C/ W, then the junction temperatures will be 63°C and δP = 0.007(63 – 25) = 0.27. The required RDS(ON) for the MOSFET can now be calculated: P-Ch RDS(ON) = 5(0.25) = 0.3Ω 3.3(1)2 (1.27) The P-channel requirement can be met by a Si9430DY. Note that the most stringent requirement for the Schottky diode is with VOUT = 0 (i.e., short circuit). During a continuous short circuit, the worst-case Schottky diode dissipation rises to: ) ) 1 3.3 = 102kHz 1– 2.61µs 4.5 3.3(0.125Ω)(1A)2(1.27) = 116mW 4.5 This last step is necessary to assure that the power dissipation and junction temperature of the P-channel are not exceeded. Troubleshooting Hints Since efficiency is critical to LTC1147 series applications, it is very important to verify that the circuit is functioning correctly in both continuous and Burst Mode operation. The waveform to monitor is the voltage on the timing capacitor Pin 2. In continuous mode (ILOAD > IBURST) the voltage on the CT pin should be a sawtooth with a 0.9VP-P swing. This voltage should never dip below 2V as shown in Figure 6a. When load currents are low (ILOAD < IBURST) Burst Mode operation occurs. The voltage on the CT pin now falls to ground for periods of time as shown in Figure 6b. During this time the LTC1147 series are in sleep mode with the quiescent current reduced to 160µA. The inductor current should also be monitored. Look to verify that the peak-to-peak ripple current in continuous mode operation is approximately the same as in Burst Mode operation. PD = ISC(AVG)(VD) 3.3V With the 0.1Ω sense resistor ISC(AVG) = 1A will result, increasing the 0.4V Schottky diode dissipation to 0.4W. CIN will require an RMS current rating of at least 0.5A at temperature, and COUT will require an ESR of 0.1Ω for optimum efficiency. Now allow VIN to drop to its minimum value. At lower input voltages the operating frequency will decrease and the P-channel will be conducting most of the time, causing the power dissipation to increase. At VIN(MIN) = 4.5V, the frequency will decrease and the P-channel will be conducting most of the time causing its power dissipation to increase. At VIN(MIN) = 4.5V: 0V Figure 6a. Continuous Mode Operation CT Waveform 3.3V 0V LTC1147 • F06 Figure 6b. Burst Mode Operation CT Waveform If Pin 2 is observed falling to ground at high output currents, it indicates poor decoupling or improper grounding. Refer to the Board Layout Checklist. sn1147 1147fds 12 LTC1147-3.3 LTC1147-5/LTC1147L U U W U APPLICATIO S I FOR ATIO Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC1147 series. These items are also illustrated graphically in the layout diagram of Figure 7. Check the following in your layout: 1. Are the signal and power grounds segregated? The LTC1147 ground (Pin 7) must return separately to a) the power and b) signal grounds. The power ground (a) returns to the source anode of the Schottky diode and (–) plate of CIN, which should have lead lengths as short as possible. The signal ground (b) connects to the (–) plate of COUT. 3. Are the SENSE – and SENSE + leads routed together with minimum PC trace spacing? The 1000pF capacitor between Pins 4 and 5 should be as close as possible to the LTC1147. 4. Does the (+) plate of CIN connect to the source of the P-channel MOSFET as closely as possible? This capacitor provides the AC current to the P-channel MOSFET. 5. Is the input decoupling capacitor (0.1µF/1µF) connected closely between VIN (Pin 1) and ground (Pin 7)? This capacitor carries the MOSFET driver peak currents. 6. On fixed output versions, is the SHDN (Pin 6) actively pulled to ground during normal operation? The SHDN pin is high impedance and must not be allowed to float. 2. Does the LTC1147 SENSE – (Pin 4) connect to a point close to RSENSE and the (+) plate of COUT? + BOLD LINES INDICATE HIGH CURRENT PATHS P-CH L + VIN RSENSE D1 CIN + + COUT – VOUT – 1µF + LTC1147-3.3 LTC1147-5 (LTC1147L) 1 2 3 390pF 3300pF 4 VIN CT ITH SENSE – PDRIVE 8 GND 7 SHDN (VFB) 6 SENSE + 5 R1 R2 100pF OUTPUT DIVIDER REQUIRED WITH ADJUSTABLE VERSION ONLY 1k 1000pF SHUTDOWN LTC1147 • F07 Figure 7. LTC1147 Layout Diagram (See Board Layout Checklist) For additional High Efficiency application circuits see Application Note 54. sn1147 1147fds 13 LTC1147-3.3 LTC1147-5/LTC1147L U TYPICAL APPLICATIO S 3.3V Low Dropout High Efficiency Regulator VIN 3.5V TO 12V + CIN 47µF 16V Si9433DY D1 MBRS130LT3 0.1µF L* 10µH LTC1147L-3.3 1 2 3 CT 120pF CC 3300pF 4 RC 1k PDRIVE VIN CT GND ITH SHDN SENSE – SENSE + 8 7 6 SHUTDOWN 5 + RSENSE** 0.068Ω 0.01µF COUT 100µF 10V AVX VOUT 3.3V/1.25A *SUMIDA CDR74B-100LC **IRC LRC-LR2010-01-R068-F LTC1147 • F08 Precision Constant Current Source VIN 10V TO 14V + Si3455DV D1 MBRS130LT3 0.1µF 3 CC 3300pF 4 RC 1k 8 7 CT GND ITH SHDN (VFB) 6 SENSE + 5 SENSE – 0.01µF 100pF R1 1.2k RSENSE** 0.22Ω R2 6.8k + COUT 100µF 16V AVX VOUT *SUMIDA CDR74-221 **IRC LRC-LR2010-01-R068-F VIN = 12.6V 0.8 IOUT (A) 2 CT 300pF PDRIVE VIN 1.0 L* 220µH LTC1147L 1 IOUT vs VOUT For Figure 9 CIN 33µF 25V 0.6 0.4 0.2 0 0 2 6 4 VOUT (V) 8 10 LTC1147 F10 LTC1147 • F09 sn1147 1147fds 14 LTC1147-3.3 LTC1147-5/LTC1147L U TYPICAL APPLICATIO S 2.5V/2A Regulator VIN 3.5V TO 12V + Si9433DY CIN 15µF 25V × 2 D1 MBRD330 0.1µF L* 10µH LTC1147L 1 2 3 CT 120pF CC 3300pF 4 RC 1k PDRIVE VIN 8 7 CT GND ITH SHDN (VFB) 6 SENSE + 5 SENSE – R1 49.9k 1% I00pF RSENSE** 0.05Ω 0.01µF R2 49.9k 1% + COUT 220µF 10V × 2 AVX 2.5V/2A *COILTRONICS CTX10-4 **IRC LR2512-01-0R050-G U PACKAGE DESCRIPTIO LTC1147 • F11 Dimensions in inches (millimeters) unless otherwise noted. N8 Package 8-Lead PDIP (Narrow 0.300) (LTC DWG # 05-08-1510) 0.300 – 0.325 (7.620 – 8.255) 0.009 – 0.015 (0.229 – 0.381) ( +0.035 0.325 –0.015 +0.889 8.255 –0.381 ) 0.045 – 0.065 (1.143 – 1.651) 0.130 ± 0.005 (3.302 ± 0.127) 0.065 (1.651) TYP 0.100 ± 0.010 (2.540 ± 0.254) 0.400* (10.160) MAX 8 7 6 5 1 2 3 4 0.255 ± 0.015* (6.477 ± 0.381) 0.125 (3.175) 0.020 MIN (0.508) MIN 0.018 ± 0.003 (0.457 ± 0.076) N8 1197 *THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS. MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm) sn1147 1147fds Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 15 LTC1147-3.3 LTC1147-5/LTC1147L U TYPICAL APPLICATION 3.3V/2A Output High Efficiency Regulator VIN 4V TO 14V + Si4431DY 0.1µF 1 2 3 CT 220pF CC 3300pF 4 RC 1k PDRIVE VIN GND LTC1147-3.3 SHDN ITH CT SENSE + SENSE – L* D1 20µH MBRS130LT3 8 7 6 SHUTDOWN COUT 220µF 10V AVX 5 + 0.01µF RSENSE** 0.05Ω *COILTRONICS CTX20-4 **KRL SP-1/2-A1-OR050 LTC1147 • F12 U PACKAGE DESCRIPTIO CIN 22µF 25V × 2 VOUT 3.3V/2A Dimensions in inches (millimeters) unless otherwise noted. S8 Package 8-Lead Plastic Small Outline (Narrow 0.150) (LTC DWG # 05-08-1610) 0.189 – 0.197* (4.801 – 5.004) 0.010 – 0.020 × 45° (0.254 – 0.508) 0.008 – 0.010 (0.203 – 0.254) 0.053 – 0.069 (1.346 – 1.752) 8 0.004 – 0.010 (0.101 – 0.254) 7 6 5 0°– 8° TYP 0.016 – 0.050 0.406 – 1.270 0.014 – 0.019 (0.355 – 0.483) 0.050 (1.270) TYP 0.150 – 0.157** (3.810 – 3.988) 0.228 – 0.244 (5.791 – 6.197) *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE SO8 0996 1 2 3 4 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LTC1142 Dual High Efficiency Synchronous Step-Down Switching Regulator Dual Version of LTC1148 LTC1143 Dual High Efficiency Step-Down Switching Regulator Controller Dual Version of LTC1147 LTC1147 High Efficiency Step-Down Switching Regulator Controller Nonsynchronous, 8-Lead, VIN ≤ 16V LTC1148 High Efficiency Step-Down Switching Regulator Controller Synchronous, VIN ≤ 20V LTC1149 High Efficiency Step-Down Switching Regulator Synchronous, VIN ≤ 48V, for Standard Threshold FETs LTC1159 High Efficiency Step-Down Switching Regulator Synchronous, VIN ≤ 40V for Logic Level MOSFETS LTC1174 High Efficiency Step-Down and Inverting DC/DC Converter 0.5A Switch, VIN ≤ 18.5V, Comparator LTC1265 High Efficiency Step-Down DC/DC Converter 1.2A Switch, VIN ≤ 13V, Comparator LTC1267 Dual High Efficiency Synchronous Step-Down Switching Regulators Dual Version of LTC1159 LTC1435 High Efficiency Low Noise Synchronous Step-Down Switching Regulator 16-Pin Narrow SO/SSOP; Constant Frequency sn1147 1147fds 16 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com LT/TP 0698 REV D 2K • PRINTED IN USA LINEAR TECHNOLOGY CORPORATION 1993