STMICROELECTRONICS L5994A

L5994
L5994A
ADJUSTABLE TRIPLE OUTPUT
POWER SUPPLY CONTROLLER
FEATURE
■ DUAL PWM CONTROLLERS ADJUSTABLE
1.9V to 5.3V(Section1)
1.6V to 3.5V(Section2)
■ AUXILIARY DRIVER FOR LINEAR
REGULATOR
■ CURRENT MODE CONTROL USING A LOW
SENSE RESISTOR
■ DUAL SYNC RECTIFIERS DRIVERS
■ "ONE SHOT" FEATURE (L5994A only)
■ 96% EFFICIENCY ACHIEVABLE
■ 50µ[email protected] STAND BY CONSUMPTION
■ 4.75V TO 25V OPERATING SUPPLY VOLTAGE
■ EXCELLENT LOAD TRANSIENT RESPONSE
■ "PULSE SKIPPING" FUNCTION
■ OUTPUT UNDER VOLTAGE SHUTDOWN
■ ADAPTATIVE ANTI SHOOT-THROUGH
CONTROL
■ OVER/UNDER VOLTAGE DETECTION
■ POWER GOOD SIGNALS
■ SEPARATED DISABLE
■ THERMAL SHUTDOWN
TQFP32
ORDERING NUMBERS: L5994
L5994A
APPLICATIONS
■ NOTEBOOK AND SUB NOTEBOOK
COMPUTERS
■
1.8V AND 2.5V I/O SUPPLY
■
WORDPAD
■
INTERNET APPLIANCE
DESCRIPTION
The device provides a dual PWM controller and a
linear driver controller that can support the complete
power management in mobile equipment with high
efficiency.
Figure 1. System Block Diagram
2.5V
5V
to
25V
L5994
SYNC
POWER
SECTION
1.8V
12V LDO
5.1V LDO
POWER
MANAGEMENT
& SYSTEM
SUPERVISOR
µP
MEMORY
PERIPHERALS
3.39V REF
POWER GOOD
D98IN862
April 2002
1/26
L5994 - L5994A
DESCRIPTION (continued)
The device produces an adjustable regulated voltage in both sections and a linear regulated voltage with an
external bipolar such as for PCMCIA applications.
The auxiliary linear driver is able to source up to 1A for 12V bus and is also possible to use it for the regulation
of 2.5V from 5V bus.
Synchronous rectification and pulse skipping mode for the buck sections optimise the overall efficiency over a
wide load current range.
The two high performance PWM output sections are monitored for over voltage, under voltage and over current
conditions.
A POWER GOOD signal is provided for each section.
On detection of a fault, the relevant POWER GOOD signal is generated and a specific shutdown procedure
takes place to prevent physical damage and data corruption.
A disable function allows to manage the output power sections separately, optimising the quiescent consumption of the IC in stand-by conditions.
The internal architecture is a current mode that allows to have fast transient response without compromise the
efficiency due to the ultra low sense resistor.
Under voltage shutdown is forced in case of short circuit in one of the two sections.
The drivers are provided of an adaptative anti cross conduction system for high output current application
ABSOLUTE MAXIMUM RATINGS
Symbol
Parameters
Vin
Power Supply Voltage
Vi
Maximum pin voltage to pins 1,24,25,32
Value
Unit
0 to 27
V
-0.3 to (Vin + 0.3)
V
Value
Unit
60
°C/W
-40 to 140
°C
THERMAL DATA
Symbol
RTH j-amb
Tj
Parameters
Thermal Resistance Junction to Ambient
Operative Junction Temperature Range
H2GATE
H2SRC
R2GATE
PGND
PREG5
R1GATE
H1SRC
H1GATE
PIN CONNECTIONS (Top views)
32 31 30 29 28 27 26 25
H1STRAP
1
24
H2STRAP
VIN
2
23
PWROK2
SREG5
3
22
VFBLIN
V5SW
4
21
VDRLIN
V1SNS
5
20
V2SNS
I1SNS
6
19
I2SNS
COMP1
7
18
COMP2
SOFT1
8
17
SOFT2
RUN2
OSC
NOSKIP
SGND
VREF
RUN1
CRST
2/26
PWROK1
9 10 11 12 13 14 15 16
D98IN863B
L5994 - L5994A
BLOCK DIAGRAM
SOFT1
SOFT2
8
SLOPE
I1SNS
V1SNS
COMP1
6
+
ERROR SUMMING
+
5
I1SNS
+
H1GATE
H1SRC
R1GATE
PREG5
W5SW
VIN
32
Hside
LOGIC
AND
SHOOTTHROUGH
CONTROL
31
REG5
30
Lside
VREF
SGND
V1SNS
V2SNS
20
18
+
+
+
-
-
VREF
+
-
-
LOGIC
AND
SHOOTTHROUGH
CONTROL
PULSE SKIPPING
COMPARATOR
+
+
-
-
24
Hside
ZERO CROSSING
COMPARATOR
+
I2SNS
V2SNS
COMP2
-
OVER CURRENT
COMPARATOR
PULSE SKIPPING
COMPARATOR
19
-
I2SNS
ZERO CROSSING
COMPARATOR
SLOPE
+
25
26
PREG5
Lside
27
28
H2STRAP
H2GATE
H2SRC
R2GATE
PGND
29
LINEAR
REGULATOR
4
21
2
+
OVERVOLT
COMPARATOR
-
SREG5
+
ERROR SUMMING
OVER CURRENT
COMPARATOR
VREF
1
SOFT
∑
-
H1STRAP
SOFT
∑
-
7
17
12
VREF BUFFER
-
V5SW
+
SWITCH
COMPARATOR
3
UNDERVOLT
COMPARATOR
VREF
4.7V
+
-
OSCILLATOR &
SYNCHRONIZATION
VREF
13
16
RUN2
10
14
RUN1 PWROK1
11
NOSKIP
REG12
LDO
22
POWER MANAGEMENT
&
SYSTEM SUPERVISOR
9
VDRLIN
15
VFBLIN
OSC
23
D98IN864A
CRST PWROK2
ELECTRICAL CHARACTERISTICS (Vin = 12V; Tj = 25°C; Vosc = GND; unless otherwise specified.)
Symbol
Parameter
Test Condition
Min.
Typ.
Max.
Unit
25
V
1.1
1.4
mA
50
60
90
100
µA
µA
1.81
1.89
1.97
V
37
50
63
mV
DC CHARACTERISTIC
Vin
Input supply voltage
V1out,4V
I2
Operating quiescent current
R1GATE= R2GATE = OPEN
H1GATE = H2GATE = OPEN
RUN1= RUN2 = SREG5
(DRIVERS OFF)
I2
Stand by current
RUNQ=RUN2=GND
NOSKIP=GND
Vin=12V
Vin=20V
4.75
SECTION 1 PWM CONTROLLER
V1out
V1SNS feedback voltage
V in = 5V to 20V
NOSKIP = REG5
Vi1sns-Vv1sns = 0 to 40mA
V6-V5
Overcurrent Threshold current
VSOFT1 = 3.1V
V6-V5
Pulse skipping mode threshold
voltage
Vin > 6.8V
Vin < 5.8V
13
6.5
mV
mV
3/26
L5994 - L5994A
ELECTRICAL CHARACTERISTICS (continued)
Symbol
V5
Parameter
Test Condition
Min.
Typ.
Max.
Unit
Over voltage threshold ON
V1SNS
2.09
2.15
2.21
V
Under voltage threshold ON
V1SNS
1.66
1.71
1.76
V
1.566
1.64
1.714
V
37
50
63
mV
SECTION 2 PWM CONTROLLER
V2SNS feedback voltage
Vin = 5V to 20V
NOSKIP = REG5
Vi2sns-Vv2sns = 0 to 40mA
V19-V20
Overcurrent Threshold current
VSOFT2 = 3.1V
V19-V20
Pulse skipping mode threshold
voltage
Vin > 6.8V
V2out
V20
13
mV
Over voltage threshold ON
V2SNS
1.843
1.9
1.957
V
Under voltage threshold ON
V2SNS
1.451
1.496
1.541
V
255
170
300
200
345
230
KHz
2.6
V
PWM CONTROLLERS CHARACTERISTICS
fOSC
Switching frequency accuracy
V15
Voltage range for 300KHz
operation
Td
Dead time
HS rise
LS rise
Tov
Over voltage propagation time
V1SNS to PWROK or
V2SNS to PWROK
1.25
µs
Tuv
Under voltage propagation time
V1SNS to PWROK or
V2SNS to PWROK
1.5
µs
V5,V20
Output UVLO threshold latched
Respect output voltage
I8,I17
V8,V17
OSC = 2.5V
OSC = GND OR 5V
2.4
Soft start current
50
30
ns
65
70
75
%
3.2
4
4.8
µA
Soft start clamp voltage
3.1
V
HIGH AND LOW SIDE GATE DRIVER (BOTH SECTIONS)
I25,I27
I32,I30
Source output peak current
Cload = 3.3nF
1
A
Sink output peak current
C load = 3.3nF
1
A
RH
RdsON resistance
Driver out high
2.1
4
Ohm
RL
RdsON resistance
Driver out low
1.5
3
Ohm
Output high voltage
HSTRAP = PREG5
Isource = 10mA;
HSOURCE = GND
5.3
5.61
V
VOH
4/26
4.4
L5994 - L5994A
ELECTRICAL CHARACTERISTICS (continued)
Symbol
VOL
Parameter
Output low voltage
Test Condition
Min.
Typ.
Max.
Unit
HSTRAP = PREG5
Isource = 10mA;
HSOURCE = GND
0.5
V
30
mA
2.56
V
20
V
2
µA
LINEAR REGULATOR DRIVER SECTION
Imax
Driver current
VFBLIN < 2.5V
Vref
VFBLIN threshold
VDRLIN = 15V
VDRLIN
Input voltage range
IFBLIN
Bias Current
VCP
2.44
2.5
4.5
Input voltage clamp
Iclamp = 100µA
16
“One Shot” activation
threshold (L5994A only)
VDRLIN falling
12.88
V
13.7
“One Shot” Pulse (L5994A only)
14.52
V
1.5
µs
5.5
V
INTERNAL REGULATOR DRIVER SECTION
V29
VREG5 output voltage
V in = 7 to 25V
5.0
I29
Totale current capability
VPREG5 = 5.3V
Vin = 7V
25
70
Swith-over threshold voltage
Vin > 7V, V5SW fall
4.2
4.53
4.8
V
V12
Reference voltage
I load = 1mA
Vin = 5 to 20V
Iload = 1 to 5mA
2.45
2.425
2.5
2.5
2.55
2.575
V
I12
Source current at reference
voltage
V12 = Vref-0.3V
15
mA
V5SW
5.3
mA
POWER GOOD AND ENABLE FUNCTION
V16,V11
RUN1, RUN2, enable voltage
High level
V16,V11
RUN1, RUN2, disable voltage
Low level
Power good delay
CRST=100nF
115
Shutdown delay time before LS
activation (except for over voltage
fault)
CRST=100nF
115
CRST timing rate
V12 = Vref-0.3V
Power good high level
I POWEROK = 40µA
Power good low level
I POWEROK = 320µA
T10
T27,T30
2.4
V
0.8
V
147
180
ms
147
180
ms
1.47
ms/nF
4.1
V
0.4
V
SYNCHRONIZATION
V15
Synchronization pulse width
Fsw = 1MHz
Synchronization input voltage
(falling edge)
IPOWEROK = 320µA
400
ns
5
V
5/26
L5994 - L5994A
PIN DESCRIPTION
N°
Name
Function
1
H1STRAP
Section 1 bootstrap capacitor connection. A bootstrap capacitor must be connected between
this pin and pin H1SRC to supply the H1GATE driver.
2
Vin
3
SREG5
Internal logic supply. It must be connected to PREG5 pin through a R-C low-pass filter.
4
V5SW
Alternative supply voltage for the internal 5V regulator. If its voltage is greater than 4.5V, the
internal regulator is supplied via this pin. If left floating, the regulator is supplied through the
Vin pin.
5
V1SNS
This pin connects the (-) input of section 1 current sense comparator. It must be connected
downstream the external RSENSE resistor.
6
I1SNS
This pin connects the (+) input of section 1 current sense comparator. It must be connected
upstream the external RSENSE resistor.
7
COMP1
Feedback input for section 1. It must be connected directly or through a resistor divider to the
output of the section 1.
8
SOFT1
Soft start connection for section 1. The soft start time is programmed by an external capacitor
connected between this pin and SGND. Approximatively 0.7ms/nF.
9
CRST
Used for start-up and shut-down timing. A capacitor must be connected between this pin and
SGND defining a time of 1.47ms/nF.
10
PWROK1
11
RUN1
Control input to enable / disable the section 1. A high logic level (>2.4V) enables this section,
a low level (<0.8V) shuts it down.
12
Vref
Internal 2.5V high accuracy voltage generator. It can source 5mA to external load. Bypass to
SGND with a 4.7uF electrolytic capacitor to reduce noise.
13
SGND
Signal ground. Reference for internal logic circuitry. It must be routed separately from high
current returns.
14
NOSKIP
Pulse skipping mode control. A high logic level (>2.4V) disables pulse skipping at low load
current, a low level (<0.8V) enables it.
15
OSC
Oscillator frequency control: connect to 2.5V to select 300kHz operation, connect to ground or
to 5V for 200kHz operation.
A proper external signal can synchronise the oscillator
16
RUN2
Control input to enable / disable the section 2. A high logic level (>2.4V) enables this section,
a low level (<0.8V) shuts it down.
17
SOFT2
Soft start connection for section 2. The soft start time is programmed by an external capacitor
connected between this pin and SGND. Approximatively 0.7ms/nF.
18
COMP2
Feedback input for section 2. It must be connected directly or through a resistor divider to the
output of the section 2.
19
I2SNS
This pin connects the (+) input of section 2 current sense comparator. It must be connected
upstream the external RSENSE resistor.
20
V2SNS
This pin connects the (-) input of section 2 current sense comparator. It must be connected
downstream the external RSENSE resistor.
6/26
Device supply voltage.
Power Good open drain diagnostic signal. This output is in high inpedence when sect. 1 is
enabled and running properly after a delay defined by the CRST capacitor. When not used
may be left floating.
L5994 - L5994A
PIN DESCRIPTION (continued)
N°
Name
Function
21
VDRLIN
Linear regulator driver connection. It must be connected to the base pin of an external PNP
transistor and, through a resistor, to its emitter in order to supply the internal driver.
If no linear regulation is implemented, it may be left floating.
22
VFBLIN
Feedback input for the linear regulator. It must be connected directly, or through a resistor
divider, to the linear regulated output. If no linear regulation is implemented, it may be left
floating
23
PWROK2
Power Good open drain diagnostic signal. This output is in high inpedence when sect. 2 is
enabled and running properly after a delay defined by the CRST capacitor. When not used
may be left floating.
24
H2STRAP
Section 2 bootstrap capacitor connection. A bootstrap capacitor must be connected between
this pin and pin H2SRC to supply the H2GATE driver.
25
H2GATE
Gate driver for the section 2, high side NMOS
26
H2SRC
Section 2 High side NMOS source connection.
27
R2GATE
Gate driver for the section 2, low side NMOS (synchronuos rectifier)
28
PGND
Current return for power mosfet driver for both sections. Connect to the low side mosfet
sources pin. It must be routed separately from signal current returns
29
PREG5
5V internal regulator output. Used mainly to supply the bootstrap capacitors and the internal
circuitry connected to SREG5 via a low-pass filter.
30
R1GATE
Gate driver for the section 1, low side NMOS (synchronuos rectifier)
31
H1SRC
Section 1 High side NMOS source connection.
32
H1GATE
Gate driver for the section 1, high side NMOS
Detailed Functional Description
In the device block diagram six fundamental functional blocks can be identified:
·
1.9V to 5.1V step-down PWM switching regulator (section 1, pins 1, 4 to 8, 30 to 32);
·
1.66V to 3.3V step-down PWM switching regulator (section 2, pins 17 to 20, 24 to 27);
·
Linear regulator driver for an external PNP transistor (pins 21,22);
·
5V low drop-out linear regulator (pin 29);
·
2.5V reference voltage generator (pin 12);
·
Power Management section (pins 9 to 11, 14,16).
The chip is supplied through pin Vin (2), typically by a battery pack or the output of an AC-DC adapter, with a
voltage that can range from 5V to 25V. The return of the bias current of the device is the signal ground pin SGND
(13), which references the internal logic circuitry. The drivers of the external MOSFET's have their separate current return, namely the power ground pin PGND (28). Take care of keeping separate the routes of signal ground
and the power ground pin when laying out the PCB (see "Layout and grounding" section). The two PWM regulators share the internal oscillator, programmable or synchronizable through pin OSC (15).
PWM Regulators
Each PWM regulator includes control circuitry as well as gate-drive circuits for a step-down DC-DC converter
in buck topology using synchronous rectification and current mode control.
The two regulators are independent and almost identical. As one can see in the Block Diagram, they share only
7/26
L5994 - L5994A
the oscillator and the internal supply and differ for the pre-set output voltages.
Each converter can be turned on and off independently: RUN1 and RUN2 are control inputs which disable the
relevant section when a low logic level (below 0.8 V) is applied and enable its operation with a high logic level
(above 2.4 V). When both inputs are low the device is in stand-by condition and its current consumption is extremely reduced (less than 120mA over the entire input voltage range).
The device is able to regulate the desired output voltage in two different ways: classic PWM operation and Pulse
Skip operation (see the relevant sections).
Oscillator
The oscillator, which does not require any external timing component, controls the PWM switching frequency.
This can be either 200 or 300 kHz, depending on the logic state of the control pin OSC, or else can be synchronized by an external oscillator.
If the OSC pin is grounded or connected to pin PREG5 (5V) the oscillator works at 200kHz. By connecting the
OSC pin to a 2.5 V voltage, 300 kHz operation will be selected. Moreover, if pin OSC is fed with an external
signal like the one shown in fig. 2, the oscillator will be synchronized by its falling edges.
Considering the spread of the oscillator, synchronization can be guaranteed for frequencies above 230kHz.
Even though a maximum frequency value is in practice imposed by efficiency considerations it should be noticed
that increasing frequency too much arises problems (noise, subharmonic oscillation, etc.) without significant
benefits in terms of external component size reduction and better dynamic performance.
The oscillator imposes a time interval (300 ns min.), during which the high-side MOSFET is definitely OFF, to
recharge the bootstrap capacitor (see "MOSFET's Drivers" section). This, implies a limit on the maximum duty
cycle (88.5%@fSW=300kHz, 92.6%@fsw=200kHz, worst case) which, in turn, imposes a limit on the minimum
operating input voltage.
Figure 2. Synchronization signal and operation
OSC
300ns min.
5V
0V
t
H1GATE
t
H2GATE
t
PWM Operation
The control loop does not employ a traditional error amplifier in favour of an error summing comparator which
sums the reference voltage, the feedback signal, the voltage drop across an external sense resistor and a slope
compensation ramp (to avoid subharmonic oscillation with duty cycles greater than 50%) with the appropriate
signs.
With reference to the schematic of fig. 3, the output latch of both controllers is set by every pulse coming from
8/26
L5994 - L5994A
the oscillator. That turns off the low-side MOSFET (synchronous rectifier) and, when the low-side gate voltage
falls below 0.3V to prevent cross-conduction, turns on the high-side one, thus allowing energy to be drawn from
the input source and stored in the inductor.
The error summing, by comparing the above mentioned signals, determines the moment in which the output
latch is to be reset. The high-side MOSFET is then turned off and the synchronous rectifier is turned on when
the voltage on the high-side MOSFET source falls below 2V to prevent cross-conduction, thus making the inductor current recirculate. The high side mosfet is in any case turned off on the clock signal falling edge: this is
the reason why the duty cicle is limited in its maximum value.
The reached state is maintained until the next oscillator pulse.
The open-loop transfer function of such a kind of control system, under the assumption of an ideal slope compensation, is:
Ro
1 + s ⋅ ESR ⋅ C O
F ( s ) = A ⋅ -------------------------- ⋅ --------------------------------------------------------------------------------------2 ⋅ R s ens e ( 1 + s ⋅ R o ⋅ C o ) ⋅ ( 1 + s ⋅ R F ⋅ C F )
where A is the gain of the error summing comparator, which is 2 by design.
The system is inherently very fast since it tends to correct output voltage deviations nearly on a cycle-by-cycle
basis. Actually, in case of line or load changes, few switching cycles can be sufficient for the transient to expire.
The operation above illustrated is modified during particular or anomalous conditions. Leaving out other circumstances (described in "Protections" section) for the moment, consider when the load current is low enough or
during the first switching cycles at start-up: the inductor current may become discontinuous, so it is zero during
the last part of each cycle. In such a case, a "zero current comparator" detects the event and turns off the synchronous rectifier, avoiding inductor current reversal and reproducing the natural turn-off of a diode when reverse biased. This allows to increase the efficiency in ligth load. Both MOSFET's stay in off state until the next
oscillator pulse.
Figure 3. Control loop.
HSTRAP
Vin
`CLOCK
S
Q
L
Rsense
REG5
R
NQ
Co
Ro
Rf
+
ERROR
+
SUMMING +
-
ESR
SLOPE
COMPENSATION
VREF
Cf
Synchronous Rectification
Very high efficiency is achieved at high load current with the synchronous rectification technique, which is particularly advantageous because of the low output voltage. The low-side MOSFET, that is the synchronous rectifier, is selected with a very low on-resistance, so that the paralleled Schottky diode is not turned on, except for
the small time in which neither MOSFET is conducting. The effect is a considerable reduction of power loss during the recirculation period.
9/26
L5994 - L5994A
Although the Schottky might appear to be redundant, it is not in a system where a very high efficiency is required. In fact, its lower threshold prevents the lossy body-diode of the synchronous rectifier MOSFET from turning on during the above mentioned dead-time. Both conduction and reverse recovery losses are cut down and
efficiency can improve of 1-2% in some cases. Besides a small diode is sufficient since it conducts for a very
short time.
See the "Power Management" section to see how both synchronous rectifiers are used to ensure zero voltage
output in stand-by conditions or in case of overvoltage.
Pulse-skipping operation
To achieve high efficiency at light load current as well, under this condition the regulators change their operation
(unless this feature is disabled): they abandon PWM and enter the so-called pulse-skipping mode, in which a
single switching cycle takes place every many oscillator periods.
The "light load condition" is detected when the voltage across the external sense resistor (VRSENSE) does not
exceed the pulse skipping threshold (13mV typ.) while the high-side MOSFET is conducting. When the reset
signal of the output latch comes from the error summing comparator while VRSENSE is below this value, it is ignored and the actual reset is driven as soon as VRSENSE reaches the pulse skipping threshold. This gives some
extra energy that maintains the output voltage above its nominal value for a while. The oscillator pulses now set
the output latch only when the feedback signal indicates that the output voltage has fallen below its nominal
value. In this way, most of oscillator pulses are skipped and the resulting switching frequency is much lower, as
expressed by the following relationship:
2
R S ENSE
V OUT

f PS = K ⋅ ---------------------- ⋅ I OUT ⋅ V OUT ⋅  1 – ---------------
L
V IN 

where K = 3.2×103 and fPS is in Hz. As a result, the losses due to switching and to gate-drive, which mostly
account for power dissipation at low output power, are considerably reduced.
The section 1 can work with the input voltage very close to the output one (i.e. the output voltage is 5V), where
the current waveform may be so flat to prevent pulse-skipping from being activated. To avoid this, the pulseskipping threshold (of section 1 only) is roughly halved at low input voltages (V IN < 6.8V). Under this condition,
in the above formula the constant K becomes 12.8×103.
When in pulse-skipping, the output voltage is some ten mV higher than in PWM mode, just because of its mode
of operation. If this "load regulation" effect is undesirable for any reason, the pulse skipping feature can be disabled (see "Power Management" section) to the detriment of efficiency at light load.
MOSFET's Drivers
To get the gate-drive voltage for the high-side N-channel MOSFET a bootstrap technique is employed. A capacitor is alternately charged through a diode from the 5V PREG5 line when the high-side MOSFET is OFF and
then connected to its gate-source leads by the internal floating driver to turn the MOSFET on. The PREG5 line
is used to drive the synchronous rectifier as well, and therefore the use of low-threshold MOSFET's (the socalled "logic-level" devices) is highly recommended.
The drivers are of "dynamic" type, which means they do not give origin to current consumption when they are
in static conditions (ON or OFF), but only during transitions. This feature is aimed at minimizing the power consumption of the device even during stand-by when both low-side MOSFET's are ON.
Adaptative anti shoot-through protection is implemented to prevent cross-conduction: the low side mosfet turn
on is disbled until the HSRC pin is above 2V and, in the same way, the high side mosfet turn on is disabled until
the RGATE pin is above 0.3V. During the time in which both mosfets are in off state, the recirculation of the
current is insured by the schottky diode. The resulting dead time depends on the mosfets used and on the current flowing in the inductor; in this way many kinds of mosfets may be used and cross conduction is avoided.
10/26
L5994 - L5994A
Protections
Each converter is fully protected against fault conditions. A monitoring system checks for overvoltages of the
output, quickly disabling the interested converter in case such an event occurs. This condition is latched and to
allow the device to start again either the supply voltages have to be removed or the relative RUNx pin has to be
driven low.
Also the undervoltage conditions are detected: a light undervoltage (90% of the programmed value) only causes
the relative PWROKx to be driven low while an hard undervolatge (70% of the programmed value) causes interruption of the operation of both converters. This is a protection against short circuits.
PWROKx signals (at pin 10 and 23) reveals the anomaly of the relative section (output voltage not within the
±10% of the programmed voltage) with a low output level. If the chip overheats (above 135°C typ.) the device
stops operating as long as the temperature falls below a safe value (105°C typ.). The overtemperature condition
is signalled by a low level on both PWROKx as well.
A current limitation comparator prevents from excessive current in case of overload. It intervenes as the voltage
VRSENSE exceeds 50mV, turning off the high-side switch before the error summing does. By the way, this also
gives the designer the ability to program the maximum operating current by selecting an appropriate sense resistor. This pulse-by-pulse limitation gives a quasi-constant current characteristic.
Linear Driver
The Linear driver is capable of sinking up to 60mA from an external PNP transistor through the pin VDRLIN
considering the typical application circuit shown in fig. 4. The internal comparator is supplied by the same pin
VDRLIN which accepts voltages included in the range from 4.5V to 20V. If the application works with input voltages that allows the regulation, the supply for the regulator can be obtained directly from the input source (VIN).
If such is not the case and is not available an additional input voltage, the most convenient way to get the supply
is to use an auxiliary winding on one of the two sections inductor with a catch diode, DS, and a filter capacitor,
CS, as shown in fig. 5. This winding delivers energy to pin VDRLIN during the recirculation period of each switching cycle with a voltage determined by the turns ratio n and little dependent on the input voltage.
Figure 4. Linear regulator supply with auxiliary winding
Vcc
2.5V
VDRLIN
Vout
VFBLIN
L5994
In case the section with the auxiliary winding is working at full load and the linear regulator is lightly loaded, the
voltage at pin VDRLIN can exceed the expected value. In fact, DS and CS act as a peak-holding circuit and
VDRLIN is influenced by the voltage spikes at switching transients. An internal clamp limits the voltage on the
VDRLIN pin at a maximum value of 16V, but, in case of intervention, the chip power dissipation will rise.The
linear driver is always active as long as PREG5 and VREF are present on the chip (see the relevant section); it
11/26
L5994 - L5994A
works in order to obtain a voltage on the VFBLIN pin of about 2.5V. In this way, the minimum regulated voltage
is of 2.5V, obtained connecting directly the VFBLIN pin to the output, while the maximum is of about the supply
voltage minus the bipolar PNP VceSAT.
For a correct operation of the regulator, the voltage at pin VDRLIN must not be too low. The flyback connection
of the two windings ensures a well regulated voltage, provided if there is good magnetic coupling. The coupled
inductors configuration, however, is not able to sustain the auxiliary voltage if the main output is lightly loaded:
the secondary voltage drops and the system goes out of regulation.
The additional winding may be implemented with L5994 if the relative section is loaded enough.
To overcome this problem, in L5994A, when the VDRLIN voltage falls below a certain threshold (13.7V ±5%)
because of too light a load on the section 2, the relevant synchronous rectifier is turned on for 1.5 µs max. during
the interval in which the inductor current is zero ("one-shot" feature, see fig. 6). In this way, the inductor current
reverses and draws from the output capacitor energy which is forward transferred to the auxiliary output.
Since that the linear driver is supplied from the VDRLIN pin, if the linear regulator is not necessary for the application,
leave floating this pin implies that the linear driver is not supplied and so no power is wasted (L5994 only).
The linear regulator is active, if at the least one of the two runx signal is asserted
Figure 5. "One Shot" pulse to substain VDRLIN voltage
H1GATE
t
1.5µs
L1GATE
t
IL
t
V13IN
13.7V
t
+5V Linear Regulator and +2.5V Reference Voltage Generator
The 5V low drop-out regulator powers directly the MOSFET drivers and it is externally available through pin
PREG5. A low pass filter is connected between PREG5 pin and SREG5 pin from who all the internal circuitry is
powered. The introduction of this R-C network is useful to minimize noise effects.
The typical external use of this generator is to charge the bootstrap capacitors used to produce the gate-drive
voltage for the high-side MOSFET's of both PWM converters.
At start-up and when the 5V section is not operating, this regulator is powered by the chip input voltage. To reduce power consumption, the linear regulator is turned off and the PREG5 pin is internally connected to the 5V
PWM regulator output via V5SW pin, when the 5V PWM regulator is active and its output voltage is above the
switchover threshold, 4.5V. This happens when V5SW pin is connected to the section 1 output regulating 5V.
In any case, if V5SW is above 4.5V, the internal regulator is turned off and PREG5 is powered through this pin.
12/26
L5994 - L5994A
The 5V regulator is always active, even if both PWM regulators are disabled, as long as power is applied to the
chip.
The 2.5V reference voltage generator, provides comparison levels for threshold detection and device operation.
It is allowed to source up to 5mA to an external load from its buffered output, externally available through pin
VREF.
The reference voltage generator is active if at least one of the two RUNx signal is asserted.
If either PREG5 or VREF does not deliver the correct voltage, the device is shut down.
Figure 6. Controlled timing sequencies
RUN2
RUN2
t
t
VOUT2
VOUT2
t
t
H2GATE
RUN1
t
t
R2GATE
VOUT1
t
CRST
t
t
CRST
PWROK2
t
t
PWROK1
RUN1
t
t
PWROK2
VOUT1
t
t
a) TURN-ON TIMING SEQUENCE
H1GATE
t
RUN2
R1GATE
t
VOUT2
t
PWROK1
t
t
b) SHUTDOWN TIMING SEQUENCE (1)
H2GATE
t
R2GATE
VOUT2
t
CRST
t
H2GATE
t
PWROK2
t
R2GATE
t
t
CRST
RUN1
t
VOUT1
PWROK
t
H1GATE
t
VOUT1
t
R1GATE
t
H1GATE
t
PWROK1
t
R1GATE
t
c) SHUTDOWN TIMING SEQUENCE (2)
t
d) OVP TIMING SEQUENCE
13/26
L5994 - L5994A
Power management
The device is provided with some control pins suitable to perform some functions which are commonly used or
sometimes required in battery-operated equipment. Besides, it features controlled timing sequences in case of
turn-on/off and device shutdown for a safe and reliable behaviour under all conditions.
As above mentioned, RUN1 and RUN2 pins allow to disable separately both PWM converters by means of logic
signals (likely coming from a µP) as mentioned earlier.
NOSKIP can disable the pulse-skipping feature: when it is held high neither of the PWM regulators is allowed
to enter this kind of operation.
The two PWROKx signals, one for each section, drive low immediately when the relative PWM regulator output
falls below its own undervoltage (light or hard) threshold or when it is disabled. It is high when the relative regulator runs properly.
A capacitor connected between CRST and ground fixes a time, in the order of 1.5msec/nF, which delays the
transition low-high of PWROKx. This happens at start-up of each section or after recovering an undervoltage
condition using the relative RUNx command. The delay starts from the moment in which the output voltage has
reached its correct value.
The same delay intervenes also in another circumstance: when a section is disabled (because its RUNx is driven low or owing to a thermal shutdown), the relevant synchronous rectifier is turned on after the above delay in
order to make sure that the load is no longer supplied.
This delay, however, does not intervene in case of overvoltage: the synchronous rectifier is immediately turned
on after the shutdown, thus acting as a built-in "crowbar".
All these timing sequences are illustrated in fig. 7.
Design procedure
Basically, the application circuit topology is fixed, and the design procedure concerns only the selection of the
component values suitable for the voltage and current requirements of the specific application.
The design data one needs to know are therefore:
■ Input voltage range: the minimum (VINMIN ) and the maximum (VINMAX ) voltage under which the
application is expected to operate;
■
Maximum load current for each of the two sections;
■
Iout1 for the section 1;
■
Iout2 for the section 2;
■
Output voltage and current for the linear regulator, setted for 12V 50mA using an additional winding on
the 5.1V section;
■
Maximum peak-to-peak ripple amplitude of the output voltage for each switching section:
■
Vrpp1 for the section 1;
■
Vrpp2 for the section 2;
■
The operating frequency fSW (200kHz / 300kHz or externally synchronized).
It is worth doing some preliminary considerations. The selection of the switching frequency depends on the requirements of the application. If the aim is to minimize the size of the external components, 300kHz will be chosen. For low input voltage applications 200kHz is preferred, since it leads to a higher maximum duty cycle.
As for the switching regulators, the inductance value of the output filter affects the inductor current ripple: the
higher the inductance the lower the ripple. This implies a lower current sense resistor value (for a given IOUT),
lower core losses and a lower output voltage ripple (for a given output capacitor) but, on the other hand, more
copper losses and a worse transient behaviour due to load changes. Usually the maximum ripple peak-to-peak
amplitude (which occurs at V INMAX) is chosen between 15% and 50% of the full load current. It is convenient to
introduce a ripple factor coefficient, RF, that is therefore a number between 0.15 and 0.5.
As for the linear regulator, its input voltage VDRLIN should not fall below 12V and therefore the auxiliary winding, if used, should be dimensioned to get this voltage with a certain margin (say, 13-14V). Conversely, an higher input voltage leads to higher losses inside the PNP transistor, to the detriment of efficiency, and to higher
total current on the +5V inductor. Besides it implies a higher turns ratio and therefore a worse magnetic coupling,
which affect energy transfer during flyback.
14/26
L5994 - L5994A
+3.3V Inductor
To define the inductor, it is necessary to determine firstly the inductance value. Its minimum value is given by:
3.3 – ( V INM AX – 3.3 )
L3M IN = --------------------------------------------------------------------V INMAX ⋅ f SW ⋅ I OUT3 ⋅ RF
and a value L3 > L3MIN should be selected. Core geometry selection is connected to the requirements of the
specific application in terms of space utilization and other practical issues like ease of mounting, availability and
so on. As to the material, the choice should be directed towards ferrite, molypermalloy or KoolMµ, to achieve
high efficiency. These materials provide low core losses (ferrite in particular), so that the design can be concentrated on preventing saturation and limiting copper losses. Saturation must be avoided even at maximum peak
current:
3.3 ⋅ ( V INM AX – 3.3 )
I L 3PK = I OUT3 + -----------------------------------------------------⋅L ⋅V
2⋅f
SW
3
INM AX
To limit copper losses, the winding DC resistance, RL, should be as low as possible (in the range of mΩ). AC
losses can usually be neglected. A practical criterion to minimize DC resistance could be to use the largest wire
that fits the selected core.
Anyway the best solution, whenever possible, is to use an off-the-shelf inductor which meets the requirements
in terms of inductance and maximum DC current. Nowadays there is a broad range of products offered by manufacturer, also for surface mount assemblies.
+5.1V Transformer
The primary winding carries the secondary power as well, thus the total primary average current is:
V INLIN ⋅ I OUT12
I TO T5 = I OUT 5 + ----------------------------------------5.1
where VDRLIN is the voltage generated during the recirculation of the primary and fed into the input of the +12V
linear regulator. The turns ratio 1:n of the transformer is chosen so that VDRLIN is above 13V. To reduce the
turns ratio in order to minimize stray parameters, the secondary is referred to the 5.1V output, and therefore the
minimum value is given by:
V INLIN – 5.1 + V f
η MIN = -------------------------------------------5.1
where Vf is the forward drop across the rectifier (assume 1V to be conservative). Make sure the secondary is
connected with the proper polarity (see fig. 4). The minimum primary inductance value can be expressed as:
2
5.1 ⋅ ( V IN – 5.1 )
3
L 5pm in = --- ⋅ --------------------------------------------------------------------------------------------------------------------------------------------4 V IN ⋅ f SW ⋅ [ I T O T5 ⋅ RF ⋅ ( V IN – 5.1 ) – η ⋅ V IN ⋅ I OUT12 ]
where RF, to get positive values for L5PMIN, must satisfy the inequality:
η ⋅ V IN ⋅ I OUT12
RF > -----------------------------------------------I T OT 5 ⋅ ( V IN – 5.1 )
and where VIN can be either VINMIN or VINMAX, whichever gives the higher value for L5PMIN. With a primary inductance L5P > L5PMIN the primary peak current, which must not saturate the magnetic core, will be:
5.1 ⋅ ( V INMA X – 5.1 )
I 5 PK = I 5 TO T + ---------------------------------------------------------- + η ⋅ I OUT12
⋅L ⋅V
2⋅f
SW
5P
INM AX
As to the transformer realization, the considerations regarding to the +3.3V inductor can be here repeated.
15/26
L5994 - L5994A
Power MOSFET's and Schottky Diodes
Since the gate drivers of the device are powered by a 5V bus , the use of logic-level MOSFET's is highly recommended, especially for high current applications. Their breakdown voltage V (BR)DSS must be greater than
VINMAX with a certain margin, so the selection will address 20V or 30V devices.
The RDS(ON) can be selected once the allowable power dissipation has been established. By selecting identical
power MOSFET's as the main switch and the synchronous rectifier, the total power they dissipate does not depend on the duty cycle. Thus, if PON is this power loss (few percent of the rated output power), the required
RDS(ON) (@ 25 °C) can be derived from:
P ON
R DS ( ON ) = ---------------------------------------------2
I OUT ⋅ ( 1 + α ⋅ ∆T )
where Iout is either ITOT5 or IOUT3, according to the section under consideration, a is the temperature coefficient
of RDS(ON) (typically, a = 5 · 10-3 °C-1 for these low-voltage classes) and ∆T the admitted temperature rise. It is
worth noticing, however, that generally the lower RDS(ON), the higher is the gate charge Qg, which leads to a
higher gate drive consumption. In fact, each switching cycle, a charge Qg moves from the input source to
ground, resulting in an equivalent drive current:
Ig = Qg · fSW
which affects efficiency at low load currents. Besides, this current is drawn from the PREG5 line whose source
capability, ISRC (25mA min), must not be exceeded, thus a further constraint concernes the MOSFET total gate
charge (@VGS = 5V):
I SRC
Q g ≤ ----------------4 ⋅ fSW
assuming four identical MOSFET's.
The Schottky diode to be placed in parallel to the synchronous rectifier must have a reverse voltage VRRM greater than VINMAX. Since it conducts for less than 5% of the switching period, the current rating can be much lower
than Iout. The selection criterion should be:
Vt(Schottky) < Vt(body-diode)@I = ILPK
Sense Resistors
The sense resistor of each section is selected according to their respective maximum output current. The current sense comparator limits the inductor peak current and therefore the maximum DC output current is the peak
value less half of the peak-to-peak ripple. The intervention threshold is set at 50mV for both sections, thus the
resistor values should be:
50
R SENSE5 = -------------- [ mΩ ]
I L5 PK
50
R SENSE 5 – -------------- [ mΩ ]
I L5 PK
Since the comparator threshold that triggers pulse-skipping mode is 11mV, the output current at which the system enters this kind of operation is approximately one fourth of the maximum output current. The sense resistors
values are in the low milliohms thus it is important to take correctly the current sense signals. Make sure that
the Kelvin connections between the current sense pins of the IC and the sense resistor do not carry the output
current.
16/26
L5994 - L5994A
Input Capacitors
A pulsed current (with zero average value) flows through the input capacitor of a buck converter. The AC component of this current is quite high and dissipates a considerable amount of power on the ESR of the capacitor:
V OUT ⋅ ( V IN – V OUT )
2
Pc IN = ESR ⋅ I OUT ⋅ -----------------------------------------------------2
V IN
It is easy to find that PCIN has a maximum equal to IOUT/2 (@ VIN=2×VOUT, that is, 50% duty cycle). The
input capacitor of each section, therefore, should be selected for a RMS ripple current rating as high as half the
respective maximum output current. The capacitance value is not very important but in reality a minimum value
must be ensured for stability reasons. In fact, switching regulators exhibit a negative input impedance that, at
low frequencies, is:
2
V IN
Z IN ( DC ) = -------------------------------V OUT ⋅ I OUT
thus, if the impedance of the power source is not well below the absolute value of ZIN(DC) at frequencies up to
the bandwidth of the regulator control loop, there is the possibility for oscillations. To ensure stability, the following condition must be satisfied:
LEQ
C IN » ----------------------------------------------ESR IN ⋅ Z IN ( DC )
where L EQ is the inductance of the circuit upstream the switching regulator input and ESRIN is related to the
input capacitor itself.The use of high performance electrolytic capacitors is recommended. If a higher cost is of
no concern, OS-CON capacitors are an excellent choice because they offer the smallest size for a given ESR
or current rating. Tantalum capacitors do not tolerate pulsed current, so their use is not advisable.
Output Capacitors
The output capacitor selection is based on the output voltage ripple requirements. This ripple is related to the
current ripple through the inductor and is almost entirely due to the ESR of the output capacitor. Therefore, the
goal is to achieve an ESR lower than a certain value, regardless of the actual capacitance value.
The maximum current ripple of the +3.3V section is:
∆IL3 = 2 · (IL3PK - IOUT3)
considering the values obtained in the paragraph "+3.3V inductor". As for the +5.1V, the maximum ripple is given by:
V IN
3 5.1 ⋅ ( V IN – 5.1 )
∆I L5 = η ⋅ I OUT 12 ⋅ ----------------------- + --- ⋅ ----------------------------------------V IN – 5.1 4 f SW ⋅ L3 P ⋅ V IN
where VIN is VINMIN or VINMAX, as selected in the "+5.1 transformer" section.
Anyhow, the maximum ESR will be:
V RPKX
ESR X ≤ -----------------∆I LX
where the subscript x refers to either section.
In pulse-skipping operation, the capacitive component of the output ripple is comparable to the resistive one,
17/26
L5994 - L5994A
thus both should be considered:
ESR X
(R )
V RPP X = 0.011 ⋅ -------------------------R SENS EX
(R)
V RPP X = 3.1 ⋅ 10
–6
LX
1
1
1
- ⋅  ---------------------------------------- – ---------------
⋅ ------------------- ⋅ ------------------------V

C OUTX R 2
–
V
V
INM IN
OUT
OUT
SENS EX
If specification on the output ripple under pulse-skipping condition is also given, COUTX and ESRX must comply
with it as well.
Further constraints on the minimum output capacitance can arise from specifications regarding the maximum
undershoot, ∆V-OUT, or overshoot, ∆V+OUT, due to a step-load change ∆IOUT:
2
L ⋅ ∆I OUT
C OUT > ---------------------------------------------------------------------------------------∆V OUT ⋅ ( V INM IN ⋅ D M AX – V OUT )
2
L ⋅ ∆I OUT
C OUT > ------------------------------------+
∆V OUT ⋅ V OUT
whichever is greater, and where Dmax is the maximum duty cycle and the quantities are relevant either to the
+3.3V or +5.1V section. High performance capacitors should be employed to reduce the capacitance needed
for a given ESR, to avoid paralleling several parts with a considerable waste of space. Although excellent electrolytic capacitors are available, OS-CON or tantalums may be preferred especially if very compact design is
required, or in case of surface mount assemblies. Multilayer ceramic capacitors with extremely low ESR are now
available, but they have a large spread of the capacitance value, so they should be paralleled with another more
stable, high-ESR capacitor.
Miscellaneous Components
The feedback loop has virtually unlimited bandwidth, thus a filter is necessary to make the system insensitive
to the switching frequency ripple and, in general, to prevent noise from disturbing the correct operation of the
error summing comparator. Anyway, the cut-off frequency of this filter can be very high, so that line and load
transient response is extremely fast. This filter is a simple R-C type where resistance and capacitance can be
chosen for a typical 3dB cut-off frequency of 60kHz.
As to the bootstrap diodes, even though small signal diodes might be effectively used, it is preferable to employ
low-power Schottky rectifiers, since that increase slightly the gate drive voltage, in favour of efficiency. The bootstrap capacitor can be a 100nF film capacitor.
The soft-start capacitors determine the time during which the current limitation circuit moves gradually the setpoint from zero up to 50mV in order to limit the current inflow at start-up. This ramp lasts approximately 1 ms
per nF of soft-start capacitance (10 to 100 nF typical values), but the actual time necessary to the output voltage
to reach the steady-state value depends on the load current and the output filter capacitance.
There are some critical points of the IC that may require by-pass capacitors to prevent noise from disturbing the
circuit. These points are the reference voltage VREF, the IC supply pin VIN, the PREG5 line and the alternative
supply pin V5SW. Use film capacitors suitable for AC decoupling.
External PNP bipolar transistor
As the output of the auxiliary winding on the 5.1V section is dimensioned for 13V, considering an output voltage
of 12V the power loss across the external PNP transistor is of:
PLOSS = (VIN - VOUT) · IOUT
The collector-emitter breakdown voltage must be greater than the one delivered by the transformer on the 5.1V
section and this is true also for the collector-base junction. A small signal trnsistor is enough for the considered
application.
18/26
L5994 - L5994A
Transformer Catch Diode (L5994A only)
The diode which steers the current generated by the secondary winding of the +5.1V transformer should be a
fast-recovery one, with a breakdown voltage greater than:
V RR = ( V INLIN – 5.1 ) + η ⋅ ( V INM AX – 5.1 )
with a certain safety margin. The diode has to withstand a pulsed current whose peak value is approximately:
V INMIN
I 13PK – I OUT12 ⋅ --------------------------------V INM IN – 5.1
while its RMS value is given by:
V INM IN
I 13RM S = I OUT12 ⋅ --------------------------------V INM IN – 5.1
The DC value is obviously IOUT12.
Transformer Filter Capacitors
The most stringent requirement on the input filter capacitor (connected between V13IN and ground) is its RMS
ripple current rating, which should be at least:
5.1
I 13AC = I OUT12 ⋅ --------------------------------V INM IN – 5.1
The working voltage should be higher than the voltage generated when the regulator is lightly loaded. Also for
this part the use of high quality electrolytic or OS-CON capacitors is advised.
Layout and Grounding
The electrical design is only the first step in the development of a switching converter. Since currents ranging
from mA to some A, both DC and switched, live together on the same circuitboard, the PCB layout is vital for a
correct operation of the circuit but is not an easy task.
A proper layout process generally includes careful component placing, proper grounding, correct traces routing,
and appropriate trace widths. Fortunately, since low voltages are involved in this kind of applications, isolation
requirements are of no concern.
Referring to literature for a detailed analysis of this matter, only few important points will be here reminded.
1) All current returns (signal ground, power ground, etc.) should be mutually isolated and should be connected only at a single ground point. Ground planes may be extremely useful both to arrange properly
current returns and to minimize radiation (see next 2 points), even though they cannot solve every problem
2) Noise coupling between adjacent circuitry can be reduced minimizing the area of the loop where current
flows. This is particularly important where there are high pulsed currents, that is the circuit including the
input filter capacitor, the power switch, the synchronous rectifier and the output capacitor. The next priority should be given to the gate drive circuits.
3) Magnetic field radiation (and stray inductance) can be reduced by keeping all traces which carry
switched currents as short as possible.
4) The Kelvin-connected traces of current sense should be kept short and close together.
5) For high current paths, the traces could be doubled on the other side of the PCB whenever possible: this
will reduce both the resistance and the inductance of the wiring.
6) In general, traces carrying signal currents should run far from traces carrying pulsed currents or with
quickly swinging voltages. From this viewpoint, particular care should be taken of the high impedance
paths (feedback input, current sense traces...). It could be a good idea to route signal traces on one PCB
19/26
L5994 - L5994A
side and power traces on the other side.
7) Use heavy copper traces: this will reduce their resistance, increasing overall efficiency and will improve
their heat-sinking ability.
L5994 Evaluation Kit
The L5994 Evaluation kit is a fully assembled and tested demonstration board that implements a standard application circuit, configured according to the following specifications:
Input Voltage Range: 5V to 25V;
3.3V Output: Iout3 = 3A;
5.1V Output: Iout5 = 3A;
12V Output: Iout12 = 50mA;
Switching frequency: fSW = 300kHz.
Figure 7. L5994 Evaluation kit
VIN
R16
R33
C6
C2
C1
VIN
C7
GND
C8
SREG5
D3
Vin2
C28
C3
C4
D4
2
H1STRAP
3
1
24
32
25
H2STRAP
R17
Q3
Q1
H2GATE
H1GATE
R21
R23
D6
C26
C17
C16
H2SRC
H1SRC
VOUT1
R1GATE
Q2
R2GATE
R22
L5994
PGND
29
28
6
19
I1SNS
V1SNS
C18
I2SNS
R7
VDRLIN
5
21
4
20
7
18
V5SW
R28
V2SNS
COMP1
R6
GND
C31
G2
+5VIN
R20
D2
R24
G1
R34
C13
C14
C15
27
30
D1
C30
R5
R3
R4
Q4
PREG5
GND
VOUT2
L1
C9
C10
C11
R19
L2
26
31
R1
R2
Vin2
COMP2
C19
C20
R8
GND
R15
R13
R12
R11
R10
R9
8
10
OSC
CRST
9
16
VREF
12
14
RUN1
11
S1
S3
S4
C27
17
15
RUN2
S2
R26
SOFT2
NOSKIP
NOSKIP
R25
SOFT1
PWROK1
PWROK1
LIN
22
23
R14
VOUT3
VFBLIN
PWROK2
PWROK2
C21
C23
C22
C24
13
SGND
D98IN866B
20/26
L5994 - L5994A
The electrical schematic, illustrated in fig. 7, shows that some pull-up/down resistor are added to the components strictly needed in a real application. Along with a quad dip-switch, they allow to set manually the logic signals that control the chip operation. These signals are in the present case:
- Switch 1: RUN1 (0= 5.1V OFF, 1= 5.1V ON)
- Switch 2: OSC (0= 200kHz, 1=300kHz)
- Switch 3: NOSKIP (0= pulse-skipping ON, 1= pulse-skipping OFF)
- Switch 4: RUN2 (0= 3.3V OFF, 1= 3.3V ON)
Please note that as long as each regulator is disabled, the relevant low-side MOSFET is in ON state. Hence, if
the load is capable of sourcing current, it will be short-circuited to ground through the choke and the low-side
MOS.
Although the default switching frequency is 300kHz (switch 2 set on 1) and the passive components have been
selected for this frequency, the demo board will work satisfactorily at 200kHz as well. Actually, at 200kHz the
regulators exhibit the maximum efficiency and the maximum extension of the input voltage range downwards.
On the other hand, the output ripple is greater and the dynamic behaviour slightly worse.
The demonstration board, as it is, does not provide an interface for synchronization. Anyway, it is possible to
synchronize the oscillator (with an appropriate signal: 5V amplitude pulses, spaced out by 400ns min.), provided
the switch is set on 1, simply by feeding the signal into the middle of the divider R8-R9. In this way, synchronization can be achieved at a frequency higher than 300kHz. To synchronize the oscillator to a frequency between
200kHz and 300kHz, heavier interventions on the board are needed.
Pulse-skipping operation is enabled by default in order to maximize efficiency also in low load current range.
The transition between PWM and pulse-skipping occurs approximately below 1A, however there is a region in
which the two operation modes coexist rather than a definite boundary. That can be seen on the scope as an
irregularity of the waveforms but does not have much influence both on output ripple and efficiency.
Those who do not appreciate asynchronous operation of the pulse-skipping mode can disable it for both regulators, by setting switch 3 on 1. That maintains PWM operation up to very low output currents where, however,
the regulation becomes incompatible with the switching frequency. This means that the minimum ON-time of
the high-side MOSFET is too long for the thruput energy level at the operating frequency. Thus the control system begins skipping conduction cycles to avoid the output voltage drifting upwards.
21/26
L5994 - L5994A
Component list
Table shows the complete L5994/ L5994A Evaluation Kit parts list. Critical components characteristics are given in detail.
Resistors
R1,R2,R3,R4
0.02
1W - 1% DALE WSL-2512
SMD - 2512
R5
R6
5.6k
3.3k
0.1W - 1%
0.1W - 1%
SMD - 0603
SMD - 0603
R7,R8
3.9k
0.1W - 1%
SMD - 0603
R9,R15
47k
0.1W
SMD - 0603
R10,R11,R12,R13,R14
1M
0.1W
SMD - 0603
R16
R17,R18,R19,R20
4.7
N.C.
1/4W
0.1W
SMD - 1206
SMD - 0603
R21,R22,R23,R24
2
1/4W
SMD - 0603
R25
8.2K
1/4W
SMD - 0603
R26
R27
2.2K
0
1/4W
1/4W
SMD – 0603
SMD – 0603
R28
1K
1/4W
SMD - 0603
4.7
1/4W
SMD - 1206
47µ
SANYO - OS-CON 25V 25SC47M
Radial 10 - 5
R33,R34
Capacitors C1,C4
C2,C6
1µ
Ceramic 25V
SMD - 1206
C3,C7,C8,C20
1µ
N.C.
Ceramic
SMD - 1206
TANTD
330µ
100n
22n
KEMET 10V - T510
Ceramic
Ceramic
TANTD
SMD - 0603
SMD - 0603
4.7µ
100n
Tantalium 16V
SMD - 3528
C22,C24
Ceramic
SMD - 0603
C25
N.C.
Radial 8 - 2.5
SMD - 0603
SMD - 0603
C9,C15
C10,C11,C13,C14
C16,C17,C23
C18,C19
C21,C27
Magnetics
15µ
SANYO - OS-CON 25V 25SC15M
C28,C29
C30,C31
T1
N.C.
47n
Ceramic
Ceramic
10µ
Transpower Technologies
TTI5870
(only for L5994A)
L1
10µ
SUMIDA CDR125-100 (only for
L5994)
10µ
SI4410
SI4410
BC807
SUMIDA CDR125-100
L2
Transistors Q1,Q2
Q3,Q4
Q5
Diodes
IC
Jumper
22/26
Radial 8 - 2.5
C26 (L5994A only)
D1,D2
D3,D4
D5
D6 (L5994A only)
U1
G1
G2,G3,G4
STPS340S
BAR18
N.C.
STPR120A
L5994/
L5994A
SHORTED
OPEN
SO8
SO8
SOT23
SMC
SOT23
SMA
SMA
TQFP32
L5994 - L5994A
Figure 8. PCB and component layout
23/26
L5994 - L5994A
Figure 9. Demo Board Efficiency vs Output
Current
EFF.
(%)
D96IN420
Figure 12. Demo Board Efficiency vs Output
Current
EFF.
(%)
D96IN423A
Vin=6V
Vin=6V
90
90
Vin=20V
80
80
Vin=20V
Vin=15V
70
70
VO=5.1V
fSW=200KHz
RUN3=GND
NOSKIP=GND
60
VO=3.3V
fSW=300KHz
RUN5=GND
NOSKIP=GND
Vin=15V
60
50
50
0.001
0.005 0.01
0.05 0.1
0.5 1
5 IO(A)
Figure 10. Demo Board Efficiency vs Output
Current
EFF.
(%)
D96IN421
Vin=6V
0.001
0.005 0.01
0.05 0.1
0.5 1
5 IO(A)
Figure 13. Demo Board Overall Efficiency (Iout3
= 3A, REG12 = OPEN, OSC = GND)
EFF.
(%)
D97IN581
93
90
Vin=6V
92
Vin=20V
80
91
Vin=15V
90
70
VO=5.1V
fSW=300KHz
RUN3=GND
NOSKIP=GND
60
89
Vin=20V
88
50
0.001
0.005 0.01
0.05 0.1
0.5 1
5 IO(A)
Figure 11. Demo Board Efficiency vs Output
Current
EFF.
(%)
D96IN422A
87
0.001
0.005 0.01
0.05 0.1
0.5 1
5 IOUT5(A)
Figure 14. Demo Board Overall Efficiency (Iout5 =
3A, REG12 = OPEN, OSC = GND)
EFF.
(%)
Vin=6V
D97IN582
Vin=6V
90
93
Vin=20V
80
92
70
91
VO=3.3V
fSW=200KHz
RUN5=GND
NOSKIP=GND
Vin=15V
60
Vin=20V
90
50
0.001
24/26
0.005 0.01
0.05 0.1
0.5 1
5 IO(A)
89
0.001
0.005 0.01
0.05 0.1
0.5 1
5IOUT3(A)
L5994 - L5994A
mm
DIM.
MIN.
inch
TYP.
MAX.
A
MIN.
TYP.
1.60
A1
0.05
A2
1.35
B
0.30
C
0.09
0.063
0.15
0.002
0.006
1.40
1.45
0.053
0.055
0.057
0.37
0.45
0.012
0.015
0.018
0.20
0.004
0.008
D
9.00
0.354
D1
7.00
0.276
D3
5.60
0.220
e
0.80
0.031
E
9.00
0.354
E1
7.00
0.276
E3
5.60
0.220
L
0.45
0.60
L1
0.75
OUTLINE AND
MECHANICAL DATA
MAX.
0.018
0.024
1.00
0.030
0.039
TQFP32
0°(min.), 7°(max.)
K
D
A
D1
A2
D3
24
A1
17
25
16
0.10mm
.004
B
E
E1
B
E3
Seating Plane
9
32
8
1
C
L
L1
e
K
TQFP32
25/26
L5994 - L5994A
Information furnished is believed to be accurate and reliable. However, STMicroelectronics assumes no responsibility for the consequences
of use of such information nor for any infringement of patents or other rights of third parties which may result from its use. No license is granted
by implication or otherwise under any patent or patent rights of STMicroelectronics. Specifications mentioned in this publication are subject
to change without notice. This publication supersedes and replaces all information previously supplied. STMicroelectronics products are not
authorized for use as critical components in life support devices or systems without express written approval of STMicroelectronics.
The ST logo is a registered trademark of STMicroelectronics
 2002 STMicroelectronics - All Rights Reserved
STMicroelectronics GROUP OF COMPANIES
Australia - Brazil - Canada - China - Finland - France - Germany - Hong Kong - India - Israel - Italy - Japan -Malaysia - Malta - Morocco Singapore - Spain - Sweden - Switzerland - United Kingdom - United States.
http://www.st.com
26/26