ETC CS5422/D

CS5422
Dual Out-of-Phase
Synchronous
Buck Controller
with Current Limit
The CS5422 is a dual N–channel synchronous buck regulator
controller. It contains all the circuitry required for two independent
buck regulators and utilizes the V2 control method to achieve the
fastest possible transient response and best overall regulation, while
using the least number of external components. The CS5422 features
out–of–phase synchronization between the channels, reducing the
input filter requirement. The CS5422 also provides undervoltage
lockout, Soft Start, built in adaptive FET non–overlap and hiccup
mode overcurrent protection. The part is available in a 16 Lead SO
Narrow or 24 Lead Fused SO Wide package allowing the designer to
minimize solution size.
SO–16
D SUFFIX
CASE 751B
16
1
SO–24L
DWF SUFFIX
CASE 751E
24
1
PIN CONNECTIONS AND
MARKING DIAGRAMS
1
GATE(H)1
GATE(L)1
GND
BST
IS+1
IS–1
VFB1
COMP1
16
GATE(H)2
GATE(L)2
VCC
ROSC
IS+2
IS–2
VFB2
COMP2
1 SO–24L 24
A
WL, L
YY, Y
WW, W
CS5422
AWLYYWW
GATE(H)1
GATE(L)1
PGND
BST
LGND
LGND
LGND
LGND
IS+1
IS–1
VFB1
COMP1
SO–16
CS5422
AWLYWW
Features
• V2 Control Topology
• Hiccup Mode Overcurrent Protection
• 150 ns Transient Response
• Programmable Soft Start
• 100% Duty Cycle for Enhanced Transient Response
• 150 kHz to 600 kHz Programmable Frequency Operation
• Switching Frequency Set by Single Resistor
• Out–Of–Phase Synchronization Between the Channels Reduces the
Input Filter Requirement
• Undervoltage Lockout
• Internally Fused Leads available in 24 Lead SO Wide Package
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GATE(H)2
GATE(L)2
VCC
ROSC
LGND
LGND
LGND
LGND
IS+2
IS–2
VFB2
COMP2
= Assembly Location
= Wafer Lot
= Year
= Work Week
ORDERING INFORMATION
Device
 Semiconductor Components Industries, LLC, 2001
May, 2001 – Rev. 5
1
Package
Shipping
CS5422GD16
SO–16
48 Units/Rail
CS5422GDR16
SO–16
2500 Tape & Reel
CS5422GDWF24
SO–24L
31 Units/Rail
CS5422GDWFR24
SO–24L
1000 Tape & Reel
Publication Order Number:
CS5422/D
CS5422
12 V
R10
Q5
2N3904
220
1.5 V/10 A
+
C11–C13
3 × 680 µF/4.0 V
L2
1.3 µH
Q4
MTD3302
R3
2.0 k
R4
C7
0.1 µF
2.0 k
C15
0.1 µF
R7
5.0 k ± 1.0%
C17
100 pF
R8
10 k ± 1.0%
D1
C4
0.2 µF
18 V
MA3180
C3
1.0 µF
Q3
MTD3302
MBR0530T MBR0530T1
D2
C5
Q1
MTD3302
4
VCC
BST
1
16
GATE(H)2 GATE(H)1
GATE(L)2 GATE(L)1
12 IS+2
9
Q2
MTD3302
COMP1
2.0 k
0.1 µF R2
2.0 k
8
10
C14
0.1 µF
GND
3
R5
VFB1 7
R9
30.9 k
8.0 k ± 1.0%
R6
10 k±1.0%
C16
100 PF
Figure 1. Application Diagram, 12 V to 1.5 V/10 A and 1.8 V/10 A Converter
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2
+
C8–C10
3 × 680 µF/4.0 V
C6
IS–1 6
COMP2
1.8 V/10 A
1.3 µH
R1
IS+1 5
IS–2
VFB2
13 R
OSC
L1
2
CS5422
11
C1–C2
2 × 220 µF
0.2 µF
14
15
+
CS5422
ABSOLUTE MAXIMUM RATINGS*
Rating
Value
Unit
150
°C
–65 to +150
°C
ESD Susceptibility (Human Body Model)
2.0
kV
Package Thermal Resistance, SO–16:
Junction–to–Case, RθJC
Junction–to–Ambient, RθJA
28
115
°C/W
°C/W
Package Thermal Resistance, SO–24L:
Junction–to–Case, RθJC
Junction–to–Ambient, RθJA
9.0
55
°C/W
°C/W
230 peak
°C
Operating Junction Temperature, TJ
Storage Temperature Range, TS
Lead Temperature Soldering:
Reflow: (SMD styles only) (Note 1.)
1. 60 second maximum above 183°C.
*The maximum package power dissipation must be observed.
ABSOLUTE MAXIMUM RATINGS
Pin Symbol
Pin Name
VMAX
VMIN
ISOURCE
ISINK
VCC
IC Power Input
16 V
–0.3 V
N/A
1.5 A peak
200 mA DC
COMP1, COMP2
Compensation Capacitor for
Channel 1 or 2
4.0 V
–0.3 V
1.0 mA
1.0 mA
VFB1, VFB2
Voltage Feedback Input for
Channel 1 or 2
5.0 V
–0.3 V
1.0 mA
1.0 mA
BST
Power Input for GATE(H)1, 2
20 V
–0.3 V
N/A
1.5 A peak
200 mA DC
ROSC
Oscillator Resistor
4.0 V
–0.3 V
1.0 mA
1.0 mA
GATE(H)1, GATE(H)2
High–Side FET Driver
for Channel 1 or 2
20 V
–0.3 V
1.5 A peak
200 mA DC
1.5 A peak
200 mA DC
GATE(L)1, GATE(L)2
Low–Side FET Driver for
Channel 1 or 2
16 V
–0.3 V
1.5 A peak
200 mA DC
1.5 A peak
200 mA DC
GND
Ground
0V
0V
1.5 A peak
200 mA DC
N/A
IS+1, IS+2
Positive Current Sense for
Channel 1 or 2
6.0 V
–0.3 V
1.0 mA
1.0 mA
IS–1, IS–2
Negative Current Sense for
Channel 1 or 2
6.0 V
–0.3 V
1.0 mA
1.0 mA
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CS5422
ELECTRICAL CHARACTERISTICS (0°C < TA < 70°C; 0°C < TJ < 125°C; ROSC = 30.9 k, CCOMP1,2 = 0.1 µF,
10.8 V < VCC < 13.2 V; 10.8 V < BST < 20 V, CGATE(H)1,2 = CGATE(L)1,2 = 1.0 nF, unless otherwise specified.)
Characteristic
Test Conditions
Min
Typ
Max
Unit
–
0.5
1.6
µA
0
–
1.1
V
Error Amplifier
VFB1(2) Bias Current
VFB1(2) = 0 V
VFB1(2) Input Range
–
COMP1,2 Source Current
COMP1,2 = 1.2 V to 2.5 V; VFB1(2) = 0.8 V
15
30
60
µA
COMP1,2 Sink Current
COMP1,2 = 1.2 V; VFB1(2) = 1.2 V
15
30
60
µA
Reference Voltage 1(2)
COMP1 = VFB1; COMP2 = VFB2
0.980
1.000
1.020
V
COMP1,2 Max Voltage
VFB1(2) = 0.8 V
3.0
3.3
–
V
COMP1,2 Min Voltage
VFB1(2) = 1.2 V
–
0.25
0.35
V
Open Loop Gain
–
–
95
–
dB
Unity Gain Band Width
–
–
40
–
kHz
PSRR @ 1.0 kHz
–
–
70
–
dB
Transconductance
–
–
32
–
mmho
Output Impedance
–
–
2.5
–
MΩ
GATE(H) and GATE(L)
High Voltage (AC)
Measure: VCC – GATE(L)1,2;
BST – GATE(H)1,2; Note 2.
–
0
0.5
V
Low Voltage (AC)
Measure:GATE(L)1,2 or GATE(H)1,2; Note 2.
–
0
0.5
V
Rise Time
1.0 V < GATE(L)1,2 < VCC – 1.0 V
1.0 V < GATE(H)1,2 < BST – 1.0 V,
BST ≤ 14 V
–
20
50
ns
Fall Time
VCC – 1.0 > GATE(L)1,2 > 1.0 V
BST – 1.0 > GATE(H)1,2 > 1.0 V,
BST ≤ 14 V
–
15
50
ns
GATE(H) to GATE(L) Delay
GATE(H)1,2 < 2.0 V, GATE(L)1,2 > 2.0 V
BST ≤ 14 V
20
40
70
ns
GATE(L) to GATE(H) Delay
GATE(L)1,2 < 2.0 V, GATE(H)1,2 > 2.0 V;
BST ≤ 14 V
20
40
70
ns
GATE(H)1(2) and GATE(L)1(2)
pull–down.
Resistance to GND
Note 2.
50
125
280
kΩ
Transient Response
COMP1,2 = 1.0 V, VFB1(2) = 0 to 1.2 V
–
150
300
ns
PWM Comparator Offset
VFFB1(2) = 0 V; Increase COMP1,2 until
GATE(H)1,2 starts switching
0.30
0.45
0.60
V
Artificial Ramp
Duty cycle = 50%, Note 2.
40
70
100
mV
Minimum Pulse Width
Note 2.
–
–
300
ns
PWM Comparator
Oscillator
Switching Frequency
ROSC = 61.9 k; Measure GATE(H)1; Note 2.
112
150
188
kHz
Switching Frequency
ROSC = 30.9 k; Measure GATE(H)1
224
300
376
kHz
Switching Frequency
ROSC = 15.1 k; Measure GATE(H)1; Note 2.
450
600
750
kHz
ROSC Voltage
ROSC = 30.9 k, Note 2.
0.970
1.000
1.030
V
–
180
–
°
Phase Difference
–
2. Guaranteed by design, not 100% tested in production.
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CS5422
ELECTRICAL CHARACTERISTICS (continued) (0°C < TA < 70°C; 0°C < TJ < 125°C; ROSC = 30.9 k, CCOMP1,2 = 0.1 µF,
10.8 V < VCC < 13.2 V; 10.8 V < BST < 20 V, CGATE(H)1,2 = CGATE(L)1,2 = 1.0 nF, unless otherwise specified.)
Characteristic
Test Conditions
Min
Typ
Max
Unit
Supply Currents
VCC Current
COMP1,2 = 0 V (No Switching)
–
13
17
mA
BST Current
COMP1,2 = 0 V (No Switching)
–
3.5
6.0
mA
Undervoltage Lockout
Start Threshold
GATE(H) Switching; COMP1,2 charging
7.8
8.6
9.4
V
Stop Threshold
GATE(H) not switching; COMP1,2 discharging
7.0
7.8
8.6
V
Hysteresis
Start–Stop
0.5
0.8
1.5
V
0 V < IS+ 1(2) < 5.5 V, 0 V < IS– 1(2) < 5.5 V
55
70
85
mV
–
0.20
0.25
0.30
V
Hiccup Mode Overcurrent Protection
OVC Comparator Offset Voltage
Discharge Threshold
IS+ 1(2) Bias Current
0 V < IS+ 1(2) < 5.5 V
–1.0
0.1
1.0
µA
IS– 1(2) Bias Current
0 V < IS– 1(2) < 5.5 V
–1.0
0.1
1.0
µA
0
–
5.5
V
OVC Common Mode Range
–
OVC Latch COMP1 Discharge Current
COMP1 = 1.0 V
2.0
5.0
8.0
µA
OVC Latch COMP2 Discharge Current
COMP2 = 1.0 V
0.3
1.2
3.5
mA
5.0
6.0
7.0
–
COMP1 Charge/Discharge
Ratio in OVC
–
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CS5422
PACKAGE PIN DESCRIPTION
PACKAGE PIN #
SO–16
SO–24L
PIN SYMBOL
1
1
GATE(H)1
High Side Switch FET driver pin for channel 1.
2
2
GATE(L)1
Low Side Synchronous FET driver pin for channel 1.
3
–
GND
–
3
PGND
4
4
BST
–
5–8, 17–20
LGND
5
9
IS+1
Positive input for channel 1 overcurrent comparator.
6
10
IS–1
Negative input for channel 1 overcurrent comparator.
7
11
VFB1
Error amplifier inverting input for channel 1.
8
12
COMP1
Channel 1 Error Amp output. PWM Comparator reference
input. A capacitor to LGND provides Error Amp compensation. The same capacitor provides Soft Start timing for channel 1. This pin also disables the channel 1 output when pulled
below 0.3 V.
9
13
COMP2
Channel 2 Error Amp output. PWM Comparator reference
input. A capacitor to LGND provides Error Amp compensation
and Soft Start timing for channel 2. Channel 2 output is disabled when this pin is pulled below 0.3 V.
10
14
VFB2
Error amplifier inverting input for channel 2.
11
15
IS–2
Negative input for channel 2 overcurrent comparator.
12
16
IS+2
Positive input for channel 2 overcurrent comparator.
13
21
ROSC
Oscillator frequency pin. A resistor from this pin to ground
sets the oscillator frequency.
14
22
VCC
15
23
GATE(L)2
Low Side Synchronous FET driver pin for channel 2.
16
24
GATE(H)2
High Side Switch FET driver pin for channel 2.
FUNCTION
Ground pin for all circuitry contained in the IC. This pin is
internally bonded to the substrate of the IC.
Ground pin for the FET drivers.
Power input for GATE(H)1 and GATE(H)2 pins.
Ground pin for the internal control circuitry. The pin is internally bonded to the substrate of the IC.
Input Power supply pin.
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CS5422
VCC
ROSC
BIAS
+
IS+1
−
CLK1
BST
OSC
−
CLK2
S
Reset
Dominant
70 mV
−
70 mV
−
+
FAULT
−
Set
Dominant
+
− 0.25 V
R
GATE(L)1
BST
−
+
S
PWM
Comparator 2
GATE(H)2
Reset
Dominant
FAULT
VCC
GATE(L)2
R
RAMP2
E/A OFF
−
+
E/A OFF
0.45 V
1.2 mA
−
E/A1
−
+
1.0 V
VCC
RAMP1
0.45 V
R
5.0 µA
GATE(H)1
+
Q FAULT
S
−
+
−
+
IS–2
PWM
Comparator 1
+
IS+2
RAMP2
RAMP1
BST
+
+
IS–1
−
+
VCC
8.6 V
−
7.8 V
CURRENT
SOURCE
GEN
+
FAULT
PGND
(24 Lead
pkg. only)
E/A2
1.0 V
VFB1
COMP1
VFB2
Figure 2. Block Diagram
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COMP2
GND
LGND
(24 Lead pkg. only) (16 Lead pkg. only)
CS5422
APPLICATIONS INFORMATION
THEORY OF OPERATION
time to the output load step is not related to the crossover
frequency of the error signal loop.
The error signal loop can have a low crossover frequency,
since the transient response is handled by the ramp signal
loop. The main purpose of this ‘slow’ feedback loop is to
provide DC accuracy. Noise immunity is significantly
improved, since the error amplifier bandwidth can be rolled
off at a low frequency. Enhanced noise immunity improves
remote sensing of the output voltage, since the noise
associated with long feedback traces can be effectively
filtered.
Line and load regulation is drastically improved because
there are two independent control loops. A voltage mode
controller relies on the change in the error signal to
compensate for a deviation in either line or load voltage.
This change in the error signal causes the output voltage to
change corresponding to the gain of the error amplifier,
which is normally specified as line and load regulations. A
current mode controller maintains a fixed error signal during
line transients, since the slope of the ramp signal changes in
this case. However, regulation of load transients still requires
a change in the error signal. The V2 method of control
maintains a fixed error signal for both line and load variation,
since the ramp signal is affected by both line and load.
The stringent load transient requirements of modern
microprocessors require the output capacitors to have very
low ESR. The resulting shallow slope in the output ripple
can lead to pulse width jitter and variation caused by both
random and synchronous noise. A ramp waveform
generated in the oscillator is added to the ramp signal from
the output voltage to provide the proper voltage ramp at the
beginning of each switching cycle. This slope compensation
increases the noise immunity particularly at higher duty
cycle (above 50%).
The CS5422 is a dual power supply controller that utilizes
the V2 control method. Two synchronous V2 buck regulators
can be built using a single controller. The fixed–frequency
architecture, driven from a common oscillator, ensures a
180° phase differential between channels.
V2 Control Method
The V2 method of control uses a ramp signal that is
generated by the ESR of the output capacitors. This ramp is
proportional to the AC current through the main inductor
and is offset by the DC output voltage. This control scheme
inherently compensates for variation in either line or load
conditions, since the ramp signal is generated from the
output voltage itself. The V2 method differs from traditional
techniques such as voltage mode control, which generates an
artificial ramp, and current mode control, which generates
a ramp using the inductor current.
–
GATE(H)
PWM
+
GATE(L)
RAMP
Slope
Compensation
Output
Voltage
Error
Amplifier
VFB
–
COMP
Error
Signal
+
Reference
Voltage
Figure 3. V2 Control with Slope Compensation
The V2 control method is illustrated in Figure 3. The
output voltage generates both the error signal and the ramp
signal. Since the ramp signal is simply the output voltage, it
is affected by any change in the output, regardless of the
origin of that change. The ramp signal also contains the DC
portion of the output voltage, allowing the control circuit to
drive the main switch to 0% or 100% duty cycle as required.
A variation in line voltage changes the current ramp in the
inductor, which causes the V2 control scheme to compensate
the duty cycle. Since any variation in inductor current
modifies the ramp signal, as in current mode control, the V2
control scheme offers the same advantages in line transient
response.
A variation in load current will affect the output voltage,
modifying the ramp signal. A load step immediately changes
the state of the comparator output, which controls the main
switch. The comparator response time and the transition
speed of the main switch determine the load transient
response. Unlike traditional control methods, the reaction
Start Up
The CS5422 features a programmable Soft Start function,
which is implemented through the Error Amplifier and the
external Compensation Capacitor. This feature prevents
stress to the power components and overshoot of the output
voltage during start–up. As power is applied to the regulator,
the CS5422 Undervoltage Lockout circuit (UVL) monitors
the IC’s supply voltage (VCC). The UVL circuit prevents the
MOSFET gates from switching until VCC exceeds the 8.6 V
threshold. A hysteresis function of 800 mV improves noise
immunity. The Compensation Capacitor connected to the
COMP pin is charged by a 30 µA current source. When the
capacitor voltage exceeds the 0.45 V offset of the PWM
comparator, the PWM control loop will allow switching to
occur. The upper gate driver GATE(H) is activated turning
on the upper MOSFET. The current then ramps up through
the main inductor and linearly powers the output capacitors
and load. When the regulator output voltage exceeds the
COMP pin voltage minus the 0.45 V PWM comparator
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CS5422
level. Since the inductor current cannot be changed
instantaneously, regulation is maintained by the output
capacitors during the time required to slew the inductor
current. For better transient response, several high
frequency and bulk output capacitors are usually used.
offset threshold and the artificial ramp, the PWM
comparator terminates the initial pulse.
VIN
8.6 V
Out–of–Phase Synchronization
VCOMP
In out–of–phase synchronization, the turn–on of the
second channel is delayed by half the switching cycle. This
delay is supervised by the oscillator, which supplies a clock
signal to the second channel which is 180° out of phase with
the clock signal of the first channel.
The advantages of out–of–phase synchronization are
many. Since the input current pulses are interleaved with one
another, the overlap time is reduced. The effect of this
overlap reduction is to reduce the input filter requirement,
allowing the use of smaller components. In addition, since
peak current occurs during a shorter time period, emitted
EMI is also reduced, thereby reducing shielding
requirements.
0.45 V
VFB
GATE(H)1
GATE(H)2
UVLO
STARTUP
tS
NORMAL OPERATION
Figure 4. Idealized Waveforms
Normal Operation
During normal operation, the duty cycle of the gate drivers
remains approximately constant as the V2 control loop
maintains the regulated output voltage under steady state
conditions. Variations in supply line or output load
conditions will result in changes in duty cycle to maintain
regulation.
Overvoltage Protection
Overvoltage Protection (OVP) is provided as a result of
the normal operation of the V2 control method and requires
no additional external components. The control loop
responds to an overvoltage condition within 150 ns, turning
off the upper MOSFET and disconnecting the regulator
from its input voltage. This results in a crowbar action to
clamp the output voltage preventing damage to the load. The
regulator remains in this state until the overvoltage
condition ceases.
Gate Charge Effect on Switching Times
When using the onboard gate drivers, the gate charge has
an important effect on the switching times of the FETs. A
finite amount of time is required to charge the effective
capacitor seen at the gate of the FET. Therefore, the rise and
fall times rise linearly with increased capacitive loading,
according to the following graphs.
Average Fall Time
Hiccup Overcurrent Protection
A lossless hiccup mode short circuit protection feature is
provided on the chip. The only external component required
is the COMP1 capacitor. Any overcurrent condition results
in the immediate shutdown of both output phases.
A comparator between the IS+ and IS– on each output
phase detects a short circuit when the voltage difference
between the two pins exceeds 70 mV and sets the fault latch.
The fault latch immediately turns off the error amplifier and
discharges both COMP capacitors. The capacitor connected
to COMP1 is discharged through a 5.0 µA current sink in
order to provide timing for the reset cycle. When COMP1
has fallen below 0.25 V, a comparator resets the fault latch
and error amplifier 1 begins to charge COMP1 with a 30 µA
source current. When COMP1 exceeds the feedback voltage
plus the PWM Comparator offset voltage, the normal
switching cycle will resume.
If the short circuit condition persists through the restart
cycle, the overcurrent reset cycle will repeat itself until the
short circuit is removed, resulting in small hiccup output
pulses while the COMP capacitor charges, as shown in
Figure 6.
Average Rise Time
90
Fall/Rise Time (ns)
80
70
60
50
40
30
20
10
0
0
1
2
3
4
5
6
7
8
Load (nF)
Figure 5. Average Rise and Fall Times
Transient Response
The 150 ns reaction time of the control loop provides fast
transient response to any variations in input voltage and
output current. Pulse–by–pulse adjustment of duty cycle is
provided to quickly ramp the inductor current to the required
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CS5422
Selecting Feedback Divider Resistors
VOUT
R1
VFB
R2
Figure 7. Selecting Feedback Divider Resistors
The feedback pins (VFB1(2)) are connected to external
resistor dividers to set the output voltages. The error
amplifier is referenced to 1.0 V and the output voltage is
determined by selecting resistor divider values. Resistor R1
is selected based on a design trade–off between efficiency
and output voltage accuracy. The output voltage error can be
estimated due to the bias current of the error amplifier
neglecting resistor tolerance:
Figure 6. Hiccup Overcurrent Protection
Output Enable
On/Off control of the regulator outputs can be
implemented by pulling the COMP pins low. The COMP
pins must be driven below the 0.45 V PWM comparator
offset voltage in order to disable the switching of the GATE
drivers.
6 R1
Error% 1 10
100%
1
R2 can be sized after R1 has been determined:
VOUT
1
1
R2 R1
DESIGN GUIDELINES
Definition of the design specifications
Calculating Duty Cycle
The output voltage tolerance can be affected by any or all
of the following reasons:
1. buck regulator output voltage setpoint accuracy;
2. output voltage change due to discharging or charging
of the bulk decoupling capacitors during a load
current transient;
3. output voltage change due to the ESR and ESL of the
bulk and high frequency decoupling capacitors,
circuit traces, and vias;
4. output voltage ripple and noise.
Budgeting the tolerance is left up to the designer who must
take into account all of the above effects and provide an
output voltage that will meet the specified tolerance at the
load.
The designer must also ensure that the regulator
component temperatures are kept within the manufacturer’s
specified ratings at full load and maximum ambient
temperature.
The duty cycle of a buck converter (including parasitic
losses) is given by the formula:
Duty Cycle D VOUT (VHFET VL)
VIN VLFET VHFET VL
where:
VOUT = buck regulator output voltage;
VHFET = high side FET voltage drop due to RDS(ON);
VL = output inductor voltage drop due to inductor wire
DC resistance;
VIN = buck regulator input voltage;
VLFET = low side FET voltage drop due to RDS(ON).
Selecting the Switching Frequency
Selecting the switching frequency is a trade–off between
component size and power losses. Operation at higher
switching frequencies allows the use of smaller inductor and
capacitor values. Nevertheless, it is common to select lower
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CS5422
frequency operation because a higher frequency results in
lower efficiency due to MOSFET gate charge losses.
Additionally, the use of smaller inductors at higher
frequencies results in higher ripple current, higher output
voltage ripple, and lower efficiency at light load currents.
The value of the oscillator resistor is designed to be
linearly related to the switching period. If the designer
prefers not to use Figure 8 to select the necessary resistor, the
following equation quite accurately predicts the proper
resistance for room temperature conditions.
ROSC where:
LMIN = minimum inductance value;
VIN(MIN) = minimum design input voltage;
VOUT = output voltage;
fSW = switching frequency;
ISW(MAX) – maximum design switch current.
The inductor ripple current can then be determined:
V
(1 D)
IL OUT
L fSW
where:
∆IL = inductor ripple current;
VOUT = output voltage;
L = inductor value;
D = duty cycle.
fSW = switching frequency
The designer can now verify if the number of output
capacitors will provide an acceptable output voltage ripple
(1.0% of output voltage is common). The formula below is
used:
21700 fSW
2.31fSW
where:
ROSC = oscillator resistor in kΩ;
fSW = switching frequency in kHz.
800
Frequency (kHz)
700
VOUT
IL ESRMAX
600
500
Rearranging we have:
400
ESRMAX 300
200
100
10
20
30
40
50
VOUT
IL
where:
ESRMAX = maximum allowable ESR;
∆VOUT = 1.0% × VOUT = maximum allowable output
voltage ripple ( budgeted by the designer );
∆IL = inductor ripple current;
VOUT = output voltage.
The number of output capacitors is determined by:
60
ROSC (k)
Figure 8. Switching Frequency
Selection of the Output Inductor
Number of capacitors The inductor should be selected based on its inductance,
current capability, and DC resistance. Increasing the
inductor value will decrease output voltage ripple, but
degrade transient response. There are many factors to
consider in selecting the inductor including cost, efficiency,
EMI and ease of manufacture. The inductor must be able to
handle the peak current at the switching frequency without
saturating, and the copper resistance in the winding should
be kept as low as possible to minimize resistive power loss.
There are a variety of materials and types of magnetic
cores that could be used for this application. Among them
are ferrites, molypermalloy cores (MPP), amorphous and
powdered iron cores. Powdered iron cores are very
commonly used. Powdered iron cores are very suitable due
to its high saturation flux density and have low loss at high
frequencies, a distributed gap and exhibit very low EMI.
The minimum value of inductance which prevents
inductor saturation or exceeding the rated FET current can
be calculated as follows:
ESRCAP
ESRMAX
where:
ESRCAP = maximum ESR per capacitor (specified in
manufacturer’s data sheet).
The designer must also verify that the inductor value
yields reasonable inductor peak and valley currents (the
inductor current is a triangular waveform):
I
IL(PEAK) IOUT L
2
where:
IL(PEAK) = inductor peak current;
IOUT = load current;
∆IL = inductor ripple current.
I
IL(VALLEY) IOUT L
2
where:
IL(VALLEY) = inductor valley current.
(VIN(MIN) VOUT)VOUT
LMIN fSW VIN(MIN) ISW(MAX)
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CS5422
Selection of the Output Capacitors
Similarly, the maximum allowable ESL is calculated from
the following formula:
These components must be selected and placed carefully
to yield optimal results. Capacitors should be chosen to
provide acceptable ripple on the regulator output voltage.
Key specifications for output capacitors are their ESR
(Equivalent Series Resistance), and ESL (Equivalent Series
Inductance). For best transient response, a combination of
low value/high frequency and bulk capacitors placed close
to the load will be required.
In order to determine the number of output capacitors the
maximum voltage transient allowed during load transitions
has to be specified. The output capacitors must hold the
output voltage within these limits since the inductor current
can not change with the required slew rate. The output
capacitors must therefore have a very low ESL and ESR.
The voltage change during the load current transient is:
ESLMAX Selection of the Input Inductor
A common requirement is that the buck controller must
not disturb the input voltage. One method of achieving this
is by using an input inductor and a bypass capacitor. The
input inductor isolates the supply from the noise generated
in the switching portion of the buck regulator and also limits
the inrush current into the input capacitors upon power up.
The inductor’s limiting effect on the input current slew rate
becomes increasingly beneficial during load transients. The
worst case is when the load changes from no load to full load
(load step), a condition under which the highest voltage
change across the input capacitors is also seen by the input
inductor. The inductor successfully blocks the ripple current
while placing the transient current requirements on the input
bypass capacitor bank, which has to initially support the
sudden load change.
The minimum inductance value for the input inductor is
therefore:
t
VOUT IOUT ESL ESR TR
t
COUT
where:
∆IOUT / ∆t = load current slew rate;
∆IOUT = load transient;
∆t = load transient duration time;
ESL = Maximum allowable ESL including capacitors,
circuit traces, and vias;
ESR = Maximum allowable ESR including capacitors
and circuit traces;
tTR = output voltage transient response time.
The designer has to independently assign values for the
change in output voltage due to ESR, ESL, and output
capacitor discharging or charging. Empirical data indicates
that most of the output voltage change (droop or spike
depending on the load current transition) results from the
total output capacitor ESR.
The maximum allowable ESR can then be determined
according to the formula:
ESRMAX V
LIN (dIdt)MAX
where:
LIN = input inductor value;
∆V = voltage seen by the input inductor during a full load
swing;
(dI/dt)MAX = maximum allowable input current slew rate.
The designer must select the LC filter pole frequency so
that at least 40 dB attenuation is obtained at the regulator
switching frequency. The LC filter is a double–pole network
with a slope of –2.0, a roll–off rate of –40 dB/dec, and a
corner frequency:
VESR
IOUT
fC where:
∆VESR = change in output voltage due to ESR (assigned
by the designer)
Once the maximum allowable ESR is determined, the
number of output capacitors can be found by using the
formula:
Number of capacitors VESL t
I
1
2 LC
where:
L = input inductor;
C = input capacitor(s).
SELECTION OF THE POWER FET
FET Basics
ESRCAP
ESRMAX
The use of the MOSFET as a power switch is propelled by
two reasons: 1) Its very high input impedance; and 2) Its very
fast switching times. The electrical characteristics of a
MOSFET are considered to be those of a perfect switch.
Control and drive circuitry power is therefore reduced.
Because the input impedance is so high, it is voltage driven.
The input of the MOSFET acts as if it were a small capacitor,
which the driving circuit must charge at turn on. The lower
the drive impedance, the higher the rate of rise of VGS, and
the faster the turn–on time. Power dissipation in the
where:
ESRCAP = maximum ESR per capacitor (specified in
manufacturer’s data sheet).
ESRMAX = maximum allowable ESR.
The actual output voltage deviation due to ESR can then
be verified and compared to the value assigned by the
designer:
VESR IOUT ESRMAX
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CS5422
switching MOSFET consists of 1) conduction losses, 2)
leakage losses, 3) turn–on switching losses, 4) turn–off
switching losses, and 5) gate–transitions losses. The latter
three losses are proportional to frequency.
The most important aspect of FET performance is the
Static Drain–To–Source On–Resistance (RDS(ON)), which
affects regulator efficiency and FET thermal management
requirements. The On–Resistance determines the amount of
current a FET can handle without excessive power
dissipation that may cause overheating and potentially
catastrophic failure. As the drain current rises, especially
above the continuous rating, the On–Resistance also
increases. Its positive temperature coefficient is between
+0.6%/°C and +0.85%/°C. The higher the On–Resistance
the larger the conduction loss is. Additionally, the FET gate
charge should be low in order to minimize switching losses
and reduce power dissipation.
Both logic level and standard FETs can be used.
Voltage applied to the FET gates depends on the
application circuit used. Both upper and lower gate driver
outputs are specified to drive to within 1.5 V of ground when
in the low state and to within 2.0 V of their respective bias
supplies when in the high state. In practice, the FET gates
will be driven rail–to–rail due to overshoot caused by the
capacitive load they present to the controller IC.
where:
PSWH(ON) = upper MOSFET switch–on losses;
PSWH(OFF) = upper MOSFET switch–off losses;
VIN = input voltage;
IOUT = load current;
tRISE = MOSFET rise time (from FET manufacturer’s
switching characteristics performance curve);
tFALL = MOSFET fall time (from FET manufacturer’s
switching characteristics performance curve);
T = 1/fSW = period.
The total power dissipation in the switching MOSFET can
then be calculated as:
PHFET(TOTAL) PRMS(H) PSWH(ON) PSWH(OFF)
where:
PHFET(TOTAL) = total switching (upper) MOSFET losses;
PRMS(H) = upper MOSFET switch conduction Losses;
PSWH(ON) = upper MOSFET switch–on losses;
PSWH(OFF) = upper MOSFET switch–off losses;
Once the total power dissipation in the switching FET is
known, the maximum FET switch junction temperature can
be calculated:
TJ TA [PHFET(TOTAL) RJA]
where:
TJ = FET junction temperature;
TA = ambient temperature;
PHFET(TOTAL) = total switching (upper) FET losses;
RΘJA = upper FET junction–to–ambient thermal resistance.
Selection of the Switching (Upper) FET
The designer must ensure that the total power dissipation
in the FET switch does not cause the power component’s
junction temperature to exceed 150°C.
The maximum RMS current through the switch can be
determined by the following formula:
IRMS(H) IL(PEAK)2 (IL(PEAK) IL(VALLEY))
IL(VALLEY)2 D
Selection of the Synchronous (Lower) FET
The switch conduction losses for the lower FET can be
calculated as follows:
PRMS(L) IRMS2 RDS(ON)
[IOUT (1 D)]2 RDS(ON)
3
where:
PRMS(L) = lower MOSFET conduction losses;
IOUT = load current;
D = Duty Cycle;
RDS(ON) = lower FET drain–to–source on–resistance.
The synchronous MOSFET has no switching losses,
except for losses in the internal body diode, because it turns
on into near zero voltage conditions. The MOSFET body
diode will conduct during the non–overlap time and the
resulting power dissipation (neglecting reverse recovery
losses) can be calculated as follows:
where:
IRMS(H) = maximum switching MOSFET RMS current;
IL(PEAK) = inductor peak current;
IL(VALLEY) = inductor valley current;
D = duty cycle.
Once the RMS current through the switch is known, the
switching MOSFET conduction losses can be calculated:
PRMS(H) IRMS(H)2 RDS(ON)
where:
PRMS(H) = switching MOSFET conduction losses;
IRMS(H) = maximum switching MOSFET RMS current;
RDS(ON) = FET drain–to–source on–resistance
The upper MOSFET switching losses are caused during
MOSFET switch–on and switch–off and can be determined
by using the following formula:
PSWL VSD ILOAD non–overlap time fSW
where:
PSWL = lower FET switching losses;
VSD = lower FET source–to–drain voltage;
ILOAD = load current;
PSWH PSWH(ON) PSWH(OFF)
V IOUT (tRISE tFALL)
IN
6T
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CS5422
Non–overlap time = GATE(L)–to–GATE(H) or
GATE(H)–to–GATE(L) delay (from CS5422 data sheet
Electrical Characteristics section);
fSW = switching frequency.
The total power dissipation in the synchronous (lower)
MOSFET can then be calculated as:
The junction temperature of the control IC is primarily a
function of the PCB layout, since most of the heat is removed
through the traces connected to the pins of the IC.
Current Sensing
The current supplied to the load can be sensed easily using
the IS+ and IS– pins for the output. These pins sense a
voltage, proportional to the output current, and compare it to
a fixed internal voltage threshold. When the differential
voltage exceeds 70 mV, the internal overcurrrent protection
system goes into hiccup mode. Two methods for sensing the
current are available.
Sense Resistor. A sense resistor can be added in series
with the inductor. When the voltage drop across the sense
resistor exceeds the internal voltage threshold of 70 mV, a
fault condition is set.
The sense resistor is selected according to:
PLFET(TOTAL) PRMS(L) PSWL
where:
PLFET(TOTAL) = Synchronous (lower) FET total losses;
PRMS(L) = Switch Conduction Losses;
PSWL = Switching losses.
Once the total power dissipation in the synchronous FET
is known the maximum FET switch junction temperature
can be calculated:
TJ TA [PLFET(TOTAL) RJA]
RSENSE 0.070 V
ILIMIT
where:
TJ = MOSFET junction temperature;
TA = ambient temperature;
PLFET(TOTAL) = total synchronous (lower) FET losses;
RΘJA = lower FET junction–to–ambient thermal resistance.
In a high current supply, the sense resistor will be a very
low value, typically less than 10 mΩ. Such a resistor can be
either a discrete component or a PCB trace. The resistance
value of a discrete component can be more precise than a
PCB trace, but the cost is also greater.
Setting the current limit using an external sense resistor is
very precise because all the values can be designed to
specific tolerances. However, the disadvantage of using a
sense resistor is its additional constant power loss and heat
generation.
Inductor ESR. Another means of sensing current is to use
the intrinsic resistance of the inductor. A model of an
inductor reveals that the windings of an inductor have an
effective series resistance (ESR).
The voltage drop across the inductor ESR can be
measured with a simple parallel circuit: an RC integrator. If
the value of RS1 and C are chosen such that:
Control IC Power Dissipation
The power dissipation of the IC varies with the MOSFETs
used, VCC, and the CS5422 operating frequency. The
average MOSFET gate charge current typically dominates
the control IC power dissipation.
The IC power dissipation is determined by the formula:
PCONTROL(IC) ICC1VCC1 IBSTVBST PGATE(H)1
PGATE(L)1 PGATE(H)2 PGATE(L)2
where:
PCONTROL(IC) = control IC power dissipation;
ICC1 = IC quiescent supply current;
VCC1 = IC supply voltage;
PGATE(H) = upper MOSFET gate driver (IC) losses;
PGATE(L) = lower MOSFET gate driver (IC) losses.
The upper (switching) MOSFET gate driver (IC) losses
are:
L R C
S1
ESR
then the voltage measured across the capacitor C will be:
VC ESR ILIM
PGATE(H) QGATE(H) fSW VBST
Selecting Components. Select the capacitor C first. A
value of 0.1 µF is recommended. The value of RS1 can be
selected according to:
where:
PGATE(H) = upper MOSFET gate driver (IC) losses;
QGATE(H) = total upper MOSFET gate charge at VCC;
fSW = switching frequency;
The lower (synchronous) MOSFET gate driver (IC)
losses are:
RS1 1
ESR C
Typical values for inductor ESR range in the low m;
consult manufacturer’s datasheet for specific details.
Selection of components at these values will result in a
current limit of:
PGATE(L) QGATE(L) fSW VCC
where:
PGATE(L) = lower MOSFET gate driver (IC) losses;
QGATE(L) = total lower MOSFET gate charge at VCC;
fSW = switching frequency;
ILIM 0.070 V
ESR
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14
CS5422
L
VCC
ESR
COMP
GATE(H)
Co
C
RS1
CCOMP
CS5422
GATE(L)
R2
C1
R1
IS+
IS–
GATE(L)
Figure 9. Inductor ESR Current Sensing
Figure 10. Small RC Filter Provides the
Proper Voltage Ramp at the Beginning of
Each On–Time Cycle
Given an ESR value of 3.5 mΩ, the current limit becomes
20 A. If an increased current limit is required, a resistor
divider can be added.
The advantages of setting the current limit by using the
winding resistance of the inductor are that efficiency is
maximized and heat generation is minimized. The tolerance
of the inductor ESR must be factored into the design of the
current limit. Finally, one or two more components are
required for this approach than with resistor sensing.
The artificial voltage ramp created by the slope
compensation scheme results in improved control loop
stability provided that the RC filter time constant is smaller
than the off–time cycle duration (time during which the
lower MOSFET is conducting). It is important that the series
combination of R1 and R2 is high enough in resistance to
avoid loading the GATE(L) pin. Also, C1 should be very
small (less than a few nF) to avoid heating the part.
Adding External Slope Compensation
Today’s voltage regulators are expected to meet very
stringent load transient requirements. One of the key factors
in achieving tight dynamic voltage regulation is low ESR.
Low ESR at the regulator output results in low output
voltage ripple. The consequence is, however, that very little
voltage ramp exists at the control IC feedback pin (VFB),
resulting in increased regulator sensitivity to noise and the
potential for loop instability. In applications where the
internal slope compensation is insufficient, the performance
of the CS5422–based regulator can be improved through the
addition of a fixed amount of external slope compensation
at the output of the PWM Error Amplifier (the COMP pin)
during the regulator off–time. Referring to Figure 8, the
amount of voltage ramp at the COMP pin is dependent on the
gate voltage of the lower (synchronous) FET and the value
of resistor divider formed by R1and R2.
To Synchronous
FET
EMI MANAGEMENT
As a consequence of large currents being turned on and off
at high frequency, switching regulators generate noise as a
consequence of their normal operation. When designing for
compliance with EMI/EMC regulations, additional
components may be added to reduce noise emissions. These
components are not required for regulator operation and
experimental results may allow them to be eliminated. The
input filter inductor may not be required because bulk filter
and bypass capacitors, as well as other loads located on the
board will tend to reduce regulator di/dt effects on the circuit
board and input power supply. Placement of the power
component to minimize routing distance will also help to
reduce emissions.
LAYOUT GUIDELINES
When laying out the CPU buck regulator on a printed
circuit board, the following checklist should be used to
ensure proper operation of the CS5422.
1. Rapid changes in voltage across parasitic capacitors
and abrupt changes in current in parasitic inductors
are major concerns for a good layout.
2. Keep high currents out of sensitive ground
connections.
3. Avoid ground loops as they pick up noise. Use star or
single point grounding.
–t
R2
VSLOPECOMP VGATE(L) (1 e )
R1 R2
where:
VSLOPECOMP = amount of slope added;
VGATE(L) = lower MOSFET gate voltage;
R1, R2 = voltage divider resistors;
t = tON or tOFF (switch off–time);
τ = RC constant determined by C1 and the parallel
combination of R1, R2 neglecting the low driver
output impedance.
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CS5422
9. Place the switching MOSFET as close to the input
capacitors as possible.
10. Place the output capacitors as close to the load as
possible.
11. Place the COMP capacitor as close as possible to the
COMP pin.
12. Connect the filter components of the following pins:
ROSC, VFB, VOUT, and COMP to the GND pin with a
single trace, and connect this local GND trace to the
output capacitor GND.
13. Place the VCC bypass capacitors as close as possible
to the IC.
14. Place the ROSC resistor as close as possible to the
ROSC pin.
15. Include provisions for 100–100pF capacitor across
each resistor of the feedback network to improve
noise immunity and add COMP.
16. Assign the output with lower duty cycle to channel 2,
which has better noise immunity.
4. For high power buck regulators on double–sided
PCB’s a single ground plane (usually the bottom) is
recommended.
5. Even though double sided PCB’s are usually
sufficient for a good layout, four–layer PCB’s are the
optimum approach to reducing susceptibility to
noise. Use the two internal layers as the power and
GND planes, the top layer for power connections and
component vias, and the bottom layers for the noise
sensitive traces.
6. Keep the inductor switching node small by placing
the output inductor, switching and synchronous FETs
close together.
7. The MOSFET gate traces to the IC must be short,
straight, and wide as possible.
8. Use fewer, but larger output capacitors, keep the
capacitors clustered, and use multiple layer traces
with heavy copper to keep the parasitic resistance
low.
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CS5422
PACKAGE DIMENSIONS
SO–16
D SUFFIX
CASE 751B–05
ISSUE J
NOTES:
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2. CONTROLLING DIMENSION: MILLIMETER.
3. DIMENSIONS A AND B DO NOT INCLUDE
MOLD PROTRUSION.
4. MAXIMUM MOLD PROTRUSION 0.15 (0.006)
PER SIDE.
5. DIMENSION D DOES NOT INCLUDE DAMBAR
PROTRUSION. ALLOWABLE DAMBAR
PROTRUSION SHALL BE 0.127 (0.005) TOTAL
IN EXCESS OF THE D DIMENSION AT
MAXIMUM MATERIAL CONDITION.
–A–
16
9
–B–
1
P
8 PL
0.25 (0.010)
8
B
M
S
G
R
K
F
X 45 C
–T–
SEATING
PLANE
J
M
D
16 PL
0.25 (0.010)
M
T B
S
A
DIM
A
B
C
D
F
G
J
K
M
P
R
MILLIMETERS
MIN
MAX
9.80
10.00
3.80
4.00
1.35
1.75
0.35
0.49
0.40
1.25
1.27 BSC
0.19
0.25
0.10
0.25
0
7
5.80
6.20
0.25
0.50
INCHES
MIN
MAX
0.386
0.393
0.150
0.157
0.054
0.068
0.014
0.019
0.016
0.049
0.050 BSC
0.008
0.009
0.004
0.009
0
7
0.229
0.244
0.010
0.019
S
SO–24L
DWF SUFFIX
CASE 751E–04
ISSUE E
–A–
24
13
–B–
12X
P
0.010 (0.25)
1
NOTES:
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2. CONTROLLING DIMENSION: MILLIMETER.
3. DIMENSIONS A AND B DO NOT INCLUDE
MOLD PROTRUSION.
4. MAXIMUM MOLD PROTRUSION 0.15 (0.006)
PER SIDE.
5. DIMENSION D DOES NOT INCLUDE DAMBAR
PROTRUSION. ALLOWABLE DAMBAR
PROTRUSION SHALL BE 0.13 (0.005) TOTAL IN
EXCESS OF D DIMENSION AT MAXIMUM
MATERIAL CONDITION.
M
B
M
12
24X
D
J
0.010 (0.25)
M
T A
S
B
S
F
R
C
–T–
SEATING
PLANE
M
22X
G
K
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17
X 45 DIM
A
B
C
D
F
G
J
K
M
P
R
MILLIMETERS
MIN
MAX
15.25
15.54
7.40
7.60
2.35
2.65
0.35
0.49
0.41
0.90
1.27 BSC
0.23
0.32
0.13
0.29
0
8
10.05
10.55
0.25
0.75
INCHES
MIN
MAX
0.601
0.612
0.292
0.299
0.093
0.104
0.014
0.019
0.016
0.035
0.050 BSC
0.009
0.013
0.005
0.011
0
8
0.395
0.415
0.010
0.029
CS5422
Notes
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CS5422
Notes
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CS5422
V2 is a trademark of Switch Power, Inc.
ON Semiconductor and
are trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes
without further notice to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular
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