MP2108 2A, 6V, 740KHz Synchronous Buck Converter The Future of Analog IC Technology DESCRIPTION FEATURES The MP2108 is a 2A, 740KHz synchronous buck converter designed for low voltage applications requiring high efficiency. It is capable of providing output voltages as low as 0.9V, and integrates top and bottom switches to minimize power loss and component count. The 740KHz switching frequency allows for small filtering components, further reducing the solution size. • • • • • • • • • • The MP2108 includes cycle-by-cycle current limiting and under voltage lockout. Internal power switches, combined with the tiny 10-pin MSOP or 3mm x 3mm QFN packages, provide a solution requiring a minimum of board space. QFN package is recommended if output current is higher than 1.5A. EVALUATION BOARD REFERENCE 2A Output Current Synchronous Rectification Internal 160mΩ and 190mΩ Power Switches Input Range of 2.6V to 6V Over 95% Efficiency Under Voltage Lockout Protection Soft-Start Operation Thermal Shutdown Internal Current Limit (Source & Sink) Tiny 10-Pin MSOP and 3x3 QFN Packages APPLICATIONS • • • • SOHO Routers, PCMCIA Cards, Mini PCI Handheld Computers, PDAs Cell phones, Digital Still and Video Cameras Small LCD Displays “MPS” and “The Future of Analog IC Technology” are Registered Trademarks of Monolithic Power Systems, Inc. Board Number Dimensions EV2108DQ/DK-00A 2.5”X x 2.0”Y x 0.5”Z TYPICAL APPLICATION 10nF INPUT 2.6V to 6V 100 OFF ON 6 8 10nF 2.2nF VIN BST LX RUN MP2108 SS COMP VREF FB SGND PGND 9 5 10nF 4 80 3 7 OUTPUT 2.5V / 2A EFFICIENCY (%) 10 VOUT=2.5V 90 1 2 VOUT=1.2V 70 60 50 40 30 20 10 0 0.01 MP2108 Rev 1.1 10/2/2006 VIN=3.3V 0.1 1 LOAD CURRENT (A) www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 10 1 MP2108 – 2A, 6V, 740KHz SYNCHRONOUS BUCK CONVERTER PACKAGE REFERENCE TOP VIEW TOP VIEW BST 1 10 RUN VREF VIN 2 9 VREF LX 3 8 COMP PGND 4 7 FB SGND 5 6 SS BST 1 10 RUN VIN 2 9 LX 3 8 COMP PGND 4 7 FB SGND 5 6 SS EXPOSED PAD ON BACKSIDE Part Number* MP2108DK * Package MSOP10 Temperature Part Number** Package*** Temperature MP2108DQ QFN10 (3mm x 3mm) –40°C to +85°C –40°C to +85°C ** For Tape & Reel, add suffix –Z (eg. MP2108DK–Z) For RoHS compliant packaging, add suffix –LF (eg. MP2108DK–LF–Z) *** For Tape & Reel, add suffix –Z (eg. MP2108DQ–Z) For RoHS compliant packaging, add suffix –LF (eg. MP2108DQ–LF–Z) Recommended for output currents higher than 1.5A ABSOLUTE MAXIMUM RATINGS (1) Thermal Resistance Input Supply Voltage VIN ............................. 6.5V LX Voltage VLX ..................... –0.3V to VIN + 0.3V BST to LX Voltage ......................... –0.3V to +6V Voltage on All Other Pins............... –0.3V to +6V Storage Temperature............... –55°C to +150°C MSOP10 ................................ 150 ..... 65... °C/W QFN10 (3mm x 3mm) ............. 50 ...... 12... °C/W Recommended Operating Conditions (2) (3) θJA θJC Notes: 1) Exceeding these ratings may damage the device. 2) The device is not guaranteed to function outside of its operating conditions. 3) Measured on approximately 1” square of 1 oz copper. Input Supply Voltage VIN ...................... 2.6V to 6 Output Voltage VOUT ...........................0.9V to 5V Operating Temperature.............. –40°C to +85°C ELECTRICAL CHARACTERISTICS VIN = 5.0V, TA = +25°C, unless otherwise noted. Parameter Input Voltage Range Input Undervoltage Lockout Input Undervoltage Lockout Hysteresis Shutdown Supply Current Operating Supply Current VREF Voltage RUN Input Low Voltage RUN Input High Voltage RUN Hysteresis RUN Input Bias Current MP2108 Rev 1.1 10/2/2006 Symbol Condition VIN VREF VIL VHL Min 2.6 VRUN ≤ 0.3V VRUN > 2V, VFB = 1.1V VIN = 2.6V to 6V Typ 2.2 Units V V 100 mV 0.5 1.2 2.4 Max 6 1.0 1.8 0.4 1.5 100 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 1 µA mA V V V mV µA 2 MP2108 – 2A, 6V, 740KHz SYNCHRONOUS BUCK CONVERTER ELECTRICAL CHARACTERISTICS (continued) VIN = 5.0V, TA = +25°C, unless otherwise noted. Parameter Oscillator Switching Frequency Maximum Duty Cycle Minimum On Time Error Amplifier Voltage Gain Transconductance COMP Maximum Output Current FB Regulation Voltage FB Input Bias Current Soft-Start Soft-Start Current Output Switch On-Resistance Switch On Resistance Synchronous Rectifier On Resistance Switch Current Limit (Source) Synchronous Rectifier Current Limit (Sink) Thermal Shutdown Symbol Condition fSW DMAX TON VFB = 0.7V Min Typ Max Units 620 85 740 920 200 KHz % ns 400 450 ±40 895 –100 V/V µA/V µA mV nA AVEA GEA VFB IFB 875 VFB = 0.9V ISS VIN = 5V VIN = 3V VIN = 5V VIN = 3V 2.5 915 2 µA 190 280 160 230 3.5 mΩ mΩ mΩ mΩ A 350 mA 160 °C PIN FUNCTIONS Pin # Name 1 BST 2 VIN 3 LX 4 PGND 5 SGND 6 SS 7 FB 8 COMP 9 VREF 10 RUN MP2108 Rev 1.1 10/2/2006 Description Power Switch Boost. BST powers the gate of the high-side N-Channel power MOSFET switch. Connect a 10nF or greater capacitor between BST and LX. Internal Power Input. VIN supplies the power to the MP2108 through the internal LDO regulator. Bypass VIN to PGND with a 10µF or greater capacitor. Connect VIN to the input source voltage. Output Switching Node. LX is the source of the high-side N-Channel switch and the drain of the low-side N-Channel switch. Connect the output LC filter between LX and the output. Power Ground. PGND is the source of the N-Channel MOSFET synchronous rectifier. Connect PGND to SGND as close to the MP2108 as possible. Signal Ground. Soft-Start Input. Place a capacitor from SS to SGND to set the soft-start period. The MP2108 sources 2µA from SS to the soft-start capacitor at start-up. As the voltage at SS rises, the feedback threshold voltage increases to limit inrush current at startup. Feedback Input. FB is the inverting input of the internal error amplifier. Connect a resistive voltage divider from the output voltage to FB to set the output voltage. Compensation Node. COMP is the output of the error amplifier. Connect a series RC network to compensate the regulation control loop. Internal 2.4V Regulator Bypass. Connect a 10nF capacitor between VREF and SGND to bypass the internal regulator. Do not apply any load to VREF. On/Off Control Input. Drive RUN high to turn on the MP2108, drive RUN low to turn the MP2108 off. For automatic startup, connect RUN to VIN via a 100kΩ pull-up resistor. www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 3 MP2108 – 2A, 6V, 740KHz SYNCHRONOUS BUCK CONVERTER TYPICAL PERFORMANCE CHARACTERISTICS Circuit of Figure 2, VIN = 5V, VOUT = 2.5V, L1 = 5µH, C1 = 10µF, C2 = 22µF, TA = +25°C, unless otherwise noted. Efficiency vs Load Current 100 Peak Current vs Duty Cycle 70 VOUT=1.2V 60 PEAK CURRENT (A) 80 EFFICIENCY (%) 5.0 VOUT=3.3V 90 VOUT=2.5V 50 40 30 20 10 0.1 1 LOAD CURRENT (A) 4.0 3.5 3.0 2.5 VIN=5V 0 0.01 4.5 2.0 10 Feedback Voltage vs Die Temperature 0.902 0.900 0.898 0.896 0.894 0.892 0.890 0.888 0.886 0.884 -50 -25 0 25 50 75 100 125 150 DIE TEMPERATURE (OC) MP2108 Rev 1.1 10/2/2006 20 40 60 80 DUTY CYCLE (%) 100 Switching Frequency vs Die Temperature SWITCHING FREQUENCY (KHz) FEEDBACK VOTLAGE (V) 0.904 0 830 810 790 770 750 730 710 690 670 650 630 -50 -25 0 25 50 75 100 125 DIE TEMPERATURE (OC) www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 4 MP2108 – 2A, 6V, 740KHz SYNCHRONOUS BUCK CONVERTER TYPICAL PERFORMANCE CHARACTERISTICS (continued) Circuit of Figure 2, VIN = 5V, VOUT = 2.5V, L1 = 5µH, C1 = 10µF, C2 = 22µF, TA = +25°C, unless otherwise noted. 0.5A-1A Step Resistive Load Full Load No Load VOUT AC Coupled 10mV/div. Load Transient Steady State Operation Steady State Operation VOUT AC Coupled 10mV/div. IL 1A/div. VOUT AC Coupled 200mV/div. IL 2A/div. VSW 5V/div. IL 1A/div. VSW 5V/div. 400ns/div. 400ns/div. Start-up through Enable Start-up through Enable Shut-down through Enable Full Load No Load No Load VOUT 2V/div. VOUT 1V/div. VOUT 2V/div. VEN 2V/div. VEN 5V/div. VEN 2V/div. 1ms/div. 100ms/div. 400ms/div. Shut-down through Enable Full Load VOUT 2V/div. VEN 2V/div. 200ms/div. MP2108 Rev 1.1 10/2/2006 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 5 MP2108 – 2A, 6V, 740KHz SYNCHRONOUS BUCK CONVERTER OPERATION VIN OFF ON RUN 10 ENABLE CKT & LDO REGULATOR Vdr CURRENT SENSE AMPLIFIER + -BST VREF 2.4V PWM COMPARATOR 9 C6 GATE DRIVE REGULATOR C1 2 VIN 2.6V to 6V 1 + -- C7 LX CONTROL LOGIC Vdr L1 VOUT 3 C2 720KHz OSCILLATOR RAMP VBP CURRENT LIMIT COMPARATOR + -- UVLO & THERMAL SHUTDOWN R2 + -- PGND SS 4 C5 6 -FB GM -ERROR AMPLIFIER VFB 0.9V CURRENT LIMIT THRESHOLD 5 R1 8 SGND C4 7 + COMP R3 C3 Figure 1—Functional Block Diagram MP2108 Rev 1.1 10/2/2006 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 6 MP2108 – 2A, 6V, 740KHz SYNCHRONOUS BUCK CONVERTER The average inductor current is controlled by the voltage at COMP, which in turn, is controlled by the output voltage. Thus the output voltage controls the inductor current to satisfy the load. The MP2108 measures the output voltage through an external resistive voltage divider and compares it to the internal 0.9V reference to generate the error voltage at COMP. The current-mode regulator uses the voltage at COMP and compares it to the inductor current to regulate the output voltage. The use of current-mode regulation improves transient response and control loop stability. Since the high-side N-Channel MOSFET requires voltage above VIN to drive its gate, a bootstrap capacitor from LX to BST is required to drive the high-side MOSFET gate. When LX is driven low (through the low-side MOSFET), the BST capacitor is internally charged. The voltage at BST is applied to the high-side MOSFET gate to turn it on. Voltage is maintained until the high-side MOSFET is turned off and the low-side MOSFET is turned on, and the cycle repeats. Connect a 10nF or greater capacitor from BST to SW to drive the high-side MOSFET gate. At the beginning of each cycle, the high-side N-Channel MOSFET is turned on, forcing the inductor current to rise. The current at the drain of the high-side MOSFET is internally measured and converted to a voltage by the current sense amplifier. That voltage is compared to the error voltage at COMP. When the inductor current rises sufficiently, the PWM comparator turns off the high-side switch and turns on the low-side switch; forcing the inductor current to decrease. APPLICATION INFORMATION C7 10nF INPUT 2.6V to 6V 10 6 8 C5 10nF 1 2 C4 OPEN C3 2.2nF VIN BST RUN SS LX MP2108 COMP VREF C6 10nF 9 FB 3 OUTPUT 2.5V / 2A 7 SGND PGND 5 4 Figure 2—Typical Application Circuit Internal Low-Dropout Regulator The internal power to the MP2108 is supplied from the input voltage (VIN) through an internal 2.4V low-dropout linear regulator, whose output is VREF. Bypass VREF to SGND with a 10nF or greater capacitor for proper operation. The internal regulator can not supply more current than is required to operate the MP2108. Therefore, do not apply any external load to VREF. MP2108 Rev 1.1 10/2/2006 Soft-Start The MP2108 includes a soft-start timer that slowly ramps the output voltage at startup to prevent excessive current at the input. When power is applied to the MP2108, and RUN is asserted. A 2µA internal current source charges the external capacitor at SS. As the capacitor charges, the voltage at SS rises. The MP2108 internally limits the feedback threshold voltage at FB to that of the voltage at SS. This forces the output voltage to rise at the same www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 7 MP2108 – 2A, 6V, 740KHz SYNCHRONOUS BUCK CONVERTER rate as the voltage at SS, forcing a linear output voltage ramp from 0V to the desired regulation voltage during soft-start. The soft-start period is determined by the equation: t SS = 0.45 × C5 Where C5 (in nF) is the soft-start capacitor from SS to GND, and tSS (in ms) is the soft-start period. Determine the capacitor required for a given soft-start period by the equation: C5 = 2.22 × t SS Use values for C5 between 10nF and 22nF to set the soft-start period between 4ms and 10ms. Setting the Output Voltage (see Figure 2) Set the output voltage by selecting the resistive voltage divider ratio. The voltage divider drops the output voltage to the 0.9V feedback voltage. Use 10kΩ for the low-side resistor of the voltage divider. Determine the high-side resistor by the equation: ⎛V ⎞ R2 = ⎜⎜ OUT − 1⎟⎟ × R1 0 . 9 V ⎝ ⎠ Where R2 is the high-side resistor, R1 is the low-side resistor and VOUT is the output voltage. Selecting the Input Capacitor The input current to the step-down converter is discontinuous, so a capacitor is required to supply the AC current to the step-down converter while maintaining the DC input voltage. A low ESR capacitor is required to keep the noise at the IC to a minimum. Ceramic capacitors are preferred, but tantalum or low ESR electrolytic capacitors are also an option. The capacitor can be electrolytic, tantalum or ceramic. Because it absorbs the input switching current, it must have an adequate ripple current rating. Use a capacitor with RMS current rating greater than 1/2 of the DC load current. For stable operation, place the input capacitor as close to the IC as possible. A smaller high quality 0.1µF ceramic capacitor may be placed closer to the IC with the larger capacitor placed further away. MP2108 Rev 1.1 10/2/2006 If using this technique, it is recommended that the larger capacitor be a tantalum or electrolytic type. All ceramic capacitors should be placed close to the IC. For most applications, a 10µF ceramic capacitor will work. Selecting the Output Capacitor The output capacitor (C2) is required to maintain the DC output voltage. Low ESR capacitors are preferred to keep the output voltage ripple to a minimum. The characteristics of the output capacitor also affect the stability of the regulation control system. Ceramic, tantalum, or low ESR electrolytic capacitors are recommended. The output voltage ripple is: VRIPPLE = ⎛ VOUT V × ⎜1 − OUT f SW × L ⎜⎝ VIN ⎞ ⎞ ⎛ 1 ⎟ ⎟⎟ × ⎜ R ESR + ⎜ ⎟ 8 f C 2 × × SW ⎠ ⎝ ⎠ Where VRIPPLE is the output voltage ripple, fSW is the switching frequency, VIN is the input voltage and RESR is the equivalent series resistance of the output capacitors. Choose an output capacitor to satisfy the output ripple requirements of the design. A 22µF ceramic capacitor is suitable for most applications. Selecting the Inductor The inductor is required to supply constant current to the output load while being driven by the switched input voltage. A larger value inductor results in less ripple current that in turn results in lower output ripple voltage. However, the larger value inductor is likely to have a larger physical size and higher series resistance. Choose an inductor that does not saturate under the worst-case load conditions. A good rule for determining the inductance is to allow peak-to-peak ripple current to be approximately 30% to 40% of the maximum load current. Make sure that the peak inductor current (the load current plus half the peak-topeak inductor ripple current) is below 2.5A to prevent loss of regulation due to the current limit. www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 8 MP2108 – 2A, 6V, 740KHz SYNCHRONOUS BUCK CONVERTER Calculate the required inductance value by the equation: L= VOUT × (VIN − VOUT ) VIN × f SW × ∆I Where ∆I is the peak-to-peak inductor ripple current. It is recommended to choose ∆I to be 30%~40% of the maximum load current. Compensation The system stability is controlled through the COMP pin. COMP is the output of the internal transconductance error amplifier. A series capacitor-resistor combination sets a pole-zero combination to control the characteristics of the control system. The DC loop gain is: ⎛ V A VDC = ⎜⎜ FB ⎝ VOUT ⎞ ⎟ × A VEA × G CS × R LOAD ⎟ ⎠ Where VFB is the feedback voltage, 0.9V, AVEA is the transconductance error amplifier voltage gain, 400 V/V and GCS is the current sense transconductance, (roughly the output current divided by the voltage at COMP), 4.5A/V. RLOAD is the load resistance: R LOAD = VOUT I OUT Where IOUT is the output load current. The system has 2 poles of importance, one is due to the compensation capacitor (C3), and the other is due to the load resistance and the output capacitor (C2), where: fP1 = GEA 2π × A VEA × C3 P1 is the first pole, and GEA is the error amplifier transconductance (450µA/V) and fP 2 = If large value capacitors with relatively high equivalent-series-resistance (ESR) are used, the zero due to the capacitance and ESR of the output capacitor can be compensated by a third pole set by R3 and C4. The pole is: fP3 = 1 2π × R3 × C4 The system crossover frequency (the frequency where the loop gain drops to 1dB or 0dB) is important. Set the crossover frequency below one tenth of the switching frequency to insure stable operation. Lower crossover frequencies result in slower response and worse transient load recovery. Higher crossover frequencies degrade the phase and/or gain margins and can result in instability. Table 1—Compensation Values for Typical Output Voltage/Capacitor Combinations VOUT C2 1.8V 22µF Ceramic 2.5V 22µF Ceramic 3.3V 22µF Ceramic 47µF Tantalum 1.8V (300mΩ) 47µF Tantalum 2.5V (300mΩ) 47µF Tantalum 3.3V (300mΩ) R3 C3 C4 6.8kΩ 9.1kΩ 12kΩ 3.3nF 2.2nF 1.8nF None None None 13kΩ 2nF 1nF 18kΩ 1.2nF 750pF 24kΩ 1nF 560pF Choosing the Compensation Components The values of the compensation components listed in Table 1 yields a stable control loop for the given output voltage and capacitor. To optimize the compensation components for conditions not listed in Table 1, use the following procedure. 1 2π × R LOAD × C2 The system has one zero of importance, due to the compensation capacitor (C3) and the compensation resistor (R3). The zero is: f Z1 = MP2108 Rev 1.1 10/2/2006 1 2π × R3 × C3 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 9 MP2108 – 2A, 6V, 740KHz SYNCHRONOUS BUCK CONVERTER Choose the compensation resistor to set the desired crossover frequency. Determine the value by the following equation: 2π × C2 × f C VOUT R3 = × G EA × G CS VFB Where GEA is the EA transconductance (450µA/V) and fC is the desired crossover frequency (preferably 33KHz). Choose the compensation capacitor to set the zero below one fourth of the crossover frequency. Determine the value by the following equation: C3 > C4 = π × C2 × R ESR × f SW > 1 where RESR is the equivalent series resistance of the output capacitor. capcacitor is C2 × R ESR(max) R3 Where RESR(MAX) is the maximum ESR of the output capacitor. External Boost Diode It is recommended that an external boost diode be added to help improve the regulator efficiency. The diode can be a low cost diode such as an IN4148 or BAT54. 5V 2 π × R3 × f C Determine if the second compensation capacitor, C4 is required. It is required if the ESR zero of the output capacitor happens at less than half of the switching frequency or: MP2108 Rev 1.1 10/2/2006 The second compensation determined by the equation: BST 1 10nF MP2108 LX BOOST DIODE 3 Figure 3—External Boost Diode This diode is also recommended for high duty cycle operation (when VOUT >65%). VIN www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 10 MP2108 – 2A, 6V, 740KHz SYNCHRONOUS BUCK CONVERTER PACKAGE INFORMATION MSOP10 MP2108 Rev 1.1 10/2/2006 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 11 MP2108 – 2A, 6V, 740KHz SYNCHRONOUS BUCK CONVERTER QFN10 (3mm x 3mm) NOTICE: The information in this document is subject to change without notice. Users should warrant and guarantee that third party Intellectual Property rights are not infringed upon when integrating MPS products into any application. MPS will not assume any legal responsibility for any said applications. MP2108 Rev. 1.1 10/2/2006 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 12