Order this document by MC12179/D The MC12179 is a monolithic Bipolar synthesizer integrating the high frequency prescaler, phase/frequency detector, charge pump, and reference oscillator/buffer functions. When combined with an external loop filter and VCO, the MC12179 serves as a complete PLL subsystem. Motorola’s advanced MOSAIC V technology is utilized for low power operation at a 5.0 V supply voltage. The device is designed for operation up to 2.8 GHz for high frequency applications such as CATV down converters and satellite receiver tuners. • • • • • • • • • 500 – 2800 MHz SINGLE CHANNEL FREQUENCY SYNTHESIZER SEMICONDUCTOR TECHNICAL DATA 2.8 GHz Maximum Operating Frequency Low Power Supply Current of 3.5 mA Typical, Including ICC and IP Currents Supply Voltage of 5.0 V Typical Integrated Divide by 256 Prescaler On–Chip Reference Oscillator/Buffer – 2.0 to 11 MHz Operation When Driven From Reference Source – 5.0 to 11 MHz Operation When Used With a Crystal Digital Phase/Frequency Detector with Linear Transfer Function 8 1 Balanced Charge Pump Output D SUFFIX PLASTIC PACKAGE CASE 751 (SO–8) Space Efficient 8–Lead SOIC Operating Temperature Range of –40 to 85°C For additional information on calculating the loop filter components, an InterActiveApNote document containing software (based on a Microsoft Excel spreadsheet) and an Application Note is available. Please order DK306/D from the Motorola Literature Distribution Center. PIN CONNECTIONS MOSAIC V, Mfax and InterActiveApNote are trademarks of Motorola, Inc. MAXIMUM RATINGS (Note 1) Parameter Symbol Value Unit Power Supply Voltage, Pin 2 VCC –0.5 to 6.0 Vdc Power Supply Voltage, Pin 7 VP VCC to 6.0 Vdc Storage Temperature Range Tstg –65 to 150 °C NOTES: 1. Maximum Ratings are those values beyond which damage to the device may occur. Functional operation should be restricted to the Recommended Operating Conditions as identified in the Electrical Characteristics table. 2. ESD data available upon request. OSCin 1 8 OSCout VCC 2 7 VP Gnd 3 6 PDout Fin 4 5 GndP Block Diagram OSCin Crystal Oscillator OSCout Fin fr Phase/Frequency Detector Prescaler ÷256 (Top View) Charge Pump PDout ORDERING INFORMATION fv Device Operating Temperature Range Package MC12179D TA = –40° to +85°C SO–8 Motorola, Inc. 1997 Rev 3 MC12179 ELECTRICAL CHARACTERISTICS (VCC = 4.5 to 5.5 V; VP = VCC to 5.5 V; TA = –40 to 85°C, unless otherwise noted.) Characteristic Supply Current for VCC Supply Current for VP Operating Frequency fINmax fINmin Operating Frequency Crystal Mode External Oscillator OSCin Input Sensitivity Fin Input Sensitivity External Oscillator OSCin Symbol Min Typ Max Unit ICC – 3.1 5.6 mA Note 1 Condition IP – 0.4 1.3 mA Note 1 FIN 2800 – – – – 500 MHz Note 2 FOSC 5 2 – – 11 11 MHz Note 3 Note 4 VIN 200 – 1000 mVP–P Note 2 Note 4 VOSC 500 – 2200 mVP–P Output Source Current5 (PDout) IOH –2.8 –2.2 –1.6 mA VP = 4.5 V, VPDout = VP/2 Output Sink Current5 (PDout) IOL 1.6 2.2 2.8 mA VP = 4.5 V, VPDout = VP/2 Output Leakage Current (PDout) IOZ – 0.5 15 nA VP = 5.0 V, VPDout = VP/2 NOTES: 1. VCC and VP = 5.5 V; FIN = 2.56 GHz; FOSC = 10 MHz crystal; PDout open. 2. AC coupling, FIN measured with a 1000 pF capacitor. 3. Assumes C1 and C2 (Figure 1) limited to ≤30 pF each including stray and parasitic capacitances. 4. AC coupling to OSCin. 5. Refer to Figure 15 and Figure 16 for typical performance curves over temperature and power supply voltage. PIN FUNCTION DESCRIPTION 2 Pin Symbol I/O Function 1 OSCin I Oscillator Input — An external parallel–resonant, fundamental crystal is connected between OSCin and OSCout to form an internal reference oscillator (crystal mode). External capacitors C1 and C2, as shown in Figure 1, are required to set the proper crystal load capacitance and oscillator frequency. For an external reference oscillator, an external signal is AC–coupled to the OSCin pin with a 1000 pF coupling capacitor, with no connection to OSCout. In either mode, a resistor with a nominal value of 50 kΩ MUST be placed across the OSCin and OSCout pins for proper operation. 2 VCC — Positive Power Supply. Bypass capacitors should be placed as close as possible to the pin and be connected directly to the ground plane. 3 Gnd — Ground. 4 Fin I 5 GndP — Ground — For charge pump circuitry. 6 PDout O Single ended phase/frequency detector output (charge pump output). Three–state current sink/source output for use as a loop error signal when combined with an external low pass filter. The phase/frequency detector is characterized by a linear transfer function. 7 VP — Positive power supply for charge pump. VP MUST be equal or greater than VCC. Bypass capacitors should be placed as close as possible to the pin and be connected directly to the ground plane. 8 OSCout O Oscillator output, for use with an external crystal as shown in Figure 1. Prescaler Input — The VCO signal is AC coupled into the Fin pin. MOTOROLA RF/IF DEVICE DATA MC12179 Figure 1. MC12179 Expanded Block Diagram +5.0 V C1 +5.0 V 2 VCC 1 OSCin 8 OSCout C2 NOTE: External 50 kΩ resistor across Pins 1 and 8 is necessary in either crystal or driven mode. 4 VCO 1000 pF VP Crystal Oscillator fr Phase/Frequency Detector 6 Charge Pump To Loop Filter PDout fv Fin 7 Prescaler ÷256 GND 3 GNDP 5 PHASE CHARACTERISTICS The phase comparator in the MC12179 is a high speed digital phase/frequency detector circuit. The circuit determines the “lead” or “lag” phase relationship and time difference between the leading edges of the VCO (fv) signal and the reference (fr) input. The detector can cover a range of ±2π radian of fv/fr phase difference. The operation of the charge pump output is shown in Figure 2. fr lags fv in phase OR fv>fr in frequency When the phase of fr lags that of fv or the frequency of fv is greater than fr, the Do output will sink current. The pulse width will be determined by the time difference between the two rising edges. fr leads fv in phase OR fv<fr in frequency When the phase of fr leads that of fv or the frequency of fv is less than fr, the Do output will source current. The pulse width will be determined by the time difference between the two rising edges. fr = fv in phase and frequency When the phase and frequency of fr and fv are equal, the charge pump will be in a quiet state, except for current spikes when signals are in phase. This situation indicates that the loop is in lock and the phase comparator will maintain the loop in its locked state. Figure 2. Phase/Frequency Detector and Charge Pump Waveforms H fr (OSCin) L H fv (Fin ÷256) L Sourcing Current Pulse Z Sinking Current Pulse PDout H = High voltage level; L = Low voltage level; Z = High impedance NOTES: Phase difference detection range: ∼ –2π to 2π Kp–Charge Pump Gain MOTOROLA RF/IF DEVICE DATA mA [ |Isource4| p) |Isink| + |2.2| )4p|–2.2| + p1.1radian 3 MC12179 APPLICATIONS INFORMATION The MC12179 is intended for applications where a fixed local oscillator is required to be synthesized. The prescaler on the MC12179 operates up to 2.8GHz which makes the part ideal for many satellite receiver applications as well as applications in the 2nd ISM (Industrial, Scientific, and Medical) band which covers the frequency range of 2400MHz to 2483MHz. The part is also intended for MMDS (Multi–channel Multi–point Distribution System) block downconverter applications. Below is a typical block diagram of the complete PLL. Figure 3. Typical Block Diagram of Complete PLL MC12179 PLL External Ref 10.0 MHz VCO φ/Freq Det Charge Pump Loop Filter 2560.00 MHz Since the MC12179 is realized with an all–bipolar ECL style design, the internal oscillator circuitry is different from more traditional CMOS oscillator designs which realize the crystal oscillator with a modified inverter topology. These CMOS designs typically excite the crystal with a rail–to–rail signal which may overdrive the crystal resulting in damage or unstable operation. The MC12179 design does not exhibit these phenomena because the swing out of the OSCout pin is less than 600mV. This has the added advantage of minimizing EMI and switching noise which can be generated by rail–to–rail CMOS outputs. The OSCout output should not be used to drive other circuitry. The oscillator buffer in the MC12179 is a single stage, high speed, differential input/output amplifier; it may be considered to be a form of the Pierce oscillator. A simplified circuit diagram is seen in Figure 4. Figure 4. Simplified Crystal Oscillator/Buffer Circuit ÷P 256 As can be seen from the block diagram, with the addition of a VCO, a loop filter, and either an external oscillator or crystal, a complete PLL sub–system can be realized. Since most of the PLL function is integrated into the MC12179, the user’s primary focus is on the loop filter design and the crystal reference circuit. Figure 13 and Figure 14 illustrate typical VCO spectrum and phase noise characteristics. Figure 17 and Figure 18 illustrate the typical input impedance versus frequency for the prescaler input. Crystal Oscillator Design The MC12179 is used as a multiply–by–256 PLL circuit which transfers the high stability characteristic of a low frequency reference source to the high frequency VCO in the PLL loop. To facilitate this, the device contains an input circuit which can be configured as a crystal oscillator or a buffer for accepting an external signal source. In the external reference mode, the reference source is AC–coupled into the OSCin input pin. The input level signal should be between 500–2200 mVpp. When configured with an external reference, the device can operate with input frequencies down to 2MHz, thus allowing the circuit to control the VCO down to 512 MHz. To optimize the phase noise of the PLL when used in this mode, the input signal amplitude should be closer to the upper specification limit. This maximizes the slew rate of the input signal as it switches against the internal voltage reference. In the crystal mode, an external parallel–resonant fundamental mode crystal is connected between the OSCin and OSCout pins. This crystal must be between 5.0 MHz and 11 MHz. External capacitors, C1 and C2 as shown in Figure 1, are required to set the proper crystal load capacitance and oscillator frequency. The values of the capacitors are dependent on the crystal chosen and the input capacitance of the device and any stray board capacitance. In either mode, a 50kΩ resistor must be connected between the OSCin and the OSCout pins for proper device operation. The value of this resistor is not critical so a 47kΩ or 51kΩ ±10% resistor is acceptable. 4 VCC OSCout OSCin To Phase/ Frequency Detector Bias Source OSCin drives the base of one input of an NPN transistor differential pair. The non–inverting input of the differential pair is internally biased. OSCout is the inverted input signal and is buffered by an emitter follower with a 70 µA pull–down current and has a voltage swing of about 600 mVpp. Open loop output impedance is about 425Ω. The opposite side of the differential amplifier output is used internally to drive another buffer stage which drives the phase/frequency detector. With the 50 kΩ feedback resistor in place, OSCin and OSCout are biased to approximately 1.1V below VCC. The amplifier has a voltage gain of about 15 dB and a bandwidth in excess of 150 MHz. Adherence to good RF design and layout techniques, including power supply pin decoupling, is strongly recommended. A typical crystal oscillator application is shown in Figure 1. The crystal and the feedback resistor are connected directly between OSCin and OSCout, while the loading capacitors, C1 and C2, are connected between OSCin and ground, and OSCout and ground respectively. It is important to understand that as far as the crystal is concerned, the two loading capacitors are in series (albeit through ground). So when the crystal specification defines a specific loading capacitance, this refers to the total external (to the crystal) capacitance seen across its two pins. This capacitance consists of the capacitance contributed by the amplifier (IC and packaging), layout capacitance, and the series combination of the two loading capacitors. This is illustrated in the equation below: MOTOROLA RF/IF DEVICE DATA MC12179 CI C2 + CAMP ) CSTRAY ) C1 C1 ) C2 Provided the crystal and associated components are located immediately next to the IC, thus minimizing the stray capacitance, the combined value of CAMP and CSTRAY is approximately 5pF. Note that the location of the OSCin and OSCout pins at the end of the package, facilitates placing the crystal, resistor and the C1 and C2 capacitors very close to the device. Usually, one of the capacitors is in parallel with an adjustable capacitor used to trim the frequency of oscillation. It is important that the total external (to the IC) capacitance seen by either OSCin or OSCout, be no greater than 30pF. In operation, the crystal oscillator will start up with the application of power. If the crystal is in a can that is not grounded it is often possible to monitor the frequency of oscillation by connecting an oscilloscope probe to the can; this technique minimizes any disturbance to the circuit. If a malfunction is indicated, a high impedance, low capacitance, FET probe may be connected to either OSCin or OSCout. Signals typically seen at those points will be very nearly sinusoidal with amplitudes of roughly 300 to 600 mVpp. Some distortion is inevitable and has little bearing on the accuracy of the signal going to the phase detector. Loop Filter Design Because the device is designed for a non–frequency agile synthesizer (i.e., how fast it tunes is not critical) the loop filter design is very straight forward. The current output of the charge pump allows the loop filter to be realized without the need of any active components. The preferred topology for the filter is illustrated below in Figure 5. Figure 5. Loop Filter Xtl Osc Ph/Frq Det Chrg Pump Kp ÷256 N VCO Ro Co Rx Kv Ca Cx MC12179 The Ro/Co components realize the primary loop filter. Ca is added to the loop filter to provide for reference sideband suppression. If additional suppression is needed, the Rx/Cx realizes an additional filter. In most applications, this will not be necessary. If all components are used, this results in a 4th order PLL, which makes analysis difficult. To simplify this, the loop design will be treated as a 2nd order loop (Ro/Co) and additional guidelines are provided to minimize the influence of the other components. If more rigorous analysis is needed, mathematical/system simulation tools can be used. MOTOROLA RF/IF DEVICE DATA Component Guideline Ca <0.1 × Co Rx >10 × Ro Cx <0.1 × Co The focus of the design effort is to determine what the loop’s natural frequency, ωo, should be. This is determined by Ro, Co, Kp, Kv, and N. Because Kp, Kv, and N are given, it is only necessary to calculate values for Ro and Co. There are 3 considerations in selecting the loop bandwidth: 1) Maximum loop bandwidth for minimum tuning speed 2) Optimum loop bandwidth for best phase noise performance 3) Minimum loop bandwidth for greatest reference sideband suppression Usually a compromise is struck between these 3 cases, however, for the fixed frequency application, minimizing the tuning speed is not a critical parameter. To specify the loop bandwidth for optimal phase noise performance, an understanding of the sources of phase noise in the system and the effect of the loop filter on them is required. There are 3 major sources of phase noise in the phase–locked loop – the crystal reference, the VCO, and the loop contribution. The loop filter acts as a low–pass filter to the crystal reference and the loop contribution equal to the total divide–by–N ratio. This is mathematically described in Figure 10. The loop filter acts as a high–pass filter to the VCO with an in–band gain equal to unity. This is described in Figure 11. The loop contribution includes the PLL IC, as well as noise in the system; supply noise, switching noise, etc. For this example, a loop contribution of 15 dB has been selected, which corresponds to data in Figure 14. The crystal reference and the VCO are characterized as high–order 1/f noise sources. Graphical analysis is used to determine the optimum loop bandwidth. It is necessary to have noise plots from the manufacturer. This method provides a straightforward approximation suitable for quickly estimating the optimal bandwidth. The loop contribution is characterized as white–noise or low–order 1/f noise given in the form of a noise factor which combines all the noise effects into a single value. The phase noise of the Crystal Reference is increased by the noise factor of the PLL IC and related circuitry. It is further increased by the total divide–by–N ratio of the loop. This is illustrated in Figure 6. The point at which the VCO phase noise crosses the amplified phase noise of the Crystal Reference is the point of the optimum loop bandwidth. In the example of Figure 6, the optimum bandwidth is approximately 15 KHz. 5 MC12179 Figure 6. Graphical Analysis of Optimum Bandwidth –60 15kHz/2.5 or 6kHz (37.7krads) with a damping coefficient, ζ ≈ 1. T(s) is the transfer function of the loop filter. Optimum Bandwidth –70 Figure 8. Design Equations for the 2nd Order System –80 VCO dB –90 T(s) –100 20*log(256) –110 –120 –130 Crystal Reference –150 10 100 1k 10k 100k 1M Hz Figure 7. Closed Loop Frequency Response for ζ = 1 Natural Frequency 10 3dB Bandwidth 0 –10 dB –20 –30 –40 –50 –60 0.1 1 10 Hz 100 ǒ Ǔ NCo K pK v 1k To simplify analysis further a damping factor of 1 will be selected. The normalized closed loop response is illustrated in Figure 7 where the loop bandwidth is 2.5 times the loop natural frequency (the loop natural frequency is the frequency at which the loop would oscillate if it were unstable). Therefore the optimum loop bandwidth is s2 )1 ) RoCos ) 1 ǒǓ) + ǒ Ǔ )ǒ Ǔ ) 2z wo s 1 2 w o2 s 1 2z wo s 1 ǒ Ǔ+ǒ Ǔ³ +Ǹ ³ [ǒ Ǔ Ǔ³ +ǒ Ǔ +ǒ Ǔ³ +ǒ NCo KpKv 15dB NF of the Noise Contribution from Loop –140 + RoCos RoCo 1 wo2 2z wo wo z Kp Kv NCo woRoCo 2 Co Ro KpKv Nwo2 2z woCo In summary, follow the steps given below: Step 1: Plot the phase noise of crystal reference and the VCO on the same graph. Step 2: Increase the phase noise of the crystal reference by the noise contribution of the loop. Step 3: Convert the divide–by–N to dB (20log 256 – 48 dB) and increase the phase noise of the crystal reference by that amount. Step 4: The point at which the VCO phase noise crosses the amplified phase noise of the Crystal Reference is the point of the optimum loop bandwidth. This is approximately 15 kHz in Figure 6. Step 5: Correlate this loop bandwidth to the loop natural frequency and select components per Figure 8. In this case the 3.0 dB bandwidth for a damping coefficient of 1 is 2.5 times the loop’s natural frequency. The relationship between the 3.0 dB loop bandwidth and the loop’s “natural” frequency will vary for different values of ζ. Making use of the equations defined above in a math tool or spread sheet is useful. To aid in the use of such a tool the equations are summarized in Figures 9 through 11. Figure 9. Loop Parameter Relations Let: NCo KpKv Let: Ca + w1 2 o + aCo , , RoCo Cx + w2zo + bCo , A + 1 ) a , and B +1)a)b + w13 , RxCx + w14 , Ro(Ca ) Cx) + w15 K3w3 + wo , K4w4 + wo , K5w5 + wo Let: RoCo Let: 6 MOTOROLA RF/IF DEVICE DATA MC12179 Figure 10. Transfer Function for the Crystal Noise in the Frequency Plane T(jw) +N@ ǒ ) j ǒ 2z ww Ǔ * B ww Ǔ ) j ǒ 2z ww * (AK4 ) K5) ww 1 1 ) 4 K3K4 w 4 wo o 2 o o2 3 o3 Ǔ Figure 11. Transfer Function for the VCO Noise in the Frequency Plane T(jw) + ǒ ǒ 1 * B ww Ǔ * j ǒ (AK4 ) K5) ww Ǔ * B ww Ǔ ) j ǒ 2z ww * (AK4 ) K5) ww 4 K3K4 w 4 ) K3K4 ww wo 4 o4 2 3 o2 o3 2 o o2 3 o3 Ǔ overall transfer function of the loop filter. To use these equations in determining the overall transfer function of a PLL multiply the filter’s impedance by the gain constant of the phase detector then multiply that by the filter’s transfer function (which is unity in the 2nd and 3rd order cases below). Appendix: Derivation of Loop Filter Transfer Function The purpose of the loop filter is to convert the current from the phase detector to a tuning voltage for the VCO. The total transfer function is derived in two steps. Step 1 is to find the voltage generated by the impedance of the loop filter. Step 2 is to find the transfer function from the input of the loop filter to its output. The “voltage” times the “transfer function” is the Figure 12. Overall Transfer Function of the PLL For the 2nd Order PLL: Vp Vt Ro ZLF(s) Co TLF(s) For the 3rd Order PLL: Vp Vt Ro Ca Co ZLF(s) + VVpt(s) +1 , (s) Vp(s) + Kp(s)ZLF(s) + C R CRso2C)os(C) 1) C )s o o a TLF(s) For the 4th Order PLL: + RoCCooss) 1 + VVpt(s) +1 , (s) Vp o Vp(s) a + Kp(s)ZLF(s) Vt Ro Ca Rx Cx Co ZLF(s) + C R C R C s3 ) [ (C ) (RC o)RCosC))1)C(RRxC(Cxs ))1)C ) ] s2 ) (C ) C ) C )s o o a x x o a x x o o x a o a x TLF(s) + VVpt(s) + (RxCx1s ) 1) (s) MOTOROLA RF/IF DEVICE DATA , Vp(s) + Kp(s)ZLF(s) 7 MC12179 Figure 13. VCO Output Spectrum with MC12179, VCC = 5.0 V (ECLiPTEK 8.9 MHz Crystal and ZCOM 2500 VCO) NOTE: Spurs can be reduced further by narrowing the loop bandwidth of the PLL loop filter and/or adding an extra filter (Rx/Cx) Figure 14. Typical Phase Noise Plot, 2200 MHz VCO (With the MC12179 in a Closed Loop) HP 3048A CARRIER 2200MHz 0 –25 dBc/Hz –50 –75 –100 –125 –150 –170 1k 10k 100k 1M 10M 40M L(f) [dBc/Hz] vs f[Hz] 8 MOTOROLA RF/IF DEVICE DATA MC12179 Figure 15. Typical Charge Pump Current versus Temperature (VCC = Vpp = 5.0 V) 2.5 2.0 SINK 1.5 –40°C +25°C +85°C Sink/Source Current (mA) 1.0 0.5 0.0 –0.5 –1.0 –1.5 SOURCE –2.0 –2.5 0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 Voltage at PDout (V) Figure 16. Typical Charge Pump Current versus Voltage (T = 25°C) 2.5 2.0 SINK 1.5 4.5V VCC/VPP 5.0V VCC/VPP 5.5V VCC/VPP Sink/Source Current (mA) 1.0 0.5 0.0 –0.5 –1.0 –1.5 SOURCE –2.0 –2.5 0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 Voltage at PDout (V) MOTOROLA RF/IF DEVICE DATA 9 MC12179 Figure 17. Typical Real Input Impedance versus Input Frequency (For the Fin Input) 100 80 R (Ohms) 60 40 20 0 250 500 750 1000 1250 1500 1750 2000 2250 2500 2750 Frequency (MHz) Figure 18. Typical Imaginary Input Impedance versus Input Frequency (For the Fin Input) 50 25 0 –25 jX (Ohms) –50 –75 –100 –125 –150 –175 –200 –225 –250 250 500 750 1000 1250 1500 1750 2000 2250 2500 2750 Frequency (MHz) 10 MOTOROLA RF/IF DEVICE DATA MC12179 OUTLINE DIMENSIONS D SUFFIX PLASTIC PACKAGE CASE 751-06 (SO–8) ISSUE T D A 8 NOTES: 1. DIMENSIONING AND TOLERANCING PER ASME Y14.5M, 1994. 2. DIMENSIONS ARE IN MILLIMETER. 3. DIMENSION D AND E DO NOT INCLUDE MOLD PROTRUSION. 4. MAXIMUM MOLD PROTRUSION 0.15 PER SIDE. 5. DIMENSION B DOES NOT INCLUDE DAMBAR PROTRUSION. ALLOWABLE DAMBAR PROTRUSION SHALL BE 0.127 TOTAL IN EXCESS OF THE B DIMENSION AT MAXIMUM MATERIAL CONDITION. C 5 0.25 H E M B M 1 4 h B X 45 _ e q A C SEATING PLANE L 0.10 A1 B 0.25 M C B S A S DIM A A1 B C D E e H h L q MILLIMETERS MIN MAX 1.35 1.75 0.10 0.25 0.35 0.49 0.19 0.25 4.80 5.00 3.80 4.00 1.27 BSC 5.80 6.20 0.25 0.50 0.40 1.25 0_ 7_ Motorola reserves the right to make changes without further notice to any products herein. Motorola makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does Motorola assume any liability arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation consequential or incidental damages. “Typical” parameters which may be provided in Motorola data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. Motorola does not convey any license under its patent rights nor the rights of others. Motorola products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications intended to support or sustain life, or for any other application in which the failure of the Motorola product could create a situation where personal injury or death may occur. Should Buyer purchase or use Motorola products for any such unintended or unauthorized application, Buyer shall indemnify and hold Motorola and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that Motorola was negligent regarding the design or manufacture of the part. Motorola and are registered trademarks of Motorola, Inc. Motorola, Inc. is an Equal Opportunity/Affirmative Action Employer. Mfax is a trademark of Motorola, Inc. How to reach us: USA / EUROPE / Locations Not Listed: Motorola Literature Distribution; P.O. Box 5405, Denver, Colorado 80217. 1–303–675–2140 or 1–800–441–2447 JAPAN: Nippon Motorola Ltd.: SPD, Strategic Planning Office, 141, 4–32–1 Nishi–Gotanda, Shagawa–ku, Tokyo, Japan. 03–5487–8488 Customer Focus Center: 1–800–521–6274 Mfax: [email protected] – TOUCHTONE 1–602–244–6609 ASIA/PACIFIC: Motorola Semiconductors H.K. Ltd.; 8B Tai Ping Industrial Park, Motorola Fax Back System – US & Canada ONLY 1–800–774–1848 51 Ting Kok Road, Tai Po, N.T., Hong Kong. 852–26629298 – http://sps.motorola.com/mfax/ HOME PAGE: http://motorola.com/sps/ MOTOROLA RF/IF DEVICE DATA◊ MC12179/D 11