Dual Bipolar/JFET, Audio Operational Amplifier OP275* FEATURES Excellent Sonic Characteristics Low Noise: 6 nV/Hz Low Distortion: 0.0006% High Slew Rate: 22 V/s Wide Bandwidth: 9 MHz Low Supply Current: 5 mA Low Offset Voltage: 1 mV Low Offset Current: 2 nA Unity Gain Stable SOIC-8 Package PDIP-8 Package PIN CONNECTIONS 8-Lead Narrow-Body SOIC (S Suffix) OUT A 1 8 V+ –IN A 2 7 OUT B 6 –IN B +IN A 3 V– 4 OP275 5 +IN B 8-Lead PDIP (P Suffix) OUT A 1 –IN A +IN A V– OP275 8 V+ 2 7 OUT B 3 6 –IN B 4 5 +IN B APPLICATIONS High Performance Audio Active Filters Fast Amplifiers Integrators GENERAL DESCRIPTION The OP275 is the first amplifier to feature the Butler Amplifier front end. This new front end design combines both bipolar and JFET transistors to attain amplifiers with the accuracy and low noise performance of bipolar transistors, and the speed and sound quality of JFETs. Total Harmonic Distortion plus Noise equals that of previous audio amplifiers, but at much lower supply currents. A very low l/f corner of below 6 Hz maintains a flat noise density response. Whether noise is measured at either 30 Hz or 1 kHz, it is only 6 nVHz. The JFET portion of the input stage gives the OP275 its high slew rates to keep distortion low, even when large output swings are required, and the 22 V/µs slew rate of the OP275 is the fastest of any standard audio amplifier. Best of all, this low noise and high speed are accomplished using less than 5 mA of supply current, lower than any standard audio amplifier. Improved dc performance is also provided with bias and offset currents greatly reduced over purely bipolar designs. Input offset voltage is guaranteed at 1 mV and is typically less than 200 µV. This allows the OP275 to be used in many dc-coupled or summing applications without the need for special selections or the added noise of additional offset adjustment circuitry. The output is capable of driving 600 loads to 10 V rms while maintaining low distortion. THD + Noise at 3 V rms is a low 0.0006%. The OP275 is specified over the extended industrial (–40°C to +85°C) temperature range. OP275s are available in both plastic DIP and SOIC-8 packages. SOIC-8 packages are available in 2500-piece reels. Many audio amplifiers are not offered in SOIC-8 surface-mount packages for a variety of reasons; however, the OP275 was designed so that it would offer full performance in surface-mount packaging. *Protected by U.S. Patent No. 5,101,126. REV. C Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781/329-4700 www.analog.com Fax: 781/326-8703 © 2004 Analog Devices, Inc. All rights reserved. OP275–SPECIFICATIONS ELECTRICAL CHARACTERISTICS (@ V = 15.0 V, T = 25C, unless otherwise noted.) S Parameter Symbol AUDIO PERFORMANCE THD + Noise Voltage Noise Density en Current Noise Density Headroom in INPUT CHARACTERISTICS Offset Voltage VOS Input Bias Current IB Input Offset Current IOS Input Voltage Range Common-Mode Rejection Ratio VCM CMRR Large Signal Voltage Gain AVO Offset Voltage Drift VOS/T A Conditions Min VIN = 3 V rms, RL = 2 k, f = 1 kHz f = 30 Hz f = 1 kHz f = 1 kHz THD + Noise 0.01%, RL = 2 k, VS = ±18 V –40°C TA +85°C VCM = 0 V VCM = 0 V, –40°C TA +85°C VCM = 0 V VCM = 0 V, –40°C TA +85°C VCM = ±10.5 V, –40°C TA +85°C RL = 2 k RL = 2 k, –40°C TA +85°C RL = 600 –10.5 Typ % nVHz nVHz pAHz >12.9 dBu 100 100 2 2 80 250 175 106 RL = 2 k RL = 2 k, –40°C TA +85°C RL = 600 , VS = ±18 V –13.5 –13 ±13.9 ±13.9 +14, –16 POWER SUPPLY Power Supply Rejection Ratio PSRR VS = ±4.5 V to ±18 V VS = ±4.5 V to ±18 V, –40°C TA +85°C VS = ±4.5 V to ±18 V, VO = 0 V, RL = , –40°C TA +85°C VS = ±22 V, VO = 0 V, RL = , –40°C TA +85°C 85 111 RL = 2 k 15 Supply Voltage Range VS DYNAMIC PERFORMANCE Slew Rate Full-Power Bandwidth Gain Bandwidth Product Phase Margin Overshoot Factor SR BWP GBP Øm VIN = 100 mV, AV = +1, RL = 600 , CL = 100 pF 1 1.25 350 400 50 100 +10.5 mV mV nA nA nA nA V dB V/mV V/mV V/mV µV/°C 200 2 VO ISY Unit 0.006 7 6 1.5 OUTPUT CHARACTERISTICS Output Voltage Swing Supply Current Max +13.5 +13 V V V dB 80 dB 4 ±4.5 22 5 mA 5.5 ±22 mA V 9 62 V/µs kHz MHz Degrees 10 % Specifications subject to change without notice. –2– REV. C OP275 ABSOLUTE MAXIMUM RATINGS1 Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±22 V Input Voltage2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±22 V Differential Input Voltage2 . . . . . . . . . . . . . . . . . . . . . . . ±7.5 V Output Short-Circuit Duration to GND3 . . . . . . . . . . Indefinite Storage Temperature Range P, S Packages . . . . . . . . . . . . . . . . . . . . . . . –65°C to +150°C Operating Temperature Range OP275G . . . . . . . . . . . . . . . . . . . . . . . . . . . . –40°C to +85°C Junction Temperature Range P, S Packages . . . . . . . . . . . . . . . . . . . . . . . –65°C to +150°C Lead Temperature Range (Soldering, 60 sec) . . . . . . . . . .300°C Package Type JA4 JC Unit 8-Lead Plastic DIP (P) 8-Lead SOIC (S) 103 158 43 43 °C/W °C/W NOTES 1 Absolute maximum ratings apply to packaged parts, unless otherwise noted. 2 For supply voltages greater than ±22 V, the absolute maximum input voltage is equal to the supply voltage. 3 Shorts to either supply may destroy the device. See data sheet for full details. 4 JA is specified for the worst-case conditions, i.e., JA is specified for device in socket for PDIP packages; JA is specified for device soldered in circuit board for SOIC packages. ORDERING GUIDE Model Temperature Range Package Description Package Option OP275GP OP275GS OP275GS-REEL OP275GS-REEL7 OP275GSZ* OP275GSZ-REEL* OP275GSZ-REEL7* –40°C to +85°C –40°C to +85°C –40°C to +85°C –40°C to +85°C –40°C to +85°C –40°C to +85°C –40°C to +85°C 8-Lead PDIP 8-Lead SOIC 8-Lead SOIC 8-Lead SOIC 8-Lead SOIC 8-Lead SOIC 8-Lead SOIC N-8 R-8 R-8 R-8 R-8 R-8 R-8 *Z = Pb-free part. CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000V readily accumulate on the human body and test equipment and can discharge without detection. Although the OP275 features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality. REV. C –3– OP275–Typical Performance Characteristics 25 1500 5 0 –5 –10 –VOM –15 1000 500 +GAIN RL = 600 5 10 15 20 SUPPLY VOLTAGE – V 25 135 20 90 MARKER 15 309.058Hz PHASE (A/R) 90.606Deg 45 0 0 –10 –45 –20 –90 10k 100k 1M PHASE – Degrees 30 CLOSED-LOOP GAIN – dB 40 40 10 80 60 40 20 1M TPC 7. Common-Mode Rejection vs. Frequency –180 10M 100k 20 VS = 15V TA = 25C AVCL = +100 10M VS = 15V TA = 25C 50 AVCL = +10 10 0 1M TPC 3. Closed-Loop Gain and Phase, AV = +1 AVCL = +1 AVCL = +1 40 AVCL = +10 30 AVCL = +100 20 –10 10 10k 100k 1M 10M FREQUENCY – Hz 0 100 100M 80 GAIN +PSRR 80 VS = 15V TA = 25C –PSRR 40 20 0 10 1M 10M 100 100 60 10k 100k FREQUENCY – Hz 1k TPC 6. Closed-Loop Output Impedance vs. Frequency OPEN-LOOP GAIN – dB POWER SUPPLY REJECTION – dB VS = 15V TA = 25C 10k 100k FREQUENCY – Hz –40 FREQUENCY – Hz 120 1k –135 TPC 5. Closed-Loop Gain vs. Frequency 120 0 100 –90 –30 30 –30 1k TPC 4. Open-Loop Gain, Phase vs. Frequency 100 –20 10k 100 –20 10M 0 –45 60 FREQUENCY – Hz COMMON-MODE REJECTION – dB 75 50 VS = 15V TA = 25C 50 0 25 50 TEMPERATURE – C 45 TPC 2. Open-Loop Gain vs. Temperature MARKER 15 309.059Hz MAG (A/H) 60.115dB 60 –25 90 10 –10 IMPEDANCE – 0 135 20 0 –GAIN RL = 600 250 0 –50 TPC 1. Output Voltage Swing vs. Supply Voltage GAIN – dB –GAIN RL = 2k 750 180 VS = 15V TA = 25C 30 +GAIN RL = 2k –20 –25 40 100 1k 10k 100k FREQUENCY – Hz 1M TPC 8. Power Supply Rejection vs. Frequency –4– 60 40 PHASE VS = 15V RL = 2k TA = 25C 0 45 Ø m = 58 90 20 135 0 180 –20 225 –40 270 –60 1k 10k 100k 1M 10M FREQUENCY – Hz PHASE – Degrees +VOM 10 VS = 15V VO = 15V PHASE – Degrees 1250 OPEN-LOOP GAIN – V/mV OUTPUT VOLTAGE SWING – V 15 GAIN – dB TA = 25C RL = 2k 20 100M TPC 9. Open-Loop Gain, Phase vs. Frequency REV. C OP275 100 16 AVCL = +1 NEGATIVE EDGE 90 60 Øm 9 55 GBW 8 50 80 OVERSHOOT – % 10 70 60 –25 0 25 50 TEMPERATURE – C 50 40 30 VS = 15V RL = 2k VIN = 100mV p-p 20 0 40 100 75 TPC 10. Gain Bandwidth Product, Phase Margin vs. Temperature 200 300 400 100 LOAD CAPACITANCE – pF 10 TA = 25C VS = 15V AVCL = +1 RL = 2k 4.5 TA = +85C TA = +25C 4.0 TA = –40C 3.5 5 0 1k 10k 100k 1M FREQUENCY – Hz 3.0 10M TPC 13. Maximum Output Swing vs. Frequency 10 15 20 SUPPLY VOLTAGE – V 200 150 100 50 –25 0 25 50 TEMPERATURE – C 75 100 TPC 16. Input Bias Current vs. Temperature 8 +VOM 6 4 TA = 25C VS = 15V 2 1k LOAD RESISTANCE – 10k VS = 15V 110 100 90 SINK 80 70 60 50 40 SOURCE 30 20 –50 25 –25 0 25 50 TEMPERATURE – C 75 100 TPC 15. Short-Circuit Current vs. Temperature 500 VS = 15V TA = 25C 4 VS = 15V –40 C to +85 C 400 BASED ON 920 OP AMPS 3 300 UNITS CURRENT NOISE DENSITY – pA/ Hz 250 REV. C 5 5 300 0 –50 0 TPC 14. Supply Current vs. Supply Voltage VS = ±15V 10 TPC 12. Maximum Output Voltage vs. Load Resistance ABSOLUTE OUTPUT CURRENT – mA 15 –VOM 120 25 20 12 0 100 500 5.0 SUPPLY CURRENT – mA MAXIMUM OUTPUT SWING – V 0 14 TPC 11. Small Signal Overshoot vs. Load Capacitance 30 INPUT BIAS CURRENT – nA AVCL = +1 POSITIVE EDGE 10 7 –50 MAXIMUM OUTPUT SWING – V 65 PHASE MARGIN – Degrees GAIN BANDWIDTH PRODUCT – MHz 11 2 1 200 100 10 100 1k FREQUENCY – Hz TPC 17. Current Noise Density vs. Frequency –5– 100k 0 0 1 2 3 4 5 6 7 TCVOS – V/C 8 9 TPC 18. TCVOS Distribution 10 OP275 10 200 160 VS = 15V TA = 25C 50 45 6 +0.1% +0.01% STEP SIZE – V 4 120 UNITS TA = 25C VS = 15V 8 80 2 0 –2 –4 40 –0.01% –0.1% –6 40 SLEW RATE – V/s BASED ON 920 OP AMPS 35 –SR 30 +SR 25 –8 0 –500–400–300–200–100 0 100 200 300 400 500 INPUT OFFSET VOLTAGE – V –10 TPC 19. Input Offset (VOS) Distribution 0 100 200 300 400 CAPACITIVE LOAD – pF 500 TPC 21. Slew Rate vs. Capacitive Load 50 VS = 15V RL = 2k 35 30 45 TA = 25C SLEW RATE – V/s SLEW RATE – V/s 20 100 200 300 400 500 600 700 800 900 SETTLING TIME – ns TPC 20. Step Size vs. Settling Time 40 25 20 15 10 VS = 15V RL = 2k –SR 100 90 40 35 30 +SR 10 0% 25 5 0 0 0 0.2 0.4 0.6 0.8 1.0 20 –50 5V –25 DIFFERENTIAL INPUT VOLTAGE – V TPC 22. Slew Rate vs. Differential Input Voltage 0 25 50 TEMPERATURE – C 75 100 TPC 23. Slew Rate vs. Temperature TPC 24. Negative Slew Rate RL = 2 k , VS = ±15 V, AV = +1 CH A: 80.0 V FS MKR: 6.23 nV/ Hz 100 90 200ns 10.0 V/DIV 100 90 10 10 0% 0% 5V 200ns TPC 25. Positive Slew Rate RL = 2 k , VS = ±15 V, AV = +1 50mV 100ns TPC 26. Small Signal Response RL = 2 k , VS = ±5 V, AV = +1 –6– 0 Hz MKR: 1 000 Hz 2.5 kHz BW: 15.0 MHz TPC 27. Voltage Noise Density vs. Frequency VS = ±15 V REV. C OP275 APPLICATIONS Circuit Protection THD + NOISE – % 0.010 OP275 has been designed with inherent short-circuit protection to ground. An internal 30 resistor, in series with the output, limits the output current at room temperature to ISC+ = 40 mA and ISC– = –90 mA, typically, with ±15 V supplies. However, shorts to either supply may destroy the device when excessive voltages or currents are applied. If it is possible for a user to short an output to a supply for safe operation, the output current of the OP275 should be design-limited to ±30 mA, as shown in Figure 1. VS = 18V RL = 600 0.001 0.0001 0.5 1 10 OUTPUT SWING – V rms Figure 4. Headroom, THD + Noise vs. Output Amplitude (V rms); RLOAD = 600 , VSUP = ±18 V Total Harmonic Distortion Total Harmonic Distortion + Noise (THD + N) of the OP275 is well below 0.001% with any load down to 600 . However, this is dependent upon the peak output swing. In Figure 2, the THD + Noise with 3 V rms output is below 0.001%. In Figure 3, THD + Noise is below 0.001% for the 10 k and 2 k loads but increases to above 0.1% for the 600 load condition. This is a result of the output swing capability of the OP275. Notice the results in Figure 4, showing THD versus VIN (V rms). This figure shows that the THD + Noise remains very low until the output reaches 9.5 V rms. This performance is similar to competitive products. The output of the OP275 is designed to maintain low harmonic distortion while driving 600 loads. However, driving 600 loads with very high output swings results in higher distortion if clipping occurs. A common example of this is in attempting to drive 10 V rms into any load with ±15 V supplies. Clipping will occur and distortion will be very high. To attain low harmonic distortion with large output swings, supply voltages may be increased. Figure 5 shows the performance of the OP275 driving 600 loads with supply voltages varying from ±18 V to ±20 V. Notice that with ±18 V supplies the distortion is fairly high, while with ±20 V supplies it is a very low 0.0007%. RFB 0.0001 FEEDBACK RX 332 – A1 + VOUT 0.001 THD – % A1 = 1/2 OP275 Figure 1. Recommended Output Short-Circuit Protection RL = 600 VOUT = 10V rms @ 1kHz 0.01 0.010 THD + NOISE – % RL = 600, 2k, 10k VS = 15V VIN = 3V rms AV = +1 0.1 0 17 18 19 20 SUPPLY VOLTAGE – V 21 22 0.001 0.0005 Figure 5. THD + Noise vs. Supply Voltage 20 100 1k 10k FREQUENCY – Hz 20k Noise The voltage noise density of the OP275 is below 7 nV/Hz from 30 Hz. This enables low noise designs to have good performance throughout the full audio range. Figure 6 shows a typical OP275 with a 1/f corner at 2.24 Hz. Figure 2. THD + Noise vs. Frequency vs. RLOAD THD + NOISE – % 1 600 0.1 CH A: 80.0V FS MKR: 45.6V/ Hz AV = +1 VS = 18V VIN = 10V rms 80kHz FILTER 0.010 10.0V/DIV 2k 0.001 10k 0.0001 20 100 1k FREQUENCY – Hz 10k 20k Figure 3. THD + Noise vs. RLOAD; VIN =10 V rms 0Hz MKR: 2.24Hz 10Hz BW: 0.145Hz Figure 6. 1/f Noise Corner, VS = ±15 V, AV = 1000 REV. C –7– OP275 Noise Testing prevent phase reversal; however, they will not prevent this effect from occurring in noninverting applications. For these applications, the fix is a simple one and is illustrated in Figure 9. A 3.92 k resistor in series with the noninverting input of the OP275 cures the problem. For audio applications, the noise density is usually the most important noise parameter. For characterization, the OP275 is tested using an Audio Precision, System One. The input signal to the Audio Precision must be amplified enough to measure it accurately. For the OP275, the noise is gained by approximately 1020 using the circuit shown in Figure 7. Any readings on the Audio Precision must then be divided by the gain. In implementing this test fixture, good supply bypassing is essential. RFB* – VOUT + VIN RL RS 3.92k 100 909 *RFB 2k IS OPTIONAL A OP37 OP275 Overload or Overdrive Recovery OUTPUT B 909 100 100 Figure 9. Output Voltage Phase Reversal Fix OP37 909 Overload or overdrive recovery time of an operational amplifier is the time required for the output voltage to recover to a rated output voltage from a saturated condition. This recovery time is important in applications where the amplifier must recover quickly after a large abnormal transient event. The circuit shown in Figure 10 was used to evaluate the OP275’s overload recovery time. The OP275 takes approximately 1.2 ms to recover to VOUT = +10 V and approximately 1.5 µs to recover to VOUT = –10 V. 4.42k 490 Figure 7. Noise Test Fixture Input Overcurrent Protection The maximum input differential voltage that can be applied to the OP275 is determined by a pair of internal Zener diodes connected across its inputs. They limit the maximum differential input voltage to ±7.5 V. This is to prevent emitter-base junction breakdown from occurring in the input stage of the OP275 when very large differential voltages are applied. However, to preserve the OP275’s low input noise voltage, internal resistances in series with the inputs were not used to limit the current in the clamp diodes. In small signal applications, this is not an issue; however, in applications where large differential voltages can be inadvertently applied to the device, large transient currents can flow through these diodes. Although these diodes have been designed to carry a current of ±5 mA, external resistors as shown in Figure 8 should be used in the event that the OP275’s differential voltage were to exceed ±7.5 V. 1.4k 1.4k 2 R2 10k 2 3 VIN 4V p-p @100Hz – A1 1 VOUT + RS 909k RL 2.43k A1 = 1/2 OP275 Figure 10. Overload Recovery Time Test Circuit Measuring Settling Time The design of OP275 combines a high slew rate and a wide gain bandwidth product to produce a fast settling (tS < 1 µs) amplifier for 8- and 12-bit applications. The test circuit designed to measure the settling time of the OP275 is shown in Figure 11. This test method has advantages over false-sum node techniques in that the actual output of the amplifier is measured, instead of an error voltage at the sum node. Common-mode settling effects are exercised in this circuit in addition to the slew rate and bandwidth effects measured by the false-sum node method. Of course, a reasonably flat-top pulse is required as the stimulus. – OP275 3 R1 1k 6 + Figure 8. Input Overcurrent Protection The output waveform of the OP275 under test is clamped by Schottky diodes and buffered by the JFET source follower. The signal is amplified by a factor of 10 by the OP260 and then Schottky-clamped at the output to prevent overloading the oscilloscope’s input amplifier. The OP41 is configured as a fast integrator, which provides overall dc offset nulling. Output Voltage Phase Reversal Since the OP275’s input stage combines bipolar transistors for low noise and p-channel JFETs for high speed performance, the output voltage of the OP275 may exhibit phase reversal if either of its inputs exceeds its negative common-mode input voltage. This might occur in very severe industrial applications where a sensor or system fault might apply very large voltages on the inputs of the OP275. Even though the input voltage range of the OP275 is ±10.5 V, an input voltage of approximately –13.5 V will cause output voltage phase reversal. In inverting amplifier configurations, the OP275’s internal 7.5 V input clamping diodes will High Speed Operation As with most high speed amplifiers, care should be taken with supply decoupling, lead dress, and component placement. Recommended circuit configurations for inverting and noninverting applications are shown in Figures 12 and 13. –8– REV. C OP275 16V–20V – + +15V 0.1F 1k 2N4416 + – V– D1 D2 0.1F 1/2 OP260AJ 1F RF 2k 10k – – – IC2 10k 16V–20V RG 222 2N2222A 5V 1N4148 OUTPUT (TO SCOPE) D4 + V+ D3 DUT + 1k RL + 750 15k SCHOTTKY DIODES D1–D4 ARE HEWLETT-PACKARD HP5082-2835 IC1 IS 1/2 OP260AJ IC2 IS PMI OP41EJ –15V Figure 11. OP275’s Settling Time Test Fixture CFB +15V 10F + 0.1F RFB – 2 3 VIN 8 – 1/2 1 OP275 + 4 0.1F RS VOUT RL Figure 14. Compensating the Feedback Pole Attention to Source Impedances Minimizes Distortion Since the OP275 is a very low distortion amplifier, careful attention should be given to source impedances seen by both inputs. As with many FET-type amplifiers, the p-channel JFETs in the OP275’s input stage exhibit a gate-to-source capacitance that varies with the applied input voltage. In an inverting configuration, the inverting input is held at a virtual ground and, as such, does not vary with input voltage. Thus, since the gate-to-source voltage is constant, there is no distortion due to input capacitance modulation. In noninverting applications, however, the gate-to-source voltage is not constant. The resulting capacitance modulation can cause distortion above 1 kHz if the input impedance is greater than 2 k and unbalanced. Figure 12. Unity Gain Follower +15V 10F + 0.1F 10pF 4.99k 4.99k 3 8 – VOUT 2k –15V 2 CIN + 10F + VIN CS 1/2 OP275 + 4 1 VOUT 2k RG RF 2.49k 0.1F 10F + – VIN Figure 13. Unity Gain Inverter VOUT + * RS = RG//RF IF RG//RF > 2k FOR MINIMUM DISTORTION In inverting and noninverting applications, the feedback resistance forms a pole with the source resistance and capacitance (RS and CS) and the OP275’s input capacitance (CIN), as shown in Figure 14. With RS and RF in the kilohm range, this pole can create excess phase shift and even oscillation. A small capacitor, CFB, in parallel and RFB eliminates this problem. By setting RS (CS + CIN) = RFBCFB, the effect of the feedback pole is completely removed. REV. C OP275 RS* –15V Figure 15. Balanced Input Impedance to Minimize Distortion in Noninverting Amplifier Circuits Figure 15 shows some guidelines for maximizing the distortion performance of the OP275 in noninverting applications. The best way to prevent unwanted distortion is to ensure that the parallel combination of the feedback and gain setting resistors (RF and RG) is less than 2 k. Keeping the values of these resistors small has the added benefits of reducing the thermal noise of the circuit –9– OP275 and dc offset errors. If the parallel combination of RF and RG is larger than 2 k, then an additional resistor, RS, should be used in series with the noninverting input. The value of RS is determined by the parallel combination of RF and RG to maintain the low distortion performance of the OP275. Driving Capacitive Loads The OP275 was designed to drive both resistive loads to 600 and capacitive loads of over 1000 pF and maintain stability. While there is a degradation in bandwidth when driving capacitive loads, the designer need not worry about device stability. The graph in Figure 16 shows the 0 dB bandwidth of the OP275 with capacitive loads from 10 pF to 1000 pF. 10 9 8 6 5 4 3 2 1 A 3-Pole, 40 kHz Low-Pass Filter The closely matched and uniform ac characteristics of the OP275 make it ideal for use in GIC (Generalized Impedance Converter) and FDNR (Frequency-Dependent Negative Resistor) filter applications. The circuit in Figure 18 illustrates a linear-phase, 3-pole, 40 kHz low-pass filter using an OP275 as an inductance simulator (gyrator). The circuit uses one OP275 (A2 and A3) for the FDNR and one OP275 (A1 and A4) as an input buffer and bias current source for A3. Amplifier A4 is configured in a gain of 2 to set the pass band magnitude response to 0 dB. The benefits of this filter topology over classical approaches are that the op amp used in the FDNR is not in the signal path and that the filter’s performance is relatively insensitive to component variations. Also, the configuration is such that large signal levels can be handled without overloading any of the filter’s internal nodes. As shown in Figure 19, the OP275’s symmetric slew rate and low distortion produce a clean, well behaved transient response. 0 800 R1 95.3k 1000 Figure 16. Bandwidth vs. CLOAD VIN High Speed, Low Noise Differential Line Driver The circuit in Figure 17 is a unique line driver widely used in industrial applications. With ±18 V supplies, the line driver can deliver a differential signal of 30 V p-p into a 2.5 k load. The high slew rate and wide bandwidth of the OP275 combine to yield a full power bandwidth of 130 kHz while the low noise front end produces a referred-to-input noise voltage spectral density of 10 nV/Hz. – A2 3 + R1 2k 2 A1 1 R2 787 1 – A2 + 2 3 R3 1.82k 5 6 R6 4.12k A3 C4 2200pF 7 5 R7 100k R9 1k 6 A4 7 VOUT R8 1k C3 2200pF A1, A4 = 1/2 OP275 A2, A3 = 1/2 OP275 R9 50 VO1 R5 1.82k R11 1k R4 2k R7 2k Figure 18. A 3-Pole, 40 kHz Low-Pass Filter VO2 – VO1 = VIN 1 P1 10k R2 2k R5 2k R6 2k 6 A1 = 1/2 OP275 C1 2200pF R4 1.87k – 3 + VIN 1 C2 2200pF R3 2k 2 2 – A1 3 + – 400 600 CLOAD – pF + 200 – 0 + BANDWIDTH – MHz 7 The design is a transformerless, balanced transmission system where output common-mode rejection of noise is of paramount importance. Like the transformer based design, either output can be shorted to ground for unbalanced line driver applications without changing the circuit gain of 1. Other circuit gains can be set according to the equation in the diagram. This allows the design to be easily set to noninverting, inverting, or differential operation. – A3 5 + A2, A3 = 1/2 OP275 100 90 7 R8 2k R10 50 R12 1k VOUT 10V p-p 10kHz VO2 10 R3 GAIN = R1 SET R2, R4, R5 = R1 AND R6, R7, R8 = R3 0% Figure 17. High Speed, Low Noise Differential Line Driver SCALE: VERTICAL–2V/ DIV HORIZONTAL–10s/ DIV Figure 19. Low-Pass Filter Transient Response –10– REV. C OP275 OP275 SPICE Model * * Node assignments * noninverting input * inverting input * positive supply * negative supply * output ** .SUBCKT OP275 1 2 99 50 34 * * INPUT STAGE & POLE AT 100 MHz * R3 5 51 2.188 R4 6 51 2.188 CIN 1 2 3.7E-12 CM1 1 98 7.5E-12 CM2 2 98 7.5E-12 C2 5 6 364E-12 I1 97 4 100E-3 IOS 1 2 1E-9 EOS 9 3 POLY(1) 26 28 0.5E-3 1 Q1 5 2 7 QX Q2 6 9 8 QX R5 7 4 1.672 R6 8 4 1.672 D1 2 36 DZ D2 1 36 DZ EN 3 1 10 0 1 GN1 0 2 13 0 1E-3 GN2 0 1 16 0 1E-3 * EREF 98 0 28 0 1 EP 97 0 99 0 1 EM 51 0 50 0 1 * * VOLTAGE NOISE SOURCE * DN1 35 10 DEN DN2 10 11 DEN VN1 35 0 DC 2 VN2 0 11 DC 2 * * CURRENT NOISE SOURCE * DN3 12 13 DIN DN4 13 14 DIN VN3 12 0 DC 2 VN4 0 14 DC 2 * * CURRENT NOISE SOURCE * DN5 15 16 DIN DN6 16 17 DIN VN5 15 0 DC 2 VN6 0 17 DC 2 * * GAIN STAGE & DOMINANT POLE AT 32 Hz * R7 18 98 1.09E6 C3 18 98 4.55E-9 G1 98 18 5 6 4.57E-1 V2 97 19 1.35 V3 20 51 1.35 D3 18 19 DX D4 20 18 DX REV. C * POLE/ZERO PAIR AT 1.5 MHz/2.7 MHz * R8 21 98 1E-3 R9 21 22 1.25E-3 C4 22 98 47.2E-12 G2 98 21 18 28 1E-3 * * POLE AT 100 MHz * R10 23 98 1 C5 23 98 1.59E-9 G3 98 23 21 28 1 * * POLE AT 100 MHz * R11 24 98 1 C6 24 98 1.59E-9 G4 98 24 23 28 1 * * COMMON-MODE GAIN NETWORK WITH ZERO AT 1 kHz * R12 25 26 1E6 C7 25 26 1.5915E-12 R13 26 98 1 E2 25 98 POLY(2) 1 98 2 98 0 2.50 2.50 * * POLE AT 100 MHz * R14 27 98 1 C8 27 98 1.59E-9 G5 98 27 24 28 1 * * OUTPUT STAGE * R15 28 99 100E3 R16 28 50 100E3 C9 28 50 1E-6 ISY 99 50 1.85E-3 R17 29 99 100 R18 29 50 100 L2 29 34 1E-9 G6 32 50 27 29 10E-3 G7 33 50 29 27 10E-3 G8 29 99 99 27 10E-3 G9 50 29 27 50 10E-3 V4 30 29 1.3 V5 29 31 3.8 F1 29 0 V4 1 F2 0 29 V5 1 D5 27 30 DX D6 31 27 DX D7 99 32 DX D8 99 33 DX D9 50 32 DY D10 50 33 DY * * MODELS USED * .MODEL QX PNP(BF=5E5) .MODEL DX D(IS=1E-12) .MODEL DY D(IS=1E-15 BV=50) .MODEL DZ D(IS=1E-15 BV=7.0) .MODEL DEN D(IS=1E-12 RS=4.35K KF=1.95E-15 AF=1) .MODEL DIN D(IS=1E-12 RS=268 KF=1.08E-15 AF=1) .ENDS –11– OP275 OUTLINE DIMENSIONS 8-Lead Standard Small Outline Package [SOIC] (S Suffix) (R-8) Dimensions shown in millimeters and (inches) 4.00 (0.1574) 3.80 (0.1497) 8 5 1 4 6.20 (0.2440) 5.80 (0.2284) 1.27 (0.0500) BSC 0.25 (0.0098) 0.10 (0.0040) COPLANARITY SEATING 0.10 PLANE C00298–0–2/04(C) 5.00 (0.1968) 4.80 (0.1890) 1.75 (0.0688) 1.35 (0.0532) 0.51 (0.0201) 0.31 (0.0122) 0.50 (0.0196) 45 0.25 (0.0099) 8 0.25 (0.0098) 0 1.27 (0.0500) 0.40 (0.0157) 0.17 (0.0067) COMPLIANT TO JEDEC STANDARDS MS-012AA CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN 8-Lead Plastic Dual-in-Line Package [PDIP] (P Suffix) (N-8) Dimensions shown in inches and (millimeters) 0.375 (9.53) 0.365 (9.27) 0.355 (9.02) 8 5 1 4 0.295 (7.49) 0.285 (7.24) 0.275 (6.98) 0.325 (8.26) 0.310 (7.87) 0.300 (7.62) 0.100 (2.54) BSC 0.180 (4.57) MAX 0.150 (3.81) 0.130 (3.30) 0.110 (2.79) 0.022 (0.56) 0.018 (0.46) 0.014 (0.36) 0.015 (0.38) MIN SEATING PLANE 0.060 (1.52) 0.050 (1.27) 0.045 (1.14) 0.150 (3.81) 0.135 (3.43) 0.120 (3.05) 0.015 (0.38) 0.010 (0.25) 0.008 (0.20) COMPLIANT TO JEDEC STANDARDS MO-095AA CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN Revision History Location Page 2/04—Data Sheet changed from REV. B to REV. C. Changes to ABSOLUTE MAXIMUM RATINGS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 Changes to ORDERING GUIDE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 Updated OUTLINE DIMENSIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12 1/03—Data Sheet changed from REV. A to REV. B. Deleted WAFER TEST LIMITS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 Edits to ABSOLUTE MAXIMUM RATINGS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 Edits to ORDERING GUIDE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 Deleted DICE CHARACTERISTICS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 Updated OUTLINE DIMENSIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12 –12– REV. C