19-5966; Rev 0; 6/11
EVALUATION KIT AVAILABLE
MAX15112
High-Efficiency, 12A, Current-Mode Synchronous
Step-Down Regulator with Integrated Switches
General Description
Features
The MAX15112 high-efficiency, current-mode step-down
regulator with integrated power switches operates from
2.7V to 5.5V and delivers up to 12A of output current in a
small 2mm x 3mm package. The MAX15112 offers excellent efficiency with skip mode capability at light-load conditions, yet provides unmatched efficiency under heavy
load conditions. The combination of small size and high
efficiency makes this device suitable for both portable
and nonportable applications.
S Continuous 12A Output Current Over Temperature
S ±1% Feedback Accuracy Over Load, Line, and
Temperature
S Operates from 2.7V to 5.5V Supply
S Input Undervoltage Lockout
S Adjustable Output Range from 0.6V Up to 0.94 x VIN
S Programmable Soft-Start
S Factory-Trimmed 1MHz Switching Frequency
The MAX15112 utilizes a current-mode control architecture
with a high-gain transconductance error amplifier, which
allows a simple compensation scheme and enables a
cycle-by-cycle current limit with fast response to line and
load transients. A factory-trimmed switching frequency of
1MHz (PWM operation) allows for a compact, all-ceramic
capacitor design.
S Stable with Low-ESR Ceramic Output Capacitors
S Safe-Startup into a Prebiased Output
S External Reference Input
S Selectable Skip Mode Option for Improved
Efficiency at Light Loads
S Enable Input/PGOOD Output Allows Sequencing
Integrated switches with low on-resistance ensure high
efficiency at heavy loads while minimizing critical inductances. The MAX15112’s simple layout and footprint
assure first-pass success in new designs.
S Remote Ground Sense for Improved Accuracy
S Thermal and Overcurrent Protection
S Tiny 2.10mm x 3.05mm, 24-Bump WLP Package
Other features of the MAX15112 include a capacitorprogrammable soft-start to reduce inrush current, safe
startup into a prebiased output, an enable input, and a
power-good output for power sequencing.
Applications
Notebooks
Servers
Distributed Power Systems
The regulator is available in a 24-bump (4 x 6), 2.10mm
x 3.05mm WLP package, and is fully specified over the
-40NC to +85NC extended temperature range.
DDR Memory
Base Stations
Ordering Information appears at end of data sheet.
Typical Operating Circuits
ON
BST
EN
OFF
CBST
SKIP
LOUT
VOUT
LX
AIN
VIN = 2.7V TO 5.5V
MAX15112
COUT
R1
GSNS
IN
CIN
FB
RPULL
PGOOD
PGOOD
SS/REFIN
COMP
GND
CSS
R2
CCC
RC
CC
PWM MODE OPERATION
Typical Operating Circuits continued at end of data sheet.
For related parts and recommended products to use with this part, refer to: www.maxim-ic.com/MAX15112.related
����������������������������������������������������������������� Maxim Integrated Products 1
For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642,
or visit Maxim’s website at www.maxim-ic.com.
MAX15112
High-Efficiency, 12A, Current-Mode Synchronous
Step-Down Regulator with Integrated Switches
ABSOLUTE MAXIMUM RATINGS
IN, PGOOD to GND.................................................-0.3V to +6V
EN, COMP, FB, SS/REFIN, GSNS, SKIP,
LX to GND...............................................-0.3V to (VIN + 0.3V)
LX to GND (for 10ns).........................................-2V to (VIN + 2V)
LX to GND (for 50ns).........................................-1V to (VIN + 1V)
BST to LX..................................................................-0.3V to +6V
BST to GND............................................................-0.3V to +12V
BST to IN..................................................................-0.3V to +6V
LX Continuous Current (Note 1).......................................... Q15A
Output Short-Circuit Duration . ..................................Continuous
Continuous Power Dissipation
WLP (derate 53.85mW/NC above +70NC)......................2.15W
Operating Temperature Range........................... -40NC to +85NC
Junction Temperature (Note 2)........................................+110NC
Storage Temperature Range............................. -65NC to +150NC
Bump Reflow Temperature (Note 3)................................+260NC
Note 1: LX has internal clamp diodes to GND and IN. Applications that forward bias these diodes must take care not to exceed
the IC’s package power dissipation limits.
Note 2: Limit the junction temperature to +110NC for continuous operation at maximum output current.
Note 3: The WLP package is constructed using a unique set of package techniques that impose a limit on the thermal profile the
device can be exposed to during board-level solder attach and rework. This limit permits only the use of the solder profiles recommended in the industry-standard specification JEDEC 020A, paragraph 7.6, Table 3 for IR/VPR and convection
reflow. Preheating is required. Hand or wave soldering is not allowed.
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect device reliability.
DC ELECTRICAL CHARACTERISTICS
(VIN = 5V, see the Typical Operating Circuits, TA = -40NC to +85NC. Typical values are at TA = +25NC, unless otherwise noted.) (Note 4)
PARAMETER
IN Voltage Range
SYMBOL
CONDITIONS
VIN
IN Supply Current
IIN
MIN
TYP
2.7
MAX
UNITS
5.5
V
VEN = VIN, VFB = 0.5V, no switching
4.6
7
mA
IN Shutdown Current
ISHDN
VEN = 0V
0.01
3
FA
IN Undervoltage Lockout
Threshold
VUVLO
VIN rising, LX starts switching
2.6
2.68
V
VIN falling, LX stops switching
200
mV
1.1
mS
IN Undervoltage Lockout
Threshold Hysteresis
ERROR AMPLIFIER
Transconductance
Voltage Gain
gM
AVEA
90
FB Setpoint Voltage
VFB
Over line, load, and temperature
0.594
FB Input Bias Current
IFB
VFB = 0.6V
-500
COMP to Current-Sense
Transconductance
gMC
COMP Clamp Low Voltage
Slope Compensation Ramp
Amplitude
VFB = 0.65V, VSS/REFIN = 0.6V
VSLOPE
0.600
dB
0.606
V
+500
nA
80
A/V
0.91
V
130
mV
����������������������������������������������������������������� Maxim Integrated Products 2
MAX15112
High-Efficiency, 12A, Current-Mode Synchronous
Step-Down Regulator with Integrated Switches
DC ELECTRICAL CHARACTERISTICS (continued)
(VIN = 5V, see the Typical Operating Circuits, TA = -40NC to +85NC. Typical values are at TA = +25NC, unless otherwise noted.) (Note 4)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
GROUND SENSE
GSNS Output Current
VSS/REFIN = 0.6V, VGSNS = 0V
52
High-side switch
24
Low-side switch, sinking
24
Low-side switch, sourcing
24
FA
POWER SWITCHES
Current-Limit Threshold
LX Leakage Current
VEN = 0V
BST Leakage Current
BST On-Resistance
VEN = 0V
RON_BST
IBST = 50mA
LX RMS Output Current
A
3
FA
3
FA
0.63
I
12
A
OSCILLATOR
Switching Frequency
fSW
Maximum Duty Cycle
DMAX
Minimum Controllable On-Time
850
1000
PWM mode
94
Skip mode
85
tON
1150
kHz
%
70
ns
ENABLE FUNCTIONALITY
EN Input High Threshold
VIH
VEN rising
EN Input Low Threshold
VIL
VEN falling
EN Input Leakage Current
1.4
V
+1
FA
1.4
V
0.4
V
-1
SKIP FUNCTIONALITY (Note 5)
SKIP Input High Threshold
VSKIP rising
SKIP Input Low Threshold
VSKIP falling
0.4
SKIP Pulldown Resistor
Minimum LX On-Current in Skip
Mode
Zero-Crossing LX Threshold
V
230
kI
3
A
0.5
A
SOFT-START AND PREBIAS FUNCTIONALITY
Soft-Start Current
ISS
VSS/REFIN = 0.45V, sourcing
SS/REFIN Discharge Resistance
RSS
ISS/REFIN = 10mA, sinking
SS/REFIN Prebias Mode Stop
Voltage
VSS/REFIN rising
6.8
10
12.5
FA
7
I
0.58
V
SS/REFIN External Reference
Input Range
VIN 2.5
V
HICCUP MODE
Number of Consecutive CurrentLimit Events to Hiccup Mode
Hiccup Mode Timeout
NHIC
8
Events
1024
Clock
Cycles
����������������������������������������������������������������� Maxim Integrated Products 3
MAX15112
High-Efficiency, 12A, Current-Mode Synchronous
Step-Down Regulator with Integrated Switches
DC ELECTRICAL CHARACTERISTICS (continued)
(VIN = 5V, see the Typical Operating Circuits, TA = -40NC to +85NC. Typical values are at TA = +25NC, unless otherwise noted.) (Note 4)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
0.514
0.529
0.542
V
POWER-GOOD OUTPUT
PGOOD Threshold
VFB falling, PGOOD deasserts
PGOOD Threshold Hysteresis
VFB rising
25
18
PGOOD Output Voltage Low
VPG_OL
IPGOOD = 5mA, VEN = 0V
PGOOD Leakage Current
IPG_LK
VPGOOD = 5.5V, VFB = 0.65V
TSHDN
Die temperature rising
mV
50
mV
1
FA
THERMAL SHUTDOWN
Thermal Shutdown Threshold
Thermal Shutdown Hysteresis
+150
NC
20
NC
Note 4: All devices are 100% production tested at TA = +25NC. Limits over the operating temperature range are guaranteed by design.
Note 5: Connect SKIP to EN for skip mode functionality. Leave SKIP unconnected or connect to GND for PWM mode functionality.
Typical Operating Characteristics
(VIN = 5V, VOUT = 1.5V, ILOAD = 4A, CSS = 33nF, see the Typical Operating Circuits, TA = +25°C, unless otherwise noted.)
EFFICIENCY (%)
70
VOUT = 1.5V
60
50
VOUT = 1.2V
40
70
VOUT = 2.5V
60
VOUT = 1.8V
50
VOUT = 1.2V
40
30
30
20
20
10
10
0
1
10
LOAD CURRENT (A)
VOUT = 2.5V
95
90
85
VOUT = 1.8V
80
VOUT = 1.5V
VOUT = 1.2V
75
70
65
VOUT = 1.5V
60
55
0
0.1
100
MAX15112 toc03
VOUT = 3.3V
80
EFFICIENCY (%)
VOUT = 2.5V
VOUT = 1.8V
90
EFFICIENCY vs. LOAD CURRENT
(VIN = 3.3V, SKIP MODE)
EFFICIENCY (%)
90
80
100
MAX15112 toc01
100
EFFICIENCY vs. LOAD CURRENT
(VIN = 5V, PWM MODE)
MAX15112 toc02
EFFICIENCY vs. LOAD CURRENT
(VIN = 3.3V, PWM MODE)
50
0.1
1
10
LOAD CURRENT (A)
100
0.1
1
10
100
LOAD CURRENT (A)
����������������������������������������������������������������� Maxim Integrated Products 4
MAX15112
High-Efficiency, 12A, Current-Mode Synchronous
Step-Down Regulator with Integrated Switches
Typical Operating Characteristics (continued)
(VIN = 5V, VOUT = 1.5V, ILOAD = 4A, CSS = 33nF, see the Typical Operating Circuits, TA = +25°C, unless otherwise noted.)
SWITCHING FREQUENCY
vs. INPUT VOLTAGE (PWM, IOUT = 0A)
EFFICIENCY vs. LOAD CURRENT
(VIN = 5V, SKIP MODE)
85
VOUT = 1.8V
80
VOUT = 1.5V
75
70
65
60
TA = +85°C
1060
1040
1020
TA = +25°C
1000
980
960
TA = -40°C
940
920
55
900
50
1
0.1
10
2.5
100
3.0
LOAD CURRENT (A)
OUTPUT VOLTAGE (V)
1.504
IOUT = 0A
1.502
IOUT = 6A
1.500
1.498
1.496
IOUT = 12A
1.494
4.5
5.0
5.5
REFERENCED TO VIN = 4V
0.15
0.10
0.05
VOUT = 1.2V
0
-0.05
-0.10
VOUT = 1.8V
-0.15
-0.20
1.492
2.5
3.0
3.5
4.0
4.5
5.0
5.5
2.5
3.0
3.5
4.0
OUTPUT VOLTAGE
vs. LOAD CURRENT (PWM, VOUT = 1.5V)
1.508
1.506
5.5
MAX15112 toc09
VOUT
20mV/div
AC-COUPLED
1.504
VIN = 3.3V
VIN = 5V
5.0
LOAD-TRANSIENT RESPONSE
(VIN = 5V, PWM, 1A /µs)
MAX15112 toc08
1.510
1.502
4.5
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
0.20
OUTPUT VOLTAGE ERROR (%)
MAX15112 toc06
1.506
4.0
OUTPUT VOLTAGE ERROR
vs. INPUT VOLTAGE (PWM, IOUT = 6A)
OUTPUT VOLTAGE
vs. INPUT VOLTAGE (PWM)
1.508
3.5
INPUT VOLTAGE (V)
MAX15112 toc07
EFFICIENCY (%)
90
1080
MAX15112 toc05
VOUT = 2.5V
SWITCHNG FREQUENCY (kHz)
95
1100
VOUT = 3.3V
MAX15112 toc04
100
1.500
6A
1.498
IOUT
2A/div
1.496
1.494
0.1A
1.492
1.490
0
2
4
6
8
LOAD CURRENT (A)
10
12
40µs/div
����������������������������������������������������������������� Maxim Integrated Products 5
MAX15112
High-Efficiency, 12A, Current-Mode Synchronous
Step-Down Regulator with Integrated Switches
Typical Operating Characteristics (continued)
(VIN = 5V, VOUT = 1.5V, ILOAD = 4A, CSS = 33nF, see the Typical Operating Circuits, TA = +25°C, unless otherwise noted.)
LOAD-TRANSIENT RESPONSE
(VIN = 3.3V, PWM, 1A /µs)
LOAD-TRANSIENT RESPONSE
(VIN = 5V, SKIP, 1A /µs)
MAX15112 toc10
MAX15112 toc11
VOUT
20mV/div
AC-COUPLED
VOUT
50mV/div
AC-COUPLED
6A
IOUT
2A/div
4A
IOUT
2A/div
0.1A
0.1A
40µs/div
40µs/div
LOAD-TRANSIENT RESPONSE
(VIN = 3.3V, SKIP, 1A /µs)
LOAD-TRANSIENT RESPONSE
(VIN = 5V, PWM, 1A /µs)
MAX15112 toc12
MAX15112 toc13
VOUT
50mV/div
AC-COUPLED
VOUT
50mV/div
AC-COUPLED
4A
IOUT
2A/div
11A
IOUT
5A/div
0.1A
40µs/div
1A
40µs/div
SWITCHING WAVEFORMS
(VIN = 5V, IOUT = 12A, PWM)
LOAD-TRANSIENT RESPONSE
(VIN = 3.3V, PWM, 1A /µs)
MAX15112 toc15
MAX15112 toc14
VOUT
20mV/div
AC-COUPLED
VOUT
50mV/div
AC-COUPLED
VLX
5V/div
11A
IOUT
5A/div
ILX
5A/div
1A
0A
40µs/div
400ns/div
����������������������������������������������������������������� Maxim Integrated Products 6
MAX15112
High-Efficiency, 12A, Current-Mode Synchronous
Step-Down Regulator with Integrated Switches
Typical Operating Characteristics (continued)
(VIN = 5V, VOUT = 1.5V, ILOAD = 4A, CSS = 33nF, see the Typical Operating Circuits, TA = +25°C, unless otherwise noted.)
SWITCHING WAVEFORMS
(VIN = 3.3V, IOUT = 12A, PWM)
SWITCHING WAVEFORMS
(VIN = 5V, IOUT = 1A, SKIP)
MAX15112 toc16
MAX15112 toc17
VOUT
20mV/div
AC-COUPLED
VOUT
20mV/div
AC-COUPLED
VLX
2V/div
VLX
2V/div
ILX
5A/div
ILX
2A /div
0A
400ns/div
400ns/div
SOFT-START WAVEFORMS
(PWM, IOUT = 6A)
SHUTDOWN WAVEFORMS (IOUT = 6A)
MAX15112 toc18
MAX15112 toc19
VEN
5V/div
VEN
5V/div
VOUT
500mV/div
VPGOOD
5V/div
VOUT
500mV/div
IOUT
5A /div
VPGOOD
5V/div
IOUT
5A/div
100µs/div
400µs/div
SOFT-START WAVEFORMS
(IOUT = 1A, SKIP)
SHUTDOWN CURRENT
vs. INPUT VOLTAGE (VEN = 0V)
MAX15112 toc20
VPGOOD
5V/div
IOUT
1A /div
SHUTDOWN CURRENT (nA)
VOUT
500mV/div
MAX15112 toc21
100
VEN
5V/div
80
60
40
20
0
400µs/div
2.5
3.0
3.5
4.0
4.5
5.0
5.5
INPUT VOLTAGE (V)
����������������������������������������������������������������� Maxim Integrated Products 7
MAX15112
High-Efficiency, 12A, Current-Mode Synchronous
Step-Down Regulator with Integrated Switches
Typical Operating Characteristics (continued)
(VIN = 5V, VOUT = 1.5V, ILOAD = 4A, CSS = 33nF, see the Typical Operating Circuits, TA = +25°C, unless otherwise noted.)
RMS INPUT CURRENT
vs. INPUT VOLTAGE (PWM)
SHORT-CIRCUIT HICCUP MODE
MAX15112 toc22
IOUT
10A /div
230
RMS INPUT CURRENT (mA)
IIN
5A /div
MAX15112 toc23
250
VOUT
1V/div
210
190
170
150
130
110
90
70
SHORT CIRCUIT ON OUTPUT
50
1ms/div
2.5
3.0
3.5
4.0
4.5
5.0
5.5
INPUT VOLTAGE (V)
FEEDBACK VOLTAGE
vs. TEMPERATURE (NO LOAD, PWM)
SOFT-START WAVEFORMS
(EXTERNAL REFIN, PWM MODE)
MAX15112 toc25
MAX15112 toc24
0.606
0.605
0.604
0.603
VSS/REFIN
500mV/div
VOUT
1V/div
VFB (V)
0.602
0.601
NO LOAD
0.600
0.599
0.598
0.597
ILX
5A /div
0.596
0.595
0.594
VPGOOD
5V/div
-40
-15
10
35
60
85
400µs/div
TEMPERATURE (°C)
STARTING INTO 1V PREBIASED OUTPUT
(PWM, IOUT = 4A)
SOFT-START WAVEFORMS
(EXTERNAL REFIN, SKIP MODE)
MAX15112 toc26
MAX15112 toc27
VEN
5V/div
VSS/REFIN
500mV/div
VOUT
1V/div
VOUT
500mV/div
1V
NO LOAD
ILX
5A /div
ILX
5A /div
VPGOOD
5V/div
VPGOOD
5V/div
400µs/div
400µs/div
����������������������������������������������������������������� Maxim Integrated Products 8
MAX15112
High-Efficiency, 12A, Current-Mode Synchronous
Step-Down Regulator with Integrated Switches
Typical Operating Characteristics (continued)
(VIN = 5V, VOUT = 1.5V, ILOAD = 4A, CSS = 33nF, see the Typical Operating Circuits, TA = +25°C, unless otherwise noted.)
STARTING INTO 1V PREBIASED OUTPUT
(PWM, NO LOAD)
STARTING INTO 1V PREBIASED OUTPUT
(SKIP MODE, NO LOAD)
MAX15112 toc28
MAX15112 toc29
VEN
5V/div
1V
VEN
5V/div
1V
VOUT
500mV/div
ILX
2A /div
ILX
2A /div
VPGOOD
5V/div
VPGOOD
5V/div
400µs/div
400µs/div
STARTING INTO A PREBIASED OUTPUT
HIGHER THAN SET OUTPUT
INPUT CURRENT IN SKIP MODE
vs. OUTPUT VOLTAGE (NO LOAD)
MAX15112 toc30
VOUT
500mV/div
ILX
5A /div
4.5
INPUT CURRENT (mA)
1.5V
MAX15112 toc31
5.0
VEN
5V/div
1.8V
4.0
VIN = 5V
3.5
3.0
2.5
VIN = 3.3V
2.0
VSS/REFIN
500mV/div
10ms/div
VOUT
500mV/div
1.5
1.0
0.6
1.0
1.4
1.8
2.2
2.6
OUTPUT VOLTAGE (V)
����������������������������������������������������������������� Maxim Integrated Products 9
MAX15112
High-Efficiency, 12A, Current-Mode Synchronous
Step-Down Regulator with Integrated Switches
Pin Configuration
TOP VIEW
(BUMP ON THE BOTTOM)
+
MAX15112
GND
LX
LX
BST
PGOOD
GSNS
A1
A2
A3
A4
A5
A6
GND
LX
LX
IN
N.C.
FB
B1
B2
B3
B4
B5
B6
GND
LX
LX
IN
SKIP
SS/REFIN
C1
C2
C3
C4
C5
C6
GND
GND
AIN
IN
EN
COMP
D1
D2
D3
D4
D5
D6
WLP
Pin Description
PIN
NAME
FUNCTION
Ground Connection. GND is the source terminal of the internal low-side switch. Connect all GND
bumps to a component-side PCB copper ground plane at a single point near the input bypass
capacitor return terminal.
A1, B1, C1,
D1, D2
GND
A2, A3, B2,
B3, C2, C3
LX
A4
BST
A5
PGOOD
A6
GSNS
B4, C4, D4
IN
B5
N.C
No Connection. Do not connect.
B6
FB
Feedback Input. Connect FB to the center tap of an external resistor divider from the output to the
output capacitor return terminal to set the output voltage from 0.6V to 0.94 x VIN.
C5
SKIP
Skip-Mode Selector Input. Connect SKIP to EN for skip-mode operation. Connect SKIP to GND or leave
unconnected for continuous mode operation. Do not change the state of SKIP when EN is high.
C6
SS/REFIN
Soft-Start and External Voltage Reference Input. Connect a capacitor from SS/REFIN to GND to set
the soft-start delay. See the Setting the Soft-Start Time section for more information. To use SS/REFIN
as an external voltage reference, apply a voltage ranging from 0V to (VIN - 2.5V) to SS/REFIN to
externally control the soft-start time and feedback voltage.
D3
AIN
Filtered Input Voltage
D5
EN
Enable Input. Drive EN high to enable the MAX15112. Connect EN to IN for always-on operation.
D6
COMP
Inductor Connection. Connect LX to the switching side of the inductor. LX is high impedance when
the MAX15112 is in shutdown mode.
Boost Input for the High-Side Switch Driver. Connect a capacitor from BST to LX.
Power-Good Open-Drain Output. PGOOD asserts high when VFB is above 0.554V (typ) and
deasserts when VFB falls below 0.529V (typ).
Remote Ground-Sense Input. Connect GSNS to the ground terminal of the load and to the bottom of
the feedback resistors.
Input Power Supply. Bypass IN to GND with at least two 22FF low-ESR ceramic capacitors with sufficient ripple current ratings.
Error Amplifier Output. Connect the compensation network from COMP to GND. See the
Compensation Design Guidelines section for more information.
���������������������������������������������������������������� Maxim Integrated Products 10
MAX15112
High-Efficiency, 12A, Current-Mode Synchronous
Step-Down Regulator with Integrated Switches
Functional Diagram
SKIP
EN
AIN
IN
BIAS
GENERATOR
CURRENT-SENSE
AMPLIFIER
SKPM
SKIP MODE
LOGIC
EN LOGIC,
IN UVLO, THERMAL
SHDN
LX
VOLTAGE
REFERENCE
MAX15112
0.58V
HIGH-SIDE
CURRENT LIMIT
0.6V
LX
PREBIAS
ABOVE
FORCED
PWM
START
SS/REFIN
10µA
SS/REFIN BUFFER
BST
ERROR
AMPLIFIER
AV = 1
GSNS
IN
PWM
COMPARATOR
gM
CONTROL
LOGIC
FB
COMP
LX
C
PGOOD
CK
IN
554mV, RISING
529mV, FALLING
GND
COMPENSATION
RAMP
OSCILLATOR
RAMP
GENERATOR
CK
LOW-SIDE SOURCE-SINK
CURRENT-LIMIT AND ZEROCROSSING COMPARATOR
SOURCE
SINK
ZX
SKPM
���������������������������������������������������������������� Maxim Integrated Products 11
MAX15112
High-Efficiency, 12A, Current-Mode Synchronous
Step-Down Regulator with Integrated Switches
Detailed Description
The MAX15112 high-efficiency, current-mode switching
regulator delivers up to 12A of output current. The regulator provides output voltages from 0.6V up to 0.94 x VIN
from 2.7V to 5.5V input supplies, making the device ideal
for on-board point-of-load applications.
The MAX15112 delivers current-mode control architec
ture using a high-gain transconductance error amplifier.
The current-mode control architecture facilitates easy
compensation design and ensures cycle-by-cycle cur
rent limit with fast response to line and load transients.
The regulator features a 1MHz fixed switching frequency,
allowing for all-ceramic capacitor designs and fast transient responses. The high operating frequency minimizes
the size of external components.
The regulator offers a selectable skip mode functionality
to reduce current consumption and achieve a higher
efficiency at light output loads. Integrated switches
ensure high efficiency at heavy loads while minimizing
critical inductances.
The MAX15112 features PWM current-mode control,
allowing for an all-ceramic capacitor solution. The regulator offers capacitor-programmable soft-start to reduce
input inrush current. The device safely starts up into a
prebiased output. The MAX15112 includes an enable
input and open-drain PGOOD output for sequencing with
other devices.
Controller Function—PWM Logic
The controller logic block is the central processor that
determines the duty cycle of the high-side MOSFET
under different line, load, and temperature conditions.
Under normal operation, where the current-limit and
temperature protection are not triggered, the controller
logic block takes the output from the PWM comparator
and generates the driver signals for both high-side and
low-side MOSFETs. The control logic block controls the
break-before-make logic and all the necessary timing.
The high-side MOSFET turns on at the beginning of the
oscillator cycle and turns off when the COMP voltage
crosses the internal current-mode ramp waveform. The
internal ramp is the sum of the compensation ramp and
the current-mode ramp derived from the inductor current
(current-sense block). The high-side MOSFET also turns
off if either the maximum duty cycle (94%, typ) or the current limit is reached. The low-side MOSFET turns on for
the remainder of the oscillation cycle.
Starting into a Prebiased Output
The MAX15112 can soft-start into a prebiased output
without discharging the output capacitor. In safe prebiased startup, both low-side and high-side MOSFETs
remain off to avoid discharging the prebiased output.
PWM operation starts when the voltage on SS/REFIN
crosses the voltage on FB.
The MAX15112 can start into a prebiased voltage higher
than the nominal set point without abruptly discharging the output. Forced PWM operation starts when the
SS/REFIN voltage reaches 0.58V (typ), forcing the converter to start. The low-side current limit is increased over
350µs to the maximum from the first LX pulse. When the
low-side sink current-limit threshold of 24A is reached,
the low-side switch turns off before the end of the clock
period and the high-side switch turns on until one of the
following conditions is satisfied:
U High-side source current hits the reduced high-side
current limit (24A, typ); in this case, the high-side
switch is turned off for the remaining time of the clock
period.
U The clock period ends.
Reduced high-side current limit is activated to recirculate
the current into the high-side power switch rather than
into the internal high-side body diode.
Low-side sink current limit is provided to protect the
low-side switch from excessive reverse current during
prebiased operation.
Enable Input and Power-Good
(PGOOD) Output
The MAX15112 features independent enable control and a
power-good signal that allows for flexible power sequencing. Drive the enable input (EN) high to enable the regulator, or connect EN to IN for always-on operation.
Power good (PGOOD) is an open-drain output that
asserts when VFB is above 554mV (typ) and deasserts
low if VFB is below 529mV (typ).
Programmable Soft-Start (SS/REFIN)
The MAX15112 utilizes a soft-start feature to slowly
ramp up the regulated output voltage to reduce input
inrush current during startup. Connect a capacitor from
SS/REFIN to GND to set the startup time (see the Setting
the Soft-Start Time section for capacitor selection details).
���������������������������������������������������������������� Maxim Integrated Products 12
MAX15112
High-Efficiency, 12A, Current-Mode Synchronous
Step-Down Regulator with Integrated Switches
Error Amplifier
A high-gain transconductance error amplifier provides
accuracy for the voltage-feedback loop regulation.
Connect the necessary compensation network between
COMP and GND (see the Compensation Design
Guidelines section). The error-amplifier transconductance is 1.1mS (typ). COMP clamp low is set to 0.91V
(typ), just below the slope ramp compensation valley,
helping COMP to rapidly return to the correct set point
during load and line transients.
Ground-Sense Amplifier
The MAX15112 features a ground-sense amplifier to prevent output voltage droop under heavy load conditions.
Connect GSNS to the negative terminal of the load output
capacitor to properly Kelvin-sense the output ground.
Route the GSNS trace away from the switching nodes.
PWM Comparator
The PWM comparator compares the COMP voltage to
the current-derived ramp waveform (COMP voltage to LX
current transconductance value is 80A/V, typ). To avoid
instability due to subharmonic oscillations when the duty
cycle is around 50% or higher, a slope compensation
ramp is added to the current-derived ramp waveform.
The compensation ramp slope is designed to ensure
stable operation at any duty cycle up to 94%.
Overcurrent Protection and Hiccup Mode
When the converter output is shorted or the device is
overloaded, each high-side MOSFET current-limit event
turns off the high-side MOSFET and turns on the low-side
MOSFET. On each current-limit event (either high-side
or low-side) a 3-bit counter is incremented. The counter
is reset after three consecutive switching cycles that do
not reach the current limit. If the current-limit condition
persists, the counter fills up reaching eight events. The
control logic then keeps the low-side MOSFET turned on
until the inductor current is fully discharged to avoid high
currents circulating through the low-side body diode.
The control logic turns off both high-side and low-side
MOSFETs and waits for the hiccup period (1024 clock
cycles, typ) before attempting a new soft-start sequence.
The hiccup mode is also enabled during soft-start time.
Thermal Shutdown Protection
The MAX15112 contains an internal thermal sensor that
limits the total power dissipation to protect the device in
the event of an extended thermal fault condition. When
the die temperature exceeds +150NC (typ), the thermal
sensor shuts down the device, turning off the DC-DC
converter to allow the die to cool. After the die tempera
ture falls by 20NC (typ), the device restarts.
Skip Mode Operation
The MAX15112 features selectable skip mode operation
when SKIP is connected to EN. When in skip mode, the
LX output becomes high impedance when the inductor
current falls below 0.5A (typ). The inductor current does
not become negative. If during a clock cycle the inductor
current falls below the 0.5A threshold (during off-time),
the low-side turns off. At the next clock cycle, if the
output voltage is above set point, the PWM logic keeps
both high-side and low-side MOSFETs off. If instead the
output voltage is below the set point, the PWM logic
drives the high-side on until a reduced current limit
threshold (3A, typ) is reached. In this way the system can
skip cycles, reducing the frequency of operation, and
switches only as needed to service load at the cost of
an increase in output voltage ripple (see the Skip Mode
Frequency and Output Ripple section). In skip mode,
power dissipation is reduced and efficiency is improved
at light loads because power MOSFETs do not switch at
every clock cycle.
The MAX15112 automatically enters continuous mode
regardless of the state of SKIP when the load current
increases beyond the skip mode current limit.
Do not change the state of SKIP when EN is high.
���������������������������������������������������������������� Maxim Integrated Products 13
MAX15112
High-Efficiency, 12A, Current-Mode Synchronous
Step-Down Regulator with Integrated Switches
Applications Information
Setting the Output Voltage
The MAX15112 output voltage is adjustable from 0.6V
up to 94% of VIN by connecting FB to the center tap of
a resistor-divider between the output and GND (see the
Typical Operating Circuits). Choose R1 and R2 values
so that the DC errors due to the FB input bias current
(Q500nA) do not affect the output voltage accuracy.
With lower value resistors the DC error is reduced, but
the amount of power consumed in the resistor-divider
increases. R2 values between 1kI and 20kI are acceptable (see Table 1 for typical values). Once R2 is chosen,
calculate R1 using:
R1=R2 × (VOUT /VFB ) -1
where the feedback threshold voltage VFB = 0.6V (typ).
When regulating for an output of 0.6V in skip mode, short
FB to OUT and keep R2 connected from FB to GND.
Inductor Selection
A high-valued inductor results in reduced inductor-ripple
current, leading to a reduced output-ripple voltage.
However, a high-valued inductor results in either a larger
physical size or a high series resistance (DCR) and a
lower saturation current rating. Typically, choose an
inductor value to produce a current ripple, DIL, equal to
30% of load current. Choose the inductor with the following formula:
L=
V
VOUT
× 1- OUT
fSW × LIR × ILOAD
VIN
where fSW is the fixed 1MHz switching frequency, and
LIR is the desired inductor current ratio (typically 0.3). In
addition, the peak inductor current, IL_PK, must always
be below the 24A high-side current-limit and the inductor
saturation current rating, IL_SAT. Ensure that the following
relationship is satisfied:
IL_PK = ILOAD +
1
∆IL(P-P) < min (24A, IL_SAT )
2
where:
∆IL(P-P) =
(VIN − VOUT )
x
L x fSW
VOUT
VIN
Input Capacitor Selection
For a step-down converter, the input capacitor, CIN,
helps to keep the DC input voltage steady, in spite of
discontinuous input AC current. Use low-ESR capacitors
to minimize the voltage ripple due to ESR.
Size CIN using the following formula:
CIN =
ILOAD
V
× OUT
fSW × ∆VIN_RIPPLE
VIN
where DVIN_RIPPLE is the maximum-allowed input-ripple
voltage across the input capacitors and is recommend
ed to be less than 2% of the minimum input voltage,
fSW is the switching frequency (1MHz), and ILOAD is the
output load. The impedance of the input capacitor at
the switching frequency should be less than that of the
input source so high-frequency switching currents do not
pass through the input source, but are instead shunted
through the input capacitor.
Ensure that the input capacitor can accommodate the
input-ripple current requirement imposed by the switching currents. The RMS input-ripple current is given by:
12
V
OUT × (VIN - VOUT )
IRMS =
×I
LOAD
VIN
where IRMS is the input RMS ripple current.
Use multiple capacitors in parallel to meet the RMS current rating requirement.
���������������������������������������������������������������� Maxim Integrated Products 14
MAX15112
High-Efficiency, 12A, Current-Mode Synchronous
Step-Down Regulator with Integrated Switches
Output Capacitor Selection
The key selection parameters for the output capacitor are
capacitance, ESR, ESL, and voltage-rating requirements.
These affect the overall stability, output-ripple voltage,
and transient response of the DC-DC converter. The output ripple occurs due to variations in the charge stored in
the output capacitor, the voltage drop due to the capacitor’s ESR, and the voltage drop due to the capacitor’s
ESL. Estimate the output-voltage ripple due to the output
capacitance, ESR, and ESL as follows:
VRIPPLE = VRIPPLE(C) + VRIPPLE(ESR) + VRIPPLE(ESL)
where the output ripple due to output capacitance, ESR,
and ESL is:
VRIPPLE(C) =
∆IP −P
8 × C OUT × fSW
VRIPPLE(ESR) = ∆IP −P × ESR
and VRIPPLE(ESL) can be approximated as an inductive
divider from LX to GND:
VRIPPLE (ESL) = VLX ×
ESL
ESL
= VIN ×
L
L
where VLX swings from VIN to GND.
The peak-to-peak inductor current (DIP-P) is:
VOUT
VIN
(VIN − VOUT ) ×
∆IP −P =
L × fSW
When using ceramic capacitors, which generally have
low-ESR, DVRIPPLE(C) dominates. When using electrolytic capacitors, DVRIPPLE(ESR) dominates. Use ceramic
capacitors for low ESR and low ESL at the switching frequency of the converter. The ripple voltage due to ESL is
negligible when using ceramic capacitors.
As a general rule, a smaller inductor-ripple current results
in less output-ripple voltage. Since inductor-ripple current depends on the inductor value and input voltage, the
output-ripple voltage decreases with larger inductance
and increases with higher input voltages. However, the
inductor-ripple current also impacts transient-response
performance, especially at low VIN to VOUT differentials.
Low inductor values allow the inductor current to slew
faster, replenishing charge removed from the output filter
capacitors by a sudden load step.
Load-transient response also depends on the selected
output capacitance. During a load transient, the output
instantly changes by ESR x ∆ILOAD. Before the controller
can respond, the output deviates further, depending on
the inductor and output capacitor values. After a short
time, the controller responds by regulating the output
voltage back to the predetermined value.
Use higher COUT values for applications that require
light-load operation or transition between heavy load and
light load, triggering skip mode, causing output undershooting or overshooting. When applying the load, limit
the output undershooting by sizing COUT according to
the following formula:
∆ILOAD
C OUT =
3fCO × ∆VOUT
where ∆ILOAD is the total load change, fCO is the unitygain bandwidth (or zero-crossing frequency), and ∆VOUT
is the desired output undershooting. When removing the
load and entering skip mode, the device cannot control
output overshooting, since it has no sink current capability; see the Skip Mode Frequency and Output Ripple
section to properly size COUT under this circumstance.
A worst-case analysis in sizing the minimum output
capacitance takes the total energy stored in the inductor
into account, as well as the allowable sag/soar (undershoot/overshoot) voltage as follows:
C OUT (MIN) =
C OUT(MIN) =
(
L × I 2 OUT(MAX) − I 2 OUT(MIN)
(VFIN + VSOAR )
2
2
− V INIT
(
L × I 2 OUT(MAX) − I 2 OUT(MIN)
2
V INIT − (VFIN − VSAG )
2
) , voltage soar (overshoot)
) , voltage sag (undershoot)
where IOUT(MAX) and IOUT(MIN) are the initial and final
values of the load current during the worst-case load
dump, VINIT is the initial voltage prior to the transient,
VFIN is the steady-state voltage after the transient, VSOAR
is the allowed voltage soar (overshoot) above VFIN, and
VSAG is the allowable voltage sag below VFIN. The terms
(VFIN + VSOAR) and (VFIN - VSAG) represent the maximum/minimum transient output voltage reached during
the transient, respectively.
Use these equations for initial output-capacitor selection.
Determine final values by testing a prototype or an evaluation circuit under the worst-case conditions.
���������������������������������������������������������������� Maxim Integrated Products 15
MAX15112
High-Efficiency, 12A, Current-Mode Synchronous
Step-Down Regulator with Integrated Switches
Skip Mode Frequency and Output Ripple
Enable skip mode in battery-powered systems for high
efficiency at light loads. In skip mode the switching frequency (fSKIP), as illustrated in Figure 1, is calculated as
follows:
fSKIP =
1
t ON + t OFF1 + t OFF2
and:
t OFF2 =
1
1 I SKIP_LIMIT
+
− ILOAD
L × I SKIP_LIMIT ×
×
V
V
V
2
IN
OUT
OUT
t OFF2 =
ILOAD
Output ripple in skip mode is:
where:
L × I SKIP_LIMIT
VOUT_RIPPLE =
+ R ESR_COUT
C OUT × (VIN - VOUT )
L
× I SKIP_LIMIT
t ON =
VIN − VOUT
t OFF1 =
∆Q OUT
ILOAD
× (I SKIP_LIMIT - ILOAD )
L × I SKIP_LIMIT
VOUT
IL
ISKIP-LIMIT
ILOAD
tON
tOFF1
tOFF2 = n × tCK
VOUT
VOUT_RIPPLE
Figure 1. Skip Mode Waveform
���������������������������������������������������������������� Maxim Integrated Products 16
MAX15112
High-Efficiency, 12A, Current-Mode Synchronous
Step-Down Regulator with Integrated Switches
Compensation Design Guidelines
The MAX15112 uses a fixed-frequency, peak currentmode control scheme to provide easy compensation
and fast transient response. The inductor peak current
is monitored on a cycle-by-cycle basis and compared to
the COMP voltage (output of the voltage error amplifier).
The regulator’s duty cycle is modulated based on the
inductor’s peak current value. This cycle-by-cycle control of the inductor current emulates a controlled current
source. As a result, the inductor’s pole frequency is shifted beyond the gain bandwidth of the regulator. System
stability is provided with the addition of a simple series
capacitor-resistor from COMP to GND. This pole-zero
combination serves to tailor the desired response of the
closed-loop system. The basic regulator loop consists of
a power modulator (composed of the regulator’s pulse-
FEEDBACK
DIVIDER
R LOAD × I L
VOUT
=
= R LOAD × G MOD
VCOMP
IL
G MOD
where IL is the average inductor current, GMOD is the
power modulator’s transconductance, and RLOAD is the
equivalent load resistance value.
POWER MODULATOR
ERROR AMPLIFIER
COMPENSATION
RAMP
VOUT
R1
width modulator, compensation ramp, control circuitry,
MOSFETs, and inductor), the capacitive output filter and
load, an output feedback divider, and a voltage-loop
error amplifier with its associated compensation circuitry.
See Figure 2 for a graphical representation. The power
modulator’s transfer function with respect to VCOMP is:
*CFF
C
FB
OUTPUT FILTER
AND LOAD
VIN
gMC
COMP
VFB
QHS
IL
L
CONTROL
LOGIC
VCOMP
gM
R2
PWM
COMPARATOR
RC
ROUT
QLS
DCR
VOUT
IOUT
ESR
RLOAD
COUT
CC
VCOMP
GMOD
VOUT
IL
ROUT =
REF
10
AVEA(dB)/20
gM
NOTE: THE GMOD STAGE SHOWN ABOVE MODELS THE AVERAGE CURRENT OF THE INDUCTOR, IL,
INJECTED INTO THE OUTPUT LOAD, IOUT, e.g., IL = IOUT.
SUCH CAN BE USED TO SIMPLIFY/MODEL THE MODULATION/CONTROL/POWER STAGE
CIRCUITRY SHOWN WITHIN THE BOXED AREA.
*CFF IS OPTIONAL, DESIGNED TO EXTEND THE REGULATOR’S
GAIN BANDWIDTH AND INCREASED PHASE MARGIN FOR SOME
LOW-DUTY CYCLE APPLICATIONS.
Figure 2. Peak Current-Mode Regulator Transfer Model
���������������������������������������������������������������� Maxim Integrated Products 17
MAX15112
High-Efficiency, 12A, Current-Mode Synchronous
Step-Down Regulator with Integrated Switches
The peak current-mode controller’s modulator gain is
attenuated by the equivalent divider ratio of the load
resistance and the current-loop gain. GMOD becomes:
G MOD = gMC ×
1
R LOAD
1+
× K × (1- D) - 0.5
fSW x L S
where RLOAD = VOUT/IOUT(MAX), fSW is the switching
frequency, L is the output inductance, D is the duty cycle
(VOUT/VIN), and KS is the slope compensation factor
calculated as:
V
×f
× L × gMC
K S = 1 + SLOPE SW
VIN - VOUT
where VSLOPE = 130mV and gMC = 80A/V.
The power modulator’s dominant pole is a function of the
parallel effects of the load resistance and the currentloop gain’s equivalent impedance. Assuming that ESR
of the output capacitor is much smaller than the parallel
combination of the load and the current loop, fPMOD can
be calculated as:
fPMOD =
where:
G FF (s) =
sC CR C + 1
10 AVEA(dB)/20
sC C
+1
gM
G FILTER (s) = R LOAD
GEA (s) = 10 AVEA(dB)/20 ×
sC OUTESR + 1
×
SW
1
where Q C =
π × [K S × (1- D) - 0.5]
The dominant poles and zeros of the transfer loop gain
are:
fZMOD = fZESR =
fP1 <<
fP2 =
1
2π × C OUT × ESR
The total system transfer can be written as:
GAIN(s) = G FF (s) × G EA (s) × GMOD (DC)
× G FILTER (s) × G SAMPLING (s)
-1
K × (1- D) - 0.5
1
sC OUT
+ S
+1
2
R
2π × fSW × L
π
×
LOAD
1
G SAMPLING (s) =
+1
2
s
s
+
(π × f ) 2 π × fSW × Q C
[K S × (1- D) - 0.5]
1
+
2π × C OUT × R LOAD 2π × fSW × L × C OUT
The power modulator zero is:
sC FFR1 + 1
R2
×
R1 + R2 sC FF (R1||R2) + 1
gM
2π × C C × 10 AVEA(dB)/20
1
1
K × (1- D) - 0.5
2π × C OUT
+ S
fSW × L
RLOAD
f
fP3 = SW
2
1
fZ1 =
2π × C CR C
fZ2 =
-1
1
2π × C OUTESR
The order of pole occurrence is:
fP1 < fP2 < fZ1 < fCO < fP3 < fZ2
���������������������������������������������������������������� Maxim Integrated Products 18
MAX15112
High-Efficiency, 12A, Current-Mode Synchronous
Step-Down Regulator with Integrated Switches
Figure 3 shows a graphical representation of the asymptotic system closed-loop response, including the dominant pole and zero locations.
2) Select RC using the transfer-loop’s fourth asymptote
gain equal to unity (assuming fCO > fP1, fP2, and fZ1).
RC becomes:
The loop response’s fourth asymptote (in bold, Figure 3)
is the one of interest in establishing the desired crossover
frequency (and determining the compensation component values). A lower crossover frequency provides for
stable closed-loop operation at the expense of a slower
load and line-transient response. Increasing the crossover frequency improves the transient response at the
(potential) cost of system instability. A standard rule of
thumb sets the crossover frequency P 1/5 to 1/10 of the
switching frequency.
R LOAD × K S[(1- D) - 0.5]
1 +
L × fSW
R1 + R2
×
RC =
R2
gM × gMC × R LOAD
1
× 2π × fCO × C OUT × ESR +
K S[(1- D) - 0.5]
1
+
R LOAD
L × fSW
where KS is calculated as:
Closing the Loop: Designing
the Compensation Circuitry
1) Select the desired crossover frequency. Choose fCO
equal to 1/10th of fSW, or fCO @ 100kHz.
V
×f
× L × gMC
K S = 1 + SLOPE SW
VIN - VOUT
and gM = 1.1mS, gMC = 80A/V, and VSLOPE = 130mV.
1ST ASYMPTOTE
R2 x (R1 + R2)-1 x 10AVEA(dB)/20 x gMC x RLOAD x {1 + RLOAD x [KS x (1 – D) – 0.5] x (L x fSW)-1}-1
dB
2ND ASYMPTOTE
R2 x (R1 + R2)-1 x gM x (2GCC)-1 x gMC x RLOAD x {1 + RLOAD x [KS x (1 – D) – 0.5] x (L x fSW)-1}-1
GAIN
3RD ASYMPTOTE
R2 x (R1 + R2)-1 x gM x (2GCC)-1 x gMC x RLOAD x {1 + RLOAD x [KS x (1 – D) – 0.5] x (L x fSW)-1}-1
x (2GCOUT x {RLOAD-1 + [KS(1 – D) – 0.5] x (L x fSW)-1}-1)-1
4TH ASYMPTOTE
R2 x (R1 + R2)-1 x gM x RC x gMC x RLOAD x {1 + RLOAD x [KS x (1 – D) – 0.5] x (L x fSW)-1}-1
x (2GCOUT x {RLOAD-1 + [KS(1 – D) – 0.5] x (L x fSW)-1}-1)-1
3RD POLE
2ND ZERO
0.5 x fSW (2GCOUTESR)-1
UNITY
1ST ZERO
(2GCCRC)-1
1ST POLE
[2GCC(10AVEA(dB)/20
x gM-1)]-1
FREQUENCY
fCO
2ND POLE
fPMOD*
5TH ASYMPTOTE
R2 x (R1 + R2)-1 x gM x RC x gMC x RLOAD x {1 + RLOAD x [KS x (1 – D) – 0.5] x (L x fSW)-1}-1
x [(2GCOUT x {RLOAD-1 + [KS(1 – D) – 0.5] x (L x fSW)-1}-1)-1 x (0.5 x fSW)2 x (2Gf)-2
NOTE:
ROUT = 10AVEA(dB)/20 x gM-1
*fPMOD = [2GCOUT x (ESR + {RLOAD-1 + [KS(1 – D) – 0.5] x (L x fSW)-1}-1)]-1
WHICH FOR
ESR << {RLOAD-1 + [KS(1 – D) – 0.5] x (L x fSW)-1}-1
BECOMES
fPMOD = [2GCOUT x {RLOAD-1 + [KS(1 – D) – 0.5] x (L x fSW)-1}-1]-1
fPMOD = (2GCOUT x RLOAD)-1 + [KS(1 – D) – 0.5] x (2GCOUT x L x fSW)-1
6TH ASYMPTOTE
R2 x (R1 + R2)-1 x gM x RC x gMC x RLOAD x {1 + RLOAD x [KS x (1 – D) – 0.5] x (L x fSW)-1}-1
x ESR x {RLOAD-1 + [KS(1 – D) – 0.5] x (L x fSW)-1}-1 x (0.5·fSW)2 x (2Gf)-2
Figure 3. Asymptotic Loop Response of Peak Current-Mode Regulator
���������������������������������������������������������������� Maxim Integrated Products 19
MAX15112
High-Efficiency, 12A, Current-Mode Synchronous
Step-Down Regulator with Integrated Switches
3) Select CC. CC is determined by selecting the desired
first system zero, fZ1, based on the desired phase
margin. Typically, setting fZ1 below 1/5th of fCO provides sufficient phase margin.
CC ≥
ISS, the soft-start current, is 10FA (typ) and VFB is the
0.6V (typ) output feedback voltage threshold. When
using large COUT capacitance values, the high-side
current limit can trigger during the soft-start period. To
ensure the correct soft-start time, tSS, choose CSS large
enough to satisfy:
5
2π fCO × R C
C SS >> C OUT ×
Optionally, for low duty-cycle applications, the addition
of a phase-leading capacitor (CFF in Figure 2) helps mitigate the phase lag of the damped half-frequency double
pole. Adding a second zero near to but below the desired
crossover frequency increases both the closed-loop
phase margin and the regulator’s unity-gain bandwidth
(crossover frequency). Select the capacitor as follows:
C FF =
VOUT × I SS
(24A - ILOAD ) × VFB
An external tracking reference with steady-state value
between 0V and (VIN - 2.5V) can be applied to SS/REFIN.
In this case, connect an RC network from the external
tracking reference and SS/REFIN, as shown in Figure 4.
The recommended value for RSS is approximately 330I.
RSS is needed to ensure that, during hiccup period,
SS/REFIN can be pulled down internally.
1
2π × fCO × (R1 || R2)
Using CFF, the zero-pole order is adjusted as follows:
VREF_EXT
fP1 < fP2 < fZ1 < 1/ [2πC FFR1]
RSS
SS/REFIN
CSS
< 1/ 2πC FF (R1 || R2) < fP3 < fZ2
MAX15112
Setting the Soft-Start Time
The soft-start feature ramps up the output voltage slowly,
reducing input inrush current during startup. Size the CSS
capacitor to achieve the desired soft-start time, tSS, using:
Figure 4. RC Network for External Reference at SS/REFIN
Design Examples
I ×t
C SS = SS SS
VFB
Table 1 provides values for various outputs based on the
typical operating circuit.
Table 1. Suggested Component Values (see the Typical Operating Circuits)
VOUT (V)
L (µH)
LIR (A/A)
(VIN = 3.3V)
LIR (A/A)
(VIN = 5V)
C15 (pF)
R3 (kI)
C14 (pF)
R1 (kI)
R2 (kI)
0.8
0.18
0.28
0.31
3300
5.23
22
0.74
2.21
1.2
0.22
0.29
0.35
3300
5.23
22
2.21
2.21
1.5
0.22
0.31
0.40
3300
5.23
22
3.32
2.21
1.8
0.22
0.31
—
3300
5.23
22
4.42
2.21
1.8
0.36
—
0.27
3300
5.23
22
4.42
2.21
2.5
0.22
0.23
—
3300
5.23
22
6.98
2.21
2.5
0.36
—
0.29
3300
5.23
22
6.98
2.21
3.3
0.36
—
0.26
3300
5.23
22
9.95
2.21
���������������������������������������������������������������� Maxim Integrated Products 20
MAX15112
High-Efficiency, 12A, Current-Mode Synchronous
Step-Down Regulator with Integrated Switches
Power Dissipation
The MAX15112 is available in a 24-bump WLP package
and can dissipate up to 2.15W at TA = +70NC. When
the die temperature exceeds +150NC, the thermal shut
down protection is activated (see the Thermal Shutdown
Protection section).
6) Ensure all feedback connections are short and direct.
Place the feedback resistors and compensation components as close as possible to the IC.
7) Route high-speed switching nodes (such as LX and
BST) away from sensitive analog areas (such as FB
and COMP).
Layout Procedure
Careful PCB layout is critical to achieve clean and stable
operation. It is highly recommended to duplicate the
MAX15112 Evaluation Kit layout for optimum performance. The MAX15112 EV kit board has a small, quiet,
ground-shape SGND on the back side below the IC. This
ground is the return for the control circuitry, especially
the return of the compensation components. This SGND
is returned to the IC ground through vias close to the
ground bumps of the IC. If deviation is necessary, follow
these guidelines for good PCB layout:
1) Connect a single ground plane immediately adjacent
to the GND bumps of the IC.
2) Place capacitors on IN and SS/REFIN as close as
possible to the IC and the corresponding pad using
direct traces.
3) Keep the high-current paths as short and wide as
possible. Keep the path of switching current short
and minimize the loop area formed by LX, the output
capacitors, and the input capacitors.
4) An electrolytic capacitor is strongly recommended for
damping when there is significant distance between
the input power supply and the MAX15112.
5) Connect IN, LX, and GND separately to a large copper
area to help cool the IC to further improve efficiency.
Ordering Information
PART
MAX15112EWG+
TEMP RANGE
PIN-PACKAGE
-40NC to +85NC
24 WLP
+Denotes a lead(Pb)-free/RoHS-compliant package.
Chip Information
PROCESS: BiCMOS
Package Information
For the latest package outline information and land patterns
(footprints), go to www.maxim-ic.com/packages. Note that a
“+”, “#”, or “-” in the package code indicates RoHS status only.
Package drawings may show a different suffix character, but
the drawing pertains to the package regardless of RoHS status.
PACKAGE
TYPE
PACKAGE
CODE
OUTLINE
NO.
LAND
PATTERN NO.
24 WLP
(2.1mm x
3.05mm)
W242A3Z+1
21-0538
Refer to
Application
Note 1891
���������������������������������������������������������������� Maxim Integrated Products 21
MAX15112
High-Efficiency, 12A, Current-Mode Synchronous
Step-Down Regulator with Integrated Switches
Typical Operating Circuits (continued)
EN < 0.4V = OFF
1.4V < EN < VIN = ON
CONNECT SKIP TO EN TO ENABLE SKIP MODE
CONNECT SKIP TO GND FOR PWM MODE
PGOOD
D5
C5
A5
D3
R5
100kI
C2
22µF
SKIP
C1
22µF
C19
10µF
D4
C4
B5
LX
AIN
LX
IN
LX
IN
U1
IN
MAX15112
LX
LX
N.C.
GND
C6
D6
C16
0.033µF
470I
SS/REFIN
COMP
R3
5.23kI
±1%
C14
22pF
C15
3300pF
S
R7
4.7I
C13
0.47µF
PGOOD
LX
B4
C3
22µF
BST
2.2µF
R12
10I
VIN
2.7V TO 5.5V
EN
A4
GND
GND
GND
GND
GSNS
A3
C18
1500pF
A2
C3
C2
B3
B2
D2
S
D1
L1
0.22µH
A1
C1
C7
47µF
B1
C8
47µF
C9
47µF
C10
47µF
C20
10µF
VOUT
1.5V
0 TO 12A
A6
R
S
FB
R8
1I
B6
R1
3.32kI
±1%
R2
2.21kI
±1%
SMALL-SIGNAL GND (SGND)
S 20-MIL TRACE ON THE BOTTOM THAT CONNECTS TO PGND ONLY ON
COMPONENT LAYER AT VIA NEXT TO U1.
R
REMOTE SENSE GND (RGND)
R TRACE TO REMOTE SENSE THE GND VOLTAGE AT THE LOAD.
CONNECT TO PGND ONLY AT THE LOAD.
POWER GND (PGND)
TOP LAYER GND FLOOD, SYSTEM GND.
���������������������������������������������������������������� Maxim Integrated Products 22
MAX15112
High-Efficiency, 12A, Current-Mode Synchronous
Step-Down Regulator with Integrated Switches
Revision History
REVISION
NUMBER
REVISION
DATE
0
6/11
DESCRIPTION
Initial release
PAGES
CHANGED
—
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied.
Maxim reserves the right to change the circuitry and specifications without notice at any time. The parametric values (min and max limits) shown in the Electrical
Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance.
Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600
© 2011
Maxim Integrated Products 23
Maxim is a registered trademark of Maxim Integrated Products, Inc.