MAXIM MAX8505

19-2992; Rev 0; 10/03
ILABLE
N KIT AVA
EVALUATIO
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
6m
m
-QSOP
16
General Description
The MAX8505 step-down regulator operates from a 2.6V
to 5.5V input and generates an adjustable output voltage
from 0.8V to 0.85 ✕ VIN at up to 3A. With a 2.6V to 5.5V
bias supply, the input voltage can be as low as 2.25V.
The MAX8505 integrates power MOSFETs and
operates at 1MHz/500kHz switching frequency to
provide a compact design. Current-mode pulse-widthmodulated (PWM) control simplifies compensation with
ceramic or polymer output capacitors and provides
excellent transient response.
The MAX8505 features 1% accurate output over load,
line, and temperature variations. Adjustable soft-start is
achieved with an external capacitor. During the
soft-start period, the voltage-regulation loop is active.
This limits the voltage dip when the active devices,
such as microprocessors or ASICs connected to the
MAX8505’s output, apply a sudden load current step
upon passing their undervoltage thresholds.
Features
♦ Saves Space—4.9mm x 6mm Footprint, 1µH
Inductor, 47µF Ceramic Output Capacitor
♦ Input Voltage Range
2.6V to 5.5V
Down to 2.25V with Bias Supply
♦ 0.8V to 0.85 ✕ VIN, 3A Output
♦ Ceramic or Polymer Capacitors
♦ ±1% Output Accuracy Over Load, Line, and
Temperature
♦ Fast Transient Response
♦ Adjustable Soft-Start
♦ In-Regulation Soft-Start Limits Output-Voltage
Dips at Power-On
♦ POK Monitors Output Voltage
The MAX8505 features current-limit, short-circuit, and
thermal-overload protection and enables a rugged
design. Open-drain power-OK (POK) monitors the
output voltage.
Applications
µP/ASIC/DSP/FPGA Core and I/O Supplies
Ordering Information
PART
MAX8505EEE
TEMP RANGE
PIN-PACKAGE
-40°C to +85°C
16 QSOP
Chipset Supplies
Server, RAID, and Storage Systems
Functional Diagram appears at end of data sheet.
Network and Telecom Equipment
Pin Configuration
Typical Operating Circuit
TOP VIEW
LX 1
16 LX
IN 2
15 PGND
BST
IN
LX
14 LX
LX 3
IN 4
INPUT
2.6V TO 5.5V
MAX8505
MAX8505
13 PGND
PGND
VCC
BST 5
12 GND
VCC 6
11 REF
POK 7
10 FB
CTL 8
9
COMP
OUTPUT
0.8V TO
0.85 x VIN
3A
FB
COMP
ENABLE
CTL
POWER-OK
POK
REF
GND
QSOP
________________________________________________________________ Maxim Integrated Products
For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at
1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com.
1
MAX8505
m x .9 m
4
MAX8505
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
ABSOLUTE MAXIMUM RATINGS
Operating Temperature Range
MAX8505EEE...................................................-40°C to +85°C
Storage Temperature Range .............................-65°C to +150°C
Junction Temperature ......................................................+150°C
Lead Temperature (soldering, 10s) .................................+300°C
CTL, FB, IN, VCC to GND .........................................-0.3V to +6V
COMP, REF, POK to GND ..........................-0.3V to (VCC + 0.3V)
BST to LX..................................................................-0.3V to +6V
PGND to GND .......................................................-0.3V to +0.3V
Continuous Power Dissipation (TA = +70°C)
16-Pin QSOP (derate 12.5mW/°C above +70°C).......1000mW
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(VIN = VCC = VCTL = +3.3V, VFB = 0.8V, VCOMP = 1.25V, CREF = 0.01µF, TA = 0°C to +85°C, unless otherwise noted.)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
IN AND VCC
IN Voltage Range
VIN
2.25
VCC
V
VCC Voltage Range
VCC
2.6
5.5
V
IN Supply Current
IIN
Switching with no load
VCC Supply Current
ICC
Switching with no load
Total Shutdown Current into IN
and VCC
VCC Undervoltage Lockout
Threshold
ISHDN
UVLOth
VIN = 3.3V
6
VIN = 5.5V
10
VCC = 3.3V
3
VCC = 5.5V
6
VIN = VCC = VBST - VLX = 5.5V, VCTL = 0V,
VLX = 0
When LX starts/stops
switching
VCC rising
VCC falling
10
10
20
50
2.40
2.55
mA
mA
µA
V
2.2
2.35
0.792
0.800
0.808
13
100
Ω
25
30
µA
REF
REF Voltage
VREF
IREF = 0µA, VIN = VCC = 2.6V to 5.5V
REF Shutdown Resistance
From REF to GND, VCTL = 0V
REF Soft-Start Current
VREF = 0.4V
Soft-Start Ramp Time
Output from 0% to 100%, CREF = 0.01µF to
1µF
20
32
V
ms/µF
FB
FB Regulation Voltage
VIN = 2.6V to 5.5V
FB Input Bias Current
VFB = 0.7V
Maximum Output Current
VIN = VCC = 3.3V, VOUT = 1.2V,
L = 1µH/5.9mΩ (Note 1)
IOUT_MAX
FB Threshold for POK Transition
FB rising or falling
FB to POK Delay
FB rising or falling
0.792
0.800
0.808
V
0.01
0.1
µA
3
A
FB high
10.5
12
13.5
FB low
-13.5
-12
-10.5
50
%
µs
COMP
COMP Transconductance
From FB to COMP
Gain from FB to COMP
VCOMP = 1.25V to 1.75V
2
60
100
80
_______________________________________________________________________________________
160
µS
dB
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
(VIN = VCC = VCTL = +3.3V, VFB = 0.8V, VCOMP = 1.25V, CREF = 0.01µF, TA = 0°C to +85°C, unless otherwise noted.)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
COMP Clamp Voltage, Low
VIN = VCC = 2.6V, 3.3V, 5.5V, VFB = 0.9V
0.45
0.75
1.00
V
COMP Clamp Voltage, High
VIN = VCC = 2.6V, 3.3V, 5.5V, VFB = 0.7V
1.7
1.9
2.1
V
COMP Shutdown Resistance
From COMP to GND, VCTL = 0V
13
100
Ω
VIN = VBST - VLX = 3.3V
38
74
VIN = VBST - VLX = 2.6V
42
LX (All LX Outputs Connected Together)
LX On-Resistance, High
LX On-Resistance, Low
LX Current-Sense Transresistance
LX Current-Limit Threshold
RT
VIN = VBST - VLX = 3.3V
38
VIN = VBST - VLX = 2.6V
42
From LX to COMP
Sourcing, Typical Application Circuit
Sinking, VIN = VCC = 2.6V to 5.5V
0.068
0.086
VIN = VCC = 5.5V,
VCTL = 0
LX Switching Frequency
VIN = VCC = 2.6V, 3.3V,
5.5V
LX Minimum Off-Time
VIN = VCC = 2.6V, 3.3V, 5.5V
LX Maximum Duty Cycle
VIN = VCC = 2.6V, 3.3V,
5.5V
LX Minimum Duty Cycle
VIN = VCC = 2.6V, 3.3V,
5.5V
0.104
4.6
5.6
6.6
-4.3
-2.6
-1.0
LX = 5.5V
LX Leakage Current
74
100
LX = 0V
-100
CTL = VCC
0.85
1
1.15
CTL = 2/3VCC
0.44
0.5
0.56
95
110
135
500kHz
90
94
1MHz
84
89
mΩ
mΩ
Ω
A
µA
MHz
ns
%
500kHz
5
8
1MHz
10
15
300
400
%
SLOPE COMPENSATION
Slope Compensation
Extrapolated to 100% duty cycle
245
mV
BST
BST Shutdown Supply Current
(VBST - VLX) = VIN =
VCC = 5.5V, VCTL = 0
VLX = 5.5V
10
VLX = 0V
10
LX open
10
µA
CTL
CTL Input Threshold
VIN = VCC = 2.6V,
3.3V, 5.5V
For 1MHz
80
For 500kHz
55
70
For shutdown
CTL Input Current
VCTL = 0V or 5.5V, VIN = VCC = 5.5V
45
-1
% of
VCC
+1
µA
100
mV
0.001
1
µA
50
100
µs
POK (Power-OK)
POK Output Voltage, Low
VFB = 0.6V or 1.0V, IPOK = 2mA
POK Leakage Current
VPOK = 5.5V
POK Fault Delay Time
From FB to POK, any threshold
25
25
THERMAL SHUTDOWN
Thermal-Shutdown Threshold
Thermal-Shutdown Hysteresis
When LX stops switching TJ rising
+170
°C
20
°C
_______________________________________________________________________________________
3
MAX8505
ELECTRICAL CHARACTERISTICS (continued)
MAX8505
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
ELECTRICAL CHARACTERISTICS
(VIN = VCC = VCTL = +3.3V, VFB = 0.8V, VCOMP = 1.25V, CREF = 0.01µF, TA = -40°C to +85°C, unless otherwise noted.) (Note 2)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
IN AND VCC
IN Voltage Range
VIN
VCC Voltage Range
2.25
VCC
V
2.6
5.5
V
IN Supply Current
IIN
Switching with no load
VIN = 3.3V
10
mA
VCC Supply Current
ICC
Switching with no load
VCC = 3.3V
10
mA
50
µA
Total Shutdown Current into IN
and VCC
VCC Undervoltage Lockout
Threshold
REF
REF Voltage
ISHDN
UVLOth
VREF
VIN = VCC = VBST - VLX = 5.5V, VCTL = 0V,
VLX = 0
When LX starts/stops
switching
VCC rising
VCC falling
IREF = 0µA, VIN = VCC = 2.6V to 5.5V
REF Shutdown Resistance
From REF to GND, VCTL = 0V
REF Soft-Start Current
VREF = 0.4V
2.55
2.2
0.791
0.808
V
V
100
Ω
20
30
µA
0.791
0.808
V
0.1
µA
FB
FB Regulation Voltage
VFB
FB Input Bias Current
Maximum Output Current
VIN = 2.6V to 5.5V
VFB = 0.7V
IOUT_MAX
FB Threshold for POK Transition
VIN = VCC = 3.3V, VOUT = 1.2V,
L = 1µH/5.9mΩ (Note 1)
FB rising or falling
3
A
FB high
10.5
13.5
FB low
-13.5
-10.5
%
COMP
COMP Transconductance
From FB to COMP
60
160
µS
COMP Clamp Voltage, Low
VIN = VCC = 2.6V, 3.3V, 5.5V, VFB = 0.9V
0.45
1.00
V
COMP Clamp Voltage, High
VIN = VCC = 2.6V, 3.3V, 5.5V, VFB = 0.7V
1.7
2.1
V
COMP Shutdown Resistance
From COMP to GND, VCTL = 0V
100
Ω
LX On-Resistance, High
VIN = VBST - VLX = 3.3V
74
mΩ
LX On-Resistance, Low
VIN = VBST - VLX = 3.3V
LX (All LX Outputs Connected Together)
LX Current-Sense Transresistance
LX Current-Limit Threshold
RT
74
mΩ
0.068
0.104
Ω
Sourcing, Typical Application Circuit
4.6
5.6
Sinking, VIN = VCC = 2.6V to 5.5V
-4.3
-1.0
From LX to COMP
LX Leakage Current
VIN = VCC = 5.5V,
VCTL = 0
LX = 5.5V
LX = 0V
-100
LX Switching Frequency
VIN = VCC = 2.6V,
3.3V, 5.5V
CTL = VCC
0.85
1.15
CTL = 2/3 ✕ VCC
0.44
0.56
LX Minimum Off-Time
VIN = VCC = 2.6V, 3.3V, 5.5V
95
135
4
100
_______________________________________________________________________________________
A
µA
MHz
ns
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
(VIN = VCC = VCTL = +3.3V, VFB = 0.8V, VCOMP = 1.25V, CREF = 0.01µF, TA = -40°C to +85°C, unless otherwise noted.) (Note 2)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
LX Maximum Duty Cycle
VIN = VCC = 2.6V, 3.3V,
5.5V
500kHz
90
1MHz
84
LX Minimum Duty Cycle
VIN = VCC = 2.6V, 3.3V,
5.5V
500kHz
8
1MHz
15
UNITS
%
%
SLOPE COMPENSATION
Slope Compensation
Extrapolated to 100% duty cycle
245
406
mV
BST
BST Shutdown Supply Current
(VBST - VLX) = VIN =
VCC = 5.5V, VCTL = 0
VLX = 5.5V
10
VLX = 0V
10
LX open
10
µA
CTL
CTL Input Threshold
VIN = VCC = 2.6V,
3.3V, 5.5V
For 1MHz
80
For 500kHz
55
For shutdown
CTL Input Current
VCTL = 0V or 5.5V, VIN = VCC = 5.5V
70
45
-1
% of
VCC
+1
µA
100
mV
1
µA
100
µs
POK (Power-OK)
POK Output Voltage, Low
VFB = 0.6V or 1.0V, IPOK = 2mA
POK Leakage Current
VPOK = 5.5V
POK Fault Delay Time
From FB to POK, any threshold
25
Note 1: Under normal operating conditions, COMP moves between 1.25V and 2.15V as the duty cycle changes from 10% to 90%
and peak inductor current changes from 0 to 3A. Maximum output current is related to peak inductor current, inductor value
input voltage, and output voltage by the following equations:
IOUT _ MAX =
ILIM − (1 − D) × t S × VOUT / 2L
1 + (1 − D) × t S × (RNLS + RL ) / 2L
where VOUT = output voltage; ILIM = current limit of high-side switch; tS = switching period; RL = ESR of inductor; RNLS =
on-resistance of low-side switch; L = inductor. Equations for ILIM and D are shown as follows:
ILIM = ILIM _ DC100 + VSW
1− D
RT
where ILIM_DC100 = current limit at D = 100%; RT = transresistance from LX to COMP; VSW = slope compensation (310mV
±20%); D = duty cycle:
V
+ I (R
+ RL )
D = OUT O NLS
VIN + IO (RNLS − RNHS )
where VOUT = output voltage; VIN = input voltage; IO = output current; RL = ESR of inductor; RNHS = on-resistance of highside switch; RNLS = on-resistance of low-side switch. See the Typical Application Circuit for external components.
Note 2: Specifications to -40°C are guaranteed by design and not production tested.
Note 3: LX has internal clamp diodes to PGND and IN pins 2 and 4. Applications that forward bias these diodes should take care
not to exceed the IC’s package power dissipation limits.
Note 4: When connected together, the LX output is designed to provide 3.5ARMS current.
_______________________________________________________________________________________
5
MAX8505
ELECTRICAL CHARACTERISTICS (continued)
Typical Operating Characteristics
(Typical values are at VIN = VCC = VCTL = 5V, VOUT = 1.2V, IOUT = 3A, and TA = +25°C, unless otherwise noted.)
EFFICIENCY vs. OUTPUT CURRENT
(VIN = VCC = 3.3V, fSW = 500kHz)
A
90
A
90
B
70
A: VOUT = 3.3V
B: VOUT = 2.5V
C: VOUT = 1.2V
D: VOUT = 0.8V
D
80
70
A: VOUT = 2.5V
B: VOUT = 1.8V
C: VOUT = 1.2V
D: VOUT = 0.8V
60
50
C
EFFICIENCY (%)
EFFICIENCY (%)
D
1
2
0
1
OUTPUT CURRENT (A)
2
4
3
C
80
2
3
70
MAX8505 toc05
1.05
1.03
FREQUENCY (MHz)
B
A: VOUT = 1.8V
B: VOUT = 1.2V
C: VOUT = 0.8V
+85°C
1.01
+25°C
0.99
0.97
50
-40°C
0.95
0
1
2
4
3
2.5
3.0
3.5
OUTPUT CURRENT (A)
+85°C
5.5
5.0
OUTPUT LOAD REGULATION
+25°C
500
A: VOUT = 0.8V
B: VOUT = 1.2V
C: VOUT = 1.8V
D: VOUT = 2.5V
5
-∆VOUT (mV)
510
490
4.5
6
MAX8505 toc06
530
520
4.0
INPUT VOLTAGE (V)
FREQUENCY vs. INPUT VOLTAGE AND
TEMPERATURE
FREQUENCY (kHz)
1
OUTPUT CURRENT (A)
MAX8505 toc04
A
90
EFFICIENCY (%)
0
FREQUENCY vs. INPUT VOLTAGE AND
TEMPERATURE
100
4
C
D
B
3
A
2
-40°C
480
1
470
0
VIN = VCC = 3.3V
2.5
3.0
3.5
4.0
4.5
INPUT VOLTAGE (V)
6
A: VOUT = 2.5V
B: VOUT = 1.8V
C: VOUT = 1.2V
D: VOUT = 0.8V
OUTPUT CURRENT (A)
EFFICIENCY vs. OUTPUT CURRENT
(VIN = 2.5V, VCC = 5V, fSW = 1MHz)
60
70
50
4
3
C
D
60
50
0
80
MAX8505 toc07
60
A
90
B
B
C
80
100
MAX8505 toc03
100
MAX8505 toc01
100
EFFICIENCY vs. OUTPUT CURRENT
(VIN = VCC = 3.3V, fSW = 1MHz)
MAX8505 toc02
EFFICIENCY vs. OUTPUT CURRENT
(VIN = VCC = 5V, fSW = 1MHz)
EFFICIENCY (%)
MAX8505
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
5.0
5.5
0
1
2
3
OUTPUT CURRENT (A)
_______________________________________________________________________________________
4
4
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
SHUTDOWN SUPPLY CURRENT
vs. INPUT VOLTAGE
MAX8505 toc09
5.3
CURRENT LIMIT (A)
0.2
fSW = 1MHz
5.4
0.3
0.1
5.2
5.1
5.0
4.9
4.8
4.7
4.6
fSW = 1MHz
0
4.5
4.0
4.5
5.0
5.5
0.8
1.3
1.8
2.3
OUTPUT VOLTAGE (V)
OUTPUT SHORT-CIRCUIT CURRENT
vs. INPUT VOLTAGE
GND-MEASURED TEMPERATURE
vs. OUTPUT CURRENT
4.5
3.5
120
GND-MEASURED TEMPERATURE (°C)
fSW = 1MHz
3.3
2.8
INPUT VOLTAGE (V)
MAX8505 toc10
5.5
3.5
MAX8505 toc11
3.0
2.5
OUTPUT SHORT-CIRCUIT CURRENT (A)
CURRENT LIMIT vs. OUTPUT VOLTAGE
5.5
MAX8505 toc08
SHUTDOWN SUPPLY CURRENT (mA)
0.4
TA = +85°C
100
TA = +25°C
80
60
40
VIN = 5V,
VOUT = 1.5V
20
TA = -40°C
0
2.5
2.5
3.0
3.5
4.0
4.5
5.5
5.0
3.25
3.50
3.75
4.00
OUTPUT CURRENT (A)
REFERENCE VOLTAGE
vs. TEMPERATURE
TRANSIENT RESPONSE
(VIN = 5V, VOUT = 1.2V)
MAX8505 toc13
MAX8505 toc12
0.810
REFERENCE VOLTAGE (V)
3.00
INPUT VOLTAGE (V)
OUTPUT VOLTAGE
AC-COUPLED
100mV/div
0.805
0.800
2.25A
OUTPUT
0.75A CURRENT
1A/div
0
0.795
fSW = 1MHz
0.790
-40
-15
10
35
60
85
110
40µs/div
TEMPERATURE (°C)
_______________________________________________________________________________________
7
MAX8505
Typical Operating Characteristics (continued)
(Typical values are at VIN = VCC = VCTL = 5V, VOUT = 1.2V, IOUT = 3A, and TA = +25°C, unless otherwise noted.)
MAX8505
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
Typical Operating Characteristics (continued)
(Typical values are at VIN = VCC = VCTL = 5V, VOUT = 1.2V, IOUT = 3A, and TA = +25°C, unless otherwise noted.)
TRANSIENT RESPONSE
(VIN = 3.3V, VOUT = 1.2V)
SWITCHING WAVEFORM
(VIN = 5V, VOUT = 1.2V, IOUT = 2.5A)
MAX8505 toc14
MAX8505 toc15
VLX
2V/div
OUTPUT VOLTAGE
AC-COUPLED
100mV/div
INDUCTOR CURRENT
AC-COUPLED
2A/div
2.25A
OUTPUT
0.75A CURRENT
1A/div
0
40µs/div
VOUT
AC-COUPLED
20mV/div
200ns/div
SOFT-START/SHUTDOWN WAVEFORM
(VIN = 3.3V, VOUT = 1.2V, IOUT = 3A, CREF = 0.068µF)
TRANSIENT RESPONSE DURING SOFT-START
MAX8505 toc16
MAX8505 toc17
VOUT
100mV/div
VOUT
500mV/div
VCTRL
5V/div
INPUT CURRENT
1A/div
IOUT
2A/div
VPOK
5V/div
400µs/div
8
100µs/div
_______________________________________________________________________________________
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
PIN
NAME
FUNCTION
1, 3, 14, 16
LX
Inductor Connection. Connect an inductor between these pins and the regulator output. All LX pins must
be connected together externally. Connect a 3300pF ceramic capacitor from LX to PGND.
2, 4
IN
Power-Supply Inputs. Ranges from 2.6V to 5.5V. Bypass with two ceramic 22µF capacitors to GND. All IN
pins must be connected together externally.
5
BST
Bootstrapped Voltage Input. High-side driver supply pin. Bypass to LX with a 0.1µF capacitor. Charged
from IN with an external Schottky diode.
6
VCC
Supply Voltage and Gate-Drive Supply for Low-Side Driver. Decouple with a 10Ω resistor and bypass to
GND with 0.1µF.
7
POK
Power-OK Output. Open-drain output of a window comparator that pulls POK low when the FB pin is
outside the 0.8V ±12% range.
8
CTL
Output Control. When at GND, the regulator is off. When at VCC, the regulator is operating at 1MHz. For a
500kHz application, raise the pin to 2/3 VCC.
9
COMP
Regulator Loop Compensation. Connect a series RC network to GND. This pin is pulled to GND when the
output is shut down, or in UVLO or thermal shutdown.
10
FB
Feedback Input. This pin regulates to 0.8V. Use an external resistive-divider from the output to set the
output voltage.
11
REF
Place a capacitor at this pin to set the soft-start time. This pin goes to 0V when the part is shut down.
12
GND
Ground
13, 15
PGND
Power Ground. Connect this pin to GND at a single point.
Detailed Description
The MAX8505 is a high-efficiency synchronous buck
regulator capable of delivering up to 3A of output
current. It operates in PWM mode at a high fixed
frequency of 500kHz or 1MHz, thereby reducing
external component size. The MAX8505 operates from
a 2.6V to 5.5V input voltage and can produce an output
voltage from 0.8V to 0.85 ✕ VIN.
Controller Block Function
The MAX8505 step-down converter uses a PWM
current-mode control scheme. An open-loop comparator
compares the voltage-feedback error signal against the
sum of the amplified current-sense signal and the slope
compensation ramp. At each rising edge of the internal
clock, the internal high-side MOSFET turns on until the
PWM comparator trips. During this on-time, current ramps
up through the inductor, sourcing current to the output
and storing energy in the inductor. The current-mode
feedback system regulates the peak inductor current as a
function of the output-voltage error signal. Since the average inductor current is nearly the same as the peak
inductor current, the circuit acts as a switch-mode
transconductance amplifier. To preserve inner-loop
stability and eliminate inductor staircasing, a slope-
compensation ramp is summed into the main PWM comparator. During the second half of the cycle, the internal
high-side N-channel MOSFET turns off, and the internal
low-side N-channel MOSFET turns on. The inductor
releases the stored energy as its current ramps down
while still providing current to the output. The output
capacitor stores charge when the inductor current
exceeds the load current, and discharges when the
inductor current is lower, smoothing the voltage across
the load. Under overload conditions, when the inductor
current exceeds the current limit (see the Current Limit
section), the high-side MOSFET does not turn on at the
rising edge of the clock and the low-side MOSFET
remains on to let the inductor current ramp down.
Current Sense
An internal current-sense amplifier produces a current
signal proportional to the voltage generated by the highside MOSFET on-resistance and the inductor current
(RDS(ON) ✕ ILX). The amplified current-sense signal and
the internal slope-compensation signal are summed
together into the comparator’s inverting input. The PWM
comparator turns off the internal high-side MOSFET
when this sum exceeds the feedback voltage from the
voltage-error amplifier.
_______________________________________________________________________________________
9
MAX8505
Pin Description
MAX8505
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
Current Limit
Soft-Start
The MAX8505 offers both high-side and low-side
current limits. The high-side current limit monitors the
inductor peak current and the low-side current limit
monitors the inductor valley current. Current-limit thresholds are 6A (typ) for high side and 3.8A (typ) for low
side. If the output inductor current exceeds the highside current limit during its on-time, the high-side MOSFET turns off and the synchronous rectifier turns on. The
inductor current is continuously monitored during the
on-time of the low-side MOSFET. If the inductor current
is still above the low-side current limit at the moment of
the next clock cycle, the high-side MOSFET is not
turned on and the low-side MOSFET is kept on to continue discharging the output inductor current. Once the
inductor current is below the low-side current limit, the
high-side MOSFET is turned on at the next clock cycle.
If the inductor current stays less than the high-side current limit during the minimum on duty ratio, the normal
operation resumes at the next clock cycle. Otherwise,
the current-limit operation continues.
To reduce input transient currents during startup, a programmable soft-start is provided. The soft-start time is
given by:
VCC Decoupling
Due to the high switching frequency and tight output
tolerance (1%), decouple V CC from IN with a 10Ω
resistor and bypass to GND with a 0.1µF capacitor.
Place the capacitor as close to VCC as possible.
Bootstrap (BST)
Gate-drive voltage for the high-side N-channel switch is
generated by a bootstrapped capacitor boost circuit.
The bootstrapped capacitor is connected between the
BST pin and LX. When the low-side N-channel MOSFET
is on, it forces LX to ground and charges the capacitor
to VIN through diode D1. When the low-side N-channel
MOSFET turns off and the high-side N-channel MOSFET
turns on, LX is pulled to VIN. D1 prevents the capacitor
from discharging into VIN and the voltage on the bootstrapped capacitor is boosted above VIN. This provides
the necessary voltage for the high driver. A Schottky
diode should be used for D1.
Frequency Selection/Enable (CTL)
The MAX8505 includes a frequency selection circuit to
allow it to run at 500kHz or 1MHz. The operating frequency is selected through a control input, CTL, which
has three input threshold ranges that are ratiometric to
the input supply voltage. When CTL is driven to GND, it
acts like an enable pin, switching the output off. When
the CTL input is driven to >0.8 ✕ VCC, the MAX8505 is
enabled with 1MHz switching. When the CTL input is
between 0.55 ✕ VCC and 0.7 ✕ VCC, the part operates
at 500kHz. When the CTL input is <0.45 x VCC, the
device is in shutdown.
10
t SOFT _ START = CREF ×
0.8V
25µA
A minimum capacitance of 0.01µF at REF is recommended to reduce the susceptibility to switching noise.
Power-OK (POK)
The MAX8505 also includes an open-drain POK output
that indicates when the regulator output is within ±12%
of its nominal output. If the output voltage moves
outside this range, the POK output is pulled to ground.
Since this comparator has no hysteresis on either
threshold, a 50µs delay time is added to prevent the
POK output from chattering between states. The POK
should be pulled to VIN or another supply voltage less
than 5.5V through a resistor.
UVLO
If VCC drops below +2.25V, the UVLO circuit inhibits
switching. Once VCC rises above +2.35V, the UVLO
clears, and the soft-start sequence activates.
Thermal Protection
Thermal-overload protection limits total power dissipation in the device. When the junction temperature
exceeds T J = +170°C, a thermal sensor forces the
device into shutdown, allowing the die to cool. The thermal sensor turns the device on again after the junction
temperature cools by 20°C, resulting in a pulsed output
during continuous overload conditions. Following a
thermal-shutdown condition, the soft-start sequence
begins anew.
Design Procedure
Duty Cycle
The equation below shows how to calculate the resulting duty cycle when series losses from the inductor and
internal switches are accounted for:
VOUT + IOUT (RNLS + RL ) VOUT + IOUT (RNLS + RL )
=
VIN + IOUT (RNLS − RNHS )
VIN
if RNLS = RNHS
D=
where V OUT = output voltage; V IN = input voltage;
IOUT = output current (3A maximum); RL = ESR of the
inductor; RNHS = on-resistance of the high-side switch;
and RNLS = on-resistance of the low-side switch.
______________________________________________________________________________________
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
V

R2 = R3 ×  OUT − 1
V
 REF

VRIPPLE = VRIPPLE(C)2 + VRIPPLE(ESR)2 + VRIPPLE(ESL)2
where the output ripples due to output capacitance,
ESR, and ESL are:
VRIPPLE(C) =
IP − P
8 × COUT × fS
VRIPPLE(ESR) = IP−P × ESR
where VREF = 0.8V.
Inductor Design
When choosing the inductor, the key parameters are
inductor value (L) and peak current (I PEAK ). The
following equation includes a constant, denoted as LIR,
which is the ratio of peak-to-peak inductor AC current
(ripple current) to maximum DC load current. A higher
value of LIR allows smaller inductance but results in
higher losses and ripple. A good compromise between
size and losses is found at approximately 20% to 30%
ripple-current to load-current ratio (LIR = 0.20 to 0.30):
V
× (1 − D)
L = OUT
IOUT × LIR × fS
where fS is the switching frequency and
(I
−I
)
LIR = 2 × PEAK OUT
IOUT
Choose an inductor with a saturation current at least as
high as the peak inductor current. Additionally, verify
the peak inductor current does not exceed the current
limit. The inductor selected should exhibit low losses at
the chosen operating frequency.
Output Capacitor Design and Output Ripple
The key selection parameters for the output capacitor
are capacitance, ESR, ESL, and the voltage rating
requirements. These affect the overall stability, output
ripple voltage, and transient response of the DC-DC
converter. The output ripple occurs due to variations in
the charge stored in the output capacitor, the voltage
drop due to the capacitor’s ESR, and the voltage drop
due to the capacitor’s ESL. Calculate the output voltage
ripple due to the output capacitance, ESR, and ESL as:
I
I
VRIPPLE(ESL) = P − P × ESL or P − P × ESL,
t ON
t OFF
or, whichever is greater.
The ESR is the main contribution to the output voltage
ripple.
IP-P, the peak-to-peak inductor current, is:
IP − P =
(VIN − VOUT ) VOUT
×
fS × L
VIN
Use these equations for initial capacitor selection,
but determine final values by testing a prototype or
evaluation circuit. As a rule, a smaller ripple current
results in less output voltage ripple. Since the inductor
ripple current is a factor of the inductor value, the
output voltage ripple decreases with larger inductance.
Use ceramic capacitors for their low ESR and ESL at the
switching frequency of the converter. The low ESL of
ceramic capacitors makes ripple voltages negligible.
Load-transient response depends on the selected
output capacitor. During a load transient, the output
instantly changes by ESR ✕ ILOAD. Before the controller
can respond, the output deviates further, depending on
the inductor and output capacitor values. After a short
time (see Transient Response in the Typical Operating
Characteristics), the controller responds by regulating the
output voltage back to its nominal state. The controller
response time depends on the closed-loop bandwidth,
the inductor value, and the slew rate of the transconductance amplifier. A higher bandwidth yields a faster
response time, thus preventing the output from deviating
further from its regulating value.
______________________________________________________________________________________
11
MAX8505
Output Voltage Selection
The output voltage of the MAX8505 can be adjusted
from 0.8V to 85% of the input voltage at 500kHz or up
to 80% of the input voltage at 1MHz. This is done by
connecting a resistive-divider (R2 and R3) between the
output and the FB pin (see the Typical Operating
Circuit). For best results, keep R3 below 50kΩ and
select R2 using the following equation:
MAX8505
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
Input Capacitor Design
The input capacitor reduces the current peaks drawn
from the input power supply and reduces switching
noise in the IC. The impedance of the input capacitor at
the switching frequency should be less than that of the
input source so high-frequency switching currents do
not pass through the input source but instead are
shunted through the input capacitor. A high source
impedance requires larger input capacitance. The
input capacitor must meet the ripple current requirement imposed by the switching currents. The RMS
input ripple current is given by:
IRIPPLE = ILOAD ×
VOUT × (VIN − VOUT )
VIN2
where IRIPPLE is the input RMS ripple current.
For customized compensation networks that increase
stability or transient response, the simplified loop gain
can be described by the equation:
AVOL =
VFB
× gmERR × ROERR ×
VOUT


s × CCOMP × RCOMP + 1
 (s × C
×
×
R
+
×
s
×
C
×
R
+
1
)
(
1
)


COMP
COMP
PARA
COMP


RL
s × COUT × RESR + 1
×
R T  s × COUT × RL + 1 
where:
gmERR (COMP transconductance) = 100µmho
ROERR (output resistance of transconductance
amplifier) = 20MΩ
Use sufficient input bypass capacitance to ensure that
the absolute maximum voltage rating of the MAX8505 is
not exceeded in any condition. When input supply is
not located close to the MAX8505, a bulk bypass input
capacitor may be needed.
CCOMP (compensation capacitor at COMP pin)
RT (current-sense transresistance) = 0.086Ω
Compensation Design
The double pole formed by the inductor and output
capacitor of most voltage-mode controllers introduces
a large phase shift, which requires an elaborate
compensation network to stabilize the control loop.
The MAX8505 controller utilizes a current-mode control
scheme that regulates the output voltage by forcing
the required current through the external inductor,
eliminating the double pole caused by the inductor
and output capacitor, and greatly simplifying the
compensation network. A simple type 1 compensation
with single compensation resistor (R1) and compensation capacitor (C8) create a stable and high-bandwidth
loop (see the Typical Operating Circuit).
An internal transconductance error amplifier compensates the control loop. Connect a series resistor and
capacitor between COMP (the output of the error amplifier) and GND to form a pole-zero pair. The external
inductor, internal current-sensing circuitry, output capacitor, and external compensation circuit determine the loop
stability. Choose the inductor and output capacitor based
on performance, size, and cost. Additionally, select the
compensation resistor and capacitor to optimize controlloop stability. The component values shown in the Typical
Operating Circuit yield stable operation over a broad
range of input-to-output voltages.
COUT (output capacitor)
RESR (series resistance of COUT)
12
CPARA (parasitic capacitance at COMP pin) = 10pF
RL (load resistor)
s = j2πf
In designing the compensation circuit, select an appropriate converter bandwidth (fC) to stabilize the system
while maximizing transient response. This bandwidth
should not exceed 1/10 of the switching frequency. Use
100kHz as a reasonable starting point. Calculate
CCOMP based on this bandwidth using the following
equation:
I
× RT × (R3 + R2) × 2π × fC × COUT
RCOMP = OUT
VOUT × gmERR × R2
where R2 and R3 are the feedback resistors.
Calculate CCOMP to cancel out the pole created by RL
and COUT using the following equation;
CCOMP = RL ×
COUT
RCOMP
______________________________________________________________________________________
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
PC Board Layout Considerations
Careful PC board layout is critical to achieve clean and
stable operation. The switching power stage requires
particular attention. Follow these guidelines for good
PC board layout:
1) Place decoupling capacitors as close to the IC as
possible. Keep power ground plane (connected to
PGND) and signal ground plane (connected to
GND) separate.
4) If possible, connect IN, LX, and PGND separately to
a large copper area to help cool the IC to further
improve efficiency and long-term reliability.
5) Ensure all feedback connections are short and
direct. Place the feedback resistors as close to the
IC as possible.
2) Connect input and output capacitors to the power
ground plane; connect all other capacitors to the
signal ground plane.
6) Route high-speed switching nodes away from sensitive analog areas (FB, COMP).
Chip Information
TRANSISTOR COUNT: 3352
PROCESS: BiCMOS
Typical Application Circuit
D1
(CENTRAL CMOSH-3)
L1
1µH
(FDV3H-IRON)
BST
VIN
2.6V TO 5.5V
C2
2 x 22µF
(10V CERAMIC)
C7
0.1µF
IN
R7
10Ω
LX
MAX8505
FB
R6
20kΩ
C9
COUT
3300pF
47µF
(6.3V CERAMIC)
R2
11.3kΩ
PGND
VCC
C5
0.1µF
VOUT
1.2V
R1
51kΩ
COMP
CTL
REF
POWER-OK
POK
GND
C6
0.01µF
C8
220pF
R3
22.6kΩ
______________________________________________________________________________________
13
MAX8505
3) Keep the high-current paths as short and wide as
possible. Keep the path of switching current short
and minimize the loop area formed by the high-side
MOSFET, the low-side MOSFET, and the input
capacitors. Avoid vias in the switching paths.
Applications Information
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
MAX8505
Functional Diagram
REFERENCE
1.25V
POK
MAX8505
50µs
N
VCC
FB
UVLO
BST
GND
IN
25µA
REF
PWM
LX
25µA
FB
PGND
GND
14
COMP
CTL
______________________________________________________________________________________
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
QSOP.EPS
PACKAGE OUTLINE, QSOP .150", .025" LEAD PITCH
21-0055
E
1
1
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 ____________________ 15
© 2003 Maxim Integrated Products
Printed USA
is a registered trademark of Maxim Integrated Products.
MAX8505
Package Information
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information,
go to www.maxim-ic.com/packages.)