MAXIM MAX1952ESA-18

19-2622; Rev 0; 10/02
1MHz, All-Ceramic, 2.6V to 5.5V Input,
1.5A PWM Step-Down DC-to-DC Regulators
Features
The MAX1951/MAX1952 high-efficiency, DC-to-DC
step-down switching regulators deliver up to 1.5A of
output current. The devices operate from an input voltage range of 2.6V to 5.5V and provide an output voltage from 0.8V to VIN, making the MAX1951/MAX1952
ideal for on-board postregulation applications. The
MAX1951 total output error is less than 1% over load,
line, and temperature.
The MAX1951/MAX1952 operate at a fixed frequency of
1MHz with an efficiency of up to 94%. The high operating
frequency minimizes the size of external components.
Internal soft-start control circuitry reduces inrush current.
Short-circuit and thermal-overload protection improve
design reliability.
♦ Compact 0.385in2 Circuit Footprint
The MAX1951 provides an adjustable output from 0.8V
to VIN, whereas the MAX1952 has a preset output of
1.8V. Both devices are available in a space-saving 8-pin
SO package.
♦ Internal Digital Soft-Soft
♦ 10µF Ceramic Input and Output Capacitors, 2µH
Inductor
♦ Efficiency Up to 94%
♦ 1% Output Accuracy Over Load, Line, and
Temperature (MAX1951)
♦ Guaranteed 1.5A Output Current at +85°C
♦ Operate from 2.6V to 5.5V Supply
♦ Adjustable Output from 0.8V to VIN (MAX1951)
♦ Preset Output of 1.8V (1.5% Accuracy) (MAX1952)
♦ Short-Circuit and Thermal-Overload Protection
Ordering Information
Applications
ASIC/DSP/µP/FPGA Core and I/O Voltages
PART
Set-Top Boxes
PINPACKAGE
TEMP RANGE
Cellular Base Stations
MAX1951ESA
Networking and Telecommunications
MAX1952ESA-18 -40°C to +80°C 8 SO
Pin Configuration
-40°C to +80°C 8 SO
TOP VIEW
REF
GND
2
3
MAX1951
MAX1952
FB 4
IN
7
LX
6
PGND
5
COMP
Fixed 1.8V
OUTPUT
0.8V TO VIN, 1.5A
IN
8
Adj 0.8V to VIN
Typical Operating Circuit
INPUT
2.6V TO 5.5V
VCC 1
OUTPUT
LX
MAX1951
VCC
COMP
PGND
OFF
FB
REF
GND
ON
SO
OPTIONAL
________________________________________________________________ Maxim Integrated Products
For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at
1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com.
1
MAX1951/MAX1952
General Description
MAX1951/MAX1952
1MHz, All-Ceramic, 2.6V to 5.5V Input,
1.5A PWM Step-Down DC-to-DC Regulators
ABSOLUTE MAXIMUM RATINGS
IN, VCC to GND ........................................................-0.3V to +6V
COMP, FB, REF to GND .............................-0.3V to (VCC + 0.3V)
LX to Current (Note 1).........................................................±4.5A
PGND to GND .............................................Internally Connected
Continuous Power Dissipation (TA = +85°C)
8-Pin SO (derate 12.2mW/°C above +70°C)................976mW
Operating Temperature Range
MAX195_ ESA..................................................-40°C to +85°C
Junction Temperature Range ............................-40°C to +150°C
Storage Temperature Range .............................-65°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
Note 1: LX has internal clamp diodes to PGND and IN. Applications that forward bias these diodes should take care not to exceed
the IC’s package power dissipation limits.
ELECTRICAL CHARACTERISTICS
(VIN = VCC = 3.3V, PGND = GND, FB in regulation, CREF = 0.1µF, TA = 0°C to +85°C, unless otherwise noted. Typical values are at
TA = +25°C.)
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
5.5
V
IN AND VCC
IN Voltage Range
2.6
Supply Current
Switching with no load, LX floating
Shutdown Current
COMP = GND
VCC Undervoltage Lockout
Threshold
When LX starts/stops switching
VIN = 5.5V
VCC rising
VCC falling
6
10
mA
0.5
1.0
mA
2.35
2.5
2
2.25
1.96
2
V
REF
REF Voltage
IREF = 0, VIN = 2.6V to 5.5V
2.03
V
REF Load Regulation
IREF = 0 to 40µA, VIN = 2.6V to 5.5V
0.01
0.2
%
REF Line Regulation
IREF = 20µA, VIN = 2.6V to 5.5V
0.01
0.4
%
REF Shutdown Resistance
From REF to GND, COMP = GND
12
22
Ω
COMP
MAX1951
40
60
80
MAX1952
26.7
40
53.3
COMP Transconductance
From FB to COMP, VCOMP = 1.25V
COMP Clamp Voltage, Low
VIN = 2.6V to 5.5V, VFB = 1.3V
0.6
1
1.2
COMP Clamp Voltage, High
VIN = 2.6V to 5.5V, VFB = 1.1V
1.97
2.15
2.28
V
COMP Shutdown Resistance
From COMP to GND, VIN = 2V
15
30
Ω
COMP Shutdown Threshold
When LX starts/stops switching
0.6
1
COMP Startup Current
COMP rising
COMP falling
0.17
0.4
COMP = GND
15
25
Output Voltage Range
(MAX1951)
When using external feedback resistors to drive FB
0.8
FB Regulation Voltage
(Error Amp Only)
VCOMP = 1V to 2V,
IOUT = 0 to 1.5A
40
µS
V
V
µA
FB
VIN = 2.6V to 5.5V
MAX1951
0.787
0.795
0.803
VIN = 2.8V to 5.5V
MAX1952
1.773
1.8
1.827
18
FB Input Resistance
MAX1952
13
FB Input Bias Current
MAX1951
-0.1
2
VIN
_______________________________________________________________________________________
V
V
28
kΩ
+0.1
µA
1MHz, All-Ceramic, 2.6V to 5.5V Input,
1.5A PWM Step-Down DC-to-DC Regulators
(VIN = VCC = 3.3V, PGND = GND, FB in regulation, CREF = 0.1µF, TA = 0°C to +85°C, unless otherwise noted. Typical values are at
TA = +25°C.)
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
266
mΩ
206
mΩ
0.24
0.35
Ω
3.1
4.5
LX
LX On-Resistance, PMOS
LX On-Resistance, NMOS
VIN = 5V
116
VIN = 3.3V
140
VIN = 2.6V
163
VIN = 5V
93
VIN = 3.3V
106
VIN = 2.6V
116
LX Current-Sense
Transimpedance
From LX to COMP, VIN = 2.6V to 5.5V
LX Current-Limit Threshold
Duty cycle = 100%, VIN = 2.6V to 5.5V
LX Leakage Current
VIN = 5.5V
LX Switching Frequency
VIN = 2.6V to 5.5V
0.85
LX Maximum Duty Cycle
VCOMP = 1.5V, LX = high-Z, VIN = 2.6V to 5.5V
100
LX Minimum Duty Cycle
VCOMP = 1V, VIN = 2.6V to 5.5V
0.16
High side
2.4
Low side
-0.6
VLX = 5.5V
LX = GND
10
-10
1
1.1
A
µA
MHz
%
15
%
THERMAL CHARACTERISTICS
Thermal-Shutdown Threshold
When LX starts/stops switching
TJ rising
160
TJ falling
145
°C
ELECTRICAL CHARACTERISTICS
(VIN = VCC = 3.3V, PGND = GND, FB in regulation, CREF = 0.1µF, TA = -40°C to +85°C, unless otherwise noted.) (Note 2)
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
IN AND VCC
IN Voltage Range
5.5
V
Supply Current
Switching with no load, VIN = 5.5V
2.6
10
mA
Shutdown Current
COMP = GND
1
mA
VCC Undervoltage Lockout
Threshold
When LX starts/stops switching
VCC rising
VCC falling
2.5
1.95
V
REF
REF Voltage
IREF = 0, VIN = 2.6V to 5.5V
2.03
V
REF Load Regulation
IREF = 0 to 40µA, VIN = 2.6V to 5.5V
1.95
0.2
%
REF Line Regulation
IREF = 20µA, VIN = 2.6V to 5.5V
0.4
%
REF Shutdown Resistance
From REF to GND, COMP = GND
22
Ω
_______________________________________________________________________________________
3
MAX1951/MAX1952
ELECTRICAL CHARACTERISTICS (continued)
MAX1951/MAX1952
1MHz, All-Ceramic, 2.6V to 5.5V Input,
1.5A PWM Step-Down DC-to-DC Regulators
ELECTRICAL CHARACTERISTICS (continued)
(VIN = VCC = 3.3V, PGND = GND, FB in regulation, CREF = 0.1µF, TA = -40°C to +85°C, unless otherwise noted.) (Note 2)
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
COMP
MAX1951
40
80
MAX1952
26.7
53.3
COMP Transconductance
From FB to COMP, VCOMP = 1.25V
COMP Clamp Voltage, Low
VIN = 2.6V to 5.5V, VFB = 1.3V
0.6
1.2
V
COMP Clamp Voltage, High
VIN = 2.6V to 5.5V, VFB = 1.1V
1.97
2.28
V
COMP Shutdown Resistance
From COMP to GND, VIN = 2V
30
Ω
COMP rising
1.2
µS
COMP Shutdown Threshold
When LX starts/stops switching
COMP Startup Current
COMP = GND
14
40
µA
Output Voltage Range
(MAX1951)
When using external feedback resistors to drive FB
0.8
VIN
V
FB Regulation Voltage
(Error Amp Only)
VCOMP = 1V to 2V, VIN = 2.6V to 5.5V
MAX1951
0.783
0.807
MAX1952
1.764
1.836
FB Input Resistance
From FB to GND
COMP falling
0.17
V
FB
V
MAX1952
10
30
kΩ
MAX1951
-0.1
+0.1
µA
LX On-Resistance, PMOS
266
mΩ
LX On-Resistance, NMOS
LX Current Sense
206
mΩ
From LX to COMP, VIN = 2.6V to 5.5V
0.16
0.35
Ω
LX Current-Limit Threshold
Duty cycle = 100%, VIN = 2.6V to 5.5V, high side
2.4
4.5
A
FB Input Bias Current
LX
LX Leakage Current
VIN = 5.5V
VLX = 5.5V
LX = GND
10
-10
LX Switching Frequency
VIN = 2.6V to 5.5V
0.8
LX Maximum Duty Cycle
VCOMP = 1.5V, LX = Hi-Z, VIN = 2.6V to 5.5V
100
Note 2: Specifications to -40°C are guaranteed by design and not production tested.
Note 3: The LX output is designed to provide 2A RMS current.
4
_______________________________________________________________________________________
1.1
µA
MHz
%
1MHz, All-Ceramic, 2.6V to 5.5V Input,
1.5A PWM Step-Down DC-to-DC Regulators
70
VOUT = 1.5V
50
40
30
1.994
VOUT = 1.8V
70
60
VOUT = 1.5V
50
40
30
VOUT = 0.8V
20
1.995
VOUT = 0.8V
20
10
MAX1951 toc03
80
EFFICIENCY (%)
VOUT = 2.5V
60
TA = +85°C
1.993
TA = +25°C
1.992
1.991
TA = -40°C
1.990
10
0
0
100
10
1000
10,000
1.989
100
10
LOAD CURRENT (mA)
1000
10,000
1.05
1.00
TA = +25°C
0.95
TA = -40°C
0.85
0.80
2.6
3.1
3.6
10
4.1
4.6
5.1
VOUT = 2.5V
2
1
0
-1
-2
-3
-4
-5
-6
5.6
VOUT = 0.8V
0
25
30
35
40
0.4
VOUT = 3.3V
VOUT = 1.8V
0.8
1.6
1.2
LOAD CURRENT (A)
LOAD TRANSIENT RESPONSE
LOAD TRANSIENT RESPONSE
MAX1951 toc06
MAX1951 toc07
OUTPUT VOLTAGE:
100mV/div,
AC-COUPLED
OUTPUT VOLTAGE:
100mV/div,
AC-COUPLED
OUTPUT CURRENT:
0.5A/div
OUTPUT CURRENT:
0.5A/div
VIN = 5V
VOUT = 2.5V
IOUT = 0.5 TO 1A
40µs/div
20
6
5
4
3
INPUT VOLTAGE (V)
0
15
REF OUTPUT CURRENT (µA)
MAX1951 toc05
TA = +85°C
OUTPUT VOLTAGE DEVIATION (mV)
MAX1951 toc04
1.15
0.90
5
OUTPUT VOLTAGE DEVIATION
vs. LOAD CURRENT
1.20
1.10
0
LOAD CURRENT (mA)
SWITCHING FREQUENCY
vs. INPUT VOLTAGE
SWITCHING FREQUENCY (MHz)
EFFICIENCY (%)
80
VOUT = 2.5V
90
REF VOLTAGE
vs. REF OUTPUT CURRENT
REF VOLTAGE (V)
VOUT = 3.3V
90
100
MAX 1951 toc01
100
EFFICIENCY vs. LOAD CURRENT
(VCC = VIN = 3.3V)
MAX 1951 toc02
EFFICIENCY vs. LOAD CURRENT
(VCC = VIN = 5V)
VIN = 3.3V
VOUT = 1.5V
IOUT = 0.5 TO 1A
0
40µs/div
_______________________________________________________________________________________
5
MAX1951/MAX1952
Typical Operating Characteristics
(Typical values are at VIN = VCC = 5V, VOUT = 1.5V, IOUT = 1.5A, and TA = +25°C, unless otherwise noted. See Figure 2.)
Typical Operating Characteristics (continued)
(Typical values are at VIN = VCC = 5V, VOUT = 1.5V, IOUT = 1.5A, and TA = +25°C, unless otherwise noted. See Figure 2.)
SWITCHING WAVEFORMS
SOFT-START WAVEFORMS
MAX1951 toc08
MAX1951 toc09
INDUCTOR CURRENT
1A/div
VCOMP
2V/div
0
VLX
5V/div
0
OUTPUT VOLTAGE
1V/div
OUTPUT VOLTAGE
10mV/div,
AC-COUPLED
VIN = 3.3V
VOUT = 1.8V
ILOAD = 1.5A
VIN = VCC = 3.3V
VOUT = 2.5V
ILOAD = 1.5A
200ns/div
1ms/div
SOFT-START WAVEFORMS
SHUTDOWN WAVEFORMS
MAX1951 toc10
MAX1951 toc11
0
VCOMP
2V/div
0
VLX
5V/div
VCOMP
2V/div
OUTPUT VOLTAGE
0.5V/div
VIN = VCC = 3.3V
VOUT = 2.5V
ILOAD = 1.5A
VIN = VCC = 3.3V
VOUT = 0.8V
0
1ms/div
20µs/div
SHUTDOWN CURRENT
vs. INPUT VOLTAGE
MAX1951 toc12
1.0
0.9
SHUTDOWN CURRENT (mA)
MAX1951/MAX1952
1MHz, All Ceramic, 2.6V to 5.5V Input,
1.5A PWM Step-Down DC-to-DC Regulators
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
INPUT VOLTAGE (V)
6
_______________________________________________________________________________________
OUTPUT VOLTAGE
1V/div
1MHz, All-Ceramic, 2.6V to 5.5V Input,
1.5A PWM Step-Down DC-to-DC Regulators
PIN NAME
FUNCTION
1
VCC
Supply Voltage. Bypass with 0.1µF capacitor to
ground and 10Ω resistor to IN.
2
REF
Reference Bypass. Bypass with 0.1µF
capacitor to ground.
3
GND
Ground
4
FB
Feedback Input. Connect to the output to
regulate using the internal feedback resistor
string (MAX1952). Connect an external resistordivider from the output to FB and GND to set
the output to a voltage between 0.8V and VIN
(MAX1951).
5
Regulator Compensation. Connect series RC
network to GND. Pull COMP below 0.17V to
COMP shut down the regulator. COMP = GND when
VIN is less than 2.25V (see the Compensation
and Shutdown Mode section)
6
Power Ground. Internally connected to GND.
PGND Keep power ground and signal ground planes
separate.
7
8
LX
Inductor Connection. Connect an inductor
between LX and the regulator output.
IN
Power-Supply Voltage. Input voltage range
from 2.6V to 5.5V. Bypass with a 10µF (min)
ceramic capacitor to GND and a 10Ω resistor
to VCC.
Detailed Description
The MAX1951/MAX1952 high-efficiency switching regulators are small, simple, DC-to-DC step-down converters
capable of delivering up to 1.5A of output current. The
devices operate in pulse-width modulation (PWM) at a
fixed frequency of 1MHz from a 2.6V to 5.5V input voltage
and provide an output voltage from 0.8V to VIN, making
the MAX1951/MAX1952 ideal for on-board postregulation applications. The high switching frequency allows
for the use of smaller external components, and internal
synchronous rectifiers improve efficiency and eliminate
the typical Schottky free-wheeling diode. Using the onresistance of the internal high-side MOSFET to sense
switching currents eliminates current-sense resistors,
further improving efficiency and cost. The MAX1951
total output error over load, line, and temperature (0°C
to +85°C) is less than 1%.
Controller Block Function
The MAX1951/MAX1952 step-down converters use a
PWM current-mode control scheme. An open-loop comparator compares the integrated voltage-feedback signal
against the sum of the amplified current-sense signal and
the slope compensation ramp. At each rising edge of the
internal clock, the internal high-side MOSFET turns on
until the PWM comparator trips. During this on-time, current ramps up through the inductor, sourcing current to
the output and storing energy in the inductor. The currentmode feedback system regulates the peak inductor current as a function of the output voltage error signal. Since
the average inductor current is nearly the same as the
peak inductor current (<30% ripple current), the circuit
acts as a switch-mode transconductance amplifier. To
preserve inner-loop stability and eliminate inductor staircasing, a slope-compensation ramp is summed into the
main PWM comparator. During the second half of the
cycle, the internal high-side P-channel MOSFET turns off,
and the internal low-side N-channel MOSFET turns on.
The inductor releases the stored energy as its current
ramps down while still providing current to the output. The
output capacitor stores charge when the inductor current
exceeds the load current, and discharges when the
inductor current is lower, smoothing the voltage across
the load. Under overload conditions, when the inductor
current exceeds the current limit (see the Current Limit
section), the high-side MOSFET does not turn on at the
rising edge of the clock and the low-side MOSFET
remains on to let the inductor current ramp down.
Current Sense
An internal current-sense amplifier produces a current
signal proportional to the voltage generated by the
high-side MOSFET on-resistance and the inductor current (RDS(ON) x ILX). The amplified current-sense signal
and the internal slope compensation signal are
summed together into the comparator’s inverting input.
The PWM comparator turns off the internal high-side
MOSFET when this sum exceeds the output from the
voltage-error amplifier.
Current Limit
The internal high-side MOSFET has a current limit of 3.1A
(typ). If the current flowing out of LX exceeds this limit,
the high-side MOSFET turns off and the synchronous
rectifier turns on. This lowers the duty cycle and causes
the output voltage to droop until the current limit is no
longer exceeded. A synchronous rectifier current limit of
-0.6A (typ) protects the device from current flowing into
LX. If the negative current limit is exceeded, the synchronous rectifier turns off, forcing the inductor current to flow
_______________________________________________________________________________________
7
MAX1951/MAX1952
Pin Description
MAX1951/MAX1952
1MHz, All-Ceramic, 2.6V to 5.5V Input,
1.5A PWM Step-Down DC-to-DC Regulators
POSITIVE AND NEGATIVE CURRENT LIMITS
VCC
OSC
IN
CLOCK
CURRENT SENSE
PWM
CONTROL
LX
RAMP GEN SLOPE
COMP
CLAMP
ERROR
SIGNAL
THERMAL
SHUTDOWN
gm
COMP
SOFT-START/
UVLO
PGND
FB
DAC
REF
2V
REF
BANDGAP
REF
1.25V
MAX1951
GND
Figure 1. Functional Diagram
through the high-side MOSFET body diode, back to the
input, until the beginning of the next cycle or until the
inductor current drops to zero. The MAX1951/MAX1952
utilize a pulse-skip mode to prevent overheating during
short-circuit output conditions. The device enters pulseskip mode when the FB voltage drops below 300mV, limiting the current to 3A (typ) and reducing power
dissipation. Normal operation resumes upon removal of
the short-circuit condition.
VCC Decoupling
Due to the high switching frequency and tight output
tolerance (1%), decouple VCC with a 0.1µF capacitor
connected from VCC to GND, and a 10Ω resistor connected from VCC to IN. Place the capacitor as close to
VCC as possible.
Soft-Start
The MAX1951/MAX1952 employ digital soft-start circuitry
to reduce supply inrush current during startup conditions.
When the device exits undervoltage lockout (UVLO), shutdown mode, or restarts following a thermal-overload
event, or the external pulldown on COMP is released, the
digital soft-start circuitry slowly ramps up the voltages at
REF and FB (see the Soft-Start Waveforms in the Typical
Operating Characteristics).
8
Undervoltage Lockout
If V CC drops below 2.25V, the UVLO circuit inhibits
switching. Once V CC rises above 2.35V, the UVLO
clears, and the soft-start sequence activates.
Compensation
and Shutdown Mode
The output of the internal transconductance voltage
error amplifier connects to COMP. The normal operation
voltage for COMP is 1V to 2.2V. To shut down the
MAX1951/MAX1952, use an NPN bipolar junction
transistor or a very low output capacitance open-drain
MOSFET to pull COMP to GND. Shutdown mode causes
the internal MOSFETs to stop switching, forces LX to a
high-impedance state, and shorts REF to GND.
Release COMP to exit shutdown and initiate the softstart sequence.
Thermal-Overload Protection
Thermal-overload protection limits total power dissipation
in the device. When the junction temperature exceeds TJ
= +160°C, a thermal sensor forces the device into shutdown, allowing the die to cool. The thermal sensor turns
the device on again after the junction temperature cools
by 15°C, resulting in a pulsed output during continuous
overload conditions. Following a thermal-shutdown condition, the soft-start sequence begins.
_______________________________________________________________________________________
1MHz, All-Ceramic, 2.6V to 5.5V Input,
1.5A PWM Step-Down DC-to-DC Regulators
Output Voltage Selection: Adjustable
(MAX1951) or Preset (MAX1952)
The MAX1951 provides an adjustable output voltage
between 0.8V and VIN. Connect FB to output for 0.8V
output. To set the output voltage of the MAX1951 to a
voltage greater than VFB (0.8V typ), connect the output
to FB and GND using a resistive divider, as shown in
Figure 2a. Choose R2 between 2kΩ and 20kΩ, and set
R3 according to the following equation:
R3 = R2 x [(VOUT / VFB) – 1]
The MAX1951 PWM circuitry is capable of a stable minimum duty cycle of 18%. This limits the minimum output
voltage that can be generated to 0.18 ✕ VIN. Instability
may result for VIN/VOUT ratios below 0.18.
The MAX1952 provides a preset output voltage.
Connect the output to FB, as shown in Figure 2b.
Output Inductor Design
Use a 2µH inductor with a minimum 2A-rated DC current for most applications. For best efficiency, use an
inductor with a DC resistance of less than 20mΩ and a
saturation current greater than 3A (min). See Table 2
for recommended inductors and manufacturers. For
most designs, derive a reasonable inductor value
(LINIT) from the following equation:
LINIT = VOUT x (VIN - VOUT) / (VIN x LIR x IOUT(MAX) x fSW)
where fSW is the switching frequency (1MHz typ) of the
oscillator. Keep the inductor current ripple percentage
LIR between 20% and 40% of the maximum load current for the best compromise of cost, size, and performance. Calculate the maximum inductor current as:
IL(MAX) = (1 + LIR / 2) x IOUT(MAX)
Check the final values of the inductor with the output
ripple voltage requirement. The output ripple voltage is
given by:
VRIPPLE = VOUT x (VIN - VOUT) x ESR / (VIN x LFINAL x fSW)
where ESR is the equivalent series resistance of the
output capacitors.
Input Capacitor Design
The input filter capacitor reduces peak currents drawn
from the power source and reduces noise and voltage
ripple on the input caused by the circuit’s switching.
The input capacitor must meet the ripple current
requirement (IRMS) imposed by the switching currents
defined by the following equation:
IRMS = (1/ VIN ) ×
(IOUT2 × VOUT × (VIN − VOUT ))
For duty ratios less than 0.5, the input capacitor RMS
current is higher than the calculated current. Therefore,
use a +20% margin when calculating the RMS current
at lower duty cycles. Use ceramic capacitors for their
low ESR, equivalent series inductance (ESL), and lower
cost. Choose a capacitor that exhibits less than 10°C
temperature rise at the maximum operating RMS current for optimum long-term reliability.
After determining the input capacitor, check the input
ripple voltage due to capacitor discharge when the
high-side MOSFET turns on. Calculate the input ripple
voltage as follows:
VIN_RIPPLE = (IOUT x VOUT) / (fSW x VIN x CIN)
Keep the input ripple voltage less than 3% of the input
voltage.
Output Capacitor Design
The key selection parameters for the output capacitor
are capacitance, ESR, ESL, and the voltage rating
requirements. These affect the overall stability, output
ripple voltage, and transient response of the DC-to-DC
converter. The output ripple occurs due to variations in
the charge stored in the output capacitor, the voltage
drop due to the capacitor’s ESR, and the voltage drop
due to the capacitor’s ESL. Calculate the output voltage
ripple due to the output capacitance, ESR, and ESL as:
VRIPPLE = VRIPPLE(C) + VRIPPLE(ESR) + VRIPPLE(ESL)
where the output ripple due to output capacitance,
ESR, and ESL is:
VRIPPLE(C) = IP-P / (8 x COUT x fSW)
VRIPPLE(ESR) = IP-P x ESR
VRIPPLE(ESL) = (IP-P / tON) x ESL or (IP-P / tOFF) x ESL,
whichever is greater
and IP-P the peak-to-peak inductor current is:
IP-P = [ (VIN – VOUT ) / fSW x L) ] x VOUT / VIN
Use these equations for initial capacitor selection, but
determine final values by testing a prototype or evaluation circuit. As a rule, a smaller ripple current results in
less output voltage ripple. Since the inductor ripple
current is a factor of the inductor value, the output
voltage ripple decreases with larger inductance. Use
ceramic capacitors for their low ESR and ESL at the
switching frequency of the converter. The low ESL of
ceramic capacitors makes ripple voltages negligible.
Load transient response depends on the selected
output capacitor. During a load transient, the output
instantly changes by ESR x ILOAD. Before the controller
can respond, the output deviates further, depending on
the inductor and output capacitor values. After a short
time (see the Load Transient Response graph in the
_______________________________________________________________________________________
9
MAX1951/MAX1952
Design Procedure
MAX1951/MAX1952
1MHz, All-Ceramic, 2.6V to 5.5V Input,
1.5A PWM Step-Down DC-to-DC Regulators
Typical Operating Characteristics), the controller
responds by regulating the output voltage back to its
nominal state. The controller response time depends on
the closed-loop bandwidth. A higher bandwidth yields
a faster response time, thus preventing the output from
deviating further from its regulating value.
Compensation Design
The double pole formed by the inductor and output
capacitor of most voltage-mode controllers introduces a
large phase shift, which requires an elaborate compensation network to stabilize the control loop. The MAX1951/
MAX1952 utilize a current-mode control scheme that regulates the output voltage by forcing the required current
through the external inductor, eliminating the double pole
caused by the inductor and output capacitor, and greatly
simplifying the compensation network. A simple type 1
compensation with single compensation resistor (R1) and
compensation capacitor (C2) creates a stable and highbandwidth loop.
An internal transconductance error amplifier compensates the control loop. Connect a series resistor and
capacitor between COMP (the output of the error amplifier) and GND to form a pole-zero pair. The external
inductor, internal current-sensing circuitry, output
capacitor, and the external compensation circuit determine the loop system stability. Choose the inductor and
output capacitor based on performance, size, and cost.
Additionally, select the compensation resistor and
capacitor to optimize control-loop stability. The component values shown in the typical application circuit
(Figure 2) yield stable operation over a broad range of
input-to-output voltages.
The basic regulator loop consists of a power modulator,
an output feedback divider, and an error amplifier. The
power modulator has DC gain set by gmc x RLOAD,
with a pole-zero pair set by RLOAD, the output capacitor (COUT), and its ESR. The following equations define
the power modulator:
Modulator gain:
GMOD = ∆VOUT / ∆VCOMP = gmc x RLOAD
Modulator pole frequency:
fpMOD = 1 / (2 x π x COUT x (RLOAD+ESR))
Modulator zero frequency:
fzESR = 1 / (2 x π x COUT x ESR)
where, RLOAD = VOUT / IOUT(MAX), and gmc = 4.2S.
The feedback divider has a gain of GFB = VFB / VOUT,
where VFB is equal to 0.8V. The transconductance error
amplifier has a DC gain, GEA(DC), of 70dB. The compensation capacitor, C2, and the output resistance of
the error amplifier, R OEA (20MΩ), set the dominant
10
pole. C2 and R1 set a compensation zero. Calculate the
dominant pole frequency as:
fpEA = 1 / (2πx CC x ROEA)
Determine the compensation zero frequency is:
fzEA = 1 / (2π x CC x RC)
For best stability and response performance, set the
closed-loop unity-gain frequency much higher than the
modulator pole frequency. In addition, set the closedloop crossover unity-gain frequency less than, or equal
to, 1/5 of the switching frequency. However, set the
maximum zero crossing frequency to less than 1/3 of
the zero frequency set by the output capacitance and
its ESR when using POSCAP, SPCAP, OSCON, or other
electrolytic capacitors.The loop-gain equation at the
unity-gain frequency is:
GEA(fc) x GMOD(fc) x VFB / VOUT = 1
where GEA(fc) = gmEA x R1, and GMOD(fc) = gmc x
RLOAD x fpMOD/fC, where gmEA = 60µS.
R1 calculated as:
R1 = VOUT x K / (gmEA x VFB x GMOD(fc))
where K is the correction factor due to the extra phase
introduced by the current loop at high frequencies
(>100kHz). K is related to the value of the output
capacitance (see Table 1 for values of K vs. C). Set the
error-amplifier compensation zero formed by R1 and C2
at the modulator pole frequency at maximum load. C2
is calculated as follows:
C2 = (2 x VOUT x COUT / (R1 x IOUT(MAX))
As the load current decreases, the modulator pole also
decreases; however, the modulator gain increases
accordingly, resulting in a constant closed-loop unitygain frequency. Use the following numerical example to
calculate R1 and C2 values of the typical application
circuit of Figure 2a.
Table 1. K Value
DESCRIPTION
COUT (µF)
10
22
K
Values are for output inductance from 1.2µH
0.55 to 2.2µH. Do not use output inductors larger
0.47 than 2.2µH. Use fC = 200kHz to calculate R1.
VOUT = 1.5V
IOUT(MAX) = 1.5A
COUT = 10µF
RESR = 0.010Ω
gmEA = 60µS
______________________________________________________________________________________
1MHz, All-Ceramic, 2.6V to 5.5V Input,
1.5A PWM Step-Down DC-to-DC Regulators
particular attention. Follow these guidelines for good
PC board layout:
1) Place decoupling capacitors as close to the IC as
possible. Keep power ground plane (connected to
PGND) and signal ground plane (connected to
GND) separate.
2) Connect input and output capacitors to the power
ground plane; connect all other capacitors to the
signal ground plane.
3) Keep the high-current paths as short and wide as
possible. Keep the path of switching current (C1 to IN
and C1 to PGND) short. Avoid vias in the switching
paths.
then:
R1 = VO x K / (gmEA x VFB x GMOD(fc)) = (1.5 x 0.55) /
(60 ×10-6 × 0.8 × 0.33) ≈ 51.1kΩ (1%)
C2 = (2 x VOUT × COUT) / (RC × IOUT(max) )
= (2 × 1.25 × 10 × 10-6) / (51.1k × 1.5)
≈ 209pF, choose 220pF, 10%
4) If possible, connect IN, LX, and PGND separately to
a large copper area to help cool the IC to further
improve efficiency and long-term reliability.
5) Ensure all feedback connections are short and
direct. Place the feedback resistors as close to the
IC as possible.
6) Route high-speed switching nodes away from sensitive analog areas (FB, COMP).
Applications Information
PC Board Layout Considerations
Careful PC board layout is critical to achieve clean and
stable operation. The switching power stage requires
L1
2µH
2.6V TO 5.5V
IN
R4
10Ω
C5
0.1µF
MAX1951ESA
COMP
C1
10µF
GND
REF
PGND
OFF
C2
220pF
ON
R3
14.7kΩ
1%
FB
VCC
R1
51.1kΩ
1.5V AT 1.5A
LX
Q1
R5
10kΩ
C3
0.1µF
R2
16.9kΩ
1%
C4
10µF
GND
OPTIONAL
SHUTDOWN
CONTROL
OUTPUT
COMPONENT VALUES
VOLTAGE (V) R1 (kΩ) R2 (kΩ) R3 (kΩ) C2 (pF)
220
SHORT
OPEN
0.8
33.2
220
14.7
16.9
1.5
51.1
220
30
14
2.5
82.5
220
75
24
3.3
110
Figure 2a. MAX1951 Adjustable Output Typical Application Circuit
______________________________________________________________________________________
11
MAX1951/MAX1952
gmc = 4.2S
fSWITCH = 1MHz
RLOAD = VOUT / IOUT(MAX) = 1.5V / 1.5 A = 1Ω
fpMOD = [1 / (2π x COUT x (RLOAD + RESR)]
= [1 / (2 x π ×10 ×10-6 x (1 + 0.01)] = 15.76kHz.
fzESR = [1/(2π xCOUT RESR)]
= [1 / (2 x π × 10 ×10-6 × 0.01)] = 1.59MHz.
For 2µH output inductor, pick the closed-loop unitygain crossover frequency (fC) at 200kHz. Determine the
power modulator gain at fC:
GMOD(fc) = gmc × RLOAD × fpMOD / fC = 4.2 × 1 ×
15.76kHz / 200kHz = 0.33
MAX1951/MAX1952
1MHz, All-Ceramic, 2.6V to 5.5V Input,
1.5A PWM Step-Down DC-to-DC Regulators
L1
2µH
2.6V TO 5.5V
IN
R4
10Ω
C5
0.1µF
1.8V AT 1.5A
LX
MAX1952ESA-18
FB
VCC
R1
68kΩ
COMP
C1
10µF
PGND
REF
C4
10µF
GND
OFF
C2
220pF
ON
Q1
R5
10kΩ
C3
0.1µF
GND
OPTIONAL
SHUTDOWN
CONTROL
Figure 2b. MAX1952 Fixed-Output Typical Application Circuit
12
______________________________________________________________________________________
1MHz, All-Ceramic, 2.6V to 5.5V Input,
1.5A PWM Step-Down DC-to-DC Regulators
COMPONENT (FIGURE 2)
FUNCTION
L1
Output inductor
C1
Input filtering capacitor
10µF ±20%, 6.3V X5R capacitor
Taiyo Yuden JMK316BJ106ML or
TDK C3216X5R0J106MT
C2
Compensation capacitor
220pF ±10%, 50V capacitor
Murata GRM39X7R221K050AD or
Taiyo Yuden UMK107CH221KZ
DESCRIPTION
2µH ±20% inductor
Sumida CDRH4D28-1R8 or
Toko A915AY-2R0M
C3
Reference bypass capacitor
0.1µF ±20%, 16V X7R capacitor
Taiyo Yuden EMK107BJ104MA,
TDK C1608X7R1C104K, or
Murata GRM 39X7R104K016AD
C4
Output filtering capacitor
10µF ±20%, 6.3V X5R capacitor
Taiyo Yuden JMK316BJ106ML or
TDK C3216X5R0J106MT
C5
VCC bypass capacitor
0.1µF ±20%, 16V X7R capacitor
Taiyo Yuden EMK107BJ104MA,
TDK C1608X7R1C104K, or
Murata GRM 39X7R104K016AD
R1
Loop compensation resistor
Figure 2a
R2
Feedback resistor
Figure 2a
R3
Feedback resistor
Figure 2a
R4
Bypass resistor
10Ω ±5% resistor
R5
Shutdown transistor base current bias (optional)
10kΩ ±5% resistor
Q1
Shutdown transistor (optional)
Table 3. Component Suppliers
MANUFACTURER
MAX1951/MAX1952
Table 2. External Components List
NPN bipolar junction transistor
Fairchild MMBT3904
Zetex FMMT413
Chip Information
PHONE
FAX
Murata
650-964-6321
650-964-8165
Sumida
847-545-6700
847-545-6720
Taiyo Yuden
800-348-2496
847-925-0899
TDK
847-803-6100
847-803-6296
Toko
1-800-PIK-TOKO
408-943-9790
TRANSISTOR COUNT: 2500
PROCESS: BiCMOS
______________________________________________________________________________________
13
Package Information
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information
go to www.maxim-ic.com/packages.)
DIM
A
A1
B
C
e
E
H
L
N
E
H
INCHES
MILLIMETERS
MAX
MIN
0.069
0.053
0.010
0.004
0.014
0.019
0.007
0.010
0.050 BSC
0.150
0.157
0.228
0.244
0.016
0.050
MAX
MIN
1.35
1.75
0.10
0.25
0.35
0.49
0.19
0.25
1.27 BSC
3.80
4.00
5.80
6.20
0.40
SOICN .EPS
MAX1951/MAX1952
1MHz, All-Ceramic, 2.6V to 5.5V Input,
1.5A PWM Step-Down DC-to-DC Regulators
1.27
VARIATIONS:
1
INCHES
TOP VIEW
DIM
D
D
D
MIN
0.189
0.337
0.386
MAX
0.197
0.344
0.394
MILLIMETERS
MIN
4.80
8.55
9.80
MAX
5.00
8.75
10.00
N MS012
8
AA
14
AB
16
AC
D
A
B
e
C
0 -8
A1
L
FRONT VIEW
SIDE VIEW
PROPRIETARY INFORMATION
TITLE:
PACKAGE OUTLINE, .150" SOIC
APPROVAL
DOCUMENT CONTROL NO.
21-0041
REV.
B
1
1
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
14 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600
© 2002 Maxim Integrated Products
Printed USA
is a registered trademark of Maxim Integrated Products.