LTC3803-5 Constant Frequency Current Mode Flyback DC/DC Controller in ThinSOT U FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ DESCRIPTIO VIN and VOUT Limited Only by External Components 4.8V Undervoltage Lockout Threshold Operating Junction Temperature from –40°C to 150°C Adjustable Slope Compensation Internal Soft-Start Constant Frequency 200kHz Operation ±1.5% Reference Accuracy Current Mode Operation for Excellent Line and Load Transient Response No Minimum Load Requirement Low Quiescent Current: 240µA Low Profile (1mm) SOT-23 Package U APPLICATIO S ■ ■ ■ ■ 42V and 12V Automotive Power Supplies Telecom Power Supplies Auxiliary/Housekeeping Power Supplies Power Over Ethernet The LTC®3803-5 is a constant frequency current mode flyback controller optimized for driving 4.5V and 6V-rated N-channel MOSFETs in high input voltage applications. The LTC3803-5 operates from inputs as low as 5V. Constant frequency operation is maintained down to very light loads, resulting in less low frequency noise generation over a wide range of load currents. Slope compensation can be programmed with an external resistor. The LTC3803-5 provides ±1.5% output voltage accuracy and consumes only 240µA of quiescent current. Groundreferenced current sensing allows LTC3803-5-based converters to accept input supplies beyond the LTC3803-5’s absolute maximum VCC. For simplicity, the LTC3803-5 can be powered from a high VIN through a resistor, due to its internal 8V shunt regulator. An internal undervoltage lockout shuts down the IC when the input voltage falls below 3.2V, guaranteeing at least 3.2V of gate drive to the external MOSFET. The LTC3803-5 is available in a low profile (1mm) 6-lead SOT-23 (ThinSOTTM) package. , LTC and LT are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. ThinSOT is a trademark of Linear Technology Corporation. U TYPICAL APPLICATIO Efficiency and Power Loss vs Output Power Dual Output Wide Input Range Converter VPH5-0155 7.5k LTC3803-5 PHM25NQ10T 1µF 100V ITH/RUN NGATE GND 8.06k VFB VCC SENSE 1µF 100V 4.7k B3100 0.012Ω 85 VIN = 12V 2.5 80 2.0 VIN = 24V 75 6.5V/1.2A 70 47µF 10V 65 0.1µF ALL CAPACITORS ARE X7R, TDK 38035 TA01 1.5 1.0 VIN = 48V 60 57.6k 3.0 VIN = 8V POWER LOSS (W) 22µF 10V MMBTA42 10nF 90 13V/0.3A 20mA MIN LOAD 1µF 100V 3x 22k PDZ9.1B 10MQ100N EFFICIENCY (%) VIN 6V TO 50V 0.5 VIN = 12V 0 2 6 8 4 OUTPUT POWER (W) 10 0 12 38035 TA01b 38035f 1 LTC3803-5 W U U U W W W ABSOLUTE MAXIMUM RATINGS PACKAGE/ORDER INFORMATION (Note 1) VCC to GND (Current Fed) .................... 25mA into VCC* NGATE Voltage ......................................... – 0.3V to VCC VFB, ITH/RUN Voltages ..............................– 0.3V to 3.5V SENSE Voltage ........................................... – 0.3V to 1V NGATE Peak Output Current (<10µs) ........................ 1A Operating Junction Temperature Range (Note 2) LTC3803E-5 ....................................... – 40°C to 85°C LTC3803H-5 (Note 3) ....................... – 40°C to 150°C Storage Temperature Range ................. – 65°C to 150°C Lead Temperature (Soldering, 10 sec).................. 300°C *LTC3803-5 internal clamp circuit self regulates VCC voltage to 8V. ORDER PART NUMBER TOP VIEW LTC3803HS6-5 LTC3803ES6-5 6 NGATE ITH/RUN 1 5 VCC GND 2 VFB 3 4 SENSE S6 PART MARKING S6 PACKAGE 6-LEAD PLASTIC TSOT-23 TJMAX = 150°C, θJA = 165°C/W LTBMH LTBPF Consult LTC Marketing for parts specified with wider operating temperature ranges. ELECTRICAL CHARACTERISTICS LTC3803E-5: The ● indicates specifications which apply over the full –40°C to 85°C operating junction temperature range, otherwise specifications are at TJ = 25°C. VCC = 5V, unless otherwise noted. (Note 2) SYMBOL VTURNON VTURNOFF VHYST VCLAMP1mA VCLAMP25mA ICC VITHSHDN IITHSTART VFB PARAMETER VCC Turn On Voltage VCC Turn Off Voltage VCC Hysteresis VCC Shunt Regulator Voltage VCC Shunt Regulator Voltage Input DC Supply Current Normal Operation Undervoltage Shutdown Threshold (at ITH/RUN) Start-Up Current Source Regulated Feedback Voltage gm ∆VO(LINE) ∆VO(LOAD) Error Amplifier Transconductance Output Voltage Line Regulation Output Voltage Load Regulation IFB fOSC DCON(MIN) DCON(MAX) tRISE tFALL VIMAX ISLMAX tSFST VFB Input Current Oscillator Frequency Minimum Switch On Duty Cycle Maximum Switch On Duty Cycle Gate Drive Rise Time Gate Drive Fall Time Peak Current Sense Voltage Peak Slope Compensation Output Current Soft-Start Time CONDITIONS ● ● VTURNON – VTURNOFF ICC = 1mA, VITH/RUN = 0V ICC = 25mA, VITH/RUN = 0V (Note 4) VITH/RUN = 1.3V VCC = VTURNON – 100mV VCC = VTURNON + 100mV VITH/RUN = 0V 0°C ≤ TJ ≤ 85°C (Note 5) –40°C ≤ TJ ≤ 85°C (Note 5) ITH/RUN Pin Load = ±5µA (Note 5) VTURNOFF < VCC < VCLAMP (Note 5) ITH/RUN Sinking 5µA (Note 5) ITH/RUN Sourcing 5µA (Note 5) (Note 5) VITH/RUN = 1.3V VITH/RUN = 1.3V, VFB = 0.8V VITH/RUN = 1.3V, VFB = 0.8V CLOAD = 3000pF CLOAD = 3000pF RSL = 0 (Note 6) (Note 7) ● ● ● MIN 4 3.3 0.05 6.2 6.3 ● ● ● ● 0.12 0.07 0.788 0.780 200 170 70 ● 90 TYP 4.8 4 0.8 8 8.1 MAX 5.7 4.9 240 40 0.28 0.34 0.800 0.800 333 0.1 3 3 10 200 6.5 80 40 40 100 5 0.7 350 90 0.45 0.8 0.812 0.816 500 9.9 10.3 50 230 8.5 90 115 UNITS V V V V V µA µA V µA V V µA/V mV/V mV/µA mV/µA nA kHz % % ns ns mV µA ms 38035f 2 LTC3803-5 ELECTRICAL CHARACTERISTICS LTC3803H-5: The ● indicates specifications which apply over the full –40°C to 150°C operating junction temperature range, otherwise specifications are at TA = 25°C. VCC = 5V, unless otherwise noted. (Notes 2, 3) SYMBOL VTURNON VTURNOFF VHYST VCLAMP1mA VCLAMP25mA ICC VITHSHDN IITHSTART VFB PARAMETER VCC Turn On Voltage VCC Turn Off Voltage VCC Hysteresis VCC Shunt Regulator Voltage VCC Shunt Regulator Voltage Input DC Supply Current Normal Operation Undervoltage Shutdown Threshold (at ITH/RUN) Start-Up Current Source Regulated Feedback Voltage gm ∆VO(LINE) ∆VO(LOAD) Error Amplifier Transconductance Output Voltage Line Regulation Output Voltage Load Regulation IFB fOSC DCON(MIN) DCON(MAX) tRISE tFALL VIMAX ISLMAX tSFST VFB Input Current Oscillator Frequency Minimum Switch On Duty Cycle Maximum Switch On Duty Cycle Gate Drive Rise Time Gate Drive Fall Time Peak Current Sense Voltage Peak Slope Compensation Output Current Soft-Start Time CONDITIONS ● ● VTURNON – VTURNOFF ICC = 1mA, VITH/RUN = 0V ICC = 25mA, VITH/RUN = 0V (Note 4) VITH/RUN = 1.3V VCC = VTURNON – 100mV VCC = VTURNON + 100mV VITH/RUN = 0V 0°C ≤ TJ ≤ 85°C (Note 5) –40°C ≤ TJ ≤ 150°C (Note 5) ITH/RUN Pin Load = ±5µA (Note 5) VTURNOFF < VCC < VCLAMP (Note 5) ITH/RUN Sinking 5µA (Note 5) ITH/RUN Sourcing 5µA (Note 5) (Note 5) VITH/RUN = 1.3V VITH/RUN = 1.3V, VFB = 0.8V VITH/RUN = 1.3V, VFB = 0.8V CLOAD = 3000pF CLOAD = 3000pF RSL = 0 (Note 6) (Note 7) Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: The LTC3803H-5 is guaranteed to meet specifications from –40°C to 150°C. The LTC3803E-5 is guaranteed to meet specifications from 0°C to 85°C with specifications over the –40°C to 85°C temperature range assured by design, characterization and correlation with statistical process controls. Junction temperature (TJ) is calculated from the ambient temperature TA and the power dissipation PD in the LTC3803-5 using the formula: TJ = TA + (PD • 230°C/W) Note 3: High junction temperatures degrade operating lifetimes. Operating ● ● ● MIN 3.9 3.2 0.05 6.2 6.3 ● ● ● ● 0.08 0.07 0.788 0.780 200 170 70 ● 85 TYP 4.8 4 0.8 8 8.1 MAX 5.7 4.9 240 40 0.28 0.34 0.800 0.800 333 0.1 3 3 10 200 6.5 80 40 40 100 5 0.7 350 100 0.45 1 0.812 0.820 500 10.4 10.7 50 230 8.5 90 115 UNITS V V V V V µA µA V µA V V µA/V mV/V mV/µA mV/µA nA kHz % % ns ns mV µA ms lifetime at junction temperatures greater than 125°C is derated to 1000 hours. Note 4: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. Note 5: The LTC3803-5 is tested in a feedback loop that servos VFB to the output of the error amplifier while maintaining ITH/RUN at the midpoint of the current limit range. Note 6: Peak current sense voltage is reduced dependent on duty cycle and an optional external resistor in series with the SENSE pin (RSL). For details, refer to the programmable slope compensation feature in the Applications Information section. Note 7: Guaranteed by design. 38035f 3 LTC3803-5 U W TYPICAL PERFOR A CE CHARACTERISTICS 812 808 VFB VOLTAGE (mV) VFB VOLTAGE (mV) 808 804 800 796 812 TA = 25°C VCC ≤ VCLAMP1mA 804 800 796 792 792 788 –50 –30 –10 10 30 50 70 90 110 130 150 TEMPERATURE (°C) 788 4.0 220 VCC = 5V 4.5 5.0 5.5 6.0 6.5 7.0 VCC SUPPLY VOLTAGE (V) 788 7.5 0 205 200 195 190 10 15 ICC (mA) 20 220 TA = 25°C 210 205 200 195 190 185 185 180 –50 –30 –10 10 30 50 70 90 110 130 150 TEMPERATURE (°C) 180 TA = 25°C 215 210 205 200 195 190 185 180 4.0 4.5 6.5 7.0 5.0 5.5 6.0 VCC SUPPLY VOLTAGE (V) 38035 G04 7.5 0 5 15 10 ICC (mA) 38035 G05 VCC Undervoltage Lockout Thresholds vs Temperature 10.5 5.5 10.0 20 25 38035 G06 VCC Shunt Regulator Voltage vs Temperature 6.0 25 Oscillator Frequency vs VCC Shunt Regulator Current OSCILLATOR FREQUENCY (kHz) 210 5 38035 G03 215 OSCILLATOR FREQUENCY (kHz) OSCILLATOR FREQUENCY (kHz) 796 Oscillator Frequency vs Supply Voltage 215 ICC Supply Current vs Temperature 300 VCC = 5V VITH/RUN = 1.3V 280 9.5 VTURNON VCC (V) VOLTS 800 38035 F02 Oscillator Frequency vs Temperature 5.0 804 792 38035 G01 220 TA = 25°C 808 VFB VOLTAGE (mV) VCC = 5V SUPPLY CURRENT (µA) 812 Reference Voltage vs VCC Shunt Regulator Current Reference Voltage vs Supply Voltage Reference Voltage vs Temperature 4.5 9.0 ICC = 25mA 8.5 ICC = 1mA 4.0 8.0 VTURNOFF 260 240 220 3.5 7.5 3.0 –50 –30 –10 10 30 50 70 90 110 130 150 TEMPERATURE (°C) 7.0 –50 –30 –10 10 30 50 70 90 110 130 150 TEMPERATURE (°C) 3803 G07 38035 G08 200 –50 –30 –10 10 30 50 70 90 110 130 150 TEMPERATURE (°C) 38035 G08 38035f 4 LTC3803-5 U W TYPICAL PERFOR A CE CHARACTERISTICS Start-Up ICC Supply Current vs Temperature ITH/RUN Start-Up Current Source vs Temperature 500 VCC = VTURNON – 0.1V 1000 SHUTDOWN THRESHOLD (mV) 50 40 30 20 10 ITH/RUN PIN CURRENT SOURCE (nA) 450 60 400 350 300 250 200 150 100 50 0 –50 –30 –10 10 30 50 70 90 110 130 150 TEMPERATURE (°C) 0 –50 –30 –10 10 30 50 70 90 110 130 150 TEMPERATURE (°C) Peak Current Sense Voltage vs Temperature 120 VCC = VTURNON + 0.1V 900 VITH/RUN = 0V 800 700 600 500 400 300 200 100 0 –50 –30 –10 10 30 50 70 90 110 130 150 TEMPERATURE (°C) 3803 G11 38035 G10 38035 G12 Soft-Start Time vs Temperature 1.4 VCC = 5V 115 VCC = 5V 1.2 110 SOFT-START TIME (ms) SENSE PIN VOLTAGE (mV) START-UP SUPPLY CURRENT (µA) 70 ITH/RUN Shutdown Threshold vs Temperature 105 100 95 90 1.0 0.8 0.6 0.4 0.2 85 80 –50 –30 –10 10 30 50 70 90 110 130 150 TEMPERATURE (°C) 38035 G13 0 –50 –30 –10 10 30 50 70 90 110 130 150 TEMPERATURE (°C) 38035 G14 38035f 5 LTC3803-5 U U U PI FU CTIO S ITH/RUN (Pin 1): This pin performs two functions. It serves as the error amplifier compensation point as well as the run/shutdown control input. Nominal voltage range is 0.7V to 1.9V. Forcing this pin below 0.28V causes the LTC3803-5 to shut down. In shutdown mode, the NGATE pin is held low. SENSE (Pin 4): This pin performs two functions. It monitors switch current by reading the voltage across an external current sense resistor to ground. It also injects a current ramp that develops slope compensation voltage across an optional external programming resistor. GND (Pin 2): Ground Pin. VCC (Pin 5): Supply Pin. Must be closely decoupled to GND (Pin 2). VFB (Pin 3): Receives the feedback voltage from an external resistive divider across the output. NGATE (Pin 6): Gate Drive for the External N-Channel MOSFET. This pin swings from 0V to VCC. W BLOCK DIAGRA 5 VCC 0.3µA 0.28V 800mV REFERENCE VCC SHUNT REGULATOR + SHUTDOWN COMPARATOR VCC < VTURNON – SHUTDOWN SOFTSTART CLAMP + – ERROR AMPLIFIER CURRENT COMPARATOR 3 GND 2 VCC R + VFB UNDERVOLTAGE LOCKOUT Q S – 20mV 1.2V 200kHz OSCILLATOR SWITCHING LOGIC AND BLANKING CIRCUIT GATE DRIVER NGATE SLOPE COMP CURRENT RAMP SENSE 1 6 4 ITH/RUN 38035 BD 38035f 6 LTC3803-5 U OPERATIO The LTC3803-5 is a constant frequency current mode controller for flyback, SEPIC and DC/DC boost converter applications in a tiny ThinSOT package. The LTC3803-5 is designed so that none of its pins need to come in contact with the input or output voltages of the power supply circuit of which it is a part, allowing the conversion of voltages well beyond the LTC3803-5’s absolute maximum ratings. Main Control Loop Due to space limitations, the basics of current mode DC/DC conversion will not be discussed here; instead, the reader is referred to the detailed treatment in Application Note 19, or in texts such as Abraham Pressman’s Switching Power Supply Design. Please refer to the Block Diagram and the Typical Application on the front page of this data sheet. An external resistive voltage divider presents a fraction of the output voltage to the VFB pin. The divider must be designed so that when the output is at the desired voltage, the VFB pin voltage will equal the 800mV from the internal reference. If the load current increases, the output voltage will decrease slightly, causing the VFB pin voltage to fall below 800mV. The error amplifier responds by feeding current into the ITH/RUN pin. If the load current decreases, the VFB voltage will rise above 800mV and the error amplifier will sink current away from the ITH/RUN pin. The voltage at the ITH/RUN pin commands the pulse-width modulator formed by the oscillator, current comparator and RS latch. Specifically, the voltage at the ITH/RUN pin sets the current comparator’s trip threshold. The current comparator monitors the voltage across a current sense resistor in series with the source terminal of the external MOSFET. The LTC3803-5 turns on the external power MOSFET when the internal free-running 200kHz oscillator sets the RS latch. It turns off the MOSFET when the current comparator resets the latch or when 80% duty cycle is reached, whichever happens first. In this way, the peak current levels through the flyback transformer’s primary and secondary are controlled by the ITH/RUN voltage. Since the ITH/RUN voltage is increased by the error amplifier whenever the output voltage is below nominal, and decreased whenever output voltage exceeds nominal, the voltage regulation loop is closed. For example, whenever the load current increases, output voltage will decrease slightly, and sensing this, the error amplifier raises the ITH/RUN voltage by sourcing current into the ITH/RUN pin, raising the current comparator threshold, thus increasing the peak currents through the transformer primary and secondary. This delivers more current to the load, bringing the output voltage back up. The ITH/RUN pin serves as the compensation point for the control loop. Typically, an external series RC network is connected from ITH/RUN to ground and is chosen for optimal response to load and line transients. The impedance of this RC network converts the output current of the error amplifier to the ITH/RUN voltage which sets the current comparator threshold and commands considerable influence over the dynamics of the voltage regulation loop. Start-Up/Shutdown The LTC3803-5 has two shutdown mechanisms to disable and enable operation: an undervoltage lockout on the VCC supply pin voltage, and a forced shutdown whenever external circuitry drives the ITH/RUN pin low. The LTC38035 transitions into and out of shutdown according to the state diagram (Figure 1). LTC3803-5 SHUT DOWN VCC < VTURNOFF (NOMINALLY 4V) V > VITHSHDN VITH/RUN < VITHSHDN ITH/RUN AND VCC > VTURNON (NOMINALLY 0.28V) (NOMINALLY 4.8V) LTC3803-5 ENABLED 38035 F01 Figure 1. Start-Up/Shutdown State Diagram 38035f 7 LTC3803-5 U OPERATIO The undervoltage lockout (UVLO) mechanism prevents the LTC3803-5 from trying to drive a MOSFET with insufficient VGS. The voltage at the VCC pin must exceed VTURNON (nominally 4.8V) at least momentarily to enable LTC3803-5 operation. The VCC voltage is then allowed to fall to VTURNOFF (nominally 4V) before undervoltage lockout disables the LTC3803-5. The ITH/RUN pin can be driven below VSHDN (nominally 0.28V) to force the LTC3803-5 into shutdown. An internal 0.3µA current source always tries to pull this pin towards VCC. When the ITH/RUN pin voltage is allowed to exceed VSHDN, and VCC exceeds VTURNON, the LTC3803-5 begins to operate and an internal clamp immediately pulls the ITH/RUN pin up to about 0.7V. In operation, the ITH/RUN pin voltage will vary from roughly 0.7V to 1.9V to represent current comparator thresholds from zero to maximum. Internal Soft-Start An internal soft-start feature is enabled whenever the LTC3803-5 comes out of shutdown. Specifically, the ITH/RUN voltage is clamped and is prevented from reaching maximum until roughly 0.7ms has passed. This allows the input and output currents of LTC3803-5based power supplies to rise in a smooth and controlled manner on start-up. Powering the LTC3803-5 In the simplest case, the LTC3803-5 can be powered from a high voltage supply through a resistor. A built-in shunt regulator from the VCC pin to GND will draw as much current as needed through this resistor to regulate the VCC voltage to around 8V as long as the VCC pin is not forced to sink more than 25mA. This shunt regulator is always active, even when the LTC3803-5 is in shutdown, since it serves the vital function of protecting the VCC pin from seeing too much voltage. The VCC pin must be bypassed to ground immediately adjacent to the IC pins with a ceramic or tantalum capacitor. Proper supply bypassing is necessary to supply the high transient currents required by the MOSFET gate driver. 10µF is a good starting point. Adjustable Slope Compensation The LTC3803-5 injects a 5µA peak current ramp out through its SENSE pin which can be used for slope compensation in designs that require it. This current ramp is approximately linear and begins at zero current at 6.5% duty cycle, reaching peak current at 80% duty cycle. Additional details are provided in the Applications Information section. 38035f 8 LTC3803-5 U W U U APPLICATIO S I FOR ATIO Many LTC3803-5 application circuits can be derived from the topology shown in Figure 2. The LTC3803-5 itself imposes no limits on allowed power output, input voltage VIN or desired regulated output voltage VOUT; these are all determined by the ratings on the external power components. The key factors are: Q1’s maximum drain-source voltage (BVDSS), on-resistance (RDS(ON)) and maximum drain current, T1’s saturation flux level and winding insulation breakdown voltages, CIN and COUT’s maximum working voltage, ESR, and maximum ripple current ratings, and D1 and RSENSE’s power ratings. VIN D1 T1 VOUT • RVCC CIN LPRI LSEC COUT • 5 CVCC 1 CC 2 VCC ITH/RUN NGATE LTC3803-5 GND SENSE 6 4 VFB R1 3 Q1 RSL RSENSE R2 38035 F02 Figure 2. Typical LTC3803-5 Application Circuit TRANSFORMER DESIGN CONSIDERATIONS Transformer specification and design is perhaps the most critical part of applying the LTC3803-5 successfully. In addition to the usual list of caveats dealing with high frequency power transformer design, the following should prove useful. Turns Ratios Due to the use of the external feedback resistor divider ratio to set output voltage, the user has relative freedom in selecting transformer turns ratio to suit a given application. Simple ratios of small integers, e.g., 1:1, 2:1, 3:2, etc. can be employed which yield more freedom in setting total turns and mutual inductance. Simple integer turns ratios also facilitate the use of “off-the-shelf” configurable transformers such as the Coiltronics VERSA-PACTM series in applications with high input to output voltage ratios. For example, if a 6-winding VERSA-PAC is used with three windings in series on the primary and three windings in parallel on the secondary, a 3:1 turns ratio will be achieved. Turns ratio can be chosen on the basis of desired duty cycle. However, remember that the input supply voltage plus the secondary-to-primary referred version of the flyback pulse (including leakage spike) must not exceed the allowed external MOSFET breakdown rating. SELECTING FEEDBACK RESISTOR DIVIDER VALUES Leakage Inductance The regulated output voltage is determined by the resistor divider across VOUT (R1 and R2 in Figure 2). The ratio of R2 to R1 needed to produce a desired VOUT can be calculated: Transformer leakage inductance (on either the primary or secondary) causes a voltage spike to occur after the output switch (Q1) turn-off. This is increasingly prominent at higher load currents, where more stored energy must be dissipated. In some cases a “snubber” circuit will be required to avoid overvoltage breakdown at the MOSFET’s drain node. Application Note 19 is a good reference on snubber design. R2 = VOUT – 0.8 V • R1 0.8 V Choose resistance values for R1 and R2 to be as large as possible in order to minimize any efficiency loss due to the static current drawn from VOUT, but just small enough so that when VOUT is in regulation, the error caused by the nonzero input current to the VFB pin is less than 1%. A good rule of thumb is to choose R1 to be 80k or less. A bifilar or similar winding technique is a good way to minimize troublesome leakage inductances. However, remember that this will limit the primary-to-secondary breakdown voltage, so bifilar winding is not always practical. VERSA-PAC is a trademark of Coiltronics, Inc. 38035f 9 LTC3803-5 U W U U APPLICATIO S I FOR ATIO CURRENT SENSE RESISTOR CONSIDERATIONS The external current sense resistor (RSENSE in Figure 2) allows the user to optimize the current limit behavior for the particular application. As the current sense resistor is varied from several ohms down to tens of milliohms, peak switch current goes from a fraction of an ampere to several amperes. Care must be taken to ensure proper circuit operation, especially with small current sense resistor values. For example, a peak switch current of 5A requires a sense resistor of 0.020Ω. Note that the instantaneous peak power in the sense resistor is 0.5W and it must be rated accordingly. The LTC3803-5 has only a single sense line to this resistor. Therefore, any parasitic resistance in the ground side connection of the sense resistor will increase its apparent value. In the case of a 0.020Ω sense resistor, one milliohm of parasitic resistance will cause a 5% reduction in peak switch current. So the resistance of printed circuit copper traces and vias cannot necessarily be ignored. PROGRAMMABLE SLOPE COMPENSATION The LTC3803-5 injects a ramping current through its SENSE pin into an external slope compensation resistor (RSL in Figure 2). This current ramp starts at zero right after the NGATE pin has been high for the LTC3803-5’s minimum duty cycle of 6.5%. The current rises linearly towards a peak of 5µA at the maximum duty cycle of 80%, shutting off once the NGATE pin goes low. A series resistor (RSL) connecting the SENSE pin to the current sense resistor (RSENSE) thus develops a ramping voltage drop. From the perspective of the SENSE pin, this ramping voltage adds to the voltage across the sense resistor, effectively reducing the current comparator threshold in proportion to duty cycle. This stabilizes the control loop against subharmonic oscillation. The amount of reduction in the current comparator threshold (∆VSENSE) can be calculated using the following equation: ∆VSENSE = Duty Cycle – 6.5% • 5µA • RSL 73.5% Note: LTC3803-5 enforces 6.5% < Duty Cycle < 80%. A good starting value for RSL is 5.9k, which gives a 30mV drop in current comparator threshold at 80% duty cycle. Designs not needing slope compensation may replace RSL with a short circuit. VCC SHUNT REGULATOR An internal shunt regulator allows the LTC3803-5 to be powered through a single dropping resistor from VIN to VCC, in conjunction with a bypass capacitor, CVCC, that closely decouples VCC to GND (see Figure 3). The shunt regulator can draw up to 25mA through the VCC pin to GND to drop enough voltage across RVCC to regulate VCC to around 8V. For applications where VIN is low enough such that the static power dissipation in RVCC is acceptable, using the VCC shunt regulator is the simplest way to power the LTC3803-5. EXTERNAL PREREGULATOR The circuit in Figure 4 shows another way to power the LTC3803-5. An external series preregulator consisting of series pass transistor Q1, Zener diode D1, and bias resistor RB brings VCC above the VCC turn-on threshold, enabling the LTC3803-5. 8V TO 75 VIN VIN RVCC LTC3803-5 RB 100k LTC3803-5 VCC VCC CVCC Q1 MMBTA42 D1 6.8V GND CVCC 0.1µF GND 38035 F04 38035 F03 Figure 3. Powering the LTC3803-5 Via the Internal Shunt Regulator Figure 4. Powering the LTC3803-5 with an External Preregulator 38035f 10 LTC3803-5 U TYPICAL APPLICATIO S 2W Isolated Housekeeping Telecom Converter BAS516 PRIMARY SIDE 10V, 100mA OUTPUT T1 • 2.2µF 1µF VIN 36V TO 75V • 22k 806Ω 2.2µF BAS516 9.2k 1nF BAS516 1k 1 LTC3803-5 6 ITH/RUN NGATE 2 5 3 VCC GND VFB SENSE 220k • SECONDARY SIDE 10V, 100mA OUTPUT SECONDARY SIDE GROUND FDC2512 T1: PULSE ENGINEERING PA0648 OR TYCO TTI8698 5.6k 4 1µF 0.1Ω PRIMARY GROUND 38035 TA03 U PACKAGE DESCRIPTIO S6 Package 6-Lead Plastic TSOT-23 (Reference LTC DWG # 05-08-1636) 0.62 MAX 2.90 BSC (NOTE 4) 0.95 REF 1.22 REF 3.85 MAX 2.62 REF 1.4 MIN 2.80 BSC 1.50 – 1.75 (NOTE 4) PIN ONE ID RECOMMENDED SOLDER PAD LAYOUT PER IPC CALCULATOR 0.30 – 0.45 6 PLCS (NOTE 3) 0.95 BSC 0.80 – 0.90 0.20 BSC 0.01 – 0.10 1.00 MAX DATUM ‘A’ 0.30 – 0.50 REF 0.09 – 0.20 (NOTE 3) 1.90 BSC S6 TSOT-23 0302 NOTE: 1. DIMENSIONS ARE IN MILLIMETERS 2. DRAWING NOT TO SCALE 3. DIMENSIONS ARE INCLUSIVE OF PLATING 4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR 5. MOLD FLASH SHALL NOT EXCEED 0.254mm 6. JEDEC PACKAGE REFERENCE IS MO-193 38035f Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 11 LTC3803-5 U TYPICAL APPLICATIO S 90% Efficient Synchronous Flyback Converter VIN 36V TO 72V • Q2 CIN 270k Synchronous Flyback 5VOUT Synchronous Flyback 3.3VOUT VOUT* 3.3V 1.5A T1 92 91 CO • 91 D1 2 8.06k 3 ITH/RUN 6 GATE LTC3803-5 5 VCC GND VFB SENSE 25.5k* RFB VOUT T1: PULSE ENGINEERING PA1006 Q1: FAIRCHILD FDC2512 Q2: VISHAY Si9803 4 Q1 90 • 0.1µF 560 5k 1µF 10V 90 EFFICIENCY (%) 33k 1 EFFICIENCY (%) 1n 89 89 88 87 38035 TA04a RCS 86 RCS: VISHAY OR IRC, 80mΩ D1: PHILIPS BAS516 CIN: TDK 1µF, 100V, X5R *FOR 5V OUTPUT CHANGE CO: TDK 100µF, 6.3V, X5R RFB TO 42.2k 88 0.5 1.5 1.0 OUTPUT CURRENT (A) 2.0 38035 TA04b 85 0.5 1.0 1.5 2.0 OUTPUT CURRENT (A) 2.5 38035 TA04c RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT®1425 Isolated Flyback Switching Regulator with No External Power Devices No Optoisolator or “Third Winding” Required, Up to 6W Output LT1725 General Purpose Isolated Flyback Controller No Optoisolator Required, VIN and VOUT Limited Only by External Power Components LTC1772 SOT-23 Constant Frequency Current Mode Step-Down DC/DC Controller 550kHz Switching Frequency, 2.4V to 9.8V VIN Range LTC1871 Wide Input Range, No RSENSETM Current Mode Flyback, Boost and SEPIC Controller Adjustable Switching Frequency, Programmable Undervoltage Lockout, Optional Burst Mode® Operation at Light Load LTC1872 SOT-23 Constant Frequency Current Mode Boost DC/DC Controller 550kHz Switching Frequency, 2.4V to 9.8V VIN Range LT1950 Current Mode PWM Controller Controller for Forward Converters from 30W to 300W LT1952 Current Mode PWM Controller Synchronous Controller for Forward Converters from 30W to 500W LT3420 Photoflash Capacitor Charger with Automatic Refresh Specialized Flyback Charges High Voltage Photoflash Capacitors Quickly and Efficiently LT3468/LT3468-1 Photoflash Capacitor Charger in 5-Pin SOT-23 Minimal Component Count, Uses Small Transformers; VIN from 2.5V to 16V LTC3806 Synchronous Flyback Controller High Efficiency (89%); Multiple Output with Excellent Cross Regulation LTC4441 6A N-Channel MOSFET Driver Gate Drive Adjustable from 5V to 8V, Adjustable Blanking Prevents Ringing, 10-Lead MSSOP Package Burst Mode is a registered trademark of Linear Technology Corporation. No RSENSE is a trademark of Linear Technology Corporation. 38035f 12 Linear Technology Corporation LT/TP 1104 1K • PRINTED IN THE USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2004