LT3436 3A, 800kHz Step-Up Switching Regulator U FEATURES DESCRIPTIO ■ The LT®3436 is an 800kHz monolithic boost switching regulator. A high efficiency 3A, 0.1Ω switch is included on the die together with all the control circuitry required to complete a high frequency, current-mode switching regulator. Current-mode control provides fast transient response and excellent loop stability. ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ Constant 800kHz Switching Frequency Wide Operating Voltage Range: 3V to 25V High Efficiency 0.1Ω/3A Switch 1.2V Feedback Reference Voltage ±2% Overall Output Voltage Tolerance Uses Low Profile Surface Mount External Components Low Shutdown Current: 11µA Synchronizable from 1MHz to 1.4MHz Current-Mode Control Constant Maximum Switch Current Rating at All Duty Cycles* Available in a Small Thermally Enhanced TSSOP-16 Package New design techniques achieve high efficiency at high switching frequencies over a wide operating range. A low dropout internal regulator maintains consistent performance over a wide range of inputs from 24V systems to LiIon batteries. An operating supply current of 1mA maintains high efficiency, especially at lower output currents. Shutdown reduces quiescent current to 11µA. Maximum switch current remains constant at all duty cycles. Synchronization capability allows an external logic level signal to increase the internal oscillator from 1MHz to 1.4MHz. U APPLICATIO S ■ ■ ■ ■ DSL Modems Portable Computers Battery-Powered Systems Distributed Power Full cycle-by-cycle switch current limit protection and thermal shutdown are provided. High frequency operation allows the reduction of input and output filtering components and permits the use of tiny chip inductors. The LT3436 is available in an exposed pad, 16-pin TSSOP package. , LTC and LT are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. *Protectd by U.S. Patents including 6535042, 6611131, 6498466 U TYPICAL APPLICATIO Efficiency vs Load Current 5V to 12V Boost Converter 90 VIN = 5V VOUT = 12V 3.9µH INPUT 5V 4.7µF CERAMIC VSW VIN OPEN OR HIGH = ON LT3436 SHDN SYNC OUTPUT 12V 0.9A† GND 90.9k VC FB 10nF 470pF 10k 1% 22µF CERAMIC 80 75 70 65 4.7k †MAXIMUM OUTPUT CURRENT IS SUBJECT TO THERMAL DERATING. EFFICIENCY (%) 85 B220A 60 3436 TA01 0 0.1 0.2 0.3 0.4 0.5 0.6 LOAD CURRENT (A) 0.7 0.8 3436 TA01b 3436fa 1 LT3436 W U U U W W W ABSOLUTE MAXIMUM RATINGS PACKAGE/ORDER INFORMATION (Note 1) ORDER PART NUMBER TOP VIEW Input Voltage .......................................................... 25V Switch Voltage ......................................................... 35V SHDN Pin ............................................................... 25V FB Pin Current ....................................................... 1mA SYNC Pin Current .................................................. 1mA Operating Junction Temperature Range (Note 2) LT3436E .......................................... – 40°C to 125°C Storage Temperature Range ................ – 65°C to 150°C Lead Temperature (Soldering, 10 sec)................. 300°C GND 1 16 GND VIN 2 15 NC SW 3 14 SYNC SW 4 GND 5 12 FB GND 6 11 SHDN NC 7 10 NC GND 8 9 17 LT3436EFE 13 VC FE PART MARKING GND 3436EFE FE PACKAGE 16-LEAD PLASTIC TSSOP EXPOSED PAD IS GND (PIN 17), MUST BE SOLDERED TO PCB TJMAX = 125°C, θJA = 45°C/W, θJC(PAD) = 10°C/W Consult LTC Marketing for parts specified with wider operating temperature ranges. ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VC = 0.8V, SHDN, SYNC and switch open unless otherwise noted. PARAMETER CONDITION MIN ● Recommended Operating Voltage TYP 3 MAX 25 UNITS V ● 3 4 6 Oscillator Frequency 3.3V < VIN < 25V ● 640 800 960 kHz Switch On Voltage Drop ISW = 3A ● 330 550 mV VIN Undervoltage Lockout (Note 3) ● 2.6 2.73 V VIN Supply Current ISW = 0A ● 1 1.3 mA VIN Supply Current/ISW ISW = 3A Shutdown Supply Current VSHDN = 0V, VIN = 25V, VSW = 25V Maximum Switch Current Limit 2.47 15 3V < VIN < 25V, 0.4V < VC < 0.9V FB Input Current mA/A 11 25 45 µA µA ● Feedback Voltage A 1.182 1.176 1.2 ● 1.218 1.224 V V ● 0 – 0.2 – 0.4 µA 150 350 FB to VC Voltage Gain 0.4V < VC < 0.9V FB to VC Transconductance ∆IVC = ±10µA ● 500 850 1300 µMho VC Pin Source Current VFB = 1V ● – 85 – 120 – 165 µA VC Pin Sink Current VFB = 1.4V ● 70 110 165 µA VC Pin to Switch Current Transconductance VC Pin Minimum Switching Threshold Duty Cycle = 0% VC Pin 3A ISW Threshold Maximum Switch Duty Cycle VC = 1.2V, ISW = 350mA VC = 1.2V, ISW = 1A SHDN Threshold Voltage SHDN Input Current (Shutting Down) SHDN = 60mV Above Threshold SHDN Threshold Current Hysteresis SHDN = 100mV Below Threshold SYNC Pin Resistance A/V 0.3 V 0.9 V % % 85 80 90 ● ● 1.28 1.35 1.42 V ● –7 –10 –13 µA 4 7 10 µA 1.5 2.2 V 1.4 MHz SYNC Threshold Voltage SYNC Input Frequency 4.8 1 ISYNC = 1mA 20 kΩ 3436fa 2 LT3436 ELECTRICAL CHARACTERISTICS Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: The LT3436E is guaranteed to meet performance specifications from 0°C to 125°C. Specifications over the – 40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. Note 3: Minimum input voltage is defined as the voltage where the internal regulator enters lockout. Actual minimum input voltage to maintain a regulated output will depend on output voltage and load current. See Applications Information. U W TYPICAL PERFORMANCE CHARACTERISTICS FB Voltage Switch On Voltage Drop 920 450 TA = 125°C 400 SWITCH VOLTAGE (mV) 1.210 1.205 1.200 1.195 1.190 OSCILLATOR FREQUENCY (kHz) 1.215 FB VOLTAGE (V) Oscillator Frequency 500 1.220 TA = 25°C 350 300 250 TA = –40°C 200 150 100 1.185 25 50 75 100 125 0 0.5 TEMPERATURE (°C) 1.5 2.0 1.0 SWITCH CURRENT (A) 2.5 SHDN Threshold 3.0 SHDN Supply Current TA = 25°C SHDN = 0V SHDN INPUT CURRENT (µA) VIN CURRENT (µA) 1.34 10 8 6 4 1.32 0 75 100 125 TEMPERATURE (°C) 3436 G04 50 75 100 125 –10 SHUTTING DOWN –8 –6 –4 STARTING UP –2 2 50 25 SHDN Input Current 1.38 1.36 0 –12 12 SHDN THRESHOLD (V) 740 3436 G03 14 25 770 3436 G02 1.40 0 800 TEMPERATURE (°C) 3436 G01 1.30 –50 –25 830 680 –50 –25 0 0 860 710 50 1.180 –50 –25 890 0 5 10 15 20 INPUT VOLTAGE (V) 25 30 3436 G05 0 –50 –25 50 25 75 0 TEMPERATURE (°C) 100 125 3436 G06 3436fa 3 LT3436 U W TYPICAL PERFOR A CE CHARACTERISTICS SHDN Supply Current TA = 25°C VIN = 15V 4.0 TA = 25°C 3.5 VIN CURRENT (µA) 1000 200 150 100 800 MINIMUM INPUT VOLTAGE 600 400 50 200 0 0 40 TA = 25°C SWITCH CURRENT 3.0 30 2.5 2.0 20 1.5 1.0 10 FB INPUT CURRENT (µA) VIN CURRENT (µA) 250 Current Limit Foldback Input Supply Current 1200 SWITCH PEAK CURRENT (A) 300 0.5 0 0.2 0.4 0.6 0.8 1.0 SHDN VOLTAGE (V) 1.2 1.4 0 5 10 15 20 INPUT VOLTAGE (V) 3436 G07 25 30 3436 G08 0 0 0.2 1.0 0.4 0.6 0.8 FEEDBACK VOLTAGE (V) 0 1.2 3436 G09 U U U PIN FUNCTIONS GND (Pins 1, 5, 6, 8, 9, 16, 17): Short GND pins 1, 5, 6,8, 9, 16 and the exposed pad (pin 17) on the PCB. The GND is the reference for the regulated output, so load regulation will suffer if the “ground” end of the load is not at the same voltage as the GND of the IC. This condition occurs when the load current flows through the metal path between the GND pins and the load ground point. Keep the ground path short between the GND pins and the load and use a ground plane when possible. Keep the path between the input bypass and the GND pins short. The exposed pad should be attached to a large copper area to improve thermal performance. VIN (Pin 2): This pin powers the internal circuitry and internal regulator. Keep the external bypass capacitor close to this pin. SW (Pins 3, 4): The switch pin is the collector of the onchip power NPN switch and has large currents flowing through it. Keep the traces to the switching components as short as possible to minimize radiation and voltage spikes. SHDN (Pin 11): The shutdown pin is used to turn off the regulator and to reduce input drain current to a few microamperes. The 1.35V threshold can function as an accurate undervoltage lockout (UVLO), preventing the regulator from operating until the input voltage has reached a predetermined level. Float or pull high to put the regulator in the operating mode. FB (Pin 12): The feedback pin is used to set output voltage using an external voltage divider that generates 1.2V at the pin with the desired output voltage. If required, the current limit can be reduced during start up when the FB pin is below 0.5V (see the Current Limit Foldback graph in the Typical Performance Characteristics section). An impedance of less than 5kΩ at the FB pin is needed for this feature to operate. VC (Pin 13): The VC pin is the output of the error amplifier and the input of the peak switch current comparator. It is normally used for frequency compensation, but can do double duty as a current clamp or control loop override. This pin sits at about 0.3V for very light loads and 0.9V at maximum load. SYNC (Pin 14): The sync pin is used to synchronize the internal oscillator to an external signal. It is directly logic compatible and can be driven with any signal between 20% and 80% duty cycle. The synchronizing range is equal to initial operating frequency, up to 1.4MHz. See Synchronization section in Applications Information for details. When not in use, this pin should be grounded. 3436fa 4 LT3436 W BLOCK DIAGRAM The LT3436 is a constant frequency, current-mode boost converter. This means that there is an internal clock and two feedback loops that control the duty cycle of the power switch. In addition to the normal error amplifier, there is a current sense amplifier that monitors switch current on a cycle-by-cycle basis. A switch cycle starts with an oscillator pulse which sets the RS flip-flop to turn the switch on. When switch current reaches a level set by the inverting input of the comparator, the flip-flop is reset and the switch turns off. Output voltage control is obtained by using the output of the error amplifier to set the switch current trip point. This technique means that the error amplifier commands current to be delivered to the output rather than voltage. A voltage fed system will have low phase shift up to the resonant frequency of the inductor and output capacitor, then an abrupt 180° shift will occur. The current fed system will have 90° phase shift at a much lower frequency, but will not have the additional 90° shift until well beyond the LC resonant frequency. This makes it much easier to frequency compensate the feedback loop and also gives much quicker transient response. A comparator connected to the shutdown pin disables the internal regulator, reducing supply current. INPUT 2.5V BIAS REGULATOR INTERNAL VCC SLOPE COMP Σ 0.3V 800kHz OSCILLATOR SYNC + – SHUTDOWN COMPARATOR 7µA + SW S DRIVER CIRCUITRY RS FLIP-FLOP CURRENT COMPARATOR R Q1 POWER SWITCH CURRENT SENSE AMPLIFIER VOLTAGE GAIN = 40 – + 1.35V – 0.005Ω SHDN – VC ERROR AMPLIFIER gm = 850µMho FB + 3µA 1.2V GND 3436 F01 Figure 1. Block Diagram 3436fa 5 LT3436 U U W U APPLICATIONS INFORMATION FB RESISTOR NETWORK The suggested resistance (R2) from FB to ground is 10k 1%. This reduces the contribution of FB input bias current to output voltage to less than 0.2%. The formula for the resistor (R1) from VOUT to FB is: R1 = R2(VOUT − 1. 2) 1.2 − R2(0.2µA) VSW LT3436 OUTPUT ERROR AMPLIFIER + 1.2V FB + AVX TPS, Sprague 593D 0.1 to 0.3 0.7 to 1.1 0.1 to 0.3 0.7 to 1.1 0.2 (typ) 0.5 (typ) C Case Size 3436 F02 GND Figure 2. Feedback Network OUTPUT CAPACITOR Step-up regulators supply current to the output in pulses. The rise and fall times of these pulses are very fast. The output capacitor is required to reduce the voltage ripple this causes. The RMS ripple current can be calculated from: IRIPPLE(RMS) = IOUT Table 1. Surface Mount Solid Tantalum Capacitor ESR and Ripple Current E Case Size ESR (Max, Ω ) Ripple Current (A) D Case Size R2 10k VC Tantalum capacitors are usually chosen for their bulk capacitance properties, useful in high transient load applications. ESR rather than absolute value defines output ripple at 800kHz. Values in the 22µF to 100µF range are generally needed to minimize ESR and meet ripple current ratings. Care should be taken to ensure the ripple ratings are not exceeded. AVX TPS, Sprague 593D R1 – to 22µF range. Since the absolute value of capacitance defines the pole frequency of the output stage, an X7R or X5R type ceramic, which have good temperature stability, is recommended. (VOUT − VIN ) / VIN The LT3436 will operate with both ceramic and tantalum output capacitors. Ceramic capacitors are generally chosen for their small size, very low ESR (effective series resistance), and good high frequency operation, reducing output ripple voltage. Their low ESR removes a useful zero in the loop frequency response, common to tantalum capacitors. To compensate for this, the VC loop compensation pole frequency must typically be reduced by a factor of 10. Typical ceramic output capacitors are in the 4.7µF AVX TPS INPUT CAPACITOR Unlike the output capacitor, RMS ripple current in the input capacitor is normally low enough that ripple current rating is not an issue. The current waveform is triangular, with an RMS value given by: IRIPPLE(RMS) = 0.29(VIN )(VOUT − VIN ) (L)( f)(VOUT ) At higher switching frequency, the energy storage requirement of the input capacitor is reduced so values in the range of 2.2µF to 10µF are suitable for most applications. Y5V or similar type ceramics can be used since the absolute value of capacitance is less important and has no significant effect on loop stability. If operation is required close to the minimum input voltage required by either the output or the LT3436, a larger value may be necessary. This is to prevent excessive ripple causing dips below the minimum operating voltage resulting in erratic operation. 3436fa 6 LT3436 U W U U APPLICATIONS INFORMATION INDUCTOR CHOICE AND MAXIMUM OUTPUT CURRENT When choosing an inductor, there are 2 conditions that limit the minimum inductance; required output current, and avoidance of subharmonic oscillation. The maximum output current for the LT3436 in a standard boost converter configuration with an infinitely large inductor is: IOUT (MAX) = 3A VIN • η VOUT Where η = converter efficiency (typically 0.87 at high current). As the value of inductance is reduced, ripple current increases and IOUT(MAX) is reduced. The minimum inductance for a required output current is given by: LMIN = VIN (VOUT – VIN ) ⎛ (V )(I )⎞ 2VOUT (f)⎜ 3 – OUT OUT ⎟ VIN • η ⎠ ⎝ The second condition, avoidance of subharmonic oscillation, must be met if the operating duty cycle is greater than 50%. The slope compensation circuit within the LT3436 prevents subharmonic oscillation for inductor ripple currents of up to 1.4AP-P, defining the minimum inductor value to be: The recommended minimum inductance is: LMIN = (VIN )2 (VOUT – VIN ) 0.4(VOUT )2 (IOUT )(f) The inductor value may need further adjustment for other factors such as output voltage ripple and filtering requirements. Remember also, inductance can drop significantly with DC current and manufacturing tolerance. The inductor must have a rating greater than its peak operating current to prevent saturation resulting in efficiency loss. Peak inductor current is given by: ILPEAK = (VOUT )(IOUT ) VIN (VOUT − VIN ) + VIN • η 2VOUT (L)(f) Also, consideration should be given to the DC resistance of the inductor. Inductor resistance contributes directly to the efficiency losses in the overall converter. Suitable inductors are available from Coilcraft, Coiltronics, Dale, Sumida, Toko, Murata, Panasonic and other manufactures. Table 2 PART NUMBER VIN (VOUT – VIN ) 1.4VOUT (f) These conditions define the absolute minimum inductance. However, it is generally recommended that to prevent excessive output noise, and difficulty in obtaining stability, the ripple current is no more than 40% of the average inductor current. Since inductor ripple is: IP −P RIPPLE = VIN (VOUT – VIN ) VOUT (L)(f) ISAT(DC) (Amps) DCR (Ω) HEIGHT (mm) 2.2 2.4 0.07 2.9 1.5 1.6 0.043 1.8 Coilcraft DO1608C-222 Sumida CDRH3D16-1R5 LMIN = VALUE (µH) CDRH4D18-1R0 1.0 1.7 0.035 2.0 CDC5D23-2R2 2.2 2.2 0.03 2.5 CR43-1R4 1.4 2.5 0.056 3.5 CDRH5D28-2R6 2.6 2.6 0.013 3.0 CDRH6D38-3R3 3.3 3.5 0.02 4.0 CDRH6D28-3R0 3.0 3.0 0.024 3.0 (D62F)847FY-2R4M 2.4 2.5 0.037 2.7 (D73LF)817FY-2R2M 2.2 2.7 0.03 3.0 Toko 3436fa 7 LT3436 U U W U APPLICATIONS INFORMATION CATCH DIODE The suggested catch diode (D1) is a B220A Schottky. It is rated at 2A average forward current and 20V reverse voltage. Typical forward voltage is 0.5V at 2A. The diode conducts current only during switch off time. Peak reverse voltage is equal to regulator output voltage. Average forward current in normal operation is equal to output current. SHUTDOWN AND UNDERVOLTAGE LOCKOUT LT3436 7µA 1.35V 3µA VCC SHDN C1 VH − VL 7µA 1.35V (VH − 1.35V) + 3µA R1 VH – Turn-on threshold VL – Turn-off threshold Example: switching should not start until the input is above 4.75V and is to stop if the input falls below 3.75V. VH = 4.75V VL = 3.75V 4.75V − 3.75V = 143k 7µA 1.35V R2 = = 50.4k (4.75V − 1.35V) + 3µA 143k R1 = IN R1 R1 = R2 = Figure 4 shows how to add undervoltage lockout (UVLO) to the LT3436. Typically, UVLO is used in situations where the input supply is current limited, or has a relatively high source resistance. A switching regulator draws constant power from the source, so source current increases as source voltage drops. This looks like a negative resistance load to the source and can cause the source to current limit or latch low under low source voltage conditions. UVLO prevents the regulator from operating at source voltages where these problems might occur. INPUT shutdown pin can be used. The threshold voltage of the shutdown pin comparator is 1.35V. A 3µA internal current source defaults the open pin condition to be operating (see Typical Performance Graphs). Current hysteresis is added above the SHDN threshold. This can be used to set voltage hysteresis of the UVLO using the following: R2 GND 3436 F04 Figure 4. Undervoltage Lockout An internal comparator will force the part into shutdown below the minimum VIN of 2.6V. This feature can be used to prevent excessive discharge of battery-operated systems. If an adjustable UVLO threshold is required, the Keep the connections from the resistors to the SHDN pin short and make sure that the interplane or surface capacitance to the switching nodes are minimized. If high resistor values are used, the SHDN pin should be bypassed with a 1nF capacitor to prevent coupling problems from the switch node. 3436fa 8 LT3436 U W U U APPLICATIONS INFORMATION SYNCHRONIZATION The SYNC pin, is used to synchronize the internal oscillator to an external signal. The SYNC input must pass from a logic level low, through the maximum synchronization threshold with a duty cycle between 20% and 80%. The input can be driven directly from a logic level output. The synchronizing range is equal to initial operating frequency up to 1.4MHz. This means that minimum practical sync frequency is equal to the worst-case high self-oscillating frequency (960kHz), not the typical operating frequency of 800kHz. Caution should be used when synchronizing above 1.1MHz because at higher sync frequencies the amplitude of the internal slope compensation used to prevent subharmonic switching is reduced. Higher inductor values will tend to eliminate this problem. See Frequency Compensation section for a discussion of an entirely different cause of subharmonic switching before assuming that the cause is insufficient slope compensation. Application Note 19 has more details on the theory of slope compensation. LAYOUT CONSIDERATIONS As with all high frequency switchers, when considering layout, care must be taken to achieve optimal electrical, thermal and noise performance. For maximum efficiency, switch rise and fall times are typically in the nanosecond range. To prevent noise both radiated and conducted, the high speed switching current path, shown in Figure 5, must be kept as short as possible. This is implemented in the suggested layout of Figure 6. Shortening this path will also reduce the parasitic trace inductance of approximately 25nH/inch. At switch off, this parasitic inductance produces a flyback spike across the LT3436 switch. When operating at higher currents and output voltages, with poor layout, this spike can generate voltages across the LT3436 that may exceed its absolute maximum rating. A ground plane should always be used under the switcher circuitry to prevent interplane coupling and overall noise. The VC and FB components should be kept as far away as possible from the switch node. The LT3436 pinout has been designed to aid in this. The ground for these components should be separated from the switch current path. Failure to do so will result in poor stability or subharmonic like oscillation. Board layout also has a significant effect on thermal resistance. The exposed pad is the copper plate that runs under the LT3436 die. This is the best thermal path for heat out of the package. Soldering the pad onto the board will reduce die temperature and increase the power capability of the LT3436. Provide as much copper area as possible around this pad. Adding multiple solder filled feedthroughs under and around the pad to the ground plane will also help. Similar treatment to the catch diode and inductor terminations will reduce any additional heating effects. L1 D1 C3 VOUT SW LT3436 VIN HIGH FREQUENCY SWITCHING PATH C1 LOAD GND 3436 F05 Figure 5. High Speed Switching Path 3436fa 9 LT3436 U U W U APPLICATIONS INFORMATION L1 3.9µH D1 B220A INPUT 5V C3 4.7µF CERAMIC OPEN OR HIGH = ON LT3436 SHDN SYNC OUTPUT 12V 0.8A† VSW VIN R1 90.9k VC GND FB C2 10nF R3 4.7k C4 470pF C1 22µF CERAMIC R2 10k 1% †MAXIMUM OUTPUT CURRENT IS SUBJECT TO THERMAL DERATING. INPUT L1 GND R3 C3 C2 C4 KEEP FB AND VC COMPONENTS AWAY FROM HIGH FREQUENCY, HIGH INPUT COMPONENTS D1 U1 MINIMIZE LT3436, C1, D1 LOOP C1 R2 R1 GND VOUT KELVIN SENSE VOUT PLACE FEEDTHROUGHS AROUND GROUND PIN FOR GOOD THERMAL CONDUCTIVITY SOLDER EXPOSED GROUND PAD TO BOARD Figure 6. Typical Application and Suggested Layout (Topside Only Shown) 3436fa 10 LT3436 U W U U APPLICATIONS INFORMATION The inductor must have a rating greater than its peak operating current to prevent saturation resulting in efficiency loss. Peak inductor current is given by: thermal resistance number and add in worst-case ambient temperature: (VOUT )(IOUT ) VIN (VOUT − VIN ) + VIN • η 2VOUT (L)(f) Also, consideration should be given to the DC resistance of the inductor. Inductor resistance contributes directly to the efficiency losses in the overall converter. If a true die temperature is required, a measurement of the SYNC to GND pin resistance can be used. The SYNC pin resistance across temperature must first be calibrated, with no device power, in an oven. The same measurement can then be used in operation to indicate the die temperature. THERMAL CALCULATIONS FREQUENCY COMPENSATION Power dissipation in the LT3436 chip comes from four sources: switch DC loss, switch AC loss, drive current, and input quiescent current. The following formulas show how to calculate each of these losses. These formulas assume continuous mode operation, so they should not be used for calculating efficiency at light load currents. Loop frequency compensation is performed on the output of the error amplifier (VC pin) with a series RC network. The main pole is formed by the series capacitor and the output impedance (≈500kΩ) of the error amplifier. The pole falls in the range of 2Hz to 20Hz. The series resistor creates a “zero” at 1kHz to 5kHz, which improves loop stability and transient response. A second capacitor, typically one-tenth the size of the main compensation capacitor, is sometimes used to reduce the switching frequency ripple on the VC pin. VC pin ripple is caused by output voltage ripple attenuated by the output divider and multiplied by the error amplifier. Without the second capacitor, VC pin ripple is: ILPEAK = (VOUT − VIN ) VOUT (V )(I ) = OUT OUT VIN DC, duty cycle = ISW Switch loss: ( ) PSW = (DC )(ISW )2 (RSW ) + 17n(ISW ) VOUT ( f) VIN loss: (VIN )(ISW )(DC ) + 1mA(VIN ) 50 RSW = Switch resistance (≈ 0.16Ω hot) PVIN = Example: VIN = 5V, VOUT = 12V and IOUT = 0.8A Total power dissipation = 0.34 + 0.31 + 0.11 + 0.005 = 0.77W Thermal resistance for LT3436 package is influenced by the presence of internal or backside planes. With a full plane under the package, thermal resistance will be about 40°C/W. To calculate die temperature, use the appropriate TJ = TA + θJA (PTOT) VC Pin Ripple = 1.2(VRIPPLE)(gm)(RC) (VOUT) VRIPPLE = Output ripple (VP–P) gm = Error amplifier transconductance (≈850µmho) RC = Series resistor on VC pin VOUT = DC output voltage To prevent irregular switching, VC pin ripple should be kept below 50mVP–P. Worst-case VC pin ripple occurs at maximum output load current and will also be increased if poor quality (high ESR) output capacitors are used. The addition of a 150pF capacitor on the VC pin reduces switching frequency ripple to only a few millivolts. A low value for RC will also reduce VC pin ripple, but loop phase margin may be inadequate. 3436fa 11 LT3436 U TYPICAL APPLICATIO S Load Disconnects in Shutdown D3 1N4148 L1 3.9µH VIN 5V OFF ON D2 1N4148 D1 B220A C5 0.1µF VSW VIN C3 4.7µF C6 0.1µF FB VC C2 10nF R3 4.7k VOUT 12V C7 0.8A 22µF Q1 C1 Si2306DS 4.7µF R1 90.9k LT3436 SHDN SYNC GND R4 1M R2 10k 1% C4 470pF LT3436 • TA02 3V to 20VIN 5VOUT SEPIC with Either Two Inductors or a Transformer D1 B220A L1 CDRH6D28-100 VIN 3V TO 20V + C1 OPT VIN SW C7 1µF, X5R, 25V CERAMIC C5 OPT C6 OPT R1 31.6K 1% FB SHDN SHDN LT3436 SYNC SYNC VC GND GND C1 4.7µF X5R 25V CERAMIC C3 10nF C4 470pF C2 22µF X5R 10V CERAMIC L2 CDRH6D28-100 R2 10K 1% R3 2.2k GND GND OPTION: REPLACE L1, L2 WITH TRANSFORMER CTX5-1A, CTX8-1A, CTX10-2A Maximum Load Current Increases with Input Voltage 100 1.8 90 1.6 80 1.4 70 1.2 1.0 0.8 12VIN 3.3VIN 60 5VIN 50 40 0.6 30 0.4 20 0.2 10 0 3436 TA02b Efficiency 2.0 EFFICIENCY (%) MAXIMUM LOAD CURRENT (A) VOUT 5V 0 0 2 4 6 8 10 12 14 16 18 20 VIN (V) 3436 TA02c 0 500 1.0k 1.5k LOAD CURRENT (mA) 2.0k 3436 TA02d 3436fa 12 LT3436 U TYPICAL APPLICATIO S 4V-9VIN to 5VOUT SEPIC Converter** VIN** 4V TO 9V L1A* 15µH VIN OFF ON VSW SHDN • C1 4.7µF 20V FB GND R2 31.6k 1% C2 4.7µF LT3436 + D1 B220A • VC + L1B* 15µH R1 2.2k C4 15nF R3 10k 1% C5 470pF IOUT 0.84A 1.03A 1.18A 1.29A 1.50A C3 47µF 10V LT3436 • TA03 †MAX I OUT * COILTRONICS CTX15-4 ** INPUT VOLTAGE MAY BE GREATER OR LESS THAN OUTPUT VOLTAGE VOUT† 5V VIN 4V 5V 6V 7V 9V Boost Converter Drives Luxeon III 1A 3.6V White LED with 70% Efficiency 0.05Ω 1% VIN 3.3V TO 4.2V 1A CONSTANT CURRENT LXHL-PW09 EMITTER VOUT = VIN + VLED UPS120 L1 49.9Ω 1% VIN VIN SHDN LT3436 LED ON + SW LT1783 FB – SYNC VC GND GND 4.7µF X5R 6.3V CERAMIC VOUT Q1 Q2 78.7k 22µF X5R 10V CERAMIC 0.1µF 8.2k 4.99k 1.21k 1% GND 3436 TA03b Q1: MMBT2222A Q2: FMMT3906 L1: CDRH6D28-3R0 3436fa 13 LT3436 U TYPICAL APPLICATIO S Single Li-Ion Cell to 5V D1 B220A L1 4.7µH VOUT 5V VSW R1 31.6k 1% FB + VIN OFF ON SHDN LT3436 + SINGLE Li-Ion CELL + C1 10µF VC GND C4 47µF 10V R2 10k 1% C2 3.3nF R3 1.5k C3 470pF LT3436 • TA04 IOUT VIN 1.5A 2.7V 1.86A 3.3V 2.0A 3.6V SEPIC Converter Drives 5W LumiLEDs Luxeon V White LEDs at 70% Efficiency D1 B130A L1 VIN 3.6V TO 17V VOUT CCOUP 2.2µF, X5R, 25V CERAMIC D2 L2 VIN LED ON VIN SHDN LT3436 + SW FB – R5 23.7k SYNC VC GND GND C1 4.7µF X5R 25V CERAMIC 700mA LT1783 C4 0.1µF Q1 8.2k R7 124k VOUT R6 4.99k R4 1k 1% R2 0.068Ω 1% C2 22µF X5R 16V CERAMIC GND 3436 TA04b Q1: DIODES, INC. MMBT2222A L1: CDRH6D28 10µH 1.7A L2: CDRH4D28 10µH 1A D2: LUMILEDS LXHL-PW03 EMITTER OR LXHL-LW6C STAR 3436fa 14 LT3436 U PACKAGE DESCRIPTION FE Package 16-Lead Plastic TSSOP (4.4mm) (Reference LTC DWG # 05-08-1663) Exposed Pad Variation BB 4.90 – 5.10* (.193 – .201) 3.58 (.141) 3.58 (.141) 16 1514 13 12 1110 6.60 ±0.10 9 2.94 (.116) 4.50 ±0.10 2.94 6.40 (.116) (.252) BSC SEE NOTE 4 0.45 ±0.05 1.05 ±0.10 0.65 BSC 1 2 3 4 5 6 7 8 RECOMMENDED SOLDER PAD LAYOUT 4.30 – 4.50* (.169 – .177) 0.09 – 0.20 (.0035 – .0079) 0.50 – 0.75 (.020 – .030) NOTE: 1. CONTROLLING DIMENSION: MILLIMETERS MILLIMETERS 2. DIMENSIONS ARE IN (INCHES) 3. DRAWING NOT TO SCALE 0.25 REF 1.10 (.0433) MAX 0° – 8° 0.65 (.0256) BSC 0.195 – 0.30 (.0077 – .0118) TYP 0.05 – 0.15 (.002 – .006) FE16 (BB) TSSOP 0204 4. RECOMMENDED MINIMUM PCB METAL SIZE FOR EXPOSED PAD ATTACHMENT *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.150mm (.006") PER SIDE 3436fa Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 15 LT3436 U TYPICAL APPLICATIO High Voltage Laser Power Supply 0.01µF 5kV 1800pF 10kV 47k 5W 1800pF 10kV 8 11 L1 1 4 5 HV DIODES 3 2 LASER + 2.2µF Q1 0.47µF 150Ω L2 10µH MUR405 VIN 12V TO 25V Q2 VSW 10k VIN + 10k FB LT3436 2.2µF VC 0.1µF VIN 1N4002 (ALL) 190Ω 1% GND + 10µF LT3436 • TA05 L1 = TBD Q1, Q2 = ZETEX ZTX849 0.47µF = WIMA 3X 0.15µF TYPE MKP-20 HV DIODES = SEMTECH-FM-50 LASER = HUGHES 3121H-P COILTRONICS (407) 241-7876 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1310 1.5A (ISW), 4.5 MHz, High Efficiency Step-Up DC/DC Converter with PLL VIN = 2.75V to 18V, VOUT(MAX) = 35V, IQ = 12mA, ISD = <1µA, MSE Package LT1370/LT1370HV 6A (ISW), 500kHz, High Efficiency Step-Up DC/DC Converter VIN = 2.7V to 30V, VOUT(MAX) = 35V/42V, IQ = 4.5mA, ISD = <12µA, DD, TO220-7 Packages LT1371/LT1371HV 3A (ISW), 500kHz, High Efficiency Step-Up DC/DC Converter VIN = 2.7V to 30V, VOUT(MAX) = 35V/42V, IQ = 4mA, ISD = <12µA, DD,TO220-7,S20 Packages LT1613 550mA (ISW), 1.4MHz, High Efficiency Step-Up DC/DC Converter 90% Efficiency, VIN = 0.9V to 10V, VOUT(MAX) = 34V, IQ = 3mA, ISD = <1µA, ThinSOT Package LT1618 1.5A (ISW), 1.25MHz, High Efficiency Step-Up DC/DC Converter 90% Efficiency, VIN = 1.6V to 18V, VOUT(MAX) = 35V, IQ = 1.8mA, ISD = <1µA, MS Package LT1946/LT1946A 1.5A (ISW), 1.2MHz/2.7MHz, High Efficiency Step-Up DC/DC Converter VIN = 2.45V to 16V, VOUT(MAX) = 34V, IQ = 3.2mA, ISD = <1µA, MS8 Package LT1961 1.5A (ISW), 1.25MHz, High Efficiency Step-Up DC/DC Converter 90% Efficiency, VIN = 3V to 25V, VOUT(MAX) = 35V, IQ = 0.9mA, ISD = 6µA, MS8E Package LTC3400/LTC3400B 600mA (ISW), 1.2MHz, Synchronous Step-Up DC/DC Converter 92% Efficiency, VIN = 0.85V to 5V, VOUT(MAX) = 5V, IQ = 19µA/300µA, ISD = <1µA, ThinSOT Package LTC3401 1A (ISW), 3MHz, Synchronous Step-Up DC/DC Converter 97% Efficiency, VIN = 0.5V to 5V, VOUT(MAX) = 6V, IQ = 38µA, ISD = <1µA, MS Package LTC3402 2A (ISW), 3MHz, Synchronous Step-Up DC/DC Converter 97% Efficiency, VIN = 0.5V to 5V, VOUT(MAX) = 6V, IQ = 38µA, ISD = <1µA, MS Package 3436fa 16 Linear Technology Corporation LT/LWI/LT 0505 REV A • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2003