MAXIM MAX1883

MAX1778/
MAX1880–MAX1885
Quad-Output TFT LCD DC-DC
Converters with Buffer
General Description
The MAX1778/MAX1880–MAX1885 multiple-output
DC-DC converters provide the regulated voltages
required by active matrix thin-film transistor (TFT) liquid
crystal displays (LCD) in a low-profile TSSOP package.
One high-power step-up converter and two low-power
charge pumps convert the 2.7V to 5.5V input voltage
into three independent output voltages. A built-in linear
regulator and VCOM buffer complete the power-supply
requirements.
The main step-up converter accurately generates an
externally set output voltage up to 13V that can supply
the display’s row/column drivers. The converter’s high
switching frequency and current-mode PWM architecture provide fast transient response and allow the use
of small low-profile inductors and ceramic capacitors.
The low-power BiCMOS control circuitry and internal
14V switch (0.35Ω N-channel MOSFET) enable efficiencies up to 91%.
The dual low-power charge pumps (MAX1778/
MAX1880/MAX1881/MAX1882 only) independently regulate one positive output (VPOS) and one negative output (V NEG ). These low-power outputs use external
diode and capacitor stages (as many stages as
required) to regulate output voltages up to +40V and
-40V. A unique control scheme minimizes output ripple
as well as capacitor sizes for both charge pumps.
A resistor-programmable, 40mA, low-dropout linear
regulator (MAX1778/MAX1881/MAX1883/MAX1884
only) provides preregulation or postregulation for any of
the supplies. For higher current applications, an external transistor can be added. Additionally, the VCOM
buffer provides a high current output that is ideal for
driving the capacitive backplane of TFT LCD panels.
The VCOM buffer’s output voltage is preset with an
internal 50% resistive-divider or can be externally
adjusted for other voltages.
The MAX1778/MAX1880–MAX1885 are protected
against output undervoltage and thermal overload conditions by a latched fault detection circuit that shuts
down the device. All devices are available in the ultrathin TSSOP package (1.1mm max height).
Applications
Features
o 500kHz/1MHz Current-Mode PWM Step-Up
Regulator
Up to +13V Main High-Power Output
±1% Accurate
High Efficiency (91%)
o Dual Regulated Charge-Pump Outputs
(MAX1778/MAX1880–MAX1882 only)
Up to +40V Positive Charge-Pump Output
Up to -40V Negative Charge-Pump Output
o Low-Dropout 40mA Linear Regulator
(MAX1778/MAX1881/MAX1883/MAX1884 only)
Up to +15V LDO Input
o Optional Higher Current with External Transistor
o 2.7V to 5.5V Input Supply
o Internal Supply Sequencing and Soft-Start
o Power-Ready Output
o Adjustable Fault-Detection Latch
o Thermal Protection (+160°C)
o 0.1µA Shutdown Current
o 0.7mA IN Quiescent Current
o Ultra-Small External Components
o Thin TSSOP Package (1.1mm max height)
Ordering Information
TEMP RANGE
PIN-PACKAGE
MAX1778EUG
PART
-40°C to +85°C
24 TSSOP
MAX1778EUG+
-40°C to +85°C
24 TSSOP
MAX1880EUG
-40°C to +85°C
24 TSSOP
MAX1880EUG/V+
-40°C to +85°C
24 TSSOP
MAX1881EUG
-40°C to +85°C
24 TSSOP
MAX1882EUG
-40°C to +85°C
24 TSSOP
MAX1883EUP
-40°C to +85°C
20 TSSOP
MAX1884EUP
-40°C to +85°C
20 TSSOP
MAX1885EUP
-40°C to +85°C
20 TSSOP
+Denotes a lead(Pb)-free/RoHS-compliant package.
/V denotes an automotive qualified part.
TFT LCD Notebook Displays
TFT LCD Desktop Monitor Panels
Typical Operating Circuit appears at end of data sheet.
Pin Configurations and Selector Guide appear at end of
data sheet.
For pricing, delivery, and ordering information, please contact Maxim Direct
at 1-888-629-4642, or visit Maxim’s website at www.maximintegrated.com.
19-1979; Rev 2; 10/12
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
ABSOLUTE MAXIMUM RATINGS
IN, SHDN, TGND, FLTSET to GND...........................-0.3V to +6V
DRVN to GND .........................................-0.3V to (VSUPN + 0.3V)
DRVP to GND..........................................-0.3V to (VSUPP + 0.3V)
PGND to GND.....................................................................±0.3V
RDY, SUPB to GND ................................................-0.3V to +14V
LX, SUPP, SUPN to PGND .....................................-0.3V to +14V
SUPL to GND..........................................................-0.3V to +18V
LDOOUT to GND ....................................-0.3V to (VSUPL + 0.3V)
INTG, REF, FB, FBN, FBP to GND ...............-0.3V to (VIN + 0.3V)
FBL to GND .............-0.3V to the lower of (VSUPL + 0.3V) or +6V
BUFOUT, BUF+, BUF- to GND ...............-0.3V to (VSUPB + 0.3V)
Continuous Power Dissipation (TA = +70°C)
20-Pin TSSOP (derate 10.9mW/°C above +70°C) ......879mW
24-Pin TSSOP (derate 12.2mW/°C above +70°C) ......975mW
Operating Temperature Range
MAX1778EUG, MAX1883EUP ........................-40°C to +85°C
Junction Temperature ......................................................+150°C
Storage Temperature Range .............................-65°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(VIN = +3.0V, SHDN = IN, VSUPP = VSUPN = VSUPB = VSUPL = 10V, LDOOUT = FBL, BUF- = BUFOUT, BUF+ = FLTSET = TGND =
PGND = GND, CREF = 0.22µF, CBUF = 1µF, TA = 0°C to +85°C. Typical values are at TA = +25°C, unless otherwise noted.)
PARAMETER
Input Supply Range
Input Undervoltage Threshold
IN Quiescent Supply Current
SUPP Quiescent Current
SUPN Quiescent Current
2
SYMBOL
CONDITIONS
MAX
UNITS
5.5
V
2.4
2.6
V
MAX1778/MAX1880/
MAX1883 (fOSC = 1MHz)
0.7
1
MAX1881/MAX1882/
MAX1884/MAX1885
(fOSC = 500kHz)
0.6
1
MAX1778/MAX1880
(fOSC = 1MHz)
0.4
0.7
MAX1881/MAX1882
(fOSC = 500kHz)
0.3
0.5
MAX1778/MAX1880
(fOSC = 1MHz)
0.4
0.7
MAX1881/MAX1882
(fOSC = 500kHz)
0.3
0.5
VIN
VUVLO
IIN
ISUPP
ISUPN
MIN
TYP
2.7
VIN rising, 40mV hysteresis (typ)
VFB = VFBP
= 1.5V, VFBN
= -0.2V
VFBP = 1.5V
VFBN = -0.2V
2.2
mA
mA
mA
IN Shutdown Current
VSHDN = 0, VIN = 5V
0.1
10
µA
SUPP Shutdown Current
VSHDN = 0, VSUPP = 13V,
MAX1778/MAX1880/MAX1881/MAX1882
0.1
10
µA
SUPN Shutdown Current
VSHDN = 0, VSUPN = 13V,
MAX1778/MAX1880/MAX1881/MAX1882
0.1
10
µA
SUPL Shutdown Current
VSHDN = 0, VSUPL = 13V
MAX1778/MAX1881/MAX1883/MAX1884
0.1
10
µA
SUPB Shutdown Current
VSHDN = 0, VSUPB = 13V
6
13
µA
Maxim Integrated
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
ELECTRICAL CHARACTERISTICS (continued)
(VIN = +3.0V, SHDN = IN, VSUPP = VSUPN = VSUPB = VSUPL = 10V, LDOOUT = FBL, BUF- = BUFOUT, BUF+ = FLTSET = TGND =
PGND = GND, CREF = 0.22µF, CBUF = 1µF, TA = 0°C to +85°C. Typical values are at TA = +25°C, unless otherwise noted.)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
13
V
MAIN STEP-UP CONVERTER
Main Output Voltage Range
VMAIN
FB Regulation Voltage
VFB
FB Input Bias Current
IFB
Operating Frequency
fOSC
VIN
Integrator enabled, CINTG = 1000pF
1.234
Integrator disabled (INTG = REF)
1.220
1.280
-50
+50
nA
VFB = 1.25V, INTG = GND
0.85
1
1.15
MAX1881/MAX1882/MAX1884/MAX1885
425
500
575
kHz
80
85
91
%
ILX = 0 to 200mA,
VMAIN = 10V
Integrator enabled,
CINTG = 1000pF
0.01
Integrator disabled
(INTG = REF)
0.2
%
0.1
Integrator Transconductance
LX Leakage Current
ILX
ILIM
0.35
0.7
Ω
VLX = 13V
0.01
20
µA
0.38
0.5
0.275
Phase II = soft-start (1024/fOSC)
0.75
Phase III = soft-start (1024/fOSC)
1.12
Phase IV = fully on (after 3072/fOSC)
1.15
Maximum RMS LX Current
Soft-Start Period
tSS
FB Fault Trip Level
Power-up to the end of Phase III
fCHP
FBP Regulation Voltage
VFBP
FBP Input Bias Current
IFBP
DRVP PCH On-Resistance
RPCH(ON)
DRVP NCH On-Resistance
RNCH(ON)
FBP Fault Trip Level
Maxim Integrated
1.85
1
A
3072 / fOSC
s
Falling edge, FLTSET = GND
1.07
1.1
1.14
0.955
0.99
1.025
2.7
13
1.2
VFBP = 1.5V
1.3
V
+50
nA
10
Ω
2
4
Ω
kΩ
20
A
0.1
Rising edge
V
5
-50
VFBP = 1.2V
VFBP = 1.3V
1.25
V
Hz
0.5 x fOSC
Maximum RMS DRVP Current
FBP Power-Ready Trip Level
1.5
A
Falling edge, FLTSET = 1V
POSITIVE CHARGE PUMP (MAX1778/MAX1880/MAX1881/MAX1882 only)
VSUPP
SUPP Input Supply Range
Operating Frequency
µS
ILX = 100mA
Phase I = soft-start (1024/fOSC)
LX Current Limit
%/V
317
RLX(ON)
V
MAX1778/MAX1880/MAX1883
Line Regulation
LX Switch On-Resistance
1.260
MHz
Oscillator Maximum Duty
Cycle
Load Regulation
1.247
1.09
1.125
1.16
Falling edge, FLTSET = GND
1.08
1.11
1.16
Falling edge, FLTSET = 1V
0.955
0.99
1.025
V
V
3
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
ELECTRICAL CHARACTERISTICS (continued)
(VIN = +3.0V, SHDN = IN, VSUPP = VSUPN = VSUPB = VSUPL = 10V, LDOOUT = FBL, BUF- = BUFOUT, BUF+ = FLTSET = TGND =
PGND = GND, CREF = 0.22µF, CBUF = 1µF, TA = 0°C to +85°C. Typical values are at TA = +25°C, unless otherwise noted.)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
NEGATIVE CHARGE PUMP (MAX1778/MAX1880/MAX1881/MAX1882 only)
SUPN Input Supply Range
Operating Frequency
fCHP
FBN Regulation Voltage
VFBN
FBN Input Bias Current
IFBN
DRVN PCH On-Resistance
DRVN NCH On-Resistance
2.7
VSUPN
-50
VFBN = 0
0
-50
RPCH(ON)
RNCH(ON)
13
0.5 x fOSC
VFBN = +50mV
V
Hz
+50
mV
+50
nA
5
10
Ω
2
4
Ω
VFBN = -50mV
20
kΩ
FBN Power-Ready Trip Level
Falling edge
80
125
165
mV
FBN Fault Trip Level
Rising edge
80
140
190
mV
15
V
4
4.3
V
µA
0.1
Maximum RMS DRVN Current
A
LOW-DROPOUT LINEAR REGULATOR (MAX1778/MAX1881/MAX1883/MAX1884 only)
SUPL Input Supply Range
VSUPL
SUPL Undervoltage Lockout
SUPL Quiescent Current
Dropout Voltage (Note 1)
FBL Regulation Voltage
4.5
Rising edge, 50mV hysteresis (typ)
ISUPL
VDROP
VFBL
3.8
ILDO = 100µA
LDO is set to
regulate at 9V
120
220
ILDO = 40mA
130
300
ILDO = 5mA
70
VSUPL = 10V, LDO regulating at 9V,
ILDO = 15mA
1.235
1.25
mV
1.265
V
LDO Load Regulation
VSUPL = 10V, LDO regulating at 9V,
ILDO = 100µA to 40mA
1.2
%
LDO Line Regulation
VSUPL = 4.5V to 15V, FBL = LDOOUT,
ILDO = 15mA
0.02
%/V
FBL Input Bias Current
LDO Current Limit
IFBL
ILDOLIM
VFBL = 1.25V
VSUPL = 10V, VLDOOUT = 9V, VFBL = 1.2V
+0.8
µA
130
220
mA
13
V
420
850
µA
+10
µA
-0.8
40
VCOM BUFFER
SUPB Input Supply Range
VSUPB
SUPB Quiescent Current
ISUPB
4.5
VSUPB = 13V
BUFOUT Leakage Current
Power-Supply Rejection Ratio
PSRR
Input Common-Mode Voltage
Range
VCM
Common-Mode Rejection Ratio
Input Bias Current
Input Offset Current
Gain Bandwidth Product
4
-10
CMRR
IBIAS
IOS
GBW
VSUPB = 4.5V to 13V, VCM = 2.25V
85
|VOS| < 10mV
1.2
VCM = 1.2V to 8.8V
8.8
-100
VCM = 5V
-100
V
dB
75
VCM = 5V
CBUF = 1µF
dB
98
-10
+100
+100
13
nA
nA
kHz
Maxim Integrated
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
ELECTRICAL CHARACTERISTICS (continued)
(VIN = +3.0V, SHDN = IN, VSUPP = VSUPN = VSUPB = VSUPL = 10V, LDOOUT = FBL, BUF- = BUFOUT, BUF+ = FLTSET = TGND =
PGND = GND, CREF = 0.22µF, CBUF = 1µF, TA = 0°C to +85°C. Typical values are at TA = +25°C, unless otherwise noted.)
PARAMETER
Output Voltage
Input Offset Voltage
SYMBOL
VBUFOUT
VOS
CONDITIONS
BUF+ = GND
VSUPB = 4.5V to 13V,
VCM = 1.2V to
(VSUPB - 1.2V)
MIN
TYP
MAX
IBUFOUT = 0
4.99
5.01
IBUFOUT = ±5mA
4.97
5.03
IBUFOUT = ±45mA
4.93
5.07
IBUFOUT = ±5mA
-30
+30
IBUFOUT = ±45mA
-70
+70
V
mV
Output Voltage Swing High
VOH
IBUFOUT = -45mA, ∆VOS = 1V
Output Voltage Swing Low
VOL
IBUFOUT = +45mA, ∆VOS = 1V
9
9.6
0.4
Peak Buffer Output Current
V
1
±150
BUF+ Dual Mode™ Threshold
Voltage
UNITS
Falling edge, 20mV hysteresis (typ)
V
mA
80
125
170
mV
1.231
1.25
1.269
V
0.9
1.05
1.2
V
0.9
V
1
µA
0.85 x
VREF
V
170
mV
nA
REFERENCE
Reference Voltage
VREF
-2µA < IREF < 50µA
Reference Undervoltage
Threshold
LOGIC SIGNALS
SHDN Input Low Voltage
SHDN Input High Voltage
SHDN Input Current
2.1
ISHDN
V
0.01
0.67 x
VREF
FLTSET Input Voltage Range
FLTSET Threshold Voltage
Rising edge, 25mV hysteresis (typ)
80
125
FLTSET Input Current
VFLTSET = 1V
0.1
50
RDY Output Low Voltage
ISINK = 2mA
0.25
0.5
V
RDY Output High Leakage
V RDY = 13V
0.01
1
µA
Thermal Shutdown
Rising temperature
160
°C
Dual Mode is a trademark of Maxim Integrated Products, Inc.
Maxim Integrated
5
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
ELECTRICAL CHARACTERISTICS
(VIN = +3.0V, SHDN = IN, VSUPP = VSUPN = VSUPB = VSUPL = 10V, LDOOUT = FBL, BUF- = BUFOUT, BUF+ = FLTSET = TGND =
PGND = GND, CREF = 0.22µF, CBUF = 1µF, TA = -40°C to +85°C, unless otherwise noted.) (Note 2)
PARAMETER
SYMBOL
Input Supply Range
VIN
Input Undervoltage
Threshold
VUVLO
IN Quiescent Supply
Current
SUPP Quiescent Current
SUPN Quiescent Current
IIN
ISUPP
ISUPN
CONDITIONS
VIN Rising, 40mV hysteresis (typ)
VFB =
VFBP = 1.5V,
VFBN = -0.2V
VFBP = 1.5V
VFBN = -0.2V
MIN
MAX
UNITS
2.7
5.5
V
2.2
2.6
V
MAX1778/MAX1880/
MAX1883 (fOSC = 1MHz)
1
MAX1881/MAX1882/MAX1884/
MAX1885 (fOSC = 500kHz)
1
mA
MAX1778/MAX1880
(fOSC = 1MHz)
0.7
MAX1881/MAX1882
(fOSC = 500kHz)
0.5
MAX1778/MAX1880
(fOSC = 1MHz)
0.7
MAX1881/MAX1882
(fOSC = 500kHz)
0.5
mA
mA
IN Shutdown Current
VSHDN = 0, VIN = 5V
10
µA
SUPP Shutdown Current
VSHDN = 0, VSUPP = 13V,
MAX1778/MAX1880/MAX1881/MAX1882
10
µA
SUPN Shutdown Current
VSHDN = 0, VSUPN = 13V,
MAX1778/MAX1880/MAX1881/MAX1882
10
µA
SUPL Shutdown Current
VSHDN = 0, VSUPL = 13V,
MAX1778/MAX1881/MAX1883/MAX1884
10
µA
SUPB Shutdown Current
VSHDN = 0, VSUPB = 13V
13
µA
V
MAIN STEP-UP CONVERTER
Main Output Voltage Range
VMAIN
FB Regulation Voltage
VFB
FB Input Bias Current
IFB
Operating Frequency
FOSC
VIN
13
Integrator enabled, CINTG = 1000pF
1.223
1.269
Integrator disabled (INTG = REF)
1.21
1.29
VFB = 1.25V, INTG = GND
-50
+50
nA
MAX1778/MAX1880/MAX1883
0.75
1.25
MHz
MAX1881/MAX1882/MAX1884/MAX1885
375
625
kHz
79
91
%
0.7
Ω
20
µA
Oscillator Maximum Duty
Cycle
LX Switch On-Resistance
LX Leakage Current
LX Current Limit
FB Fault Trip Level
6
RLX(ON)
ILX
ILIM
V
ILX = 100mA
VLX = 13V
Phase I = soft-start (1024/fOSC)
0.275
0.525
Phase IV = fully on (after 3072/fOSC)
1.1
2.05
Falling edge, FLTSET = GND
1.07
1.14
A
V
Maxim Integrated
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
ELECTRICAL CHARACTERISTICS (continued)
(VIN = +3.0V, SHDN = IN, VSUPP = VSUPN = VSUPB = VSUPL = 10V, LDOOUT = FBL, BUF- = BUFOUT, BUF+ = FLTSET = TGND =
PGND = GND, CREF = 0.22µF, CBUF = 1µF, TA = -40°C to +85°C, unless otherwise noted.) (Note 2)
PARAMETER
SYMBOL
CONDITIONS
MIN
MAX
UNITS
2.7
13
V
POSITIVE CHARGE PUMP (MAX1778/MAX1880/MAX1881/MAX1882 only)
SUPP Input Supply Range
VSUPP
FBP Regulation Voltage
VFBP
FBP Input Bias Current
IFBP
DRVP PCH On-Resistance
RPCH(ON)
DRVP NCH On-Resistance
RNCH(ON)
FBP Power-Ready Trip Level
VFBP = 1.5V
1.2
1.3
V
-50
+50
nA
10
Ω
VFBP = 1.2V
4
VFBP = 1.3V
20
Rising edge
1.09
Ω
kΩ
1.16
V
NEGATIVE CHARGE PUMP (MAX1778/MAX1880/MAX1881/MAX1882 only)
SUPN Input Supply Range
FBN Regulation Voltage
FBN Input Bias Current
DRVN PCH On-Resistance
DRVN NCH On-Resistance
VSUPN
2.7
13
V
VFBN
-50
+50
mV
-50
+50
nA
IFBN
VFBN = 0
10
Ω
4
Ω
165
mV
4.5
15
V
3.8
4.3
V
RPCH(ON)
RNCH(ON)
FBN Power-Ready Trip Level
VFBN = +50mV
VFBN = -50mV
20
Falling edge
80
kΩ
LOW DROPOUT LINEAR REGULATOR (MAX1778/MAX1881/MAX1883/MAX1884 only)
SUPL Input Supply Range
VSUPL
SUPL Undervoltage Lockout
Rising edge, 50mV hysteresis (typ)
SUPL Quiescent Current
ISUPL
ILDO = 100µA
240
µA
Dropout Voltage (Note 1)
VDROP
LDO regulating to 9V, ILDO = 40mA
330
mV
VFBL
VSUPL = 10V, LDO regulating to 9V,
ILDO = 15mA
1.265
V
FBL Regulation Voltage
1.222
LDO Load Regulation
VSUPL = 10V, LDO regulating to 9V,
ILDO = 100µA to 40mA
1.2
%
LDO Line Regulation
VSUPL = 4.5V to 15V, FBL = LDOOUT,
ILDO = 15mA
0.02
%/V
-1.2
+1.2
µA
40
260
mA
4.5
13
V
850
µA
-10
+10
µA
1.2
8.8
V
FBL Input Bias Current
LDO Current Limit
IFBL
ILDOLIM
VFBL = 1.25V
VSUPL = 10V, VLDOOUT = 9V, VFBL = 1.2V
VCOM BUFFER
SUPB Input Supply Range
VSUPB
SUPB Quiescent Current
ISUPB
BUFOUT Leakage Current
Input Common-Mode Voltage
Maxim Integrated
VCM
VSUPB = 13V
|VOS| < 10mV
7
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
ELECTRICAL CHARACTERISTICS (continued)
(VIN = +3.0V, SHDN = IN, VSUPP = VSUPN = VSUPB = VSUPL = 10V, LDOOUT = FBL, BUF- = BUFOUT, BUF+ = FLTSET = TGND =
PGND = GND, CREF = 0.22µF, CBUF = 1µF, TA = -40°C to +85°C, unless otherwise noted.) (Note 2)
PARAMETER
Input Bias Current
Input Offset Current
Output Voltage
Input Offset Voltage
SYMBOL
CONDITIONS
MIN
MAX
UNITS
IBIAS
VCM = 5V
-500
+500
nA
IOS
VCM = 5V
-500
+500
nA
IBUFOUT = 0
4.988
5.012
IBUFOUT = ±5mA
4.97
5.03
IBUFOUT = ±45mA
4.93
5.07
IBUFOUT = ±5mA
-30
+30
IBUFOUT = ±45mA
-70
+70
VBUFOUT
VOS
BUF+ = GND
VSUPB = 4.5V to 13V
VCM = 1.2V to
(VSUPB - 1.2V)
mV
Output Voltage Swing High
VOH
IBUFOUT = -45mA, ∆VOS = 1V
Output Voltage Swing Low
VOL
IBUFOUT = +45mA, ∆VOS = 1V
BUF+ Dual-Mode
Threshold Voltage
V
Falling edge, 20mV hysteresis (typ)
9
V
1
V
80
170
mV
1.223
1.269
V
0.9
1.2
V
REFERENCE
Reference Voltage
VREF
-2µA < IREF < 50µA
Reference Undervoltage
Threshold
LOGIC SIGNALS
SHDN Input Low Voltage
0.9
SHDN Input High Voltage
SHDN Input Current
2.1
I SHDN
1
FLTSET Input Voltage Range
0.74 x VREF
FLTSET Threshold Voltage
Rising edge, 25mV hysteresis (typ)
FLTSET Input Current
V
V
0.85 x VREF
80
µA
V
170
mV
VFLTSET = 1V
50
nA
RDY Output Low Voltage
ISINK = 2mA
0.5
V
RDY Output High Leakage
V RDY = 13V
1
µA
Note 1: Dropout voltage is defined as the VSUPL - VLDOOUT, when VSUPL is 100mV below the set value of VLDOOUT.
Note 2: Specifications to -40°C are guaranteed by design, not production tested.
8
Maxim Integrated
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC-DC
Converters with Buffer
Typical Operating Characteristics
(Circuit of Figure 1, V IN = +3.3V, SHDN = IN, V MAIN = V SUPP = V SUPN = V SUPB = V SUPL = 8V, BUF- = BUFOUT,
BUF+ = FLTSET = TGND = PGND = GND, TA = +25°C.)
MAIN 8V OUTPUT EFFICIENCY
vs. LOAD CURRENT
VIN = 5V
8.00
90
EFFICIENCY (%)
7.96
CINTG = 470pF
RCOMP = 24kΩ
CCOMP = 470pF
7.92
70
200
400
600
VOUT = 8V
RCOMP = 24kΩ
CCOMP = 470pF
CINTG = 470pF
VIN = 5V
11.84
200
FIGURE 8
CINTG = 470pF
11.76
400
600
0
800
100
200
300
400
500
600
IOUT (mA)
MAIN 12V OUTPUT EFFICIENCY
vs. LOAD CURRENT
STEP UP CONVERTERS
SWITCHING FREQUENCY vs. INPUT VOLTAGE
POSITIVE CHARGE-PUMP OUTPUT VOLTAGE
vs. LOAD CURRENT
70
60
FIGURE 8
VOUT = 12V
CINTG = 470pF
50
MAX1778 toc05
20.2
1.05
1.00
300
400
500
VSUPP = 8V
19.6
0.95
0.90
VSUPP = 7.5V
VSUPP = 7V
19.4
19.2
0.80
200
19.8
0.85
40
100
VSUPP = 10V
20.0
1.10
VPOS (V)
VIN = 3.3V
MAX1778
1.15
SWITCHING FREQUENCY (MHz)
MAX1778 toc04
90
1.20
MAX1778 toc06
IOUT (mA)
VIN = 5V
0
12.00
IOUT (mA)
100
80
0
800
VIN = 3.3V
11.92
60
40
0
2.5
600
3.0
3.5
4.0
4.5
5.0
0
5.5
5
10
15
20
IOUT (mA)
VIN (V)
IPOS (mA)
POSITIVE CHARGE-PUMP EFFICIENCY
vs. LOAD CURRENT
MAXIMUM POSITIVE CHARGE-PUMP
OUTPUT VOLTAGE vs. SUPPLY VOLTAGE
NEGATIVE CHARGE-PUMP OUTPUT VOLTAGE
vs. LOAD CURRENT
VSUPP = 7.5V
80
35
VPOS (V)
70
VSUPP = 10V
60
50
40
VPOS = 20V
30
5
10
INEG (mA)
15
20
VSUPN = 6V
VSUPN = 7V
IPOS = 1mA
25
IPOS = 10mA
20
-4.96
-4.98
15
-5.00
10
-5.02
VSUPN = 8V
-5.04
5
0
-4.92
-4.94
30
VSUPP = 8V
-4.90
MAX1778 toc09
VSUPP = 7V
90
40
MAX1778 toc07
100
VNEG (V)
EFFICIENCY (%)
12.08
VIN = 3.3V
50
7.88
EFFICIENCY (%)
80
12.16
MAX1778 toc08
VOUT (V)
8.04
VIN = 5V
VOUT (V)
VIN = 3.3V
12.24
MAX1778 toc02
8.08
100
MAX1778 toc01
8.12
MAIN 12V OUTPUT VOLTAGE
vs. LOAD CURRENT
MAX1778 toc03
MAIN 8V OUTPUT VOLTAGE
vs. LOAD CURRENT
2
4
6
8
VSUPP (V)
10
12
14
0
10
20
INEG (mA)
30
40
9
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
Typical Operating Characteristics (continued)
(Circuit of Figure 1, V IN = +3.3V, SHDN = IN, V MAIN = V SUPP = V SUPN = V SUPB = V SUPL = 8V, BUF- = BUFOUT,
BUF+ = FLTSET = TGND = PGND = GND, TA = +25°C.)
-4
VSUPN = 6V
1.26
-6
60
VNEG (V)
VSUPN = 7V
70
INEG = 10mA
-8
VREF (V)
80
INEG = 1mA
VSUPN = 8V
1.25
-10
50
1.24
-12
40
30
1.23
-14
0
10
20
INEG (mA)
30
2
40
4
6
8
10
12
0
14
20
40
60
80
100
VSUPN (V)
IREF (µA)
STEP-UP CONVERTER LOAD-TRANSIENT
RESPONSE
STEP-UP CONVERTER LOAD-TRANSIENT
RESPONSE WITHOUT INTEGRATOR
STEP-UP CONVERTER LOAD-TRANSIENT
RESPONSE (1µs PULSES)
MAX1778 toc13
MAX1778 toc14
MAX1778 toc15
200mA
0.5A
200mA
A
A
A
0
0
0
8.1V
8.1V
8.0V
B
8.0V
7.9V
B
8.0V
0.5A
1A
C
0
C
C
0
0
40µs/div
A. IMAIN = 20mA to 200mA, 200mA/div
B. VMAIN = 8V, 100mV/div
C. INDUCTOR CURRENT, 1A/div
CINTG = 1000pF
B
7.9V
1A
7.9V
1A
10
MAX1778 toc12
90
1.27
MAX1778 toc11
VNEG = -5V
EFFICIENCY (%)
-2
MAX1778 toc10
100
REFERENCE VOLTAGE
vs. REFERENCE LOAD CURRENT
MAXIMUM NEGATIVE CHARGE-PUMP
OUTPUT VOLTAGE vs. SUPPLY VOLTAGE
NEGATIVE CHARGE-PUMP EFFICIENCY
vs. LOAD CURRENT
40µs/div
A. IMAIN = 20mA to 200mA, 200mA/div
B. VMAIN = 8V, 100mV/div
C. INDUCTOR CURRENT, 1A/div
INTG = REF
4µs/div
A. IMAIN = 0 to 500mA, 500mA/div
B. VMAIN = 8V, 100mV/div
C. INDUCTOR CURRENT, 500mA/div
Maxim Integrated
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
Typical Operating Characteristics (continued)
(Circuit of Figure 1, V IN = +3.3V, SHDN = IN, V MAIN = V SUPP = V SUPN = V SUPB = V SUPL = 8V, BUF- = BUFOUT,
BUF+ = FLTSET = TGND = PGND = GND, TA = +25°C.)
STEP-UP CONVERTER
SOFT-START (LIGHT LOAD)
RIPPLE VOLTAGE WAVEFORMS
STEP-UP CONVERTER
SOFT-START (HEAVY LOAD)
MAX1778 toc17
MAX1778 toc16
MAX1778 toc18
2V
A
A
8V
20V
A
0
0
8V
8V
B
6V
B
-5V
2V
B
6V
4V
4V
0.5A
1.0A
C
C
0
C
0.5A
0
1ms/div
A. VSHDN = O TO 2V, 2V/div
B. VMAIN = 8V, 2V/div
C. INDUCTOR CURRENT, 500mA/div
RLOAD = 400Ω
1µs/div
A. VMAIN = 8V, IMAIN = 200mA, 10mV/div
B. VNEG = -5V, INEG = 10mA, 20mV/div
C. VPOS = 20V, IPOS = 5mA, 20mV/div
1ms/div
A. VSHDN = O TO 2V, 2V/div
B. VMAIN = 8V, 2V/div
C. INDUCTOR CURRENT, 500mA/div
RLOAD = 20Ω
POWER-UP SEQUENCE
(CIRCUIT OF FIGURE 10)
POWER-UP SEQUENCE
MAX1778 toc19
4V
2V
2V
POWER-UP INTO SHORT-CIRCUIT
(CIRCUIT OF FIGURE 10)
MAX1778 toc20
MAX1778 toc21
A
A
0
0
0
20V
B
0
10V
C
5V
10V
C
0
D
0
B
0
D
-5V
0
A
2V
20V
5V
B
4V
10V
C
5V
E
-10V
0
2ms/div
A. VSHDN = O TO 2V, 2V/div
B. RDY, 5V/div
C. POSITIVE CHARGE PUMP = VPOS = 20V, RLOAD = 4kΩ, 10V/div
D. STEP-UP CONVERTER: VMAIN = 8V, RLOAD = 40Ω, 10V/div
E. NEGATIVE CHARGE PUMP: VNEG = -5V, RLOAD = 500Ω, 10V/div
Maxim Integrated
1ms/div
A. RDY, 2V/div
B. POSITIVE CHARGE PUMP, VPOS(SYS) = 20V, 10V/div
C. STEP-UP CONVERTER: VMAIN(SYS) = 8V, 10V/div
D. NEGATIVE CHARGE PUMP, VNEG = -5V, -5V/div
100µs/div
A. RDY, 2V/div
B. GATE OF N-CH MOSFET, 5V/div
C. STEP-UP CONVERTER, VMAIN(START) = 8V, 5V/div
VMAIN(SYS) = GND
11
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
Typical Operating Characteristics (continued)
(Circuit of Figure 1, V IN = +3.3V, SHDN = IN, V MAIN = V SUPP = V SUPN = V SUPB = V SUPL = 8V, BUF- = BUFOUT,
BUF+ = FLTSET = TGND = PGND = GND, TA = +25°C.)
LDO OUTPUT VOLTAGE
vs. LDO OUTPUT CURRENT
(INTERNAL LINEAR REGULATOR)
5.02
VLDOOUT (V)
4.95
ILDOOUT = 40mA
4.90
4.85
5.08
5.06
5.00
5.04
4.98
5.02
VLDO (V)
5.00
5.10
MAX1778 toc23
ILDOOUT = 0
4.96
ILDOOUT = 0
5.00
ILDOOUT = 40mA
4.98
4.96
4.94
4.94
4.80
4.92
4.75
4.90
4
6
8
10
12
4.92
4.90
0.01
VSUPL (V)
0.1
1
10
-15
VLDOOUT = 5V
4.0
VLDOOUT = 5V
3.5
ISUPL - ILDOOUT (mA)
160
35
60
85
LDO SUPPLY CURRENT
vs. LDO OUTPUT CURRENT
(INTERNAL LINEAR REGULATOR)
MAX1778 toc25
200
10
TEMPERATURE (°C)
ILDOOUT (mA)
DROPOUT VOLTAGE
vs. LDO LOAD CURRENT
(INTERNAL LINEAR REGULATOR)
VSUPL - VLDOOUT (mV)
-40
100
120
80
MAX1778 toc26
VLDOOUT (V)
5.04
MAX1778 toc22
5.05
LDO OUTPUT VOLTAGE vs. TEMPERATURE
(INTERNAL LINEAR REGULATOR)
MAX1778 toc24
LDO OUTPUT VOLTAGE
vs. LDO INPUT VOLTAGE
(INTERNAL LINEAR REGULATOR)
3.0
2.5
2.0
1.5
1.0
40
0.5
0
0
0
10
20
ILDOOUT (mA)
12
30
40
0
10
20
30
40
ILDOOUT (mA)
Maxim Integrated
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
Typical Operating Characteristics (continued)
(Circuit of Figure 1, V IN = +3.3V, SHDN = IN, V MAIN = V SUPP = V SUPN = V SUPB = V SUPL = 8V, BUF- = BUFOUT,
BUF+ = FLTSET = TGND = PGND = GND, TA = +25°C.)
MAX1778 toc28
CLDOOUT = 1µF
4OmA
10
CLDOOUT ESR (Ω)
PSRR (dB)
80
MAX1778 toc29
100
MAX1778 toc27
100
LOAD-TRANSIENT RESPONSE
(INTERNAL LINEAR REGULATOR)
REGION OF STABLE CLDOOUT ESR
vs. LOAD CURRENT
POWER-SUPPLY REJECTION RATIO
vs. FREQUENCY
60
40
A
5.00V
B
1
STABLE REGION
0.1
20
0
CLDOOUT = 4.7µF
ILDOOUT = 40mA
4.96V
0.01
0
1
10
100
0
1000
FREQUENCY (kHz)
LOAD-TRANSIENT RESPONSE NEAR
DROPOUT (INTERNAL LINEAR REGULATOR)
10
20
ILDOOUT (mA)
100µs/div
40
30
A. ILDO = 100µA TO 40mA, 40mA/div
B. VLDO = 5V, 20mV/div
VSUPL = VLDO + 500mV
INTERNAL LINEAR-REGULATOR
STARTUP
INTERNAL LINEAR-REGULATOR
RIPPLE REJECTION
MAX1778 toc32
MAX1778 toc31
MAX1778 toc30
4OmA
5.0V
A
0
A
0
A
2V
B
8.0V
5.00V
B
4V
2V
B
C
0
1.0A
0.5A
C
0
4.94V
4V
2V
100µs/div
A. ILDO = 100µA TO 40mA, 40mA/div
B. VLDO = 5V, 20mV/div
VIN = VLDO + 100mV
Maxim Integrated
10µs/div
A. VLDOOUT = 5V, ILDOOUT = 40mA, 10mV/div
B. VMAIN = VSUPL = 8V, 200mV/div
C. IMAIN = 0 TO 750mA, 500mA/div
400µs/div
A. VSHDN = 0 TO 2V, 2V/div
B. VLDOOUT = 5V, RLDOOUT = 125Ω, 2V/div
C. VMAIN = 8V, RMAIN = 40Ω, 2V/div
13
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
Typical Operating Characteristics (continued)
(Circuit of Figure 1, V IN = +3.3V, SHDN = IN, V MAIN = V SUPP = V SUPN = V SUPB = V SUPL = 8V, BUF- = BUFOUT,
BUF+ = FLTSET = TGND = PGND = GND, TA = +25°C.)
2.53
ILDO = 0
2.51
ILDO = 750mA
2.49
FIGURE 7
2.53
VLDO (V)
VLDO (V)
2.55
MAX1778 toc33
2.55
LINEAR-REGULATOR OUTPUT VOLTAGE
vs. LOAD CURRENT
(EXTERNAL LINEAR REGULATOR)
EXTERNAL LINEAR-REGULATOR
LOAD-TRANSIENT RESPONSE
MAX1778 toc35
MAX1778 toc34
LINEAR-REGULATOR OUTPUT VOLTAGE
vs. INPUT VOLTAGE
(EXTERNAL LINEAR REGULATOR)
50mA
A
2.51
2.55V
2.49
2.50V
B
2.45V
2.47
2.47
250mA
FIGURE 7
2.45
2.5
3.0
3.5
4.0
4.5
5.0
0.1
5.5
1
10
VIN (V)
100
EXTERNAL LINEAR-REGULATOR
RIPPLE REJECTION
A. ILDO = 50mA TO 250mA, 200mA/div
B. VLDO = 2.5V, 50mV/div
FIGURE 7
1.5
1A
0.5A
C
VSUPB = 13V
0.5
VCM = VSUPB / 2
0.6
∆VOS (mV)
7.8V
VSUPB = 4.5V
∆VOS (mV)
B
1.0
MAX1778 toc37
2.5
A
8.0V
INPUT OFFSET VOLTAGE DEVIATION
vs. BUFFER SUPPLY VOLTAGE
INPUT OFFSET VOLTAGE DEVIATION
vs. COMMON-MODE VOLTAGE
MAX1778 toc36
2.5V
100µs/div
1000
ILDO (mA)
-0.5
MAX1778 toc38
2.45
0.2
-0.2
-0.6
-1.5
0
-2.5
10µs/div
A. VLDO = 2.5V, ILDO = 200mA, 10mV/div
B. VMAIN = VSUPL = 8V, 200mV/div
C. IMAIN = 0 TO 750mA, 500mA/div
FIGURE 7
14
-1.0
0
2
4
6
8
VCM (V)
10
12
14
4
6
8
10
12
14
VSUPB (V)
Maxim Integrated
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
Typical Operating Characteristics (continued)
(Circuit of Figure 1, V IN = +3.3V, SHDN = IN, V MAIN = V SUPP = V SUPN = V SUPB = V SUPL = 8V, BUF- = BUFOUT,
BUF+ = FLTSET = TGND = PGND = GND, TA = +25°C.)
VSUPB = 13V
VCM = VSUPB/2
0.6
-0.2
-0.6
0
0.2
4
-0.6
2
10
35
60
85
0
-40
-15
TEMPERATURE (°C)
10
35
60
85
0
2
4
TEMPERATURE (°C)
BUFFER INPUT BIAS CURRENT
vs. TEMPERATURE
BUFFER INPUT BIAS CURRENT
vs. BUFFER SUPPLY VOLTAGE
12
MAX1778 toc42
10
VCM = VSUPB/2
11
10
12
14
0.50
VSUPB = 13V
0.46
ISUPB (mA)
IBIAS (nA)
IBIAS (nA)
9
8
7
6
8
BUFFER SUPPLY CURRENT
vs. COMMON-MODE VOLTAGE
10
8
6
VCM (V)
MAX1778 toc43
-15
VSUPB = 4.5V
-0.2
0
-40
6
MAX1778 toc44
0.2
VSUPB = 13V
8
IBIAS (nA)
∆VOS (mV)
∆VOS (mV)
0.6
10
MAX1778 toc39
VSUPB = 13V
VCM = VSUPB/2
BUFFER INPUT BIAS CURRENT
vs. COMMON-MODE VOLTAGE
1.0
MAX1778 toc39
1.0
INPUT OFFSET VOLTAGE DEVIATION
vs. TEMPERATURE
MAX1778 toc41
INPUT OFFSET VOLTAGE DEVIATION
vs. TEMPERATURE
0.42
VSUPB = 4.5V
0.38
6
0.34
5
VCM = VSUPB / 2
4
4
6
8
10
12
0.30
-40
14
-15
10
35
60
85
2
4
6
8
10
12
TEMPERATURE (°C)
VCM (V)
BUFFER SUPPLY CURRENT
vs. BUFFER SUPPLY VOLTAGE
NO-LOAD BUFFER SUPPLY CURRENT
vs. TEMPERATURE
VCOM BUFFER
SMALL-SIGNAL RESPONSE
0.46
VSUPB = 13V
VCM = VSUPB/2
0.9
0.8
ISUPB (mA)
0.38
4.05V
4.00V
0.7
0.42
A
3.95V
0.6
0.5
0.4
4.05V
0.3
0.34
B
4.00V
0.2
3.95V
0.1
VCM = VSUPB/2
14
MAX1778 toc47
1.0
MAX1778 toc45
0.50
ISUPB (mA)
0
VSUPB (V)
MAX1778 toc46
4
0
0.30
4
6
8
10
VSUPB (V)
Maxim Integrated
12
14
-40
-15
10
35
TEMPERATURE (°C)
60
85
4µs/div
A. VBUF+ = 3.95V TO 4.05V, 50mV/div
B. BUFOUT = BUF-, 50mV/div
CBUF = 1µF, VSUPB = 8V
15
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
Typical Operating Characteristics (continued)
(Circuit of Figure 1, V IN = +3.3V, SHDN = IN, V MAIN = V SUPP = V SUPN = V SUPB = V SUPL = 8V, BUF- = BUFOUT,
BUF+ = FLTSET = TGND = PGND = GND, TA = +25°C.)
VCOM BUFFER
LARGE-SIGNAL RESPONSE
VCOM BUFFER
LOAD-TRANSIENT RESPONSE
VCOM BUFFER
LOAD-TRANSIENT RESPONSE
MAX1778 toc48
MAX1778 toc50
MAX1778 toc49
4.50V
500mA
200mA
0
A
4.00V
3.50V
4.50V
A
-200mA
-500mA
4.2V
4.5V
4.0V
B
4.00V
B
4.0V
B
3.5V
3.8V
8.0V
3.50V
A
0
C
10µs/div
8.0V
C
4µs/div
4µs/div
A. VBUF+ = 3.50V TO 4.50V, 0.5V/div
B. BUFOUT = BUF-, 0.5V/div
CBUF = 1µF, VSUPB = 8V
A. IBUFOUT = 400mA PULSES, 500mA/div
B. BUFOUT = BUF-, 0.5V/div
C. VMAIN = 8V, 100mV/div
VSUPB = VMAIN, BUF+ = GND, CBUF = 1µF
A. IBUFOUT = 200mA PULSES, 200mA/div
B. BUFOUT = BUF-, 200mV/div
C. VMAIN = 8V, 50mV/div
VSUPB = VMAIN, BUF+ = GND, CBUF = 1µF
VCOM BUFFER STARTUP
VCOM BUFFER STARTUP
MAX1778 toc51
4V
2V
MAX1778 toc51
4V
A
0
2V
A
0
4V
4V
B
2V
0
B
2V
0
8.1V
C
7.8V
8.1V
C
7.8V
100µs/div
A. RDY, 2V/div
B. BUFOUT = BUF-, CBUF = 1µF, 2V/div
C. VSUPB = VMAIN = 8V, IMAIN = 20mA, 200mV/div
BUF+ = GND
16
100µs/div
A. RDY, 2V/div
B. BUFOUT = BUF-, CBUF = 1µF, 2V/div
C. VSUPB = VMAIN = 8V, IMAIN = 20mA, 200mV/div
BUF+ = GND
Maxim Integrated
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC-DC
Converters with Buffer
Pin Description
PIN
MAX1778
MAX1881
MAX1880
MAX1882
MAX1883
MAX1884
MAX1885
NAME
1
1
1
1
FB
2
2
2
2
INTG
3
3
3
3
FUNCTION
Main Step-Up Regulator Feedback Input. Regulates to 1.25V
nominal. Connect a resistive divider from the output (VMAIN) to FB
to analog ground (GND).
Main Step-Up Integrator Output. When using the integrator,
connect 1000pF to analog ground (GND). To disable the
integrator, connect INTG to REF.
IN
Main Supply Voltage. The supply voltage powers the control
circuitry for all the regulators and can range from 2.7V to 5.5V.
Bypass with a 0.1µF capacitor between IN and GND, as close to
the pins as possible.
4
4
4
4
BUF+
VCOM Buffer (Operational Transconductance Amplifier) Positive
Feedback Input. Connect to GND to select the internal resistive
divider that sets the positive input to half the amplifier’s supply
voltage (VBUF+ = V SUPB /2).
5
5
5
5
BUF-
VCOM Buffer (Operational Transconductance Amplifier) Negative
Feedback Input
6
6
6
6
SUPB
VCOM Buffer (Operational Transconductance Amplifier) Supply
Voltage
7
7
7
7
BUFOUT
VCOM Buffer (Operational Transconductance Amplifier) Output
8
8
8
8
GND
Analog Ground. Connect to power ground (PGND) underneath the
IC.
9
9
9
9
REF
Internal Reference Bypass Terminal. Connect a 0.22µF ceramic
capacitor from REF to analog ground (GND). External load
capability up to 50µA.
10
10
—
—
FBP
Positive Charge-Pump Regulator Feedback Input. Regulates to
1.25V nominal. Connect a resistive divider from the positive
charge-pump output (VPOS) to FBP to analog ground (GND).
11
11
—
—
FBN
Negative Charge-Pump Regulator Feedback Input. Regulates to
0V nominal. Connect a resistive divider from the negative chargepump output (VNEG) to FBN to the reference (REF).
12
12
10
10
SHDN
Active-Low Shutdown Control Input. Pull SHDN low to force the
controller into shutdown. If unused, connect SHDN to IN for normal
operation. A rising edge on SHDN clears the fault latch.
SUPL
Low-Dropout Linear Regulator Input Voltage. Can range from 4.5V
to 15V. Bypass with a 1µF capacitor to GND (see Capacitor
Selection and Regulator Stability). Connect both input pins
together externally.
13
Maxim Integrated
—
11
—
17
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
Pin Description (continued)
PIN
MAX1778
MAX1881
MAX1880
MAX1882
MAX1883
MAX1884
MAX1885
NAME
FUNCTION
Linear Regulator Output. Sources up to 40mA. Bypass to GND with
a ceramic capacitor determined by:
18
14
—
12
—
LDOOUT
15
—
13
—
FBL
Voltage Setting Input. Connect a resistive divider from the linear
regulator output (VLDOOUT) to FBL to analog ground (GND).
⎛ ILDOOUT(MAX) ⎞
CLDOOUT ≥ 0.5ms X ⎜
⎟
⎝ VLDOOUT ⎠
16
16
14
14
FLTSET
Fault Trip-Level Set Input. Connect to a resistive divider between
REF and GND to set the main step-up converter’s and positive
charge pump’s fault thresholds between 0.67 x VREF and 0.85 x
VREF. Connect to GND for the preset fault threshold (0.9 x VREF).
17
17
—
—
SUPN
Negative Charge-Pump Driver Supply Voltage. Bypass to power
ground (PGND) with a 0.1µF capacitor.
18
18
—
—
DRVN
Negative Charge-Pump Driver Output. Output high level is VSUPN
and low level is PGND.
19
19
—
—
SUPP
Positive Charge-Pump Driver Supply Voltage. Bypass to power
ground (PGND) with a 0.1µF capacitor.
20
20
—
—
DRVP
Positive Charge-Pump Driver Output. Output high level is VSUPP
and low level is PGND
21
21
17
17
PGND
Power Ground. Connect to analog ground (GND) underneath the
IC.
22
22
18
18
LX
Main Step-Up Regulator Power MOSFET N-Channel Drain. Place
output diode and output capacitor as close as possible to PGND.
23
23
19
19
TGND
24
24
20
20
RDY
Active-Low, Open-Drain Output. Indicates all outputs are ready.
On-resistance is 125Ω (typ).
—
13–15
15, 16
11–13,
15, 16
N.C.
No Connection. Not internally connected.
Must be connected to ground.
Maxim Integrated
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
L1
6.8µH
INPUT
VIN = 3.3V
CIN
4.7µF
RRDY
100kΩ
C1
0.22µF
MAIN
(8V)
R8
49.9kΩ
LX
SHDN
FB
RDY
SUPL
LDOOUT
TO LOGIC
LDO
VLDOOUT = 5V CLDO
4.7µF
IN
R7
150kΩ
SUPB
SUPN
SUPP
MAX1778
MAIN
VMAIN = 8V
COUT
(2) 4.7µF
R2
274kΩ
R2
49.9kΩ
C4
0.1µF
C5
1.0µF
DRVP
C2
0.1µF
FBL
C4
0.1µF
C7
1.0µF
DRVN
POSITIVE
VPOS = 20V
R3
750kΩ
NEGATIVE
VNEG = -5V
FBN
C3
1.0µF
FBP
R5
200kΩ
R6
49.9kΩ
CREF
0.22µF
R4
49.9kΩ
REF
INTG
FLTSET
PGND
BUFOUT
BUFBUF+
GND
TGND
CBUF
1.0µF
BUFFER OUTPUT
VBUFOUT = VSUPB/2
Figure 1. Typical Application Circuit
Detailed Description
The MAX1778/MAX1880–MAX1885 are highly efficient
multiple-output power supplies for thin-film transistor
(TFT) liquid crystal display (LCD) applications. The
devices contain one high-power step-up converter, two
low-power charge pumps, an operational transconductance amplifier (VCOM buffer), and a low-dropout linear
regulator. The primary step-up converter uses an internal
N-channel MOSFET to provide maximum efficiency and
to minimize the number of external components. The output voltage of the main step-up converter (VMAIN) can
be set from VIN to 13V with external resistors.
The dual charge pumps (MAX1778/MAX1880–MAX1882
only) independently regulate a positive output (VPOS)
and a negative output (VNEG). These low-power outputs
use external diode and capacitor stages (as many
stages as required) to regulate output voltages from 40V to +40V. A unique control scheme minimizes output
ripple as well as capacitor sizes for both charge pumps.
Maxim Integrated
A resistor-programmable 40mA linear regulator
(MAX1778/MAX1881/MAX1883/MAX1884 only) can
provide preregulation or postregulation for any of the
supplies. For higher current applications, an external
transistor can be added.
Additionally, the VCOM buffer provides a high current
output that is ideal for driving capacitive loads, such as
the backplane of a TFT LCD panel. The positive feedback input features dual-mode operation, allowing this
input to be connected to an internal 50% resistivedivider between the buffer’s supply voltage and
ground, or externally adjusted for other voltages.
Also included in the MAX1778/MAX1880–MAX1885 is a
precision 1.25V reference that sources up to 50µA,
logic shutdown, soft-start, power-up sequencing,
adjustable fault detection, thermal shutdown, and an
active-low, open-drain ready output.
19
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
Main Step-up Controller
Changes in the feedback voltage-error signal shift the
switch-current trip level, consequently modulating the
MOSFET duty cycle.
Under very light loads, an inherent switchover to pulseskipping takes place (Figure 3). When this occurs, the
controller skips most of the oscillator pulses in order to
reduce the switching frequency and gate charge losses. When pulse-skipping, the step-up controller initiates
a new switching cycle only when the output voltage
drops too low. The n-channel MOSFET turns on, allowing the inductor current to ramp up until the multi-input
comparator trips. Then, the MOSFET turns off and the
diode turns on, forcing the inductor current to ramp
down. When the inductor current reaches zero, the
diode turns off, so the inductor stops conducting
current. This forces the threshold between pulse-skipping and PWM operation to coincide with the boundary
between continuous and discontinuous inductorcurrent operation:
During normal pulse-width modulation (PWM) operation, the MAX1778/MAX1880–MAX1885 main step-up
controllers switch at a constant frequency of 500kHz or
1MHz (see the Selector Guide), allowing the use of lowprofile inductors and output capacitors. Depending on
the input-to-output voltage ratio, the controller regulates
the output voltage and controls the power transfer by
modulating the duty cycle (D) of each switching cycle:
D ≈
VMAIN - VIN
VMAIN
On the rising edge of the internal clock, the controller
sets a flip-flop when the output voltage is too low, which
turns on the n-channel MOSFET (Figure 2). The inductor current ramps up linearly, storing energy in a magnetic field. Once the sum of the feedback voltage error
amplifier, slope-compensation, and current-feedback
signals trip the multi-input comparator, the MOSFET
turns off, the flip-flop resets, and the diode (D1) turns
on. This forces the current through the inductor to ramp
back down, transferring the energy stored in the magnetic field to the output capacitor and load. The
MOSFET remains off for the rest of the clock cycle.
1 ⎛ VIN ⎞
ILOAD(CROSSOVER) ≈
2 ⎜⎝ VMAIN ⎟⎠
L1
⎛ VMAIN - VIN ⎞
⎜
⎟
fOSCL
⎝
⎠
VIN
(2.7V TO 5.5V)
OSC
(80% DUTY)
MAX1778
MAX1880
MAX1881
MAX1882
MAX1883
MAX1884
MAX1885
2
CIN
LX
D1
VMAIN
(UP TO 13V)
S
COUT
R
Q
ILIM
PWM
COMPARATOR
R1
PGND
ILIM
COMPARATOR
RCOMP
(OPTIONAL)
FB
CCOMP
(OPTIONAL)
gm
ERROR
AMPLIFIER
REF
VREF
1.25V
INTG
R2
CREF
GND
( )V
VMAIN = 1 + R1
R2
VREF = 1.25V
REF
CINTG
Figure 2. Main Step-Up Converter Block Diagram
20
Maxim Integrated
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
Dual Charge-Pump Regulator (MAX1778/
MAX1880–MAX1882 Only)
The switching waveforms appear noisy and asynchronous when light loading causes pulse-skipping operation; this is a normal operating condition that improves
light-load efficiency.
The MAX1778/MAX1880–MAX1882 controllers contain
two independent low-power charge pumps (Figure 4).
One charge pump inverts the input voltage and provides a regulated negative output voltage. The second
charge pump doubles the input voltage and provides a
regulated positive output voltage. The controllers
contain internal p-channel and n-channel MOSFETs to
control the power transfer. The internal MOSFETs
switch at a constant frequency (fCHP = fOSC/2).
INDUCTOR CURRENT
IPEAK
Positive Charge Pump
During the first half-cycle, the n-channel MOSFET turns
on and charges flying capacitor CX(POS) (Figure 4).
This initial charge is controlled by the variable
n-channel on-resistance. During the second half-cycle,
the n-channel MOSFET turns off and the p-channel
MOSFET turns on, level shifting CX(POS) by VSUPP volts.
This connects C X(POS) in parallel with the reservoir
capacitor COUT(POS). If the voltage across COUT(POS)
plus a diode drop (VPOS + VDIODE) is smaller than the
level-shifted flying capacitor voltage (V CX(POS) +
VSUPP), charge flows from CX(POS) to COUT(POS) until
the diode (D3) turns off.
ILOAD
TIME
tON
tOFF
Figure 3. Discontinuous-to-Continuous Conduction Crossover
Point
MAX1778
MAX1880
MAX1881
MAX1882
SUPP
VSUPP
2.7V TO 13V
SUPN
VSUPN
2.7V TO 13V
OSC
D2
VSUPD
CX(POS)
DRVP
DRVN
D3
CX(NEG)
D4
D5
R3
FBP
VPOS
R5
FBN
VNEG
COUT(POS)
COUT(NEG)
R4
R6
VREF
1.25V
REF
( )V
VPOS = 1 + R3
R4
VREF = 1.25V
REF
GND
PGND
CREF
0.22µF
( )
VNEG = - R5 VREF
R6
VREF = 1.25V
Figure 4. Low-Power Charge Pump Block Diagram
Maxim Integrated
21
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
Negative Charge Pump
During the first half-cycle, the p-channel MOSFET turns
on, and flying capacitor C X(NEG) charges to VSUPN
minus a diode drop (Figure 4). During the second
half-cycle, the p-channel MOSFET turns off, and the
n-channel MOSFET turns on, level shifting CX(NEG).
This connects CX(NEG) in parallel with reservoir capacitor COUT(NEG). If the voltage across COUT(NEG) minus
a diode drop is greater than the voltage across
CX(NEG), charge flows from COUT(NEG) to CX(NEG) until
the diode (D5) turns off. The amount of charge transferred to the output is controlled by the variable
n-channel on-resistance.
Low-Dropout Linear Regulator (MAX1778/
MAX1881/MAX1883/MAX1884 Only)
The MAX1778/MAX1881/MAX1883/MAX1884 contain a
low-dropout linear regulator (Figure 5) that uses an
internal PNP pass transistor (QP) to supply loads up to
40mA. As illustrated in Figure 5, the 1.25V reference is
connected to the error amplifier, which compares this
reference with the feedback voltage and amplifies the
difference. If the feedback voltage is higher than the
reference voltage, the controller lowers the base
current of QP, which reduces the amount of current to
the output. If the feedback voltage is too low, the
device increases the pass transistor base current,
which allows more current to pass to the output and
increases the output voltage. However, the linear regulator also includes an output current limit to protect the
internal pass transistor against short circuits.
The low-dropout linear regulator monitors and controls
the pass transistor’s base current, limiting the output
current to 130mA (typ). In conjunction with the thermal
overload protection, this current limit protects the
output, allowing it to be shorted to ground for an indefinite period of time without damaging the part.
VCOM Buffer
The MAX1778/MAX1880–MAX1885 include a VCOM
buffer, which uses an operational transconductance
amplifier (OTA) to provide a current output that is ideal
for driving capacitive loads, such as the backplane of a
TFT LCD panel. The unity-gain bandwidth of this
current-output buffer is:
GBW = gm/COUT
where gm is the amplifier’s transconductance. The
bandwidth is inversely proportional to the output
capacitor, so large capacitive loads improve stability;
however, lower bandwidth decreases the buffer’s transient response time. To improve the transient response
MAX1778
MAX1881
MAX1883
MAX1884
SUPL
CSUPL
VSUPL
4.5V TO 15V
THERMAL
SENSOR
CURRENT
LIMIT
QP
VLDOOUT
1.25V TO (VSUPL - 0.3V)
LDOOUT
CLDOOUT
R7
FBL
ERROR
AMPLIFIER
R8
VREF
1.25V
(
VLDOOUT = 1 +
GND
)
R7
VREF
R8
VREF = 1.25V
Figure 5. Low-Dropout Linear Regulator Block Diagram
22
Maxim Integrated
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
SUPB
MAX1778
MAX1880
MAX1881
MAX1882
MAX1883
MAX1884
MAX1885
VSUPB
4.5V TO 13V
VBUFOUT
1.2V TO (VSUPB - 1.2V)
CBUF
BUFOUT
gm
BUF-
R
R11
BUF+
R
R12
125mV
GND
VBUFOUT =
( R11R12+ R12 )V
SUPB
Figure 6. VCOM Buffer Block Diagram
times, the amplifier’s transconductance increases as
the output current increases (see the Typical Operating
Characteristics).
The VCOM buffer’s positive feedback input features
dual mode operation. The buffer’s output voltage can
be internally set by a 50% resistive divider connected
to the buffer’s supply voltage (SUPB), or the output voltage can be externally adjusted for other voltages.
Shutdown (SHDN)
A logic-low level on SHDN shuts down all of the converters and the reference. When shut down, the supply
current drops to 0.1µA to maximize battery life, and the
reference is pulled to ground. The output capacitance,
feedback resistors, and load current determine the rate
at which each output voltage decays. A logic-level high
on SHDN power activates the MAX1778/
MAX1880–MAX1885 (see the Power-Up Sequencing
section). Do not leave SHDN floating. If unused, connect SHDN to IN. A logic-level transition on SHDN
clears the fault latch.
Power-Up Sequencing
Upon power-up or exiting shutdown, the MAX1778/
MAX1880–MAX1885 start a power-up sequence. First,
the reference powers up. Then, the main DC-DC stepup converter powers up with soft-start enabled. The linear regulator powers up at the same time as the main
step-up converter; however, the power sequence and
Maxim Integrated
ready output signal are not affected by the regulation of
the linear regulator. While the main step-up converter
powers up, the output of the PWM comparator remains
low (Figure 2), and the step-up converter charges the
output capacitors, limited only by the maximum duty
cycle and current-limit comparator. When the step-up
converter approaches its nominal regulation value and
the PWM comparator’s output changes states for the
first time, the negative charge pump turns on. When the
negative output voltage reaches approximately 90% of
its nominal value (VFBN < 110mV), the positive charge
pump starts up. Finally, when the positive output voltage reaches 90% of its nominal value (VFBP > 1.125V),
the active-low ready signal (RDY) goes low (see the
Power Ready section), and the VCOM buffer powers
up. The MAX1883–MAX1885 do not contain the charge
pumps, but the power-up sequence still contains the
charge pumps’ startup logic, which appears as a delay
(2 4096/fOSC) between the step-up converter reaching regulation and when the ready signal and VCOM
buffer are activated.
Soft-Start
For the main step-up regulator, soft-start allows a gradual increase of the current-limit level during startup to
reduce input surge currents. The MAX1778/MAX1880–
MAX1885 divide the soft-start period into four phases.
During the first phase, the controller limits the current
limit to only 0.38A (see the Electrical Characteristics),
approximately a quarter of the maximum current limit
23
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
(ILX(MAX)). If the output does not reach regulation within
1ms, soft-start enters phase II, and the current limit is
increased by another 25%. This process is repeated for
phase III. The maximum 1.5A (typ) current limit is
reached within 3072 clock cycles or when the output
reaches regulation, whichever occurs first (see the
startup waveforms in the Typical Operating
Characteristics).
For the charge pumps (MAX1778/MAX1880–MAX1882
only), soft-start is achieved by controlling the rate of
rise of the output voltage. Both charge-pump output
voltages are controlled to be in regulation within 4096
clock cycles, irregardless of output capacitance and
load, limited only by the charge pump’s output impedance. Although the MAX1883–MAX1885 controllers do
not include the charge pumps, the soft-start logic still
contains the 4096 clock cycle startup periods for both
charge pumps.
Fault Trip Level (FLTSET)
The MAX1778/MAX1880–MAX1885 feature dual-mode
operation to allow operation with either a preset fault
trip level or an adjustable trip level for the step-up converter and positive charge-pump outputs. Connect
FLTSET to GND to select the preset 0.9 VREF fault
threshold. The fault trip level can also be adjusted by
connecting a voltage-divider from REF to FLTSET
(Figure 8). For greatest accuracy, the total load on the
reference (including current through the negative
charge-pump feedback resistors) should not exceed
50µA so that VREF is guaranteed to be in regulation
(see the Electrical Characteristics). Therefore, select
R10 in the 100kΩ to 1MΩ range, and calculate R9 with
the following equation:
R9 = R10 [(VREF/VFLTSET) - 1]
where VREF = 1.25V, and VFLTSET can range from 0.67
x VREF to 0.85 x VREF. FLTSET’s input bias current has
a maximum value of 50nA. For 1% error, the current
through R10 should be at least 100 times the FLTSET
input bias current (IFLTSET).
Fault Condition
Once RDY is low, if the output of the main regulator or
either low-power charge pump falls below its fault
detection threshold, or if the input drops below its
undervoltage threshold, then RDY goes high impedance
and all outputs shut down; however, the reference
remains active. After removing the fault condition, toggle
shutdown (below 0.8V) or cycle the input voltage (below
0.2V) to clear the fault latch and reactivate the device.
The reference fault threshold is 1.05V. For the step-up
converter and positive charge-pump, the fault trip level is
24
set by FLTSET (see the Fault Trip Level (FLTSET) section). For the negative charge pump, the fault threshold
measured at the charge-pump’s feedback input (FBN) is
140mV (typ).
Power Ready (RDY)
RDY is an open-drain output. When the power-up
sequence for the main step-up converter and lowpower charge pumps has properly completed, the 14V
MOSFET turns on and pulls RDY low with a 125Ω (typ)
on-resistance. If a fault is detected on any of these
three outputs, the internal open-drain MOSFET appears
as a high impedance. Connect a 100kΩ pullup resistor
between RDY and IN for a logic-level output.
Voltage Reference (REF)
The voltage at REF is nominally 1.25V. The reference can
source up to 50µA with good load regulation (see the
Typical Operating Characteristics). Connect a 0.22µF
ceramic bypass capacitor between REF and GND.
Thermal-Overload Protection
Thermal-overload protection limits total power dissipation in the MAX1778/MAX1880–MAX1885. When the
junction temperature exceeds TJ = +160°C, a thermal
sensor activates the fault protection, which shuts down
the controller, allowing the IC to cool. Once the device
cools down by 15°C, toggle shutdown (below 0.8V) or
cycle the input voltage (below 0.2V) to clear the fault
latch and reactivate the controller. Thermal-overload
protection protects the controller in the event of fault
conditions. For continuous operation, do not exceed
the absolute maximum junction-temperature rating of
TJ = +150°C.
Operating Region and Power Dissipation
The MAX1778/MAX1880–MAX1885s’ maximum power
dissipation depends on the thermal resistance of the IC
package and circuit board, the temperature difference
between the die junction and ambient air, and the rate
of any airflow. The power dissipated in the device
depends on the operating conditions of each regulator
and the buffer.
The step-up controller dissipates power across the
internal n-channel MOSFET as the controller ramps up
the inductor current. In continuous conduction, the
power dissipated internally can be approximated by:
2
2⎤
⎡⎛
⎞
I
V
1 ⎛ VIND ⎞ ⎥
PSTEP − UP ≈ ⎢⎜ MAIN MAIN ⎟ +
⎢⎝
VIN
12 ⎜⎝ fOSCL ⎟⎠ ⎥
⎠
⎣
⎦
× RDS(ON)D
Maxim Integrated
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
where IMAIN includes the primary load current and the
input supply currents for the charge pumps (see the
Charge-Pump Input Power and Efficiency
Considerations section), linear regulator, and VCOM
buffer.
The linear regulator generates an output voltage by dissipating power across an internal pass transistor, so
the power dissipation is simply the load current times
the input-to-output voltage differential:
PMAX = (TJ(MAX) - TA)/(θJB + θBA)
where TJ - TA is the temperature difference between
the controller’s junction and the surrounding air, θJB (or
θJC) is the thermal resistance of the package to the
board, and θBA is the thermal resistance from the PCB
to the surrounding air.
Design Procedure
PLDO(INT) = ILDO (VSUPL - VLDO )
Main Step-Up Converter
When driving an external transistor, the internal linear
regulator provides the base drive current. Depending
on the external transistor’s current gain (β) and the
maximum load current, the power dissipated by the
internal linear regulator can still be significant:
[
]
I
PLDO(INT) = LDO VSUPL - (VLDO + 0.7V )
β
= ILDOOUT (VSUPL - VLDOOUT )
R1 = R2 [(VMAIN/VREF) - 1]
where VREF = 1.25V. VMAIN can range from VIN to 13V.
The charge pumps provide regulated output voltages
by dissipating power in the low-side n-channel
MOSFET, so they could be modeled as linear regulators followed by unregulated charge pumps. Therefore,
their power dissipation is similar to a linear regulator:
[
IPOS [(VSUPP - 2VDIODE )N +
PNEG = INEG (VSUPN - 2VDIODE )N - VNEG
PPOS =
]
VSUPD - VPOS
]
where N is the number of charge-pump stages, VDIODE
is the diodes’ forward voltage, and V SUPD is the
positive charge-pump diode supply (Figure 4).
The VCOM buffer’s power dissipation depends on the
capacitive load (C LOAD) being driven, the peak-topeak voltage change (VP-P) across the load, and the
load’s switching rate:
PBUF = VP - PCLOADfLOADVSUPB
To find the total power dissipated in the device, the
power dissipated by each regulator and the buffer must
be added together:
PTOTAL = PSTEP - UP + PLDO(INT)
+ PNEG + PPOS + PBUF
The maximum allowed power dissipation is 975mW (24pin TSSOP)/879mW (20-pin TSSOP) or:
Maxim Integrated
Output-Voltage Selection
Adjust the output voltage by connecting a voltagedivider from the output (VMAIN) to FB to GND (see the
Typical Operating Circuit). Select R2 in the 10kΩ to
50kΩ range. Calculate R1 with the following equations:
Inductor Selection
Inductor selection depends upon the minimum required
inductance value, saturation rating, series resistance, and
size. These factors influence the converter’s efficiency,
maximum output load capability, transient response time,
and output-voltage ripple. For most applications, values
between 4.7µH and 22µH work best with the controller’s
switching frequency (Tables 1 and 2).
The inductor value depends on the maximum output
load the application must support, input voltage, output
voltage, and switching frequency. With high inductor
values, the MAX1778/MAX1880–MAX1885 source higher output currents, have less output ripple, and enter
continuous conduction operation with lighter loads;
however, the circuit’s transient response time is slower.
On the other hand, low-value inductors respond faster
to transients, remain in discontinuous conduction operation, and typically offer smaller physical size for a
given series resistance and current rating. The equations provided here include a constant LIR, which is the
ratio of the peak-to-peak AC inductor current to the
average DC inductor current. For a good compromise
between the size of the inductor, power loss, and
output-voltage ripple, select an LIR of 0.3 to 0.5. The
inductance value is then given by:
2
⎛ VIN(MIN) ⎞ ⎛ VMAIN - VIN(MIN) ⎞ ⎛ 1 ⎞
LMIN = ⎜
⎟η
⎟ ⎜
⎟ ⎜
⎝ VMAIN ⎠ ⎝ IMAIN(MAX)fOSC ⎠ ⎝ LIR ⎠
25
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
where η is the efficiency, fOSC is the oscillator frequency (see the Electrical Characteristics ), and I MAIN
includes the primary load current and the input supply
currents for the charge pumps (see the Charge-Pump
Input Power and Efficiency Considerations section), linear regulator, and VCOM buffer. Considering the typical application circuit, the maximum average DC load
current (IMAIN(MAX)) is 300mA with an 8V output. Based
on the above equations and assuming 85% efficiency,
the inductance value is then chosen to be 4.7µH.
The inductor’s saturation current rating should exceed
the peak inductor current throughout the normal operating range. The peak inductor current is then given by:
⎛ IMAIN(MAX) VMAIN ⎞ ⎛
LIR ⎞ ⎛ 1 ⎞
IPEAK = ⎜
⎟⎜ ⎟
⎟ ⎜1 +
⎝
V
2 ⎠ ⎝ η⎠
IN(MIN)
⎝
⎠
Under fault conditions, the inductor current can reach
up to 1.85A (I LIM(MAX) ), see the Electrical
Characteristics). However, the controller’s fast currentlimit circuitry allows the use of soft-saturation inductors
while still protecting the IC.
The inductor’s DC resistance can significantly affect
efficiency due to the power loss in the inductor. The
power loss due to the inductor’s series resistance (PLR)
can be approximated by the following equation:
⎛I
X VMAIN ⎞
PLR ≅ RL ⎜ MAIN
⎟
VIN
⎠
⎝
2
where RL is the inductor’s series resistance. For best performance, select inductors with resistance less than the
internal n-channel MOSFET on-resistance (0.35Ω typ).
Use inductors with a ferrite core or equivalent. To minimize radiated noise in sensitive applications, use a
shielded inductor.
Output Capacitor
Output capacitor selection depends on circuit stability
and output-voltage ripple. A 10µF ceramic capacitor
works well in most applications (Tables 1 and 2).
Additional feedback compensation is required (see the
Feedback Compensation section) to increase the margin for stability by reducing the bandwidth further. In
cases where the output capacitance is sufficiently large,
additional feedback compensation is not necessary.
26
Output-voltage ripple has two components: variations
in the charge stored in the output capacitor with each
LX pulse, and the voltage drop across the capacitor’s
equivalent series resistance (ESR) caused by the
current into and out of the capacitor:
VRIPPLE = VRIPPLE(C) + VRIPPLE(ESR)
VRIPPLE(ESR) ≈ IPEAKRESR(COUT) , AND
⎛V
− VIN ⎞ ⎛ IMAIN ⎞
VRIPPLE(C) ≈ ⎜ MAIN
⎟⎜ C
⎟
VMAIN
⎝
⎠ ⎝ OUT fOSC ⎠
where I PEAK is the peak inductor current (see the
Inductor Selection section). For ceramic capacitors, the
output-voltage ripple is typically dominated by VRIPPLE(C). The voltage rating and temperature characteristics of the output capacitor must also be considered.
Feedback Compensation
For stability, add a pole-zero pair from FB to GND in the
form of a compensation resistor (RCOMP) in series with
a compensation capacitor (CCOMP), as shown in Figure
2. Select RCOMP to be half the value of R2, the low-side
feedback resistor.
Integrator Capacitor
The MAX1778/MAX1880–MAX1885 contain an internal
current integrator that improves the DC load regulation
but increases the peak-to-peak transient voltage (see
the load-transient waveforms in the Typical Operating
Characteristics). For highly accurate DC load regulation, enable the current integrator by connecting a
470pF (ƒ OSC = 1MHz)/1000pF (ƒ OSC = 500kHz)
capacitor to INTG. To minimize the peak-to-peak transient voltage at the expense of DC regulation, disable
the integrator by connecting INTG to REF. When using
the MAX1883–MAX1885, connect a 100kΩ resistor to
GND when disabling the integrator.
Input Capacitor
The input capacitor (CIN) in step-up designs reduces
the current peaks drawn from the input supply and
reduces noise injection. The value of CIN is largely
determined by the source impedance of the input supply. High source impedance requires high input capacitance, particularly as the input voltage falls. Since
step-up DC-DC converters act as “constant-power”
loads to their input supply, input current rises as input
voltage falls. A good starting point is to use the same
capacitance value for CIN as for COUT.
Maxim Integrated
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
Rectifier Diode
Use a Schottky diode with an average current rating
equal to or greater than the peak inductor current, and
a voltage rating at least 1.5 times the main output voltage (VMAIN).
charge pump’s output impedance can be approximated using the following equation:
⎛
⎞
1
RTX = 2(RPCH(ON) + RNCH(ON) ) + ⎜
⎟
⎝ CX fCHP ⎠
⎛
⎞
1
+⎜
⎟
⎝ COUT fCHP ⎠
Charge Pumps (MAX1778/ MAX1880/
MAX1881/MAX1882 Only)
Selecting the Number of Charge-Pump Stages
The number of charge-pump stages required to regulate the output voltage depends on the supply voltage,
output voltage, load current, switching frequency, the
diode’s forward voltage drop, and ceramic capacitor
values.
For positive charge-pump outputs, the number of
required stages can be determined by:
⎛
⎞
VPOS - VSUPD
NPOS ≥ ⎜
⎟
⎝ VSUPP - 1.1(2VDIODE + RTXILOAD ) ⎠
where VSUPD is the positive charge-pump diode supply
(Figure 4), VDIODE is the diode’s forward voltage drop,
and RTX is the charge pump’s output impedance. The
where the charge pump’s switching frequency (fCHP) is
equal to 0.5 x fOSC, the p-channel MOSFET’s on-resistance (RPCH(ON)) is 10Ω, and the n-channel MOSFET’s
on-resistance (R NCH(ON )) is 4Ω (see the Electrical
Characteristics).
For negative charge-pump outputs, the number of
required stages can be determined by:
⎛
⎞
VNEG
NNEG ≥ ⎜
⎟
⎝ VSUPN - 1.1(2VDROP + RTXILOAD ) ⎠
where NNEG is rounded up to the nearest integer.
Table 1. MAX1778/MAX1880/MAX1883 Component Values (fOSC = 1MHz)
VIN
CIRCUIT 1
CIRCUIT 2
CIRCUIT 3
CIRCUIT 4
CIRCUIT 5
3.3V
3.3V
3.3V
5V
5V
VMAIN
9V
9V
9V
12V
12V
IMAIN(MAX)
100mA
200mA
200mA
220mA
220mA
VNEG
-5V
-5V
-5V
-5V
-5V
INEG
2mA
5mA
5mA
5mA
5mA
VPOS
24V
24V
24V
24V
24V
IPOS
2mA
5mA
5mA
5mA
5mA
L
2.2µH
4.7µH
4.7µH
6.8µH
6.8µH
IPEAK
>1A
>1A
>1A
>1A
>1A
COUT
4.7µF
10µF
20µF
10µF
20µF
R1
309kΩ
309kΩ
309kΩ
429kΩ
429kΩ
R2
49.9kΩ
49.9kΩ
49.9kΩ
49.9kΩ
49.9kΩ
RCOMP
None
None
39kΩ*
None
20kΩ*
CCOMP
None
None
100pF*
None
200pF*
*RCOMP and CCOMP are connected between the step-up converter’s output (VMAIN) and FB.
Maxim Integrated
27
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
Table 2. MAX1881/MAX1882/MAX1884/MAX1885 Component Values (fOSC = 500kHz)
CIRCUIT 6
CIRCUIT 7
CIRCUIT 8
CIRCUIT 9
3.3V
3.3V
3.3V
3.3V
VIN
VMAIN
9V
9V
9V
9V
IMAIN(MAX)
100mA
100mA
200mA
200mA
VNEG
-5V
-5V
-5V
-5V
5mA
INEG
2mA
2mA
5mA
VPOS
24V
24V
24V
24V
IPOS
2mA
2mA
5mA
5mA
10µH
L
4.7µH
10µH
10µH
IPEAK
>1A
>1A
>1A
>1A
COUT
4.7µF
10µF
10µF
20µF
R1
309kΩ
309kΩ
309kΩ
309kΩ
R2
49.9kΩ
49.9kΩ
49.9kΩ
49.9kΩ
RCOMP
None
None
None
20kΩ*
CCOMP
None
None
None
200pF*
*RCOMP and CCOMP are connected between the step-up converter’s output (VMAIN) and FB.
Table 3. Component Suppliers
SUPPLIER
PHONE
FAX
INDUCTORS
Coilcraft
847-639-6400
847-639-1469
Coiltronics
561-241-7876
561-241-9339
Sumida USA
847-956-0666
847-956-0702
TOKO
847-297-0070
847-699-1194
AVX
803-946-0690
803-626-3123
KEMET
408-986-0424
408-986-1442
SANYO
619-661-6835
619-661-1055
Taiyo Yuden
408-573-4150
408-573-4159
Central
Semiconductor
516-435-1110
516-435-1824
International
Rectifier
310-322-3331
310-322-3332
Motorola
602-303-5454
602-994-6430
Nihon
847-843-7500
847-843-2798
Zetex
516-543-7100
516-864-7630
CAPACITORS
Charge-Pump Input Power and
Efficiency Considerations
The charge pumps in the MAX1778/MAX1880–MAX1882
provide regulated output voltages by controlling the
voltage drop across the low-side n-channel MOSFET, so
they can be modeled as linear regulators followed by an
unregulated charge pump when determining the input
power requirements and efficiency.
The charge pump only provides charge to the output
capacitor during half the period (50% duty cycle), so
the input current is a function of the number of stages
and the load current:
ISUPP = IPOS (N + 1)
DIODES
28
for the positive charge pump, and:
ISUPP = IPOS (N + 1)
for the negative charge pump, where N is the number
of charge-pump stages.
The efficiency characteristics of the MAX1778/
MAX1880–MAX1882 regulated charge pumps are similar to a linear regulator. It is dominated by quiescent
current at low output currents and by the input voltage
Maxim Integrated
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
at higher output currents (see the Typical Operating
Characteristics). So the maximum efficiency can be
approximated by:
ηPOS ≅
for the negative charge pump, where N is the stage
number in which the flying capacitor appears, and
V SUPD is the positive charge pump’s diode supply
(Figure 4). For example, the two-stage positive charge
pump in the typical application circuit (Figure 1) where
VSUPP = VSUPD = 8V contains two flying capacitors.
The flying capacitor in the first stage (C4) requires a
voltage rating over 12V. The flying capacitor in the
second stage (C6) requires a voltage rating over 24V.
VPOS
VSUPD + VSUPPN
for the positive charge pump, and:
ηNEG ≅
VNEG
VSUPNN
for the negative charge pump, where V SUPD is the
positive charge pump’s diode supply (Figure 4).
Output-Voltage Selection
Adjust the positive output voltage by connecting a voltage-divider from the output (VPOS) to FBP to GND (see
the Typical Operating Circuit ). Adjust the negative
output voltage by connecting a voltage-divider from the
output (VNEG) to FBN to REF. Select R4 and R6 in the
50kΩ to 100kΩ range. Higher resistor values improve
efficiency at low output current but increase outputvoltage error due to the feedback input bias current. For
the negative charge pump, higher resistor values also
reduce the load on the reference, which should not
exceed 50µA for greatest accuracy (including current
through the FLTSET resistors) to guarantee that VREF
remains in regulation (see the Electrical Characteristics).
Calculate the remaining resistors with the following
equations:
R3 = R4 [(VPOS/VREF) - 1]
R5 = R6 |VNEG/VREF|
where VREF = 1.25V. VPOS can range from VSUPP to
40V, and VNEG can range from 0V to -40V.
Flying Capacitor
Increasing the flying capacitor (CX) value increases the
output current capability. Above a certain point,
increasing the capacitance has a negligible effect
because the output current capability becomes dominated by the internal switch resistance and the diode
impedance. The flying capacitor’s voltage rating must
exceed the following:
[
]
VCXN(POS) > 1.5 VSUPD + VSUPP (N - 1)
for the positive charge pump, and:
Maxim Integrated
VCXN(NEG) > 1.5(VSUPNN)
Charge-Pump Output Capacitor
Increasing the output capacitance or decreasing the
ESR reduces the output ripple voltage and the peak-topeak transient voltage. With ceramic capacitors, the
output-voltage ripple is dominated by the capacitance
value. Use the following equation to approximate the
required capacitor value:
COUT ≥
ILOAD
fCHP VRIPPLE
where f CHP is typically f OSC /2 (see the Electrical
Characteristics).
Charge-Pump Input Capacitor
Use a bypass capacitor with a value equal to or greater
than the flying capacitor. Place the capacitor as close
as possible to the IC. Connect directly to power ground
(PGND).
Charge-Pump Rectifier Diodes
Use Schottky diodes with a current rating equal to or
greater than two times the average charge-pump input
current, and a voltage rating at least 1.5 times VSUPP
for the positive charge pump and V SUPN for the
negative charge pump.
Low-Dropout Linear Regulator (MAX1778/
MAX1881/MAX1883/MAX1884 Only)
Output-Voltage Selection
Adjust the linear-regulator output voltage by connecting
a voltage-divider from LDOOUT to FBL to GND
(Figure 5). Select R8 in the 5kΩ to 50kΩ range. Calculate
R7 with the following equation:
R7 = R8 [(VLDOOUT/VFBL) - 1]
where VFBL = 1.25V, and VLDOOUT can range from
1.25V to (VSUPL - 300mV). FBL’s input bias current is
0.8µA (max). For less than 0.5% error due to FBL input
bias current (IFBL), R8 must be less than 8kΩ.
29
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
Capacitor Selection and Regulator Stability
Capacitors are required at the input and output of the
MAX1778/MAX1881/MAX1883/MAX1884 for stable
operation over the full temperature range and with load
currents up to 40mA. Connect a 1µF input bypass
capacitor (CSUPL) between SUPL and ground to lower
the source impedance of the input supply. Connect a
ceramic capacitor between LDOOUT and ground,
using the following equation to determine the lowest
value required for stable operation:
⎛ ILDOOUT(MAX)
CLDOOUT ≥ 0.5ms X ⎜
⎜ VLDOOUT
⎝
⎞
⎟
⎟
⎠
For example, with a 5V linear regulator output voltage
and a maximum 40mA load, use at least 4µF of output
capacitance. Applications that experience high-current
load pulses may require more output capacitance.
The ESR of the linear regulator’s output capacitor
(CLDOOUT) affects stability and output noise. Use output
capacitors with an ESR of 0.1Ω or less to ensure stability
and optimum transient response. Surface-mount ceramic capacitors are good for this purpose. Place CSUPL
and CLDOOUT as close as possible to the linear regulator to minimize the impact of PCB trace inductance.
External Pass Transistor
For applications where the linear regulator currents
exceed 40mA or where the power dissipation in the IC
needs to be reduced, an external npn transistor can be
used. In this case, the internal LDO only provides the
necessary base drive while the external npn transistor
supports the load, so most of the power dissipation occurs
across the external transistor’s collector and emitter.
Selection of the external npn transistor is based on
three factors: the package’s power dissipation, the current gain (β), and the collector-to-emitter saturation voltage (VCE(SAT)). First, the maximum power dissipation
should not exceed the transistor’s package rating:
P = (VCOLLECTOR − VLDO ) x ILOAD(MAX)
Once the appropriate package type is selected,
consider the npn transistor’s current gain. Since the
internal LDO cannot source more than 40mA (min), the
transistor’s current gain must be high enough at the
lowest collector-to-emitter voltage to support the
maximum output load:
βMIN ≥
30
For stable operation, place a capacitor (CLDOOUT) and
a minimum load resistor (R5) at the output of the internal linear regulator (the base of the external transistor)
to set the dominant pole:
⎛ 1 ⎞
CLDOOUT ≥ 0.5ms⎜
⎟
⎝ VLDO ⎠
ILOAD(MAX) ⎞
⎛V
+ 0.7V
x ⎜ LDO
+
⎟
R5
βMIN
⎝
⎠
Since the LDO cannot sink current, a minimum pulldown resistor (R5) is required at the base of the npn
transistor to sink leakage currents and improve the
high-to-low load-transient response. Under no-load
conditions, leakage currents from the internal pass
transistor supply the output capacitor (CLDOOUT), even
when the transistor is off. As the leakage currents
increase over temperature, charge can build up on
C LDOOUT , making the linear regulator’s output rise
above its set point. Therefore, R5 must sink at least
100µA to guarantee proper regulation. Additionally, the
minimum load current provided by R5 improves the
high-to-low load transients by lowering the impedance
seen by CLDOOUT after the transient occurs. Therefore,
if large load transients are expected, select R5 so that
the minimum load current is 10% of the transistor’s
maximum base current:
R5 =
VLDO + 0.7V
= 0.1
ILDOOUT(MIN)
⎡ (V
+ 0.7V)βMIN
⎢ LDO
ILOAD(MAX)
⎢⎣
Alternatively, output capacitance placed on the external
linear regulator’s output (the emitter) adds a second pole
that could destabilize the regulator. A capacitive-divider
from the transistor’s base to the feedback input (C2 and
C3, Figure 7) circumvents this second pole by adding a
pole-zero pair. Furthermore, to minimize excessive overshoot, the capacitive-divider’s ratio must be the same as
the resistive-divider’s ratio. Once the output capacitor is
selected, using the following equations to determine the
required capacitive-divider values:
CLDO ⎛
R4 ⎞
⎜1 +
⎟
100 ⎝
R3 ⎠
C2
R4
V
=
= REF
C2 + C3
R3 + R4
VLDO
C2 + C3 ≥
ILOAD(MAX) - 40mA
40mA
Maxim Integrated
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
Input-to-Output (Dropout) Voltage and Startup
A linear regulator’s minimum input-to-output voltage differential (dropout voltage) determines the lowest useable supply voltage. Because the MAX1778/
MAX1881/MAX1883/MAX1884 use an internal pnp transistor (or external npn transistor), their dropout voltage
is a function of the transistor’s collector-to-emitter saturation voltage (see the Typical Operating
Characteristics). The linear regulator’s quiescent current increases when in dropout.
The internal linear regulator tries to start up once its
supply voltage (VSUPL) exceeds 4V. When the linear
regulator powers up, the linear regulator may be in
dropout if the linear regulator’s output set voltage is
higher than its input supply voltage. Therefore, during
this brief period, the linear regulator draws additional
supply current until the input supply voltage exceeds
the output set voltage plus the pass transistor’s saturation voltage (VLDO(SET) + VCE(SAT)).
VCOM Buffer (Operational
Transconductance Amplifier)
Buffer Output Voltage and Capacitor Selection
The positive input (BUF+) features dual-mode operation. Connect BUF+ to GND for the preset VSUPB/2 output voltage, set by an internal 50% resistive-divider.
Adjust the amplifier’s output voltage by connecting a
INPUT
VIN = 3.3V
voltage-divider from SUPB to BUF+ to GND (Figure 6).
Select R12 in the 10kΩ to 100kΩ range. Calculate R11
with the following equation:
⎡⎛ V
⎤
⎞
R11 = R12⎢⎜ SUPB ⎟ - 1⎥
⎢⎣⎝ VBUF + ⎠
⎥⎦
where VSUPB can range from 4.5V to 13V, and VBUF+
can range from 1.2V to (VSUPB - 1.2V). Connect a minimum 1µF ceramic capacitor from BUFOUT to ground.
PCB Layout and Grounding
Careful PCB layout is extremely important for proper
operation. Follow the following guidelines for good PCB
layout:
1) Place the main step-up converter output diode and
output capacitor less than 0.2in (5mm) from the LX
and PGND pins with wide traces and no vias.
2) Separate analog ground and power ground. The
ground connections for the step-up converter’s and
charge pump’s input and output capacitors should
be connected to the power ground plane. The linear regulator’s and VCOM buffer’s input and output
capacitors should be connected to a separate
power-ground path, star-connected to the PGND
pin to minimize voltage drops. When using multi-
L1
6.8µH
CIN
4.7µF
IN
C1
0.22µF
SHDN
LX
MAIN
VMAIN = 8V
COUT
(2) 4.7µF
R1
274kΩ
FB
MAX1778
MAX1883
(MAX1881)*
(MAX1884)* SUPL
LDOOUT
INTG
R2
49.9kΩ
Q1
R5
1.5kΩ
REF
CREF
0.22µF
CLDOIN
1µF
CLDOOUT
4.7µF
C2
0.01µF
R3
49.9kΩ
CLDO
1µF
LDO
VLDO = 2.5V
FBL
PGND
GND
C3
0.01µF
R4
49.9kΩ
Figure 7. External Linear Regulator
Maxim Integrated
31
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
layer boards, the top layer should contain the boost
regulator and charge-pump power ground plane,
and the inner layer should contain the analog
ground plane and power-ground plane/path for the
VCOM buffer and LDO. Connect all three ground
planes together at one place near the PGND pin.
3) Locate all feedback resistive-dividers as close as
possible to their respective feedback pins. The voltage-divider’s center trace should be kept short.
Avoid running any feedback trace near the LX
switching node or the charge-pump drivers. The
resistive-dividers’ ground connections should be to
analog ground (GND).
between fast-charging nodes on the top layer and
high-impedance nodes on the bottom layer. The
fast-charging nodes, such as the LX and chargepump driver nodes, should not have any other
traces or ground planes near by.
5) Keep the charge-pump circuitry as close as possible to the IC, using wide traces and avoiding vias
when possible. Place 0.1µF ceramic bypass
capacitors near the charge-pump input pins (SUPP
and SUPN) to the PGND pin.
6) To maximize output power and efficiency and minimize output ripple voltage, use extra-wide, powerground traces, and solder the IC’s power-ground
pin directly to it.
Refer to the MAX1778/MAX1880–MAX1885 evaluation
kit for an example of proper board layout.
4) When using multilayer boards, separate the top signal layer and bottom signal layer with a ground
plane between to eliminate capacitive coupling
INPUT
VIN = 5V
L1
10µH
CIN
(2) 4.7µF
C1
0.22µF
RRDY
100kΩ
TO LOGIC
IN
LX
SHDN
FB
R1
86.6kΩ
RDY
LDO
VLDO = 3.3V
LDOOUT
CLDOOUT
4.7µF
C6
1µF
R8
10kΩ
R7
16.4kΩ
R8
1.5kΩ
C4
0.1µF
MAX1778
DRVP
C6
0.01µF
FBL
C7
0.01µF
NEGATIVE
VNEG = -8V
SUPB
SUPN
SUPP
FBP
R4
49.9kΩ
DRVN
C2
0.1µF
FBN
C3
1.0µF
MAIN
VMAIN = 12V
CCOMP
470pF
R2
10kΩ
SUPL
Q1
RCOMP
4.7kΩ
COUT
(2) 10µF
R5
316kΩ
R6
49.9kΩ
R3
750kΩ
CBUF
1.0µF
FLTSET
REF
CREF
0.22µF
BUFOUT
BUF-
INTG
PGND
BUF+
GND
TGND
C5
1.0µF
POSITIVE
VPOS = 20V
BUFFER OUTPUT
VBUFOUT = VSUPB/2
R9
30kΩ
REF
R10
100kΩ
Figure 8. 5V Input Monitor Application
32
Maxim Integrated
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
Applications Information
Larger output capacitors with higher voltage ratings
allow configurations with output voltages above 10V.
Additionally, physically larger inductors with less series
resistance and higher saturation ratings provide more
output current and higher efficiency.
Low-Profile Components
Notebook applications generally require low-profile
components, potentially limiting the circuit’s performance. For example, low-profile inductors typically
have lower saturation ratings and more series resistance, limiting output current and efficiency. Low-profile
capacitors have lower voltage ratings for a given
capacitance value, so 3.3µF low-profile capacitors with
voltage ratings greater than 10V were not available at
the time of publication.
Input Voltage Above and
Below the Output Voltage
Combining the step-up converter and linear regulator
as shown in Figure 9 provides output-voltage regulation
above and below the input voltage. Supplied by the
step-up converter, the linear regulator output provides
a constant output voltage (VLDO). When the input voltage exceeds the main step-up converter’s nominal output voltage, the controller stops switching but the linear
regulator maintains the output voltage. When the input
voltage drops below the output voltage, the step-up
Desktop Monitors
Monitor applications do not have the same component
height restrictions associated with laptops, allowing
more flexibility in component selection (Figure 8).
POWER INPUT
VBATT = 10V TO 15V
INPUT
VIN = 3.3V TO 5V
L1
6.8µH
CIN
4.7µF
LX
IN
C1
0.1µF
FB
RDY
SUPL
BUFOUT
CBUF
1.0µF
BUFC2
0.1µF
R5
475kΩ
SUPB
SUPN
SUPP
R7
470kΩ
R9
6.8kΩ
LDO
VLDO = 13V
CLDO
(2) 3.3µF
R8
49.9kΩ
C4
0.1µF
REF
DRVP
R9
30kΩ
R10
100kΩ
FBP
FLTSET
CINTG
470pF
C7
0.1µF
FBL
R6
49.9kΩ
CREF
0.22µF
CLDOOUT
3.3µF
C6
0.1µF
FBN
C3
1.0µF
Q1
LDOOUT
MAX1778
DRVN
NEGATIVE
VNEG = -12V
R2
49.9kΩ
SHDN
RRDY
100kΩ
TO LOGIC
BUFFER OUTPUT
VBUFOUT = VSUPB/2
COUT
(3) 3.3µF
R1
511kΩ
INTG
PGND
BUF+
GND
TGND
R4
49.9kΩ
R3
909kΩ
C5
1.0µF
POSITIVE
VPOS = 24V
Figure 9. Input Voltage Above and Below the Output Voltage
Maxim Integrated
33
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
INPUT
VIN = 3.3V
L1
6.8µH
CIN
4.7µF
STARTUP MAIN
VMAIN(START) = 8V
C1
0.22µF
SHDN
C8
3.3µF
COUT
(2) 3.3µF
R1
274kΩ
LX
IN
SYSTEM MAIN
VMAIN(SYS) = 8V
FB
R2
49.9kΩ
C10
0.1µF
SUPP
C4
0.1µF
MAX1778
C5
1.0µF
DRVP
C6
0.1µF
INTG
R7
10kΩ
C7
1.0µF
REF
CREF
0.22µF
R3
750kΩ
R9
30kΩ
FBP
FLTSET
R4
49.9kΩ
R10
100kΩ
STARTUP
POSITIVE
VPOS(START) = 20V
SYSTEM
POSITIVE
VPOS(SYS) = 20V
Q3
Q2
INPUT
VIN = 3.3V
RDY
TGND
PGND
GND
RRDY
5.1kΩ
R8
100kΩ
Figure 10. Power-Up Sequencing and Fault Protection
converter steps up the input voltage so that the linear
regulator will not drop out. Therefore, to guarantee that
the external pass transistor does not saturate, the stepup converter’s output voltage must be set above the linear regulator’s output voltage plus the transistor’s
saturation rating (VMAIN ≥ VLDO + VSAT).
Power-Up Sequencing and Fault Protection
The MAX1778/MAX1880–MAX1885’s fault protection
cannot be activated until the power-up sequence is
successfully completed and the power-ready output
goes low. Therefore, faults on the main output or positive charge-pump output could damage the controller
or external components. Additional fault protection can
be added as shown in Figure 10. The external MOSFET
and pnp transistor isolate the positive outputs during
startup. When the controller finishes the power-up
sequence, the power-ready output goes low, turning on
34
the pnp transistor. Any fault on the positive chargepump output pulls down the charge pump’s output voltage and triggers the fault protection; otherwise, the
MOSFET’s gate slow charges. Once the MOSFET turns
on, any faults on the main step-up converter’s output
pull down the main output voltage and trigger the fault
protection.
VCOM Buffer Startup
The VCOM buffer does not include soft-start. Therefore,
once the VCOM buffer turns on, it draws high surge
currents while charging the output capacitance. In
some applications, the buffer’s high startup surge
current could potentially trip the fault-detection circuit,
forcing the controller to shut down. In these cases,
adding a soft-start resistive-divider between SUPB and
BUFOUT reduces the startup surge current and voltage
drops associated with this load (Figure 11), as shown in
Maxim Integrated
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
INPUT
VIN = 3.3V
L1
6.8µH
CIN
4.7µF
LX
IN
C1
0.22µF
SHDN
R2
49.9kΩ
SUPB
REF
CREF
0.22µF
R1
274kΩ
FB
MAX1778
INTG
MAIN
VMAIN = 8V
COUT
(2) 4.7µF
BUF-
R3
10kΩ
CSUPB
1.0µF
BUFFER OUTPUT
VBUFOUT = VSUPB/2
BUFOUT
BUF+
PGND
GND
R4
10kΩ
CBUF
1.0µF
[( VV
R3 = R4
SUPB
BUFOUT
) -1]
Figure 11. VCOM Buffer Soft-Start
the Typical Operating Characteristics. Set the resistivedivider to precharge BUFOUT, matching the buffer’s
output set voltage:
⎡⎛ V
⎤
⎞
R 3 = R4 ⎢⎜ SUPB ⎟ − 1⎥
⎢⎣⎝ VBUFOUT ⎠
⎥⎦
These resistor values are selected to charge the output
capacitor close to the output set voltage before the
buffer starts up:
CBUFOUT (R3 || R4) ≈
Maxim Integrated
5000
fOSC
Selector Guide
PART
STEP-UP
SWITCHING
FREQUENCY
(Hz)
DUAL
CHARGE
PUMPS
LINEAR
REGULATOR
MAX1778
1M
Yes
Yes
MAX1880
1M
Yes
No
MAX1881
500k
Yes
Yes
MAX1882
500k
Yes
No
MAX1883
1M
No
Yes
MAX1884
500k
No
Yes
MAX1885
500k
No
No
35
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
Typical Operating Circuit
MAIN
INPUT
TO LOGIC
LDO OUTPUT
IN
LX
SHDN
FB
RDY
SUPL
LDOOUT
SUPB
SUPN
SUPP
FBL
MAX1778
DRVP
DRVN
NEGATIVE
FBP
POSITIVE
FBN
REF
BUFOUT
BUF-
BUFFER OUTPUT
BUF+
INTG
PGND
36
FLTSET
GND
TGND
Maxim Integrated
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
Pin Configurations
TOP VIEW
FB 1
+
24 RDY
INTG 2
23 TGND
IN 3
22 LX
21 PGND
BUF+ 4
MAX1778
MAX1881
BUF- 5
FB 1
+
INTG 2
23 TGND
IN 3
22 LX
21 PGND
BUF+ 4
MAX1880
MAX1882
20 DRVP
20 DRVP
BUF- 5
SUPB 6
19 SUPP
SUPB 6
19 SUPP
BUFOUT 7
18 DRVN
BUFOUT 7
18 DRVN
GND 8
17 SUPN
GND 8
17 SUPN
REF 9
16 FLTSET
REF 9
16 FLTSET
FBP 10
15 FBL
FBP 10
15 N.C.
FBN 11
14 LDOOUT
FBN 11
14 N.C.
SHDN 12
13 N.C.
SHDN 12
13 SUPL
TSSOP
TOP VIEW
TSSOP
+
+
FB 1
20 RDY
INTG 2
19 TGND
IN 3
18 LX
BUF+ 4
BUF- 5
SUPB 6
MAX1883
MAX1884
BUFOUT 7
FB 1
19 TGND
IN 3
17 PGND
BUF+ 4
16 N.C.
BUF- 5
15 N.C.
SUPB 6
14 FLTSET
20 RDY
INTG 2
18 LX
17 PGND
MAX1885
16 N.C.
15 N.C.
BUFOUT 7
14 FLTSET
GND 8
13 FBL
GND 8
13 N.C.
REF 9
12 LDOOUT
REF 9
12 N.C.
SHDN 10
11 N.C.
SHDN 10
11 SUPL
TSSOP
Maxim Integrated
24 RDY
TSSOP
37
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
Package Information
Chip Information
TRANSISTOR COUNT: 3739
38
For the latest package outline information and land patterns (footprints), go to www.maximintegrated.com/packages. Note that a
“+”, “#”, or “-” in the package code indicates RoHS status only.
Package drawings may show a different suffix character, but the
drawing pertains to the package regardless of RoHS status.
PACKAGE
TYPE
PACKAGE
CODE
OUTLINE NO.
LAND
PATTERN NO.
20 TSSOP
U20-2
21-0066
90-0116
24 TSSOP
U24-1
21-0066
90-0118
Maxim Integrated
MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
Revision History
REVISION
NUMBER
REVISION
DATE
2
10/12
DESCRIPTION
Added MAX1880EUG/V+ to Ordering Information
PAGES
CHANGED
1
Maxim Integrated cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim Integrated product. No circuit patent
licenses are implied. Maxim Integrated reserves the right to change the circuitry and specifications without notice at any time. The parametric values (min and
max limits) shown in the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance.
Maxim Integrated 160 Rio Robles, San Jose, CA 95134 USA 1-408-601-1000 ________________________________ 39
© 2012 Maxim Integrated Products, Inc.
Maxim Integrated and the Maxim Integrated logo are trademarks of Maxim Integrated Products, Inc.

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