LT3513 2MHz High Current 5-Output Regulator for TFT-LCD Panels FEATURES DESCRIPTION n The LT®3513 5-output adjustable switching regulator provides power for large TFT-LCD panels. The 38-pin 5mm × 7mm QFN device can generate a 3.3V or 5V logic supply along with the triple output supply required for the TFT-LCD panel. A lower voltage secondary logic supply may also be generated with the addition of an external NPN driven by the internal linear regulator. A step-down regulator provides a low voltage output, VLOGIC, with up to 1.2A of current while capable of operating from a wide input range of 4.5V to 30V. A high power step-up converter, a lower power step-up converter and an inverting converter provide the three independent output voltages: AVDD, VON and VOFF required by the LCD panel. A high-side PNP provides delayed turn-on of the VON signal and can handle up to 30mA. Protection circuitry ensures that VON is disabled if any of the four outputs are more than 10% below the programmed voltage. n n n n n n n n n n 4.5V to 30V Input Voltage Range Four Integrated Switches: 2.2A Buck, 1.5A Boost, 0.25A Boost, 0.25A Inverter (Guaranteed Minimum Current Limit) External NPN LDO Driver Fixed Frequency, Low Noise Outputs Inductor Current Sense for Buck Soft-Start for All Outputs Externally Programmable VON Delay Three Integrated Schottky Diodes PGOOD Pin for AVDD Output Disconnect PanelProtectTM Circuitry Disables VON Upon Fault Thermally Enhanced 38-Lead 5mm × 7mm QFN Package APPLICATIONS n n n Automotive TFT-LCD Displays Large TFT-LCD Desktop Monitors Flat Panel Televisions L, LT, LTC and LTM are registered trademarks of Linear Technology Corporation. PanelProtect is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. TYPICAL APPLICATION VIN 8V TO 16V 10μH VLOGIC 5V 10μF 178k 53.6k 100k 10k 10μF 0.47μF VOFF –10V 20mA 2.2μF 60.4k UVLO LDOPWR VIN 10μH SW2 SW4 69.8k PGOOD D4 BIAS BOOST 4.7μH VLOGIC 5V 0.5A RUN-SS2 RUN-SS3/4 CT 0.22μF 47nF SW1 SENSE+ SENSE– FB1 30.1k 22μF Start-Up Waveforms RUN-SS1 NFB4 10k AVDD 8V 80mA FB2 15nF 15nF 15nF LT3513 VON_CLK VON_CLK VON VONSINK 10k VON 22V 20mA 232k E3 6.8μH VLDO 3.3V 0.5A SW3 FB3 BD FB5 VC1 42.2k 10μF 7.5k 10k 2.7nF VC2 GND 4.7k VC3 30k 4.7nF 1.5nF VC4 VE3 20V/DIV VON 20V/DIV 165k VLOGIC 5V 10k RUN/SS 2V/DIV VLOGIC 5V/DIV AVDD 10V/DIV VOFF 10V/DIV IIN(AVG) 1A/DIV 0.47μF 2.2μF 5ms/DIV 3513 TA01b 13k 2.2nF 3513fa 1 LT3513 ABSOLUTE MAXIMUM RATINGS PIN CONFIGURATION (Note 1) FB1 GND SW1 SW1 VIN VIN UVLO TOP VIEW 38 37 36 35 34 33 32 FB5 1 31 SENSE+ VC1 2 30 SENSE– RUN-SS3/4 3 29 BIAS FB3 4 28 BOOST RUN-SS2 5 27 LDOPWR SW3 6 26 BD 39 E3 7 25 SW4 VON 8 24 D4 VONSINK 9 23 NFB4 VON_CLK 10 22 RUN-SS1 PGOOD 11 21 VC4 20 VC2 VC3 12 FB2 BIAS GND SW2 SW2 CT 13 14 15 16 17 18 19 GND VIN, LDOPWR Voltage ...............................................32V UVLO Voltage ............................................................32V SW2, SW3, SW4 Voltage ..........................................40V E3 Pin Voltage ...........................................................40V VON, VONSINK Voltage ................................................40V PGOOD Voltage .........................................................40V D4 Voltage ........................................................ 1V, –40V BOOST Voltage .........................................................37V BOOST Over SW1 .......................................................8V SENSE+, SENSE– Voltage ..........................................10V VON_CLK Voltage ........................................................10V BIAS, BD Voltage ......................................................10V CT Pin Voltage .............................................................5V RUN-SS1, RUN-SS2, RUN-SS3/4 Voltage ..................5V FB1, FB2, FB3, FB5 Voltage .........................................5V NFB4 Voltage ...................................................... 5V, –5V VC1, VC2, VC3, VC4 Voltage ..........................................5V Junction Temperature ........................................... 125°C Operating Temperature Range (Note 2).. –40°C to 125°C Storage Temperature Range................... –65°C to 125°C UHF PACKAGE 38-LEAD (5mm s 7mm) PLASTIC QFN TJMAX = 125°C, θJA = 34°C/W, θJC = 1°C/W EXPOSED PAD (PIN 39) IS GND, MUST BE SOLDERED TO PCB ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT3513EUHF#PBF LT3513EUHF#TRPBF 3513 38-Lead (5mm × 7mm) Plastic QFN –40°C to 125°C LT3513IUHF#PBF LT3513IUHF#TRPBF 3513 38-Lead (5mm × 7mm) Plastic QFN –40°C to 125°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, BIAS = 3V, unless otherwise noted. PARAMETER CONDITIONS MIN Minimum Input Voltage Quiescent Current Not Switching VRUNSS1 = 0V RUN-SS1, RUN-SS2, RUN-SS3/4 Pin Current RUN-SS1= RUN-SS2 = RUN-SS3 = RUN-SS4 = 0.4V 7.5 30 RUN-SS1, RUN-SS2, RUN-SS3/4 Threshold l BIAS Pin Voltage to Begin RUN-SS2, RUN-SS3/4 BIAS Pin Current TYP l BIAS = 3.1V, All Switches Off MAX UNITS 4.5 V 12 65 mA μA 2 μA 0.8 V 2.25 2.7 V 16.5 20 mA 3513fa 2 LT3513 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, BIAS = 3V, unless otherwise noted. PARAMETER CONDITIONS MIN TYP MAX UNITS FB Threshold Offset to Begin CT Charge (Note 3) 90 125 160 mV CT Pin Current Source All FB Pins = 1.5V 16 20 25 μA CT Threshold to Power VON All FB Pins = 1.5V 1 1.1 1.2 V 200 400 VON Switch Drop VON Current = 30mA Maximum VON Current VE3 = 30V l VON_CLK Input Voltage High 30 50 1.5 V VON_CLK Input Voltage Low VONSINK Voltage On VONSINK Current = 1μA ● Master Oscillator Frequency ● 1.90 1.80 mV mA 2 0.3 V 1.2 V 2.12 2.22 MHz MHz Foldback Switching Frequency All FB = 0V 200 kHz UVLO Pin Threshold UVLO Pin Voltage Rising 1.25 V UVLO Pin Hysteresis Current VUVLO = 1V PGOOD Threshold Offset PGOOD Sink Current PGOOD Connected to 40V Through 100k PGOOD Pin Leakage VPGOOD = 40V 3.4 3.9 4.4 μA 90 125 160 mV 4 mA 1 μA 1.235 1.255 1.265 V V 0.01 0.03 %/V 30 200 nA Switch 1 (2.2A Buck) FB1 Voltage ● FB1 Voltage Line Regulation 4.5V < VIN < 32V FB1 Pin Bias Current (Note 4) 1.215 1.205 ● Error Amplifier 1 Voltage Gain Error Amplifier 1 Transconductance ΔI = 10μA ● Maximum Duty Cycle 75 2.2 250 V/V 220 μmhos 85 % Switch 1 Current Limit Duty Cycle = 35% (Note 5) Switch 1 VCESAT ISW = 1.5A 3 3.5 Switch 1 Leakage Current FB1 = 1.5V 0.1 10 μA Minimum BOOST Voltage Above SW1 Pin ISW = 1.5A (Note 6) 1.8 2.5 V BOOST Pin Current ISW = 1.5A 30 50 mA BOOST Schottky Diode Drop I = 170mA 700 430 A mV mV Switch 2 (1.5A BOOST) FB2 Voltage ● FB2 Voltage Line Regulation 4.5V < VIN < 32V FB2 Pin Bias Current (Note 4) 1.20 1.19 ● Error Amplifier 2 Voltage Gain Error Amplifier 2 Transconductance ΔI = 10μA Switch 2 Current Limit 1.5 1.22 1.24 1.25 V V 0.01 0.03 %/V 30 200 V/V 220 μmhos 1.85 Switch 2 VCESAT ISW2 = 1.2A 360 Switch 2 Leakage Current FB2 = 1.5V 0.1 BIAS Pin Current Due to SW2 ISW2 = 1.2A Maximum Duty Cycle (SW2) ● 75 nA 250 2.3 A mV 1 μA 45 mA 90 % 3513fa 3 LT3513 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, BIAS = 3V, unless otherwise noted. PARAMETER CONDITIONS MIN TYP MAX UNITS 1.20 1.19 1.22 1.24 1.25 V V 0.01 0.03 %/V 30 200 nA Switch 3 (250mA BOOST) FB3 Voltage ● FB3 Voltage Line Regulation 4.5V < VIN < 32V FB3 Pin Bias Current (Note 4) ● Error Amplifier 3 Voltage Gain Error Amplifier 3 Transconductance ΔI = 10μA Switch 3 Current Limit 0.25 250 V/V 220 μmhos 0.3 Switch 3 VCESAT ISW3 = 0.2A 200 Switch 3 Leakage Current FB3 = 1.5V 0.1 BIAS Pin Current Due to SW3 ISW3 = 0.2A 18 ● Maximum Duty Cycle (SW3) Schottky Diode Drop 84 I = 170mA 0.38 A mV 1 μA mA 88 % 900 mV Switch 4 (250mA Inverter) NFB4 Voltage ● NFB4 Voltage Line Regulation 4.5V < VIN < 32V NFB4 Pin Bias Current (Note 4) –1.205 –1.215 ● Error Amplifier 4 Voltage Gain Error Amplifier 4 Transconductance ΔI = 10μA Switch 4 Current Limit 0.28 –1.180 –1.155 –1.145 0.01 0.03 %/V 5 16 μA 200 V/V 220 μmhos 0.3 Switch 4 VCESAT ISW4 = 0.2A 200 Switch 4 Leakage Current NFB4 = –1.5V 0.1 BIAS Pin Current Due to SW4 ISW4 = 0.2A 18 Maximum Duty Cycle (SW4) Schottky Diode Drop (D4) 84 I = 170mA V V 0.40 A mV 1 μA mA 88 % 700 mV NPN LDO FB5 Voltage ● 0.61 0.6 0.625 0.63 0.65 8 10 Base Drive Current FB5 = 0.5V 6 LDOPWR Minimum Voltage BD = 3.5V 4.5 Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LT3513E is guaranteed to meet specified performance from 0°C to 125°C junction temperature. Specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LT3513I is guarenteed over the full –40°C to 125°C operating junction temperature range. Note 3: The CT pin is held low until FB1, FB2, FB3 and NFB4 all ramp above the FB threshold offset. V V mA V Note 4: Current flows out of FB1, FB2, FB3 and NFB4. Note 5: Current limit is guaranteed by design and/or correlation to static test. Slope compensation reduces current limit at higher duty cycles. Note 6: This is the minimum voltage across the boost capacitor needed to guarantee full saturation of the internal power switch. Note 7: This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed the maximum operating junction temperature range when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. 3513fa 4 LT3513 TYPICAL PERFORMANCE CHARACTERISTICS Maximum Output Current for VLOGIC = 3.3V Start and Run VLOGIC = 3.3V SW1 Current Limit vs Duty Cycle 3.0 2.6 8 SW1 CURRENT LIMIT vs DUTY CYCLE 2.4 7 2.5 2.2 L = 2.4μH 2.0 1.8 1.6 MINIMUM 5 1.5 2 1 0 1.2 5 0 10 25 20 15 35 VIN (V) 45 55 DUTY CYCLE (%) 0 0.001 75 65 BOOST Pin Current SW3 Current Limit SW2 Current Limit 500 2.4 450 20 SW CURRENT LIMIT (mA) 2.3 SW CURRENT LIMIT (A) BOOST CURRENT (mA) 60 30 2.2 2.1 2.0 1.9 1.8 1.7 10 0 500 1000 1500 2000 2500 SWITCH CURRENT (mA) 1.5 –40 3000 350 300 250 200 100 –50 –30 –10 10 30 50 70 90 110 AMBIENT TEMPERATURE (°C) 60 35 110 10 85 –15 AMBIENT TEMPERATURE (°C) 3513 G05 3513 G04 SW4 Current Limit 3513 G06 SW1 VCESAT 500 1000 450 900 SW2 VCESAT 600 500 800 400 300 250 VCE2SAT (mV) 700 350 VCESAT (mV) SW CURRENT LIMIT (mA) 400 150 1.6 0 600 500 400 200 150 400 300 200 300 200 1 3513 G03 2.5 70 40 0.01 0.1 LOAD CURRENT (A) 3513 G02 3513 G01 50 VIN(MIN) RUN 4 3 1.0 0.5 1.4 VIN(MIN) START 6 2.0 VIN (V) SW1 CURRENT LIMIT (A) L = 4.3μH IOUT(MAX) (A) TA = 25°C, unless otherwise noted. 100 100 100 –50 –30 –10 10 30 50 70 90 110 AMBIENT TEMPERATURE (°C) 3513 G07 0 0 0 500 1000 1500 2000 SW1 CURRENT (mA) 2500 3000 3513 G08 0 400 800 1200 ISW2 (mA) 1600 2000 3513 G09 3513fa 5 LT3513 TYPICAL PERFORMANCE CHARACTERISTICS SW3 VCESAT TA = 25°C, unless otherwise noted. SW4 VCESAT 300 350 250 300 VON Current Limit 45 40 150 ION LIMIT (mA) 200 VCESAT (mV) VCE3SAT (mV) 35 250 200 150 100 100 30 25 20 15 10 50 50 0 0 0 50 100 150 200 250 ISW3 (mA) 300 350 5 0 50 100 150 200 ISW (mA) 250 300 3513 G10 0 350 0 5 10 15 20 VE3 (V) 25 3513 G11 Oscillator Frequency 3513 G12 Frequency Foldback 2.5 35 30 Reference Voltage 2500 1.25 2000 1.24 FREQUENCY (MHz) 2.3 2.2 2.1 2.0 1.9 REFERENCE VOLTAGE (V) SWITCHING FREQUENCY (kHz) 2.4 1500 1000 1.23 1.22 1.21 500 1.8 1.7 –50 0 50 0 100 AMBIENT TEMPERATURE (°C) 0 3513 G13 BIAS Pin Current EFFICIENCY (%) BIAS CURRENT (mA) ISW2 = 0A ISW3 = 0A ISW4 = 0A 20 10 0 –50 100 90 90 80 80 70 60 50 100 TEMPERATURE (°C) 150 3513 G16 70 60 50 50 0 110 Efficiency, VLOGIC = 5V 100 EFFICIENCY (%) L2 = 10μH L3 = 10μH L4 = 10μH 30 60 35 10 85 TEMPERATURE (°C) 3513 G15 Efficiency, AVDD = 13V 40 –15 3513 G14 60 50 1.20 –40 150 300 450 600 750 900 1050 1200 VFB (mV) 40 1 100 200 300 400 LOAD CURRENT (mA) 500 3513 G17 40 100 300 500 700 900 1100 1300 1500 IOUT (mA) 3513 G18 3513fa 6 LT3513 TYPICAL PERFORMANCE CHARACTERISTICS VUVLO vs Temperature 1.33 8 1.32 660 UVLO FOR START 7 1.31 6 5 4 3 Reference Voltage for FB5, LDO 670 1.30 UVLO MINIMUM FOR RUN 1.29 1.28 2 0 –50 0 25 –25 75 100 50 AMBIENT TEMPERATURE (°C) 125 1.26 –50 650 640 630 620 610 1.27 1 REFERENCE VOLTAGE (mV) 9 UVLO (V) BASE CURRENT LIMIT OF INTERNAL PNP (mA) LDO Current Limit vs Temperature TA = 25°C, unless otherwise noted. 0 50 100 AMBIENT TEMPERATURE (oC) 3513 G19 3513 G20 600 –40 –15 60 35 85 10 TEMPERATURE (°C) 110 3513 G21 PIN FUNCTIONS FB5 (Pin 1): Feedback Pin. Tie the resistor tap to this pin and set the output of the LDO according to VLDO = 0.625 • (1 + R14/R15). Reference designators refer to Figure 1. VC1 (Pin 2): Control Voltage and Compensation Pin for Internal Error Amplifier. Connect a series RC from this pin to ground to compensate switching regulator 1. RUN-SS3/4 (Pin 3): Run/Soft-Start Pin. This is the softstart pin for switching regulators 3 and 4. Place a soft-start capacitor here to limit start-up inrush current and output voltage ramp rate. When the BIAS pin reaches 2.8V, a 2μA current source charges the capacitor. When the voltage at this pin reaches 0.8V, switches 3 and 4 turn on and begin switching. For slower start-up use a larger capacitor. For complete shutdown tie RUN-SS3/4 to ground. FB3 (Pin 4): Feedback Pin. Tie the resistor tap to this pin and set VON according to VON = 1.23V • (1 + R8/R9) – 150mV. Reference designators refer to Figure 1. RUN-SS2 (Pin 5): Run/Soft-Start Pin. This is the soft-start pin for switching regulator 2. Place a soft-start capacitor here to limit start-up inrush current and output voltage ramp rate. When the BIAS pin reaches 2.25V, a 2μA current source charges the capacitor. When the voltage at this pin reaches 0.8V, switch 2 turns on and begins switching. For slower start-up use a larger capacitor. For complete shutdown tie RUN-SS2 to ground. SW3 (Pin 6): Switch Node. The SW3 pin is the collector of the internal NPN bipolar transistor for switching regulator 3. Minimize trace area at this pin to keep EMI down. E3 (Pin 7): This is switching regulator 3’s output and the emitter of the output disconnect PNP. Tie the output capacitor and resistor divider here. VON (Pin 8): This is the delayed output for switching regulator 3. VON reaches its programmed voltage after the internal CT timer times out. Protection circuitry ensures VON is disabled if any of the four outputs are more than 10% below normal voltage. This output is also disabled when VON_CLK is high. VONSINK (Pin 9): This is an open-collector output controlled by the VON_CLK pin. When VON_CLK is low, this pin draws no current and when VON_CLK is high, this pin draws current. VON_CLK (Pin 10): This pin controls the output disconnect device and the open collector of VONSINK. When this pin is low, the VON pin is enabled and the VONSINK pin is a high impedance. When this pin is high, the VON pin is disabled and the VONSINK pin sinks current to ground. 3513fa 7 LT3513 PIN FUNCTIONS PGOOD (Pin 11): Power Good Comparator Output. This is the open collector output of the power good comparator and can be used in conjunction with an external P-channel MOSFET to provide output disconnect for AVDD as shown in Figure 2. When switcher 2’s output reaches approximately 90% of its programmed voltage, PGOOD will be pulled to ground. This will pull down on the gate of the MOSFET, connecting AVDD. A 100k pull-up resistor between the source and the gate of the P-channel MOSFET keeps it off when switcher 2’s output is low. VC3 (Pin 12): Control Voltage and Compensation Pin for Internal Error Amplifier. Connect a series RC from this pin to ground to compensate switching regulator 3. CT (Pin 13): Timing Capacitor Pin. This is the input to the VON timer and programs the time delay from all four feedback pins reaching 1.125V to VON turning on. The CT capacitor value can be set using the equation C = (20μA • tDELAY)/1.1V. GND (Pins 14, 17, 33): Ground. SW2 (Pins 15, 16): Switch Node. The SW2 pin is the collector of the internal NPN bipolar transistor for switching regulator 2. Minimize trace area at this pin to keep EMI down. BIAS (Pins 18, 29): The BIAS pin is used to improve efficiency when operating at higher input voltages. Connecting this pin to the output of switching regulator 1 forces most of the internal circuitry to draw its operating current from VLOGIC rather than VIN. The drivers of switches 2, 3, 4 and 5 are supplied by BIAS. Switches 2, 3, 4 and 5 will not switch until the BIAS pin reaches approximately 2.7V. Both BIAS pins must be tied to VLOGIC. FB2 (Pin 19): Feedback Pin. Tie the resistor divider tap to this pin and set AVDD according to AVDD = 1.23V • (1 + R5/R6). Reference designators refer to Figure 2. VC2 (Pin 20): Control Voltage and Compensation Pin for Internal Error Amplifier. Connect a series RC from this pin to ground to compensate switching regulator 2. VC4 (Pin 21): Control Voltage and Compensation Pin for Internal Error Amplifier. Connect a series RC from this pin to ground to compensate switching regulator 4. RUN-SS1 (Pin 22): Run/Soft-Start Pin. This is the soft-start pin for switching regulator 1. Place a soft-start capacitor here to limit start-up inrush current and output voltage ramp rate. When power is applied to the VIN pin, a 2μA current source charges the capacitor. When the voltage at this pin reaches 0.8V, switch 1 turns on and begins switching. For slower start-up use a larger capacitor. For complete shutdown tie RUN-SS1 to ground. NFB4 (Pin 23): Negative Feedback Pin. Tie the resistor divider tap to this pin and set VOFF according to VOFF = –1.18 • (1 + R3/R4). Reference designators refer to Figure 2. D4 (Pin 24): Internal Schottky Diode Pin. This pin is the anode of an internal Schottky diode with the other end connected to ground. This Schottky diode is used in generating the VOFF output. SW4 (Pin 25): Switch Node. The SW4 pin is the collector of the internal NPN bipolar transistor for switching regulator 4. Minimize trace area at this pin to keep EMI down. BD (Note 26): NPN LDO Base Drive. This pin controls the base of the external NPN LDO transistor. LDOPWR (Pin 27): Input Voltage for LDO Driver. This pin supplies the current for the NPN LDO base. This pin can be connected to VIN. To save power at high VIN voltages, the pin can alternatively be connected to the AVDD supply. BOOST (Pin 28): The BOOST pin is used to provide a drive voltage higher than VIN to the switch 1 drive circuit. An internal Schottky diode is connected between BIAS and BOOST. A capacitor needs to be connected between BOOST and SW1. SENSE– (Pin 30) Negative Current Sense Input. This pin (along with the SENSE+ pin) is used to sense the inductor current for the buck switching regulator. 3513fa 8 LT3513 PIN FUNCTIONS SENSE+ (Pin 31) Positive Current Sense Input. This pin (along with the SENSE– pin) is used to sense the inductor current for the buck switching regulator. FB1 (Pin 32): Feedback Pin. Tie the resistor divider tap to this pin and set VLOGIC according to VLOGIC = 1.23V • (1 + R1/R2). Reference designators refer to Figure 2. SW1 (Pins 34, 35): Switch Node. The SW1 pins are the emitter of the internal NPN bipolar power transistor for switching regulator 1. These points must be tied together for proper operation. Connect these pins to the inductor, catch diode and boost capacitor. VIN (Pins 36, 37): Input Voltage. This pin supplies current to the internal circuitry of the LT3513. This pin must be locally bypassed with a capacitor. UVLO (Pin 38): Undervoltage Lockout. A resistor divider connected to VIN is tied to this pin to program the minimum input voltage at which the LT3513 will operate. This pin is compared to the internal 1.25V reference. When UVLO is less than 1.25V, the switching regulators are not allowed to operate (the RUN/SS pins are still used to turn on each switching regulator). When this pin falls below 1.25V, 3.9μA will be pulled from the pin to provide programmable hysteresis for UVLO. Exposed Pad (Pin 39): Ground. The Exposed Pad of the package provides both electrical contact to ground and good thermal contact to the printed circuit board. The Exposed Pad must be soldered to the circuit board for proper operation. 3513fa 9 LT3513 BLOCK DIAGRAM LDOPWR + – 27 + gm 26 1 BD VON_CLK 0.625V VONSINK MASTER OSCILLATOR 2MHz FB5 VIN 36,37 11 19 13 PGOOD VC1 – + BOOST S Q DRIVER SW1 34, 35 SENSE+ 1.1V UVLO + – 1.234V + 1.1V + – +gm CURRENT SENSE AMP VC2 – + 31 30 20 BIAS 18, 29 SW2 15, 16 R FOLDBACK OSCILLATOR 28 VIN SENSE– CT 2 BIAS R SLOPE COMP/ ONE-SHOT FB2 + + 9 VON_CLK FB2 1.25V 38 1.234V + – – – – 32 FB1 INTERNAL REGULATOR AND REFERENCE + – 22 RUN-SS1 – +gm 10 S Q DRIVER UVLO 3μA VC4 4 RUN-SS2 1.18V SW3 LOCKOUT BIAS + – 5 RUN-SS3/4 100k – + + – 3 + – +gm BIAS S SW2 LOCKOUT FB3 1.23V + + – GND 14,17,33 FOLDBACK OSCILLATOR 25 D4 VC3 BIAS 8 SW4 – + VON 24 12 SW3 6 R S 23 Q DRIVER – +gm NFB4 R FOLDBACK OSCILLATOR 2.7V 100k 21 Q DRIVER 1.1V E3 VON_CLK Figure 1 7 3513 F01 3513fa 10 LT3513 OPERATION The LT3513 is a highly integrated power supply IC containing four separate switching regulators and a low dropout linear regulator (LDO). Switching regulator 1 is a stepdown 2.5A regulator with inductor current sense and an integrated boost Schottky diode. Switching regulator 2 can be configured as a step-up or SEPIC converter and has a 1.2A switch. Switching regulator 3 consists of a step-up regulator with a 0.25A switch as well as an integrated Schottky diode. Switching regulator 4 is a negative regulator with a switch current limit of 0.25A and an integrated Schottky diode. Linear regulator 5 is capable of providing 8mA of current to the base of an external NPN transistor. The regulators share common circuitry including input source, voltage reference and master oscillator. Operation can be best understood by referring to the Block Diagram as shown in Figure 1. If the RUN-SS1 pin is pulled to ground, the LT3513 is shut down and draws 30μA from the input source tied to VIN. An internal 2μA current source charges the external softstart capacitor, generating a voltage ramp at this pin. If the RUN-SS1 pin exceeds 0.6V, the internal bias circuits turn on, including the internal regulator, reference and 2MHz master oscillator. The master oscillator generates four clock signals, one for each of the switching regulators. Switching regulator 1 will only begin to operate when the RUN-SS1 pin reaches 0.8V. Switcher 1 generates VLOGIC, which must be tied to the BIAS pin. When BIAS reaches 2.8V, the NPNs pulling down on the RUN-SS2 and RUN-SS3/4 pins turns off, allowing an internal 2μA current source to charge the external capacitors tied to RUN-SS2 and RUN-SS3/4 pins. When the voltage on RUN-SS2 reaches 0.8V, switcher 2 is enabled. Correspondingly, when the voltage on RUN-SS3/4 reaches 0.8V, switchers 3 and 4 are enabled. AVDD, E3 and VOFF will then begin rising at a rate determined by the capacitors tied to the RUN-SS2 and RUN-SS3/4 pins. When all four switching outputs reach 90% of their programmed voltages, the NPN pulling down on the CT pin will turn off, and an internal 20μA current source will charge the external capacitor tied to the CT pin. When the CT pin reaches 1.1V, the output disconnect PNP turns on, connecting VON to E3. In the event of any of the four outputs dropping below 10% of their programmed voltage, PanelProtect circuitry pulls the CT pin to GND, disabling VON. A power good comparator monitors AVDD and turns on when FB2 is at or above 90% of its regulated value. The output is an open-collector transistor that is off when the output is out of regulation, allowing an external resistor to pull the pin high. This pin can be used with a P-channel MOSFET that functions as an output disconnect for AVDD. The four switchers are current mode regulators. Instead of directly modulating the duty cycle of the power switch, the feedback loop controls the peak current in the switch during each cycle. Compared to voltage mode control, current mode control improves loop dynamics and provides cycle-by-cycle current limit. RUN-SS 2V/DIV VLOGIC 5V/DIV IL1 1A/DIV SS-234 2V/DIV AVDD 10V/DIV IL2 500μA/DIV PGOOD 20V/DIV 5ms/DIV 3513 F02a (2a) VSS3/4 2V/DIV VOFF 10V/DIV IL4 500mA/DIV VE3 20V/DIV IL3 500mA/DIV VCT 2V/DIV VON 20V/DIV 5ms/DIV 3513 F02b (2b) Figure 2. LT3513 Power-Up Sequence. (Traces from Both Photos are Synchnonized to the Same Trigger) 3513fa 11 LT3513 OPERATION All four switchers employ a constant-frequency current mode control scheme. Switcher 1, the step-down regulator, differs slightly from the others with inductor current sense. Instead of monitoring the current at the switch, current nodes are used to measure the current through the inductor. Inductor current sense does not suffer from minimum on-time problems, therefore always keeping the switch current limited with any input-to-output voltage ratio. Switcher 1 is always synchronized to the master oscillator. The other three switchers each have their own slave oscillator. The slave oscillator reduces the frequency when the feedback voltage dips below 0.75V and decreases linearly below the threshold as shown in the Performance Characteristics’ Frequency Foldback plot. Other than these two differences, the control loop is similar in all four switchers. A pulse from the master oscillator for switcher 1 or a pulse from the slave oscillator for the other three switchers sets the RS latch and turns on the internal NPN bipolar power switch. Current in the switch and the external inductor begins to increase. When this current exceeds a level determined by the voltage at VC, the current comparator resets the latch, turning off the switch. The current in the inductor flows through the Schottky diode and begins to decrease. The cycle begins again at the next pulse from the oscillator. In this way, the voltage on the VC pin controls the current through the inductor to the output. The internal error amplifier regulates the output by continually adjusting the VC pin voltage. The threshold for switching on the VC pin is 0.8V, and an active clamp of 1.8V limits the VC voltage. Switchers 2, 3 and 4 also contain an independent current limit not dependent on VC or duty cycle. Switcher 1’s current limit is controlled by the VC voltage and varies with duty cycle. All four switchers also use slope compensation to ensure stability with the current mode scheme at duty cycles above 50%. The RUN-SS1, RUN-SS2 and RUN-SS3/4 pins control the rate of rise of the feedback pins. The switch driver for SW1 operates either from VIN or from the BOOST pin. An external capacitor and an integrated Schottky diode are used to generate a voltage at the BOOST pin that is higher than the input supply. This allows the driver to saturate the internal bipolar NPN power switch for efficient operation. INPUT VOLTAGE RANGE STEP-DOWN CONSIDERATION The minimum operating voltage of switcher 1 is determined either by the LT3513’s undervoltage lockout of ~4V or by its maximum duty cycle. A user defined undervoltage lockout may be set with the UVLO pin at a voltage higher than the internal undervoltage lockout. The duty cycle is the fraction of time that the internal switch is on and is determined by the input and output voltages: DC = VOUT + VF VIN – VSW + VF where VF is the forward voltage drop of the catch diode (~0.4V) and VSW is the voltage drop of the internal switch (~0.3V at maximum load). This leads to a minimum input voltage of: VIN(MIN) = VOUT + VF DCMAX – VF + VSW with DCMAX = 0.75. The user defined undervoltage is set by a resistor divider connected to the UVLO pin. The comparator pulls 3μA from the pin when the UVLO pin is higher than 1.25V. The hysteresis and minimum input voltage equations are as follows: VHYS = (R2 + 2k ) • 3.9µA VIN(MIN) = 1.25V R1+ R2 R1 VIN R2 UVLO 38 R1 3513 A1 3513fa 12 LT3513 OPERATION INDUCTOR SELECTION AND MAXIMUM OUTPUT CURRENT inductance is required to avoid subharmonic oscillations, see Application Note 19. A good first choice for the inductor value is: The current in the inductor is a triangle wave with an average value equal to the load current. The peak switch current is equal to the output current plus half the peak-to-peak inductor ripple current. The LT3513 limits its switch current in order to protect itself and the system from overload faults. Therefore, the maximum output current that the LT3513 will deliver depends on the switch current limit, the inductor value, and the input and output voltages. When the switch is off, the potential across the inductor is the output voltage plus the catch diode drop. This gives the peak-to-peak ripple current in the inductor: L= VOUT + VF 1.8 where VF is the voltage drop of the catch diode (~0.4V) and L is in μH. The inductor’s RMS current rating must be greater than the maximum load current and its saturation current should be at least 30% higher. For highest efficiency, the series resistance (DCR) should be less than 0.1Ω. Table 1 lists several vendors and types that are suitable. Table 1. Inductor Vendors VENDOR Coilcraft Murata TDK URL www.coilcraft.com www.murata.com www.component.tdk.com Toko www.toko.com Sumida www.sumida.com PART SERIES MSS7341 LQH55D SLF7045 SLF10145 DC62CB D63CB D75C D75F CR54 CDRH74 CDRH6D38 CR75 TYPE Shielded Open Shielded Shielded Shielded Shielded Shielded Open Open Shielded Shielded Open The optimum inductor for a given application may differ from the one indicated by this simple design guide. A larger value inductor provides a higher maximum load current, and reduces the output voltage ripple. If your load is lower than the maximum load current, then you can relax the value of the inductor and operate with higher ripple current. This allows you to use a physically smaller inductor or one with a lower DCR resulting in higher efficiency. Be aware that the maximum load current depends on input voltage. A graph in the Typical Performance Characteristics section of this data sheet shows the maximum load current as a function of input voltage and inductor value for VOUT = 3.3V. In addition, low inductance may result in discontinuous mode operation, which further reduces maximum load current. For details of maximum output current and discontinuous mode operation, see Linear Technology’s Application Note 44. Finally, for duty cycles greater than 50% (VOUT/VIN > 0.5), a minimum ΔIL = (1– DC)( VOUT + VF ) L•f where f is the switching frequency of the LT3513 and L is the value of the inductor. The peak inductor and switch current is: ISW(PK ) = ILPK = IOUT + ΔIL 2 To maintain output regulation, this peak current must be less than the LT3513’s switch current limit of ILIM. For SW1, ILIM is at least 2A at DC = 0.35, and decreases linearly to 1.5A at DC = 0.75 as shown in the Typical Performance Characteristics section. The maximum output current is a function of the chosen inductor value: IOUT(MAX ) = ILIM – ΔIL ΔI = 2.5A • (1– 0.57 • DC) – L 2 2 Choosing an inductor value so that the ripple current is small will allow a maximum output current near the switch current limit. One approach to choosing the inductor is to start with the simple rule given above, look at the available inductors and choose one to meet cost or space goals. Then use these equations to check that the LT3513 will be able to deliver the required output current. Note again that these equations assume that the inductor current is continuous. Discontinuous operation occurs when IOUT is less than ΔIL/2. 3513fa 13 LT3513 OPERATION OUTPUT CAPACITOR SELECTION For 5V and 3.3V outputs, a 10μF 6.3V ceramic capacitor (X5R or X7R) at the output results in very low output voltage ripple and good transient response. Other types and values will also work; the following discussion explores tradeoffs in output ripple and transient performance. The output capacitor filters the inductor current to generate an output with low voltage ripple. It also stores energy in order satisfy transient loads and stabilizes the LT3513’s control loop. Because the LT3513 operates at a high frequency, minimal output capacitance is necessary. In addition, the control loop operates well with or without the presence of output capacitor series resistance (ESR). Ceramic capacitors, which achieve very low output ripple and small circuit size, are therefore an option. You can estimate output ripple with the following equations: VRIPPLE = ΔIL for ceramic capacitors, and 8 • f • COUT VRIPPLE = ΔIL • ESR for electrolytic capacitors (tantalum and aluminum) where ΔIL is the peak-to-peak ripple current in the inductor. The RMS content of this ripple is very low so the RMS current rating of the output capacitor is usually not of concern. It can be estimated with the formula: IC(RMS) = ΔIL 12 Another constraint on the output capacitor is that it must have greater energy storage than the inductor; if the stored energy in the inductor transfers to the output, the resulting voltage step should be small compared to the regulation voltage. For a 5% overshoot, this requirement indicates: ⎛ I ⎞ COUT > 10 • L • ⎜ LIM ⎟ ⎝ VOUT ⎠ 2 The low ESR and small size of ceramic capacitors make them the preferred type for LT3513 applications. However, not all ceramic capacitors are the same. Many of the higher value capacitors use poor dielectrics with high temperature and voltage coefficients. In particular, Y5V and Z5U types lose a large fraction of their capacitance with applied voltage and at temperature extremes. Because loop stability and transient response depend on the value of COUT, this loss may be unacceptable. Use X7R and X5R types. Electrolytic capacitors are also an option. The ESRs of most aluminum electrolytic capacitors are too large to deliver low output ripple. Tantalum and newer, lower ESR organic electrolytic capacitors intended for power supply use are suitable, and the manufacturers will specify the ESR. Chose a capacitor with a low enough ESR for the required output ripple. Because the volume of the capacitor determines its ESR, both the size and the value will be larger than a ceramic capacitor that would give similar ripple performance. One benefit is that the larger capacitance may give better transient response for large changes in load current. Table 2 lists several capacitor vendors. Table 2. Low ESR Surface Mount Capacitors VENDOR TYPE SERIES Taiyo Yuden Ceramic X5R, X7R AVX Ceramic Tantalum X5R, X7R TPS Kemet Tantalum Ta Organic Al Organic T491, T494, T495 T520 A700 Sanyo Ta or Al Organic POSCAP Panasonic Al Organic SP CAP TDK Ceramic X5R, X7R DIODE SELECTION The catch diode (D1 from Figure 1) conducts current only during switch off time. Average forward current in normal operation can be calculated from: ID( AVG) = IOUT VIN – VOUT VIN The only reason to consider a diode with a larger current rating than necessary for nominal operation is for the worst-case condition of shorted output. The diode current will then increase to the typical peak switch current. Peak reverse voltage is equal to the regulator input voltage. 3513fa 14 LT3513 OPERATION Use a diode with a reverse voltage rating greater than the input voltage. Table 3 lists several Schottky diodes and their manufacturers. R4 should be 10k or less to avoid bias current errors. Table 3. Schottky Diodes PART NUMBER ⎛V ⎞ R3 = R4 ⎜ OUT – 1⎟ ⎝ 1.25 ⎠ VR (V) IAVE (A) VFAT 1A (mV) VF at 2A (mV) MBRM120E 20 1 530 MBRM140 40 1 550 MBRS240 40 2 MBRA340 40 3 B120 20 1 B240 40 2 500 B340A 40 3 450 On Semiconductor Regulating Negative Output Voltages The LT3513 contains an inverting op amp with a gain of 1. The NFB4 pin works just as the other FB pins. Choose the resistors according to: 450 Diodes Inc. R6 = VOUT • R5 – R5 1.25 500 R5 should be 2.5kΩ or less to avoid bias current errors. –VOUT BOOST PIN CONSIDERATIONS The minimum operating voltage of an LT3513 application is limited by the undervoltage lockout ~4V and by the maximum duty cycle. The boost circuit also limits the minimum input voltage for proper start-up. If the input voltage ramps slowly or the LT3513 turns on when the output is already in regulation, the boost capacitor may not be fully charged. Because the boost capacitor charges with the energy stored in the inductor, the circuit will rely on some minimum load current to get the boost circuit running properly. This minimum load will depend on input and output voltages. The Typical Performance Characteristics section shows a plot of the minimum load current to start as a function of input voltage for a 3.3V output. The minimum load current generally goes to zero once the circuit has started. Even without an output load current, in many cases the discharged output capacitor will present a load to the switcher that will allow it to start. R6 NFB4 22 R5 3513 A2 Duty Cycle Range The maximum duty cycle (DC) of the LT3513 switching regulator is 75% for SW2, and 84% for SW3 and SW4. The duty cycle for a given application using the step-up or charge pump topology is: DC = Regulating Positive Output Voltages The output voltage is programmed with a resistor divider between the output and the FB pin. Choose the resistors according to: VOUT The duty cycle for a given application using the inverter or SEPIC is: DC = INVERTER/STEP-UP CONSIDERATIONS VOUT – VIN VOUT VIN + VOUT The LT3513 can still be used in applications where the duty cycle, as calculated above, is greater than the maximum. However, the part must be operated in discontinuous mode so that the actual duty cycle is reduced. 3513fa 15 LT3513 OPERATION Inductor Selection Table 1 lists several inductor vendors and types that are suitable to use with the LT3513. Consult each manufacturer for detailed information and for their entire selection of related parts. Use ferrite core inductors to obtain the best efficiency, as core losses at frequencies above 1MHz are much lower for ferrite cores than for powdered-iron units. A 10μH to 22μH inductor will be the best choice for most LT3513 step-up and charge pump designs. Choose an inductor that can carry the entire switch current without saturating. For inverting and SEPIC regulators, a coupled inductor, or two separate inductors is an option. When using coupled inductors, choose one that can handle at least the switch current without saturating. If using uncoupled inductors, each inductor need only handle approximately one-half of the total switch current. A 4.7μH to 15μH coupled inductor or two 10μH to 22μH uncoupled inductors will usually be the best choice for most LT3513 inverting and SEPIC designs. Output Capacitor Selection Use low ESR (equivalent series resistance) capacitors at the output to minimize the output ripple voltage. Multilayer ceramic capacitors are an excellent choice, as they have an extremely low ESR and are available in very small packages. X7R dielectrics are preferred, followed by X5R, as these materials retain their capacitance over wide voltage and temperature ranges. A 10μF to 22μF output capacitor is sufficient for most LT3513 applications. Even less capacitance is required for outputs with |VOUT| > 20V or |IOUT| < 100mA. Solid tantalum or OS-CON capacitors will also work, but they will occupy more board area and will have a higher ESR than a ceramic capacitor. Always use a capacitor with a sufficient voltage rating. Diode Selection A Schottky diode is recommended for use with the LT3513 switcher 2 and switcher 4. The Schottky diode for switcher 3 is integrated inside the LT3513. Choose diodes for switcher 2 and switcher 4 rated to handle an average current greater than the load current and rated to handle the maximum diode voltage. The average diode current in the step-up and SEPIC is equal to the load current. Each of the two diodes in the charge pump configurations carries an average diode current equal to the load current. The ground connected diode in the charge pump is integrated into the LT3513. The maximum diode voltage in the stepup and charge pump configurations is equal to |VOUT|. The maximum diode voltage in the SEPIC and inverting configurations is VIN + |VOUT|. Input Capacitor Selection Bypass the input of the LT3513 circuit with a 4.7μF or higher ceramic capacitor of X7R or X5R type. A lower value or a less expensive Y5V type will work if there is additional bypassing provided by bulk electrolytic capacitors or if the input source impedance is low. The following paragraphs describe the input capacitor considerations in more detail. Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input capacitor is required to reduce the resulting voltage ripple at the LT3513 input and to force this switching current into a tight local loop, minimizing EMI. The input capacitor must have low impedance at the switching frequency to do this effectively and it must have an adequate ripple current rating. The input capacitor RMS current can be calculated from the step-down output voltage and current, and the input voltage: CIN(RMS) = IOUT VOUT ( VIN – VOUT ) VIN < IOUT 2 and is largest when VIN = 2VOUT (50% duty cycle). The ripple current contribution from the other channels will be minimal. Considering that the maximum load current from switcher 1 is ~3A, RMS ripple current will always be less than 1.5A. The high frequency of the LT3513 reduces the energy storage requirements of the input capacitor, so that the capacitance required is less than 10μF. The combination of small size and low impedance (low equivalent series resistance or ESR) of ceramic capacitors makes 3513fa 16 LT3513 OPERATION them the preferred choice. The low ESR results in very low voltage ripple. Ceramic capacitors can handle larger magnitudes of ripple current than other capacitor types of the same value. Use X5R and X7R types. An alternative to a high value ceramic capacitor is a lower value along with a larger electrolytic capacitor, for example a 1μF ceramic capacitor in parallel with a low ESR tantalum capacitor. For the electrolytic capacitor, a value larger than 10μF will be required to meet the ESR and ripple current requirements. Because the input capacitor is likely to see high surge currents when the input source is applied, only consider a tantalum capacitor if it has the appropriate surge current rating. The manufacturer may also recommend operation below the rated voltage of the capacitor. Be sure to place the 1μF ceramic as close as possible to the VIN and GND pins on the IC for optimal noise immunity. A final caution is in order regarding the use of ceramic capacitors at the input. A ceramic input capacitor can combine with stray inductance to form a resonant tank circuit. If power is applied quickly (for example by plugging the circuit into a live power source), this tank can ring, doubling the input voltage and damaging the LT3513. The solution is to either clamp the input voltage or dampen the tank circuit by adding a lossy capacitor (an electrolytic) in parallel with the ceramic capacitor. For details, see Application Note 88. Soft-Start and Shutdown The RUN-SS1(Run/Soft-Start) pin is used to place the switching regulators and the internal bias circuits in shutdown mode. It also provides a soft-start function, along with RUN-SS2 and RUN-SS3/4. If the RUN-SS1 pin is pulled to ground, the LT3513 enters its shutdown mode with all regulators off and quiescent current reduced to ~30μA. An internal 2μA current source pulls up on the RUN-SS1, RUN-SS2, and RUN-SS3/4 pins. If the RUN-SS1 pin reaches ~0.6V, the internal bias circuits start and the quiescent currents increase to their nominal levels. If a capacitor is tied from the RUN-SS1, RUN-SS2 or RUN-SS3/4 pins to ground, then the internal pull-up current will generate a voltage ramp on these pins. This voltage clamps the VC pin, limiting the peak switch current and therefore input current during start-up. The RUN-SS1 pin clamps VC1, the RUN-SS2 pin clamps VC1 and the RUN-SS3/4 pin clamps the VC3 and VC4 pins. A good value for the soft-start capacitors is COUT/10,000, where COUT is the value of the largest output capacitor. VON Pin Considerations The VON pin is the delayed output for switching regulator 3. When the CT pin reaches 1.1V, the output disconnect PNP turns on, connecting VON to E3. The VON pin is current limited and will protect the LT3513 and input source from a shorted output. The VON pin output is also controlled from the VON_CLK pin. When VON_CLK is low, the VON output will turn on if the CT pin is greater than 1.1V. When VON_CLK is high, greater than 1.5V, the VON output is disabled and the VONSINK open collector device turns on. If the VONSINK pin is connected to VON through a resistor, the VON voltage will decay with a high VON_CLK. VON_CLK may be synced to the horizontal scanning frequency to improve LCD image quality. Low Voltage Dropout Linear Regulator The LT3513 features an output to drive an external NPN transistor LDO to provide a lower voltage logic supply voltage. The output is capable of providing 10mA of current to the base of the NPN. The output of the LDO is controlled by the FB5 pin. Choose the resistor values according to: ⎛ V ⎞ R8 = R7 ⎜ LDO – 1⎟ ⎝ 0.625V ⎠ R8 should be 10k or less to avoid bias current errors. The internal compensation of the LDO relies on a low ESR ceramic capacitor between the values of 2.2μF and 20μF. X7R dielectrics are preferred, followed by X5R, as these materials retain their capacitance over wide voltage and temperature ranges. 3513fa 17 LT3513 OPERATION Printed Circuit Board Layout For proper operation and minimum EMI, care must be taken during printed circuit board (PCB) layout. Figure 3 shows the high current paths in the step-down regulator circuit. Note that in the step-down regulators, large, switched currents flow in the power switch, the catch diode and the input capacitor. In the step-up regulators, large, switched currents flow through the power switch, the switching diode and the output capacitor. In SEPIC and inverting regulators, the switched currents flow through the power switch, the switching diode and the tank capacitor. The loop formed by the components in the switched current path should be as small as possible. Place these components, along with the inductor and output capacitor, on the same side of the circuit board, and connect them on that layer. Place a local, unbroken ground plane below these components, and tie this ground plane to system ground at one location, ideally at the ground terminal of VIN the output capacitor C2. Additionally, keep the SW and BOOST nodes as small as possible. Thermal Considerations The PCB must provide heat sinking to keep the LT3513 cool. The Exposed Pad on the bottom of the package must be soldered to a ground plane. This ground should be tied to other copper layers below with thermal vias; these layers will spread the heat dissipated by the LT3513. Place additional vias near the catch diodes. Adding more copper to the top and bottom layers and tying this copper to the internal planes with vias can reduce thermal resistance further. With these steps, the thermal resistance from die (or junction) to ambient can be reduced to θJA = 25°C or less. With 100LFPM airflow, this resistance can fall by another 25%. Further increases in airflow will lead to lower thermal resistance. SW GND (3a) VIN SW GND (3b) VSW VIN IC1 C1 L1 SW D1 GND C2 Figure 4. Topside PCB Layout (3c) 3519 F03 Figure 3. Subtracting the Current When the Switch is On (3a) from the Current When the Switch is Off (3b) Reveals the Path of the High Frequency Switching Current (3c) Keep this Loop Small. The Voltage on the SW and BOOST Nodes will Also be Switched; Keep These Nodes as Small as Possible. Finally, Make Sure The Circuit is Shielded with a Local Ground Plane 3513fa 18 LT3513 PACKAGE DESCRIPTION UHF Package 38-Lead Plastic QFN (5mm × 7mm) (Reference LTC DWG # 05-08-1701 Rev C) 0.70 p 0.05 5.50 p 0.05 5.15 ± 0.05 4.10 p 0.05 3.00 REF 3.15 ± 0.05 PACKAGE OUTLINE 0.25 p 0.05 0.50 BSC 5.5 REF 6.10 p 0.05 7.50 p 0.05 RECOMMENDED SOLDER PAD LAYOUT APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED 5.00 p 0.10 0.75 p 0.05 PIN 1 NOTCH R = 0.30 TYP OR 0.35 s 45o CHAMFER 3.00 REF 37 0.00 – 0.05 38 0.40 p0.10 PIN 1 TOP MARK (SEE NOTE 6) 1 2 5.15 ± 0.10 7.00 p 0.10 5.50 REF 3.15 ± 0.10 (UH) QFN REF C 1107 0.200 REF 0.25 p 0.05 0.50 BSC R = 0.125 TYP R = 0.10 TYP BOTTOM VIEW—EXPOSED PAD NOTE: 1. DRAWING CONFORMS TO JEDEC PACKAGE OUTLINE M0-220 VARIATION WHKD 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.20mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE 3513fa Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 19 LT3513 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT3003 3-Channel LED Ballaster with PWM Dimming VIN: 3V to 48V, IQ = 3,000:1 True Color PWM , ISD < 5μA, MSOP10 Package LT3465/LT3465A Constant Current, 1.2MHz/2.7MHz, High Efficiency White LED Boost Regulators with Integrated Schottky Diode VIN: 2.7V to 16V, VOUT(MAX) = 34V, IQ = 1.9mA, ISD < 1μA, ThinSOT Package LT3466/LT3466-1 Dual Constant Current, 2MHz High Efficiency White LED Boost Regulators with Integrated Schottky Diode VIN: 2.7V to 24V, VOUT(MAX) = 40V, IQ = 5mA, ISD < 16μA, 3mm × 3mm DFN10 Package LT3474 36V, 1A (ILED), 2MHz Step-Down LED Driver VIN: 4V to 36V, VOUT(MAX) = 13.5V, IQ = 400:1 True Color PWM, ISD < 16μA, TSSOP16E Package LT3475 Dual 1.5A (ILED), 36V, 2MHz Step-Down LED Driver VIN: 4V to 36V, VOUT(MAX) = 13.5V, IQ = 3,000:1 True Color PWM, ISD < 1μA, TSSOP20E Package LT3476 Quad Output 1.5A, 2MHz High Current LED Driver with 1,000:1 Dimming VIN: 2.8V to 16V, VOUT(MAX) = 36V, IQ = 1,000:1 True Color PWM, ISD < 10μA, 5mm × 7mm QFN10 Package LT3478/LT3478-1 42V, 4.5A (ISW), 2.25MHz, LED Drivers with 3,000:1 True Color PWM Dimming VIN: 2.8V to 36V, VOUT(MAX) = 42V, IQ = 6.1mA, ISD < 3μA, TSSOP16E Package LT3486 Dual 1.3A, 2MHz High Current LED Driver VIN: 2.5V to 24V, VOUT(MAX) = 36V, IQ = 1,000:1 True Color PWM, ISD < 1μA, 5mm × 3mm DFN, TSSOP16E Packages LT3491 Constant Current, 2.3MHz, High Efficiency White LED Boost Regulator with Integrated Schottky Diode VIN: 2.5V to 12V, VOUT(MAX) = 27V, IQ = 2.6mA, ISD < 8μA, 2mm × 2mm DFN6, SC70 Packages LT3494/LT3494A 40V, 180mA/350mA Micropower Low Noise Boost Converters with Output Disconnect VIN: 2.3V to 16V, VOUT(MAX) = 40V, IQ = 65μA, ISD < 1μA, 3mm × 2mm DFN8 Package LT3497 Dual 2.3MHz, Full Function LED Driver with Integrated Schottkys and 250:1 True Color PWM Dimming VIN: 2.5V to 10V, VOUT(MAX) = 32V, IQ = 6mA, ISD < 12μA, 3mm × 2mm DFN10 Package LT3498 2.3MHz, 20mA LED Driver and OLED Driver with Integrated Schottkys VIN: 2.5V to 12V, VOUT(MAX) = 32V, IQ = 1.65mA, ISD < 9μA, 3mm × 2mm DFN12 Package LT3591 Constant Current, 1MHz, High Efficiency White LED Boost VIN: 2.5V to 12V, VOUT(MAX) = 40V, IQ = 4mA, ISD < 9μA, 3mm × 2mm DFN8 Package Regulator with Integrated Schottky Diode and 80:1 True Color PWM Dimming LT3595 16-Channel 48V, 2MHz Buck Mode LED Driver with 3000:1 True Color PWM Dimming TM TM VIN: 4.5V to 50V, IQ = 3,000:1 True Color PWM, ISD < 3μA, 5mm × 9mm QFN56 Package True Color PWM and ThinSOT are trademarks of Linear Technology Corporation. 3513fa 20 Linear Technology Corporation LT 1108 REV A • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2008