19-0796; Rev 0; 4/07 Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response The MAX8664 dual-output PWM controller is a low-cost, high-performance solution for systems requiring dual power supplies. It provides two individual outputs that operate 180° out-of-phase to minimize input current ripple, and therefore, capacitance requirements. Built-in drivers are capable of driving external MOSFETs to deliver up to 25A output current from each channel. The MAX8664 operates from a 4.5V to 28V input voltage source and generates output voltages from 0.6V up to 90% of the input voltage on each channel. Total output regulation error is less than ±0.8% over load, line, and temperature. The MAX8664 operates with a constant switching frequency adjustable from 100kHz to 1MHz. Built-in boost diodes reduce external component count. Digital softstart eliminates input inrush current during startup. The second output has an optional external REFIN2, facilitating tracking supply applications. Each output is capable of sourcing and sinking current, making the device a great solution for DDR applications. The MAX8664 employs Maxim’s proprietary peak voltage-mode control architecture that provides superior transient response during either load or line transients. This architecture is easily stabilized using two resistors and one capacitor for any type of output capacitors. Fast transient response requires less output capacitance, consequently reducing total system cost. The MAX8664B latches off both controllers during a fault condition, while the MAX8664A allows one controller to continue to function when there is a fault in the other controller. Features o ±0.8% Output Accuracy Over Load and Line o Operates from a Single 4.5V to 28V Supply o Simple Compensation for Any Type of Output Capacitor o Internal 6.5V Regulator for Gate Drive o Integrated Boost Diodes o Adjustable Output from 0.6V to 0.9 x VIN o Digital Soft-Start Reduces Inrush Current o 100kHz to 1MHz Adjustable Switching o 180° Out-of-Phase Operation Reduces Input Ripple Current o Overcurrent and Overvoltage Protection o External Reference Input for Second Controller o Prebiased Startup Operation Ordering Information MAX8664AEEP+ PINPACKAGE 20 QSOP PKG CODE E20-1 FAULT ACTION Independent MAX8664BEEP+ 20 QSOP E20-1 Joint PART Note: This device operates over the -40°C to +85°C operating temperature range. +Denotes lead-free package. Typical Operating Circuit IN2 Applications Desktop and Notebook PCs VL IN ILIM2 ILIM1 Graphic Cards DH2 DH1 ASIC/CPU/DSP Power Supplies BST2 BST1 LX2 LX1 DL2 DL1 Set-Top Box Power Supply OUT2 IN1 OUT1 Printer Power Supply Network Power Supply POL Power Supply MAX8664 GND FB2 REFIN2 VCC PGND FB1 PWRGD OSC/EN12 Pin Configuration appears at end of data sheet. ________________________________________________________________ Maxim Integrated Products For pricing delivery, and ordering information please contact Maxim Direct at 1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com. 1 MAX8664 General Description MAX8664 Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response ABSOLUTE MAXIMUM RATINGS IN to GND ...........................................................…-0.3V to +30V VL to GND...................................................................-0.3 to +8V IN, BST_ to VL ........................................................-0.3V to +30V VCC, FB_, PWRGD to GND.......................................-0.3V to +6V VL to VCC ....................................................................-2V to +8V PGND to GND .......................................................-0.3V to +0.6V DL_ to PGND...............................................-0.3V to (VVL + 0.3V) DH_ to PGND............................................-0.3V to (VBST_+ 0.3V) BST_ to GND.............................................................-0.3V to 38V BST_ to LX ................................................................-0.3V to +8V LX_ to PGND .................-1V (-2.5V for < 50ns transient) to +30V DH_ to LX_................................................-0.3V to (VBST_+ 0.3V) Note 1: Package mounted on a multilayer PCB. ILIM_ to GND ...............................................-0.3V to (VIN + 0.3V) ILIM_ to LX_............................................................-0.6V to +30V OSC/EN12, REFIN2 to GND .....................-0.3V to (VVCC + 0.3V) VL Continuous Current ..............................................125mARMS VCC Continuous Current..............................................10mARMS Continuous Power Dissipation (TA = +70°C) (Note 1) 20-Pin QSOP (derate 11.0mW/°C above +70°C).........884mW Operating Temperature Range ...........................-40°C to +85°C Junction Temperature ......................................................+150°C Storage Temperature Range .............................-65°C to +150°C Lead Temperature (soldering, 10s) .................................+300°C Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. ELECTRICAL CHARACTERISTICS (VIN = 12V, ROSC/EN12 to GND = 56.1kΩ, REFIN2 = VCC, TA = -40°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.) (Note 2) PARAMETER CONDITIONS MIN TYP MAX UNITS SUPPLY VOLTAGES 7.2 28.0 IN = VL = VCC 4.5 5.5 VL Output Voltage 7.2V < VIN < 28V, 0 < IVL < 60mA 6.10 6.6 6.75 V VCC Output Voltage 7.2V < VIN < 28V, 0 < ICC < 5mA 4.5 5.0 5.5 V VCC Undervoltage Lockout (UVLO) Rising 3.4 3.5 3.6 IN Supply Voltage Hysteresis 350 V V mV Standby Supply Current OSC/EN12 not connected VIN = 12V, IIN 0.095 0.2 VCC = VIN = VVL = 5V, IIN + IVL + IVCC 0.08 0.2 Operating Supply Current No switching, VFB_ = 0.65V VIN = 12V, IIN 1.4 2.5 VCC = VIN = VVL = 5V, IIN + IVL+ IVCC 1.1 1.8 mA mA REGULATOR SPECIFICATIONS Reference Accuracy FB_ Regulation Accuracy TA = 0°C to +85°C 0.5955 0.600 0.6045 TA = -40°C to +85°C 0.5930 0.600 0.6070 VREFIN2 = VVCC TA = 0°C to +85°C 0.5952 0.600 0.6048 TA = -40°C to +85°C 0.5925 0.600 0.6075 VREFIN2 = 1.000V REFIN2 to Internal Reference Switchover Threshold Not to be switched during operation 0.995 1.000 1.005 2 VVCC 0.7 VVCC 0.3 V V V REFIN2 Maximum Program Voltage 1.3 V REFIN2 Disable Threshold 50 mV 3 nA FB Input Bias Current VFB = 0.5V REFIN2 Bias Current VREFIN2 = 0.65V 3 nA FB Propagation Delay FB rising to DH falling 90 ns 2 _______________________________________________________________________________________ Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response (VIN = 12V, ROSC/EN12 to GND = 56.1kΩ, REFIN2 = VCC, TA = -40°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.) (Note 2) PARAMETER CONDITIONS MIN TYP MAX UNITS PROTECTION FEATURES Overvoltage Protection (OVP) Threshold VFB1 rising 0.75 VREFIN2 = VVCC, VFB_ rising, MAX8664B Power-Good (PWRGD) Threshold VFB1 rising, MAX8664A 0.500 Hysteresis High-Side Current-Sense Program Current (Note 3) ILIM Leakage V REFIN2 + 0.15 VFB2 rising, VREFIN2 ≤ 1.3V 0.525 0.550 5 o TA = +85 C % 60 TA = +25oC 44 50 60 TA = +25°C 0.1 1.0 TA = +85°C 0.1 High-Side Current-Sense Overcurrent Trip Adjustment Range 0.05 0.40 Internal Soft-Start Time ROSC/EN12 = 56.1kΩ, 400kHz 2.5 REFIN2 Internal Pulldown Resistance Engaged momentarily at startup 10 Thermal-Shutdown Threshold Junction temperature V µA µA V ms 20 +160 Ω °C DRIVER SPECIFICATIONS DH_ Driver Resistance Sourcing current, IDH = -50mA Sinking current, IDH = 50mA DL_ Driver Resistance Dead Time for Low-Side to High-Side Transition VVL = 6.5V 1.35 VIN = VVL = VVCC = 5V 1.55 VVL = 6.5V 0.9 VIN = VVL = VVCC = 5V 1.0 Sourcing current, IDL = -50mA VVL = 6.5V 1.3 VIN = VVL = VVCC = 5V 1.5 Sinking current, IDL = 50mA VVL = 6.5V 0.6 VIN = VVL = VVCC = 5V 0.7 DL_ falling to DH_ rising VVL = 6.5V DH_ Minimum On-Time BST Current 13 VVL = 5V 25 2.1 1.4 2 1.1 43 28 70 Ω Ω ns 108 149 ns VBST - VLX = 7V, VLX = 28V, VFB_ = 0.55V 1.25 2.3 mA OSC/EN12 not connected 0.001 µA 6 Ω Internal Boost Switch Resistance PWM CLOCK OSCILLATOR PWM Clock-Frequency Accuracy PWM Clock-Frequency Adjustment Range ROSC/EN12 = 226kΩ to 22.6kΩ OSC/EN12 Disable Current -15 +15 % 100 1000 kHz 2.5 µA 1.5 Note 2: Specifications at -40°C are guaranteed by design and not production tested. Note 3: This current linearly compensates for the MOSFET temperature coefficient. _______________________________________________________________________________________ 3 MAX8664 ELECTRICAL CHARACTERISTICS (continued) Typical Operating Characteristics (Circuit of Figure 2, 600kHz, VIN = 12V, VOUT1 = 2.5V, VOUT2 = 1.8V, TA = +25°C, unless otherwise noted.) EFFICIENCY (%) 70 60 VOUT = 2.5V 40 30 VOUT1 = 2.5V 80 70 60 VOUT1 = 1.8V 50 40 30 VOUT = 1.8V 20 20 NO LOAD ON THE OTHER REGULATOR 10 0 0.1 10 1 0.1 2.53 2.52 2.50 2.49 2.50 2.49 2.48 IOUT2 = 0A 2.46 2.45 0 2 MAX8664 toc06 VOUT2 150 100mV/div 100 5A 2.5A 2.5A 50 2.46 0 2.45 6 8 10 12 14 16 20 18 100 INPUT VOLTAGE (V) 400 700 SWITCHING FREQUENCY (kHz) LOAD TRANSIENT -3A TO +3A TO -3A (FIGURE 3) 20μs/div 1000 POWER-UP WAVEFORMS MAX8664 toc08 MAX8664 toc07 10V/div 50mV/div VOUT1 VIN 2V/div VOUT2 50mV/div -3A VOUT1 2V/div VOUT2 +3A IOUT2 -3A 5A/div 5V/div VPRWGD 100μs/div 4 10 8 OUT1 LOAD TRANSIENT (FIGURE 2) IOUT2 2.47 6 ROSC/EN12 vs. SWITCHING FRQUENCY 2.48 NO LOAD 4 OUT1 LOAD CURRENT (A) 200 ROSC/EN12 (kΩ) 8A LOAD 2.51 IOUT2 = 4A 2.51 MAX8664 toc05 2.54 IOUT2 = 8A 2.52 10 1 250 MAX8664 toc04 2.55 2.53 LOAD CURRENT (A) LOAD CURRENT (A) LINE REGULATION (600kHz, FIGURE 2) 2.54 2.47 VIN = 3.3V VVL = 5V NO LOAD ON OUT2 10 0 2.55 MAX8664 toc02 80 90 OUT1 VOLTAGE (%) 90 EFFICIENCY (%) 100 MAX8664 toc01 100 50 LOAD REGULATION (600kHz, FIGURE 2) EFFICIENCY vs. LOAD CURRENT (1MHz, FIGURE 4) MAX8664 toc03 EFFICIENCY vs. LOAD CURRENT (600kHz, FIGURE 2) OUT1 VOLTAGE (%) MAX8664 Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response 1ms/div _______________________________________________________________________________________ 2A/div Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response MAX8664 Typical Operating Characteristics (continued) (Circuit of Figure 2, 600kHz, VIN = 12V, VOUT1 = 2.5V, VOUT2 = 1.8V, TA = +25°C, unless otherwise noted.) ENABLE WAVEFORMS (FIGURE 2) POWER-DOWN WAVEFORMS MAX8664 toc10 MAX8664 toc09 VIN 10V/div ENABLE 5V/div 2V/div VOUT1 2V/div 2V/div VOUT2 2V/div 5V/div VPRWGD 5V/div VOUT1 VOUT2 VPRWGD 1ms/div 1ms/div ENABLE WAVEFORMS (FIGURE 4) MAX8664 toc12 604 VLX1 10V/div 5A/div IL1 VOUT1 1V/div VOUT2 1V/div 10V/div VLX2 FEEDBACK VOLTAGE (mV) 5V/div 605 MAX8664 toc13 MAX8664 toc11 ENABLE FEEDBACK VOLTAGE vs. TEMPERATURE SWITCHING WAVEFORMS 603 602 601 600 599 598 597 VPRWGD 5V/div IL2 5A/div 596 NO LOAD 595 400μs/div 2μs/div -40 SHORT-CIRCUIT WAVEFORMS -20 0 20 60 40 TEMPERATURE (°C) 80 100 OVERVOLTAGE PROTECTION MAX8664 toc14 MAX8664 toc15 VOUT1 5V/div VOUT1 2V/div IIN IL1 10A/div 2A/div IL1 VDH1 5A/div VPRWGD 10V/div 10V/div VDL1 5V/div 10μs/div 20μs/div _______________________________________________________________________________________ 5 Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response MAX8664 Pin Description PIN NAME 1 DH1 High-Side MOSFET Driver Output for Controller 1. Connect DH1 to the gate of the high-side MOSFET. DH1 is low in shutdown and UVLO. 2 LX1 External Inductor Connection for Controller 1. Connect LX1 to the switching node of the MOSFETs and inductor. Make sure LX1 is close to the source of the high-side MOSFET(s) to form a Kelvin connection for high-side current sensing. LX1 is high impedance during monotonic startup and shutdown. 3 BST1 Boost Capacitor Connection for the High-Side MOSFET Driver for Controller 1. Connect a 0.22µF ceramic capacitor from BST1 to LX1. 4 DL1 Low-Side MOSFET Driver Output for Controller 1. Connect DL1 to the gate of the low-side MOSFET(s) for controller 1. DL1 is low in shutdown and UVLO. 5 VL Low-Side Gate Drive Supply and Output of the 6.5V Linear Regulator. Connect a 4.7µF ceramic capacitor from VL to PGND. When using a 4.5V to 5.5V supply, connect VL to IN. VL is the input to the VCC supply. Do not load VL when IC is disabled. 6 PGND Power Ground. Connect to the power ground plane. Connect power and analog grounds at a single point near the output capacitor’s ground. 7 DL2 Low-Side MOSFET Driver Output for Controller 2. Connect DL2 to the gate of the low-side MOSFET(s) for controller 2. DL2 is low in shutdown and UVLO. 8 BST2 Boost Capacitor Connection for the High-Side MOSFET Driver for Controller 2. Connect a 0.22µF ceramic capacitor from BST2 to LX2. 9 LX2 External Inductor Connection for Controller 2. Connect LX2 to the switching node of the MOSFETs and inductor. Make sure LX2 is close to the source of the high-side MOSFET(s) to form a Kelvin connection for high-side current sensing. LX2 is high impedance during monotonic startup and shutdown. 10 DH2 High-Side MOSFET Driver Output for Controller 2. Connect DH2 to the gate of the high-side MOSFET(s) for controller 2. DH2 is low in shutdown and UVLO. 11 ILIM2 Current-Limit Set for Controller 2. Connect a resistor from the drain of the high-side MOSFET(s) to ILIM2. See the Setting the Overcurrent Threshold section. 12 FB2 Feedback Input for Controller 2. Connect FB2 to the center of a resistor-divider connected between the output of controller 2 and GND to set the desired output voltage. VFB2 regulates to VREFIN2 or the internal 0.6V reference. To use the internal reference, connect REFIN2 to VCC. REFIN2 External Reference Input for Controller 2. To use the internal 0.6V reference, connect REFIN2 to VCC. To use an external reference, connect REFIN2 through a resistor (> 1kΩ) to a reference voltage between 0V and 1.3V. An RC lowpass filter is recommended when using an external reference and soft-start is not provided by the external reference. For tracking applications, connect REFIN2 to the center of a resistor voltage-divider between the output of controller 1 and GND (see Figure 3). Connect REFIN2 to GND to disable controller 2. 13 14 6 FUNCTION Switching Frequency Set Input. Connect a 22.6kΩ to 226kΩ resistor from OSC/EN12 to GND to set the switching frequency between 1000kHz and 100kHz. Connect a switch in series with this resistor for OSC/EN12 enable/shutdown control. When the switch is open, the IC enters low-power shutdown mode. In shutdown, OSC/EN12 is internally driven to approximately 800mV. 15 IN Internal 6.5V Linear Regulator Input. Connect IN to a 7.2V to 28V supply, and connect a 0.47µF or larger ceramic capacitor from IN to PGND. When using a 4.5V to 5.5V supply, connect IN to VL. 16 GND Analog Ground. Connect to the analog ground plane. Connect the analog and power ground planes at a single point near the output capacitor’s ground. _______________________________________________________________________________________ Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response PIN NAME FUNCTION VCC Internal Analog Supply. VCC regulates to 1.5V below VVL. Connect a 1µF ceramic capacitor from VCC to GND. When using a 4.5V to 5.5V supply, connect a 10Ω resistor from VCC to IN. VCC is used to power the IC’s internal circuitry. 18 PWRGD Open-Drain Power-Good Output. PWRGD is high impedance when controllers 1 and 2 (using the internal reference) are in regulation. PWRGD is low if the outputs are out of regulation, if there is a fault condition, or if the IC is shut down. PWRGD does not reflect the status of output 2 in the MAX8664A or when REFIN2 is connected to an external reference in the MAX8664B. 19 FB1 Feedback Input for Controller 1. Connect FB1 to the center of a resistor-divider connected between the output of controller 1 and GND to set the desired output voltage. VFB1 regulates to 0.6V. 20 ILIM1 17 Current-Limit Set for Controller 1. Connect a resistor from the drain of the high-side MOSFET(s) to ILIM1. See the Setting the Overcurrent Threshold section. _______________________________________________________________________________________ 7 MAX8664 Pin Description (continued) MAX8664 Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response VCC CURRENT-LIMIT COMPARATOR UVLO CIRCUITRY BIAS GENERATOR ILIM1 BST1 50μA BST CAP CHARGING SWITCH LX1 VOLTAGE REFERENCE THERMAL EN SHUTDOWN REF REF DH1 EN SOFT-START 1 SHUTDOWN CONTROL LOGIC SHUTDOWN 1 LX1 CONTROL LOGIC SHUTDOWN 2 DL1 CLOCK 1 PWM COMPARATOR 1 FB1 SHUTDOWN 1 PGND CURRENT-LIMIT COMPARATOR VL ILIM2 0.6V BST2 50μA BST CAP CHARGING SWITCH LX2 S2 PWM COMPARATOR 2 FB2 DH2 REF2 LX2 CONTROL LOGIC S1 ENABLE2 REFIN2 DL2 50mV SOFT-START IF VREFIN2 > 2.0V OPEN S1 AND CLOSE S2. OTHERWISE, CLOSE S1 AND OPEN S2. REF IN CLOCK 2 SHUTDOWN 2 CLOCK 1 OSC/EN12 OSCILLATOR CLOCK 2 THERMAL SHUTDOWN THERMAL SHUTDOWN FB1 ENABLE 4μA PWRGD REF1 - 0.1V VL 6.5V LDO FB2 1.5V REF2 - 0.1V VCC Figure 1. Functional Diagram 8 _______________________________________________________________________________________ GND Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response The MAX8664 dual-output PWM controller is a low-cost solution for dual power-supply systems. It provides two individual outputs that operate 180° out-of-phase to minimize input capacitance requirements. Built-in drivers are capable of driving external MOSFETs to deliver up to 25A of current from each output. The MAX8664 operates from a 4.5V to a 5.5V or a 7.2V to 28V input and generates output voltages from 0.6V up to 90% of the input voltage on each channel. Total output error is less than ±0.8% over load, line, and temperature. The MAX8664 operates with a constant switching frequency adjustable from 100kHz to 1MHz. Built-in boost diodes reduce external component count. Digital softstart eliminates input inrush current during startup. The second output has an optional REFIN2 input that takes an external reference voltage, facilitating tracking supply applications. Each output is capable of sourcing and sinking current. Internal 6.5V and 5V linear regulators provide power for gate drive and internal IC functions. The MAX8664 has built-in protection against output overvoltage, overcurrent, and thermal faults. The MAX8664B latches off both controllers during a fault condition, while the MAX8664A allows one controller to continue to function when there is a fault in the other controller. The MAX8664 employs Maxim’s proprietary peak-voltage mode control architecture that provides superior transient response during either load or line transients. This architecture is easily stabilized using two resistors and one capacitor for any type of output capacitors. Fast transient response requires less output capacitance, consequently reducing total system cost. DC-DC Controller Architecture The peak-voltage mode PWM control scheme ensures stable operation, simple compensation for any output capacitor, and fast transient response. An on-chip integrator removes any DC error due to the ripple voltage. This control scheme is simple: when the output voltage falls below the regulation threshold, the error comparator begins a switching cycle by turning on the high-side switch at the rising edge of the following clock cycle. This switch remains on until the minimum on-time expires and the output voltage is in regulation or the current-limit threshold is exceeded. At this point, the low-side synchronous rectifier turns on and remains on until the rising edge of the first clock cycle after the output voltage falls below the regulation threshold. Internal Linear Regulators The internal VL low-dropout linear regulator of the MAX8664A and MAX8664B provides the 6.5V supply used for the gate drive. Connect a 4.7µF ceramic capacitor from VL to PGND. When using a 4.5V to 5.5V input supply, connect VL directly to IN. The 5V supply used to power IC functions (VCC) is generated by an internal 1.5V shunt regulator from VL. Connect a 2.2µF ceramic capacitor from VCC to GND. When using a 4.5V to 5.5V input supply, connect VCC to IN through a 10Ω resistor. High-Side Gate-Drive Supply (BST_) The gate-drive voltage for the high-side MOSFETs is generated using a flying capacitor boost circuit. The capacitor between BST_ and LX_ is charged to the VL voltage through the integrated BST_ diode during the low-side MOSFET on-time. When the low-side MOSFET is switched off, the BST_ voltage is shifted above the LX_ voltage to provide the necessary turn-on voltage (VGS) for the high-side MOSFET. The controller closes a switch between BST_ and DH_ to turn the high-side MOSFET on. Voltage Reference An internal 0.6V reference sets the feedback regulation voltage. Controller 1 always uses the internal reference. An external reference input is provided for controller 2. To use the external reference, connect a 0 to 1.3V supply to REFIN2. This facilitates tracking applications. To use the internal 0.6V reference for controller 2, connect REFIN2 to VCC. Undervoltage Lockout (UVLO) When the VCC supply voltage drops below the UVLO threshold (3.15V falling typ), the undervoltage lockout (UVLO) circuitry inhibits the switching of both controllers, and forces the DL and DH gate drivers low. When VCC rises above the UVLO threshold (3.5V rising typ), the controllers begin the startup sequence and resume normal operation. Output Overcurrent Protection When the MAX8664 detects an overcurrent condition, DH is immediately pulled low. If the overcurrent condition persists for four consecutive cycles, the controller latches off and both DH_ and DL_ are pulled low. During softstart, when FB_ is less than 300mV, the controller latches off on the first overcurrent condition. The protection circuit detects an overcurrent condition by sensing the drain-source voltage across the high-side MOSFET(s). _______________________________________________________________________________________ 9 MAX8664 Detailed Description MAX8664 Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response The threshold that trips overcurrent protection is set by a resistor connected from ILIM_ to the drain of the highside MOSFET(s). ILIM_ sinks 50µA (typ) through this resistor. When the drain-source voltage exceeds the voltage drop across this resistor during the high-side MOSFET(s) on-time, an overcurrent fault is triggered. To prevent glitches from falsely tripping the overcurrent protection, connect a filter capacitor (0.01µF typically) in parallel with the overcurrent-setting resistor. Output Overvoltage Protection (OVP) During an overvoltage event on one or both of its outputs, the MAX8664 latches off the controller. This occurs when the feedback voltage exceeds its normal regulation voltage by 150mV for 10µs. In this state, the low-side MOSFET(s) are on and the high-side MOSFET(s) are off to discharge the output. To clear the latch, cycle EN or the input power. Thermal-Overload Protection Thermal-overload protection limits total power dissipation in the MAX8664. When the junction temperature exceeds +160°C, an internal thermal sensor shuts down the device, pulling DH_ and DL_ low for both controllers. To restart the controller, cycle EN or input power. Power-Good Output (PWRGD) PWRGD is an open-drain output that is pulled low when the output voltage rises above the PWRGD upper threshold or falls below the PWRGD falling threshold. PWRGD is held low in shutdown, when VCC is below the UVLO threshold, during soft-start, and during fault conditions. PWRGD does not reflect the status of controller 2 in the MAX8664A, or when REFIN2 is connected to an external reference with either version. See Table 1 for PWRGD operation of the circuits of Figures 2–5 during fault conditions. For logic-level output voltages, connect an external pullup resistor between PWRGD and the logic power supply. A 100kΩ resistor works well in most applications. Fault-Shutdown Modes When an overvoltage or overcurrent fault occurs on one controller of the MAX8664A, the second controller continues to operate. With the MAX8664B, a fault in one controller latches off both controllers automatically, and PWRGD is pulled low. See Table 1 for the fault-shutdown modes of the circuits shown in Figures 2–5. Table 1. Fault Shutdown Modes for Circuits of Figures 2–5 CIRCUIT MAX8664A (INDEPENDENT) CONTROLLER 1 FAULT CONTROLLER 2 FAULT MAX8664B (JOINT) CONTROLLER 1 FAULT CONTROLLER 2 FAULT Figure 2, Figure 5 (Independent) Controller 2 remains on. PWRGD is pulled low. Controller 1 remains on. PWRGD remains high. Controller 2 is shut down. PWRGD is pulled low. Controller 1 is shut down. PWRGD is pulled low. Figure 3 (Tracking) Controller 2 shuts down. PWRGD is pulled low. Controller 1 remains on. PWRGD remains high. Controller 2 is shut down. PWRGD is pulled low. Controller 1 is shut down. PWRGD is pulled low. Figure 4 (Sequenced) Controller 2 shuts down. PWRGD is pulled low. Controller 1 remains on. PWRGD remains high. Controller 2 is shut down. PWRGD is pulled low. Controller 1 is shut down. PWRGD is pulled low. 10 ______________________________________________________________________________________ Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response MAX8664 C19 0.01μF INPUT 10.8V TO 13.2V ILIM1 FB1 LX1 VCC C13 0.22μF REFIN2 BST1 N2 VL C14 4.7μF C5 1500pF R3 51.1kΩ DH1 C18 1μF C4 1000μF N1 IN C17 1μF C1 10μF C20 10μF R1 2.7kΩ L1 1μH R37 3Ω C6 47μF R4 3.92kΩ C7 47μF R5 1.15kΩ C8 47μF OUT1 2.5V/8A C23 0.1μF DL1 GND MAX8664 C25 680pF PGND C16 0.01μF VCC POWER-GOOD TO SYSTEM R9 10kΩ R2 3.01kΩ PWRGD OSC/EN12 ON OFF C3 10μF N3 DH2 R10 39.2kΩ ENABLE C21 10μF ILIM2 N9 2N7002 L2 1μH LX2 C15 0.22μF R6 51.1kΩ BST2 FB2 OUT2 1.8V/8A C12 1500μF C9 47μF C10 47μF C11 47μF C22 0.1μF DL2 N4 R38 3Ω C26 680pF R7 3.92kΩ C27 0.47μF R8 1.82kΩ Figure 2. Low-Cost, 600kHz Typical Application Circuit ______________________________________________________________________________________ 11 MAX8664 Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response Table 2. Component List for Figure 2 DESIGNATION QTY DESCRIPTION C25, C26 2 680pF, 50V C0G ceramic capacitors (0603) C27 1 0.47µF ±10%, 16V ceramic capacitor (0603) L1, L2 2 1µH inductors TOKO FDV0630-1R0M N1–N4 4 n-channel MOSFETs (8-pin SO) International Rectifier IRF7821 N9 1 n-channel MOSFET (SOT23) Central 2N7002 R1 1 2.74kΩ ±1% resistor (0603) R2 1 301kΩ ±1% resistor (0603) R3, R6 2 51.1kΩ ±1% resistors (0603) R4, R7 2 3.92kΩ ±1% resistors (0603) R5 1 1.15kΩ ±1% resistor (0603) 1µF ±20%, 16V X5R ceramic capacitor (0603) R8 1 1.82kΩ ±1% resistor (0603) R9 1 10kΩ ±5% resistor (0603) 1 1µF ±20%, 6.3V X5R ceramic capacitor (0603) R10 1 39.2kΩ ±1% resistor (0603) 2 0.1µF ±20%, 16V X7R ceramic capacitors (0603) DESIGNATION QTY C1, C3, C20, C21 4 10µF ±20%, 16V X5R ceramic capacitors (1206) C4 1 1000µF ±20%, 16V electrolytic capacitor (8mm diameter, 20mm height) 2 1500pF, 50V C0G ceramic capacitors (0603) C6–C11 6 47µF ±20%, 6.3V X5R ceramic capacitors (1206) C13, C15 2 0.22µF ±10%, 25V X7R ceramic capacitors (0603) 1 4.7µF ±10%, 6.3V X5R ceramic capacitor (0805) 2 0.01µF ±10%, 50V X7R ceramic capacitors (0603) C5, C12 C14 C16, C19 C17 C18 C22, C23 12 1 DESCRIPTION R37, R38 2 3Ω ±5% resistors (0805) U1 1 MAX8664 (20-pin QSOP) ______________________________________________________________________________________ Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response MAX8664 C1 0.01μF VCC LX1 GND MAX8664 N3 DL1 N4 R4 3.57kΩ OUT1 1.8V/20A C9 470μF R5 3Ω BST1 C13 4.7μF R3 10kΩ C8 0.015μF L1 0.56μH C7 0.22μF REFIN2 VL C12 1000pF N2 R2 24.3kΩ C6 1μF R7 1kΩ N1 DH1 IN C5 1μF OUT1 R6 1kΩ C4 1000μF ILIM1 FB1 INPUT 10V TO 14V C3 10μF C2 10μF R1 3.16kΩ C10 470μF C11 10μF C14 2200pF PGND VCC C15 0.01μF R9 10kΩ POWER-GOOD TO SYSTEM PWRGD R10 44.2kΩ R8 2.74kΩ C16 10μF C17 10μF ILIM2 N5 DH2 OSC/EN12 ENABLE ON L2 0.47μH LX2 N7 2N7002 C18 0.22μF OFF R12 2Ω BST2 FB2 DL2 N6 C23 2200pF R11 14.7kΩ OUT2 0.9V/6A C19 4700pF C20 680μF C21 680μF C22 10μF R13 10kΩ Figure 3. 500kHz Tracking Circuit for DDR2 Applications ______________________________________________________________________________________ 13 MAX8664 Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response Table 3. Component List for Figure 3 DESIGNATION QTY L2 1 0.47µH, 1.2mΩ inductor TOKO FDV0603-R47M 10µF, 16V X5R ceramic capacitors N1, N2 2 1 1000µF/16V aluminum electrolytic capacitor Rubycon 16MBZ1000M n-channel MOSFETs IRLR7821 (D-Pak) N3, N4 2 n-channel MOSFETs IRLR3907Z (D-Pak) C5 1 1µF, 16V X5R ceramic capacitor N5 1 n-channel MOSFET IRF7807Z (8-pin SO) C6 1 1µF, 10V X5R ceramic capacitor 1 2 0.22µF, 10V X7R ceramic capacitors N6 C7, C18 n-channel MOSFET IRF7821 (8-pin SO) N7 1 1 0.015µF, 10V X7R ceramic capacitor n-channel MOSFET 2N7002 (SOT23) R1 1 3.16kΩ ±1% resistor (0402 or 0603) 2 470µF, 2.5V POS capacitors Sanyo 2R5TPD470M6 DESIGNATION QTY C1, C15 2 0.01µF, 10V X7R ceramic capacitors C2, C3, C16, C17 4 C4 C8 C9, C10 C11, C22 C12 2 10µF, 6.3V X5R ceramic capacitors 1 1000pF, 10V X7R ceramic capacitor DESCRIPTION R2 1 24.3kΩ ±1% resistor (0402 or 0603) R3, R13 2 10kΩ ±1% resistors (0402 or 0603) R4 1 3.57kΩ ±5% resistor (0402 or 0603) 3.0Ω ±5% resistor (0603) R5 1 R6, R7 2 1kΩ ±1% resistors (0402 or 0603) R8 1 2.74kΩ ±1% resistor (0402 or 0603) C13 1 4.7µF, 10V X5R ceramic capacitor C14, C23 2 2200pF, 25V X7R capacitors R9 1 10kΩ ±5% resistor (0402 or 0603) C19 1 4700pF, 10V X7R capacitor R10 1 44.2kΩ ±1% resistor (0402 or 0603) 2 680µF, 2.5V POS capacitors Sanyo 2R5TPD680M6 R11 1 14.7kΩ ±1% resistor (0402 or 0603) R12 1 2.0Ω ±5% resistor (0402 or 0603) 1 0.56µH, 4.6mΩ inductor Panasonic ETQP4LR56WFL C20, C21 L1 14 DESCRIPTION ______________________________________________________________________________________ Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response MAX8664 C1 0.01μF ILIM1 FB1 5V IN C5 1μF 0.6V EXT REF VCC R7 10kΩ MAX8664 REFIN2 C14 0.01μF R8 10kΩ POWER-GOOD TO SYSTEM Q1 CMST3904 N2 IRF7821 DL1 C9 47μF R6 2Ω C15 0.01μF GND R11 3.32kΩ R10 10kΩ C10 47μF OUT1 1.8V/10A C11 0.1μF INPUT 2.97V TO 3.63V ILIM2 VCC R4 3.16kΩ C13 2200pF PGND N5 2N7002 R9 47kΩ C7 0.22μF BST1 R3 10kΩ L1 0.2μH LX1 VCC C12 1μF C8 820pF R2 17.4kΩ DH1 R5 10Ω C4 10μF N1 IRF7821 VL C6 4.7μF C3 10μF C2 1μF R1 3.32kΩ C16 1μF C17 10μF C18 10μF DH2 PWRGD R12 22.6kΩ OSC/EN12 FB2 N3 IRF7821 LX2 L2 0.2μH BST2 DL2 C19 0.22μF R13 2Ω N4 IRF7821 C20 2200pF R14 17.4kΩ C21 820pF OUT2 1.2V/10A C22 47μF C23 47μF C24 0.1μF R15 10kΩ R16 6.34kΩ Figure 4. 1MHz Application Circuit with All Ceramic Capacitors and Sequenced Outputs ______________________________________________________________________________________ 15 MAX8664 Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response Table 4. Component List for Figure 4 DESIGNATION QTY L1, L2 2 0.2µH, 2.4mΩ inductors TOKO FDV0603-R20M N1–N4 4 n-channel MOSFETs IRF7821 (8-pin SO) N5 1 n-channel MOSFET 2N7002 (SOT23) Q1 1 Transistor, bipolar, npn Central CMST3904 R1, R11 2 3.32kΩ ±1% resistors (0402 or 0603) 820pF,10V X7R ceramic capacitors R2, R14 2 17.4kΩ ±1% resistors (0402 or 0603) R3, R15 2 10kΩ ±1% resistors (0402 or 0603) 4 47µF, 6.3V X5R ceramic capacitors R4 1 3.16kΩ ±1% resistor (0402 or 0603) R5 1 10.0Ω ±5% resistor (0402 or 0603) 2 0.1µF, 10V X7R ceramic capacitors R6, R13 2 2.0Ω ±5% resistors (0603) R7, R8, R10 3 10kΩ ±5% resistors (0402 or 0603) DESIGNATION QTY C1, C14, C15 2 0.01µF, 10V X7R ceramic capacitors C2, C16 2 1µF, 6.3V X5R ceramic capacitors C3, C4, C17, C18 4 10µF, 6.3V X5R ceramic capacitors C5, C12 2 1µF, 10V X5R ceramic capacitors C6 1 4.7µF, 10V X5R ceramic capacitor C7, C19 2 0.22µF, 10V X7R ceramic capacitors C8, C21 2 C9, C10, C22, C23 C11, C24 C13, C20 16 2 DESCRIPTION 2200pF, 25V X7R ceramic capacitors DESCRIPTION R9 1 47kΩ ±5% resistor (0402 or 0603) R12 1 22.6kΩ ±1% resistor (0402 or 0603) R16 1 6.34kΩ ±1% resistor (0402 or 0603) ______________________________________________________________________________________ Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response MAX8664 C1 0.01μF C2 10μF R1 2.87kΩ C3 10μF C4 OPEN ILIM1 FB1 N1 INPUT 7.2V TO 20V C5 1μF LX1 C6 1μF R3 10kΩ R5 2Ω DL1 REFIN2 VL C12 4.7μF OUT1 1.5V/10A C9 470μF N2 MAX8664 R4 5.36kΩ L1 1.43μH C7 0.22μF BST1 VCC C8 4700pF R2 40.2kΩ DH1 IN C10 10μF C11 1000pF PGND C13 0.01μF GND C14 10μF ILIM2 R6 2.26kΩ VCC C15 10μF N3 DH2 POWER-GOOD TO SYSTEM R7 10kΩ C16 0.22μF PWRGD BST2 R8 75kΩ N4 OSC/EN12 ENABLE N5 2N7002 L2 1.43μH LX2 DL2 FB2 R10 2Ω C20 1000pF C17 R9 25.5kΩ 4700pF OUT2 1.05V/8A C18 470μF C19 10μF R11 10kΩ R12 9.53kΩ Figure 5. 300kHz Circuit with 7.2V to 20V Input ______________________________________________________________________________________ 17 MAX8664 Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response Table 5. Component List for Figure 5 DESIGNATION QTY DESCRIPTION C1, C13 2 0.01µF, 10V X7R ceramic capacitors C2, C3, C14, C15 4 10µF, 25V X5R ceramic capacitors C5 1 1µF, 25V X5R ceramic capacitor C6 1 C7, C16 C8, C17 DESIGNATION QTY DESCRIPTION L1, L2 2 1.43µH, 4.52mΩ inductors Panasonic ETQP3H1E4BFA N1–N4 4 n-channel MOSFETs IRF7821 (8-pin SOs) 1µF, 10V X5R ceramic capacitor N5 1 2 0.22µF, 10V X7R ceramic capacitors n-channel MOSFET 2N7002 (SOT23) R1 1 2.87kΩ ±1% resistor (0402 or 0603) 4700pF, 10V X7R ceramic capacitors R2 1 40.2kΩ ±1% resistor (0402 or 0603) 2 R3, R11 2 10kΩ ±1% resistors (0402 or 0603) C9, C18 2 470µF/2.5V POSCAP capacitors Sanyo 2R5TPD470M6 R4 1 5.36kΩ ±1% resistor (0402 or 0603) R5, R10 2 2.0Ω ±5% resistors (1206) C10, C19 2 10µF, 6.3V X5R ceramic capacitors R6 1 2.26kΩ ±1% resistor (0402 or 0603) C11, C20 2 1000pF, 25V X7R ceramic capacitors R7 1 10kΩ ±5% resistor (0402 or 0603) C12 1 4.7µF, 10V X5R ceramic capacitor R8 1 75kΩ ±1% resistor (0402 or 0603) R9 1 25.5kΩ ±1% resistor (0402 or 0603) R12 1 9.53kΩ ±1% resistor (0402 or 0603) Power-Up and Sequencing The MAX8664 features an OSC/EN12 input that is used both for setting the switching frequency and as an enable input for both controllers. A resistor from OSC/EN12 to GND sets the switching frequency, and when OSC/EN12 is high impedance, both controllers enter low-power shutdown mode. This is easily achieved with a transistor between the resistor and GND. Figure 6a shows the startup configuration with independent outputs. With REFIN2 connected to VCC, both controllers use the internal reference. For tracking applications, connect REFIN2 to the center of a resistive voltage-divider between the output of controller 1 and GND. See Figure 6b. In this application, the output of regulator 2 tracks the output voltage of controller 1. The voltage-divider resistors set the VOUT2/VOUT1 ratio. A typical tracking application is for the VTT supply of DDR memory. Figure 6c shows one method of sequencing the outputs. Output 1 rises first. When PWRGD goes high, the transistors allow the external reference to drive REFIN2 and output 2 rises. The circuit in Figure 6d functions similarly, except the enable signal is supplied externally instead of being driven by the PWRGD signal. CHIP ENABLE VCC VOUT1 REFIN2 VOUT2 MAX8664 ON PWRGD OSC/EN12 OFF CHIP ENABLE Figure 6a. Two Independent Output Startup and Shutdown Waveforms 18 ______________________________________________________________________________________ Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response MAX8664 VOUT1 CHIP ENABLE REFIN2 VOUT1 MAX8664 VOUT2 ON OSC/EN12 OFF PWRGD CHIP ENABLE Figure 6b. Ratiometric Tracking Startup and Shutdown Waveforms VCC CHIP ENABLE EXTERNAL REF PWRGD REFIN2 VOUT1 MAX8664 VOUT2 ON PWRGD OSC/EN12 OFF CHIP ENABLE Figure 6c. Sequencing Startup and Shutdown Waveforms ______________________________________________________________________________________ 19 MAX8664 Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response VCC EXTERNAL REF CHIP ENABLE OUT2 ENABLE REFIN2 ON OFF VOUT1 OUT2 ENABLE MAX8664 VOUT2 OSC/EN12 ON PWRGD OFF CHIP ENABLE Figure 6d. Sequencing Startup and Shutdown Waveforms with System Enable 2 Signal Design Procedure Setting the Switching Frequency Connect a resistor from OSC/EN12 to GND to set the switching frequency between 100kHz and 1000kHz. Calculate the resistor value (R10 in Figures 2–5) as follows: R10 = 2.24 × 1010 (Hz) (Ω) fS Inductor Selection There are several parameters that must be examined when determining which inductor is to be used. Input voltage, output voltage, load current, switching frequency, and LIR. LIR is the ratio of inductor-current ripple to maximum DC load current (ILOAD(MAX)). A higher LIR value allows for a smaller inductor, but results in higher losses and higher output ripple. A good compromise between size and efficiency is an LIR of 0.3. Once all the parameters are chosen, the inductor value is determined as follows: L= VOUT × (VIN − VOUT ) VIN × fS × ILOAD(MAX) × LIR where fS is the switching frequency. Choose a standard value inductor close to the calculated value. The exact 20 inductor value is not critical and can be adjusted to make trade-offs among size, cost, and efficiency. Lower inductor values minimize size and cost, but they also increase the output ripple and reduce the efficiency due to higher peak currents. On the other hand, higher inductor values increase efficiency, but eventually resistive losses due to extra turns of wire exceed the benefit gained from lower AC current levels. This is especially true if the inductance is increased without also increasing the physical size of the inductor. Find a low-loss inductor having the lowest possible DC resistance that fits the allotted dimensions. The chosen inductor’s saturation current rating must exceed the peak inductor current determined as: IPEAK = ILOAD(MAX) + LIR × ILOAD(MAX) 2 Output Capacitor The key selection parameters for the output capacitor are the actual capacitance value, the equivalent series resistance (ESR), the equivalent series inductance (ESL), and the voltage-rating requirements. These parameters affect the overall stability, output voltage ripple, and transient response. The output ripple has three components: variations in the charge stored in the output capacitor, the voltage drop across the capacitor’s ESR, and ESL caused by the current into and out of the capacitor. The maximum output voltage ripple is estimated as follows: ______________________________________________________________________________________ Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response VRIPPLE(ESR) = IP−P × ESR VRIPPLE(ESL) = VRIPPLE(C) = VIN × ESL L + ESL IP−P 8 × COUT × fS output voltage instantly changes by ESR x ΔILOAD. Before the controller can respond, the output voltage deviates further depending on the inductor and output capacitor values. After a short period of time (see the Typical Operating Characteristics ), the controller responds by regulating the output voltage back to its nominal state. The controller response time depends on its closed-loop bandwidth. With a higher bandwidth, the response time is faster, thus preventing the output voltage from further deviation from its regulating value. Setting the Output Voltages and Voltage Positioning where IP-P is the peak-to-peak inductor current: V −V V IP−P = IN OUT × OUT fS × L VIN These equations are suitable for initial capacitor selection, but final values should be chosen based on a prototype or evaluation circuit. As a general rule, a smaller ripple current results in less output-voltage ripple. Since the inductor ripple current is a factor of the inductor value and input voltage, the output-voltage ripple decreases with larger inductance, and increases with higher input voltages. Ceramic, tantalum, or aluminum polymer electrolytic capacitors are recommended. The aluminum electrolytic capacitor is the least expensive; however, it has higher ESR and ESL. To compensate for this, use a ceramic capacitor in parallel to reduce the switching ripple and noise. For reliable and safe operation, ensure that the capacitor’s voltage and ripple-current ratings exceed the calculated values. Figure 7 shows the feedback network used on the MAX8664. With this configuration, a portion of the feedback signal is sensed on the switched side of the inductor (LX), and the output voltage droops slightly as the load current is increased due to the DC resistance of the inductor (DCR). This allows the load regulation to be set to match the voltage droop during a load transient (voltage positioning), reducing the peak-to-peak output voltage deviation during a load transient, and reducing the output capacitance requirements. To set the magnitude of the voltage positioning, select a value for R2 in the 8kΩ to 24kΩ range, then calculate the value of R1 as follows: ⎛ IOUT(MAX) × DCR ⎞ R1 = R2 × ⎜ − 1⎟ ⎝ ΔVOUT(MAX) ⎠ where IOUT(MAX) is the maximum output current and Δ VOUT(MAX) is the maximum allowable droop in the output voltage at full load. The response to a load transient depends on the selected output capacitors. After a load transient, the L DCR LX_ OUT ESR RLOAD R1 Cr R2 COUT FB_ R3 Figure 7. Feedback Network ______________________________________________________________________________________ 21 MAX8664 VRIPPLE = VRIPPLE(ESR) + VRIPPLE(C) + VRIPPLE(ESL) The output voltage ripple as a consequence of the ESR, ESL, and output capacitance is: MAX8664 Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response To set the no-load output voltage (VOUT), calculate the value of R3 as follows: Finally, calculate the value of Cr as follows: VOUT (VIN − VOUT ) VIN Cr = R1× fS × | (VFB _ RIPPLE − VOUT _ RIPPLE ) | ⎛ ⎞ ⎛ R1 × R2 ⎞ VFB R3 = ⎜ ⎟ ⎟⎜ ⎝ VOUT − VFB ⎠ ⎝ R1 + R2 ⎠ where VFB is the feedback regulation voltage (0.6V when using the internal reference or VREFIN2 for external reference). If the desired output voltage is equal to the reference voltage (typical for tracking applications), R3 is not installed. To achieve the lowest possible load regulation in applications where voltage positioning is not desired, R1 is not installed and R3 is calculated as follows: ⎛ ⎞ VFB R3 = ⎜ ⎟ × R2 ⎝ VOUT − VFB ⎠ Compensation To ensure stable operation, connect a compensation capacitor (Cr) across the upper feedback resistor as shown in Figure 7. To find the value of this capacitor, follow the compensation design procedure below. Choose a closed-loop bandwidth (fC) that is less than 1/3 the switching frequency (fS). Calculate the output double pole (fO) as follows: fO = 1 R + ESR 2π L × COUT × LOAD RLOAD + DCR The FB peak-to-peak voltage ripple is: R2 ⎞ ⎛ ⎛ 1+ ⎜ VOUT ⎜ R1 ⎟ × ⎜ VFB _ RIPPLE = ⎜ R2 R2 ⎟ ⎜ ⎛ DCR ⎞ fC + ⎜ 1+ ⎟ ⎝ R3 R1 ⎠ ⎜⎝ ⎜⎝1+ RLOAD ⎟⎠ × fO ⎞ ⎟ ⎟ ⎟ ⎟ ⎠ The output ripple voltage due to the ESR of the output capacitor, COUT, is: VOUT (VIN − VOUT ) V VOUT _ RIPPLE = IN × L × fS ⎛ 1 ⎞ ⎜ ESR + ⎟ ⎝ 8 × CO × fS ⎠ Target the feedback ripple in the 25mV to 60mV range. For high duty-cycle applications (> 70%), a feedback ripple of 25mV is recommended. 22 MOSFET Selection Each output of the MAX8664 is capable of driving two to four external, logic-level, n-channel MOSFETs as the circuit switch elements. The key selection parameters are: • On-resistance (RDS(ON))—the lower, the better. • Maximum Drain-to-Source Voltage (VDSS)—should be at least 20% higher than the input supply rail at the high-side MOSFET’s drain. • Gate charges (Qg, Qgd, Qgs)— the lower, the better. For a 5V input application, choose MOSFETs with rated RDS(ON) at VGS ≤ 4.5V. With higher input voltages, the internal VL regulator provides 6.5V for gate drive in order to minimize the on-resistance for a wide range of MOSFETs. For a good compromise between efficiency and cost, choose the high-side MOSFETs that have conduction losses equal to switching losses at nominal input voltage and output current. Low RDS(ON) is preferred for lowside MOSFETs. Make sure that the low-side MOSFET(s) does not spuriously turn on due to dV/dt caused by the high-side MOSFET(s) turning on, as this would result in shoot-through current and degrade the efficiency. MOSFETs with a lower Q gd / Q gs ratio have higher immunity to dV/dt. For high-current applications, it is often preferable to parallel two MOSFETs rather than to use a single large MOSFET. For proper thermal management, the power dissipation must be calculated at the desired maximum operating junction temperature, maximum output current, and worst-case input voltage. For the-low side MOSFET(s), the worst-case power dissipation occurs at the highest duty cycle (VIN(MAX)). The low-side MOSFET(s) operate as zero voltage switches; therefore, major losses are the channel conduction loss (P LSCC) and the body diode conduction loss (PLSDC): ⎛ VOUT ⎞ 2 PLSCC(MAX) = ⎜1 − ⎟ × I LOAD(MAX) × RDS(ON) V ⎝ IN(MAX) ⎠ Use RDS(ON) at TJ(MAX): PLSDC(MAX) = 2 x ILOAD(MAX) VF x tDT x fS where VF is the body diode forward-voltage drop, tDT is the dead time between high-side and low-side switching transitions (25ns typical), and fS is the switching frequency. ______________________________________________________________________________________ Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response PHSCC(MAX) = VOUT × I2LOAD(MAX) × RDS(ON) VIN(MIN) Use RDS(ON) at TJ(MAX): PHSSW(MAX) = VIN(MAX) × ILOAD(MAX) × QGD × fS IGATE where IGATE is the average DH driver output-current capability determined by: IGATE ≅ 0.5 × VVL RDS(ON)(DR) + RGATE where RDS(ON)(DR) is the DH_ driver’s on-resistance (see the Electrical Characteristics) and RGATE is the internal gate resistance of the MOSFET (~ 2Ω): PHSDR = QG × VGS × fS × RGATE RGATE + RDS(ON)(DR) where VGS ≈ VVL. The high-side MOSFET(s) do not have body diode conduction loss, unless the converter is sinking current. When sinking current, calculate this loss as PHSDC(MAX) = ILOAD(MAX) x VF x (2 x tDT + tWD) x fS, where tWD is about 130ns. Allow an additional 20% for losses due to MOSFET output capacitances and low-side MOSFET body diode reverse-recovery charge dissipated in the high-side MOSFET(s). Refer to the MOSFET data sheet for thermal resistance specifications to calculate the PCB area needed to maintain the desired maximum operating junction temperature with the above calculated power dissipations. MOSFET Snubber Circuit Fast switching transitions cause ringing because of resonating circuit parasitic inductance and capacitance at the switching nodes. This high-frequency ringing occurs at LX’s rising and falling transitions and can interfere with circuit performance and generate EMI. To dampen this ringing, a series RC snubber circuit is added across each low-side switch. Below is the procedure for selecting the value of the series RC circuit. Connect a scope probe to measure VLX_ to GND and observe the ringing frequency, fR. Find the capacitor value (connected from LX_ to GND) that reduces the ringing frequency by half. The circuit parasitic capacitance (CPAR) at LX_ is then equal to 1/3 the value of the added capacitance above. The circuit parasitic inductance (LPAR) is calculated by: LPAR = 1 (2πfR ) 2 × CPAR The resistor for critical dampening (RSNUB) is equal to 2π x fR x LPAR. Adjust the resistor value up or down to tailor the desired damping and the peak-voltage excursion. The capacitor (CSNUB) should be at least 2 to 4 times the value of the CPAR to be effective. The power loss of the snubber circuit is dissipated in the resistor (PRSNUB) and can be calculated as: PRSNUB = CSNUB × (VIN ) × fSW 2 where VIN is the input voltage and fSW is the switching frequency. Choose an RSNUB power rating that meets the specific application’s derating rule for the power dissipation calculated. Setting the Overcurrent Threshold Connect a resistor from ILIM_ to the drain of the highside MOSFET(s) to set the overcurrent protection threshold. ILIM_ sinks 50µA (typ) through this resistor. When the drain-source voltage exceeds the voltage drop across this resistor during the high-side MOSFET(s) on-time, overcurrent protection is triggered. To set the output current level where overcurrent protection is triggered (ILIMIT), calculate the value of the ILIM_ resistor as follows: R ILIM _ = RDS(ON)HS × ILIMIT 50μA where RDS(ON)HS is the maximum on-resistance of the high-side MOSFET(s) at +25°C. At higher temperatures, the ILIM current increases to compensate for the temperature coefficient of the high-side MOSFET(s). ______________________________________________________________________________________ 23 MAX8664 The high-side MOSFET(s) operate as duty-cycle control switches and have the following major losses: the channel conduction loss (PHSCC), the overlapping switching loss (PHSSW), and the drive loss (PHSDR). The maximum power dissipation could occur either at VIN(MAX) or VIN(MIN): MAX8664 Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response Input Capacitor The input filter capacitors reduce peak currents drawn from the power source and reduce noise and voltage ripple on the input caused by the circuit’s switching. The input capacitors must meet the ripple current requirement (IRMS) imposed by the switching currents. The ripple current requirement can be estimated by the following equation: IRMS = 1 VIN (IOUT1)2 × VOUT1 × (VIN − VOUT1) + (IOUT2 )2 × VOUT2 × (VIN − VOUT2 ) Choose a capacitor that exhibits less than 10°C temperature rise at the maximum operating RMS current for optimum long-term reliability. way that the high-side MOSFET’s drain is close and near the low-side MOSFET’s source. This allows the input ceramic decoupling capacitor to be placed directly across and as close as possible to the high-MOSFET’s drain and the low-side MOSFET’s source. This helps contain the high switching current within this small loop. 3) Pour an analog ground plane in the second layer underneath the IC to minimize noise coupling. 4) Connect input, output, and VL capacitors to the power ground plane; connect all other capacitors to the signal ground plane. PCB Layout Guidelines 5) Place the MOSFETs as close as possible to the IC to minimize trace inductance of the gate drive loop. If parallel MOSFETs are used, keep the trace lengths to both gates equal and short. Careful PCB layout is an important factor in achieving low switching losses and clean, stable operation. The switching power stage requires particular attention. Follow these guidelines for good PCB layout: 6) Connect the drain leads of the power MOSFET to a large copper area to help cool the device. Refer to the power MOSFET data sheet for recommended copper area. 1) A multilayer PCB is recommended. 7) Place the feedback network components as close as possible to the IC pins. Applications Information 2) Place IC decoupling capacitors as close as possible to the IC pins. Keep separate power ground and signal ground planes. Place the low-side MOSFETs near the PGND pin. Arrange the highside MOSFETs and low-side MOSFETs in such a 24 8) The current-limit setting RC should be Kelvin connected to the high-side MOSFETs’ drain. Refer to the MAX8664 evaluation kit for an example layout. ______________________________________________________________________________________ Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response Chip Information PROCESS: BiCMOS TOP VIEW DH1 1 20 ILIM1 LX1 2 19 FB1 18 PWRGD BST1 3 DL1 4 MAX8664 17 VCC 16 GND VL 5 PGND 6 15 IN 14 OSC/EN12 DL2 7 13 REFIN2 BST2 8 LX2 9 12 FB2 DH2 10 11 ILIM2 QSOP ______________________________________________________________________________________ 25 MAX8664 Pin Configuration Package Information (continued) (The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information go to www.maxim-ic.com/packages.) QSOP.EPS MAX8664 Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response PACKAGE OUTLINE, QSOP .150", .025" LEAD PITCH 21-0055 F 1 1 Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time. 26 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 © 2007 Maxim Integrated Products is a registered trademark of Maxim Integrated Products, Inc.