March 2009 - Boost Converters for Keep-Alive Circuits Draw Only 8.5μA of Quiescent Current

L DESIGN FEATURES
Boost Converters for Keep-Alive
Circuits Draw Only 8.5μA of
by Xiaohua Su
Quiescent Current
Introduction
Industrial remote monitoring systems
and keep-alive circuits spend most of
their time idle. Many of these systems
use batteries, so to maximize run time
power losses,even during low power
idle modes, must be minimized. Even
at no load, power supplies draw some
current to produce a regulated voltage
for keep-alive circuits.
The LT8410/-1 DC/DC boost
converter features ultralow quiescent
current and integrated high value
feedback resistors to minimize the
draw on the battery when electronics
are idle.
An entire boost converter takes very
little space, as shown in Figure 1.
Ultralow Quiescent Current
Low Noise Boost Converter
with Output Disconnect
When a micropower boost converter
is in regulation with no load, the
input current depends mainly on
two things—the quiescent current
(required to keep regulation) and the
output feedback resistor value. When
the output voltage is high, the output
feedback resistor can easily dissipate
more power than the quiescent current
of the IC. The quiescent current of the
LT8410/-1 is a low 8.5µA, while the
integrated output feedback resistors
have very high values (12.4M/0.4M).
This enables the LT8410/-1 to dissipate very little power in regulation
at no load. In fact, the LT8410/-1 can
regulate a 16V output at no load from
3.6V input with about 30µA of average
input current. Figures 2, 3 and 4 show
the typical quiescent and input current
in regulation with no load.
The LT8410/-1 controls power
delivery by varying both the peak
inductor current and switch off time.
This control scheme results in low
output voltage ripple as well as high
efficiency over a wide load range. As
shown in Figure 5, even with a small
0.1µF output capacitor, the output
ripple is typically less than 10mV. The
part also features output disconnect,
which disconnects the output voltage
from the input during shutdown. This
output disconnect circuit also sets a
maximum output current limit, allowing the chip survive output shorts.
An Excellent Choice for
High Impedance Batteries
A power source with high internal
impedance, such as a coin cell battery,
may show normal output voltage on
a voltmeter, but its voltage can collapse under heavy current demands.
This makes it incompatible with high
switch-current DC/DC converters.
The LT8410/-1 has an integrated
power switch and Schottky diode,
and the switch current limits are very
low (25mA for the LT8410 and 8mA
for the LT8410-1). This low switch
current limit enables the LT8410/-1
to operate very efficiently from high
impedance sources, such as coin cell
batteries, without causing inrush
current problems. Figure 6 shows
the LT8410-1 charging an electrolytic
capacitor. Without any additional external circuitry, the input current for
12
10
1000
8
6
4
2
AVERAGE INPUT CURRENT (µA)
10
QUIESCENT CURRENT (µA)
QUIESCENT CURRENT (µA)
Figure 1. The LT8410/-1 is designed
to facilitate compact board layout.
8
6
4
2
VCC = 3.6V
100
VCC = 3.6V
0
–40
0
40
80
TEMPERATURE (°C)
Figure 2. Quiescent current vs
temperature—not switching
22
120
0
0
4
8
12
VCC VOLTAGE (V)
Figure 3. Quiescent current
vs VCC voltage—not switching
16
10
0
10
20
30
OUTPUT VOLTAGE (V)
40
Figure 4. Average input current
in regulation with no load
Linear Technology Magazine • March 2009
DESIGN FEATURES L
100µH
VIN
2.5V to 16V
2.2µF
SW
CAP
VCC
VOUT
VOUT = 16V
0.1µF*
LT8410
VREF
SHDN
CHIP
ENABLE
604K
GND
0.1µF
FBP
412K
*HIGHER VALUE CAPACITOR IS REQUIRED
WHEN THE VIN IS HIGHER THAN 5V
100
10
VIN = 3.6V
6
4
2

R1 
1.30 •  1 + 
 R2 
VIN = 12V
90
8
EFFICIENCY (%)
VOUT PEAK-TO-PEAK RIPPLE (mV)
the SHDN pin below 0.3V shuts down
the part and reduces input current to
less than 1µA. When the part is on, and
the SHDN pin voltage is close to 1.3V,
0.1µA current flows out of the SHDN
pin. A programmable enable voltage
can be set up by connecting external
resistors as shown in Figure 7.
The turn-on voltage for the configuration is:
0.1µF
VIN = 5V
80
and the turn-off voltage is:
VIN = 3.6V
70

R1 
(1.24 − R3 • 10 −7 ) •  1 +  − (R1• 10 −7 )
 R2 
60
50
0
0.01
0.1
1
LOAD CURRENT (mA)
40
0.01
10
0.1
1
10
LOAD CURRENT (mA)
100
Figure 5. General purpose bias with wide input voltage and low output voltage ripple
the entire charging cycle is less than
8mA.
Tiny Footprint with
Small Ceramic Capacitors
Available in a tiny 8-pin 2mm × 2mm
DFN package, the LT8410/-1 is internally compensated and stable for
a wide range of output capacitors. For
most applications, using 0.1µF output
capacitor and 1µF input capacitor is
sufficient. An optional 0.1µF capacitor
at the VREF pin implements a soft-start
feature. The combination of small
package size and the ability to use
small ceramic capacitors enable the
VIN
2.5V to 16V
LT8410/-1 to fit almost anywhere.
Figure 1 shows the size of a circuit
similar to that shown in Figure 4,
illustrating how little board space
is required to build a full featured
LT8410/-1 application.
where R1, R2 and R3 are resistance
in Ω. Programming the turn-on/turnoff voltage is particularly useful for
applications where high source impedance power sources are used, such as
energy harvesting applications.
By connecting an external capacitor (typically 47nF to 220nF) to the
VREF pin, a soft-start feature can be
implemented. When the part is brought
continued on page 29
ENABLE VOLTAGE
SHDN Pin Comparator and
Soft-Start Reset Feature
R1
An internal comparator compares the
SHDN pin voltage to an internal voltage reference of 1.3V, giving the part
a precise turn-on voltage level. The
SHDN pin has built-in programmable
hysteresis to reject noise and tolerate
slowly varying input voltages. Driving
R3
CONNECT TO
SHDN PIN
R2
Figure 7. Programming the enable
voltage by using external resistors
L1
220µH
C1
2.2µF
TURN ON/OFF
SW
CAP
VCC
VOUT
LT8410-1
VREF
SHDN
GND
FBP
C2
1.0µF
C3
10000µF
R1
604k
R2
412k
C1: 2.2μF, 16V, X5R, 0603
C2: 1.0μF, 25V, X5R, 0603*
C3: 10000μF, Electrolytic Capacitor
C4: 0.1μF, 16V, X7R, 0402
L1: COILCRAFT LPS3008-224ML
* HIGHER CAPACITANCE VALUE IS REQUIRED FOR
C2 WHEN THE VIN IS HIGHER THAN 12V
SHDN VOLTAGE
2V/DIV
C4
0.1µF
VOUT = 16V
VOUT VOLTAGE
10V/DIV
INPUT CURRENT
5mA/DIV
INDUCTOR
CURRENT
10mA/DIV
VIN = 3.6V
20s/DIV
Figure 6. Capacitor charger with the LT8410-1 and charging waveforms
Linear Technology Magazine • March 2009
23
DESIGN FEATURES L
and expensive solution than typical
microprocessor-controlled methods.
The simplest scheme uses a resistor
divider from the VREF pin to the CTRL
pin, where the top resistor in the divider is an NTC (negative temperature
coefficient) resistor. While simple,
this method suffers from nonlinear
temperature coefficient of the NTC
resistor. A more precise method uses
a transistor network as shown in Figure 7. The PTC (Positive Temperature
Coefficient) of the CTRL pin voltage is
realized by an emitter follower of Q1
and a VBE multiplier of Q2.
Assuming:
VBE(Q1) = VBE(Q2) = VBE
dT
dT
VCTRL = VREF −
=
2mV
°C
R8
V
R7 BE
with
PTC =
56
dVCTRL R8 2mV
=
•
dT
R7 °C
52
50
48
R8
=
R7
46
44
42
40
–50
–25
0
25
50
75
TEMPERATURE (°C)
100
125
Figure 8. Temperature response
of the circuit shown in Figure 7
LT8410, continued from page 23
out of shutdown, the VREF pin is first
discharged for 70µs with a strong pull
down current, and then charged with
10µA to 1.235V. This achieves soft
start since the output is proportional
to VREF. Full soft-start cycles occur
even with short SHDN low pulses
since VREF is discharged when the
part is enabled.
In addition, the LT8410/-1 features
a 2.5V to 16V input voltage range, up
R1
=
R2
dVOUT 2mV
+
• VOUT
dT
°C
−1
2mV
• VREF
°C
to 40V output voltage and overvoltage
protection for CAP and VOUT.
Conclusion
The LT8410/-1 is a smart choice
for applications which require low
quiescent current and low input current. The ultralow quiescent current,
combined with high value integrated
feedback resistors, keeps the average
input current very low, significantly
EFFICIENCY (%)
90
=
2mV
°C
The LT3571 is a highly integrated,
compact solution to APD bias supply
design. It provides a useful feature set
and the flexibility to meet a variety of
challenging requirements, such as low
noise, fast transient response speed,
and temperature compensation. With
a high level of integration and superior performance, the LT3571 is the
natural choice for APD bias supply
design. L
extending battery operating time.
Low current limit internal switches
(8mA for the LT8410-1, 25mA for the
LT8410) make the part ideal for high
impedance sources such as coin cell
batteries. The LT8410/-1 is packed
with features without compromising
performance or ease of use and is
available in a tiny 8-pin 2mm × 2mm
package. L
The accurate programmable output
current limit of the LT3653 and
LT3663 eliminates localized heating
from an output overload, reduces the
maximum current requirements on the
power components, and makes for a
robust power supply solutions. L
VIN = 8V
VIN = 15V
80
VIN = 30V
70
60
50
40
0.1
dT
Conclusion
100
LT3653/63, continued from page 21
of handling 60V transients. Figure 4
shows the circuit efficiency at multiple
input voltages.
The current limit of the application
is set to 1.2A, therefore, the power path
components are sized to handle 1.2A
maximum. To reduce the application
footprint, the LT3663 includes internal
compensation and a boost diode. The
RUN pin, when low, puts the LT3663
into a low current shutdown mode.
VREF
VOUT
2mV
VBE +
•
°C dVOUT dT
VBE •
dVBE(Q2)
Conclusion
Given VOUT at room and dVOUT/DT,
the R1/R2 and R8/R7 can be calculated as follows
54
VAPD (V)
dVBE(Q2)
=
Simulation using LTspice always
gives a good starting point. The circuit
shown in Figure 7 is designed to have
VAPD = 50V (VOUT = 55V) at room and
dVAPD/dT = 100mV/°C (dVOUT/dT =
100mV/°C). The measured temperature response is shown in Figure 8,
which is very close to the design
target.
then the CTRL pin voltage is
60
58
=
dVBE(Q1)
dT
and
dVBE(Q1)
Resistors R5–R9 are selected to make
I(Q1) = I(Q2) ≈ 10µA, and
0.3
0.5
0.7
0.9
OUTPUT CURRENT (A)
1.1
1.3
Authors can be contacted
at (408) 432-1900
Figure 4. Efficiency of the circuit in Figure 3
Linear Technology Magazine • March 2009
29