V03N1 - FEBRUARY

LINEAR TECHNOLOGY
FEBRUARY 1993
IN THIS ISSUE . . .
COVER ARTICLE
New Switching Regulators
Maximize Efficiency ....... 1
Milton Wilcox and Randy Flatness
Editor's Page .................. 2
Richard Markell
DESIGN FEATURES
New Isolators Replace
“Optos” ........................... 3
VOLUME III NUMBER 1
New LTC1148/LTC1149
Switching Regulators
Maximize Efficiency
from Milliamps to Amps
by Milton Wilcox and
Randy Flatness
James Herr
High Speed, Precision,
Single-Supply Op Amps .. 5
William H. Gross
5V/3V, 12-Bit ADC Perfor–
mance Comparison ......... 8
William Rempfer
DESIGN IDEAS
A Simple, Efficient Laser
Power Supply ................. 13
Jim Williams
Switching Regulator
Provides ±15V from an
8V – 40V Input ................ 14
Brian Huffman
A Twelve-Bit, Micropower
Battery-Current Monitor . 16
Sammy Lum
200mA Output,
1.5V-to-5V Converter ....... 17
Introduction
The LTC1148 and LTC1149 are the
first stepdown switching regulators
that provide extremely high operating
efficiencies (typically greater than 90%)
over the entire load-current range demanded by the next generation of sophisticated notebook computers,
cellular phones, and handheld instruments. The LTC1148/LTC1149 extends
battery life by providing efficiencies,
when these devices are in sleep or
standby modes, that are nearly as
high as those for full-power operation.
Furthermore, losses are reduced to
the point where no heat sinking is
required.
100
Jim Williams
Peter Schwartz
Multi-Output 3w Power
Supply Uses 2 AA Cells ... 20
95
EFFICIENCY (%)
LT1158 H-Bridge uses
Ground-Referenced Current
Sensing for Protection .... 18
LTC1148-5
VIN = 10V
90
85
Steve Pietkiewicz
LTC1157 Switch for 3.3V
PC Card Power ................ 22
Tim Skovmand
New Device Cameos ........ 23
80
20mA
0.2A
2A
LOAD CURRENT (A)
1148_1.eps
Figure 1. Thanks to Burst ModeTM operation,
the LTC1148-5 is capable of greater than 90%
efficiency from 20mA to 2A of output current
The extremely wide operating range
is illustrated by the typical efficiency
curve of Figure 1. The LTC1148 and
LTC1149 accomplish this feat by automatically and smoothly changing from
synchronously switched current-mode
operation at high output currents to
Burst ModeTM operation at low output
currents. Members of the LTC1148/
LTC1149 family can operate from input voltages as low as 4V (LTC1148) to
as high as 48V (LTC1149). They are all
capable of 100% duty cycles for verylow-dropout operation, and all have
built-in current limiting. Line and load
transient response is excellent under
a variety of conditions, including when
making the transition from Burst ModeTM
operation to full-current operation.
The small size and high efficiency of
the LTC1148/LTC1149 family make
DC-to-DC conversion feasible in the
most restricted spaces of today’s portable electronics. Take, for example,
the problem of locally dropping 5V
to 3.3V on a logic board. In many
cases the dissipation of a linear regulator, even for this modest voltage
drop, is unacceptable, because there
is simply no way to remove the heat
from the enclosed space. A linear
regulator delivering 1A in this application would dissipate over 1.7W.
Continued on page 10
DESIGN
EDITOR'SFEATURES
PAGE
LTC Switching Regulators Break the 90%
Efficiency Barrier—The LTC Technology
Machine Marches On
by Richard Markell
This issue of Linear Technology
proudly spotlights our LTC1148/1149
switching regulators, which break the
elusive 90% efficiency barrier. Not only
do we pass the barrier but we burst
through, without stopping, across the
entire load range from milliamps to
amps. Our lead article introduces the
LTC1148/LTC1149 synchronous
switching regulators, which provide
very high efficiencies at all current
levels. In addition to the 94%-efficient
5 volt to 3.3 volt converter mentioned
in the article, we have achieved the
following efficiencies in the lab since
the article was written:
• An LTC1148 converter using 6-to12 volt input with a 5 volt output at
4 amps output current. This
converter achieves greater than
90% efficiency.
• An LTC1148 circuit that converts
6 volts to 5 volts at 1/2 amp,
achieving 96% efficiency.
• An LTC1149 24 volt to 5 volt
converter with output current of >1
amp, achieving 90% efficiency.
• A high-current circuit using the
LT1158 (featured in the February
1992 LT Magazine), which converts
5 volts to 3.3 volts. This circuit’s
efficiency is 90% at 10 amps and
88% at 15 amps.
Efficiency, by itself, is no panacea,
but higher and higher efficiencies imply less and less heat in your enclosure. This may not be a problem for a
remote data-collection system in the
Canadian North, but it certainly is a
problem in notebook and laptop computers. (If you require additional
information on these leading-edge developments in power-supply efficiencies, do not hesitate to call the
Applications Group at the LTC factory.)
Another design feature highlights
signal isolation applications and circuitry using a new isolation device
from LTC. This device, the LTC1145/
LTC1146 uses a special integratedcircuit lead frame to form the isolation
capacitors. The device uses two dice
(one driver and one receiver) coupled
through the isolation capacitors on the
opposite ends of the lead frame. The
article also describes how the design
provides sufficient high-voltage isolation for even the most critical requirements with high transient immunity.
High speed, precision operational
amplifiers are also featured in this
issue, as are new 12-bit ADCs. Six new
high-speed op amps—their circuit topologies, performance specifications,
and some application areas—are discussed at length in another Design
Feature. Analog-to-Digital converters
are featured in an article describing the
new 3.3 and 5 volt, 12-bit converters
from LTC.
Also in this issue, I am pleased to
note, is a large collection of Design
Ideas. These range from a helium-neon
laser power supply, to a multi-output
power supply from two AA batteries, to
an H-bridge driver circuit for a 15-amp
DC motor. Circuitry for PCMCIA card
power conditioning and 12-bit battery
current monitoring are also featured in
the Design Idea area. Good reading!
FAE Cameo: Tom Mosteller
LTC has eighteen Field Application
Engineers (or FAEs) spread throughout the world to assist customers in the
design and selection of circuits available from LTC. Each FAE has a system-design background and has a
working knowledge of the world from a
system designer’s point of view. All of
our FAEs are available by phone and,
in certain situations, in person, to help
with the design of your circuitry. This
space will profile one FAE per issue.
Tom Mosteller works out of LTC’s
Northeast Office. He covers the states
of Pennsylvania, Maryland, West Virginia, Virginia, and Delaware, and parts
of New York and New Jersey. Tom’s
2
expertise is in the areas of signal processing and switching power supply
design. He also has extensive experience in process control. “My greatest
challenge in the signal processing field,”
writes Tom, “was an ECG-monitor front
end that amplified millivolts while ignoring kilovolts. This monitor had an
isolated power supply, low-level amplifiers, slew-rate limiters, switchedcapacitor filters, a pressure-bridge
exciter and amplifiers, and A-to-D
converters.”
Tom also enjoys writing. He wrote
most of Application Note 50 on a portable computer while on a balcony
overlooking the beach in Lewes,
Delaware (while he was on vacation,
no less!). Tom’s byline has also appeared in PC Magazine, Circuit Cellar
Ink, and other publications.
Tom has been married to his wife
Rose for eleven years. They have one
son, Kyle, who is seven years old and in
the second grade. Tom enjoys playing
computer games and writing stories
with Kyle on the computer. (Who enjoys computer games?) He also enjoys
audio, YMCA Indian Guides, camping
and fishing with Kyle, and helping with
Rose’s software business. Tom can be
reached through LTC’s Northeast Region Sales Office as listed on the back
of this magazine.
Linear Technology Magazine • February 1993
DESIGN FEATURES
New Isolators Replace “Optos”
The LTC1145 and LTC1146 are a
new generation of signal isolators. Previously, signal isolation was accomplished by means of opto-isolators.
Light from an LED was detected across
a physical isolation barrier by either a
photo diode or transistor and converted to an electrical signal. Isolation
levels up to thousands of volts were
easily achieved.
Attempts have been made to provide
signal isolation on a single silicon die.
Problems arose due to reliability constraints of damage from ESD or overvoltage. A new technique, using a
capacitive lead frame, overcomes the
problems associated with single-package signal isolation. Further, this technique is suitable for use in thin
surface-mount packages—a solution
not available with opto-isolators. The
data rates are 200kbps for the LTC1145
and 20kbps for the LTC1146. Both
parts can sustain over 1000V across
their isolation barriers.
is powered from the input signal. This
allows interchangeability with optoisolators, since no auxiliary power
source is needed on the input side. The
driver contains a high-frequency oscillator whose differential outputs are
capacitively coupled to the receiver.
The receiver detects the presence or
absence of the driver’s oscillations and
outputs a logic 1 or 0, as appropriate.
The driver consists of a bias circuit,
a Schmitt trigger, and a differential
output oscillator to drive the isolation
capacitors. The Schmitt trigger prevents the oscillator from driving the
capacitors until the input rises to a
sufficient level (3V for the LTC1145 or
2V for the LTC1146) to provide a clean
square-wave output. Two versions of
the drivers are available, one with a
5MHz oscillator frequency (LTC1145),
and one with a 400kHz oscillator frequency (LTC1146). The 5MHz unit provides reliable data communication at
200kbps whereas the 400kHz unit can
operate at 20kbps. Speed has its price:
the LTC1145 consumes 600µA while
the LTC1146 requires only 60µA of
input current. These power levels are
significantly below LED opto-isolators.
The receiver is designed to detect
differential drive from the capacitors. A
differential input comparator with
Circuit Design
The isolator IC includes two dice
(see Figure 1). The first is a driver
designed to drive two isolation capacitors. The second contains a receiver
and filter which detect the drive signal
and provide a digital output. The driver
DRIVER
by James Herr
300mV of hysteresis detects the incoming signal while rejecting commonmode noise. The output of the
comparator triggers a one-shot circuit
with a period longer than that of the
drive frequency. The retriggerable oneshot has a constant output as long as
the input signal exists. When an input
signal is absent for one time-out period, the one-shot goes low.
Also connected to the one-shot is a
two-bit counter that works as a digital
filter. The counter is clocked from a
second oscillator running at around
10MHz. The output of the counter
requires four oscillator clock periods to
change state and is gated by the output
of the one-shot. When an input signal
triggers the one-shot, a logical 1 is
presented to the data input of the filter.
Four clock periods later the output of
the filter goes high. If the signal is
absent for four clock cycles, the filter
output returns low. This digital filter
eliminates possible erratic operation
from spikes or temporary overloads of
the input amplifier. The noise or interference signal must exist for four clock
periods of the digital filter before an
output-stage change can occur.
The LTC1145’s digital filter can be
clocked externally to lower its
effective bandwidth. The LTC1146’s
RECEIVER
VCC
TTL
BUFFER
1-SHOT
OUTPUT
11
TTL
BUFFER
DATA
OUTPUT
10
12
DATA INPUT
1
A
C
+
OSCILLATOR
E
D
B
–
D
IN
Q
OUT
FILTER
1-SHOT
F
18
9
CD
CD
GND2
GND1
OSCILLATOR
ISOLATION
BARRIER
IN
EXT OSC
8
CLOCK
DETECTOR
1145_1.eps
Figure 1. Two-chip design provides isolation barrier
Linear Technology Magazine • February 1993
3
DESIGN FEATURES
digital-filter bandwidth
can be reduced by connecting a capacitor at pin 8.
Lower filter bandwidths
are desirable to eliminate
longer noise bursts. Experimental work using this
isolator and filter has
shown no corrupted data
over a wide range of common-mode inputs.
opto-isolator replacement.
One possible application is
1 DIN
GND1 18
1 DIN
GND1 18
an isolated RS232 receiver.
The DIN pin of LTC1145 is
driven by an RS232 signal
ISOLATION
ISOLATION
BARRIER
through a 5.1kΩ resistor
BARRIER
(Figure 3). The DOUT pin of
+5V
the
LTC1145 presents
7 NC
VCC 12
7 NC
VCC 12
isolated,
TTL-compatible
8 OSCIN
OS 11
8 OSCIN
OS 11
output
signals.
The GND2
DOUT 10
TTL OUT
9 GND2
DOUT 10
9 GND2
pin of the LTC1145 is connected to the same ground
2. LTC1145/
Isolation Capacitors Figure
potential as the receiving
LTC1146 pin description
Figure 3. Isolated, low-power RS232
A specialized lead frame
end of the link. The isolator
is needed to form the isolation capaci- metal fingers and bonding posts re- can accommodate differences of up to
tors. This lead frame can be manufac- place the 5 center pins on each side of 1kV between GND1 and GND2.
tured with the same high-volume the 18 pin package, as illustrated in
Another application is an isolated,
techniques used for conventional in- Figure 2. The dielectric for the capaci- thermocouple-sensed temperature-totegrated-circuit lead frames, and at tors is the plastic package molding frequency converter (see Figure 4). The
approximately the same cost. The two compound. The material has a high output of I3 produces a 0kHz–1kHz
dice (driver and receiver) are placed at dielectric constant and a high break- pulse train in response to a 0°C to
100°C temperature excursion (see LTC
the opposite ends of the lead frame down voltage.
Application Note 45 for the details).
and coupled through the isolation
The pulses from I3 drive the DIN pin of
capacitors. The capacitance between Applications
The LTC1145/LTC1146 can be used LTC1146. The GND1 pin is connected
the input and output is on the order
of 1pF. This provides sufficient isola- in a wide range of applications where to the same ground potential as I3. The
tion in even the most critical of applica- voltage transients, differential ground DOUT pin of LTC1146 presents isolated,
tions and is suitable for handling high potentials, or high noise may be en- TTL-compatible output signals. The
countered, such as isolated serial data circuit consumes only 460µA maxivoltages with high dV/dt.
Each 1pF capacitor is formed by interfaces, isolated analog-to-digital mum, allowing it to operate from a 9V
three parallel metal fingers spaced converters for process control, iso- battery.
about 20 mils apart. The capacitors’ lated FET drivers, and low-power
5.1kΩ
RS232
IN
1145_3.eps
1145_2.eps
CONTROL
AMPLIFIER
+6V
TYPE K
THERMOCOUPLE
NC
+6V
10k
100k
k
LT1025
GND
–
+
–
6.81k*
1M
A1
LTC1049
+
"A"
R–
Q1
2N3904
0.02µF
+V
Q2
2N3906
I1
V➞ F
I2
100k
OUTPUT
0-100°C = 0-1kHz
C1
100pF
C3
0.47µF
C4
300pF
240k
+
1.5k
+6V
6.8µF
I3
DIN
LT1004-1.2
100°C
TRIM
DOUT
16
9
LTC1146
GND1
14
15
S1
390pF †
10
11
S4
C2
2
*IRC/TRW-MTR-5/+120PPM
3
1
GND2
LTC201
CHARGE
PUMP
S2
S3
6
7
8
= 74C14
† = POLYSTYRENE
FOR GENERAL PURPOSE 4mV FULL SCALE V ➞ F DELETE
THERMOCOUPLE/LT1025 PAIR AND DRIVE POINT "A."
1145_4.eps
Figure 4. Isolated temperature-to-frequency converter
4
Linear Technology Magazine • February 1993
DESIGN FEATURES
High Speed Comes to Precision,
Single-Supply Op Amps
by William H. Gross
The LT1211–LT1216 Family
LTC proudly introduces a new family of dual and quad single-supply precision op amps. These new amplifiers
are fast, with slew rates of up to 50V/µs
and gain-bandwidth products to
28MHz. All members of the family are
unity-gain stable and operate on any
single supply between 2.5V and 36V as
well as on split supplies from ±2V to
±18V. The design of an op amp involves
a trade off between slew rate and supply current consumption. Rather than
force a major compromise, three new
amplifiers were developed. Each of the
three amplifier types is offered as both
a dual and a quad; hence there are six
part numbers.
The LT1211 dual and the LT1212
quad amplifiers draw the least quiescent current in the family, only 1.3mA
per amplifier. The LT1211/LT1212
op amps have a 14MHz gain-bandwidth product and a peak slew rate of
7V/µs. They settle to 0.01% of a 10V
step in 2.2µs.
The LT1213 dual and the LT1214
quad amplifiers have twice the gainbandwidth product of the LT1211/
LT1212 amps at 28MHz. The peak slew
rate of the LT1213/LT1214 is 12V/µs
and the supply current goes up to 3mA
per amplifier. They settle to 0.01% of a
10V step in 1.2µs.
The LT1215 dual and the LT1216
quad amplifiers are the fastest in the
family, with a slew rate of 50V/µs. The
gain-bandwidth product is 23MHz, and
the quiescent supply current is 5mA
per amplifier. They settle to 0.01% of a
10V step in 480ns.
The dual op amps are available with
the industry standard pin-out in either
8-pin SO or 8-pin mini-DIP packages.
The quad op amps have the standard
pin-out in the 14 pin DIP and the same
pin-out, with two no-connect pins on
one end, in the narrow (150mil) SO16
package. LTC is the first to offer a precision quad op amp in the narrow SO
package. Precision amplifiers consist
Linear Technology Magazine • February 1993
of many transistors and require careful thermal layout; therefore precision
op amps require much larger dice than
simple commodity op amps. For this
reason precision quad op amps have
not been available in the narrow SO
package; they are simply too wide to fit.
To address this problem, LTC developed a special lead frame for the standard narrow SO16 mold. This new lead
frame maximizes the area available for
the die, but has an internal length-towidth ratio of 3 to 1. If one die were used
to make a quad op amp for this package, it would also have to have a lengthto-width ratio (aspect ratio) of 3 to 1.
There is a trend
to lower supply voltages for
analog signal processing.
Engineers can no longer
depend on having ± 15V or
± 12V supplies available.
Often, only a single + 5V
supply is available, and the
trend is to + 3.3V
High aspect ratios cause manufacturing problems that increase costs. To
solve this problem, we die-attach two
dual op amp dice to the lead frame and
bond them out with the standard pinout. This gives us an effective aspect
ratio of 3 to 1 without having to actually
make a die with that aspect ratio.
Performance Goals
There is a trend to lower supply
voltages for analog signal processing.
Engineers can no longer depend on
having ±15V or ±12V supplies available. Often, only a single +5V supply is
available, and the trend is to +3.3V.
Our design goal was to make these
single-supply op amps operate well
with supply voltages as low as 2.5V.
The introduction of LTC’s 12-bit Ato-D converters that operate on 3.3V
supplies, such as the LTC1282,
LTC1287, and LTC1289, emphasizes
the need for low voltage, 12-bit-accurate op amps. For converters operating
with a 2.5V reference, we would like an
op amp whose output swings from
ground to 2.5V. The new LTC A-to-D
converters include the sample-and-hold
on the same chip; therefore an op amp
is needed that settles cleanly after the
input hold cap is connected. For the
160kHz-sampling LTC1282, the amplifier must deliver several milliamps of
output current and settle in less than
1µs. Since the LSB in a 12-bit system
using a 2.5V reference is only 610µV,
the settling must have no thermal tails.
Process Technology
The LT1211 family of amplifiers are
manufactured on LTC’s complementary bipolar process. The low drift and
noise requirements of the amplifier,
combined with the low open-loop output impedance required to drive sampling A-to-Ds precludes using a CMOS
process. PNP transistors are used in
the input stage so the common-mode
range includes the negative supply. To
make a fast amplifier we need fast
PNPs, hence we could not use our
standard bipolar process with lateral
PNPs. The complementary process has
36V NPNs and PNPs that both have
cutoff frequencies of 600MHz. Both
transistors in this process are optimized for high gain, excellent matching, and low noise.
Circuit Topology
The circuit topology of the LT1211 is
conventional, with some high speed
innovations. The amplifier consists of
two gain stages. The first is a differential-voltage to single-ended-current
(transconductance) stage whose output drives a high-input-impedance,
inverting voltage-gain stage configured
5
DESIGN FEATURES
V+
I1
I2
I3
I4
I5
I6
BIAS
CM
Q13
–IN
Q1
L
Q14
Q15
Q3 Q4
Q2
L
+IN
Q11
OUT
RF
Q7
Q12
Q10
CF
Q8
Q16
Q5
Q9
Q6
CI
I7
CO
I8
V–
1211_1.eps
Figure 1. LT1211 simplified schematic
as a Miller integrator. An emitter follower buffers the output of the second
stage and sources current into the
load. The current in this follower is
monitored with a second loop that
provides the output sink current.
One advantage of general-purpose
op amps is that they can take large
differential inputs without excessive
input current flowing. This is because
they use lateral PNPs with emitterbase breakdown voltages greater than
36V in the input stage. Conventional
NPNs and fast PNPs have low emitterbase breakdown voltages and require
clamps across the inputs to protect
them. To make the LT1211 as easy to
use as possible, we use lateral PNP
transistors in the input stage. Referring to the simplified schematic (Figure 1), the input PNP emitter followers
(Q1 and Q2) are lateral transistors
with high breakdown. The differentialamplifier PNPs (Q3 and Q4) that convert the differential voltage to a current
are fast transistors. Because Q1 and
Q2 drive the high impedance bases of
Q3 and Q4, the base-emitter capacitance of Q1 and Q2 couples the input
signal to the faster PNPs (Q3 and Q4)
even above the cutoff frequency of the
lateral transistors.
An active load, Q5 through Q9,
maximizes the gain of the first stage for
low noise and low offset-voltage drift.
The base current of Q8 matches the
base current of Q10 for low drift. The
capacitor CI introduces a pole and a
zero in the open-loop gain by rolling off
half the input-stage gain above 1MHz.
6
This reduces the unity-gain frequency
to half the gain-bandwidth product
and therefore increases the gain and
phase margin.
The output current from the first
stage drives the second stage, consisting of Q10, Q11, and Q12. Q10 and
Q11 are emitter followers to increase
the input impedance of this second
stage. Q12 operates in a commonemitter configuration with a currentsource load for maximum voltage gain.
The capacitor CM turns the gain stage
into a Miller integrator. For good phase
margin in the amplifier, the integrator
must work well at the amplifier unitygain frequency. Since Q10 and Q11
operate at fairly low currents, they
generate significant phase shift that
limits the accuracy of the integrator. To
improve the frequency response of this
stage, we add CF and RF to feed-forward the signal around Q10 and Q11.
The output stage buffers the second
stage with emitter follower Q15 and a
current-sink circuit. In order to sink
output current and swing all the way
to the negative supply, an NPN transistor, Q16, must drive the output.
Q14’s collector current is one tenth
that of follower Q15’s and is subtracted from Q13’s emitter current.
Then Q13’s collector current is compared with current source I8; the excess drives Q16. When the current in
Q15 drops, more current drives Q16
and the amplifier sinks current. Capacitor CO stabilizes this feedback loop,
which includes common-base transistor Q13. Because the gain of the
loop is quite large, Q15 never turns off
and the open loop output impedance
stays low for overall amplifier stability.
First order, the overall DC gain of
the op amp is the input stage transconductance times the current gain of
Q10, Q11, Q12, and Q15, times the
load resistor. The transconductance
of the LT1211 first stage is 500nA/mV;
the gain of the transistors is about 100
each; with a 500Ω load the calculated
gain is twenty-five million (25V per
µV). The actual gain is about 2.5 million, due to second-order effects like
Early voltage. The IC layout is optimized to eliminate thermal feedback
that would reduce gain. The layout
also optimizes channel-to-channel
separation, which is typically 100nV/
V (140dB).
Performance
Table 1 describes the typical AC performance of each of these amplifiers.
Table 2 summarizes the DC electrical performance of the low cost grades
of these new amplifiers. The dual op
amps also have selections for improved
offset voltage and drift in the DIP
packages.
Applications
Instead of describing several known
op amp circuits, I will now discuss
some general applications. The LT1211
family of amplifiers are the optimum
solution whenever a combination of
speed and accuracy is needed on low
supply voltage.
Linear Technology Magazine • February 1993
DESIGN FEATURES
With multimedia becoming more
important, the need for CD-quality
audio amplifiers that operate on single
+5V supplies is growing. The LT1211
can handle 1Vrms at a non-inverting
gain of one while operating on single
+5V supply. The LT1211 delivers distortion-free signals over the full 20Hzto-20kHz range; THD is less than
0.001% for all signals up to 1Vrms.
Referenced to 1V, the signal-to-noise
ratio is over 110dB and the supply
current is only 2.6mA for both stereo
channels.
The LT1216 is an ideal op amp for
precision anti-aliasing filters. The
20MHz gain bandwidth supports highQ filters up to 1MHz and the high slew
rate results in a power bandwidth of
2.5MHz at 2.5VP–P. The quad is a natural for state-variable filters where three
or four op amps are needed for each
pole pair. The low offset-voltage drift of
these amplifiers ensures that self-calibrating systems are accurate between
calibrations.
The LTC1196 is an 8-bit A-to-D
converter that operates on a 3.3V supply with an external 2.5V reference.
This converter is fast; it samples the
input at a 450kHz rate. The input
range is from 10mV±5mV to 2.5V±5mV.
The LT1211/LT1213/LT1215 output
swing is guaranteed to be such that
only code “0000 0000” is missing when
the devices are operated on the same
3.3V single supply.
The LT1211 Family
of amplifiers are the
optimum solution
whenever a combination
of speed and accuracy
is needed on low
supply voltage
The LTC1282 is a 12-bit A-to-D
converter with an internal reference
that operates on a single 3.3V supply.
The nominal input range is from 0V to
2.5V. Since the LT1211 can only swing
to within 5mV of ground, the lowest
nine codes cannot be used. This is not
Table 1. Typical AC performance
AC Parameters
too bad considering that there are
4085 codes that are OK. When using
the LTC1289, (another 3.3V, 12-bit Ato-D converter) these codes can be
recovered because an external reference is used. The external reference
can be positioned a little above ground
to shift the full 4096 codes into the
output range of the LT1211.
Summary
The LT1211 family brings a level of
precision and speed to low voltage systems that was previously unavailable.
Operation is guaranteed over the full
military temperature range with single
supplies as low as 2.5V. Input offset
voltage, input bias current, open-loop
gain, and CMRR are comparable with
the best op amps available today. The
three amplifiers span a seven-to-one
range of slew rate with a four-to-one
range of supply current. The open-loop
output impedance is low and these
amps settle to microvolts in fractions of
a microsecond, making them ideal for
data-acquisition systems. These amplifiers are available in small surfacemount packages, including for the first
time the narrow SO16 for the quads.
Table 2. Guaranteed DC performance of low-cost grades
LT1211/12
LT1213/14
LT1215/16
DC Parameters
Slew rate, VS = ±15V
7V/µs
Slew rate, VS = +5V
4V/µs
Settling time, 2V to 0.01%
800ns
Settling time, 10V to 0.01%
2.2µs
Gain bandwidth product
14MHz
Unity gain cross frequency
7MHz
Phase margin
55°
Noise voltage, 0.1 to 10Hz
125nVP-P
Spot noise voltage, 10Hz
12.5nV√Hz
Spot noise voltage, 1kHz
12nV√Hz
12V/µs
8V/µs
500ns
1.2µs
28MHz
13MHz
45°
125nVP-P
10nV√Hz
10nV√Hz
50V/µs
30V/µs
250ns
480ns
23MHz
12MHz
45°
400nVP-P
15nV√Hz
12.5nV√Hz
Max offset voltage
275µV
275µV
450µV
Max offset voltage drift
6µV/°C
6µV/°C
10µV/°C
Max input offset current
30nA
40nA
120nA
Max input bias current
125nA
200nA
600nA
Min input voltage range
VS = 3.3V
0V – 1.8V
0V – 1.8V
0V – 1.3V
VS = 5.0V
0V – 3.5V
0V – 3.5V
0V – 3.0V
Min CMRR
86dB
86dB
86dB
Min output voltage swing
VS = 3.3V
0.01V – 2.5V 0.01V – 2.5V 0.01V – 2.5V
VS = 5.0V
0.01V – 4.2V 0.01V – 4.2V 0.01V – 4.2V
Min open loop gain
VS = 3.3V or 5.0V
250V/mV
250V/mV
150V/mV
VS = ±15V
1200V/mV
1200V/mV 1000V/mV
Min channel separation
128dB
128dB
130dB
Min output current
20mA
30mA
30mA
Max supply current per amp 1.8mA
3.8mA
6.6mA
Operating supply voltage
2.5V – 36V
2.5V – 36V 2.5V – 36V
Linear Technology Magazine • February 1993
LT1211/12
LT1213/14
LT1215/16
7
DESIGN FEATURES
Five and Three Volt, 12-Bit ADC
Performance Comparison
by William Rempfer
Complete ADCs Provide
Lowest Power, Highest Speed
on Single or Dual Supplies
SAMPLE RATE/POWER RATIO (kHz/mW)
12
Device
LTC1273
LTC1275/6
LTC1282
Power
Supplies
+5V
±5V
+3V
or ±3V
The LTC1282 samples at 140kHz
and typically dissipates only 12mW
from either 3V or ±3V supplies. It
digitizes 0–2.5V inputs from a single
3V supply or ±1.25V inputs from ±3V
supplies.
A complete ADC system is provided
by the on-chip sample-and-holds, precision references, and internally
trimmed clocks. The high-impedance
analog inputs are easy to drive and
can be multiplexed without buffer amplifiers. A single 5V or 3V power supply
is all that is needed to digitize unipolar
inputs. (Bipolar inputs require ±5V or
±3V supplies but the negative supply
draws only microamperes of current).
But most significant are the speed/
power ratios, which are higher than
those of any other ADC in this speed
range.
★
S/(N + D) typ
at fINPUT
70dB at 100kHz
70dB at 100kHz
68dB at 70kHz
PDISS
(typ)
75mW
75mW
12mW
5V ADCs Sample at
300kHz on 75mW of Power
The LTC1273, LTC1275, and
LTC1276 have excellent DC specs,
including ±1/2LSB linearity and
25ppm/°C full-scale drift. In addition,
they have excellent dynamic performance. As Figure 2 shows, the ADCs
typically provide 72dB of Signal to
Noise plus Distortion (11.7 effective
bits) at the maximum sample rate of
300kHz. The S/(N + D) ratio is over
70dB (11.3 effective bits) for input
frequencies up to 100kHz.
74
12
11
68
10
LTC1282
VDD = 3V
fS = 140kHz
9
8
LTC1273
VDD = 5V
fS = 300kHz
7
6
5
4
62
56
50
3
2
1
1
10
fIN (kHz)
100
1273_3.eps
8
6
Figure 2. The 300kHz LT1273 gives 70dB
S/(N + D) with 100kHz inputs, the
140kHz LTC1282 gives 68dB at Nyquist
LTC1273/5/6 ON 5V OR ±5V
★
4
AD7880 ADS7800
LTC1292
AD7870
MAX163/4/7
CS5012A
AD1674
AD678
2
0
10
100
AD7886
AD1671
CS5412
1000
SAMPLE RATE (kHz)
10000
1273_1.eps
Figure 1. The LTC 1273/5/6 and the LTC1282 have up to
45 times higher speed/power ratios than competitive ADCs
8
Sample
Rate
300kHz
300kHz
140kHz
140kHz
0
LTC1282 ON 3V OR ±3V
10
Input
Range
0–5V
±2.5V/±5V
0–2.5V
or ±1.25V
S/(N + D) (dB)
The LTC1273, LTC1275, LTC1276,
and LTC1282 provide complete A/D
solutions at previously impossible
speed/power levels. As shown in Table
1, the LTC1273, LTC1275, and
LTC1276 all have the same 300kHz
maximum sampling rate and 75mW
typical power dissipation. The LTC1273
digitizes 0-to-5V inputs from a single
5V rail. The LTC1275 and LTC1276
operate on ±5V rails and digitize ±2.5V
and ±5V inputs, respectively.
Table 1. Four new complete ADCs offer high speed and low power on single or dual 5V or 3V
supplies
ENOBS (EFFECTIVE NUMBER OF BITS)
Four new sampling A/D converters
from Linear Technology, the LTC1273,
LTC1275, LTC1276 and LTC1282,
stand out above the crowd. These new
5V and 3V 12-bit ADCs offer the best
speed/power performance available
today (see Figure 1). They also provide
precision references, internally
trimmed clocks, and fast sample-andholds. With additional features such
as single-supply operation and highimpedance analog inputs, they reduce
system complexity and cost. This article will describe the new ADCs and
discuss the performance and power
trade-offs that should be considered
in selecting a 5V or 3V A/D converter.
This 300kHz sample rate and dynamic performance comes at a power
level that is more stingy than that of
any other ADC in this speed range.
Figure 1 shows a graph of speed/
power ratios for the competitive ADCs.
The speed/power ratio is defined as
the maximum sample rate in kHz divided by the typical power dissipation
Linear Technology Magazine • February 1993
DESIGN FEATURES
in mW. The 4.0kHz/mW of the
LTC1273, LTC1275, and LTC1276 is
better than the best competitive ADC.
Table 2 shows a competitive analysis of currently available ADCs. The
LTC1273, LTC1275, and LTC1276 offer advantages over the rest in every
area, including performance, function,
and power.
Even More Power Savings:
3V ADC Samples at 140kHz
on 12mW
The low-power, 3V LTC1282 provides even more impressive speed/
power performance. As fast and dynamically accurate as many power hungry, dual- and triple-supply ADCs, this
complete 3V or ±3V sampling ADC provides extremely good performance on
only 12mW of power. DC specs include
±1/2LSB maximum linearity and the
internal reference provides 25ppm
maximum full-scale drift. Figure 2
shows 11.4 effective bits at a 140kHz
sample rate with 11.0 effective bits at
the Nyquist frequency of 70kHz. The
speed/power ratio, as shown in Figure
1, is an outstanding 11.7kHz/mW.
The LTC1282 is ideal for 3V systems
but will also find uses in 5V designs
where the lowest possible power consumption is required. It interfaces easily to 3V logic but can also talk well to
5V systems. The LTC1282 can receive
5V CMOS levels directly and its 0-to3V outputs can meet 5V TTL levels and
connect directly to 5V systems.
Performance Comparison
Table 3 compares the performance
of the new ADCs to another recent,
low-cost product, the AD1674. The 5V
LTC1273 offers three times the speed
at one fifth the power, and the 3V
device goes even further. The table
shows that using the 3V LTC1282
gives even greater savings in power
than the LTC1273, with only modest
reductions in speed, accuracy, and
noise. The power dissipation has been
reduced six times with only a 50%
reduction in speed. Linearity and drift
don’t degrade at all in going to the 3V
device. The noise of the LTC1282 is
slightly higher, due to the reduced
input span and the lower operating
current, but the converter still gives
more than 70dB typical S/(N + D).
Compared to the AD1674, the LTC1282
offers 40% higher sampling rate and
30 times lower power.
Conclusion
These new 5V and 3V ADCs offer
the best speed/power performance
available today. They also provide precision references, internally trimmed
clocks, and fast sample-and-holds.
With additional features such as singlesupply operation and high-impedance
analog inputs, they reduce system complexity and cost. For performance,
power, and cost, these new ADCs must
be considered for new designs.
Table 2. Competitive analysis of current
ADCs
LTC1273/5
Internal reference
✔
Internal clock (no crystal req’d) ✔
S/(N + D) at fIN=100kHz (typ) 70dB
High-impedance analog input
✔
Unipolar and bipolar inputs
✔
≥300kHz sample rate
✔
Single-supply operation
LTC1273
Low power (<150mW)
✔
3V upgrade path
✔
AD1674
✔
✔
68dB
AD7870/5/6
✔
✔
—
AD7800
✔
—
✔
AD678
✔
✔
64dB
ADS7880
✔
✔
69dB
✔
MAX163/4/7
✔
69dB
✔
✔
✔
✔
Table 3. Performance comparison
LTC1273
Parameter
on +5V
Power dissipation (typ) 75mW
Sample rate
300kHz
Conversion time (max) 2.7µs
INL (max)
±1/2LSB
Typical ENOBs
11.7
Linear input bandwidth
(ENOBs >11 bits)
125kHz
Linear Technology Magazine • February 1993
LTC1282
AD1674
on +3V or ±3V on +5V or ±15V
12mW
385mW
140kHz
100kHz
6µs
10µs
±1/2LSB
±1/2LSB
11.4
11.5
70kHz
100kHz
9
DESIGN FEATURES
VIN
5V
+
+
1µF
100µF
VIN
Si9430
P DRIVE
0V = NORMAL
>1.5V = SHUTDOWN
L1
50µH
SHUTDOWN
RS
50mΩ
VOUT
3.3V/2A
LTC1148-3.3
SENSE+
1000pF
ITH
3300pF
SENSE–
+
IRLR024
CT
1k
470pF
S GND
330µF
1N5817
N DRIVE
P GND
RS = SL-1R050J, KRL BANTRY (603) 668-3210
L1 = CTX50-2-MP, COILTRONICS (305) 781-8900
1148_2.eps
Figure 2. This LTC1148 5V-to-3V converter circuit achieves 94% efficiency at 1A output
current
Continued from page 1
The LTC1148 5V to 3.3V converter
shown in Figure 2 has 94% efficiency
at 1A output. In other words, the
LTC1148 dissipates only 200mW while
delivering 3.3W to the load.
Table 1 gives an overview of the
different members of the LTC1148/
LTC1149 family and several of their
applications. Each device has adjust-
able, fixed 3.3V, and fixed 5V versions
and is available in both DIP and SOIC
(narrow) surface-mount packages.
High Performance
with High Efficiency
The LTC1148/LTC1149 synchronous switching regulator controllers
use the constant off-time, current-
mode architecture shown in Figure 3.
Current-mode operation was judged
to be mandatory for its well known
advantages of clean start-up, accurate current limit, and excellent line
and load regulation. The constant offtime architecture adds to this list simplicity (neither an oscillator nor ramp
compensation is required), inherent
100% duty cycle in dropout conditions, and constant inductor-ripple
current. Because the off-time is constant, the operating frequency changes
with input voltage. For example, in an
LTC1149-5 application, the frequency
will increase 50% when VIN is doubled
from 10V to 20V.
To maximize the operating efficiency
over a wide current range, loss-reducing circuit techniques must be carefully
applied. For example, synchronous
switching (replacing the diode in a
stepdown regulator with a switched
transistor) may buy several percentage
points in efficiency at high output currents, but will cost many more if allowed
to continue at low output currents. This
is the principal reason why the
VIN
VIN
CIN
LOW
DROPOUT
10V REG
P DRIVE
VCC
L
RSENSE
VOUT
COUT
N DRIVE
LTC1149
ONLY
–
V
+
–
1-SHOT
C
+
–
t OFF
CT
G
1.25V
REF
+
1148_3.eps
Figure 3. The LTC1148/LTC1149 architecture features constant-offtime current-mode operation, synchronous switching,
and automatic transition to Burst ModeTM operation
10
Linear Technology Magazine • February 1993
DESIGN FEATURES
Table 1. LTC1148/LTC1149 features
100
LTC1148
I2 R
GATE CHARGE
EFFICIENCY/LOSS (%)
95
LTC1148
IDC
80
Continuous input
voltage ≤ 12V
Low dropout 5V
Adjustable/
multiple output
75
10mA
5V to 3.3V/
4-cell to 3.3V
90
85
30mA
100mA 300mA
LTC1148-3.3 LTC1148-5
Continuous input
voltage ≤ 48V
1A
3A
LTC1149
✔
✔
✔
✔
LTC1149-3.3 LTC1149-5
✔
✔
✔
✔
✔
✔
✔
IOUT
1148_4.eps
TM
Figure 4. Burst Mode operation limits gate
charge and DC supply current at low output
currents; synchronous switching limits losses
at high output currents
LTC1148 and LTC1149 change to
Burst ModeTM operation as the output
current drops.
The continuous mode operation is as
follows: The external P-channel
MOSFET switch is turned on at the end
of the off-time and turned off when the
inductor current has ramped up to the
current-comparator threshold. During
the off-time, the N-channel MOSFET
synchronous switch is turned on to reduce the dissipation that would otherwise occur in the diode. At the end of
the constant off-time, the P-channel
MOSFET is again turned on and the
cycle repeats. Adaptive anti-shootthrough circuitry prevents simultaneous conduction of the two MOSFETs
regardless of their relative sizes, and
gate resistors are neither required nor
recommended.
Burst ModeTM Operation—the
Low-Current Efficiency Saver
The LTC1148/LTC1149 burst operating mode is automatically invoked
when the current required by the load
is less than the minimum current supplied by continuous operation. In Burst
ModeTM operation the output voltage is
regulated via a hysteretic comparator
which, when tripped, shuts down both
MOSFET drivers and much of the control circuitry to conserve DC supply current. From the time the comparator
trips, until the lower comparator
threshold is reached, the load current
is supplied entirely by the charge stored
Linear Technology Magazine • February 1993
in the output capacitor. When the output capacitor discharges to the lower
threshold, the main loop again turns on
briefly at a low current level to recharge
the capacitor. This cycle repeats at a
progressively slower rate as the output
current is reduced.
The LTC1148/LTC1149
switching regulators not only
beat both linear regulators
and other switching
regulators in efficiency,
but also offer superior
dropout performance
Figure 4 shows how the efficiency
losses in a typical LTC1148 regulator
are apportioned as a result of the
action of Burst ModeTM operation. The
gate-charge loss deserves special
attention, since it is responsible for
much of the efficiency lost in the midcurrent region. Each time a MOSFET
gate is switched from low to high to low
again, a packet of charge, dQ, moves
from VIN to ground. The resulting
dQ/dt is a current out of VIN that is typically much larger than the DC supply
current. If Burst ModeTM operation was
not employed at low output currents,
the gate-charge loss alone would be
enough to push efficiency down to unacceptable levels. With Burst ModeTM
operation, the DC supply current
represents the lone (and unavoidable)
loss component, which continues to become a higher percentage as output
current is reduced.
Superb Start-up Manners
When starting a load from ground or
recovering from a short circuit, the
LTC1148 and LTC1149 offer superb
control of inductor current, with no
voltage overshoot when the regulated
voltage is reached. This is accomplished
by making the off-time proportional to
the output voltage as the output capacitor charges. When VOUT = 0, the off-time
is lengthened to retain tight control of
the peak inductor current at the very
low duty cycles required. As the output
voltage increases, the off-time is progressively shortened until it reaches the
normal operating point. The result is
the typical start-up behavior shown in
Figure 5.
Line regulation can be just as vital
as load regulation in battery-operated
systems. This is because plugging in a
wall adapter often increases the regulator input voltage to over double the
battery voltage. The LTC1148/
LTC1149 can handle these large steps
with no effect on the output voltage.
VOUT
1V/DIV
IIND
1A/DIV
1148_5.eps
Figure 5. When starting up, the LTC1149/
LTC1149 supply a constant current until the
regulated voltage is reached. Note the lack of
overshoot
11
DESIGN FEATURES
VIN
5.2V-12V
Very Low Dropout Operation
The LTC1148/LTC1149 switching
regulators not only beat both linear
regulators and other switching regulators in efficiency, but also offer superior dropout performance. As the input
voltage on the LTC1148 or LTC1149
drops, the loop extends the on-time for
the switch (remember, the off-time is
constant), thereby keeping the inductor ripple current constant. When VIN –
VOUT drops below approximately 1.5V,
tOFF is reduced to compensate for the
decreasing frequency. Ultimately, the
LTC1148 and LTC1149 regulators drop
out smoothly, with the P-channel
MOSFET switch turning on at 100%
duty cycle or DC.
With the switch turned on at a 100%
duty cycle, the dropout voltage is limited only by the load current multiplied
by the total DC resistance of the
MOSFET, the inductor, and the current-sense resistor. For example, the
low dropout 5V regulator shown in
Figure 6 has a total resistance in dropout of less than 200mΩ. This gives it a
dropout voltage of less than 200mV at
1A output current. Furthermore, when
the regulator is in a dropout condition,
the ground current is limited to the DC
supply current (1.5mA for the LTC1148
and 2.5mA for the LTC1149) independent of load current.
+
+
1µF
100µF
VIN
Si9430
P DRIVE
0V = NORMAL
>1.5V = SHUTDOWN
L1
62µH
SHUTDOWN
RS
50mΩ
VOUT
5V/2A
LTC1148-5
SENSE+
1000pF
ITH
3300pF
SENSE–
+
Si9410
CT
1k
470pF
S GND
330µF
1N5817
N DRIVE
P GND
1148_6.eps
RS = SL-1R050J, KRL BANTRY (603) 668-3210
L1 = CTX62-2-MP, COILTRONICS (305) 781-8900
Figure 6. The LTC1148/LTC1149 architecture provides 100% duty cycle, allowing very low
dropout operation. This LTC1148-5 circuit supports a 1A load with the input voltage only
200mV above the output
VIN
48V
+
1N4148
CAP
VIN
P GATE
0.068µF
VCC
P DRIVE
LTC1149-5
SHUTDOWN1
0V = NORMAL
>2V = SHUTDOWN
SHUTDOWN2
3300pF
1k
CT
680pF
IRFR9024
0.047µF
VCC
+
1µF
100µF
100V
MBR380
L1
75µH
RS
39mΩ
VOUT
5V/2.5A
SENSE+
1000pF
ITH
SENSE–
CT
N GATE
S GND P,R GNDS
IRFR024
+
150µF
OS-CON
1148_7.eps
High Voltage Capability
The LTC1149 versions offer an operating-voltage capability far higher
than those found in other BiCMOSbased technologies. This is because
the LTC1149 is actually a hybrid of a
BiCMOS controller chip and a 60V
bipolar companion chip that adds the
low dropout regulator and P-channel
drive level-shift circuits shown in Figure 3. The resulting combination,
available in a narrow 16-lead SOIC
package, extends LTC1148-like performance to input voltages as high as
48V (60V absolute maximum). This
allows the LTC1149 to be used in such
applications as automotive and distributed power, as well as portable
equipment operating off high-voltage
AC adapters.
12
RS = SL-1R039J, KRL BANTRY (603) 668-3210
L1 = CTX75-2-MP, COILTRONICS (305) 781-8900
Figure 7. The LTC1149 extends high efficiency operation to high input voltages. This DC-to-DC
converter achieves 87% efficiency in dropping 48V to 5V
A 48V to 5V DC-to-DC converter
that achieves 87% efficiency at 1A load
current is shown in Figure 7. In this
application the synchronous switch
plays a vital role, since the P-channel
switch duty cycle is only a little more
than 10%, meaning that the N-channel synchronous switch conducts
nearly 90% of the time in continuous
mode. Fortunately, low RDS(ON) N-channel MOSFETs that reduce losses to the
point that heat sinking is not required,
even during continuous short-circuit
operation, are readily available. The
small Schottky diode shown across the
N-channel MOSFET conducts only in
the dead-time between the conduction
of the two power MOSFETs and provides the highest possible operating
efficiency in continuous mode.
Conclusion
The LTC1148/LTC1149 family of
synchronous, stepdown switching
regulators offers breakthroughs in
the areas of low-current operating
efficiency, high-current operating efficiency, very low dropout, and wide
input voltage compliance. This performance will be vital to extending
battery life in the next generation of
portable electronics.
Linear Technology Magazine • February 1993
DESIGN IDEAS
A Simple, Efficient Laser Power Supply
by Jim Williams
Helium-neon lasers, used for a variety of tasks, are difficult loads for a
power supply. They typically need almost 10 kilovolts to start conduction,
although they require about 1500 volts
to maintain conduction at their specified operating currents. Powering a
laser usually involves some form of
start-up circuitry to generate the initial
breakdown voltage and a separate supply for sustaining conduction. Figure
1’s circuit considerably simplifies driving the laser. The start-up and sustaining functions have been combined
into a single, closed-loop current source
with over 10 kilovolts of compliance.
When power is applied, the laser does
not conduct and the voltage across the
190Ω resistor is zero. The LT1170
switching regulator FB pin sees no
feedback voltage, and its switch pin
(VSW) provides full duty cycle pulsewidth modulation to L2. Current flows
from L1’s center tap through Q1 and
Q2 into L2 and the LT1170. This current flow causes Q1 and Q2 to switch,
alternately driving L1. The 0.47µF capacitor resonates with L1, providing a
boosted, sine-wave drive. L1 provides
substantial step-up, causing about
3500 volts to appear at its secondary.
The capacitors and diodes associated
with L1’s secondary form a voltage
tripler, producing over 10 kilovolts
across the laser. The laser breaks down
and current begins to flow through it.
The 47kΩ resistor limits current and
isolates the laser’s load characteristic.
The current flow causes a voltage to
appear across the 190Ω resistor. A
filtered version of this voltage appears
at the LT1170 FB pin, closing a control
1800pF
10kV
0.01
5kV
loop. The LT1170 adjusts its pulsewidth drive to L2 to maintain the FB pin
at 1.23 volts, regardless of changes in
operating conditions. Hence, the laser
sees constant current drive, in this
case 6.5mA. Other currents are obtainable by varying the 190Ω value. The
IN4002 diode string clamps excessive
voltages when laser conduction first
begins, protecting the LT1170. The 10µF
capacitor at the VC pin frequency compensates the loop and the MUR405
maintains L1’s current flow when the
LT1170 VSW pin is not conducting. The
circuit will start and run the laser over a
9–35 volt input range with an electrical
efficiency of about 75%.
References
1. Williams, Jim. “Illumination Circuitry for Liquid
Crystal Displays” Linear Technology Application
Note 49, August 1992.
47k
5W
1800pF
10kV
8
11
L1
1
4
5
HV DIODES
3
2
0.47µF
+
LASER
2.2µF
Q2
Q1
150Ω
L2
150µH
MUR405
VIN
9V TO 35V
VSW
10k
VIN
+
FB
LT1170
2.2µF
VC
+
0.1µF
190Ω
1%
GND
10k
10µF
VIN
1N4002
(ALL)
HV DIODES = SEMTECH-FM-50
0.47µF = WIMA 3X 0.15µF TYPE MKP-20
Q1, Q2 = ZETEX ZTX-849
L1 = COILTRONICS CTX02-11128
L2 = COILTRONICS CTX150-3-52, COILTRONICS (305) 781-8900
LASER = HUGHES 3121H-P
LASER_1.eps
Figure 1. Laser power supply
Linear Technology Magazine • February 1993
13
DESIGN IDEAS
Switching Regulator Provides ±15V
Output from an 8V – 40V Input,
without a Transformer
by Brian Huffman
Many systems require that ±15V
supplies for analog circuitry be derived
from an input voltage that may be
above or below the 15V output. The
split-supply requirement is usually fulfilled by a switcher with a multiplesecondary transformer, or by multiple
switchers. An alternative approach,
shown in Figure 1, uses an LT1074
switching regulator IC, two inductors,
and a “flying” capacitor to generate a
dual-output supply that accepts a wide
range of input voltages. This solution is
particularly noteworthy because it uses
only one switching regulator IC and
does not require a transformer. Inductors are preferred over transformers
because they are readily available and
more economical.
The operating waveforms for the
circuit are shown in Figure 2. During
the switching cycle, the LT1074’s VSW
pin swings between the input voltage
(VIN) and the negative output voltage
(–VOUT). (The ability of the LT1074’s VSW
pin to swing below ground is unusual—
most other 5-pin-buck switching regulator ICs cannot do this.) Trace A shows
the waveform of the VSW pin voltage,
and Trace B is the current flowing
through the power switch.
While the LT1074 power switch is
on, current flows from the input voltage source through the switch, through
capacitor C2 and inductor L1 (Trace
C), and into the load. A portion of the
switch current also flows into inductor
L2 (Trace D). This current is used to
recharge C2 and C4 during the switchoff time to a potential equal to the
positive output voltage (+VOUT). The
current waveforms for both inductors
occur on top of a DC level.
The waveforms are virtually identical
because the inductors have identical
values, and because the same voltage
potentials are applied across them during the switching cycles.
C2
470µF, 25V
5
VR1
LT1074
VIN
8V-40V
GND
3
FB
VC
1
L2
50µH
+
D1
MUR410
C6
0.01µF
R4
20k
2
R1
3.3k
C1
1000µF
50V
+VOUT
+15V, 0.5A
+
VSW 4
VIN
L1
50µH
C7
0.01µF
R5
20k
C5
0.01µF
+VOUT = 2.21V* (1 + R2/R3) (EQ. 1)
+VOUT = –VOUT
C1 = NICHICON UPL1H102MRH
C2, C3, C4 = NICHICON UPL1E471MPH
D1, D2 = MOTOROLA MUR410
L1, L2 = COILTRONICS CTX50-2-52 (305) 781-8900
R2
7.50k
1%
C3
470µF
25V
+
C4
470µF
25V
+
R3
1.30k
1%
D2
MUR410
–VOUT
–15V, 0.5A
1074_1.eps
Figure 1. Schematic diagram for ±15V version
14
A = 20V/DIV
VSW
B = 2A/DIV
ISW, IC1
C = 1A/DIV
IL1, IC3
D = 1A/DIV
IL2
E = 1A/DIV
ID1, IC3
F = 1A/DIV
ID2, IC4
G = 1A/DIV
IC2
5µs/DIV
Figure 2. LT1074 switching waveforms
When the switch turns off, the current in L1 and L2 begins to ramp
downward, causing the voltages across
them to reverse polarity and forcing the
voltage at the VSW pin below ground.
The VSW pin voltage falls until diodes
D1 (Trace E) and D2 (Trace F) are
forward biased. During this interval,
the voltage on the VSW pin is equal to a
diode drop below the negative output
voltage (–VOUT). L2’s current then circulates between both D1 and D2, charging C2 and C4. The energy stored in L1
is used to replace the energy lost by C2
and C4 during the switch-on time.
Trace G is capacitor C2’s current waveform. Capacitor C4’s current waveform
(Trace F) is the same as diode D2’s
current less the DC component. Assuming that the forward voltage drops
of diodes D1 and D2 are equal, the
negative output voltage (–VOUT) will be
equal to the positive output voltage
(+VOUT). After the switch turns on again,
the cycle is repeated.
Linear Technology Magazine • February 1993
DESIGN IDEAS
Linear Technology Magazine • February 1993
15.3
75
15.2
70
EFFICIENCY (%)
15.1
–VOUT (V)
+IOUT = 0.5A
15.0
14.9
+IOUT = –IOUT
14.8
65
60
55
14.7
14.6
50
0 0.05 0.1 0.15 0.2 0.25 0.3 0.35 0.4 0.45 0.5
–IOUT (A)
0
5
10
15
20
25
30
35
40
INPUT VOLTAGE (V)
1074_3.eps
1074_4.eps
Figure 4. ±15V efficiency characteristics with
0.5V common load
Figure 3. –15V output dropout characteristics
75
5.7
5.6
70
5.5
EFFICIENCY (%)
–VOUT (V)
5.4
5.3
+IOUT = 1A
5.2
5.1
5.0
60
55
+IOUT = – IOUT
4.9
65
50
4.8
0
0
0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0
5
10
15
20
25
30
1074_6.eps
1074_5.eps
Figure 6. ±5V efficiency characteristics with 1A
common load
Figure 5. –5V output dropout characteristics
C2
680µF, 16V
VSW 4
VIN
VR1
LT1074
VIN
8V-40V
GND
3
FB
VC
1
L2
50µH
+
+VOUT
+5V, 1A
L1
50µH
D1
MBR360
C6
0.01µF
R4
20k
2
R1
2k
C1
1000µF
50V
40
INPUT VOLTAGE (V)
–IOUT (A)
5
35
+
Figure 3 shows the excellent regulation of the negative output voltage for
various output currents. The negative
output voltage tracks the positive supply (+VOUT) within 200mV for load variations from 50mA to 500mA. Negative
output load current should not exceed
the positive output load by more than a
factor of 4; the imbalance causes loop
instabilities. For common load conditions the two output voltages track
each other perfectly.
Another advantage of this circuit is
that inductor L1 acts as both an energy
storage element and as a smoothing
filter for the positive output (+VOUT).
The output ripple voltage has a triangular waveshape whose amplitude is
determined by the inductor ripple current (see Trace C of Figure 2) and the
ESR (Effective Series Resistance) of the
output capacitor (C3). This type of
ripple is usually small, so a post filter is
not necessary.
Figure 4 shows the efficiency for a
0.5A common load at various input
voltages. The two main loss elements
are the output diodes (D1 and D2) and
the LT1074 power switch. At low input
voltages, the efficiency drops because
the switch’s saturation voltage becomes
a higher percentage of the available
input supply.
The output voltage is controlled by
the LT1074 internal error amplifier.
This error amplifier compares a fraction of the output voltage, via the R2–
R3 divider network shown in Figure 1,
with an internal 2.21V reference voltage and then varies the duty cycle until
the two values are equal. The RC network (R1 and C5 in Figure 1) connected
to the VC pin along with the R4/R5 and
C6/C7 network provides sufficient compensation to stabilize the control loop.
Equation 1 can be used to determine
the output voltage.
Figure 5 shows the circuit's –5V
load-regulation characteristics, and
Figure 6 shows its efficiency.
Refer to the schematic diagram in
Figure 7 for modified component values to provide ±5V at 1 Amp.
C7
0.01µF
R5
20k
C5
0.033µF
+VOUT = 2.21V* (1 + R2/R3) (EQ. 1)
+VOUT = – VOUT
C1 = NICHICON UPL1H102MRH
C2, C3, C4 = NICHICON UPL1C681MPH
D1, D2 = MOTOROLA MBR360
L1, L2 = COILTRONICS CTX50-2-52 (305) 781-8900
R2
2.80k
1%
C3
680µF,
16V
+
R3
2.21k
1%
D2
MBR360
Figure 7. Schematic diagram for ±5V version
C4
680µF, 16V
+
– VOUT
–5V, 1A
1074_1a.eps
15
DESIGN IDEAS
A Twelve-Bit, Micropower
Battery-Current Monitor
by Sammy Lum
Introduction
IN
ated by using the other half of the
LTC1047 as a comparator. Its trip point
has been set to 5V plus the dropout
voltage of the LT1121. Because data is
transmitted serially to and from the
microprocessor or microcontroller, this
current-monitor circuit can be located
close to the battery.
The Battery-Current Monitor
The battery voltage of 6V to 12V is
regulated down to 5V by the LT1121
micropower regulator. A sense resistor
of 0.05Ω is placed in series with the
battery to convert the battery current
to a voltage. Full scale is designed for
2A, giving a resolution of 0.5mA with
the 12-bit ADC. The LTC1047 amplifies the voltage across the sense resistor by 25 V/V. This goes through an RC
lowpass filter before being fed into the
input of the LTC1297. The RC filter
serves two functions. First, it helps
band-limit the input noise to the ADC.
Second, the capacitor helps the
LTC1047 recover from transients due
to the switching input capacitor of the
LTC1297. The LT1004 provides the
full-scale reference for the ADC. A low
battery detection circuit has been cre-
10000
1000
10
1
0.001
0.01
0.1
1
10
100
fSAMPLE (kHz)
1297_2.eps
Figure 2. Power-supply current versus
sampling frequency for the LTC1297
0.33µF
SHDN
100kΩ
1N4148
510kΩ
100
+5V
OUT
LT1121
GND
and the available charging current
from the regulator.
AVERAGE ICC (µA)
The LTC1297 forms the core of the
micropower battery-current monitor
shown in Figure 1. This 12-bit dataacquisition system features an automatic power shutdown that is activated
after each conversion. In shutdown
the supply current is reduced to 6µA,
typically. As shown in Figure 2, the
average power-supply current of the
LTC1297 varies from milliamperes to
a few microamperes as the sampling
frequency is reduced. This circuit
draws only 190µA from a 6V to 12V
battery when the sampling frequency
is less than 10 samples per second.
Wake-up time is limited by that required by the LTC1297 (5.5µs). For
long periods of inactivity, the circuit’s
supply current can be further reduced
to 20µA by using the shutdown feature
on the LT1121. More wake-up time is
required when using this mode of shutdown. It is usually determined by the
amount of capacitance in the circuit
0.1µF
6V-12V
LOAD
750kΩ
30.1kΩ
0.05Ω
2A FULL
SCALE
VCC
CS
–
1/2
10Ω
+IN
+ LTC1047
1µF
LTC1297
+
TO AND
FROM µP
22µF
TANT
CLK
–IN
DOUT
GND
VREF
LO BATTERY
470kΩ
0.1µF
–
LT1004-2.5
1/2
+LTC1047
20MΩ
1297_1.eps
Figure 1. A micropower battery-current monitor using the LTC1297 12-bit data acquisition system
16
Linear Technology Magazine • February 1993
DESIGN IDEAS
200mA Output, 1.5V-to-5V Converter
by Jim Williams
+
stored in the 500µF capacitor, producing the circuit’s DC output. The output-divider string is set up so the
LT1073 turns off when the circuit’s
output crosses about 4.5V. Under these
conditions the LT1073 can no longer
drive L1, but the LT1170 can. When
the start-up circuit turns off, the
LT1170 VIN pin has adequate supply
voltage and it can operate. There is
some overlap between start-up loop
turn-off and LT1170 turn-on, but this
has no detrimental effect. The start-up
loop must function over a wide range of
loads and battery voltages. Start-up
currents are about 1 ampere, necessitating attention to the LT1073’s saturation and drive characteristics. The
worst case is a nearly depleted battery
and heavy output loading. Figure 2
plots input/output characteristics for
the circuit. Note that the circuit will
start into all loads with VBATT = 1.2V.
1.5VIN
L1
20µH
220µF
+
22µF
Start-up is possible down to 1.0V at
reduced loads. Once the circuit has
started, the plot shows it will drive full
200mA loads down to VBATT = 0.6V (a
very dead battery). Figure 3 graphs
efficiency at two supply voltages over a
range of output currents. Performance
is attractive, although at lower currents circuit quiescent power degrades
efficiency. Fixed junction saturation
losses are responsible for lower overall
efficiency at the lower supply voltage.
References
1. Williams, Jim and Brian Huffman. “Some Thoughts
on DC–DC converters”, pages 13–17, “1.5V to 5V
Converters.” Linear Technology Corporation Application Note 29, October 1988.
MINIMUM INPUT VOLTAGE TO MAINTAIN VOUT = 5.0V
Some 1.5V powered systems, such
as two-way survival radios, remote,
transducer-fed data-acquisition systems, and the like, require much more
power than stand-alone IC regulators
can provide. Figure 1’s design supplies
a 5V output with 200mA capacity.
The circuit is essentially a flyback
regulator. The LT1170 switching
regulator’s low saturation losses and
ease of use permit high power operation and design simplicity. Unfortunately this device has a 3V minimum
supply requirement. Bootstrapping its
supply pin from the 5V output is possible, but requires some form of startup mechanism. The 1.5V powered
LT1073 switching regulator forms a
start-up loop. When power is applied,
the LT1073 starts, causing its VSW pin
to periodically pull current through L1.
L1 responds with high-voltage flyback
pulses. These pulses are rectified and
1.4
1.2
START
1.0
RUN
0.8
0.6
0.4
0.2
0
0 20 40 60 80 100 120 140 160 180 200 220
1N5823
OUTPUT CURRENT (mA)
5VOUT
VIN
+
VSW
500µF
3.74k*
CELL_2.eps
Figure 2. Input/output data for Figure 1
LT1170
FB
GND
VC
100
90
80
1k*
6.8µF
VIN
SW1
LT1073
SW2
VIN = 1.5V
70
60
VIN = 1.2V
50
40
30
FB
GND
EFFICIENCY (%)
1k
+
240Ω*
20
VOUT = 5.0V
10
0
L1 = COILTRONICS CTX20-5-52, COILTRONICS (305) 781-8900
* = 1% METAL FILM RESISTOR
0
CELL_1.eps
20 40 60 80 100 120 140 160 180 200
OUTPUT CURRENT (mA)
CELL_3.eps
Figure 1. 200mA output, 1.5V-to-5V converter
Linear Technology Magazine • February 1993
Figure 3. Efficiency versus operating point
for Figure 1
17
DESIGN IDEAS
LT1158 H-Bridge uses GroundReferenced Current Sensing
for System Protection
The LT1158 half-bridge motor driver
incorporates a number of powerful
protection features. Some of these,
such as its adaptive gate drive, are
dedicated in function. Others are open
to a variety of uses, depending upon
application requirements. The circuit
shown in Figure 1 takes advantage of
the wide common-mode input range of
the LT1158’s FAULT comparator to
perform ground-referenced current
sensing in an H-bridge motor driver.
By using ground-referenced sensing,
protection can easily be provided
against overloaded, stalled, or shorted
motors. For overloads and stalls, the
circuit becomes a constant-current
chopper, regulating the motor’s armature current to a preset maximum
value. For shorted loads, the circuit
protects itself by operating at a very
low duty cycle until the short is cleared.
Setting Up For GroundReferenced Sensing
The circuit of Figure 1 is essentially
a straightforward LT1158 H-bridge, of
the “sign/magnitude” type. (See the
LT1158 data sheet for a description of
component functions.) In many
LT1158 applications, a current-sense
resistor is placed in each upper
MOSFET source lead. This circuit, however, senses the IR drop across one
resistor (R1) common to the sources of
both lower MOSFETs. In Figure 1,
U1’s FAULT output activates the constant-current protection mode (for
overloads and stalls), and U2’s FAULT
output indicates a shorted load. Hence,
given a maximum continuous motor
current of 15A, R1’s value is easily
determined: VSENSE(+) of U1 must exceed VSENSE(–) by the LT1158’s internal
threshold of 110mV in order for FAULT
to go low. 15A x R1 = 0.110V, so R1 =
(0.110V/15A) ≅ 7.5mΩ. The FAULT pin
18
of U2 should go low when IR1 is 24A, so
a 1.6:1 voltage divider is added at U2’s
SENSE(+) input. R2, R3, C1, and C2
filter any switching spikes which appear across R1.
Closing the Loop on Overloads
If the motor is overloaded or stalled,
its back EMF will drop, causing the
armature current to increase at a rate
determined primarily by the motor’s
inductance. Without protection, this
current could rise to a value limited
only by supply voltage and circuit resistance. The necessary protection is
provided via the feedback loop formed
by U1’s FAULT output, U3A, U4B, and
U4D. When IR1 exceeds 15A, the FAULT
pin of U1 conducts, triggering the 40µs
monostable U3A. The “Q” output of
U3A in turn forces the outputs of U4B
and U4D to a logic low state, turning off
Q1 or Q3, and turning on both Q2 and
Q4. For the time during which U3A’s
“Q” output is high, the motor current
decays through the path formed by the
motor’s resistance, plus the “on” resistance of Q2 and Q4 in series. In this
application, turning both lower MOSFETs on is preferable to forcing all four
MOSFETs off, as it provides a lowresistance recirculation path for the
motor current. This reduces motor and
supply ripple currents, as well as MOSFET dissipation. At the end of U3A’s
40µs timeout, the H-bridge turns on
again. If the overload still exists, the
current quickly builds up to the U1
“FAULT” trip point again, and the 40ms
timeout repeats. This feedback loop
holds the motor current approximately
constant at 15A for any combination of
supply voltage and duty cycle that
would otherwise cause an excess current condition. When the motors current draw falls below 15A, the circuit
resumes normal operation.
by Peter Schwartz
Opening the Loop on Shorts
In the event of a short across the
motor terminals, the current through
the H-bridge rises faster than the U1/
U3A loop can regulate it. This could
easily exceed the safe operating area
limits of the MOSFETs. The solution is
simple: when the FAULT comparator
of U2 detects that IR1 ≥ 24A, monostable
U3B is triggered. The “Q’” output of
U3B will then hold the ENABLE line of
the two LT1158s “low” for 10ms, resulting in a rapid shutdown and a very
low duty cycle. After the 10ms shutdown interval, U3B’s “Q’” output will
return high, and the bridge will be reenabled. If the motor remains shorted,
U3B is triggered again, causing another 10ms shutdown. When the short
is cleared, circuit operation returns to
that described above.
A Final Note
As a class, sign/magnitude H-bridge
systems are susceptible to MOSFET
and/or motor damage if the motor
velocity is accelerated rapidly, or the
state of the DIRECTION line is
switched while the motor is rotating.
This is especially true if the motor/
load system has high inertia. The circuit of Figure 1 is designed to provide
protection under these conditions: the
motor may be commanded to accelerate and to change direction with no
precautions. For the case of deceleration, however, it’s generally best to use
a controlled velocity profile. If a specific application requires the ability to
operate with no restrictions upon the
rate of change of duty cycle, there are
straightforward modifications to Figure 1 which allow this. Please contact
the factory for more information.
Linear Technology Magazine • February 1993
Linear Technology Magazine • February 1993
DIRECTION (“SIGN”)
PWM (“MAGNITUDE”)
+5V
U3A
+5V
4.7k
4.7k
2
3
REXT/CEXT
74HC221 Q 13
14
CEXT
1
A
4
2
Q
B
3
CLR
15
7
GND
5
FAULT
4
ENABLE
6
INPUT
3
BIAS
0.01µF
39k
10µF
16
1
12
U4A
74HC02
1
1N4148
R2
100
R3
100
33
11
12
8
9
5
6
U4D
74HC02
U4C
74HC02
U4B
74HC02
13
10
4
M
DC MOTOR
(15A CONT)
TWISTED-PAIR
MBR170
33
33
100
270
160
0.1µF
C2
0.01µF
*** PULLDOWN FOR “ENABLE” LINE IN CASE
+5V IS NOT PRESENT.
** DIODE SHOWN IS THE MOSFET’S INTEGRAL
DRAIN-BODY DIODE.
* LOW ESR CAPACITORS (SPRAGUE 673D, ETC.)
Q1-Q4 MOUNTED ON HEAT SINK
R1
0.0075
3W
Q2
Q4
IRFZ44** IRFZ44**
TWISTED-PAIR
MBR170
33
Q1
Q3
IRFZ44** IRFZ44**
1N4148
Figure 1. H-bridge motor driver with ground referenced current sensing
4.7k
0.1µF
1N4148
C1
0.01µF
1N4148
11
SENSE–
SENSE+
9
B GATE DR
8
B GATE FB
15
T GATE DR
14
T GATE FB
13
T SOURCE
BOOST
BOOST DR
LT1158
470µF*
+
1000pF
+
V+
V+
U1
+
47k
+5V
10
2
470µF*
+24V
11
12
+5V
0.047µF
SENSE–
SENSE+
9
B GATE DR
8
B GATE FB
15
T GATE DR
14
T GATE FB
13
T SOURCE
2
V+
10
V+
+5V
+
REXT/CEXT
74HC221 Q 5
6
CEXT
9
A
12
10
Q
B
11
CLR
7
7
U3B
220k
GND
10µF
0.01µF
5
FAULT
4
ENABLE
6
INPUT
3
BIAS
U2
LT1158
1
BOOST DR
16
BOOST
10k***
47k
+5V
DESIGN IDEAS
19
DESIGN IDEAS
Multi-Output Three-Watt Power Supply
Operates from Two AA Cells Steve Pietkiewicz
Portable, battery-operated microprocessor systems often have power supply requirements beyond what existing
low-voltage IC switching regulators can
deliver. Also, a multiplicity of voltages
are usually required to supply subsystems such as main logic, flash
memory VPP supply, LCD contrast,
and modem. Previous approaches to
this problem used a separate DC–DC
converter circuit for each output, increasing system cost and complexity.
The approach described in this article
combines the classic multi-output flyback topology with an LT1110 micropower, low-voltage DC–DC
converter. The negative-to-positive topology provides for operation from a
guaranteed beta of 100. Saturation
voltage of the device with a forced beta
of 50 is 250mV at 5A IC. Base drive for
Q2 is provided by Q1, whose drive is
supplied by the SW1 pin of the LT1110.
Q2 is turned off by Q3, whose base is
AC coupled to the SW1 pin by the
2200pF capacitor. Q3, a 2N2369, is a
very fast device; it pulls the base charge
out of Q2 in 50ns. Q2 is kept off by the
2200Ω base-emitter resistor R4. The
primary winding of the trans–former,
L1, functions as the regulated (main)
secondary winding during the flyback
phase. The voltage across L1 is forced
to 5V during this phase. Hence, other
indirectly regulated voltages can be
achieved by the use of secondary wind-
1.8V to 7.5V input, a key provision for
systems that must operate with either
an AC adaptor or two AA cells. Micropower circuitry reduces quiescent
current to 400µA no-load. The circuit
can provide 5V at up to 400mA, 12V at
60mA, +28V at 2mA and –5V at 50mA
from an input voltage as low as 2.0V.
The LT1110 micropower, Burst
ModeTM DC–DC converter IC functions
as the controller in the circuit (Figure
1). The LT1110 toggles its SW1 pin
when the voltage at its FB pin drops
below 220mV. The power device in the
circuit is Q2, a Zetex ZTX-849. This
remarkable device, which comes in a
small TO-92-type package, can handle
collector currents exceeding 6A with
D5
1N5818
–5V/
50mA
T1*
R5**
20mΩ
L4
R8
100
2 AA
CELLS
+
8
Q6
2N3906
R6
432k
1%
FB
GND
5
R7
21.5k
1%
*T1 = TDK EPC-15 CORE, PC40 MATERIAL, 6 MIL GAP OR
FERROXCUBE EFD 15 CORE, 3C85 MATERIAL, 6 MIL GAP
L1 = 6T #20
L2 = 7T #32
L3 = 16T #38
L4 = 6T #34
**R5 = DALE LVR-3
D3
1N5818
L3
+
IL
Q5
2N3904
IC1
LT1110
C1
150µF
OS-CON
+
R2
220
Q4
2N3904
1
VIN
C7
D4
1N4148
R9
4700
2
+
Q1
2N4403
R10
150
SW1
SW2
4
L1
L2
D2
1N5821
C6
4.7µF
+12V/
60mA
C5
33µF
+5V/
400mA
R1
220
3
+
R3
18
Q3
2N2369
R11
2200
C3
47µF
+
C4
47µF
Q2
ZETEX
ZTX-849
C2
2200pF
D1
1N4148
+28V/
2mA
R4
2200
ZETEX: (516) 543-7100
= HEAVY, HIGH CURRENT TRACES
1110_1.eps
C1, C3, C4, C5 = SANYO OS-CON
Figure 1. Multi-output power supply delivers +5V at 400mA, +12V at 60mA, –5V at 50mA and +28V at 2mA from 2AA cells
20
Linear Technology Magazine • February 1993
DESIGN IDEAS
ings with appropriate turns ratios.
The 12V output does not posses 5%
regulation from zero to full load, but a
micropower, linear, low-dropout regulator such as the LT1121 can be used
to achieve the desired voltage regulation. Negative outputs can be generated merely by reversing the phasing
of additional secondary windings, as
is done with L4 to obtain –5V output.
Feedback is accomplished by the levelshift network comprising Q6 and R6.
Q6’s collector is fed into R7, closing
the loop.
Switch-current sensing and control is essential when throwing lots of
amperes around. Variations in VIN
and tON due to manufacturing spread
can result in large peak current
changes if sensing and control are not
implemented. Many Burst ModeTM
regulators contain no provision for
current sensing, but the LT1110 is an
exception. The LT1110 switch will
turn off when the voltage at the IL pin
reaches 600mV less than the voltage
at the VIN pin. A 600mV shunt would
reduce system efficiency severely in a
2.0V input converter, so a pre-bias
voltage drop is developed by current
source Q4–Q5 flowing through resistor R8. Approximately 480mV is developed across R8, reducing the drop
across sense resistor R5 to 120mV.
This voltage drop represents 6% of
the 2V input, causing some loss of
efficiency, but the current sense function allows operation with inputs as
high as 7.5V.
Bypass capacitor C1 should be
placed close to the DC–DC converter
circuitry. The low-ESR OS-CON type
should be used. An inexpensive, highESR unit can result in poor efficiency.
The main 5V output capacitors should
also be OS-CON types. The peak cur-
5.5
6
OUTPUT
5.0
OUTPUT
5
BATTERY/OUTPUT VOLTAGE (V)
BATTERY/ OUTPUT VOLTAGE (V)
rent into these units is over 4A.
Skimping on output capacitors can
result in costly field failures. High
peak currents also necessitate careful printed-circuit layout. The highcurrent paths (highlighted in Figure
1) should be made extra-wide and as
short as possible.
Efficiency for the circuit is approximately 70% over an input range of 2
to 3.2V and .5W to 2W total output
power. The circuit will supply a 5V,
100µA “sleep” mode load for over 3
months from a pair of alkaline AA
cells (Figure 2). A 5V, 1mA load lasts
28 days with alkalines. However,
NiCad cells are recommended for the
power source, as the relatively high
internal impedance of the alkaline
cells deliver only 8% more operating
time than a pair of 600mAHr NiCad
cells when delivering 5V at 200mA
load, as detailed in Figure 3.
ILOAD = 100µA
4
BATTERY
3
2
1
4.5
4.0
NiCAD
ALKALINE
3.5
3.0
2.5
BATTERY
2.0
1.5
1.0
0
0
20
40
60
80
100
120
5 10 15 20 25 30 35 40 45 50 55 60
TIME (MINUTES)
DAYS
1110_2.eps
Figure 2. Sleep mode lifetime, 5V output, 100µA load current
Linear Technology Magazine • February 1993
0
1110_.3eps
Figure 3. Battery lifetime for ILOAD = 200mA
21
DESIGN IDEAS
LTC1157 Switch
for 3.3V PC Card Power
Computers designed to accept PC
cards—plug-in modules specified by
the Personal Computer Memory Card
International Association (PCMCIA)—
have special hardware features to accommodate these pocket-sized cards.
PCMCIA-compliant cards require power
management electronics which conform to the height restrictions of the
three standard configurations: 3.3mm,
5mm and 10.5mm. These height limitations dramatically reduce the available options for power management on
the card itself. For example, high-efficiency switching regulators to convert
the incoming 5V down to 3.3V for the
on-card 3.3V logic require relatively
large magnetics and filter capacitors,
which are not always available in packaging thin enough to meet the tight
height requirements.
One possible approach to the problem of supplying power to a 3.3V PC
by Tim Skovmand
card is to switch the input supply
voltage from 5V to 3.3V after the card
has been inserted and the attribute
ROM has informed the computer of
the card’s voltage and current
requirements. The switching regulator, housed in the computer, switches
the power supplied to the connector
from 5V to 3.3V.
A window comparator and ultra-low
drop switch on the PC card, Q1 in
Figure 1, closes after the supply voltage drops from 5V to 3.3V, ensuring
that the sensitive 3.3V logic on the card
is never powered by more than 3.6V or
less than 2.4V. A second switch, Q2, is
provided on the card to interrupt power
to 3.3V loads that can be idled when
not in use.
The built-in charge pumps in the
LTC1157 drive the gates of the low
RDS(ON) N-channel MOSFETs to 8.7V
when powered from a 3.3V supply.
(P-channels cannot be used at 3.3V
because they do not have guaranteed
RDS(ON) with VGS < 3.3V.) The LT1017
and the LTC1157 are both micropower
and are supplied by a filter, R5 and C2,
which holds the supply pins high long
enough to ensure that the MOSFET
gates are fully discharged immediately
after the card is disconnected from the
power supply. A large bleed resistor,
R6, further ensures that the high-impedance gate of Q1 is not inadvertently
charged-up when the card is removed
or when it is stored.
All of the components shown in
Figure 1 are available in thin, surfacemount packaging and occupy a very
small amount of surface area. Further,
the power dissipation is extremely low
because the LTC1157 and LT1017 are
micropower and the MOSFET switches
are very low RDS(ON).
5V
3.3V
R5
510
ATTRIBUTE
ROM
R1
150k
1%
R4
100k
3
2
R2
49.9k
1%
5
6
R3
100k
1%
+
+
Q1
MTD3055EL
C2
10µF
6.3V
R6
5.1M
8
1/2
LT1017
1
IN1
–
VS
G1
Q2
MTD3055EL
SENSITIVE
3.3V
LOGIC
LTC1157
IN2
+
1/2
LT1017
–
LT1004-1.2
G2
GND
7
SENSITIVE
3.3V
LOGIC
4
C1
0.1µF
SW ON/OFF
FROM µP
1157_1.eps
Figure 1. 3.3V PCMCIA card power switching
22
Linear Technology Magazine • February 1993
DESIGN
IDEAS
NEW DEVICE
CAMEOS
New Device Cameos
LT1116 12ns, Single-Supply,
Ground-Sensing Comparator
The LT1116 is a high-speed (12ns)
comparator capable of sensing signals
down to ground while operating from a
single +5V supply rail. The comparator
can also operate from split ±5V supply
rails, where the input common-mode
range extends from 2 volts below the
positive rail to the negative supply rail.
The LT1116 is pin-compatible with
the industry standard LT1016. Like
the LT1016, the LT1116 is stable
through its output-transition region,
which makes it easy to use over a wide
range of operating conditions. The
device’s complementary output stages
provide active drive in both directions
for increased speed when driving TTL
logic or passive loads. The LT1116 has
a latch pin for synchronizing or retaining data. Latch setup and hold times
are typically 2ns—commensurate with
the device’s propagation delay.
The LT1116’s tight offset (1mV typical) and high gain specifications
(3000V/V typical) make it an ideal
choice for high-speed applications such
as zero-crossing detectors, triggers,
sampling circuits, A/D converters, current sensing for switching regulators,
and line receivers for data communication. Linear Technology’s Application
Note 13 describes practical design techniques for high-speed comparators.
The LT1116 is available in 8-lead
SOIC and 8-pin mini-DIP packages.
The LTC1255 Dual, 24V
High-Side MOSFET Driver
The LTC1255 dual, 24V high-side
gate driver is designed to drive two
standard N-channel power MOSFETs
in a high-side switch configuration.
The LTC1255 contains two independent, on-chip charge pumps so that
less expensive, lower RDS(ON) N-channel MOSFETs can be used in place of Pchannel switches. The charge pumps
require no external components and
have been designed to be very efficient.
All of the circuitry to drive, control,
and protect the power MOSFET and
Linear Technology Magazine • February 1993
load is provided by the LTC1255. The
input is compatible with both TTL and
CMOS logic families and the standby
current with the input switched off is
only 12 microamps from a 12V supply.
The quiescent current rises to 240
microamps with the switch turned on
and the charge pump producing 24V
from a 12V supply.
The MOSFET and load are protected
by a sense circuit that trips when an
over-current condition is detected at
the drain end of the power MOSFET. A
built-in 10-microsecond delay ensures
that the LTC1255 protection circuitry
is not false-triggered by transient load
or power supply conditions. A longer
RC delay can be added externally to
accommodate loads with large transient start-up current requirements,
such as lamps or DC motors.
The 9–24V operating range of the
LTC1255 makes it the ideal choice for
many automotive and industrial
applications, as well as 8–12 cell notebook-computer battery switching applications. The LTC1255 is available in
both 8-lead SO and 8-lead DIP packaging and is rated over both the industrial and commercial operating
temperature ranges.
LT1331, LT1341, and
LT1342 RS232 Transceivers
Three new RS232 transceivers expand LTC’s line of interface circuits for
PC-compatible applications. Each of
these transceivers contains three drivers and five receivers to support the
serial interface requirements of personal computers. All three feature
±10kV ESD protection on the RS232
line pins, operate to 120kbaud, and
contain on-chip charge-pump circuitry
to allow operation from standard logic
power supplies. Pin-outs are compatible with the LT1137, and each transceiver has low-power SHUTDOWN and
DRIVER DISABLE operating modes
to allow optimization of system
power consumption based upon signal
requirements.
The LT1341 features one low-power
receiver that remains active while the
circuit is in SHUTDOWN mode. Drawing only 60µA of power, the keep-alive
receiver may be used to monitor a data
line to control system wake-up. This is
especially useful in battery-operated
systems.
The LT1342 matches LT1137A performance with the addition of 3V logicinterface capability. The circuit is ideal
for systems with both 3V and 5V power
supplies. Power consumption is 12mA
from the 5V supply and 0.1mA from
the 3V power supply. In SHUTDOWN,
power consumption drops to near zero.
The LT1331 provides more flexible
power management features for mixed
5V/3V systems, and delivers RS562
level outputs when used in 3V-only
systems. Two power supply pins, VCC
and VL, are used to power the circuit.
Either may be used at 3V or 5V. The VCC
supply powers the charge pump and
driver circuits and may be turned
off in SHUTDOWN. With VCC = 5V, full
RS232 output levels are supported.
With VCC = 3.3V, outputs are at RS562
levels. The receivers are powered from
supply VL. VL current drain is 3mA with
all receivers active or 60µA in SHUTDOWN with one receiver active.
All three circuits are available in 28pin DIP, SOIC, and SSOP packages.
For further information on the
above or any other devices mentioned in this issue of Linear Technology, use the reader service card
or call the LTC literature-service
number: (800) 637-5545. Ask for
the pertinent data sheets and application notes.
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use.
Linear Technology makes no representation that
the circuits described herein will not infringe on
existing patent rights.
23
DESIGN IDEAS
DESIGN TOOLS
World Headquarters
Applications on Disk
Linear Technology Corporation
1630 McCarthy Boulevard
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Phone: (408) 432-1900
FAX: (408) 434-0507
NOISE DISK
This IBM-PC (or compatible) progam allows the user to
calculate circuit noise using LTC op amps, determine the
best LTC op amp for a low noise application, display the
noise data for LTC op amps, calculate resistor noise, and
calculate noise using specs for any op amp.
Available at no charge.
SPICE MACROMODEL DISK
This IBM-PC (or compatible) high density diskette contains
the library of LTC op amp SPICE macromodels. The
models can be used with any version of SPICE for general
analog circuit simulations. The diskette also contains working circuit examples using the models, and a demonstration
copy of PSPICETM by MicroSim.
Available at no charge.
Technical Books
1990 Linear Databook — This 1,440 page collection
of data sheets covers op amps, voltage regulators,
references, comparators, filters, PWMs, data conversion
and interface products (bipolar and CMOS), in both commercial and military grades. The catalog features well over
300 devices.
$10.00
1992 Linear Databook Supplement — This 1248 page
supplement to the 1990 Linear Databook is a collection of
all products introduced since then. The catalog contains full
data sheets for over 140 devices. The 1992 Linear Databook
Supplement is a companion to the 1990 Linear Databook ,
which should not be discarded.
$10.00
Linear Applications Handbook — 928 pages full of
application ideas covered in depth by 40 Application Notes
and 33 Design Notes. This catalog covers a broad range of
“real world” linear circuitry. In addition to detailed, systemsoriented circuits, this handbook contains broad tutorial
content together with liberal use of schematics and scope
photography. A special feature in this edition includes a 22page section on SPICE macromodels.
$20.00
Monolithic Filter Handbook — This 232 page book comes
with a disk which runs on PCs. Together, the book and disk
assist in the selection, design and implementation of the
right switched capacitor filter circuit. The disk contains
standard filter responses as well as a custom mode. The
handbook contains over 20 data sheets, Design Notes and
Application Notes.
$40.00
SwitcherCAD Handbook — This 144 page manual, including disk, guides the user through SwitcherCAD – a
powerful PC software tool which aids in the design and
optimization of switching regulators. The program can cut
days off the design cycle by selecting topologies, calculating operating points and specifying component values and
manufacturer's part numbers.
$20.00
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1993 Linear Technology Corporation/ Printed in U.S.A./20K
Linear Technology Magazine • February 1993