Linear Technology Magazine Circuit Collection, Volume II

Application Note 66
August 1996
Linear Technology Magazine Circuit Collection, Volume II
Power Products
Richard Markell, Editor
INTRODUCTION
Application Note 66 is a compendium of “power circuits”
from the first five years of Linear Technology. The objective
is to collect the useful circuits from the magazine into
several applications notes (another, AN67, will collect
signal processing circuits into one Application Note) so
that valuable “gems” will not be lost. This Application Note
contains circuits that can power most any system you can
imagine, from desktop computer systems to micropower
systems for portable and handheld equipment. Also
included here are circuits that provide 300W or more of
power factor corrected DC from a universal input. Battery
chargers are included, some that charge several battery
types, some that are optimized to charge a single type.
MOSFET drivers, high side switches and H-bridge driver
circuits are also included, as is an article on simple thermal
analysis. With these introductory remarks, I’ll stand aside
and let the authors describe their circuits.
ARTICLE INDEX
REGULATORS—SWITCHING (BUCK)
High Power (>4A)
Big Power for Big Processors: The LTC®1430 Synchronous Regulator ............................................................. 4
Applications for the LTC1266 Switching Regulator ............................................................................................ 5
A High Efficiency 5V to 3.3V/5A Converter ......................................................................................................... 7
High Current, Synchronous Step-Down Switching Regulator ............................................................................ 8
Medium Power (1A to 4A)
1MHz Step-Down Converter Ends 455kHz IF Woes ......................................................................................... 10
High Output Voltage Buck Regulator ................................................................................................................ 11
The LTC1267 Dual Switching Regulator Controller Operates from High Input Voltages................................... 12
High Efficiency 5V to 3.3V/1.25A Converter in 0.6 Square Inches .................................................................... 13
LT ®1074/LT1076 Adjustable 0V to 5V Power Supply ....................................................................................... 14
Triple Output 3.3V, 5V and 12V High Efficiency Notebook Power Supply ........................................................ 15
The New SO-8 LTC1147 Switching Regulator Controller Offers High Efficiency in a Small Footprint ............... 17
The LT1432: 5V Regulator Achieves 90% Efficiency ........................................................................................ 20
Low Power (<1A)
Applications for the LTC1265 High Efficiency Monolithic Buck Converter ........................................................ 22
REGULATORS—SWITCHING (BOOST)
Medium Power (1A to 4A)
High Output Current Boost Regulator............................................................................................................... 24
Low Power (<1A)
Applications for the LT1372 500kHz Switching Regulator ............................................................................... 25
, LTC and LT are registered trademarks of Linear Technology Corporation.
AN66-1
Application Note 66
REGULATORS—SWITCHING (BUCK/BOOST)
±5V Converter Uses Off-the-Shelf Surface Mount Coil..................................................................................... 27
Switching Regulator Provides Constant 5V Output from 3.5V to 40V Input Without a Transformer ................ 28
Switching Regulator Provides ±15V Output from an 8V to 40V Input Without a Transformer ......................... 29
REGULATORS—SWITCHING (INVERTING)
High Efficiency 12V to – 12V Converter ............................................................................................................ 32
Regulated Charge Pump Power Supply ............................................................................................................ 34
Applications for the LTC1265 High Efficiency Monolithic Buck Converter ........................................................ 22
LTC1174: A High Efficiency Buck Converter ..................................................................................................... 35
REGULATORS—SWITCHING (FLYBACK)
Applications for the LT1372 500kHz Switching Regulator ............................................................................... 25
REGULATORS—SWITCHING (POWER FACTOR CORRECTED)
The New LT1508/LT1509 Combines Power Factor Correction and a PWM in a Single Package ...................... 37
REGULATORS—SWITCHING (DISCUSSION)
Adding Features to the Boost Topology............................................................................................................ 39
Sensing Negative Outputs ................................................................................................................................ 40
REGULATORS—SWITCHING (MICROPOWER)
3-Cell to 3.3V Buck/Boost Converter ................................................................................................................ 41
LT1111 Isolated 5V Switching Power Supply ................................................................................................... 41
Low Noise Portable Communications DC/DC Converter ................................................................................... 43
Applications for the LT1302 Micropower DC/DC Converter ............................................................................. 44
Clock-Synchronized Switching Regulator Has Coherent Noise ........................................................................ 49
Battery-Powered Circuits Using the LT1300 and LT1301 ................................................................................. 51
LTC1174: A High Efficiency Buck Converter ..................................................................................................... 35
Battery-Powered Circuits Using the LT1304 Micropower DC/DC Converter with Low-Battery Detector ........... 54
Automatic Load Sensing Saves Power in High Voltage Converter .................................................................... 57
REGULATORS—SWITCHING (MICROPOWER)
Backlight
High Efficiency EL Driver Circuit....................................................................................................................... 58
A Low Power, Low Voltage CCFL Power Supply .............................................................................................. 60
All Surface Mount EL Panel Driver Operates from 1.8V to 8V Input ................................................................. 61
A Dual Output LCD Bias Voltage Generator ...................................................................................................... 62
LCD Bias Supply............................................................................................................................................... 63
REGULATORS—SWITCHING (MICROPOWER)
Switched Capacitor
Regulated Charge Pump Power Supply ............................................................................................................ 34
REGULATORS—SWITCHING (MICROPOWER)
VPP Generator
LTC1262 Generates 12V for Programming Flash Memories Without Inductors ............................................... 64
Flash Memory VPP Generator Shuts Down with 0V Output ............................................................................. 64
AN66-2
Application Note 66
REGULATORS—LINEAR
Low Noise Wireless Communications Power Supply ....................................................................................... 65
An LT1123 Ultralow Dropout 5V Regulator ...................................................................................................... 66
REGULATORS—LINEAR
Microprocessor Power
LT1580 Low Dropout Regulator Uses New Approach to Achieve High Performance ....................................... 67
LT1585: New Linear Regulator Solves Load Transients ................................................................................... 68
BATTERY CHARGERS
Charging NiMH/NiCd or Li-Ion with the LT1510 ............................................................................................... 70
Lithium-Ion Battery Charger ............................................................................................................................. 71
Simple Battery Charger Runs at 1MHz ............................................................................................................. 73
A Perfectly Temperature Compensated Battery Charger ................................................................................... 74
A Simple 300mA NiCd Battery Charger ............................................................................................................ 75
High Efficiency (>90%) NiCd Battery Charger Circuit Programmable for 1.3A Fast Charge
or 100mA Trickle Charge.................................................................................................................................. 76
POWER MANAGEMENT
LT1366 Rail-to-Rail Amplifier Controls Topside Current .................................................................................. 78
An Isolated High Side Driver ............................................................................................................................ 79
LTC1163: 2-Cell Power Management ............................................................................................................... 80
LTC1157 Switch for 3.3V PC Card Power ........................................................................................................ 81
The LTC1157 Dual 3.3V Micropower MOSFET Driver ...................................................................................... 82
The LTC1155 Does Laptop Computer Power Bus Switching, SCSI Termination Power or
5V/3A Extremely Low Dropout Regulator ......................................................................................................... 82
A Circuit That Smoothly Switches Between 3.3V and 5V.................................................................................. 84
A Fully Isolated Quad 4A High Side Switch ...................................................................................................... 85
The LTC1153 Electronic Circuit Breaker ........................................................................................................... 86
LTC1477: 0.07Ω Protected High Side Switch Eliminates “Hot Swap” Glitching ............................................... 87
MISCELLANEOUS
Protected Bias for GaAs Power Amplifiers ....................................................................................................... 88
LT1158 H-Bridge Uses Ground Referenced Current Sensing for System Protection........................................ 89
LT1158 Allows Easy 10A Locked Antiphase Motor Control .............................................................................. 91
All Surface Mount Programmable 0V, 3.3V, 5V and 12V VPP Generator for PCMCIA ...................................... 92
A Tachless Motor Speed Controller .................................................................................................................. 93
LT1161...And Back and Stop and Forward and Rest—All with No Worries at All ............................................ 95
Simple Thermal Analysis—A Real Cool Subject for LTC Regulators ............................................................... 98
ALPHABETIC INDEX
By Major Categories ....................................................................................................................................... 101
AN66-3
Application Note 66
Regulators—Switching (Buck)
similar class processor and the input is taken from the
system 5V ±5% supply. The LTC1430 provides the precisely regulated output voltage required by the processor
without the need for an external precision reference or
trimming. Figure 1 shows a typical application with a
3.30V ±1% output voltage and a 12A output current limit.
The power MOSFETs are sized so as not to require a heat
sink under ambient temperature conditions up to 50°C.
Typical efficiency is above 91% from 1A to 10A output
current and peaks at 95% at 5A (Figure 2).
High Power (>4A)
BIG POWER FOR BIG PROCESSORS:
THE LTC1430 SYNCHRONOUS REGULATOR
by Dave Dwelley
The LTC1430 is a new switching regulator controller
designed to be configured as a synchronous buck converter with a minimum of external components. It runs at
a fixed switching frequency (nominally 200kHz) and provides all timing and control functions, adjustable current
limit and soft start, and level shifted output drivers designed to drive an all N-channel synchronous buck converter architecture. The switch driver outputs are capable
of driving multiple paralleled power MOSFETs with
submicrosecond slew rates, providing high efficiency at
very high current levels while eliminating the need for a
heat sink in most designs. The LTC1430 is usable in
converter designs providing from a few amps to over 50A
of output current, allowing it to supply 3.3V power to the
most current-hungry arrays of microprocessors.
Pentium is a registered trademark of Intel Corporation.
100
EFFICIENCY (%)
90
VCC = 5V
TA = 25°C
VOUT = 3.3V
80
70
60
50
40
0.1
A Typical 5V to 3.3V Application
The typical application for the LTC1430 is a 5V to 3.xV
converter on a PC motherboard. The output is used to
power a Pentium® processor, Pentium® Pro processor or
1
LOAD CURRENT (A)
10
AN66 F02
Figure 2. Efficiency Plot for Figure 1’s Circuit. Note That
Efficiency Peaks at a Respectable 95%
VIN
4.5V TO 5.5V
R1
16k
C1
0.1µF
C2 +
10µF
SGND
D1
1N4148
R2
100Ω
SVCC
PVCC2
IMAX
PVCC1
G1
M1B
MTD20N03HL
C3
0.1µF
M1A
MTD20N03HL
R3
1k
LTC1430
L1
2.5µH/15A
VOUT
3.3V
IFB
NC
FREQ
G2
SHUTDOWN
SHDN
+SENSE
COMP
CC*
3300pF
SS
100pF*
RC*
33k
CSS
0.01µF
SGND
SGND
VTRIM
M2
MTD20N03HL
NC
+
–SENSE
PGND
+
COUT
330µF
6.3V
×6
AN66 F01
PGND
AND SGND
CONNECTED AT
A SINGLE POINT
PGND
L1 = 6 TURNS #16 WIRE ON MICROMETALS T50-52B CORE
CIN = 4 EACH AVX TPSE 227M010R0100
COUT = 6 EACH AVX TPSE 337M006R0100
*TRIM TO OPTIMIZE TRANSIENT REPONSE
Figure 1. Typical 5V to 3.3V, 10A LTC1430 Application
AN66-4
CIN
220µF
10V
×4
Application Note 66
The 12A current limit is set by the 16k resistor R1 from
PVCC to IMAX and the 0.035Ω ON resistance of the
MTD20N03HL MOSFETs (M1A, M1B).
The 0.1µF capacitor in parallel with R1 improves power
supply rejection at IMAX, providing consistent current limit
performance when voltage spikes are present at PVCC.
Soft start time is set by CSS; the 0.01µF value shown reacts
with an internal 10µA pull-up to provide a 3ms start-up
time. The 2.5µH, 15A inductor is sized to allow the peak
current to rise to the full current limit value without
saturating. This allows the circuit to withstand extended
output short circuits without saturating the inductor core.
The inductor value is chosen as a compromise between
peak ripple current and output current slew rate, which
affects large-signal transient response. If the output load
is expected to generate large output current transients (as
large microprocessors tend to do), the inductor value will
need to be quite low, in the 1µH to 10µH range.
Loop compensation is critical for obtaining optimum
transient response with a voltage feedback system like
the LTC1430; the compensation components shown
here give good response when used with the output
capacitor values and brands shown (Figure 3). The ESR
of the output capacitor has a significant effect on the
transient response of the system. For best results use the
APPLICATIONS FOR
THE LTC1266 SWITCHING REGULATOR
by Greg Dittmer
Figures 4, 5 and 6 show the three basic circuit configurations for the LTC1266. The all N-channel circuit shown in
Figure 4 is a 3.3V/5A surface mount converter with the
internal MOSFET drivers powered from a separate supply,
PWR VIN. The VGS(ON) of the Si9410 N-channel MOSFETs
is 4.5V; thus the minimum allowable voltage for PWR VIN
is VIN(MAX) + 4.5V. At the other end, PWR VIN should be
kept under the maximum safe level of 18V, limiting VIN to
18V – 4.5V = 13.5V. The current sense resistor value is
chosen to set the maximum current to 5A according to the
formula IOUT = 100mV/RSENSE. With VIN = 5V, the 5µH
inductor and 130pF timing capacitor provide an operating
frequency of 175kHz and a ripple current of 1.25A.
20mV/DIV
5A/DIV
AN66 F03
Figure 3. Transient Response: 0A to 5A Load Step
Imposed on Figure 1’s Output
largest value, lowest ESR capacitors that will fit the
design budget and space requirements. Several smaller
capacitors wired in parallel can help reduce total output
capacitor ESR to acceptable levels. Input bypass capacitor ESR is also important to keep input supply variations
to a minimum with 10AP-P square wave current pulses
flowing into M1. AVX TPS series surface mount tantalum
capacitors and Sanyo OS-CON organic electrolytic capacitors are recommended for both input and output
bypass duty. Low cost “computer grade” aluminum
electrolytics typically have much higher series resistance
and will significantly degrade performance. Don’t count
on that parallel 0.1µF ceramic cap to lower the ESR of a
cheap electrolytic cap to acceptable levels.
Figure 5 shows an LTC1266 in the charge pump configuration designed to provide a 3.3V/10A output from a single
supply. The Si4410s are new logic level, surface mount,
N-channel MOSFETs from Siliconix that provide a mere
0.02Ω of on-resistance at VGS = 4.5V and thus provide a
10A solution with minimal components. The efficiency
plot shows that the converter is still close to 90% efficient
at 10A. Because the charge pump configuration is used,
the maximum allowable VIN is 18V/2 = 9V. Due to the high
AC currents in this circuit we recommend low ESR
OS-CON or AVX input/output capacitors to maintain efficiency and stability.
Figure 6 shows the conventional P-channel topside switch
circuit configuration for implementing a 3.3V/3A regulator. The P-channel configuration allows the widest possible supply range of the three basic circuit configurations,
AN66-5
Application Note 66
3.5V to 18V, and provides extremely low dropout, exceeding that of most linear regulators. The low dropout results
from the LTC1266’s ability to achieve a 100% duty cycle
when in P-channel mode. In N-channel mode the duty
cycle is limited to less than 100% to ensure proper startup and thus the dropout voltage for the all N-channel
converters is slightly higher.
Si9410DY
The three application circuits demonstrate the fixed 3.3V
version of the LTC1266. The LTC1266 is also available in
fixed 5V and adjustable versions. All three versions are
available in 16-pin SO packages.
VIN
3.5V TO 14V
CIN
100µF
20V
OSCON
×2
+
D1
MBRS140T3
100
VIN = 5V
2
3
4
BINH
5
6
7
CT
130pF
CC
3300pF
8
TDRIVE
BDRIVE
Si9410DY
15
PWR VIN
PGND
LTC1266-3.3
14
LBOUT
PINV
BINH
LBIN
VIN
SGND
CT
SHDN
NC
ITH
L
5µH
13
12
11
80
0.01
SHDN
10
9
SENSE +
SENSE –
RC
470Ω
90
85
+
0.1µF
PWR VIN
(SEE TEXT)
16
EFFICIENCY (%)
95
1
RSENSE
0.02Ω
1000pF
COUT
330µF
10V
×2
0.1
1
LOAD CURRENT (A)
5
AN66 F04b
Figure 4b. Efficiency for Figure 4a’s Circuit
VOUT
3.3V/5A
AN66 F04a
Figure 4a. All N-Channel 3.3V/5A Regulator with Drivers Powered
from Seperate Power VIN (PWR VIN) Supply
VIN
MBR0530T1
Si4410DY
D1
MBRS340T3
4V TO 9V
CIN
100µF
10V
OS-CON
×3
+
0.1µF
BINH
4
13
7
CC
3300pF 8
Si4410DY
95
15
PWR VIN
PGND
LTC1266-3.3
14
LBOUT
PINV
6
RC
470Ω
16
3
5
CT
220pF
BDRIVE
BINH
LBIN
VIN
SGND
CT
SHDN
ITH
SENSE –
NC
L
5µH
12
11
SHDN
9
SENSE +
RSENSE
0.01Ω
COUT
330µF
10V
×3
VOUT
3.3V
10A
Figure 5a. All N-Channel Single Supply 5V to 3.3V/10A Regulator
AN66-6
90
85
10
1000pF
VIN = 5V
EFFICIENCY (%)
2
TDRIVE
+
1
100
80
0.01
0.1
1
LOAD CURRENT (A)
10
AN66 F05b
Figure 5b. Efficiency for Figure 5a’s Circuit
AN66 F05a
Application Note 66
D1
MBRS140T3
Si9430DY
VIN
3.5V TO 18V
CIN
100µF
25V
+
100
VIN = 5V
BDRIVE
TDRIVE
16
95
Si9410DY
15
PWR VIN
PGND
LTC1266-3.3
14
3
LBOUT
PINV
EFFICIENCY (%)
1
0.1µF
2
5
6
7
CT
220pF
CC
3300pF 8
LBIN
BINH
VIN
SGND
CT
SHDN
NC
ITH
13
11
SHDN
10
9
SENSE +
SENSE –
RC
1k
90
85
12
+
4
BINH
L
10µH
RSENSE
0.033Ω
1000pF
80
0.01
COUT
220µF
10V
×2
0.1
LOAD CURRENT (A)
1
3
AN66 F06b
Figure 6b. Efficiency for Figure 6a’s Circuit
VOUT
3.3V
3A
AN66 F06a
Figure 6a. Low Dropout 3.3V/3A Complementary MOSFET Regulator
High efficiency is mandatory in these applications, since
converting 5V to 3.3V at 5A using a linear regulator would
require dissipating over 8W. This wastes power and board
space for heat sinking.
A HIGH EFFICIENCY 5V TO 3.3V/5A CONVERTER
by Randy G. Flatness
The next generation of notebook and desktop computers
is incorporating more 3.3V ICs alongside 5V devices. As
the number of devices increases, the current requirements also increase. Typically, a high current 5V supply is
already available. Thus, the problem is reduced to deriving
3.3V from 5V efficiently in a small amount of board space.
The LTC1148 synchronous switching regulator controller
accomplishes the 5V to 3.3V conversion with high efficiencies over a wide load current range. The circuit shown
in Figure 7 provides 3.3V at efficiencies greater than 90%
VIN
5V
+
C1
1µF
0V = NORMAL
>2V = SHUTDOWN
10
Q2
Si9430DY
3
SHDN
VIN
1
PDRIVE
LTC1148-3.3
SENSE +
6
R1
470Ω
C4
3300pF
4
C5
680pF
NPO
C3
100µF
20V
×2
+
C2
0.1µF
SENSE –
ITH
CT
SGND
11
NDRIVE
PGND
Q1
Si9430DY
L1
27µH
R2
0.02Ω
8
7
14
C7
0.01µF
Q3
Si9410
+
D1
MBRS140T3
C1 = TANTALUM
C3 = SANYO (OS-CON) 20SA100M ESR = 0.037Ω IRMS = 2.25A
C6 = AVX (TA) TPSE227K01R0080 ESR = 0.080Ω IRMS = 1.285A
Q1, Q2 = SILICONIX PMOS BVDSS = 20V DCRON = 0.100Ω Qg = 50nC
Q3 = SILICONIX NMOS BVDSS = 30V DCRON = 0.050Ω Qg = 30nC
VOUT D1 = MOTOROLA SCHOTTKY VBR = 30V
3.3V R2 = KRL NP-2A-C1-0R020J Pd = 3W
5A
L1 = KOOL Mµ® CORE, 16 GAUGE
C6
220µF
10V
×2
COILTRONICS (408)241-7876
KRL BANTRY (603) 668-3210
SILICONIX (800) 554-5565
KOOL Mµ IS A REGISTERED TRADEMARK OF MAGNETICS, INC.
12
AN66 F07
Figure 7. LTC1148-3.3 High Efficiency 5V to 3.3V/5A Step-Down Converter
AN66-7
Application Note 66
maximize the operating efficiency at low output currents,
Burst ModeTM operation is used to reduce switching losses.
Synchronous switching, combined with Burst Mode operation, yields very efficient energy conversion over a wide
range of load currents.
EFFICIENCY (%)
100
90
80
70
1
10
100
1000
OUTPUT CURRENT (mA)
10000
AN66 F08
Figure 8. Efficiency for 5V to 3.3V Synchronous Switcher
from 5mA to 5A (over three decades of load current). The
efficiency of the circuit in Figure 7 is plotted in Figure 8.
At an output current of 5A the efficiency is 90%; this
means only 1.8W are lost. This lost power is distributed
among RSENSE, L1 and the power MOSFETs; thus heat
sinking is not required.
The LTC1148 series of controllers use constant off-time
current mode architecture to provide clean start-up, accurate current limit and excellent line and load regulation. To
HIGH CURRENT, SYNCHRONOUS
STEP-DOWN SWITCHING REGULATOR
by Brian Huffman
The LTC1149 is a half-bridge driver designed for synchronous buck regulator applications. Normally a P- and
N-channel output stage is employed, but the P-channel
device ON resistance becomes a limiting factor at output
currents above 2A. N-channel MOSFETs are better suited
for use in high current applications, since they have a
substantially lower ON resistance than comparably priced
P-channels. The circuit shown in Figure 9 adapts the
LTC1149 to drive a half-bridge consisting of two
N-channel MOSFETs, providing efficiency in excess of
90% at an output current of 5A.
AN66-8
The top P-channel MOSFETs in Figure 7 will be on 2/3 of
the time with an input of 5V. Hence, these devices should
be carefully examined to obtain the best performance. Two
MOSFETs are needed to handle the peak currents safely
and enhance high current efficiency. The LTC1148 can
drive both MOSFETs adequately without a problem. A
single N-channel MOSFET is used as the bottom synchronous switch, which shunts the Schottky diode. Finally,
adaptive anti-shoot-though circuitry automatically prevents cross conduction between the complementary
MOSFETs which can kill efficiency.
The circuit in Figure 7 has a no-load current of only 160µA.
In shutdown mode, with Pin 10 held high (above 2V), the
quiescent current decreases to less than 20µA with all
MOSFETs held off DC. Although the circuit in Figure 7 is
specified at a 5V input voltage, the circuit will function from
4V to 15V without requiring any component substitutions.
Burst Mode is a trademark of Linear Technology Corporation.
The circuit’s operation is as follows: the LTC1149 provides
a P-drive output (Pin 4) that swings between ground and
10V, turning Q3 on and off. While Q3 is on, the N-channel
MOSFET (Q4) is off because its gate is pulled low by Q3
through D2. During this interval, the Ngate output (Pin 13)
turns the synchronous switch (Q5) on creating a low
resistance path for the inductor current.
Q4 turns on when its gate is driven above the input voltage.
This is accomplished by bootstrapping capacitor C2 off
the drain of Q4. The LTC1149 VCC output (Pin 3) supplies
a regulated 10V output that is used to charge C2 through
D1 while Q4 is off. With Q4 off, C2 charges to 5V during the
first cycle in Burst Mode operation and to 10V thereafter.
Application Note 66
VIN
12V TO 36V
+
D1
1N4148
R4
220Ω
CIN
1000µF
63V
Q1
2N3906
+
R2
10k
C1
0.1µF
2
3
+
C3
3.3µF
5
16
10
0V = NORMAL
>2V = SHUTDOWN
C4
3300pF
X7R
R1
1k
CT
820pF
NPO
VIN
PGATE
VCC
VCC
P-DRIVE
D2
1N4148
1
4
C2
0.1µF
R3
470Ω
15 SHDN2
7 I
TH
6 CT
Q4
MTP30N06EL
Q3
VN2222LL
9
SENSE+
SENSE– 8
L1
50µH
R5
100Ω
CAP LTC1149-5
SHDN1
Q2
2N2222
COUT
220µF
10V
×2
0.001µF
R6
100Ω
Q5
IRFZ34
D3
MBR160
SGND PGND RGND
12
5V
5A
+
C4
NGATE 13
11
RSENSE
0.02Ω
14
C3 (TA) LOW ESR
CIN NICHICON (AL) UPL1J102MRH, ESR = 0.027Ω, IRMS = 2.370A
COUT SANYO (OS-CON) 10SA220M, ESR = 0.035Ω, IRMS = 2.360A
Q1 PNP, BVCEO = 30V
Q2 NPN, BVCEO = 40V
Q3 SILICONIX NMOS, BVDSS = 60V, RDSON = 5Ω
Q4, Q5
D1, D2
D3
RSENSE =
L1 =
NMOS, BVDSS = 60V, RDSON = 0.05Ω
SILICON, VBR = 75V
MOTOROLA SCHOTTKY, VBR = 60V
KRL NP-2A-C1-0R020J, PD = 3W
COILTRONICS CTX50-5-52, DCR = 0.21Ω, IRON POWDER CORE
ALL OTHER CAPACITORS ARE CERAMIC
AN66 F09
Figure 9. LTC1149-5 (12V-36V to 5V/5A) Using N-Channel MOSFETs
Efficiency performance for this circuit is quite impressive.
Figure 10 shows that for a 12V input the efficiency never
drops below 90% over the 0.6A to 5A range. At higher
input voltages efficiency is reduced due to transition
losses in the power MOSFETs. For low output currents
efficiency rolls off because of quiescent current losses.
100
90
12V
EFFICIENCY (%)
When Q3 turns off, the N-channel MOSFET is turned on by
the SCR-connected NPN/PNP network (Q1 and Q2). Resistor R2 supplies Q2 with enough base drive to trigger the
SCR. Q2 then forces Q1 to turn on, supplying more base
drive to Q2. This regenerative process continues until both
transistors are fully saturated. During this period, the
source of Q4 is pulled to the input voltage. While Q4 is on,
its gate source voltage is approximately 10V, fully enhancing the N-channel MOSFET.
80
24V
70
36V
60
50
0.1
1
OUTPUT CURRENT (A)
5
AN66 F10
Figure 10. LTC1149-5 (12V-36V to 5V/5A) High Current Buck
AN66-9
Application Note 66
Regulators—Switching (Buck)
ciency buck topology switching regulator. The switch is
internally grounded, calling for the floating supply arrangement shown (D1 and C1). The circuit converts inputs
of 8V through 30V to a 5V/1A output.
Medium Power (1A to 4A)
1MHz STEP-DOWN CONVERTER
ENDS 455kHz IF WOES
by Mitchell Lee
There can be no doubt that switching power supplies and
radio IFs don’t mix. One-chip converters typically operate
in the range of 20kHz to 100kHz, placing troublesome
harmonics right in the middle of the 455kHz band. This
contributes to adverse effects such as “desensing” and
outright blocking of the intended signals. A new class of
switching converter makes it possible to mix high efficiency power supply techniques and 455kHz radio IFs
without fear of interference.
The chip’s internal oscillator operates at 1MHz for load
currents of greater than 50mA with a guaranteed tolerance
of 12% over temperature. Even wideband 455kHz IFs are
unaffected, as the converter’s operating frequency is well
over one octave distant.
Figure 12 shows the efficiency of Figure 11’s circuit. You
can expect 80% to 90% efficiency over an 8V to 16V input
range with loads of 200mA or more. This makes the circuit
suitable for 12V battery inputs (that’s how I’m using it), but
no special considerations are necessary with adapter
inputs of up to 30V.
The circuit shown in Figure 11 uses an LT1377 boost
converter operating at 1MHz to implement a high effi-
+
D1
1N5818
100µF
5
V+
4
NC
+
3
C1
2.2µF
VSW
LT1377
PFB
3.57k
2
1N4148 10Ω
NFB
VC
6
VIN = 8V
90
SHDN
SG
VO = 5V
100
8
EFFICIENCY (%)
8V TO 30V
INPUT
PG
1
1.24k
100nF
80
VIN = 12V
70
VIN = 16V
7
60
2k
4.7nF
47nF
50
0
CTX20-2P*
5V
1A
+
MBRS130
*CTX20-2P, COILTRONICS 20µH
**OS-CON, SANYO VIDEO COMPONENTS
150µF
6.3V
OSCON**
400
600
IOUT (mA)
800
1000
AN66 F12
Figure 12. Efficiency Graph of the
Circuit Shown in Figure 3
AN66 F11
Figure 11. Schematic Diagram: 1MHz LT1377-Based Boost Converter
AN66-10
200
Application Note 66
HIGH OUTPUT VOLTAGE BUCK REGULATOR
by Dimitry Goder
High efficiency step-down conversion is easy to implement using the LTC1149 as a buck switching regulator
controller. The LTC1149 features constant off-time, current mode architecture and fully synchronous rectification. Current mode operation was selected for its
well-known advantages of clean start-up, accurate current
limit and excellent transient response.
Inductor current sensing is usually implemented by placing a resistor in series with the coil, but the common mode
voltage at the LTC1149’s Sense pins is limited to 13V. If a
higher output voltage is required, the current sense resistor can be placed in the circuit’s ground return to avoid
VIN
26V TO 35V
+
common mode problems. The circuit in Figure 13 can be
used in applications that do not lend themselves to this
approach.
Figure 13 shows a special level shifting circuit (Q1 and U2)
added to a typical LTC1149 application. The LT1211, a
high speed, precision amplifier, forces the voltage across
R5 to equal the voltage across current sense resistor R8.
Q1’s drain current flows to the source, creating a voltage
across R6 proportional to the inductor current, which is
now referenced to ground. This voltage can be directly
applied to the current sense inputs of U1, the LTC1149.
C12 and C4 are added to improve high frequency noise
immunity. Maximum input voltage is now limited by the
LT1211; it can be increased if a Zener diode is placed in
parallel with C12.
C9
0.068µF
C13
R9
100Ω
C12
0.1µF
1
2
C8
0.047µF
3
4
C7
1µF
5
6
C5
220pF
7
C6
3300pF 8
R4
510Ω
P-GATE
CAP
16
15
U1
SHDN
VIN
LTC1149
14
RGND
VCC
13
N-GATE
P-DRIVE
12
VCC
PGND
11
CT
SGND
10
ITH
VFB
9
SENSE –
SENSE+
C2
1000pF
Q2
RFD15P05
Q3
RFD14N05
L1
150µH
D1
MBRS140
R8
0.05Ω
R5
100Ω
1%
R13
12k
1%
C1
D3
1N4148
R12
220k
1%
8
1
Q1
VN2222LL
C11
100pF
+
R6
100Ω
1%
24V
2A
R9
100Ω
C10
0.1µF
+
U2A
LT1211
–
3
2
4
R10
100Ω
AN66 F13
Figure 13. High Output Voltage Buck Regulator Schematic Using LTC1149
AN66-11
Application Note 66
Adjustable Output 3.6V and 5V Converter
THE LTC1267 DUAL SWITCHING REGULATOR
CONTROLLER OPERATES FROM
HIGH INPUT VOLTAGES
by Randy G. Flatness
The adjustable output LTC1267-ADJ shown in Figure 16 is
configured as a 3.6V/2.5A and 5V/2A converter. The resistor divider composed of R1 and R2 sets the output voltage
according to the formula VOUT = 1.25V (1 + R2/R1). The
input voltage range for this application is 5.5V to 28V.
Fixed Output 3.3V and 5V Converter
A fixed LTC1267 application circuit creating 3.3V/2A and
5V/2A is shown in Figure 15. The operating efficiency
shown in Figure 14 exceeds 90% for both the 3.3V and 5V
sections. The 3.3V section of the circuit in Figure 15
comprises the main switch Q1, synchronous switch Q2,
inductor L1 and current shunt RSENSE3.
100
LTC1267
VIN = 12V
5V SECTION
EFFICIENCY (%)
90
The 5V section is similar and comprises Q3, Q4, L2 and
RSENSE5. Each current sense resistor (RSENSE) monitors
the inductor current and is used to set the output current
according to the formula IOUT = 100mV/RSENSE. Advantages of current control include excellent line and load
transient rejection, inherent short-circuit protection and
controlled start-up currents. Peak inductor currents for L1
and L2 are limited to 150mV/RSENSE or 3.0A. The EXT VCC
pin is connected to the 5V output increasing efficiency at
high input voltages. The maximum input voltage is limited
by the MOSFETs and should not exceed 28V.
80
LTC1267
VIN = 12V
3.3V SECTION
70
60
0.001
0.01
0.1
OUTPUT CURRENT
1A 2A
AN66 F14
Figure 14. LTC1267 Efficiency vs Output Current
of Figure 15 Circuit
5.5V < VIN < 28V
+ CIN3
100µF
50V
VOUT3
3.3V
2A
33µF
Q1
P-CH
Si9435DY
1N4148
4
0.1µF
14
D1
MBRS140T3
1000pF
13
12
6
COUT3
220µF
10V
×2
3
1
VCC3 CAP3
PGATE3
VCC
8
5
RSENSE3
0.05Ω
3.3µF
0.15µF
0.15µF
L1
20µH
+
Q2
N-CH
Si9410DY
R SENSE,:KRL SL-C1-1/2-R050J
L1:COILTRONICS CTX20-4
L2:COILTRONICS CTX33-4
2
27
26
VIN MASTER CAP5
SHDN
28
PDRIVE5
SENSE+3
SENSE+5
LTC1267
24
SHDN3
SHDN5
NGATE3
PGND3 SGND3 CT3
11
ITH3
ITH5
15
RC5
1k
CT5
NGATE5
SGND5 PGND5
9
10
RC3
1k
16
CT3
270pF
CT5
CC3
CC5
3300pF 3300pF 270pF
20
22
100µF
50V
Q3
P-CH
Si9435DY
L2
33µH
RSENSE5
0.05Ω
D2
MBRS140T3
COUT5
220µF
10V
×2
VOUT5
5V
2A
0.1µF
18
SENSE–5 17
SENSE–3
7
1N4148
21
EXT VCC VCC5
25
PGATE5
PDRIVE3
0V = RUN
>2V = SHUTDOWN
1000pF
19
23
+
Q4
N-CH
Si9410DY
0V = RUN
>2V = SHUTDOWN
KRL (603) 668-3210
COILTRONICS (407) 241-7876
Figure 15. LTC1267 Dual Output 3.3V and 5V High Efficiency Regulator
AN66-12
+ CIN5
+
+
AN66 F15
Application Note 66
5.5V < VIN < 28V
+ CIN1
100µF
50V
1N4148
4
L1
20µH
5
0.1µF
RSENSE1
0.04Ω
13
D1
MBRS140T3
1000pF
12
11
6
+
COUT1
220µF
10V
×2
R2
100k
1%
N-CH
Si9410DY
R1
52.3k
1%
3
1
VCC1 CAP1
PGATE1
VCC
7
2
27
26
VIN MASTER CAP2
SHDN
1N4148
21
28
EXT VCC VCC2
25
PGATE2
PDRIVE1
PDRIVE2
SENSE+1
SENSE+2
LTC1267-ADJ
P-CH
Si9435DY
L2
33µH
24
SHDN1
NGATE2
10
14
ITH1
ITH2
15
RC1
1k
CT2
PGND2
VFB2
SGND2
8
9
RC1
1k
16
CT1
270pF
CC1
CT2
CC2
3300pF 3300pF 270pF
20
VOUT2
5V
2A
0.1µF
1000pF
23
22
N-CH
Si9410DY
19
D2
MBRS140T3
COUT2
220µF
10V
×2
R2
150k
1%
+
R1
49.9k
1%
100pF
100pF
0V = RUN
>2V = SHUTDOWN
RSENSE2
0.05Ω
18
SENSE–2 17
SENSE–1
NGATE1
VFB1 SGND1 CT1
100µF
50V
3.3µF
0.15µF
0.15µF
33µF
P-CH
Si9435DY
VOUT1
3.6V
2.5A
+ CIN2
+
+
KRL (603) 668-3210
COILTRONICS (407) 241-7876
R SENSE1,: KRL SL-C1-1/2-R040J
R SENSE2,: KRL SL-C1-1/2-R050J
L1: COILTRONICS CTX20-4
L2: COILTRONICS CTX33-4
AN66 F16
Figure 16. LTC1267 Dual Adjustable High Efficiency Regulator Circuit. Output Voltages Set at 3.6V and 5V
HIGH EFFICIENCY 5V TO 3.3V/1.25A CONVERTER
IN 0.6 SQUARE INCHES
by Randy G. Flatness
5V supply is already available. Thus, the problem is
reduced to deriving 3.3V from 5V at high efficiency in a
small amount of board space.
The next generation of notebook and desktop computers
will incorporate a growing number of 3.3V ICs along with
5V devices. As the number of 3.3V devices increases, the
current requirements increase. Typically, a high current
High efficiency is mandatory in these applications since
converting 5V to 3.3V at 1.25A using a linear regulator
would require dissipating over 2W. This is an unnecessary
waste of power and board space for heat sinking.
+
+
VIN
4V TO 10V
0.1µF
0V = NORMAL
>1.5V = SHUTDOWN
6
3
RC
1K
2
CC
3300pF
CT
120pF
1
VIN
8
PDRIVE
SHDN
LTC1147-3.3
5
SENSE +
SENSE –
ITH
CT
GND
KRL/BANTRY (603) 668-3210
SUMIDA (708) 956-0666
7
4
P-CH
Si9433DY
L1
10µH
CIN
47µF
16V
RSENSE
0.068Ω
0.01µF
+
VOUT
3.3V
1.5A
COUT
100µF
10V
D1
MBRS130LT3
RS: KRL SP-1/2-A1-0R068J
L: SUMIDA
CDR74
(ALT: CD54)
AN66 F17
Figure 17. High Efficiency Controller Converts 5V to 3.3V in Minimum Board Area
AN66-13
Application Note 66
The LTC1147 SO-8 switching regulator controller accomplishes the 5V to 3.3V conversion with high efficiencies
over a wide load current range. The circuit shown in Figure
17 provides 3.3V at efficiencies greater than 90% from
50mA to 1.25A. Using all surface mount components and
a low value of inductance (10µH) for L1, the circuit of
Figure 17 occupies only 0.6 square inches of PC board
area. The efficiency of the circuit in Figure 17 is plotted in
Figure 18.
efficiency; for lower cost an Si9340DY can be used at a
slight reduction in performance.
The circuit in Figure 17 has a no load current of only
160µA. In shutdown, with Pin 6 held high (above 2V), the
quiescent current is reduced to less than 20µA with the
MOSFET held off. Although the circuit in Figure 17 is
specified at a 5V input voltage the circuit will function
from 4V to 10V.
At an output current of 1.25A the efficiency is 90.4%; this
means only 0.4W are lost. This lost power is distributed
among RSENSE, L1 and the power MOSFETs; thus heat
sinking is not required.
95
90
EFFICIENCY (%)
The LTC1147 series of controllers use constant off-time
current mode architecture to provide clean start-up, accurate current limit and excellent line and load regulation. To
maximize the operating efficiency at low output currents,
Burst Mode operation is used to reduce switching losses.
Linear regulator ICs are commonly used in variable power
supplies. Common types such as the 317 can be adjusted
as low as 1.25V in single-supply applications. At low
80
LTC1147-3.3
SUMIDA CD54
VIN = 5V
75
70
65
60
1mA
The P-channel MOSFET in the circuit of Figure 17 will be
on 2/3 of the time with an input voltage of 5V. Hence, this
device should be carefully selected to obtain the best
performance. This design uses an Si9433DY for optimum
LT1074/LT1076 ADJUSTABLE 0V TO 5V
POWER SUPPLY
by Kevin Vasconcelos
LTC1147-3.3
SUMIDA CDR74
VIN = 5V
85
10mA
100mA
OUTPUT CURRENT (A)
1A 2A
AN66 F18
Figure 18. 5V to 3.3V Conversion Efficiency
output voltages power losses in these regulators can be a
problem. For example, if an output current of 1.5A is
required at 1.25V from an input of 8V, the regulator
dissipates more than 10W. Figure 19 shows a DC/DC
converter that functionally replaces a linear regulator in
this application. The converter not only eliminates power
VIN = 10V
TO 20V
C4
0.1µF
5
+
C1
330µF
35V
FB
GND
3
R4
3.01k
1%
R2
3.65k
1%
LT1076
1
+
VC
2
R1
2.7k
C2
0.01µF
D1
MBR340P
R3
10.65k
1%
C3
470µF
50V
7
6
2
+
VSW
4
VOUT
L1
CTX100-5A-52
–
VIN
3
U1
LT1006
4
R6
2.2k
5%
R5
5k
25T
R5
220
1/4W
5%
AN66 F19
L1 = COILTRONICS (407) 241-7876
Figure 19. Adjustable LT1074/LT1076 0V to 5V Power Supply
AN66-14
LT1029
Application Note 66
As R4 is driven from 0V to 5V by the buffer (U1) more or
less current is required from R2 to satisfy the loop’s desire
to hold the feedback summing point at 2.21V. This forces
the converter’s output to swing over the range of 0V to 6V.
Figure 20 shows a comparison of power losses for a linear
regulator and the circuit of Figure 19. The load current is
1.5A in both cases although the LT1076 is capable of
1.75A guaranteed output current in this application and 2A
typical. If more current is required the LT1074 can be
TRIPLE OUTPUT 3.3V, 5V AND 12V
HIGH EFFICIENCY NOTEBOOK POWER SUPPLY
by Randy G. Flatness
LTC1142 Circuit Operation
The application circuit in Figure 22 is configured to provide
output voltages of 3.3V, 5V and 12V. The current capability
of both the 3.3V and 5V outputs is 2A (2.5A peak). The
logic-controlled 12V output can provide 150mA (200mA
peak), which is ideal for flash memory applications. The
operating efficiency shown in Figure 21 exceeds 90% for
both the 3.3V and 5V sections.
The 3.3V section of the circuit in Figure 22 comprises the
main switch Q4, synchronous switch Q5, inductor L1 and
current shunt RSENSE3. The current sense resistor RSENSE
monitors the inductor current and is used to set the output
current according to the formula IOUT = 100mV/RSENSE.
Advantages of current control include excellent line and
load transient rejection, inherent short-circuit protection
and controlled start-up currents. Peak inductor currents
for L1 and T1 of the circuit in Figure 22 are limited to
150mV/RSENSE or 3.0A and 3.75A respectively.
10
LT317
8
POWER LOSS (W)
The circuit of Figure 19 employs a basic positive buck
topology with one exception: a control voltage is applied
through R4 to the feedback summing node at Pin 1 of the
LT1076 switching regulator IC, allowing the output to be
adjusted from 0V to approximately 6V. This encompasses
the 3.3V and 5V logic supply ranges as well as battery pack
combinations of one to four D cells.
substituted for the LT1076. This change accommodates
outputs up to 5A but at the expense of a heftier diode and
coil (D1, L1). An MBR735 and Coiltronics CTX50-2-52 are
recommended for 5A service.
6
4
LT1076
2
0
0
1
5
2
3
4
OUTPUT VOLTAGE (V)
AN66 F20
Figure 20. Power Loss Comparison: Linear Regulator
vs Figure 19’s Power Supply
When the output current for either regulator section drops
below approximately 15mV/RSENSE, that section automatically enters Burst Mode operation to reduce switching
losses. In this mode the LTC1142 holds both MOSFETs off
and “sleeps” at 160µA supply current while the output
capacitor supports the load. When the output capacitor
falls 50mV below its specified voltage (3.3V or 5V) the
LTC1142 briefly turns this section back on, or “bursts,” to
recharge the output capacitor. The timing capacitor pins,
100
95
90
EFFICIENCY (%)
loss as a concern, but can be adjusted for output voltages
as low as 25mV while still delivering an output current of
1.5A.
LTC1142
VIN = 8V
5V SECTION
85
80
75
70
LTC1142
VIN = 8V
3.3V SECTION
65
60
0.001
0.01
0.1
OUTPUT CURRENT (A)
2.5
1
AN66 F21
Figure 21. LTC1142 Efficiency
AN66-15
Application Note 66
+
VIN
6.5V TO 14V
VOUT3
3.3V
2A
L1
33µH
RSENSE 3
0.05Ω
22µF
25V
×2
+
Q4
Si9430DY
23
1
24
VIN3
10
VIN5
SHDN5
P-DRIVE 5
SENSE + 3
SENSE + 5
Q2
Si9430DY
28
6
Q5
Si9410DY
SENSE – 3
SENSE
N-DRIVE 3
–5
N-DRIVE 5
PGND3 SGND3 CT3
4
3
25
ITH3
CT5
ITH5
27
13
510Ω
SGND5 PGND5
11
17
18
VOUT5
5V
2A
T1
9
30µH
RSENSE 5
0.04Ω
15
100Ω
LTC1142
D1
MBRS140
100µF
10V
×2
16
P-DRIVE 3
0.01µF
+
22µF
25V
×2
1µF
2
SHDN3
+
+
0V = NORMAL
>1.5V = SHUTDOWN
1µF
1000pF
R1
100Ω
14
D2
MBRS140
20
Q3
Si9410DY
510Ω
Q1
VN2222LL
R5
18k
220µF
10V
×2
+
CT3
3300pF 3300pF CT5
390pF
200pF
12V ENABLE
0V = 12V OFF
>3V = 12V ON
(6V MAX)
L1: COILTRONICS CTX33-4
T1: DALE LPE-6562-A026
PRIMARY: SECONDARY = 1:1.8
RSENSE 3: KRL SL-1R050J
RSENSE 5: KRL SL-1R040J
COILTRONICS (407) 241-7876
DALE (605) 665-9301
KRL/BANTRY (603) 668-3210
22µF
25V
1
+
20pF
R3
649k
1% 2
R4
294k
1%
VOUT
SHDN
ADJ
LT1121
5
D3
MBRS140
22Ω
C9
22µF
35V
+
12V
150mA
1000pF
VIN
8
GND
3
AN66 F22
Figure 22. LTC1142 High Efficiency Power Supply Schematic Diagram
which go to 0V during the sleep interval, can be monitored
with an oscilloscope to observe burst action. As the load
current is decreased the circuit will burst less and less
frequently.
The timing capacitors CT3 and CT5 set the off-time according to the formula tOFF = 1.3 (104)(CT). The constant
off-time architecture maintains a constant ripple current
while the operating frequency varies only with input
voltage. The 3.3V section has an off-time of approximately 5µs, resulting in a operating frequency of 120kHz
with an 8V input. The 5V section has an off-time of 2.6µs
and a switching frequency of 140kHz with an 8V input.
Auxiliary 12V Output
The operation of the 5V section is identical to the 3.3V
section with inductor L1 replaced by transformer T1. The
12V output is derived from an auxiliary winding on the 5V
AN66-16
inductor. The output from this additional winding is rectified by diode D3 and applied to the input of an LT1121
regulator. The output voltage is set by resistors R3 and R4.
A turns ratio of 1:1.8 is used for T1 to ensure that the input
voltage to the LT1121 is high enough to keep the regulator
out of dropout mode while maximizing efficiency.
The LTC1142 synchronous switch removes the normal
limitation that power must be drawn from the primary 5V
inductor winding in order to extract power from the
auxiliary winding. With synchronous switching, the auxiliary 12V output may be loaded without regard to the 5V
primary output load, provided that the loop remains in
continuous mode operation.
When the 12V output is activated by a TTL high (6V
maximum) on the 12V enable line, the 5V section of the
LTC1142 is forced into continuous mode. A resistor
Application Note 66
divider composed of R1, R5 and switch Q1 forces an
offset, subtracting from the internal offset at Pin 14. When
this external offset cancels the built-in 25mV offset, Burst
Mode operation is inhibited.
5V output. The 100% duty cycle inherent in the LTC1142
provides low dropout operation limited only by the load
current multiplied by the sum of the resistances of the 5V
inductor, Q2 RDS(ON) and current sense resistor RSENSE5.
Auxiliary 12V Output Options
Extending the Maximum Input Voltage
The circuit of Figure 22 can be modified for operation in
low-battery count (6-cell) applications. For applications
where heavy 12V load currents exist in conjunction with
low input voltages (< 6.5V), the auxiliary winding should
be derived from the 3.3V instead of the 5V section. As the
input voltage falls, the 5V duty cycle increases to the point
when there is simply not enough time to transfer energy
from the 5V primary winding to the 12V secondary winding. For operation from the 3.3V section, a transformer
with a turns ratio of 1:3.25 should be used in place of the
33µH inductor L1. Likewise, a 30µH inductor would replace T1 in the 5V section. With these component changes,
the duty cycle of the 3.3V section is more than adequate for
full 12V load currents. The minimum input voltage in this
case will be determined only by the dropout voltage of the
The circuit in Figure 22 is designed for a 14V maximum
input voltage. The operation of the circuit can be extended
to over 18V if a few key components are changed. The
parts that determine the maximum input voltage of the
circuit are the power MOSFETs, the LTC1142 and the input
capacitors. With the LTC1142 replaced by an LTC1142HV,
an 18V typical (20V maximum) input voltage is allowable.
Since the gate drive voltages supplied by the LTC1142 and
LTC1142HV are from ground to VIN, the input voltage
must not exceed the maximum VGS of the MOSFETs. The
MOSFETs specified in Figure 22 have an absolute maximum of 20V, matching that of the LTC1142HV.1 Finally,
the input capacitor’s voltage rating will also have to be
increased above 12V.
THE NEW SO-8 LTC1147 SWITCHING REGULATOR
CONTROLLER OFFERS HIGH EFFICIENCY
IN A SMALL FOOTPRINT
by Randy Flatness
500mW. The efficiency plotted as a function of output
current is shown in Figure 24.
Introduction
The LTC1147 switching regulator controller is a high
efficiency step-down DC/DC converter. It uses the same
current mode architecture and Burst Mode operation as
the LTC1148/LTC1149 but without the synchronous
switch. Ideal for applications requiring up to 1A, the
LTC1147 shows 90% efficiencies over two decades of
output current.
High Efficiency 5V to 3.3V in a Small Area
The LTC1147 5V to 3.3V converter shown in Figure 23
has 85% efficiency at 1A output with efficiencies greater
than 90% for load currents up to 500mA. Using the
LTC1147 reduces the power dissipation to less than
1For improved efficiency, CT5 should be charged to 270pF.
+
+
VIN
(4V TO
12V)
1
VIN
0V = NORMAL 6
SHDN
PDRIVE
>1.5V = SHUTDOWN
8
LTC1147-3.3
3
RC
1k
CC
3300pF
CIN
15µF
25V
×2
0.1µF
ITH
SENSE +
CT
SENSE –
5
P-CH
Si943ODY RSENSE
0.1Ω
VOUT
3.3V
1A
L
100µH
1000pF
2
GND
CT
560pF
7
4
COUT
220µF
D1
6.3V
MBRD330
+
AN66 F23
RS = KRL SP-1/2-A1-0R100
L = COILTRONICS CTX100-4
COILTRONICS (407) 241-7876
KRL/BANTRY (603) 668-3210
Figure 23. This LTC1147 5V to 3.3V Converter Achieves
92% Efficiency at 300mA Load Current
AN66-17
Application Note 66
is the ideal application for the LTC1147. As the output
current increases the diode loss increases. At high inputto-output voltage ratios, the Schottky diode conducts
most of the time. In this situation, any loss in the diode will
have a more significant effect on efficiency and an LTC1148
might therefore be chosen.
100
LTC1147-3.3
EFFICIENCY (%)
90
VIN = 5V
80
70
60
0.001
0.01
0.1
LOAD CURRENT (A)
1
Figure 26 compares the efficiencies of LTC1147-5 and
LTC1148-5 circuits with the same inductor, timing capacitor and P-channel MOSFET. At low input voltages and 1A
output current the efficiency of the LTC1147 differs from
that of the LTC1148 by less than two percent. At lower
AN66 F24
Figure 24. The LTC1147 5V to 3.3V Converter Provides Better
Than 90% Efficiency from 20mA to 500mA of Output Current
100
I2R
The decision whether to use a nonsynchronous LTC1147
design or a fully synchronous LTC1148 design requires a
careful analysis of where losses occur. The LTC1147
switching regulator controller uses the same loss reducing techniques as the other members of the LTC1148/
LTC1149 family. The nonsynchronous design saves the
N-channel MOSFET gate drive current at the expense of
increased loss due to the Schottky diode.
Figure 25 shows how the losses in a typical LTC1147
application are apportioned. The gate-charge loss
(P-channel MOSFET) is responsible for the majority of the
efficiency lost in the midcurrent region. If Burst Mode
operation was not employed, the gate charge loss alone
would cause the efficiency to drop to unacceptable levels
at low output currents. With Burst Mode operation, the DC
supply current represents the only loss component that
increases almost linearly as output current is reduced. As
expected, the I2R loss and Schottky diode loss dominate
at high load currents.
In addition to board space, output current and input
voltage are the two primary variables to consider when
deciding whether to use the LTC1147. At low input-tooutput voltage ratios, the top P-channel switch is on most
of the time, leaving the Schottky diode conducting only a
small percentage of the total period. Hence, the power lost
in the Schottky diode is small at low output currents. This
AN66-18
95
LTC1147 IQ
SCHOTTKY DIODE
90
85
80
0.01
0.03
0.1
0.3
1
OUTPUT CURRENT (A)
3
AN66 F25
Figure 25. Low Current Efficiency is Enhanced by Burst Mode
Operation. Schottky Diode Loss Dominates at High Output
Currents
100
LTC1147-5
LTC1148-5
90
EFFICIENCY (%)
Giving Up the Synchronous Switch?
EFFICIENCY/LOSS (%)
GATE CHARGE
ILOAD = 1A
80
ILOAD = 100mA
70
60
4
6
12
8
10
INPUT VOLTAGE (V)
14
AN66 F26
Figure 26. At High Input Voltages Combined with Low Output
Currents, the Efficiency of the LTC1147 Exceeds That of the
LTC1148
Application Note 66
output currents and high input voltages the LTC1147’s
efficiency can actually exceed that of the LTC1148.
Low Dropout 5V Output Applications
Because the LTC1147 is so well-suited for low input-tooutput voltage ratio applications it is an ideal choice for
low dropout designs. All members of the LTC1148/LTC1149
family (including the LTC1147) have outstandingly low
dropout performance. As the input voltage on the LTC1147
drops, the feedback loop extends the on-time for the
(5.5V
TO 12V)
0.1µF
1
VIN
0V = NORMAL 6
8
PDRIVE
SHDN
>1.5V = SHUTDOWN
LTC1147-5
3
ITH
RC
1k
CC
3300pF
SENSE +
5
CIN
15µF
25V
×3
P-CH
Si943ODY
100
RSENSE
0.05Ω
CT
SENSE –
GND
L
50µH
CT
470pF
7
RS = KRL SL-1-C1-0R050J
L = COILTRONICS CTX50-4
COILTRONICS (407) 241-7876
KRL/BANTRY (603) 668-3210
4
D1
MBRD330
COUT
220µF
10V
×2
LTC1147-5
VOUT
5V
2A
1000pF
2
With the switch turned on at a 100% duty cycle, the
dropout is limited by the load current multiplied by the
sum of the resistances of the MOSFET, the current shunt
and the inductor. For example, the low dropout 5V regulator shown in Figure 27 has a total resistance of less than
0.2Ω. This gives it a dropout voltage of 200mV at 1A
output current. At input voltages below dropout the output
voltage follows the input. This is the circuit whose efficiency is plotted in Figure 28.
+
95
EFFICIENCY (%)
+
+ VIN
P-channel switch (off-time is constant) thereby keeping
the inductor ripple current constant. Eventually the ontime extends so far that the P-channel MOSFET is on at DC
or at a 100% duty cycle.
VIN = 6V
90
VIN = 10V
85
80
75
AN66 F27
70
1
10
100
LOAD CURRENT (mA)
1000
AN66 F28
Figure 27. The LTC1147 Architecture Provides Inherent Low
Dropout Operation. This LTC1147-5 Circuit Supports a 1A Load
with the Input Voltage Only 200mV Above the Output
Figure 28. Greater Than 90% Efficiency is Obtained for Load
Currents of 20mA to 2A (VIN = 10V)
AN66-19
Application Note 66
THE LT1432: 5V REGULATOR
ACHIEVES 90% EFFICIENCY
by Carl Nelson
critical. Ordinary 5V switchers draw quiescent currents of
5mA to 15mA for these light loads. The efficiency of a 12V
to 5V converter with 10mA supply current and 1mA load
is only 4%. Clearly, some method must be provided to
eliminate the quiescent current of the switching regulator
control section.
Power supply efficiency has become a highly visible issue
in many portable battery-powered applications. Higher
efficiency translates directly to longer useful operating
time—a potent selling point for products such as notebook computers, cellular phones, data acquisition units,
sales terminals and word processors. The “holy grail” of
efficiency for 5V outputs is 90%.
An additional requirement for some systems is full shutdown of the regulator. It would be ideal if a simple logic
signal could cause the converter to turn off and draw only
a few microamperes of current.
The combination of battery form factors, their discrete
voltage steps and the use of higher voltage wall adapters
requires a switching regulator that operates with inputs
from 6V to 30V. Both of these voltages present problems
for a MOS design because of minimum and maximum gate
voltage requirements of power MOS switches.
For a number of reasons, older designs were limited to
efficiencies of 80 to 85%. High quiescent current in the
control circuitry limited efficiency at lower output currents. Losses in the power switch, inductor and catch
diode all added up to limit efficiency at moderate-to-high
output currents. Each of these areas must be addressed in
a design that is to have high efficiency over a wide output
current range.
The LT1432 was designed to address all the requirements
described above. It is a bipolar control chip that interfaces
directly to the LT1070 family of switching regulators and
is capable of operating with 6V to 30V inputs. These ICs
have a very efficient, quasisaturating NPN switch that
mimics the resistive nature of MOS transistors with much
smaller die areas. The NPN is a high frequency device with
Some portable equipment has the additional requirement
of high efficiency at extremely light loads (1mA to 5mA).
These applications have a sleep mode in which RAM is
kept alive to retain information. The instrument may spend
days or even weeks in this mode, so battery drain is
VIN
VSW
+
C1
330µF
35V
VIN
LT1271
FB
VC
C6
0.02µF
GND
D2
1N4148
R1
680Ω
C5
0.03µF
C4
0.1µF
C3
4.7µF
TANT
VOUT
5V
3A
+
L1
50µH
R2*
0.013Ω
+
D1
MBR330P
VC
< 0.3V = NORMAL MODE
> 2.5V = SHUTDOWN
OPEN = BURST MODE
MODE INPUT
DIODE
VIN
VLIM
LT1432
MODE
200pF
V+
VOUT
GND
*R2 IS MADE FROM PC BOARD COPPER TRACES
L1 = COILTRONICS CTX 50-3-MP (3A) (407) 241-7876
Figure 29. High Efficiency 5V Buck Converter
AN66-20
AN66 F29
C2
390µF
16V
Application Note 66
an equivalent voltage and current overlap time of only
10ns. Drive to the switch is automatically scaled with
switch current, so drive losses are also low. Switch and
driver losses using an LT1271 with a 12V input and a 5V,
500mA load are only about 2%.
To reduce quiescent current losses, the LT1271 is powered from the 5V output rather than from the input voltage.
This is done by pumping the supply capacitor C3 from the
output via D2. Quick minded designers will observe that
this arrangement does not self-start; accordingly, a parallel path was included inside the LT1432 to provide power
to the IC switcher directly from the input during start-up.
Equivalent quiescent supply current is reduced to about
3.5mA with this technique.
Catch diode losses cannot be reduced with IC “tricks”
unless the diode is replaced with a synchronously driven
MOS switch. This is more expensive and still requires the
diode to avoid voltage spikes during switch nonoverlap
times. The question is, is it worth it?
The following formula was developed to calculate the
improvement in efficiency when adding a synchronous
switch.
(VIN – VOUT)(Vf – RFET • IOUT)(E)2
Efficiency change =
(VIN)(VOUT)
With VIN = 10V, VOUT = 5V, Vf (diode forward voltage) =
0.45V, RFET = 0.1Ω and IOUT = 1A the improvement in
efficiency is only 2.8%. This does not take into account
the losses associated with MOS gate drive, so real
improvement would probably be closer to 2%. The
availability of low forward voltage Schottky diodes such
as the MBR330P makes synchronous switches less
attractive than they used to be.
To achieve higher efficiency during sleep, the LT1432 has
Burst Mode operation. In this mode the LT1271 is either
driven full on, or completely shut down to its micropower
state. The LT1432 acts as a comparator with hysteresis
instead of a linear amplifier. This mode reduces equivalent
input supply current to 1.3mA with a 12V battery. Battery
life with NiCd AA cells is over 300 hours with a 1mA 5V
load. Burst Mode operation increases output ripple, especially with higher output currents, so maximum load in this
mode is 100mA.
The LT1271 normally draws about 50µA to100µA in its
shutdown state. A shutdown command to the LT1432
opens all connections to the LT1271 VIN pin so its current
drain is eliminated. This leaves only the shutdown current
of the LT1432 and the switch leakage of the LT1271, which
typically add up to less than 20µA—less than the selfdischarge rate of NiCd batteries. For many applications the
on/off function is under keystroke control. Digital chips
which draw only a few microamps are available for keystroke recognition and power control.
There is no way to design around inductor losses. These
losses are minimized by using low loss cores such as
molypermalloy or ferrite, and by sizing the core to use wire
with sufficient diameter to keep resistive losses low. The
50µH inductor shown has a core loss of 200mW with type52 powdered iron material and 28mW with molypermalloy.
For a 1A load this represents efficiency losses of 4% and
0.56% respectively—a major difference. Ferrite cores
would have even lower losses than molypermalloy, but the
“moly” has such low losses that ferrites should be chosen
for other reasons, such as height, cost, mounting and the
like. DC resistance of the inductor shown is 0.02Ω. This
represents an efficiency loss of 0.4% at 1A load and 0.8%
at 2A. Significant reduction in these resistance losses
would require a somewhat larger inductor. The choice is
yours.
The LT1432 has a high efficiency current limit with a sense
voltage of only 60mV. This has a side benefit in that printed
circuit board trace material can be used for the sense
resistor. A 3A limit requires a 0.02Ω sense resistor and
this is easily made from a small section of serpentine trace.
The 60mV sense voltage has a positive temperature coefficient that tracks that of copper so that the current limit is
flat with temperature. Foldback current limiting can be
easily implemented.
The LT1432 represents a significant improvement in high
efficiency 5V supplies that must operate over a wide range
of load currents and input voltages. Its efficiency has a
very broad peak that exceeds 90%, requiring a new
definition of the “holy grail.” Logic controlled shutdown,
millipower Burst Mode operation and efficient, accurate,
current limiting make this regulator extremely attractive
for battery-powered applications.
AN66-21
Application Note 66
Regulators—Switching (Buck)
a frequency of 100kHz. Figure 33 is the efficiency plot of
the circuit. At a load current of 100mA the efficiency is at
92%; the efficiency falls to 82% at a 1A output.
Low Power (<1A)
APPLICATIONS FOR THE LTC1265
HIGH EFFICIENCY MONOLITHIC BUCK CONVERTER
by San-Hwa Chee
2.5mm Typical-Height 5V to 3.3V Regulator
Figure 34 shows the schematic for a very thin 5V to 3.3V
converter. For the LTC1265 to be able to source 500mA
output current and yet meet the height requirement, a
small value inductor must be used. The circuit operates at
a high frequency (500kHz typically) increasing the gate
charge losses. Figure 35 is the efficiency curve for this
application.
Efficiency
Figure 30 shows a typical LTC1265-5 application circuit.
The efficiency curves for two different input voltages are
shown in Figure 31. Note that the efficiency for a 6V input
exceeds 90% over a load range from less than 10mA to
850mA. This makes the LTC1265 attractive for all battery
operated products and efficiency sensitive applications.
Positive-to-Negative Converter
Besides converting from a positive input to positive output, the LTC1265 can be configured to perform a positiveto-negative conversion. Figure 36 shows the schematic
for this application.
5V to 3.3V Converter
Figure 32 shows the LTC1265 configured for 3.3V output
with 1A output current capability. This circuit operates at
VIN
5.4V TO 12V
+
2
CIN††
68µF
20V
0.1µF
10
VIN
SW
LTC1265-5
SHDN
5
6
3900pF
*COILTRONICS CTX33-4
**KRL SL-C1-OR100J
† AVX TPSE227K010
†† AVX TPSE686k020
PGND
SGND
130pF
1k
13
7
14
L1*
33µH
12
D1
MBRS130LT3
VIN = 6V
+
11
CT
ITH
SENSE –
NC
9
1000pF
AN66 F30
COILTRONICS 407-241-7876
KRL/BANTRY 603-668-3210
COUT†
220µF
10V
VOUT = 5V
95
VIN = 9V
90
85
80
75
8
SENSE +
Figure 30. High Efficiency Step-Down Converter
AN66-22
100
RSENSE**
0.1Ω
EFFICIENCY (%)
1
PWR VIN PWR VIN
VOUT
5V
1A
70
0.01
L = 33µH
VOUT = 5V
RSENSE = 0.1Ω
CT = 130pF
0.10
LOAD CURRENT (A)
1.00
AN66 F31
Figure 31. Efficiency vs Load Current
Application Note 66
VIN
5V
2
4
3
270pF
5
6
3900pF
1k
7
PWR VIN PWR VIN
VIN
SW
LTC1265-3.3
LBIN
PGND
LBOUT
SGND
CT
SHDN
NC
ITHR
SENSE –
+
13
SENSE +
0.1µF
14
CIN†
100µF
10V
L1*
47µH
VOUT
3.3V
1A
95
0.1Ω**
+
D1
MBRS130LT1
12
100
COUT††
220µF
10V
11
10
SHUTDOWN
EFFICIENCY (%)
1
90
85
80
L1 = 47µH
VOUT = 3.3V
RSENSE = 0.1Ω
CT = 270pF
9
75
8
70
1000pF
1
AN66 F32
10
100
LOAD CURRENT (mA)
COILCRAFT 708-639-6400
KRL/BANTRY 603-668-3210
*COILCRAFT D03316-473
**KRL SL-C1-OR100J
† AVX TAJD100K010
†† AVX TAJD226K010
1000
AN66 F33
Figure 32. High Efficiency 5V to 3.3V Converter
Figure 33. Efficiency vs Load Current
VIN
5V
2
4
3
51pF
5
1k 6
3300pF
7
PWR VIN PWR VIN
VIN
SW
LTC1265-3.3
LBIN
PGND
LBOUT
SGND
CT
SHDN
NC
ITHR
SENSE –
+
13
SENSE +
0.1µF
14
CIN†
15µF
10V × 2
L1*
18µH
D1
MBRS0520LT1
12
VOUT
3.3V
500mA
0.20Ω**
90
+
11
10
SHUTDOWN
9
COUT††
22µF
6.3V
×2
85
80
L1 = 18µH
VOUT = 3.3V
RSENSE = 0.20Ω
CT = 50pF
75
8
70
1000pF
1
AN66 F34
*SUMIDA CLS62-180
**KRL SL-C1-OR200J
† AVX TAJB155K010
†† AVX TAJB225K06
95
EFFICIENCY (%)
1
SUMIDA 708-956-0666
KRL/BANTRY 603-668-3210
Figure 34. 2.5mm High 5V to 3.3V Converter (500mA Output Current)
10
100
LOAD CURRENT (mA)
500
AN66 F35
Figure 35. Efficiency vs Load Current
AN66-23
Application Note 66
VIN
3.5V TO 7.5V
SHUTDOWN
1
2
4
3
220pF
5
2200pF
1k
6
7
13
0.1µF
PWR VIN PWR VIN
VIN
14
SW
LTC1265-5
LBIN
PGND
LBOUT
SGND
CT
SHDN
SENSE –
D1
MBRS130LT3
12
RSENSE**
0.1Ω
+
CIN†
22µF
25V
×2
+
L1*
47µH
VIN (V) I OUT(MAX) (mA)
3.5
360
4.0
430
5.0
540
6.0
630
7.0
720
7.5
740
VOUT
–5V
11
† AVX TPSD226K025
†† AVX TPSD106K010
100k
10
*L1 SELECTION
MANUFACTURER
COILTRONICS
COILCRAFT
DALE
SUMIDA
**KRL SL-C1-OR100J
9
NC
ITHR
TP0610L
COUT††
100µF/10V
8
SENSE +
1000pF
PART NO.
CTX50-4
D03316-473
LPT4545-500LA
CD75-470
AN66 F36
Figure 36. Positive (3.5 to 7.5V) to Negative (– 5V) Converter
Regulators—Switching (Boost)
HIGH OUTPUT CURRENT BOOST REGULATOR
by Dimitry Goder
switch currents of up to 10A are available, providing a
convenient means for power conversion over wide input
and output voltage ranges. If higher switch currents are
required, a controller with an external power MOSFET is a
better choice.
Low voltage switching regulators are often implemented
with self-contained power integrated circuits featuring a
PWM controller and an onboard power switch. Maximum
Figure 37 shows an LTC1147-based 5V to 12V converter
with 3.5A peak output current capability. The LTC1147 is
a micropower controller that uses a constant off-time
Medium Power (1A to 4A)
VIN
5V
+
L1
15µH
C6
220µF
10V
×2
D2
BAT54
+
R5
100Ω
D1
MBR735
C7
3.3µF
R6
56k
1
2
3
C1
3300pF
C2
180pF
R1
510Ω
4
PDRIVE
VIN
CT
ITH
U1
LTC1147
SENSE –
GND
VFB
SENSE +
8
Q3
TP0610L
Q2
IRL2203
R2
11.5k
1%
Q1
VN2222LL
7
+
C5
150µF
16V
×2
5
R4
100Ω
R7
0.01Ω
2%
R3
100Ω
Figure 37. LTC1147-Based 5V to 12V Converter
AN66-24
R8
100k
1%
C4
100pF
6
C3
0.01µF
C5, C6 SANYO 0S-CON
EFFICIENCY AT 3A ≥ 90%
VOUT
12V/3A
3.5A PEAK
AN66 F37
Application Note 66
architecture, eliminating the need for external slope compensation. Current mode control allows fast transient
response and cycle-by-cycle current limiting. A maximum
voltage of only 150mV across the current-sense resistor
R7 optimizes performance for low input voltages.
When Q2 turns on, current starts building up in inductor
L1. This provides a ramping voltage across R7. When
this voltage reaches a threshold value set internally in the
LTC1147, Q2 turns off and the energy stored in L1 is
Regulators—Switching (Boost)
Low Power (<1A)
transferred to the output capacitor C5. Timing capacitor
C2 sets the operating frequency. The controller is powered from the output through R5 providing 10V of gate
drive for Q2. This reduces the MOSFET’s ON resistance
and allows efficiency to exceed 90% even at full load. The
feedback network comprising R2 and R8 sets the output
voltage. Current sense resistor R7 sets the maximum
output current; it can be changed to meet different circuit
requirements.
space. Figure 39 shows the circuit’s efficiency, which can
reach 89% on a 5V input.
APPLICATIONS FOR THE LT1372 500kHz
SWITCHING REGULATOR
by Bob Essaff
The reference voltage on the FB pin is trimmed to 1.25V
and the output voltage is set by the R1/R2 resistor divider
ratio (VOUT = VREF • (R1/R2 + 1). R3 and C2 frequency
compensate the circuit.
Boost Converter
Positive-to-Negative Flyback with Direct Feedback
The boost converter in Figure 38 shows a typical LT1372
application. This circuit converts an input voltage, which
can vary from 2.7V to 11V, into a regulated 12V output.
Using all surface mount components, the entire boost
converter consumes only 0.5 square inches of board
A unique feature of the LT1372 is its ability to directly
regulate negative output voltages. As shown in the positive-to-negative flyback converter in Figure 40, only two
resistors are required to set the output voltage. The
reference voltage on the NFB pin is – 2VREF, making
VOUT = – 2VREF • (R2/R3 + 1). Efficiency for this circuit
reaches 72% on a 5V input.
D1
MBRS120T3
L1*
4.7µH
VOUT†
12V
5
OFF
VIN
ON 4 S/S
VSW
LT1372/LT1377
+
C1**
22µF
FB
GND
6, 7
8
R1
53.6k
1%
2
+
VC
R2
6.19k
1%
1
C2
0.047µF
R3
2k
C3
0.0047µF
AN66 F38
*COILCRAFT DO1608-472 (4.7µH) OR
COILCRAFT DT3316-103 (10µH) OR
SUMIDA CD43-4R7 (4.7µH) OR
SUMIDA CD73-100KC (10µH) OR
**AVX TPSD226M025R0200
100
VIN = 5V
90
C4**
22µF
EFFICIENCY (%)
5V
80
70
60
†
MAX IOUT
L1
IOUT
4.7µH 0.25A
10µH 0.35A
Figure 38. 5V to 12V Boost Converter
50
0.01
0.1
OUTPUT CURRENT (A)
1
AN66 F39
Figure 39. 12V Output Efficiency
AN66-25
Application Note 66
Dual Output Flyback with Overvoltage Protection
Multiple-output flyback converters offer an economical
means of producing multiple output voltages, but the
power supply designer must be aware of cross regulation
issues, which can cause electrical overstress on the supply and loads. Figure 41 is a dual-output flyback converter
with overvoltage protection. Typically, in multiple-output
flyback designs only one output is voltage sensed and
regulated. The remaining outputs are “quasi-regulated” by
the turns ratios of the transformer secondary. Cross
regulation is a function of the transformer used and is a
measure of how well the quasi-regulated outputs maintain
VIN
2.7V TO 16V
C1
22µF
5
ON 4
OFF
VIN
S/S
VSW
8
D2
P6KE-15A
D3
1N4148
2 T1 4
1
3
D1
MBRS130LT3
LT1372
NC
2
FB
NFB
GND
VC
1
IOUT
0.3A
0.5A
0.75A
6, 7
C2
0.047µF
VIN
3V
5V
9V
C4
47µF
–VOUT
R2 –5V
2.49k
1%
R3
2.49k
1%
VIN = 5V
25
20
VOUT
15
10
5
0
–5
–10
AN66 F40
–VOUT
–15
T1 = COILTRONICS CTX10-2
COILTRONICS (407) 241-7876
R1
2k
C3
0.0047µF
3
30
+
VOUT (V)
+
regulation under varying load conditions. For evenly loaded
outputs, as shown in Figure 42, cross regulation can be
quite good, but when the loads differ greatly, as in the case
of a load disconnect, there may be trouble. Figure 43
shows that when only the 15V output is voltage sensed,
the – 15V quasi-regulated output exceeds – 25V when
unloaded. This can cause electrical overstress on the
output capacitor, output diode and the load when reconnected. Adding output voltage clamps is one way to fix the
problem but the circuit in Figure 41 eliminates this requirement. This circuit senses both the 15V and – 15V outputs
and prevents either from going beyond its regulating
value. Figure 44 shows the unloaded – 15V output being
held constant. The circuit’s efficiency, which can reach
79% on a 5V input, is shown in Figure 45.
–20
–25
–30
1
Figure 40. LT1372’s Positive-to-Negative Converter
with Direct Feedback
R2
1.21k
1%
30
OFF
VOUT
15V
2, 3
P6KE-20A
5
VIN
8
VSW
FB
S/S
LT1372
NFB
GND
VC
1
C2
0.047µF
C3
0.0047µF
R3
2k
3
1N4148
6, 7
5
4
8
+
+
1
MBRS140T3
6, 7
20
C5
47µF
–VOUT
R4 –15V
12.1k
1%
VOUT
15
C4
47µF
10
VOUT (V)
C1
22µF
VIN = 5V
25
MBRS140T3
T1
ON 4
AN66 F42
R1
13k
1%
2
100
Figure 42. Cross Regulation of Figure 41’s Circuit.
VOUT and – VOUT Evenly Loaded
VIN
2.7V TO 13V
+
10
OUTPUT CURRENT (mA)
5
0
–5
–10
–15
–VOUT
–20
–25
T1 = DALE LPE-4841-100MB
DALE (605) 665-9301
R5
2.49k
1%
–30
1
10
OUTPUT CURRENT (mA)
100
AN66 F41
AN66 F43
Figure 41. LT1372 Dual Output Flyback Converter
with Overvoltage Protection
AN66-26
Figure 43. Cross Regulation of Figure 41’s Circuit.
– VOUT Unloaded; Only VOUT Voltage Sensed
Application Note 66
85
30
20
EFFICIENCY (%)
10
5
0
–5
–10
VIN = 9V
80
VOUT
15
VOUT (V)
VOUT = ±15V
VIN = 5V
25
VIN = 5V
75
VIN = 3V
70
–VOUT
–15
65
–20
–25
60
–30
1
10
OUTPUT CURRENT (mA)
5
100
10
100
200
OUTPUT CURRENT (mA)
AN66 F45
AN66 F44
Figure 45. Efficiency of Dual Output Flyback Converter
in Figure 41
Figure 44. Cross Regulation of Figure 41’s Circuit.
–VOUT Unloaded; Both –VOUT and VOUT Sensed
Regulators—Switching
(Buck/Boost)
±5V CONVERTER USES OFF-THE-SHELF
SURFACE MOUNT COIL
By Mitchell Lee and Kevin Vasconcelos
VIN = 9V
FB
LT1176CS-5
VSW
VIN
VC
The circuit shown in Figure 46 fulfills a recent customer
requirement for a 9V to 12V input, 5V/800mA and
– 5V/100mA output converter. It employs a 1:1 overwinding on what is ostensibly a buck converter to provide a
– 5V output. The optimum solution would be a bifilar
wound coil with heavy gauge wire for the main 5V output
and smaller wire for the overwinding. To avoid a custom
coil design, an off-the-shelf JUMBO-PACTM quadrifilar
wound coil is used. This family of coils is wound with
GND
+
100µF
Single-output switching regulator circuits can often be
adapted to multiple output configurations with a minimum
of changes, but these transformations usually call for
custom wound inductors. A new series of standard inductors,1 featuring quadrifilar windings, allows power supply
designers to take advantage of these modified circuits but
without the risks of a custom magnetics development
program.
CTX100-5P
2k
10nF
1
8
2
7
3
6
4
5
5V
800mA
+
470µF
1N5818
+
1N5818
470µF
–5V
100mA
AN66 F46
Figure 46. 5V Buck Converter with – 5V Overwinding
1:1:1:1 sections. In the application of Figure 46, three
sections are paralleled for the main 5V winding and the
remaining section is used for the – 5V output. This concentrates the copper where it is needed most— on the high
current output.
Efficiency with the outputs loaded at 500mA and – 50mA
is over 80%. Minimum recommended load on the – 5V
output is 1mA to 2mA, and the – 5V load current must
always be less than the 5V load current.
1 JUMBO-PAC is a trademark of Coiltronics Inc. (407) 241-7876.
AN66-27
Application Note 66
SWITCHING REGULATOR PROVIDES
CONSTANT 5V OUTPUT FROM 3.5V TO 40V
INPUT WITHOUT A TRANSFORMER
by Brian Huffman
A common switching regulator requirement is to produce
a constant output voltage from an input voltage that varies
above or below the output voltage. This is particularly
important for extending battery life in battery-powered
applications. Figure 47 shows how an LT1171 switching
regulator IC, two inductors and a “flying” capacitor can
generate a constant output voltage that is independent of
input voltage variations. This is accomplished without the
use of a transformer. Inductors are preferred over transformers because they are readily available and more
economical.
The circuit in Figure 47 uses the LT1171 to control the
output voltage. A fully self-contained switching regulator
IC, the LT1171 contains a power switch as well as the
control circuitry (pulse-width modulator, oscillator, reference voltage, error amplifier and protection circuitry). The
power switch is an NPN transistor in a common-emitter
configuration; when the switch turns on, the LT1171’s
VSW pin is connected to ground. This power switch can
handle peak switch currents of up to 2.5A.
C2
150µF
50V
5
VSW
LT1171
FB
GND
+
3
C1
56µF
50V
VOUT
5V
0.5A
L2
50µH
VIN
VIN
(3.5V
TO 40V)
D1
MBR350
+
L1
50µH
4
R2
3.01k
1%
+
2
VC
C3
470µF
16V
Figure 48 shows the operating waveforms for the circuit.
In this architecture the capacitor C2 serves as the single
energy transfer device between the input voltage and
output voltage of the circuit. While the LT1171 power
switch is off, diode D1 is forward biased, providing a path
for the currents from inductors L1 and L2. Trace A shows
inductor L1’s current waveform and trace B is L2’s current
waveform. Observe that the inductor current waveforms
occur on top of a DC level. The waveforms are virtually
identical because the inductors have identical inductance
values and the same voltages are applied across them. The
current flowing through inductor L1 is not only delivered
to the load but is also used to charge C2. C2 is charged to
a potential equal to the input voltage.
When the LT1171 power switch turns on, the VSW pin is
pulled to ground and the input voltage is applied across the
inductor L1. At the same time, capacitor C2 is connected
across inductor L2. Current flows from the input voltage
source through inductor L1 and into the LT1171. Trace C
shows the voltage at the VSW pin and Trace D is the current
flowing through the power switch. The catch diode (D1) is
reverse biased and capacitor C2’s current also flows
through the switch, through ground and into inductor L2.
During this interval C2 transfers its stored energy into
inductor L2. After the switch turns off the cycle is repeated.
Another advantage of this circuit is that it draws its input
current in a triangular waveshape (see Trace A in Figure
48). The current waveshape of the input capacitor is
identical to the current waveshape of inductor L1 except
that the capacitor’s current has no DC component. This
type of ripple injects only a modest amount of noise into
the input lines because the ripple does not contain any
sharp edges.
1
R1
1k
C4
1µF
R3
1.00k
1%
AN66 F47
C1 = NICHICON (AL) UPL1H560MEH, ESR = 0.250Ω, IRMS = 360mA
C2 = NICHICON (AL) UPL1H151MPH, ESR = 0.100Ω, IRMS = 820mA
C3 = NICHICON (AL) UPL1C471MPH, ESR = 0.090Ω, IRMS = 770mA
L1, L2 = COILTRONICS CTX50-4, DCR = 0.090Ω,
COILTRONICS (407) 241-7876
A = 1A/DIV
IL1, IC1
B = 1A/DIV
IL2
C = 10V/DIV
VSW
D = 1A/DIV
ISW
EQUATION 1: VOUT = 1.25V (1 + R2/R3)
Figure 47. LT1171 Provides Constant 5V Output from
3.5V to 40V Input. No Transformer Is Required
AN66-28
5µs/DIV
AN66 F48
Figure 48. LT1171 Switching Waveforms
Application Note 66
80
1.2
75
1.0
EFFICIENCY
IOUT(MAX)
0.8
70
0.6
65
0.4
60
0.2
55
0.0
The output voltage is controlled by the LT1171 internal
error amplifier. This error amplifier compares a fraction
of the output voltage, via the R1 to R2 divider network
shown in Figure 47, with an internal 1.25V reference
voltage, and varies the duty cycle until the two values are
SWITCHING REGULATOR PROVIDES
±15V OUTPUT FROM AN 8V TO 40V INPUT
WITHOUT A TRANSFORMER
by Brian Huffman
Many systems derive ±15V supplies for analog circuitry
from an input voltage that may be above or below the 15V
output. The split supply requirement is usually fulfilled by
a switcher with a multiple-secondary transformer or by
multiple switchers. An alternative approach, shown in
Figure 50, uses an LT1074 switching regulator IC, two
inductors and a “flying” capacitor to generate a dualoutput supply that accepts a wide range of input voltages.
This solution is particularly noteworthy because it uses
only one switching regulator IC and does not require a
transformer. Inductors are preferred over transformers
because they are readily available and more economical.
The operating waveforms for the circuit are shown in
Figure 51. During the switching cycle, the LT1074’s VSW
0
5
10
15 20 25 30
INPUT VOLTAGE (V)
35
EFFICIENCY (%)
This circuit can deliver an output current of 0.5A at a 3.5V
input voltage. This rises to 1A as input voltage is increased. Above 20V, higher output currents can be achieved
by increasing the values of inductors L1 and L2. Larger
inductances store more energy, providing additional current to the load. If 0.5A of output current is insufficient, use
a higher current part, such as the LT1170.
equal. (The duty cycle is determined by multiplying the
switch ON time by the switching frequency.) The RC
network (R1 and C4 in Figure 47) connected to the VC pin
provides sufficient compensation to stabilize this control
loop. Equation 1 (see Figure 47) can be used to determine
the output voltage.
IOUT(MAX) (A)
Figure 49 shows the efficiency of this circuit for a 0.5A load
and maximum output current for various input voltages.
The two main loss elements are the output diode (D1) and
the LT1171 power switch. A Schottky diode is chosen for
its low forward voltage drop; it introduces a 10% loss,
which is relatively constant with input voltage variations.
At low input voltages the efficiency drops because the
LT1171 power switch’s saturation voltage becomes a
higher percentage of the available input supply.
50
40
AN66 F49
Figure 49. Efficiency and Load Characteristics
for Various Input Voltages
pin swings between the input voltage (VIN) and the negative output voltage (– VOUT). (The ability of the LT1074’s
VSW pin to swing below ground is unusual—most other
5-pin buck switching regulator ICs cannot do this.) Trace
A shows the waveform of the VSW pin voltage and Trace B
is the current flowing through the power switch.
While the LT1074 power switch is on, current flows from
the input voltage source through the switch, through
capacitor C2 and inductor L1 (Trace C), and into the load.
A portion of the switch current also flows into inductor L2
(Trace D). This current is used to recharge C2 and C4
during the switch OFF time to a potential equal to the
positive output voltage (VOUT). The current waveforms for
both inductors occur on top of a DC level.
The waveforms are virtually identical because the inductors have identical values and because the same voltage
potentials are applied across them during the switching
cycles.
AN66-29
Application Note 66
C2
470µF
25V
5
VIN
VSW
4
L2
50µH
GND
VIN
8V
TO 40V
C1
1000µF
50V
3
+
VC
FB
L1
50µH
+
VR1
LT1074
C6
0.01µF
D1
MUR410
R4
20k
1
2
R1
3.3k
C5
0.01µF
VOUT
15V
0.5A
C7
0.01µF
R5
20k
R2
7.50k
1%
C3
470µF
25V
+
C4
470µF
25V
+
R3
1.30k
1%
EQUATION 1: VOUT = 2.21V* (1 + R2/R3)
VOUT = – VOUT
C1 = NICHICON UPL1H102MRH
C2, C3, C4 = NICHICON UPL1E471MPH
D1, D2 = MOTOROLA MUR410
L1, L2 = COILTRONICS CTX50-2-52 (407) 241-7876
D2
MUR410
– VOUT
–15V
0.5A
AN66 F50
Figure 50. Schematic Diagram for ±15V Version
interval the voltage on the VSW pin is equal to a diode drop
below the negative output voltage (– VOUT). L2’s current
then circulates between both D1 and D2, charging C2 and
C4. The energy stored in L1 is used to replace the energy
lost by C2 and C4 during the switch ON time. Trace G is
capacitor C2’s current waveform. Capacitor C4’s current
waveform (Trace F) is the same as diode D2’s current less
the DC component. Assuming that the forward voltage
drops of diodes D1 and D2 are equal, the negative output
voltage (– VOUT) will be equal to the positive output
voltage (VOUT). After the switch turns on again the cycle
is repeated.
A = 20V/DIV
VSW
B = 2A/DIV
ISW, IC1
C = 1A/DIV
IL1, IC3
D = 1A/DIV
IL2
E = 1A/DIV
ID1, IC3
Figure 52 shows the excellent regulation of the negative
output voltage for various output currents. The negative
F = 1A/DIV
ID2, IC4
15.3
15.2
G = 1A/DIV
IC2
–VOUT (V)
15.1
IOUT = 0.5A
15.0
5µs/DIV
AN66 F51
Figure 51. LT1074 Switching Waveforms
When the switch turns off, the current in L1 and L2 begins
to ramp downward, causing the voltages across them to
reverse polarity and forcing the voltage at the VSW pin
below ground. The VSW pin voltage falls until diodes D1
(Trace E) and D2 (Trace F) are forward biased. During this
AN66-30
14.9
14.8
IOUT = –IOUT
14.7
14.6
0 0.05 0.1 0.15 0.2 0.25 0.3 0.35 0.4 0.45 0.5
–IOUT (A)
AN66 F52
Figure 52. – 15V Output Regulation Characteristics
Application Note 66
output voltage tracks the positive supply (VOUT) within
200mV for load variations from 50mA to 500mA. Negative
output load current should not exceed the positive output
load by more than a factor of 4; the imbalance causes loop
instabilities. For common load conditions the two output
voltages track each other perfectly.
in Figure 50, with an internal 2.21V reference voltage and
then varies the duty cycle until the two values are equal.
The RC network (R1 and C5 in Figure 50) connected to the
VC pin along with the R4/R5 and C6/C7 network provides
sufficient compensation to stabilize the control loop. Equation 1 can be used to determine the output voltage.
Another advantage of this circuit is that inductor L1 acts as
both an energy storage element and as a smoothing filter
for the positive output (VOUT). The output ripple voltage
has a triangular waveshape whose amplitude is determined by the inductor ripple current (see trace C of Figure
51) and the ESR (effective series resistance) of the output
capacitor (C3). This type of ripple is usually small so a post
filter is not necessary.
Figure 54 shows the circuit’s – 5V load regulation characteristics and Figure 55 shows its efficiency.
5.7
5.6
5.5
5.4
–VOUT (V)
Figure 53 shows the efficiency for a 0.5A common load at
various input voltages. The two main loss elements are the
output diodes (D1 and D2) and the LT1074 power switch.
At low input voltages, the efficiency drops because the
switch’s saturation voltage becomes a higher percentage
of the available input supply.
Refer to the schematic diagram in Figure 56 for modified
component values to provide ±5V at 1A.
5.3
IOUT = 1A
5.2
5.1
5.0
IOUT = – IOUT
4.9
4.8
The output voltage is controlled by the LT1074 internal
error amplifier. This error amplifier compares a fraction of
the output voltage, via the R2 to R3 divider network shown
0
0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0
–IOUT (A)
AN66 F54
Figure 54. – 5V Output Regulation Characteristics
75
75
70
65
EFFICIENCY (%)
EFFICIENCY (%)
70
60
65
60
55
55
50
0
5
10
15
20
25
30
35
40
50
0
INPUT VOLTAGE (V)
AN66 F53
Figure 53. ±15V Efficiency Characteristics with
0.5A Common Load
5
10
15
20
25
30
35
40
INPUT VOLTAGE (V)
AN66 F55
Figure 55. ±5V Efficiency Characteristics with 1A Common Load
AN66-31
Application Note 66
C2
680µF
16V
5
VSW
VIN
4
GND
VIN
8V
TO 40V
C1
1000µF
50V
VC
3
FB
L1
50µH
+
L2
50µH
VR1
LT1074
D1
MBR360
1
2
C7
0.01µF
C6
0.01µF
R4
20k
R5
20k
R1
2k
C5
0.033µF
+
VOUT
5V
1A
R2
2.80k
1%
C3
680µF
16V
+
C4
680µF
16V
+
R3
2.21k
1%
EQUATION 1: VOUT = 2.21V* (1 + R2/R3)
VOUT = – VOUT
C1 = NICHICON UPL1H102MRH
C2, C3, C4 = NICHICON UPL1C681MPH
D1, D2 = MOTOROLA MBR360
L1, L2 = COILTRONICS CTX50-2-52 (407) 241-7876
D2
MBR360
AN66 F56
– VOUT
–5V
1A
Figure 56. Schematic Diagram for ±5V Version
Regulators—Switching
(Inverting)
HIGH EFFICIENCY 12V TO –12V CONVERTER
by Milton Wilcox and Christophe Franklin
It is difficult to obtain high efficiencies from inverting
switching regulators because the peak switch and inductor currents must be roughly twice the output current.
Furthermore, the switch node must swing twice the input
voltage (24V for a 12V inverting converter). The adjustable
version of the LTC1159 synchronous stepdown controller
is ideally suited for this application, producing a combination of better than 80% efficiency, low quiescent current
and 20µA shutdown current.
The 1A circuit shown in Figure 57 exploits the high inputvoltage capability of the LTC1159 by connecting the controller ground pins to the – 12V output. This allows the
simple feedback divider between ground and the output
(comprising R1 and R2) to set the regulated voltage, since
the internal 1.25V reference rides on the negative output.
The inductor connects to ground via the 0.05Ω currentsense resistor.
AN66-32
A unique EXT VCC pin on the LTC1159 allows the MOSFET
drivers and control circuitry to be powered from the output
of the regulator. In Figure 57 this is accomplished by
grounding EXT VCC, placing the entire 12V output voltage
across the driver and control circuits (remember the
ground pins are at – 12V). This is permissible with the
LTC1159, which allows a maximum of 13V between the
Sense and Ground pins. During start-up or short-circuit
conditions, operating power is supplied by an internal
4.5V low dropout linear regulator. This start-up regulator
automatically turns off when the output falls below – 4.5V.
A cycle of operation begins when Q1 turns on, placing the
12V input across the inductor. This causes the inductor
current to ramp to a level set by the error amplifier in the
LTC1159. Q1 then turns off and Q2 turns on, causing the
current stored in the inductor to flow to the – 12V output.
At the end of the 5µs off-time (set by capacitor CT), Q2
turns off and Q1 resumes conduction. With a 12V input the
duty cycle is 50%, resulting in a 100kHz operating frequency.
Application Note 66
INPUT
+30%
12V –10%
+
1N4148
0.1µF
Q1
Si9435
330µF
35V
NICHICON
UPL1V331M
0.15µF
2
3
PGATE
VIN
15
6
CT
390pF
7
6800pF
8
VCC
CT
+
1N5818
SGND
ITH
VFB
SENSE –
SENSE +
MBRS140
Q2
Si9410
10
200pF
1k
100Ω
OUTPUT
–12V
1A
R1
10.5k
9
Q4
2N7002
Q3
TP0610L
L1
100µH
DALE
TJ4-100-1µ
11
1000pF
5V OR 3.3V
R2
90.9k
150µF
16V
OS-CON
×2
100Ω
0.05Ω
20k
510k
SHDN2
16
14
EXT VCC
LTC1159
4
13
PDRIVE
NGATE
3.3µF
5
12
PWR GND
VCC
0.1µF
SHUTDOWN
CAP
+
1
5.1V
1N5993
AN66 F57
Figure 57. LTC1159 Converts 12V to –12V at 1A
Complete shutdown is achieved by pulling the gate of Q3
low. Q3, which can be interfaced to either 3.3V or 5V logic,
creates a 5V shutdown signal referenced to the negative
output voltage to activate the LTC1159 Shutdown 2 pin.
Additionally, Q4 offsets the VFB pin to ensure that Q1 and
Q2 remain off during the entire shutdown sequence. In
shutdown conditions, 40µA flows in Q3 and only 20µA is
taken from the 12V input.
100
90
EFFICIENCY (%)
The LTC1159, like other members of the LTC1148 family,
automatically switches to Burst Mode operation at low
output currents. Figure 57’s circuit enters Burst Mode
operation below approximately 200mA of load current.
This maintains operating efficiencies exceeding 65% over
two decades of load current range, as shown in Figure 58.
Quiescent current (measured with no load) is 1.8mA.
80
70
60
50
10
100
OUTPUT CURRENT (mA)
1000
AN66 F58
Figure 58. Efficiency Plot of Figure 57’s Circuit
AN66-33
Application Note 66
REGULATED CHARGE PUMP POWER SUPPLY
by Tommy Wu
The circuit shown in Figure 59 uses an LTC1044A charge
pump inverter to convert a 5V input to a – 1.7V potential as
required for a certain LCD panel. Output regulation is
provided by a novel feedback scheme, which uses components Q1, R1 and R2. Without feedback the charge pump
would simply develop approximately – 5V at its output.
With feedback applied, VOUT charges in the negative
direction until the emitter of Q1 is biased by the divider
comprising R1 and R2. Current flowing in the collector
tends to slow the LTC1044A’s internal oscillator, reducing
the available output current. The output is thereby maintained at a constant voltage.
In this application less than 5mA output current is required. As a result, charge pump capacitor C1 is reduced
to 1µF from the usual 10µF. Curves of output voltage with
and without feedback are shown in Figure 60. The equivalent output impedance of the charge pump is reduced from
approximately 100Ω to 5Ω.
A variety of output voltages within the limits of the curve
in Figure 60 can be set by simply adjusting the VBE
multiplier action of Q1, R1 and R2. Tighter regulation or
a higher tolerance could be obtained by adding a reference or additional gain, at the expense of increased
complexity and cost.
–5
VIN = 5V
–4
8
7
2
+
C1
1µF
3
4
LTC1044A
5V
INPUT
1µF
+
6
Q1*
ZTX384
5
R1
47k
R2
100k
–1.7V
OUTPUT
*ZETEX (516) 543-7100
–3
–2
WITH FEEDBACK
–1
10µF
–0
+
AN66 F59
Figure 59. Regulated Charge Pump
NO FEEDBACK
OUTPUT (V)
1
0
5
10
LOAD (mA)
15
20
AN66 F60
Figure 60. Effect of Feedback on Output Voltage
AN66-34
Application Note 66
The LTC1174 is an 8-pin SO “user-friendly” step-down
converter. (A PDIP package is also available.) Only four
external components are needed to construct a complete
high efficiency converter. With no load it requires only
130µA of quiescent current; this decreases to a mere 1µA
upon shutdown. The LTC1174 is protected against output
shorts by an internal current limit, which is pin selectable
to either 340mA or 600mA. This current limit also sets the
inductor’s peak current. This allows the user to optimize
the converter’s efficiency depending upon the output
current requirement.
In dropout conditions, the internal 0.9Ω (at a supply
voltage of 9V) power P-channel MOSFET switch is turned
on continuously (DC), thereby maximizing the life of the
battery source. (Who says a switcher has to switch?) In
addition to the features already mentioned, the LTC1174
boasts a low-battery detector. All versions function down
to an input voltage of 4V and work up to an absolute
maximum of 13.5V. For extended input voltage, high
voltage parts are also available that can operate up to an
absolute maximum of 18.5V.
If higher output currents are desired Pin 7 (IPGM) can be
connected to VIN. Under this condition the maximum
load current is increased to 450mA. The resulting circuit
and efficiency curves are shown in Figures 63 and 64
respectively.
100
L = 100µH
VOUT = 5V
IPGM = 0V
95
EFFICIENCY (%)
LTC1174: A HIGH EFFICIENCY BUCK CONVERTER
by San-Hwa Chee and Randy Flatness
VIN = 6V
90
85
80
75
70
10
1
100
200
LOAD CURRENT (mA)
AN66 F62
Figure 62. Efficiency vs Load Current
VIN
9V
6
VIN
3
2
5V Output Applications
7
Figure 61 shows a practical LTC1174-5 circuit with a
minimum of components. Efficiency curves for this circuit
at two different input voltages are shown in Figure 62. Note
that the efficiency is 94% at a supply voltage of 6V and load
current of 175mA. This makes the LTC1174 attractive to all
power sensitive applications and shows clearly why switching regulators are gaining dominance over linear regulators in battery-powered devices.
VIN = 9V
LBIN
+
SHDN
100µF*
20V
8
0.1µF
LTC1174-5
1
LBOUT
VOUT
IPGM
SW
5
GND
4
100µH†
1N5818
5V
425mA
+
220µF*
10V
* SANYO OS-CON
† COILTRONICS CTX100-4
COILTRONICS (407) 241-7876
AN66 F63
Figure 63. Typical Application for Higher Output Currents
100
6
+
VIN
3
8
100µF*
20V
0.1µF
LBIN
SHDN
LTC1174-5
2
1
LBOUT
VOUT
7
IPGM
SW
5
GND
4
100µH†
+
1N5818
EFFICIENCY (%)
95
VIN
9V
90
VIN = 6V
VIN = 9V
85
80
5V
175mA
220µF*
10V
AN66 F61
* SANYO OS-CON
† COILTRONICS CTX100-4
COILTRONICS (407) 241-7876
Figure 61. Typical Application for Low Output Currents
L = 100µH
VOUT = 5V
IPGM = VIN
COILTRONICS = CTX100-4
75
70
1
10
100
500
LOAD CURRENT (mA)
AN66 F64
Figure 64. Efficiency vs Load Current
AN66-35
Application Note 66
More Applications
A 5V to 3.3V Converter
Positive-to-Negative Converter
The LTC1174-3.3 is ideal for applications that require 3.3V
at less than 450mA. A minimum board area surface mount
3.3V regulator is shown in Figure 66. Figure 67 shows that
this circuit can achieve efficiency greater than 85% for
load currents between 5mA and 450mA.
The LTC1174 can easily be set up for a negative output
voltage. The LTC1174-5 is ideal for – 5V outputs as this
configuration requires the fewest components. Figure 65
shows the schematic for this application with low-battery
detection capability. The LED will turn on at input voltages
below 4.9V. The efficiency of this circuit is 81% at an input
voltage of 5V and output current of 150mA.
+
6
VIN
INPUT VOLTAGE
4V TO 12.5V
15µF*
25V
×3
0.1µF
7
8
IPGM
SHDN
LTC1174-3.3
3
1
LBIN
VOUT
INPUT VOLTAGE
4V TO 7.5V
+
6
4.7k
+
0.1µF
VIN
270k
LOWBATTERY
INDICATOR
7
2
3
IPGM
SHDN
33µF*
16V
×2
2
LBOUT
SW
GND
4
8
LTC1174-5
1
LBOUT
VOUT
LBIN
SW
GND
4
39k
5
1N5818
+
VOUT
3.3V
450mA
33µF**
16V
×2
* (3) AVX TPSD156K025
** (2) AVX TPSD336K016
†
COILTRONICS CTX50-4
COILTRONICS (407) 241-7876
50µH†
1N5818
50µH†
5
+
* AVX TPSD336K016
† COILTRONICS CTX50-4
COILTRONICS (407) 241-7876
33µF*
16V
VOUT
×2
–5V
150mA
AN66 F66
Figure 66. 5V to 3.3V Output Application
100
AN66 F65
Figure 65. Positive to – 5V Converter
with Low-Battery Detection
90
EFFICIENCY (%)
VIN = 5V
80
70
L = 50µH
VOUT = 3.3V
IPGM = VIN
COILTRONICS = CTX50-4
60
50
1
10
100
500
LOAD CURRENT (mA)
AN66 F67
Figure 67. Efficiency vs Load Current
AN66-36
Application Note 66
90
Regulators—Switching
(Power Factor Corrected)
300W
Typical Application
80
90
60
75
40
60
20
45
0
30
–20
15
–40
100
1K
10K
FREQUENCY (Hz)
PHASE MARGIN (DEGREES)
LOOP GAIN (dB)
150W
80
75
Figure 68 shows a 24VDC, 300W, power-factor corrected,
universal input supply. The continuous, current mode
boost PFC preregulator minimizes the differential mode
input filter size required to meet European low frequency
conducted emission standards while providing a high
power factor. The 2-transistor forward converter offers
many benefits, including low peak currents, a
nondissipative snubber, 500VDC switches and automatic
core reset guaranteed by the LT1509’s 50% maximum
duty-cycle limitation. An LT1431 and inexpensive
optoisolation are used to close the loop conservatively at
3kHz with excess phase margin (see Figure 69). Figure 70
shows the output voltage’s response to a 2A to almost 10A
10
EFFICIENCY (%)
85
THE NEW LT1508/LT1509 COMBINES POWER FACTOR
CORRECTION AND A PWM IN A SINGLE PACKAGE
by Kurk Mathews
0
100K
AN66 F69
Figure 69. Bode Plot ot the Circuit Shown in Figure 68
5A/DIV
0.5A/DIV
AN66 F85
Figure 70. Figure 68’s Response to a 2A to ≈ 10A Load Step
30W
70
100
150
200
VIN (AC)
250
300
AN66 F71
Figure 71. Efficiency Curves for Figure 68’s Circuit
current step. Regulation is maintained to within 0.5V.
Efficiency curves for output powers of 30W, 150W and
300W are shown in Figure 71. The PFC preregulator alone
has efficiency numbers of between about 87% and 97%
over line and load.
Start-up of the circuit begins with the LT1509’s VCC
bypass capacitors trickle charging through 91kΩ to
16VDC, overcoming the chip’s 0.25mA typical start-up
current (VCC ≤ lockout voltage). PFC soft start is then
released, bringing up the 382VDC bus with minimal
overshoot. As the bus voltage reaches its final value, the
forward converter comes up powering the LT1431 and
closing the feedback loop. A 3-turn secondary added to
the 70-turn primary of T1 bootstraps VCC to about
15VDC, supplying the chip’s 13mA requirement as well
as about 39mA to cover the gate current of the three FETs
and high side transformer losses. A 0.15Ω sense resistor senses input current and compares it to a reference
current (IM) created by the outer voltage loop and multiplier. Thus, the input current follows the input line voltage
and changes, as necessary, in order to maintain a constant bank voltage. The forward converter sees a voltage
input of 382VDC unless the line voltage drops out, in
which case the 470µF main capacitor discharges to
250VDC before the PWM stage is shut down. Compared
to a typical off-line converter, the effective input voltage
range of the forward converter is smaller, simplifying the
design. Additionally, the higher bus voltage provides
greater hold-up times for a given capacitor size. The high
side transformer effectively delays the turn-on spike to
the end of the built-in blanking time, necessitating the
external blanking transistor.
AN66-37
20k
1%
20k
1%
9
11
GND1
3
GND2
2
IAC
OVP
4700pF
"Y"
0.1
"X"
0.047µF
14 VSENSE
0.47µF
0.0047µF
0.1
"X"
4
CSET
0.001µF
15k
15
RSET
15V
+
16
SS1
200µF
17
VCC
1.2V
1µF
FILM
–
+
+
+
Figure 68. Schematic Diagram of 300W 24VDC Output Power Factor Corrected Universal Input Supply
Danger!! Lethal Voltages Present
20
VREF
2k
18
VC
RAMP 19
GTDR2
+
330µF
35V
0.01µF
CNY17-3
2N2222A
100pF
1N5819
220Ω
2N2222A
LT1431
1k
100pF
+
C1
1µF
400V
GND-S
5
–
+
0.1µF
3
V+
20k
20Ω
10:15
TURNS
RTOP
7
REF 8
RMID
4
100Ω
OUTPUT COM
3.4k
1%
1µF
63V
FILM
+
+
IRF840
MUR150
2000pF
10Ω,
2W
0.51Ω, 2W
RG ALLEN
RPS2
(× 2)
67µH
39T 12AWG
T150-52
4700pF
"Y"
MUR150
IRF840
470µF, 50V
NICHICON
PL12,5X25
(× 3)
+
20k
2220pF
"Y"
20k
15V
1N965
(× 2)
30.1k
1%
10k
1%
24.9k
1%
VREF
10Ω
0.0022µF
2N2907
10Ω
1µF FILM
T3
470µF
450V
382VBUS
220Ω
GND-F COMP
6
2
1
COLL
2.2k
2.2k
470Ω
IRFP450
MURH860CT
(DUAL)
330Ω
FUJI
ERA82-004
0.6A/40VR
20k
15V
1N5819
20Ω
2.2µF
50V
2.2µF
50V
6
1
CAOUT GTDR1
ERA82-004
ERA82-004
T1
20k
0.047µF
7
ISENSE
0.001µF
300pF
4.02k
1%
91k
2W
T1
RT1 = KETEMA S65T SURGE GARD
T1 = COILTRONICS CTX02-12378-2, (407) 241-7876
T3 = BI TECHNOLOGY HM41-11510, (714) 447-2345
C1 = ELECTRONIC CONCEPTS 5MP12J105K
R1 = JW MILLER/FUKUSHIMA MPC71
BR1= GENERAL INSTRUMENTS KBU6J
1µF
FILM
LT1509
13
SS2
5
4.02k
1%
R1
0.15Ω
5W
8
PKLIM MOUT
10k
12
VREF
0.1µF
1.8k
BR1
VREF
0.001µF
10
RT1
VAOUT
330k
0.1
"X"
NOTE: UNLESS OTHERWISE SPECIFIED
1. ALL RESISTORS 1/4W, 5%
2. ALL CAPACITANCE VALUES IN MICROFARADS
499k
1%
499k
1%
499k
1%
382VBUS
499k
1%
VIN
90VAC TO 1M
264VAC 1/2W
+
6A
FAST
4700pF
"Y"
+
–
AN66-38
–
VIN
T2
AN66 F68
T2 7 TURNS
0.9" × 0.005" Cu
ETD44-P
LPRI = 3.1mH
G1
FEP
30DP
(DUAL)
1000pF
10Ω
1W
24VOUT
12.5A
17 TURNS
26AWG
TRI-FILAR
17 TURNS
26AWG
TRI-FILAR
Application Note 66
Application Note 66
Regulators—Switching
(Discussion)
ADDING FEATURES TO THE BOOST TOPOLOGY
by Dimitry Goder
A boost-topology switching regulator is the simplest solution for converting a 2- or 3-cell input to a 5V output.
Unfortunately, boost regulators have some inherent disadvantages, including no short-circuit protection and no
shutdown capability. In some battery-operated products,
external chargers or adapters can raise the battery voltage
to a potential higher than the 5V output. Under this
condition a boost converter cannot maintain regulation—
the high input voltage feeds through the diode to the
output.
The circuit shown in Figure 72 overcomes these problems.
An LT1301 is used as a conventional boost converter,
preserving simplicity and high efficiency in the boost
mode. Transistor Q1 adds short-circuit limiting, true shutdown and regulation when there is a high input voltage.
When the input voltage is lower than 4V and the regulator
is enabled, Q1’s emitter is driven above its base, saturating
the transistor. As a result, the voltages on C1 and C2 are
roughly the same and the circuit operates as a conventional boost regulator.
If the input voltage increases above 4V, the internal error
amplifier, acting to keep the output at 5V, boosts the
voltage on C1 to a level greater than 1V above the input.
This voltage controls Q1 to provide the desired output with
the transistor operating as a linear pass element. The
output does not change abruptly during the switch-over
between step-up and step-down modes because it is
monitored in both modes by the same error amplifier.
Figure 73 shows efficiency versus input voltage for
5V/100mA output. The break point at 4.25V is evidence of
Q1 beginning to operate in a linear mode with an attendant
roll-off of efficiency. Below 4.25V the circuit operates as a
boost regulator and maintains high efficiency across a
broad range of input voltages.
The circuit can be shut down by pulling the LT1301’s
Shutdown pin high. The LT1301 ceases switching and Q1
automatically turns off, fully disconnecting the output.
This stays true over the entire input voltage range.
Q1 also provides overload protection. When the output is
shorted the LT1301 operates in a cycle-by-cycle current
limit. The short-circuit current depends on the maximum
switch current of the LT1301 and on the Q1’s gain,
typically reaching 200mA. The transistor can withstand
overload for several seconds before heating up. For sustained faults the thermal effects on Q1 should be carefully
considered.
R1
1.5k
100
VIN
2V TO 9V
+ C3
33µF
BOOST
RANGE
MBR0520L
VOUT
5V
100mA
Q1
ZTX788B
6
VIN
SW
7
2
SHUTDOWN
4
SENSE
SELECT
LT1301
3
5
ILIM
SHDN
8
PGND
GND
+
1
C1
47µF
+
C2
100µF
R2
3.3k
LINEAR STEPDOWN RANGE
90
EFFICIENCY (%)
L1
22µH
80
70
60
50
2
3
4
5
6
7
INPUT VOLTAGE (V)
8
9
AN66 F73
AN66 F72
Figure 72. Q1 Adds Short-Circuit Limiting, True Shutdown and Regulation
When There Is a High Input Voltage to the LT1301 in Boost Mode
Figure 73. Efficiency vs Input Voltage
for 5V/100mA Output
AN66-39
Application Note 66
SENSING NEGATIVE OUTPUTS
by Dimitry Goder
Various switching regulator circuits exist to provide positive-to-negative conversion. Unfortunately, most controllers cannot sense the negative output directly; they require
a positive feedback signal derived from the negative output. This creates a problem. The circuit presented in Figure
74 provides an easy solution.
The LT1172 is a versatile switching regulator that contains
an onboard 100kHz PWM controller and a power switching transistor. Figure 74 shows the LTC1172 configured to
provide a negative output using a popular charge pump
technique. When the switch turns on, current builds up in
the inductor. At the same time the charge on C3 is
transferred to output capacitor C4. During the switch offtime, energy stored in the inductor charges capacitor C3.
A special DC level-shifting feedback circuit consisting of
Q1, Q2, and R1 to R4 senses the negative output.
Under normal conditions Q1’s base is biased at a level
about 0.6V above ground and the current through resistor
R3 is set by the output voltage. If we assume that the base
current is negligible, then R3’s current also flows through
R2, biasing Q2’s collector at a positive voltage proportional to the negative output.
+
L1
47µH
5
C2
22µF
C5
0.1µF
1
3
VIN
VSW
4
R4
1M
U1
LT1172
Q2
2N5210
VC
GND
VFB
R1
51k
2
C1
100pF
Q1
2N5210
C3
D3
33µF 1N5819
+
R2
11k R3
1% 221k
1%
L1 = 50µH, SUMIDA CD54-470
Figure 74. 10V/20V to – 24 Converter
AN66-40
C4
33µF
D2
1N5819
VOUT
–24V
100mA
+
VIN
(10V
TO 20V)
DN66 F74
Q2 is connected as a diode and is used to compensate for
Q1’s base-emitter voltage change with temperature and
collector current. Both transistors see the same collector
current and their base-emitter voltages track quite well.
Because the base-emitter voltages cancel, the voltage
across R2 also appears on the LT1172’s Feedback pin.
The resulting output voltage is given by the following
formula:
VOUT = VFB
R3
– VBE
R2
where VFB is the LT1172 internal 1.244V reference and VBE
is Q1’s base/emitter voltage (≈ 0.6V). The VBE term in the
equation denotes a minor output voltage dependency on
input voltage and temperature. However, the variation due
to this factor is usually well below 1%.
Essentially, Q1 holds its collector voltage constant by
changing its collector current and will function properly as
long as some collector current exists. This puts the
following limitation on R1: at minimum input voltage the
current through R1 must exceed the current through R2.
This is reflected by the following inequality:
R1 < R2
VIN(MIN) – VFB – VBE
VFB
If the input voltage drops below the specified limit (e.g.,
under a slow start-up condition) and Q1 turns off, R4
provides the LT1172 Feedback pin with a positive bias and
the output voltage decreases. Without R4 the Feedback
pin would not get an adequate positive signal, forcing the
LT1172 to provide excessive output voltage and resulting
in possible circuit damage.
The feedback configuration described above is simple yet
very versatile. Only resistor value changes are required for
the circuit to accommodate a variety of input and output
voltages. Exactly the same feedback technique can be
used with flyback, “Cuk” or inverting topologies, or whenever it is necessary to sense a negative output.
Application Note 66
Regulators—Switching
(Micropower)
feedback network R3 to R5 biases the low-battery comparator input (LBI) 20mV below the reference. In this
mode the circuit operates as a conventional boost converter, sensing output voltage at the FB pin.
3-CELL TO 3.3V BUCK/BOOST CONVERTER
by Dimitry Goder
Obtaining 3.3V from three 1.2V (nominal) cells is not a
straightforward task. Since battery voltage can be either
below or above the output, common step-up or step-down
converters are inadequate. Alternatives include using more
complex switching topologies, such as SEPIC, or a switching boost regulator plus a series linear pass element.
Figure 75 presents an elegant implementation of the latter
approach.
The LT1303 is a Burst Mode switching regulator that
contains control circuitry, an onboard power transistor
and a gain block. When the input voltage is below the
output, U1 starts switching and boosts the voltage across
C2 and C3 to 3.3V. The gain block turns on Q1 because the
+ C1
33µF
6
5
For input voltages derived from three NiCd or NiMH cells,
the circuit described provides excellent efficiency and the
longest battery life. At 3.6V, where the battery spends
most of its life, efficiency exceeds 91%, leaving all alternative topologies far behind.
D1
MBR0520L
L1
20µH
VIN
2.5V TO 8V
When the input voltage increases, it eventually reaches a
point where the regulator ceases switching and the input
voltage is passed unchanged to capacitor C2. The output
voltage rises until the LBI input reaches the reference
voltage of 1.25V, at which point Q1 starts operating as a
series pass element. In these conditions the circuit functions as a linear regulator with the attending efficiency rolloff at higher input voltages.
SW
VIN
7
Q1
Si9433
+ C2
R1
100k
33µF
+ C3
R2
100Ω
330µF
×2
VOUT
3.3V
300mA
R3
200k
1%
4
U1
FB
LT1303
3
2
LBO
SHDN
1
LBI
GND
PGND
R4
1.96k
1%
8
R5
121k
1%
C3: 330µF/6.3V AVX TPS
C1, C2: 33µF/20V AVX TPS
AN66 F75
Figure 75. 3-Cell to 3.3V Buck/Boost Converter
LT1111 ISOLATED 5V SWITCHING POWER SUPPLY
by Kevin R. Hoskins
Circuit Description
Many applications require isolated power supplies. Examples include remote sensing, measurement of signals
riding on high voltages, remote battery-powered equipment, elimination of ground loops and data acquisition
systems where noise elimination is vital. In each situation
the isolated circuitry needs a floating power source. In
some cases batteries or an AC line transformer can be
used for power. Alternately, the DC/DC converter shown
here creates an accurately regulated, isolated output from
a 5V source. Moreover, it eliminates the optoisolator
feedback arrangements normally associated with fully
isolated converters.
Figure 76 shows a switching power supply that generates
an isolated and accurately regulated 5V at 100mA output.
The circuit consists of an LT1111, configured as a flyback
converter, followed by an LT1121 low dropout, micropower
AN66-41
Application Note 66
linear regulator. An LTC1145 (winner of EDN’s IC Innovation of the Year Award) provides micropower isolated
feedback.
The LT1111 is a micropower device that operates on only
400µA (max). This micropower operation is important for
energy-conscious applications. It works well with surface
mount inductors such as the Coiltronics OCTA-PAC shown
in the schematic. Although the LT1111’s internal power
switch handles up to 1A, a 100Ω resistor (R1) limits the
peak switch current to approximately 650mA. This maximizes converter efficiency. One side benefit of limiting the
peak switch current is that the circuit becomes insensitive
to inductance. The circuit operates satisfactorily with an
inductance in the range of 20µH to 50µH.
It is important that capacitor C2 have low effective series
resistance (ESR) and inductance (ESL) to minimize output
ripple voltage. Although aluminum capacitors are abundant and inexpensive, they will perform poorly in this
switcher application because of their relatively high ESR
and ESL. Two good choices that meet C2’s low ESR and
ESL requirements are the AVX TPS and Sanyo OS-CONTM
capacitor series.
Circuit Operation
The LT1111 is configured to operate as a flyback converter. The voltage on the transformer’s secondary is
rectified by D2, filtered by C2 and applied to the LT1121’s
input. As the LT1121’s input voltage continues to rise, its
output will regulate at 5V. The LT1121’s input voltage
continues increasing until the differential between input
and output equals approximately 600mV. At this point Q1
begins conducting, turning on the LTC1145 isolator. The
output of the LTC1145 goes high, turning off the converter. The feedback from the LTC1145 gates the LT1111’s
oscillator, controlling the energy transmitted to the
transformer’s secondary and the LT1121’s input voltage.
The oscillator is gated on for longer periods as the LT1121’s
load current increases. Q1’s gain and the feedback through
the LTC1145 force the converter loop to maintain the
LT1121 just above dropout, resulting in the best efficiency. The LT1121 provides current limiting as well as a
tightly regulated low noise output.
OS-CON is a trademark of SANYO Electric Co., LTD.
500VRMS
ISOLATION BARRIER
+
5V
R1
100Ω
1
2
Z1
1N5355
D1
MUR120
+
C1
10µF
3
4
C2
47µF
+
C3
10µF
D2
1N5818
IC2
LT1121CZ5
IC1
LT1111
ILIM
FB
VIN
SET
SW1
A0
SW2
GND
5V
8
7
R2
30k
TR1*
6
Q1
2N3906
5
D3
1N4148
9
8
GND2 OSC IN
7
NC
1
DIN
IC3
LTC1145
DOUT
OSOUT
10
11
C5
0.1µF
VCC
12
GND1
18
*COILTRONICS CTX20-1Z
AN66 F76
Figure 76. Circuit Generates Isolated, Regulated 5V at 100mA
AN66-42
Application Note 66
LOW NOISE PORTABLE COMMUNICATIONS
DC/DC CONVERTER
by Mitchell Lee
Portable communications products pack plenty of parts
into close proximity. Digital clock noise must be eliminated not only from the audio sections but also from the
antenna, which, by the very nature of the product, is
located only inches from active circuitry. If a switching
regulator is used in the power supply, it becomes another
potential source of noise. The LTC1174 stepdown converter is designed specifically to eliminate noise at audio
frequencies while maintaining high efficiency at low
output currents.
Figure 77 shows an all surface mount solution for a 5V,
120mA output derived from five to seven NiCd or NiMH
cells. Small input and output capacitors are used to
conserve space without sacrificing reliability. In applications where it is desired, a shutdown feature is available;
otherwise, short this pin to VIN.
5- TO 7-CELL
INPUT
+
VSW
SHDN
OFF
L1
CTX33-1
VIN
MBR0520L
LTC1174CS8
IPGM
R1
91k
C2
6.8nF
+ C3
5V
120mA
OUTPUT
The interactions of load current, efficiency and operating
frequency are shown in Figure 78. High efficiency is
maintained at even low current levels, dropping below
70% at around 800µA. No-load supply current is less than
200µA, dropping to approximately 1µA in shutdown mode.
The operating frequency rises above the telephony bandwidth of 3kHz at a load of 1.2mA. Most products draw such
low load currents only in standby mode with the audio
circuits squelched, when noise is not an issue.
The frequency curve depicted in Figure 78 was measured
with a spectrum analyzer, not a counter. This ensures that
the lowest frequency noise peak is observed, rather than
a faster switching frequency component. Any tendency to
generate subharmonic noise is quickly exposed using this
measurement method.
33µF
20V
100
100
FB
GND
R2
30k
90
Figure 77. Low Noise, High Efficiency Step-Down Regulator for
Personal Communications Devices
The LTC1174’s internal switch, which is connected between VIN and VSW, is current controlled at a peak threshold of approximately 340mA. This low peak threshold is
one of the key features that allows the LTC1174 to minimize system noise compared to other chips that carry
significantly higher peak currents, easing shielding and
filtering requirements and decreasing component stress.
10
80
70
1
EFFICIENCY (%)
AN66 F77
C1 = PANASONIC SP SERIES (201-348-4630)
C3 = AVX TPS SERIES (803-946-0690)
L1 = COILTRONICS OCTA-PAK (407-241-7876)
FREQUENCY (kHz)
ON
C1
15µF
12.5V
To conserve power and maintain high efficiency at light
loads, the LTC1174 uses Burst Mode operation. Unfortunately, this control scheme can also generate audio frequency noise at both light and heavy loads. In addition to
electrical noise, acoustical noise can emanate from capacitors and coils under these conditions. A feedforward
capacitor (C2) shifts the noise spectrum up out of the
audio band, eliminating these problems. C2 also reduces
peak-to-peak output ripple, which measures approximately
30mV over the entire load range.
60
0.1
0.1
1
10
OUTPUT CURRENT (mA)
50
100
AN66 F78
Figure 78. Parameter Interaction Chart for Figure 77’s Circuit
AN66-43
Application Note 66
APPLICATIONS FOR THE LT1302
MICROPOWER DC/DC CONVERTER
by Steve Pietkiewicz
put voltage rises above the comparator threshold. Undershoot at load step is less than 5%. The circuit’s efficiency
at various input voltages is shown in Figure 81.
2- or 3-Cell to 5V Converter
Although efficiency graphs present useful information, a
more “real world” measure of converter performance
comes from battery lifetime chart recordings. Many systems require high power for a short time, for example to
spin up a hard disk or transmit a packet of data. Figures 83,
84 and 85 present battery life data with a load profile of
50mA for 9 seconds and 550mA for 1 second, as detailed
in Figure 82. At the chart speeds used, individual 10
second events are not discernable and the battery voltage
appears as a very thick line. Figure 83 shows operating life
using a 2-cell alkaline (Eveready E91) battery. Battery
voltage (pen B) drops 400mV as the output load changes
Figure 79 shows a 2- or 3-cell to 5V DC/DC converter that
can deliver up to 600mA from a 2-cell input (2V minimum)
or up to 900mA from a 3-cell input (2.7V minimum). R1
and R2 set the output voltage at 5V. The 200pF capacitor
from FB to ground aids stability; without it the FB pin can
act as an antenna and pick up dV/dt from the switch node,
causing some instability in switch current levels at heavy
loads. L1’s inductance value is not critical; a minimum of
10µH is suggested in 2-cell applications (although this
guideline is ignored in the 2-cell to 12V circuit shown
later). Lower values typically have less DC resistance and
can handle higher current. Transient response is better
with low inductance but more output current can be had
with higher values. Peak current in Burst Mode operation
increases as inductance decreases, due to the finite response time of the current sensing comparator in the
LT1302. The Coilcraft DO3316 series inductors have been
found to be excellent in terms of performance, size and
cost but their open construction results in some magnetic
flux spray; try Coiltronics’ OCTA-PAC series if EMI is a
problem. Transient response with a load step of 25mA to
525mA is detailed in Figure 80. There is no overshoot upon
load removal because switching stops entirely when out-
VOUT
100mV/DIV
AC COUPLED
ILOAD
525mA
25mA
AN66 F80
200ms/DIV
Figure 80. Transient Response of DC/DC Converter
with 2.5V Input. Load Step is 25mA to 525mA
90
88
VIN = 4V
NC
L1
10µH
6
7
C3
0.1µF
2 CELLS
D1
C1
100µF
+
+
5
ILIM
VIN
SHDN
SW
SHUTDOWN
PGND
FB
GND
4
2
CC
6800pF
82
VIN = 3V
80
78
VIN = 2V
74
VC
RC
20k
84
76
LT1302
8
1
C2
100µF
3
EFFICIENCY (%)
86
72
R1
100k
1%
R2
301k
1%
200pF
70
1
10
100
LOAD CURRENT (mA)
1000
AN66 F81
Figure 81. Efficiency of Figure 79’s Circuit
5V
600mA
OUTPUT
C1, C2 = SANYO OS-CON
COILTRONICS (407) 241-7876
L1 = COILTRONICS CTX10-3 COILCRAFT (708) 639-2361
COILCRAFT DO3316-103
D1 = MOTOROLA MBRS130LT3
1 SEC
AN66 F79
9 SEC
550mA
50mA
AN66 F82
Figure 79. 2- or 3-Cell to 5V Converter Delivers 600mA,
1A From 3.3V Supply
AN66-44
Figure 82. Load Profile for Battery Life Curves
in Figures 83, 84 and 85
Application Note 66
5
5
4
BATTERY/OUTPUT VOLTAGE (V)
BATTERY/OUTPUT VOLTAGE (V)
PEN A
OUTPUT
3
2
PEN B
BATTERY
1
3
2
PEN B
BATTERY
1
0
0
150
120
90
60
TIME (MINUTES)
30
0
`AN66 F83
Figure 83. 2-Cell Alkaline Battery to 5V Converter with Load
Profile of Figure 82 Gives 63 Minutes Operating Life. Battery Life
Decreases When 550mA Load is Applied; Impedance is 330mΩ
When Fresh. Output Voltage Drops at 550mA Load After 63
Minutes But Converter Can Still Deliver 50mA
PEN A
OUTPUT
4
3
PEN B
BATTERY
2
1
0
9
8
7
6
5
4
3
TIME (HOURS)
2
150
120
90
60
TIME (MINUTES)
30
0
`AN66 F100
Figure 85. 2-Cell NiCd Battery to 5V Converter Shows
Dramatically Lower ESR of NiCds Compared to Alkalines. Battery
Impedance Is 80mΩ. Although the 600mA Hour NiCd Has 1/4 the
Energy of 2.4A/Hr Alkalines with 50mA/550mA Loads NiCds
Outlast Alkalines by a Factor of 2.8. Low Cell Impedance is
Maintained Until the Battery Is Completely Discharged
A 3-cell alkaline battery has a significantly longer life, as
shown in Figure 84. Note that the time scale here is one
hour per inch. Usable life is about 7.3 hours, a sevenfold
improvement over the 2-cell battery. Again, battery impedance causes the battery voltage (pen B) to drop as the
load changes from 50mA to 550mA. The increasing change
between the loaded and unloaded battery voltage over
time is due to both increased current demand on the
battery as its voltage decreases and increasing battery
impedance as it is discharged.
5
BATTERY/OUTPUT VOLTAGE (V)
PEN A
OUTPUT
4
1
0
AN66 F84
Figure 84. 3-Cell Alkaline Battery to 5V Converter with Pulsed
Load Has 7.3 Hours Operating Life
from 50mA to 550mA. Battery impedance (330mΩ when
fresh) can be derived from this data. After 63 minutes the
battery voltage drops substantially below 2V when the
output load is 550mA, causing the output voltage (pen A)
to drop. The output returns to 5V when the load drops to
50mA. The LT1302’s undervoltage lockout prevents the
battery voltage from falling below 1.5V until the battery is
completely discharged (not shown on the chart).
Replacing the 2-cell alkaline with a 2-cell NiCd (AA Gates
Millennium) battery results in a surprise shown in Figure
85. Although these AA NiCd cells have one-fourth the
energy of AA alkaline cells, operating life is 2.8 times
greater with the 50mA/550mA load profile. Dramatically
lower battery impedance (80mΩ for the NiCd versus
330mΩ for the alkaline) is the cause. Battery voltage (pen
B) drops just 100mV as the output load changes from
50mA to 550mA, compared to 400mV for alkalines. Additionally, impedance stays relatively constant over the life
of the battery. This comparison clearly illustrates the
limitations of alkaline cells in high power applications.
2-Cell to 12V Converter
Portable systems with PCMCIA interfaces often require
12V at currents of up to 120mA. Figure 86’s circuit can
AN66-45
Application Note 66
NC
L1*
3.3µH
7
+
2 CELLS
C3
0.1µF
C1
100µF
D1
6
5
VIN
ILIM
C2
33µF
+
SHUTDOWN
LT1302
8
PGND
GND
+
3
SHDN
SW
4
FB
VC
1
2
RC
20k
C2
33µF
CC
0.02µF
12V
120mA
OUTPUT
*SEE TEXT C1 = AVX TPSD107M010R0100
C2 = AVX TPSD336M025R0200
R1
100k
1%
100pF
R2
866k
1%
D1 = MOTOROLA MBRS130LT3
COILCRAFT (708) 639-2361
L1 = COILCRAFT DO3316-332
FOR 3.3V/5V INPUT USE 22µH (DO3316-223)
AN66 F86
Figure 86. 2-Cell to 12V DC/DC Converter Delivers 120mA. Changing L1’s Value Allows Operation from 3.3V/5V Supply
85
90
88
VIN = 2.5V
VIN = 3V
86
EFFICIENCY (%)
EFFICIENCY (%)
80
75
VIN = 2V
70
VIN = 5V
84
82
80
78
VIN = 3.3V
76
65
74
72
60
1
70
10
100
LOAD CURRENT (mA)
1
AN66 F87
Figure 87. 2-Cell to 12V Converter Efficiency
300
LOAD CURRENT (mA)
275
250
225
200
175
150
125
100
2.00 2.20
2.40 2.60 2.80 3.00 3.20 3.40
INPUT VOLTAGE (V)
AN66 F88
Figure 88. Maximum Load Current of 2-Cell to 12V Converter
vs Input
AN66-46
10
100
LOAD CURRENT (mA)
1000
AN66 F89
Figure 89. 3.3V/5V to 12V Converter Efficiency
generate 12V at over 120mA from a 2-cell battery. Operating the converter in continuous mode requires a higher
duty cycle than the LT1302 provides, so a very low
inductance (3.3µH) must be used in order to provide
enough output current in discontinuous mode. Efficiency
for this circuit is in the 70% to 80% range, as Figure 87’s
graph shows. Battery life at this power level would be short
with a continuous load but the most common application
for this voltage/current level, flash memory programming,
has a rather low duty factor. Maximum output current
versus input voltage is shown in Figure 88. To operate this
circuit from a 3-cell battery change L1’s value to 6.8µH.
This will result in lower peak currents, improving efficiency substantially.
Application Note 66
L1
3.3µH
5V
175mA
OUTPUT
D1
220Ω
10Ω
100k
1.5V
CELL
R1
301k
1%
2N3906
(169k FOR 3.3V)
100k
IL
SET
100k
VIN
VIN
SW1
GND
FB
SW
LT1073
56.2k
1%
LT1302
AO
FB
SHDN
VC
IL
SW2
PGND
+
C1
47µF
C2
220µF
GND
100pF
20k
4.99k
1%
+
0.1µF
0.01µF
36.5k
1%
AN66 F90
L1 = COILCRAFT DO3316-332
D1 = MOTOROLA MBRS130LT3
C1 = AVX TPSD476M016R0150
C2 = AVX TPSE227M010R0100
COILCRAFT (708) 639-2361
Figure 90. Single Cell to 5V Converter Delivers 150mA. Changing R1 to 169k Provides 3.3V at 250mA
72
70
VIN = 1.5V
68
EFFICIENCY (%)
66
64
VIN = 1.2V
62
60
58
56
54
52
50
48
1
10
100
LOAD CURRENT (mA)
1000
AN66 F91
Figure 91. Single Cell to 5V Converter Efficiency
Changing L1’s value to 22µH allows the circuit to operate
from a 3.3V or 5V supply. Up to 350mA can be generated
from 3.3V; 600mA can be delivered from a 5V input.
Efficiency, pictured in Figure 89, exceeds 80% over much
of the load range and peaks at 89% with a 5V input.
Single Cell to 5V/150mA Converter
Stand alone single -cell converters can typically provide no
more than 40mA to 50mA at 5V from a single cell. When
more power is required, the LT1302 can be used in
conjunction with a single-cell device.1 Figure 90’s circuit
operates from a single cell and delivers 5V at 150mA
output. Although the LT1302 requires a minimum VIN of
2V, single-cell operation can be achieved by powering the
LT1302 from the 5V output. At start-up VOUT is equal to the
cell voltage minus a diode drop. The LT1073 initially puts
the LT1302 in its shutdown state. The LT1073 switches
L1, causing L1’s current to alternately build up and dump
into C2. When VOUT reaches approximately 2V the LT1073’s
Set pin goes above 212mV causing AO to go low. This pulls
the LT1302’s SHDN pin low, enabling it. The output, now
booted by the much higher power LT1302, quickly reaches
2.4V. When the LT1073’s FB pin reaches 212mV its
switching action stops. The brief period when the LT1073
and LT1302 are switching simultaneously has no detrimental effect. When the output reaches 5V the LT1073 has
ceased switching. Circuit efficiency is in the 60 to 70%
range as shown in Figure 91.
3-Cell to 3.3V/12V Buck/Boost Converter
Obtaining 3.3V from three cells is not a straightforward
task; a fresh battery measures over 4.5V and a fully
depleted one 2.7V. Since battery voltage can be both
1 Williams, Jim. “200mA Output, 1.5 to 5V Converter.” Linear Technology III:1
(February, 1993) p. 17.
AN66-47
Application Note 66
VIN
2.5V TO 8V
2Ω
SHUTDOWN
SHDN
T1D
T1E
4
5
+ C3
47µF
16V
D2
SW
100k
1%
+ C1
IL
GND
13V
0.1µF
LT1302
VC
200pF
6
VIN
FB
169k
1%
7
100µF
16V
PGND
D1
24k
4700pF
3
1
T1C
T1A
8
2
IN
T1B
12V
120mA
OUT
+ 22µF
9
25V
330k
1%
LT1121
+
ADJ
SHDN
+ C2
3.3µF
GND
330µF
6.3V
150k
1%
10
AN66 F92
3.3V OUTPUT
400mA
T1 =
D1, D2 =
C1 =
C2 =
C3 =
DALE LPE-6562-A069
MOTOROLA MBRS130LT3
AVX TPSE107016R0100
AVX TPSE337006R0100
AVX TPSD476016R0150
1:3:1:1:1 TURNS RATIO
DALE (605) 665-9301
Figure 92. 3-Cell to 3.3V Buck/Boost Converter with Auxiliary 12V Regulated Output
80
VIN = 3.5V
75
VIN = 2.5V
EFFICIENCY (%)
70
65
60
VIN = 4.5V
55
50
45
40
1
10
100
LOAD CURRENT (mA)
1000
AN66 F93
Figure 93. 3.3V Buck/Boost Converter Efficiency
above and below the output, common step-up (boost) or
step-down (buck) converters are inadequate. Figure 92’s
circuit provides an efficient solution to the problem using
just one magnetics component and also provides an
auxiliary 12V output. When the LT1302’s switch is on its
SW pin goes low, causing current buildup in T1D and T1E
(windings are paralleled to achieve lower DC resistance).
D1’s anode goes to – VIN because of the phasing of T1C/
T1A relative to T1D/T1E. C1 is charged to VIN. When the
AN66-48
switch opens, SW flies high to a voltage of VIN + VOUT +
VDIODE. Energy is transferred to the output by magnetic
coupling from T1D/T1E to T1C/T1A and by current flowing
through C1. During this flyback phase, T1A/T1C has 3.3V
plus a diode drop across the windings. T1B, which has a
3:1 turns ratio, has approximately 10V to 11V impressed
upon it. T1B “stands” on the 3.3V output, resulting in
about 13V to14V at the input of the LT1121 linear regulator, which then precisely regulates the 12V output. Since
this output is not directly regulated by the LT1302, it
cannot be loaded without having at least a small load on
the directly regulated 3.3V output. The LT1121 can be
turned off by pulling its SHDN pin low, isolating the load
from the output. Figure 93 shows the circuit’s efficiency
for various input voltages.
Construction Hints
The high speed, high current switching associated with
the LT1302 mandates careful attention to layout. Follow
the suggested component placement in Figure 94 for
proper operation. High current functions are separated by
the package from sensitive control functions. Feedback
Application Note 66
VIN
R2
C3
R3
2Ω
L1
+
5
4
6
3
LT1302
C1
D1
7
2
8
1
R1
200pF
SHUTDOWN
RC
CC
+
C2
VOUT
GND (BATTERY AND LOAD RETURN)
AN66 F94
Figure 94. Suggested Component Placement for LT1302
resistors R1 and R2 should be close to the Feedback pin
(Pin 4). Noise can easily be coupled into this pin if care
is not taken. If the LT1302 is operated off a 3-cell or
higher input, R3 (2Ω) in series with VIN is recommended.
This isolates the device from noise spikes on the input
voltage. Do not install R3 if the device must operate from
a 2V input, as input current will cause the LT1302’s input
voltage to go below 2V. The 0.1µF ceramic bypass
capacitor C3 (use X7R not Z5U) should be mounted as
close as possible to the package. Grounding should be
segregated as illustrated. C3’s ground trace should not
carry switch current. Run a separate ground trace up
under the package as shown. The battery and load return
should go to the power side of the ground copper.
Adherence to these rules will result in working converters
with optimum performance.
CLOCK-SYNCHRONIZED SWITCHING REGULATOR
HAS COHERENT NOISE
by Jim Williams, Sean Gold and Steve Pietkiewicz
attendant switching noise, although variable, are made
coherent with system operation.
Gated oscillator type switching regulators permit high
efficiency over extended ranges of output current. These
regulators achieve this desirable characteristic by using a
gated oscillator architecture instead of a clocked pulsewidth modulator. This eliminates the “housekeeping” currents associated with the continuous operation of
fixed-frequency designs. Gated oscillator regulators simply self-clock at whatever frequency is required to maintain the output voltage. Typically, loop oscillation frequency
ranges from a few Hertz into the kiloHertz region depending upon the load.
This asynchronous variable frequency operation seldom
creates problems; some systems, however, are sensitive
to this characteristic. The circuit in Figure 95 slightly
modifies a gated-oscillator-type switching regulator by
synchronizing its loop oscillation frequency to the system’s
clock. In this fashion the oscillation frequency and its
Circuit operation is best understood by temporarily ignoring the flip-flop and assuming that the LT1107 regulator’s
AOUT and FB pins are connected. When the output voltage
decays, the Set pin drops below VREF, causing AOUT to fall.
This causes the internal comparator to switch high, biasing the oscillator and output transistor into conduction. L1
receives pulsed drive and its flyback events are deposited
into the 100µF capacitor via the diode, restoring output
voltage. This overdrives the Set pin, causing the IC to
switch OFF until another cycle is required.
The frequency of this oscillatory cycle is load dependent
and variable. If a flip-flop is interposed in the AOUT/FB pin
path as shown, the frequency is synchronized to the
system clock. When the output decays far enough (trace
A, Figure 96) the AOUT pin (trace B) goes low. At the next
clock pulse (trace C) the flip-flop Q2 output (trace D) sets
low, biasing the comparator-oscillator. This turns on the
power switch (VSW pin is trace E), which pulses L1. L1
AN66-49
Application Note 66
VIN 2V TO 4V
47Ω
100k
5VOUT
ILIMIT
VIN
–
AOUT
L1
1N5817
VREF
+
SW1
AMPLIFIER
221k*
+
PRE1
Q1
Q1
CLR2 PRE2
100µF
VCC
D2
VREF
1.25V
74HC74
D1
OSCILLATOR
LT1107
82.5k*
SET
COMPARATOR
CLR1
CLK1
+
GND
Q2
CLK2
FB
–
100k*
SW2
GND
47k
AN66 F95
100kHz CLOCK
POWERED FROM 5V OUTPUT
L1 = 22µH COILTRONICS CTX-20-2
* = 1% METAL FILM RESISTOR
COILTRONICS (407) 241-7876
Figure 95. A Synchronizing Flip-Flop Forces Switching Regulator Noise To Be Coherent with the Clock
responds in flyback fashion, depositing its energy into the
output capacitor to maintain output voltage. This operation is similar to the previously described case except that
the sequence is forced to synchronize with the system
clock by the flip-flop’s action. Although the resulting
loop’s oscillation frequency is variable, it and all its attendant switching noise are synchronous and coherent with
the system clock.
Because of its sampled nature, this clocked loop may not
start. To ensure start-up, the flip-flop’s remaining section
is connected as a buffer. The CLR1/CLK1 line monitors
output voltage via the resistor string. If the circuit does not
start, Q1 goes high, CLR2 sets and loop operation commences. Although the circuit shown is a step-up type, any
switching regulator configuration can use this technique.
AN66-50
A = 50mV/DIV
(AC COUPLED)
B = 5V/DIV
C = 5V/DIV
D = 5V/DIV
E = 5V/DIV
AN66 F96
20µs/DIV
Figure 96. Waveforms for the Clock Synchronized Switching
Regulator. The Regulator Switches (Trace E) Only on Clock
Transitions (Trace C) Resulting in Clock Coherent Output
Noise (Trace A)
Application Note 66
BATTERY-POWERED CIRCUITS USING THE
LT1300 AND LT1301
by Steve Pietkiewicz
A = 20mV/DIV
AC COUPLED
B = 5V/DIV
5V from 2 Cells
Figure 97’s circuit provides 5V from a 2-cell input. Shutdown is effected by taking the Shutdown pin high. VIN
current drops to 10µA in this condition. This simple boost
topology does not provide output isolation, and in shutdown the load is still connected to the battery via L1 and
D1. Figure 98 shows the efficiency of the circuit with a
range of input voltages, including a fresh battery (3V) and
an “almost-dead” battery (2V). At load currents below a
few milliamperes, the 120µA quiescent current of the
device becomes significant, causing the fall off in efficiency detailed in the figure. At load currents in the 20mA
to 200mA range, efficiency flattens out in the 80% to 88%
range, depending on the input. Figure 99 details circuit
operation. VOUT is shown in trace A. The burst repetition
L1*
10µH
SHUTDOWN
2 AA
CELLS
SELECT
SHDN
VIN
SW
D1
1N5817
LT1300
NC
+
100µF
ILIM
GND
SENSE
PGND
+
C1
100µF
*SUMIDA CDS 4-100LC (708) 956-0666
COILCRAFT 3316-223 (800) 322-2645
AN66 F97
Figure 97. 2-Cell to 5V DC/DC Converter Delivers
> 200mA with a 2V Input
90
88
VIN = 4.0V
EFFICIENCY (%)
86
84
VIN = 3.0V
82
VIN = 2.5V
80
VIN = 2.0V
78
76
74
1
5V
200mA
OUTPUT
10
100
LOAD CURRENT (mA)
500
AN66 F98
Figure 98. Efficiency of Figure 112’s Circuit
C = 1A/DIV
AN66 F99
20µs/DIV
Figure 99. Burst Mode Operation In Action
pattern is clearly shown as VOUT decays, then steps back
up due to switching action. Trace B shows the voltage at
the switch node. The damped high frequency waveform at
the end of each burst is due to the inductor “ringing off,”
forming an LC tank with the switch and diode capacitance.
It is not harmful and contains far less energy than the high
speed edge that occurs when the switch turns off. Switch
current is shown in trace C. The current comparator inside
the LT1300 controls peak switch current, turning off the
switch when the current reaches approximately 1A.
Although efficiency curves present useful information, a
more important measure of battery-powered DC/DC converter performance is operating life. Figures 100 and 101
detail battery life tests with Figure 97’s circuit at load
currents of 100mA and 200mA, respectively. Operatinglife curves are shown using both Eveready E91 alkaline
cells and new L91 “Hi-Energy” lithium cells. These lithium
cells, new to the market, are specifically designed for high
drain applications. The performance advantage of lithium
is about 2:1 at 100mA load current (Figure 100), increasing to 2.5:1 at 200mA load (Figure 101). Alkaline cells
perform poorly at high drain rates because their internal
impedance ranges from 0.2Ω to 0.5Ω, causing a large
voltage drop within the cell. The alkaline cells feel quite
warm at 200mA load current, the result of I2R losses
inside the cells.
The reduced power circuit shown in Figure 102 can
generate 5V at currents up to 50mA. Here the ILIM pin is
grounded, reducing peak switch current to 400mA. Lower
profile components can be used in this circuit. The capacitors are C-case size solid tantalum and inductor L1 is the
tallest component at 3.2mm. The reduced peak current
also extends battery life, since the I2R loss due to internal
battery impedance is reduced. Figure 103 details efficiency versus load current for several input voltages and
AN66-51
Application Note 66
90
5.0
OUTPUT
88
2× E91
ALKALINE
4.0
2× L91
LITHIUM
3.0
2.5
VIN = 3V
86
3.5
EFFICIENCY (%)
OUTPUT/BATTERY VOLTAGE (V)
4.5
BATTERY
2.0
1.5
84
VIN = 2.5V
82
VIN = 2V
80
78
1.0
76
0.5
74
0
0
1
2
3
4 5 6 7
TIME (HOURS)
8
9 10 11
10
LOAD CURRENT (mA)
1
AN66 F100
AN66 F103
Figure 100. Two Eveready L91 Lithium AA Cells
Provide Approximately Twice the Life of E91 Alkaline
Cells at a 100mA Load Current
Figure 103. Efficiency of Figure 102’s Circuit
5.0
OUTPUT
OUTPUT/BATTERY VOLTAGE (V)
4.5
5.0
OUTPUT/BATTERY VOLTAGE (V)
4.5
OUTPUT
4.0
2× E91
ALKALINE
2× L91
LITHIUM
3.5
100
3.0
4.0
2× E91
ALKALINE
3.5
2× L91
LITHIUM
3.0
2.5
2.0
1.5
1.0
BATTERY
0.5
2.5
BATTERY
2.0
0
0
2
4
6
1.5
1.0
8 10 12 14 16 18 20 22 24
TIME (HOURS)
AN66 F104
0.5
Figure 104. 50mA Load and Reduced Switch Current
Are Kind to E91 AA Alkaline Battery; the Advantages
of L91 Lithium Are Not as Evident
0
0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5
TIME (HOURS)
AN66 F101
Figure 101. Doubling Load Current to 200mA Causes E91
Alkaline Battery Life to Drop by 2/3; L91 Lithium Battery
Shows 2.5:1 Difference in Operating Life
L1*
22µH
2 AA
CELLS
SHUTDOWN
SELECT
SHDN
VIN
SW
A 4-Cell Application
LT1300
+
47µF
ILIM
GND
SENSE
PGND
+
D1
MBRS140T3
5V
50mA
OUTPUT
33µF
*COILCRAFT 1608-223
AN66 F102
Figure 102. Lower Power Applications Can Use Smaller
Components. L1 Is Tallest Component at 3.1mm
AN66-52
Figure 104 shows battery life at a 50mA load. Note that the
L91 lithium battery lasts only about 40% longer than the
alkaline. The higher cost of the lithium cells makes the
alkaline cells more cost effective in this application. A pair
of Eveready AAA alkaline cells (type E92) lasts 96.6 hours
with 5mA load, very close to the rated capacity of the
battery.
A 4-cell pack is a convenient, popular battery size. Alkaline
cells are sold in 4-packs at retail stores and 4 cells usually
provide sufficient energy to keep battery replacement
frequency reasonable. Generating 5V from 4 cells, however, is a bit tricky. A fresh 4-cell pack has a terminal
voltage of 6.4V, but at the end of its life the pack’s terminal
voltage is around 3.2V; hence, the DC/DC converter must
Application Note 66
step the voltage either up or down depending on the state
of the batteries. A flyback topology with a costly custom
designed transformer could be employed but Figure 105’s
circuit gets around these problems by using a flying
capacitor scheme along with a second inductor. The
circuit also isolates the input from the output, allowing the
output to go to 0V during shutdown. The circuit can be
divided conceptually into boost and buck sections. L1 and
the LT1300 switch comprise the boost or step-up section
and L2, D1 and C3 comprise the buck or step-down
section. C2 is charged to VIN and acts as a level shift
between the two sections. The switch node toggles between ground and VIN + VOUT, and the L2/C2 diode node
toggles between –VIN and VOUT + VD. Figure 106 shows
efficiency versus load current for the circuit. All four
energy storage elements must handle power, which accounts for the lower efficiency of this circuit compared to
the simpler boost circuit in Figure 97. Efficiency is directly
L1*
27µH
C2**
+100µF
NC
5V/3.3V
4 AA
CELLS
+
ILIM
SELECT
VIN
SW
L2*
27µH
1N5817
LT1300
SHUTDOWN
C1**
100µF
SHDN
GND
SENSE
PGND
C3**
100µF
*L1, L2 = GOWANDA GA20-272K (716) 532-2234
**C1, C2, C3 = SANYO OS-CON 16SA100M (619) 661-6835
+
5V OR
3.3V
220mA
related to the ESR and DCR of the capacitors and inductors
used. Better capacitors cost more money. Better inductors
do not necessarily cost more but they do take up more
space. Worst-case RMS current through C2 occurs at
minimum input voltage and measures 0.4A at full load with
a 3V input. C2’s specified maximum RMS current must be
greater than this worst-case current. The Sanyo capacitors noted specify a maximum ESR of 0.045Ω with a
maximum ripple current rating of 2.1A. The Gowanda
inductors specify a maximum DCR of 0.058Ω.
LT1301 Outputs: 5V or 12V
The LT1301 is identical to the LT1300 in every way except
output voltage. The LT1301 can be set to a 5V or 12V
output via its Select pin. Figure 107 shows a simple 3.3V
or 5V to 12V step-up converter. It can generate 120mA at
12V from either 3.3V or 5V inputs, enabling the circuit to
provide VPP on a PCMCIA card socket. Figure 108 shows
the circuit’s efficiency. Switch voltage drop is a smaller
percentage of input voltage at 5V than at 3.3V, resulting in
the higher efficiency at 5V input.
L1*
22µH
3.3V
OR 5V
INPUT
SHUTDOWN
+
100µF
1N5817
SENSE
PGND
+
12V OUTPUT
47µF
*L1 = SUMIDA CD75-220K (708) 956-0666
AN66 F107
Figure 107. LT1301 Delivers 12V from 3.3V or 5V Input
84
90
82
88
VIN = 5V
80
86
78
EFFICIENCY (%)
EFFICIENCY (%)
VIN
SW
LT1301
ILIM
GND
AN66 F105
Figure 105. 4-Cell to 3.3V or 5V Converter Output Goes to Zero
When in Shutdown. Inductors May Have, but Do Not Require
Coupling; a Transformer or Two Separate Units Can Be Used
SELECT
SHDN
76
VIN = 3V
74
72
VIN = 4V
70
68
66
VIN = 3.3V
82
80
VIN = 5V
78
VIN = 6V
76
64
1
84
74
10
100
LOAD CURRENT (mA)
1
AN66 F106
Figure 106. Efficiency of Up/Down Converter in Figure 105
10
LOAD CURRENT (mA)
100
AN66 F108
Figure 108. Efficiency of Figure 122’s Circuit
AN66-53
Application Note 66
BATTERY-POWERED CIRCUITS USING THE
LT1304 MICROPOWER DC/DC CONVERTER
WITH LOW-BATTERY DETECTOR
by Steve Pietkiewicz
A 2-Cell to 5V Converter
A compact 2-cell to 5V converter can be constructed using
the circuit in Figure 109. Using the LT1304-5 fixed output
device eliminates the need for external voltage setting
resistors, lowering component count. As the battery voltage drops, the circuit continues to function until the
LT1304’s undervoltage lockout disables the part at approximately VIN = 1.5V. 200mA is available at a battery
voltage of 2.0V. As the battery voltage decreases below
2V, cell impedance starts to quickly increase. End-of-life is
usually assumed to be around 1.8V, or 0.9V per cell.
Efficiency is detailed in Figure 110. Micropower Burst
Mode operation keeps efficiency above 70%, even for load
current below 1mA. Efficiency reaches 85% for a 3.3V
input. Load transient response is illustrated in Figure 111.
Since the LT1304 uses a hysteretic comparator in place of
the traditional linear feedback loop, the circuit responds
immediately to changes in load current. Figure 112 details
start-up behavior without soft start circuitry (R1 and C1 in
Figure 109). Input current rises to 1A as the device is
turned on, which can cause the input supply voltage to
sag, possibly tripping the low-battery detector. Output
voltage reaches 5V in approximately 1ms. The addition of
R1 and C1 to Figure 109’s circuit limits inrush current at
start-up, providing for a smoother turn-on as indicated in
Figure 113.
VOUT 100mV/DIV
AC COUPLED
MBRS130L
22µH*
VIN
LBI
+
5V
200mA
ILOAD
200mA
0
LT1304-5
100µF
2 CELLS
SW
SENSE
100µs/DIV
LB0
GND
SHDN
IL
+
+
R1
1M
C1
1µF
100µF
SHUTDOWN
*SUMIDA CD54-220
(708) 956-0666
Figure 111. Boost Converter Load Transient Response
with VIN = 2.2V
VOUT 2V/DIV
AN66 F109
Figure 109. 2-Cell to 5V/200mA Boost Converter Takes Four
External Parts. Components with Dashed Lines Are for Soft Start
(Optional)
IIN
500mA/DIV
VSHDN
10V/DIV
1ms/DIV
90
DN66 F112
Figure 112. Start-Up Response. Input Current Rises Quickly
to 1A. VOUT Reaches 5V in Approximately 1ms. Output
Drives 20mA Load
VIN = 3.3V
80
EFFICIENCY (%)
DN66 F111
VIN = 2.5V
70
VIN = 1.8V
VOUT 2V/DIV
60
IIN
500mA/DIV
VSHDN
10V/DIV
50
40
0.1
1
10
100
LOAD CURRENT (mA)
500
AN66 F110
Figure 110. 2-Cell to 5V Converter Efficiency
AN66-54
1ms/DIV
DN66 F113
Figure 113. Start-Up Response with 1µF/1MΩ Components in
Figure 109 added. Input Current Is More Controlled. VOUT
Reaches 5V in 6ms. Output Drives 20mA Load
Application Note 66
A 4-Cell to 5V Converter
A 4-cell to 5V converter is more complex than a simple
boost converter because the input voltage can be either
above or below the output voltage. The single-ended
primary inductance converter (SEPIC) shown in Figure
114 accomplishes this task with the additional benefit of
output isolation. In shutdown conditions, the converter’s
output will go to zero, unlike the simple boost converter,
where a DC path from input to output through the inductor
and diode remains. In this circuit, peak current is limited
to approximately 500mA by the addition of 22k resistor
R1. This allows very small low profile components to be
used. The 100µF capacitors are D-case size with a height
of 2.9mm and the inductors are 3.2mm high. The circuit
can deliver 5V at up to 100mA. Efficiency is relatively flat
across the 1mA to 100mA load range.
100µF
22µH*
3.5V TO 6.5V
Super BurstTM Mode Operation: 5V/100mA DC/DC
with 15µA Quiescent Current
The LT1304’s low-battery detector can be used to control
the DC/DC converter. The result is a reduction in quiescent
current by almost an order of magnitude. Figure 116
details this Super Burst circuit. VOUT is monitored by the
LT1304’s LBI pin via resistor divider R1/R2. When LBI is
above 1.2V LBO is high, forcing the LT1304 into shutdown
mode and reducing current drain from the battery to 10µA.
When VOUT decreases enough to overcome the lowbattery detector’s hysteresis (about 35mV) LBO goes low.
Q1 turns on, pulling SHDN high and turning on the rest of
the IC. R3 limits peak current to 500mA; it can be removed
for higher output power. Efficiency is illustrated in Figure
Super Burst is a trademark of Linear Technology Corporation.
33µH*
IQ ≈ 15µA
MBR0530
SW
LBI
5V
100mA
+
47µF
22µH*
+
SHUTDOWN
100µF
LBI
1
+
330µF
FB
SHDN
GND
7
5
IL
6
R2
1.21M
47k
22k
AN66 F114
Figure 114. 4-Cell to 5V Step-Up/Step-Down Converter, Also
Known as SEPIC (Single-Ended Primary Inductance Converter).
Low Profile Components Are Used Throughout
*SUMIDA CD54-330
(708) 956-0666
AN66 F116
Figure 116. Super Burst Mode Operation 2-Cell to 5V DC/DC
Converter Draws Only 15µA Unloaded. 2 AA Alkaline Cells Will
Last for Years
85
90
80
VIN = 3V
80
VIN = 3V
EFFICIENCY (%)
VIN = 4V
75
EFFICIENCY (%)
LB0
R1
3.83M
+
100µF
*SUMIDA CD43-220
(708) 956-0666
4
SW
LT1304
2
SHDN
GND
R1
22k
2
3
VIN
2 CELLS
LT1304-5
LB0
IL
0.01µF
47k
SENSE
4 CELLS
5V
80mA
200k
Q1
2N3906
+
VIN
MBR0530
70
65
VIN = 6V
VIN = 5V
60
70
VIN = 2V
60
50
55
50
1
10
LOAD CURRENT (mA)
100
AN66 F115
Figure 115. Efficiency Plot of SEPIC Converter Shown
in Figure 114
40
0.01
0.1
1.0
10
LOAD CURRENT (mA)
100
AN66 F117
Figure 117. Super Burst Mode Operation DC/DC
Converter Efficiency
AN66-55
Application Note 66
117. The converter is approximately 70% efficient at a
100µA load, 20 points higher than the circuit of Figure 109.
Even at a 10µA load, efficiency is in the 40% to 50% range,
equivalent to 100µW to 120µW total power drain from the
battery. In contrast, Figure 109’s circuit consumes approximately 300µW to 400µW unloaded.
An output capacitor charging cycle or “burst” is shown in
Figure 118, with the circuit driving a 50mA load. The slow
response of the low-battery detector results in the high
number of individual switch cycles or “hits” within the
burst.
Figure 119 depicts output voltage at the modest load of
100µA. The burst repetition rate is around 4Hz. With the
load removed, the repetition rate drops to approximately
0.2Hz or one burst every 5 seconds. Systems that spend
a high percentage of operating time in sleep mode can
benefit from the greatly reduced quiescent power drain of
Figure 116’s circuit.
(5ns) will confirm the need for good PC board layout. The
200MHz ringing of the switch voltage is attributable to lead
inductance, switch and diode capacitance, and diode turnon time. Switch turn-on is shown in Figure 122. Transition
time is similar to that of Figure 121. Adherence to the
layout suggestions will result in working DC/DC converters with a minimum of trouble.
1Instrumentation for oscillographs of Figures 121 and 122 include Tektronix P6032 active probe,
Type 1S1 sampling unit and type 547 mainframe.
SHUTDOWN
1
2
VIN
8
LT1304
7
3
6
4
5
+ CIN
VOUT
+COUT
VOUT 100mV/DIV
AC COUPLED
VSW
5V/DIV
GND (BATTERY AND LOAD RETURN)
IL
1A/DIV
AN66 F120
VIN = 2.5V
IL = 50mA
50µs/DIV
DN66 F118
Figure 120. Suggested Layout for Best Performance. Input
Capacitor Placement as Shown Is Highly Recommended.
Switch Trace (Pin 4) Copper Area is Minimized
Figure 118. Super Burst Mode Operation in Action
VOUT 100mV/DIV
AC COUPLED
VOUT 100mV/DIV
AC COUPLED
50ms/DIV
50ms/DIV
DN66 F119
Figure 119. Super Burst Mode Operation Circuit with 100µA
Load, Burst Occurs Approximately Once Every 240ms
DN66 F121
Figure 121. LT1304 Switch Rise Time Is in the 5ns Range. These
Types of Edges Emphasize the Need for Proper PC Board Layout
Layout
The LT1304 switch turns on and off very quickly. For best
performance we suggest the component placement in
Figure 120. Improper layouts will result in poor load
regulation, especially at heavy loads. Parasitic lead inductance must be kept low for proper operation. Switch turnoff is detailed in Figure 1211. A close look at the rise time
AN66-56
VSW 1V/DIV
50ms/DIV
DN66 F122
Figure 122. Switch Fall Time. Lower Slope in Second and Third
Graticules Shows Effect of Lead and Bond Wire Inductance
Application Note 66
AUTOMATIC LOAD SENSING SAVES POWER
IN HIGH VOLTAGE CONVERTER
by Mitchell Lee
LT1107’s 320µA quiescent current the battery current is
3.5mA under no load. In standby applications this is
unacceptably high, even for two D cells.
There are a surprising number of high output voltage
applications for LTC’s micropower DC/DC converter family. These applications include electroluminescent panels,
specialized sensing tubes and xenon strobes. One of the
key features of the micropower converters is low quiescent current. Since the quiescent current is far less than
the self-discharge rate of common alkaline cells, the
traditional ON/OFF switch can be eliminated in cases
where the load is intermittent or where the load is shut
down under digital control.
A circuit consisting of transistors Q1 and Q2 was added to
reduce the standby current to an acceptable level. When a
load of more than 50µA is present, Q1 turns on, Q2 turns
off and the 9.1M resistor (R4) serves as a feedback path.
R2, R3 and R4 regulate the output at 128V.
The maximum switch voltage for many micropower devices is 50V. For higher outputs the circuit shown in Figure
123 is often recommended. It combines a boost regulator
and a charge pump tripler to produce an output voltage of
up to 150V. The output is sensed through a divider
network, which consumes a constant current of about
12µA. This doesn’t seem like much, but reflected back to
the 3V battery it amounts to over 3mA. Together with the
63V
47µF
+
6.3V
100µF
63V
100nF
R5
10k
MUR120
VIN
R1
47Ω
This automatic feedback switching scheme improves the
battery current by a factor of ten and eliminates the need
for a mechanical ON/OFF switch in applications where the
load is under digital control.
63V
100nF
+
L1
33µH
If the load current drops below 50µA, Q1 turns off and Q2
turns on, shorting out R4. With R4 out of the way, R2 and
R3 regulate the output to approximately 15V. The measured input current under this condition is only 350µA,
just slightly higher than the chip’s no-load quiescent
current. When the load returns, Q1 senses the excess
current and the output automatically rises to its nominal
value of 128V.
SW1
63V
100nF
250V*
1µF
Q2
MMBTA92
LT1107CS8
ILIM
3mA
128V
Q1
MMBTA92
63V
100nF
390pF
FB
GND SW2
R2
1M
3V
2 ALKALINE
D CELLS
R3
100k
R4
9.1M
= 1N4148 FOR ALL UNMARKED DIODES
* PANASONIC ECQ-E2105KF
AN66 F123
Figure 123. Automatic Shutdown Reduces Battery Current to 350µA
AN66-57
Application Note 66
Regulators—Switching
(Micropower)
Backlight
HIGH EFFICIENCY EL DRIVER CIRCUIT
by Dave Bell
Electroluminescent (EL) lamps are gaining popularity as
sources of LCD backlight illumination, especially in small,
handheld products. Compared with other backlighting
technologies, EL is attractive because the lamp is thin,
lightweight, rugged and can be illuminated with little
power.
EL lamps are capacitive in nature, typically exhibiting
around 3000pF/in2, and require a low frequency (50Hz to
1kHz) 120VRMS AC drive voltage. Heretofore, this has
usually been generated by a low frequency blocking oscillator using a large transformer.
Figure 124 depicts a high efficiency EL driver that can
drive a relatively large (12 in2) EL lamp using a small high
frequency transformer. The circuit is self-oscillating and
delivers a regulated triangle wave to the attached lamp.
Very high conversion efficiency may be obtained using
this circuit, even matching state-of-the-art CCFL backlights at modest brightness levels (10 to 20 foot-lamberts).
Since an EL lamp is basically a lossy capacitor, the
majority of the energy delivered to the lamp during the
charge half-cycle is stored as electrostatic energy
(1/2CV2). Overall conversion efficiency can be improved
by almost 2:1 if this stored energy is returned to the battery
during the discharge half-cycle. The circuit of Figure 124
operates as a flyback converter during the charge halfcycle, taking energy from the battery and charging the EL
capacitance. During the discharge half-cycle the flyback
converter operates in the reverse direction, taking energy
back out of the EL lamp and returning it to the battery.
Nearly 50% of the energy taken during the charge halfcycle is returned during the discharge half-cycle; hence
the 2:1 efficiency improvement.
During the charge half-cycle, the LT1303 operates as a
flyback converter at approximately 150kHz, ramping the
current in T1’s 10µH primary inductance to approximately
1A on each switching pulse. When the LT1303’s internal
AN66-58
power switch turns off, the flyback energy stored in T1 is
delivered to the EL lamp through D3 and C5. Successive
high frequency flyback cycles progressively charge the EL
capacitance until 300V is reached on the “+” side of C5. At
this point the feedback voltage present at the LT1303’s LBI
input reaches 1.25V, causing the internal comparator to
change state.
When the LT1303’s internal comparator changes state,
the open-collector driver at the LBO output is released.
This places the circuit into discharge mode and reverses
the operation of the flyback energy transfer. Q3 turns on
and removes the gate drive from Q2A, thereby disabling
switching action on the primary of T1. Flip-flop U2A is also
clocked, resulting in a high level on the Q output; this
positive feedback action keeps LBI above 1.25V. Even
though Q2A is turned off the LT1303’s SW pin still
switches into pull-up resistor R4. The resulting pulses at
the SW pin are used to clock U2B and to drive a “poor
man’s” current mode flyback converter on the secondary
of T1.
Every clock pulse to flip-flop U2B turns on Q2B and draws
current from the EL lamp through C5, T1, D2 and Q4. (Q4
must be a 600V rated MOSFET to withstand the high peak
voltages present on its drain during normal operation.)
Current ramps up through T1’s 2.25mH secondary inductance until the voltage across current sense resistor R12
reaches approximately 0.6V. At this point Q5 turns on,
providing a direct clear to U2B and thereby terminating the
pulse. Energy taken from the EL lamp and stored in T1’s
inductance is then transferred back to the battery through
D1 and T1’s primary winding. This cycle repeats at approximately 150kHz until the voltage on C5 ratchets down
to approximately 0V. Once C5 is fully discharged, the
preset input on U2A will be pulled low, forcing the voltage
on the LT1303’s LBI input to ground and initiating another
charge half-cycle.
This circuit produces a triangle voltage waveform with a
constant peak-to-peak voltage of 300V, but the frequency
of the triangle wave depends on the capacitance of the
attached EL lamp. A 12 in2 lamp has approximately 36nF
of capacitance, which results in a triangle wave frequency
of approximately 400Hz. This produces approximately
17FL of light output from a state-of-the-art EL lamp.
Because of the “constant power” nature of the charging
Application Note 66
flyback converter, light output remains relatively constant
with changes in the battery voltage. In addition, since EL
lamp capacitance decreases with age, the circuit tends to
minimize brightness reduction with lamp aging. C5, R9,
and R10 maintain a zero average voltage across the EL
lamp terminals—an essential factor for reliable lamp
operation.
thereby delivering four times as much energy (energy
stored in T1 is defined by 1/2LI2). The value of R12 must
also be reduced to 7.5Ω to increase the discharge flyback
current by the same ratio. For smaller lamps or for
brightness adjustment, the circuit may be “throttled” by
connecting the LT1303/LT1305’s FB pin to a small
current-sense resistor in the lower leg of the EL lamp.
Two options exist for EL lamps with different characteristics. Larger lamps can be supported by specifying an
LT1305 instead of the LT1303 shown in Figure 124. The
LT1305 will terminate switch cycles at 2A instead of 1A,
Not only does the depicted circuit operate very efficiently,
it takes output fault conditions in stride. The circuit, with
C5 rated at 300V, tolerates indefinite short-circuit and
open-circuit conditions across its EL lamp output pins.
C4
47µF
16V
VBATT
+
R2
2.2M
R3
2.2M
C5
4.7µF
160V
+
T1
4,5
6
1,2
10
R9
1M
10µH
Q1
2N3906
5V
D1
MBRS140T3
R6
10Ω
R4
470Ω
6
VIN
SHDN
D
Q
U2A
HC74
3
4
R1
18k
Q
5
SHDN
SW
LBI
LBO
GND
C1
220pF
C2
10pF
7
1
R5
47k
Q2A
1/2
Si9955
R7
4.7k
D3
MURS160T3
C6
0.022µF
D2
MURS160T3
R11
10Ω
U1
LT1303
FB
R10
1M
1:15
R14
10Ω
C3
0.1µF
EL
LAMP
(12IN2)
2
VBATT
Q3
2N7002
Q4
IRFRC20
PGND
8
R8
2.2k
VBATT = 5.4 TO 12V
T1 = DALE LPE5047-A132
(605) 665-9301
U2 = POWERED FROM 5V
D
Q
U2B
HC74
Q
Q2B
1/2
Si9955
R13
680Ω
R12
15Ω
Q5
2N3904
C7
1000pF
AN66 F124
Figure 124. High Efficiency EL Driver Circuit
AN66-59
Application Note 66
The Royer converter oscillates at a frequency set primarily
by T1’s characteristics (including its load) and the 0.068µF
capacitor. L1 sets the magnitude of the Q1-to-Q2 tail
current, and hence, T1’s drive level. The 1N5817 diode
maintains L1’s current flow when the LT1301’s switch is
off. The 0.068µF capacitor combines with L1’s characteristics to produce sine wave voltage drive at the Q1 and Q2
collectors. T1 furnishes voltage step-up and about 1400VP-P
appears at its secondary. Alternating current flows through
the 22pF capacitor into the lamp. On positive half cycles
the lamp’s current is steered to ground via D1. On negative
half cycles the lamp’s current flows through Q3’s collector
and is filtered by C1. The LT1301’s ILIM pin acts as a zero
summing point with about 25µA bias current flowing out
of the pin into C1. The LT1301 regulates L1’s current to
maintain equality of Q3’s average collector current, representing one-half the lamp current, and R1’s current,
represented by VA/R1. When VA is set to zero the ILIM pin’s
bias current forces about 100µA bulb current.
A LOW POWER, LOW VOLTAGE CCFL POWER SUPPLY
by Steve Pietkiewicz
Most recently published CCFL driver circuits require an
input supply of 7V to 20V and are optimized for bulb
currents of 5mA or more. This precludes lower power
operation from 2- or 3-cell batteries often used in PDAs,
palmtop computers and the like. A CCFL power supply that
operates from 2V to 6V is shown in Figure 125. This circuit
can drive a small (75mm) CCFL over a 100µA to 2mA range.
The circuit uses an LT1301 micropower DC/DC converter
IC in conjunction with a current driven Royer class converter comprising T1, Q1 and Q2. When power is applied
along with intensity adjust voltage VA, the LT1301’s ILIM
pin is driven slightly positive, causing maximum switching current to flow through the IC’s internal Switch pin
(SW). L1 conducts current, which flows from T1’s center
tap, through the transistors, into L1. L1’s current is
deposited in switched fashion to ground by the regulator’s
action.
T1
9
7
22pF
3kV
1
VIN
2V TO 6V
5
4
1Ω
3
0.068µF
120Ω
1N5817
Q1
ZTX849
SW
SENSE
+
LT1301
0.1µF
SHDN
GND
+ C1
1µF
SHUTDOWN
T1 = COILTRONICS CTX110654-1
L1 = COILCRAFT D03316-473
AN66 F125
R1
7.5k
1%
VA
0VDC TO 5VDC IN
INTENSITY ADJUST
100µA TO 2mA BULB CURRENT
Figure 125. CCFL Power Supply
AN66-60
10µF
Q3
2N3904
ILIM
PGND
CCFL
Q2
ZTX849
WIMA
MKP20
L1
47µH
SELECT
VIN
NC
2
D1
1N4148
Application Note 66
ALL SURFACE MOUNT EL PANEL DRIVER
OPERATES FROM 1.8V TO 8V INPUT
by Steve Pietkiewicz
disconnected or open. R3 provides intensity control by
varying output voltage. Intensity can also be modulated by
varying the drive signal’s frequency.
Electroluminescent (EL) panels offer a viable alternative to
LED, incandescent or CCFL backlighting systems in many
portable devices. EL panels are thin, rugged, lightweight
and consume little power. They require no diffuser and
emit an aesthetically pleasing blue-green light. EL panels,
being capacitive in nature, typically exhibit about 3000pF
per square inch of panel area and require low frequency
(50Hz to 1kHz) 120VRMS AC drive. This has traditionally
been generated using a low frequency blocking oscillator
with a transformer. Although this technique is efficient,
transformer size renders the circuit unusable in many
applications due to space constraints. Moreover, low
frequency transformers are not readily available in surface
mount form, complicating assembly.
Flyback transformer T1 (Dale LPE5047-A132) has a 10µH
primary inductance and a 1:15 turns ratio. It measures
12mm by 13.3mm and is 6.3mm high. The 1:15 turns ratio
generates high voltage at the output without exceeding the
allowable voltage on the LT1303’s Switch pin. Schottky
diode D1 is required to prevent ringing at the SW pin from
forward biasing the IC’s substrate diode. Because of T1’s
low leakage inductance the flyback spike does not exceed
22V. No snubber network is required since the LT1303 SW
pin can safely tolerate 25V. R1 and C3 provide decoupling
for the IC’s VIN pin. Feedback resistor R2 is made from
three 3.3M units in series instead of a single 10M resistor.
This lessens the possibility of output voltage reduction
due to PC board leakage shunting the resistor. Shutdown
is accomplished by bringing the IC’s SHDN pin high. For
minimum current drain in shutdown the 400Hz drive
signal should be low.
Figure 126’s circuit solves these problems by using an
LT1303 micropower switching regulator IC along with a
small surface mount transformer in a flyback topology.
The 400Hz drive signal is supplied externally. When the
drive signal is low, T1 charges the panel until the voltage
at point A reaches 240VDC. C1 removes the DC component from the panel drive, resulting in 120VDC at the panel.
When the input drive signal goes high the LT1303’s FB pin
is also pulled high, idling the IC and turning on Q1. Q1’s
collector pulls point A to ground and the panel to
– 120VDC. C2 can be added to limit voltage if the panel is
T1
1:15
VIN
1.6V
TO 8V
4, 5
+
C3
47µF
R1
10Ω
C3
0.1µF
1, 2
VIN
MUR160
6
10
D1
1N5818
SW
LT1303
SHDN
FB
GND
PGND
Figure 127 details relevant circuit waveforms with a 22nF
load (about 7 inches of panel) and a 5V input. Trace A is the
panel voltage. Trace B shows Switch pin action. The
circuit’s input current is pictured in trace C and trace D is
the 400Hz input signal. The circuit’s efficiency measures
about 77%. With a 5V input the circuit can deliver 100VRMS
at 400Hz into a 44nF load. More voltage can be obtained
at lower drive frequencies.
C2
50pF
R2
3.3M
R2
3.3M
R2
3.3M
51k
R3
25k
INTENSITY
ADJUST
A
C1
4.7µF
160V
TRACE A
200V/DIV
+
1k
TRACE B
20V/DIV
EL
PANEL
TRACE D
10V/DIV
1N4148
Q1
MPSA42
500µs/DIV
A) HIGH VOLTAGE OUTPUT
B) SWITCH PIN
C) INPUT CURRENT
D) 400Hz DRIVE
10k
OPERATE SHUTDOWN
T1 = DALE LPE5047-A132
(605) 665-9301
TRACE C
500mA/DIV
400Hz
SQUARE WAVE DRIVE
0 TO VIN
AN66 F127
Figure 127. Oscillograph of Relevant
Circuit Waveforms
AN66 F126
Figure 126. LT1303 Circuit Drives EL Panel
AN66-61
Application Note 66
A DUAL OUTPUT LCD BIAS VOLTAGE GENERATOR
by Jon A. Dutra
With the many different kinds of LCD displays available,
systems manufacturers often want the option of deciding
the polarity of the LCD bias voltage at the time of manufacturing.
The circuit in Figure 128 uses the LT1107 micropower
DC/DC converter with a single inductor. The LT1107
features an ILIM pin that enables direct control of maximum inductor current. This allows the use of a smaller
inductor without the risk of saturation. The LT1111 could
also be used with a resulting reduction in output power.
Circuit Operation
The circuit is basically an AC-coupled boost topology. The
feedback signal is derived separately from the outputs, so
loading of the outputs does not affect loop compensation.
Since there is no direct feedback from the outputs, load
regulation performance is reduced. With 28V out, from
10% to 100% load (4mA to 40mA), the output voltage
sags by about 0.65V. From 1mA to 40mA load, the output
voltage sags by about 1.4V. This is acceptable for most
displays.
Output noise is reduced by using the auxiliary gain block
(AGB) in the feedback path. This added gain effectively
reduces the hysteresis of the comparator and tends to
randomize output noise. With low ESR capacitors for C2
and C4, output noise is below 30mV over the output load
range. Output power increases with VBATTERY, from about
1.4W out with 5V in to about 2W out with 8V or more.
Efficiency is 80% over a broad output power range.
If only a positive or negative output voltage is required, the
two diodes and two capacitors associated with the unused
output can be eliminated. The 100kΩ load is required on
each output to load a parasitic voltage doubler created by
the capacitance of diodes D2 and D4. Without this minimum load, the output voltage can go up to almost 50%
above the nominal value.
Component Selection
The voltage at the Switch pin SW1 swings from 0V to VOUT
plus two diode drops. This voltage is AC coupled to the
positive output through C1 and D1 and to the negative
output through C3 and D3. The full output current flows
through C1 and C3. Most tantalum capacitors are not rated
for current flow and their use can result in field failures.
VBATTERY
4V to 16V
(OPTION
SEE TEXT)
10µF
16V
+
100k
30Ω
VIN
AO
+
10µF
16V
SW2
5V CONTRAST ADJ
1M POT
D2
+
+
C3
SW1
VO
24V TO 32V
(0mA TO 40mA)
100k
C2
D1
ILIM
D4
+
LT1107CS8
FB
C1
L1
D3
1N4148
+
VIN
3V to 12V
100k
C4
–VO
–24V TO –32V
(0mA TO 40mA)
SET
GND
1.25V
1.43M
0.01
2.32M
1N4148
10k
SHUTDOWN IN
“1” = OFF
100k
L1 = COILTRONICS CTX 50-4
C1, C2, C3 and C4 = 22µF, 35V LOW ESR
= 1N5819 or MBR140
Figure 128. LT1107 Dual Output LCD Bias Generator Schematic Diagram
AN66-62
AN67 F128
Application Note 66
Use a rated tantalum or a rated electrolytic for longer
system life. At lower output currents or higher frequencies, using monolithic ceramics is also feasible.
One could replace the 1N5819 Schottky diodes with 1N4148
types for lower cost, with a reduction in efficiency and load
regulation characteristics.
Shutdown
The circuit can be shut down in several ways. The easiest
is to pull the Set pin above 1.25V; however, this consumes
300µA in shutdown conditions. A lower power method is
to turn off VIN to the LT1107 by means of a high side switch
LCD BIAS SUPPLY
by Steve Pietkiewicz
An LCD requires a bias supply for contrast control. The
supply’s variable negative output permits adjustment of
display contrast. Relatively little power is involved, easing RF radiation and efficiency requirements. An LCD
bias generator is shown in Figure 129. In this circuit, U1
is an LT1173 micropower DC/DC converter. The 3V input
or by simply disabling a logic supply. This drops quiescent
current from the VBATTERY input below 10µA. In both cases
VOUT drops to 0V. In the event that +VOUT does not need
to drop to zero, C1 and D1 can be eliminated.
Output Voltage Adjustment
The output voltage can be adjusted from any voltage above
VBATTERY up to 46V with proper passive components.
Output voltage can be controlled by the user with DAC,
PWM or potentiometer control. By summing currents into
the feedback node, the output voltage can be adjusted
downward.
is converted to 24V by U1’s switch, L2, D1 and C1. The
Switch pin (SW1) also drives a charge pump composed
of C2, C3, D2 and D3 to generate – 24V. Line regulation
is less than 0.2% from 3.3V to 2V inputs. Although load
regulation suffers somewhat because the – 24V output is
not directly regulated, it measures 2% for loads from
1mA to 7mA. The circuit will deliver 7mA from a 2V input
at 75% efficiency.
D1
1N5818
L1*
100µH
R1
100Ω
SW1
3V
2 AA
CELLS
R4
2.21M
VIN
ILIM
U1
LT1173
+
C1
0.1µF
C2
4.7µF
FB
GND
SW2
R3
100k
D3
1N5818
D2
1N5818
D4
1N4148
OPERATE SHUTDOWN
* TOKO 262LYF-0092K
+
R2
120k
C3
AN67 F129
22µF
OUTPUT
–12V TO –24V
Figure 129. DC/DC Converter Generates LCD Bias
AN66-63
Application Note 66
Regulators—Switching
(Micropower)
and the output rises to 12V without any potentially harmful
overshoot (see Figure 131).
VPP Generator
The LTC1262 is available in both 8-pin PDIP and narrow
SO packages. With small surface mount capacitors, the
complete 12V supply takes up very little space on a printed
circuit board. In power sensitive applications, such as
PCMCIA flash cards for portable PCs, the LTC1262 shutdown current is low enough to preclude the need for
external switching devices when the system is inactive.
LTC1262 GENERATES 12V FOR PROGRAMMING
FLASH MEMORIES WITHOUT INDUCTORS
by Anthony Ng and Robert Reay
Flash memories require a 5V VCC supply and an additional
12V supply for write or erase cycles. The 12V supply can
be a system supply or be generated from the 5V supply
using a DC/DC converter circuit. Single supply flash memories (i.e., those with the 12V converter built-in) are available, but these memories have lower capacities and slower
write/erase performance and therefore are not as popular
as memories without a built-in 12V supply. Flash memories require that the 12V supply be regulated to within 5%
and not exceed the permitted maximum voltage (14V for
Intel ETOXTM memories). The LTC1262 offers a simple and
cost effective 12V programming supply to meet these
requirements.
Figure 130 shows a typical application circuit. The only
external components needed are four surface mount capacitors. The LTC1262 uses a triple charge pump technique to convert 5V to 12V. It operates from 4.75V to 5.5V
and delivers 30mA while regulating the 12V output to
within 5%. The TTL-compatible SHDN pin can be driven
directly by a microprocessor. When the SHDN pin is taken
high (or floated) the LTC1262 enters shutdown mode. In
this state the supply current of the LTC1262 is reduced to
0.5µA typical and the 12V output drops to VCC. When
SHDN is taken low, the LTC1262 leaves shutdown mode
FLASH MEMORY VPP GENERATOR
SHUTS DOWN WITH 0V OUTPUT
by Sean Gold
Nonvolatile “flash” memories require a well controlled
12V bias (VPP) for programming. The tolerance on VPP is
±5% for 12V memories. Excursions in VPP above 14V or
below – 0.3V are destructive. VPP is often generated with
a boost regulator whose output follows the input supply
when shut down. It is sometimes desirable to force VPP to
0V when the memory is not in use or is in read-only mode.
AN66-64
ETOX is a trademark of Intel Corporation.
1
0.22µF
0.22µF
2
C1–
+
SHDN
8
FROM MICROPROCESSOR
GND
LTC1262
6
3
VOUT
C2 –
4
5
VCC
C2+
C1
FLASH
MEMORY
7
VPP
+
4.7µF
+
4.7µF
VCC
VCC
(4.75V TO 5.5V)
AN66 F130
Figure 130. Typical LTC1262 Application Circuit
SHDN
5V/DIV
VOUT
2V/DIV
1ms/DIV
AN66 F131
Figure 131. LTC1262 Taken In and Out of Shutdown
The circuit in Figure 132 generates a smoothly rising 12V,
60mA supply that drops to 0V under logic control. Figure
133 illustrates the programming cycle. Shortly after driving the SHDN pin high, the LT1109-12 switching regulator
drives L1, producing high voltage pulses at the device’s
Switch pin. The 1N5818 Schottky diode rectifies these
pulses and charges a reservoir capacitor C2. Q1 functions
as a low on-resistance pass element. The 1N4148 diode
clamps Q1 for reverse voltage protection. The circuit does
not overshoot or display unruly dynamics because the
Application Note 66
regulator gets its DC feedback directly from the output at
Q1’s collector. Minor slew aberrations are due to Q1’s
switching characteristics.
Even with the additional losses introduced by Q1, efficiency is 83% with a 60mA load. Line and load regulation
are both less than 1%. Output ripple is about 100mV under
light loads. Quiescent current drops to 400µA when shut
down. All components shown in Figure 132 are available
in surface mount packages, making the circuit well suited
for flash memory cards and other applications where
minimizing PC board space is critical.
5k
L1
33µH
1N4148
1N5818
SHDN
5V/DIV
4.5V < VIN < 5.5V
0
C1
22µF
C2
22µF
VIN
SW
LT1109A-12
SHDN
VPP
5V/DIV
Q1
2N4403
SENSE
GND
SHUTDOWN PROGRAM
C3
1µF
VPP
12V
60mA
0
AN67 F133
AN66 F132
Figure 132. Boost Mode Switching Regulator with Low RON Pass
Transistor for Flash Memory Programming
Regulators—Linear
LOW NOISE WIRELESS COMMUNICATIONS
POWER SUPPLY
by Mitchell Lee and Kevin Vasconcelos
Shown in Figure 134 is a 5V power supply we designed for
a synthesizer oscillator. The original design used a
3-terminal regulator but the regulator’s voltage noise contributed to excessive phase noise in the oscillator, leading
us to this solution. Of prime importance is noise over the
10Hz to 10kHz band. Careful measurements show a 40dB
improvement over standard 3-terminal regulators.
The regulator is built around a 5V buried-Zener reference.
It is the buried Zener’s inherently low noise that makes the
finished supply so quiet. Measured over a 10Hz to 10kHz
band the 5V output contains just 7µVRMS noise at full load.
The 10Hz to 10kHz noise can be further reduced to
2.5µVRMS by adding a 100µH, 1000µF output filter. The
noise characteristics of the reference are tested and guaranteed to a maximum of 11µV over the band of interest.
An external boost transistor, the ZBD949, provides gain to
meet a 200mA output current requirement. Current limit-
Figure 133. Input and Output Waveforms for the Flash Memory
Programming Circuit
ing is achieved by ballasting the pass transistor and
clamping base drive. Although our application only requires 200mA, it is possible to extend the output current
to at least 1A by selecting an appropriate ballast resistor
and addressing attendant thermal considerations in the
pass transistor.
9V TO 12V
INPUT
+
220Ω
RED LED**
4.7Ω
47µF
ZBD949*
IN
10Ω
1/2W 5V
200mA
OUTPUT
2Ω
LT1021-5
OUT
GND
+
10µF
TANT
* ZETEX INC (516) 864-7630
** GLOWS IN CURRENT LIMIT. DO NOT OMIT
AN66 F134
Figure 134. Ultralow Noise 5V, 200mA Supply. Output Noise Is
7µVRMS Over a 10Hz to 10kHz Bandwidth. Reference Noise Is
Guaranteed Less Than 11µVRMS. Standard 3-Terminal
Regulators Have 100 Times the Noise and No Guarantees
AN66-65
Application Note 66
AN LT1123 ULTRALOW DROPOUT 5V REGULATOR
by Jim Williams and Dennis O’Neill
Switching regulator post regulation, battery-powered apparatus and other applications often require low VIN/VOUT
or low dropout linear regulators. For post regulators this
is needed for high efficiency. In battery circuits lifetime is
significantly affected by regulator dropout. The LT1123, a
new low cost reference/control IC, is designed specifically
for cost effective duty in such applications. Used in conjunction with a discrete PNP power transistor, the 3-lead
TO-92 unit allows very high performance positive regulator designs. The IC contains a 5V bandgap reference, error
amplifier, NPN Darlington driver and circuitry for current
and thermal limiting.
A low dropout example is the simple 5V circuit of Figure
135 using the LT1123 and an MJE1123 PNP transistor. In
operation, the LT1123 sinks Q1 base current through the
Drive pin to servo control the FB (feedback) pin to 5V. R1
provides pull-up current to turn Q1 off and R2 is a drive
limiter. The 10µF output capacitor (COUT) provides frequency compensation. The LT1123 is designed to tolerate
a wide range of capacitor ESR so that low cost aluminum
electrolytics can be be used for COUT. If the circuit is
located more than six inches from the input source, the
optional 10µF input capacitor (CIN) should be added.
INPUT
Q1**
+
CIN*
10µF
+
R1
600Ω
5VOUT
COUT
10µF
R2
20Ω
DRIVE
ing is practical. Excessive output current causes the
LT1123 to drive Q1 hard until the LT1123 current limits.
Maximum circuit output current is then a product of the
LT1123 current and the beta of Q1. The foldback characteristic of the LT1123’s drive current combined with the
MJE1123 beta and safe area characteristics provide reliable short-circuit limiting. Thermal limiting can also be
accomplished by mounting the active devices with good
thermal coupling.
Performance of the circuit is notable as it has lower
dropout than any monolithic regulator. Line and load
regulation are typically within 5mV and initial accuracy is
typically inside 1%. Additionally, the regulator is fully
short-circuit protected with a no load quiescent current of
1.3mA.
Figure 136 shows typical circuit dropout characteristics in
comparison with other IC regulators. Even at 5A the
LT1123/MJE1123 circuit dropout is less than 0.5V, decreasing to only 50mV at 1A. Totally monolithic regulators
cannot approach these figures, primarily because their
power transistors do not offer the MJE1123 combination
of high beta and excellent saturation. For example, dropout is ten times lower than in 138 types and significantly
better than all the other IC types. Because of Q1’s high
beta, base drive loss is only 1% to 2% of output current,
even at high output currents. This maintains high efficiency under the low VIN/VOUT conditions the circuit will
typically see. As an exercise, the MJE1123 was replaced
with a 2N4276 germanium device. This provided even
lower dropout performance but limiting couldn’t be production guaranteed without screening.
U1
FB
LT1123
3.0
GND
2.5
AN66 F135
Figure 135. The LT1123 5V Regulator Features Ultralow Dropout
Normally, such configurations require external protection
circuitry. Here, the MJE1123 has been cooperatively designed by Motorola and LTC for use with the LT1123. The
device is specified for saturation voltage for currents up to
4A with base drive equal to the minimum LT1123 drive
current specification. In addition, the MJE1123 is specified for min/max beta at high current. Because of this
factor and the defined LT1123 drive, simple current limit-
AN66-66
DROPOUT VOLTAGE (V)
*OPTIONAL (SEE TEXT)
**MOTOROLA MJE1123
LT138
2.0
1.5
LT1084
1.0
LT1123/2N4276
LT1185
0.5
LT1123/MJE1123
0
0
1
2
3
4
5
OUTPUT CURRENT (A)
AN66 F136
Figure 136. LT1123 Regulator Dropout Voltage vs Output Current
Application Note 66
Regulators—Linear
(Microprocessor Power)
Circuit Examples
LT1580 LOW DROPOUT REGULATOR
USES NEW APPROACH TO
ACHIEVE HIGH PERFORMANCE
by Craig Varga
Enter the LT1580
The LT1580 NPN regulator is designed to make use of the
higher supply voltages already present in most systems.
The higher voltage source is used to provide power for
the control circuitry and supply the drive current to the
NPN output transistor. This allows the NPN to be driven
into saturation, thereby reducing the dropout voltage by
a VBE compared to a conventional design.
The LT1580 is capable of 7A maximum with approximately 0.8V input-to-output differential. The current requirement for the control voltage source is approximately
1/100 of the output load current or about 70mA for a 7A
load.
Figure 137 shows a circuit designed to deliver 2.5V from
a 3.3V source with 5V available for the control voltage.
Figure 138 shows the response to a load step of 200mA to
4.0A. The circuit is configured with a 0.33µF Adjust pin
bypass capacitor. The performance without this capacitor
is shown in Figure 139. This difference in performance is
the reason for providing the Adjust pin on the fixed voltage
devices. A substantial savings in expensive output decoupling capacitance may be realized by adding a small
“1206-case” ceramic capacitor at this pin.
Figure 140 shows an example of a circuit with shutdown
capability. By switching the control voltage rather than the
main supply, the transistor providing the switch function
needs only a small fraction of the current handling ability
that it would need if it was switching the main supply. Also,
in most applications it is not necessary to hold the voltage
drop across the controlling switch to a very low level to
maintain low dropout performance.
5V
5
3.3V
VIN
VCONT
4
U1
LT1580
SENSE
ADJ
C3
22µF
25V
+
+
C2
220µF
10V
VOUT
1
VOUT = 2.5V
3
VCC
2
R1
110Ω
1%
+
C4
0.33µF
R2
110Ω
1%
100µF
10V
×2
+
1µF
25V
× 10
MICROPROCESSOR
SOCKET
C1
100µF
10V
VSS
RTN
AN66 F137
Figure 137. LT1580 Delivers 2.5V from 3.3V at up to 6A
AN66-67
Application Note 66
50mV/DIV
50mV/DIV
2A/DIV
2A/DIV
AN66 F138
50µs/DIV
50µs/DIV
Figure 138. Transient Response of Figure 137’s Circuit with
Adjust Pin Bypass Capacitor. Load Step Is from 200mA to 4A
AN66 F139
Figure 139. Transient Response Without Adjust Pin Bypass
Capacitor. Otherwise, Conditions Are the Same as in Figure 138
Q1
Si9407DY
5V
R3
10k
5
3.3V
VIN
VCONT
U1 SENSE
LT1580
SHUTDOWN
ADJ
C3
22µF
25V
+
+
2
C2
220µF
10V
C4
0.33µF
VOUT
4
1
VOUT = 2.5V
3
R1
110Ω
1%
C1
R2
110Ω 100µF
10V
1%
LOAD
+
RTN
AN66 F140
Figure 140. Small FET Adds Shutdown Capability to LT1580 Circuit
LT1585: NEW LINEAR REGULATOR
SOLVES LOAD TRANSIENTS
by Craig Varga
The latest hot new microprocessors have added a significant complication to the design of the power supplies that
feed them. These devices have the ability to switch from
consuming very little power to requiring several amps in
tens of nanoseconds. To add a further complication, they
are extremely intolerant of supply voltage variations. Gone
are the days of the popcorn 3-terminal regulator and the
0.1µF decoupling capacitor. The LT1585 is the first low
dropout regulator specifically designed for tight output
voltage tolerance (optimized for the latest generation
processors) and fast transient response.
AN66-68
Figure 141 shows the kind of response that can and must
be achieved if these microprocessors are to operate reliably. Figure 142 details the first several microseconds of
VOUT
50mV/DIV
IOUT
2A/DIV
AN66 F141
Figure 141. Transient Response of 200mA to 4A Load Step
Application Note 66
the transient in Figure 141. The load change in this case is
3.8A in about 20ns. Two parasitic elements dominate the
transient performance of the system. Both are controlled
by the type, quantity and location of the decoupling
capacitors in the system.
Anatomy of a Load Transient
The instantaneous droop at the leading edge of the transient is the result of the sum of the effects of the equivalent
series resistance (ESR) and the equivalent series inductance (ESL) from the output capacitor(s) terminal(s) to the
load connection. Note that these contributions also include
the lead trace parasitics from the capacitor(s) to the load.
The resistive component is simply ∆I • ESR. The droop to
point A, 23.6mV, is the ESR contribution. Calculating ESR:
23.6mV/3.8A = 0.0062Ω
The effects of inductance are predicted by the formula V =
LdI/dt. The voltage from point A to the bottom of the
trough is the inductive contribution (13.4mV). ESL is
calculated to be 0.07nH. After the load current stops rising
the inductive effects end, bringing the voltage to point B.
At this point the curve settles into a gentle droop. The
droop rate is dV/dt = I/C. There is about 1300µF of useful
capacitance on the board in this case (see Figure 143). As
the regulator output current starts to approach the new
load current, the droop rate lessens until the regulator
supplies the full load current. This is the inflection point in
the curve. Since the regulator now measures the output
voltage as being too low, it overshoots the load current and
recharges the output capacitors to the correct voltage.
Faster Regulator Means Fewer Capacitors,
Less Board Space
The regulator has one major effect on the system’s transient behavior. The faster the regulator, the less bulk
capacitance is needed to keep the droop from becoming
excessive. It is here that the advantage of the LT1585
shows up. The response time of the LT1585 is about onehalf that of the last generation 3-terminal regulators.
The response in the first several hundred nanoseconds is
controlled by the careful placement of bypass capacitors.
Figure 143 is a schematic diagram of the circuit but the
layout is critical so consult the LTC factory for circuit and
layout information.
1
5V
NOMINAL VO
C1 TO C2
220µF
10V
SANYO
OS-CON
×2
23.6mV
“A”
U1 VOUT
LT1585CT
GND
2
3
C9 TO C18
1µF
SMD
× 10
+
C3 TO C8
220µF
10V
AVX TYPE TPS
×6
LOAD
AN66 F143
Figure 143. Schematic Diagram:
LT1585 Responding to Fast Transients
13.4mV
“B”
+
VIN
AN66 F142
Figure 142. Detailed Sketch of First Few Microseconds of Transients
AN66-69
Application Note 66
Battery Chargers
For NiMH batteries, a pulsed trickle charge can be easily
implemented with a switch in series with R1; switch Q1 at
the desired rate and duty cycle. If a microcontroller is used
to control the charging, connect the DAC current-sink
output to the PROG pin.
CHARGING NiMH/NiCd OR Li-Ion WITH THE LT1510
by Chia Wei Liao
Charging NiMH or NiCd Batteries
The circuit in Figure 144 will charge battery cells with
voltages up to 20V at a full charge current of 1A (when Q1
is ON) and a trickle charge current of 100mA (when Q1 is
OFF). If the third charging level is needed, simply add a
resistor and a switch. The basic formula for charging
current is:
Charging Li-Ion Batteries
The circuit in Figure 145 will charge lithium-ion batteries
at a constant current of 1.5A until battery voltage reaches
8.4V, set by R3 and R4. It then goes into constant voltage
charging and the current slowly tapers off to zero. Q3 can
be added to disconnect R3 and R4 so they will not drain the
battery when the wall adapter is unplugged.
2.465
(2000) (when Q1 ON)
R1||R2
2.465
(2000) (when Q1 OFF)
R1
30µH
SW
C1
0.22µF
VCC
1N5819 OR
MBRD340
+
BOOST
WALL
ADAPTOR
1N5819 OR
MBRD340
10µF
PROG
1µF
LT1510S8
D1
1N914
GND
R1
50K
300Ω
0.1µF
1K
VC
SENSE
R2
5.6K
Q1
VN2222
BAT
+
+
22µF
–
ON: IBAT = 1A
OFF: IBAT = 0.1A
IBAT
2V TO 20V
AN66 F144
Figure 144. Charging NiCd or NiMH Batteries
30µH
SW
C1
0.22µF
VCC
+
1N5819 OR
MBRD340
BOOST
PROG
LT1510S16
D1
1N914
SENSE
*OPTIONAL
NOTE: PRIMARY Li-Ion
BATTERY PROTECTION
MUST BE PROVIDED
BY AN INDEPENDENT
CIRCUIT
1µF
GND
OVP
11V TO 25V
DC WALL
ADAPTOR
10µF 1N5819 OR
MBRD340
0.1µF
300Ω
1K
3.83k
VC
BAT
+
+
22µF
4.2V
–
+
Q3*
VN2222
4.2V
–
R3
59K
R4
25K
AN66 F145
Figure 145. Charging Li-Ion Batteries
AN66-70
+
–
Application Note 66
Typical Charging Algorithms
The following algorithms are representative of current
techniques:
Lithium-Ion — charge at constant voltage with current
limiting set to protect components and to avoid overloading the charging source. When the battery voltage reaches
the programmed voltage limit, current will automatically
decay to float levels. The accuracy of the float voltage is
critical for long battery life. Be aware that lithium-ion
batteries in series suffer from “walk away” because of the
required constant float voltage charging technique. “Walk
away” is a condition where the batteries in the series string
wind up in different states of the charge/discharge cycle.
They may need to be balanced by redistributing charge
from one battery to another. This phenomenon is minimized by carefully matching the batteries within a pack.
by the manufacturer. Monitor battery charge state using
voltage change with time (dV/dt), second derivative of
voltage (d2V/dt), battery pressure or some combination of
these parameters. When the battery approaches full charge,
reduce the charging current to a value (top-off) that can be
maintained for a long time without harming the battery.
After the top-off period, usually set by a simple time out,
reduce the current further to a trickle level that can be
maintained indefinitely, typically 1/10 to 1/20 of the battery capacity.
Nickel-Metal-Hydride — same as NiCd except that some
NiMH batteries cannot tolerate a continuous low level
trickle charge. Instead they require a pulsed current of
moderate value with a low duty cycle so that the average
current represents a trickle level. A typical scenario would
be one second ON and thirty seconds OFF with the current
set to thirty times desired trickle level.
Nickel-Cadmium — charge at a constant current determined by the power available or by a maximum specified
LITHIUM-ION BATTERY CHARGER
by Dimitry Goder
Lithium-ion (Li-Ion) rechargeable batteries are quickly
gaining popularity in a variety of applications. The main
reasons for the success of Li-Ion cells are higher power
density and higher terminal voltage compared to other
currently available battery technologies. The basic charging principle for a Li-Ion battery is quite simple: apply a
constant voltage source with a built-in current limit. A
depleted battery is charged with a constant current until it
reaches a specific voltage (usually 4.2V per cell), then it
floats at this voltage for an indefinite period. The main
difficulty with charging Li-Ion cells is that the floating
voltage accuracy needs to be around 1%, with 5% currentlimit accuracy. These two targets are fairly difficult to
achieve. Figure 146 shows the schematic of a full solution
for a Li-Ion charger.
The battery charger is built around the LTC1147, a high
efficiency step-down regulator controller. The IC’s constant off-time architecture and current mode control ensure circuit simplicity and fast transient response. At the
beginning of the ON cycle, P-channel MOSFET Q1 turns on
and the current ramps up in the inductor. An internal
current comparator senses the voltage proportional to the
inductor current across sense resistor R13. When this
voltage reaches a preset value, the LTC1147 turns Q1 off
for a fixed period of time set by C1. After the off-time the
cycle repeats.
To provide an accurate current limit, U3A and Q2 are used
to sense the charging current separately from the LTC1147.
U3A forces the voltage across R11 to match the average
drop across the current sense resistor R13. This voltage
sets Q2’s drain current, which flows unchanged to the
source. As a result the same voltage appears across R9,
which is now referenced to ground. Since C5 provides
high frequency filtering, U3A shifts the average value of
the output current. N-channel MOSFET Q2 ensures correct circuit operation even under short-circuit conditions
by allowing current sensing at potentials close to ground.
U3B monitors voltage across R9 and acts to keep it
constant by comparing it to the reference voltage. Diode
D3 is connected in series with U3B’s output, allowing the
circuit to operate as a current limiter. The current feedback
circuit is not active if the output current limit has not been
reached.
AN66-71
Application Note 66
VIN
(6V TO 14V)
R14
5.1k
+
Q1
Si9430
1
2
3
4
C1
270pF
VIN
PDR
CT
ITH
U1
LTC1147
VFB
VREF
C3
33µF
25V
AVX TPS
U2
LT1009-2.5
L1*
50µH
CTX50-4
8
6
D2
MURS320
R13
0.1Ω
D1
MBRS130
+
5
SENSE +
SENSE –
VOUT**
4.2V
1A MAX
GND
C4
220µF
10V
AVX TPS
R15
170k
0.25%
7
1000pF
R1
1k
R12
20k
1%
R11
20k
1%
100Ω
VIN
Q3
2N7002
R3
51k
1%
2
VREF
1
U3A
3
R4
22k
Q2
2N7002
+
5
R9
20k, 1%
6
R8
475k, 1%
LT1014
LT1014
D3
1N4148
7
U3B
–
R2
24.9k
1%
R10
100Ω
–
+
C2
3300pF
C5
0.1µF
VIN
C7
0.1µF
R5
100Ω
VREF
R7
20k, 1%
0.1µF
*L1 = CTX50-4, COILTRONICS (407) 241-7876
**PRIMARY Li-Ion BATTERY PROTECTION MUST
BE PROVIDED BY AN INDEPENDENT CIRCUIT
4
8
C6
0.1µF
+
D4
1N4148
10
U3C
11
–
9
LT1014
R6
22k
VREF
R16
249k
0.25%
AN66 F146
Figure 146. Li-Ion Battery Charger Schematic
U3C provides the voltage feedback by comparing the
output voltage to the reference. The feedback resistor ratio
[R16/(R15 + R16)] sets the output at exactly 4.2V. U3C
has a diode (D4) connected in series with its output. This
diode ensures that the voltage and current feedback circuits do not operate at the same time. The reference
voltage is supplied by the LT1009, with a guaranteed initial
tolerance of 0.2%. Together with the 0.25% feedback
AN66-72
resistors, the circuit provides less than 1% output voltage
error over temperature.
When the input voltage is not present Q3 is automatically
turned off and the feedback resistors do not discharge the
battery. Diode D2 is connected in series with the output,
preventing the battery from supplying reverse current to
the charger.
Application Note 66
voltage is less than about 4V. Under either of these
circumstances, unlimited current flows from the 5V input
supply, through D1 and Q1’s base-emitter junction, frying
at least Q1.
SIMPLE BATTERY CHARGER RUNS AT 1MHz
by Mitchell Lee
Fast switching regulators have reduced coil sizes to the
point that they are no longer the largest components on
the board. A case in point is the LT1377, which can operate
at 1MHz with inductances under 10µH.
Q2 has been added to allow full current control even when
the output voltage is less than the input voltage. In normal
operation, where the output is boosted higher than 5V, Q2
is fully on. Its gate is held at 1.25V (Pin 2 feedback voltage)
and its source is greater than 5V; hence it has no choice but
to be fully enhanced. Q2 becomes more functional when
the output voltage drops to around 4V. First of all, at 4V
input the switching regulator stops switching because
more than 50mA current flows and the feedback pin is
pulled up above 1.25V—Q1 makes sure of that. But as
Q1’s collector continues to rise, Q2 is gradually cut off, at
least to the extent necessary to starve the drain current
back to about 50mA. This action works right down to
VOUT = 0. In a short circuit Q2 dissipates about 200mW,
not too much for a surface mount MOSFET.
The circuit shown in Figure 147 was designed for a
customer who wanted to charge a 4-cell NiCd pack from
a 5V logic supply. (This circuit will work equally well with
a 3.3V input.) Clearly the circuit needs an output voltage
greater than 5V, which is handled easily by the LT1377
boost regulator. The output current is limited to approximately 50mA by a VBE current sensor (Q1/R1) controlling
the Feedback pin (2) of the LT1377. This current is perfect
for slow charging or trickle charging AA NiCd batteries.
Battery chargers are commonly subject to a number of
fault conditions, which must be addressed in the design
phase. First, what happens when the battery is disconnected? In a boost regulator the output voltage will increase without bound and blow up either the output
capacitor or switch. Some voltage limiting is necessary,
and D2 serves this purpose. If the voltage on C3 rises to
11.25V, D2 takes over the control loop at the Feedback pin.
This circuit is useful for four to six cells and the output
current can be modified somewhat by changing sense
resistor R1. A reasonable range is from very low currents
(1mA or less) up to 100mA. The current will diminish as
Q1’s VBE drops about 0.3%/°C with temperature.
Another potential calamity is an output short circuit; a
related fault results from connecting a battery pack containing one or more shorted cells, such that the terminal
L1
4.7µH
COILCRAFT
DO-1608-472
D1
MBR0520L
+
VIN = 5V
4
+
C1
22µF
10V
C2
47nF
R2
2k
5
8
VIN
VSW
SHDN/SYNC
FB
C3
100µF
16V
2
LT1377
1
VC
GND
6
R1
12Ω
Q1
2N3906
D2
10V
400mW
R3
1k
Q2
Si9400DY
50mA
(11V MAX)
GND
C4
1nF
7
AN66 F147
Figure 147. Battery Charger Schematic Diagram
AN66-73
Application Note 66
A PERFECTLY TEMPERATURE-COMPENSATED
BATTERY CHARGER
by Mitchell Lee and Kevin Vasconcelos
Battery charging circuits are usually greeted with a yawn,
but this lead-acid charger offers a combination of features
that sets it apart from all others. It incorporates a low
dropout regulator, temperature compensation, dual-rate
charging, true negative ground and consumes zero standby
current.
The LT1083 family of linear regulators exhibits dropout
characteristics of less than 1.5V as compared to 2.5V in
standard regulators. A smaller regulator drop allows for
lower input voltages and less power dissipation in the
regulator. In this application the regulator is used to control
charging voltage and limit maximum charging current.
The temperature compensation employed in this circuit,
unlike diode-based straight-line approximations, follows
the true curvature of a lead-acid cell. This prevents over or
undercharging of the battery during periods of extended
low or high ambient temperatures. Temperature compensation is conveniently provided by a Tempsistor ® as
shown in Figure 148. The Tempsistor is used to generate
a temperature-dependent current, which, in turn, adjusts
the charger’s output voltage to match that of the battery.
The match is within 100mV for a 12V battery over a range
of –10°C to 60°C. The best place for the Tempsistor is
directly under the battery with the battery resting on a pad
of styrofoam.
Q1 provides a low voltage disconnect function that reduces the charger standby current to zero. When the input
voltage (from a rectified transformer) is available, Q1 is
biased ON and Q2 is turned ON. Q2 connects the various
current paths on the output of the regulator to ground,
activating the charging circuitry. If the input voltage is
removed, Q1 and Q2 turn off, and all current paths from the
battery to ground (except for the load, of course) are
interrupted. This prevents unnecessary battery drain when
the charging source is not available.
A dual-rate charging characteristic is achieved by means
of a current-sense resistor (RS) and a sense comparator
(LT1012). If the battery charge current exceeds the floatcurrent threshold of 10mV/RS, the comparator pulls the
gate of Q3 low, increasing the output voltage by 600mV.
This sets the charging voltage to 14.4V at 25°C. After the
battery reaches full charge the current will fall below the
10mV/RS threshold and the LT1012 will short out R7,
reducing the output by 600mV to a float level of 13.8V.
Tempsistor is a registered trademark of Thermodisc Inc.
+
D1
1N4001
VIN ≥ 16.0V
+
R1
1k
C1
10µF
TANT
ADJ
C3
47µF
ALUM
TO LOAD
RS
0.2Ω
LT1086
IN
OUT
R3
300Ω
R4
12Ω
Q1
2N3906
+
C2
10µF
TANT
RTH
1K821J
R8
1k
1%
R9
124k
1%
R10
1k
12V
GELCELL
TO VIN
+
LT1012
–
R5
2210Ω
1%
R6
250Ω
R7
110Ω
R2
10k
Q2
VN2222
Q3
VN2222
RS = 10mV/ITH
= THERMODISC: 1K821J.
TEL: (616) 777-4100
R11
1M
AN66 F148
Figure 148. Battery Charger Follows Temperature Coefficient of a Lead-Acid Cell Very Accurately
AN66-74
Application Note 66
Both the float and charging voltages can be trimmed by
R6; R7 sets the 600mV difference between them.
If you want to set the trip current to an exact figure, the
current shunt RS can be calculated as RS = 10mV/ITH. For
a threshold of C/100 this reduces to RS = 1/C.
With the charging source connected, the sense resistor RS
measures only battery current. This eliminates the tendency found in some schemes for the charger to trip on
load current.
Table 1. The Regulator Should Be Chosen to Provide at Least C/4
Charging Current
Table 1 simplifies the selection of an appropriate regulator
for batteries of up to 48 Ampere-hours (Ah). The selection
is based on providing a minimum available charge current
of at least C/4, where C represents the battery’s Amperehour capacity. The next larger regulator may be required
in applications where sustained load currents of greater
than C/10 are expected.
A SIMPLE 300mA NiCd BATTERY CHARGER
by Randy G. Flatness
+
C1
22µF
25V
C2
0.1µF
8
2
VSW
0.8A
20mA
0.5Ω
LT1086
1.5A
50mA
0.2Ω
LT1085
3.2A
100mA
0.1Ω
5.5A
200mA
0.05Ω
24Ah to 48Ah
LT1083
8.0A
400mA
0.025Ω
5
L1
50µH
4
VFB
2
VOUT
3
D1
MBRS130LT3
SHDN
LBIN
LT1117
LT1084
LTC1174
3
SENSE
RESISTOR
(SHUNT)
6Ah to 12Ah
1
VIN
IPGM
3Ah to 6Ah
DEVICE
FLOAT
CURRENT
THRESHOLD
12Ah to 24Ah
6
7
≤ 3Ah
MAXIMUM
CHARGING
CURRENT
The circuit shown in Figure 149 uses an LTC1174 to
control the charging circuit. A fully self-contained switching regulator IC, the LTC1174 contains both a power
switch and the control circuitry (constant off-time controller, reference voltage, error amplifier and protection circuitry). The internal power switch is a P-channel MOSFET
transistor in a common-source configuration; consequently
when the switch turns on, the LTC1174’s VSW pin is
Low current battery charger circuits are required in
handheld products such as palmtop, pen-based and fingertip computers. The charging circuitry for these applications must use surface mount components and consume
minimal board space. The circuit shown in Figure 149
meets both of these requirements.
VIN
8V TO 12.5V
BATTERY
CAPACITY
1
LBOUT
GND
R1
182k
1%
+
D2
MBRS130LT3
C3
100µF
10V
VBATT
4 CELLS
R2
39.2k
1%
4
AN66 F149
C1 = AVX (TA) TPSD226M025R0200 ESR = 0.200 IRMS = 0.775A
C3 = AVX (TA) TPSD107M010R0100 ESR = 0.100 IRMS = 1.095A
D1, D2 = MOTOROLA SCHOTTKY VBR = 30V
L1 = COILTRONICS CTX50-2P DCR = 0.212 IDC = 0.729A TYPE 52 CORE
COILTRONICS (407) 241-7876
VOUT = 1.25V • (1 + R1/R2) = 7.0V
(VBATT + 0.6V) • 4µs
(EQ.1)
FAST CHARGE ≈ 0.6A –
2•L
Figure 149. 4-Cell, 300mA LTC1174 Battery Charger Implemented with All Surface Mount Components
AN66-75
Application Note 66
connected to the input voltage. This power switch handles
peak currents of 600mA. The LTC1174’s architecture
allows it to achieve 100% duty cycle, forcing the internal
P-channel MOSFET on 100% of the time.
When the batteries are being charged, the resistor divider
network (R1 and R2) forces the LTC1174’s Feedback pin
(VFB) below 1.25V, causing the LTC1174 to operate at the
maximum output current. An internal 0.1Ω resistor senses
HIGH EFFICIENCY (>90%) NiCd BATTERY CHARGER
CIRCUIT PROGRAMMABLE FOR 1.3A FAST CHARGE
OR 100mA TRICKLE CHARGE
by Brian Huffman
this current and sets it at approximately 300mA, according
to equation 1 (shown on the schematic). When the batteries are disconnected, the error amplifier drives the Feedback pin to 1.25V, limiting the output voltage to 7.0V.
Diode D2 prevents the batteries from discharging through
the divider network when the charger is shut down. In
shutdown mode less than 10µA of supply current is drawn
from the input supply.
with efficiency exceeding 90%. This circuit can be modified easily to handle up to eight NiCd cells.
Battery charger circuits are of universal interest to laptop,
notebook and palmtop computer manufacturers. High
efficiency is desirable in these applications to minimize the
power dissipated in the surface mount components. The
circuit shown in Figure 150 is designed to charge four
NiCd cells at a 1.3A fast charge or a 100mA trickle charge
The circuit uses an LTC1148 in a step-down configuration
to control the charge rate. The LTC1148 is a synchronous
switching regulator controller that drives external, complementary power MOSFETs using a constant off-time current
mode architecture. When the LTC1148’s P-drive output
pulls the gate of Q1 low, the P-channel MOSFET turns on
and connects one side of the inductor to the input voltage.
During this period, current flows from the input through Q1,
+
VIN
8V TO
15V
C1
1µF
C2
0.1µF
3
VIN
P-DRIVE
0V = NORMAL
>1.5V = SHUTDOWN
10
6
R1
51
“1”
TRICKLE
CHARGE
Q3
VN2222LL
R2
1k
C4
3300pF
X7R
4
C5
200pF
NPO
SHDN
1
8
SENSE +
LTC1148
7
SENSE –
ITH
VFB
CT
N-DRIVE
SGND
11
PGND
12
Q1
Si9430DY
1
4
C3
22µF
25V
×2
L1
50µH
2
3
R3
0.01Ω
D2
MBRS340T3
VOUT
C6
0.01µF
R4
274k
1%
+
9
14
C7
100pF
R5
49.9k
1%
Q2
Si9410DY
VBATT
4 CELLS
C8
220µF
10V
D1
MBRS140T3
AN66 F150
C1 = (TA)
C3 = AVX (TA) TPSD226K025R0200 ESR = 0.200 IRMS = 0.775A
C8 = AVX (TA) TPSE227M010R0100 ESR = 0.100 IRMS = 1.149A
Q1 = SILICONIX PMOS BVDSS = 20V RDSON = 0.125 CRSS = 400pF QG = 25nC θJA = 50°C/W
Q2 = SILICONIX NMOS BVDSS = 30V RDSON = 0.050 CRSS = 160pF QG = 50nC θJA = 50°C/W
D1, D2 = MOTOROLA SCHOTTKY VBR =40V
R3 = KRL SP-1/2-A1-0R100J Pd = 0.75V
L1 = COILTRONICS CTX50-4 DCR = 0.175 IDC = 1.350A KOOL Mµ CORE
VOUT = 1.25V • (1 + R4/R5) = 8.1V
FAST CHARGE = 130mV/R3 = 1.3A (EQ. 1)
TRICKLE CHARGE = 100mA (SEE FIGURE 2)
ALL OTHER CAPACITORS ARE CERAMIC
COILTRONICS (407) 241-7876
KRL (809) 668-3210
Figure 150. 4-Cell, 1.3A Battery Charger Implemented in Surface Mount Technology
AN66-76
Application Note 66
During the fast-charge interval, the resistor divider network
(R4 and R5) forces the LTC1148’s Feedback pin (VFB)
below 1.25V, causing the LTC1148 to operate at the maximum output current. R3, a 0.1Ω resistor, senses the
current and sets it at approximately 1.3A according to
equation 1 in Figure 150. When the batteries are disconnected, the error amplifier forces the Feedback pin to 1.25V,
limiting the output voltage to 8.1V. Diode D2 prevents the
batteries from discharging through the divider network
when the charger is shut down. In shutdown mode the
circuit draws less than 20µA from the input supply.
The dual rate charging is controlled by Q3, which can be
toggled between fast charge and trickle charge. The trickle
charge rate is set by resistor R1. Figure 151 is a graph
showing the value of R1 for a given trickle charge output
current. The trickle charge current can also be varied by
using an op amp to force the Threshold pin voltage within
its 0V to 2V range. Figure 152 shows the output current as
a function of Threshold pin voltage.
1400
OUTPUT CURRENT (mA)
1200
1000
800
600
400
200
0
0
1
2
R1 (kΩ)
3
4
AN66 F151
Figure 151. LTC1148 Output Current Voice
Trickle Charge Set Resistance (R1)
1400
1200
OUTPUT CURRENT (mA)
through the inductor and into the battery. When the
LTC1148 P-drive pin goes high, Q1 is turned off and the
voltage on the drain of Q1 drops until the clamp diode is
forward biased. The diode conducts for a very short period
of time, until the LTC1148 internal circuitry senses that the
P-channel is fully off, preventing the simultaneous conduction of Q1 and Q2. Then the N-drive output goes high,
turning on Q2, which shorts out D1. Now the inductor
current flows through the N-channel MOSFET instead of
through the diode, increasing efficiency. This type of switching architecture is known as synchronous rectification.
1000
800
600
400
200
0
0
1.0
0.5
1.5
THRESHOLD PIN VOLTAGE (V)
2.0
AN66 F152
Figure 152. LTC1148 Output Current
vs Forced Threshold Pin Voltage
AN66-77
Application Note 66
θHS = θJA SYSTEM – θJC FET = 55°C/5W – 1.25°C/W
Power Management
= 9.75°C/W
LT1366 RAIL-TO-RAIL AMPLIFIER
CONTROLS TOPSIDE CURRENT
by William Jett and Sean Gold
This is easily achievable with a small heat sink. When input
voltages are greater than 5V the use of a larger heat sink
or derating of the output current is necessary.
Topside Current Source
The circuit shown in Figure 153 takes advantage of the
LT1366’s rail-to-rail input range to form a wide-compliance current source. The LT1366 adjusts Q1’s gate voltage to force the voltage across the sense resistor (RSENSE)
to equal the voltage from the supply to the potentiometer’s
wiper. A rail-to-rail op amp is needed because the voltage
across the sense resistor must drop to zero when the
divided reference voltage is set to zero. Q2 acts as a
constant current sink to minimize error in the reference
voltage when the supply voltage varies.
VCC
RSENSE
0.2Ω
1k
LT1004-1.2
0.0033µF
–
100Ω
1/2 LT1366
RP
10k
+
Q1
MTP23P06
IOUT
40k
Q2
2N4340
5V < VCC < 30V
1A > ILOAD > 160mA
AN66 F153
The circuit’s supply regulation is about 0.03%/V. The
output impedance is equal to the MOSFET’s output impedance multiplied by the op amp’s open-loop gain. Degradations in current source compliance occur when the voltage
across the MOSFET’s on-resistance and the sense resistor
drops below the voltage required to maintain the desired
output current. This condition occurs when VCC – VOUT
< ILOAD (RSENSE + RON).
High Side Current Sense Amplifier
In power control it is sometimes necessary to sense load
current at low loss near the input supply. The current
sense amplifier shown in Figure 154 amplifies the voltage
across a small value sense resistor by the ratio of the
current source resistors (R2/R1). The LT1366 forces the
low power MOSFET’s gate voltage such that the sense
voltage appears across a current source resistor R1. The
resulting current in Q1’s drain is converted to a ground
referred voltage at R2. (VO = IIN RS [R2/R1])
The circuit takes advantage of the LT1366’s ability to
sense signals up to the supply rail, which permits the use
of small value, low loss sense resistors. The LT1366 and
the gain setting resistors are also biased at low current to
reduce losses in the current sense.
VCC R1
200Ω
Figure 153. Topside Current Source
The circuit can operate over a wide supply range (5V < VCC
< 30V). At low input voltage, circuit operation is limited by
the MOSFET’s gate-drive requirements. At high input
voltage, circuit operation is limited by the LT1366’s absolute maximum ratings and the output power requirements.
In this example the circuit delivers 1A at 200mV of sense
voltage. With a 5V input supply the power dissipation is
5W. For operation at 70°C ambient temperature, the
MOSFET’s heat sink must have a thermal resistance of:
AN66-78
RS
0.2Ω
–
1/2 LT1366
Q1
TPO610L
+
IIN
R2
20k
 R2 
VO = IIN R S  
 R1 
= IIN • 20Ω
AN66 F154
Figure 154. High Side Current Sense Amplifier
Application Note 66
AN ISOLATED HIGH SIDE DRIVER
by James Herr
Introduction
The LTC1146 low power digital isolator draws only 70µA
of supply current with VIN = 5V. Its low supply current
feature is well suited for battery-powered systems that
require isolation, such as an isolated high side driver. The
LTC1146A is rated at 2500VRMS and is UL approved. The
LTC1146 is intended for less stringent applications and is
rated at 500VDC.
Theory of Operation
Optoisolators available today require supply currents in
the milliampere range even for low speed operation (less
than 20kHz). This high supply current is another drain on
the battery. Figure 155 shows the alternative of using an
LTC1146A to drive an external power MOSFET (IRF840) at
speeds to 20kHz with V + = 300V.
The Input pin of LTC1146A must be driven with a signal
that swings at least 3V (referred to GND1, which is a
floating ground). The OS pin outputs a square wave
corresponding to the input signal but with a time delay.
The amplitude of the output square wave is equal to the
potential at the VCC pin. The TL4426 is a high speed
MOSFET driver used here to supply gate drive current to
the power MOSFET. The power supply to the LTC1146A
and the TL4426 is bootstrapped from a 13V supply referred to system ground. C1 supplies the current to both
the LTC1146A and the TL4426 when the power MOSFET
is being turned on. Its value should be increased when the
input signal’s ON time increases. D3 prevents the output
from swinging negative due to stray inductance. If the
output goes below ground, the gate-to-source voltage of
the IRF840 rises. This high potential could damage the
power MOSFET. The output slew rate should be limited to
1000V/µs to prevent glitches on the OS output of the
LTC1146A.
VCC = 13V V+ = 300V
D1
MUR840
R1
1.5k
+
INPUT
SIGNAL
+
VCC
C1
100µF
25V
VIN
C2
0.1µF
CER
TL4426
LTC1146A O
S
GND1
GND2
+
C3
180µF
400V
IRF840
D2
1N752A
OUTPUT
SIGNAL
GROUND
ISOLATION
BARRIER
D3
MUR1560
RL
SYSTEM GROUND
CL
AN66 F155
Figure 155. Isolated High Side Driver Schematic Diagram
AN66-79
Application Note 66
LTC1163: 2-CELL POWER MANAGEMENT
by Tim Skovmand
Schottky rectifier to charge the output capacitor to a
voltage higher than the input voltage. Unfortunately, when
the regulator is shut down, the inductor and diode remain
connected and the load may leak significant current in
standby.
The LTC1163 1.8V to 6V high side MOSFET driver allows
inexpensive N-channel switches to be used to efficiently
manage power in 2-cell systems such as palmtop computers, portable medical equipment, cellular telephones and
personal organizers.
One possible solution to this problem is to add a low
RDS(ON) MOSFET switch between the battery pack and the
input of the regulator to completely disconnect it and the
load from the battery pack. MOSFET switches, however,
cannot operate directly from 2-cell battery supplies because the gate voltage is limited to 3V with fresh cells and
1.8V when the cells are fully discharged.
Any supply voltage above 3V, such as 3.3V, 5V or 12V, can
be generated by step-up converters powered from a 2-cell
supply. Step-up regulators are typically configured as
shown in Figure 156. An inductor is connected directly to
the 2-cell battery pack and switched by a large (1A) switch.
The inductor current is then passed through a low drop
+
D1
L1
2-CELL
BATTERY
PACK
The LTC1163 solves this problem by generating gate drive
voltages that fully enhance high side N-channel switches
when powered from a 2-cell battery pack, as shown in
Figure 157. The standby current with all three drivers
switched off is typically 0.01µA. The quiescent current
rises to 85µA per channel with the input turned on and the
charge pump producing 10V (above ground) from a 3V
supply. The surface mount MOSFET switches shown are
guaranteed to be less than 0.1Ω with VGS = 5V and less
than 0.12Ω with VGS = 4V and therefore have extremely
low voltage drops.
+
SHUTDOWN
CIN
STEP-UP
SWITCHING
REGULATOR
LOAD
+
COUT
AN66 F156
Figure 156. Typical Step-Up Converter Topology
+ 2-CELL
BATTERY
PACK
IN1
CONTROL
LOGIC
OR µP
+
VS
RFD14N05LSM
OUT1
100µF
6.3V
LTC1163
IN2
RFD14N05LSM
OUT2
IN3
RFD14N05LSM
OUT3
GND
100µH
MBRS120T3
3.3V
1
22µH
22µH
MBRS120T3
MBRS120T3
12V
3
1
47Ω
1
2
3
5V
3
LT1173CS8
7
LT1109CS8-12
4
7
8
+
10µF
20V
LT1109CS8-5
4
8
8
+
22µF
16V
39k
+
24k
4
5
220µF
6.3V
AN66 F157
Figure 157. Complete 2-Cell to 3.3V, 5V and 12V Power Management System
AN66-80
Application Note 66
LTC1157 SWITCH FOR 3.3V PC CARD POWER
by Tim Skovmand
Computers designed to accept PC cards—plug-in modules specified by the Personal Computer Memory Card
International Association (PCMCIA)—have special hardware features to accommodate these pocket-sized cards.
PCMCIA-compliant cards require power management electronics that conform to the height restrictions of the three
standard configurations: 3.3mm, 5mm and 10.5mm. These
height limitations dramatically reduce the available options for power management on the card itself. For example, high efficiency switching regulators to convert the
incoming 5V down to 3.3V for the on-card 3.3V logic
require relatively large magnetics and filter capacitors,
which are not always available in packaging thin enough to
meet the tight height requirements.
One possible approach to the problem of supplying power
to a 3.3V PC card is to switch the input supply voltage from
5V to 3.3V after the card has been inserted and the
attribute ROM has informed the computer of the card’s
voltage and current requirements. The switching regulator, housed in the computer, switches the power supplied
to the connector from 5V to 3.3V.
A window comparator and ultralow drop switch on the PC
card, Q1 in Figure 158, closes after the supply voltage
drops from 5V to 3.3V, ensuring that the sensitive 3.3V
logic on the card is never powered by more than 3.6V or
less than 2.4V. A second switch, Q2, is provided on the
card to interrupt power to 3.3V loads that can be idled
when not in use.
The built-in charge pumps in the LTC1157 drive the gates
of the low RDS(ON) N-channel MOSFETs to 8.7V when
powered from a 3.3V supply. The LT1017 and the LTC1157
are both micropower and are supplied by a filter (R5 and
C2) that holds the supply pins high long enough to ensure
that the MOSFET gates are fully discharged immediately
after the card is disconnected from the power supply. A
large value bleed resistor, R6, further ensures that the high
impedance gate of Q1 is not inadvertently charged up
when the card is removed or when it is stored.
All of the components shown in Figure 158 are available in
thin, surface mount packaging and occupy a very small
amount of surface area. Further, the power dissipation is
extremely low because the LTC1157 and LT1017 are
micropower and the MOSFET switches are very low RDS(ON).
5V
3.3V
ATTRIBUTE
ROM
R1
150k
1%
+
3
2
R2
49.9k
1%
5
6
R3
100k
1%
R5
510
R4
100k
+
8
1/2
LT1017
Q1
MTD3055EL
C2
10µF
6.3V
1
IN1
–
VS
R6
5.1M
G1
LTC1157
IN2
+
SENSITIVE
3.3V
LOGIC
GND
1/2
LT1017
–
G2
Q2
MTD3055EL
4
7
SW ON/OFF
C1
FROM µP
0.1µF
LT1004-1.2
SENSITIVE
3.3V
LOGIC
AN66 F158
Figure 158. 3.3V PCMCIA Card Power Switching
AN66-81
Application Note 66
THE LTC1157 DUAL 3.3V
MICROPOWER MOSFET DRIVER
by Tim Skovmand
The LTC1157 dual micropower MOSFET driver makes it
possible to switch either supply- or ground-referenced
loads through a low RDS(ON) N-channel switch. The
LTC1157’s internal charge pump boosts the gate drive
voltage 5.4V above the positive rail (8.7V above ground),
fully enhancing a logic level, N-channel MOSFET for 3.3V
high side switching applications.
on the second channel. Slower rise and fall times are
sometimes required to reduce the start-up current demands of large supply capacitors which might otherwise
glitch the main supply.
3.3V
+
10µF
VS
IN1
µP OR
CONTROL
LOGIC
100k
LTC1157 Switches Two 3.3V Loads
Figure 160 is a schematic diagram that demonstrates the
use of the LTC1155 for switching the power buses in a
laptop computer system. The disk drive, display, printer
and the microprocessor system itself are selectively engaged via high side switching with minimum loss and are
shut down completely when not in use.
The quiescent current of the LTC1155 is designed to be
extremely low in both the OFF and ON states, so that
efficiency is preserved even when driving loads that require very little current to operate in standby, but require
much larger peak currents when in operation. This combination of a low RDS(ON) MOSFET and an efficient driver
delivers the maximum energy to the load.
AN66-82
1k
IRLR024
G2
GND
3.3V LOAD
0.1µF
+
Figure 159 illustrates how two surface mount MOSFETs
and the LTC1157 (also available in SO-8 packaging) can be
used to switch two 3.3V loads. The gate rise and fall times
are typically in the tens of microseconds, but can be
slowed by adding two resistors and a capacitor as shown
The LTC1155 is a new micropower MOSFET driver specifically designed for low voltage, high efficiency switching
applications such as those found in laptop or notebook
computers. Three applications for this versatile part are
detailed here.
3.3V LOAD
LTC1157
IN2
THE LTC1155 DOES LAPTOP COMPUTER POWER BUS
SWITCHING, SCSI TERMINATION POWER OR 5V/3A
EXTREMELY LOW DROPOUT REGULATOR
by Tim Skovmand
IRLR024
G1
LARGE
SUPPLY
CAPACITOR
AN66 F159
Figure 159. LTC1157 Used to Switch Two 3.3V Loads
VS = 4.5V TO 18V
RSENSE
0.02Ω
RDLY
300k
5A MAX
DS1
0.03Ω
MOSFET
TTL, CMOS
INPUT
+
CDLY
0.1µF
10 µF
VS
DS2
G1
LTC1155
G2
IN1
GND
IN2
CDLY
0.1µF
RDLY
300k
RSEN
0.02Ω
0.03 Ω
MOSFET
TTL, CMOS
INPUT
POWER BUS
µP
SYSTEM
DISK
DRIVE
DISPLAY
PRINTER,
ETC.
GND
AN66 F160
Figure 160. Laptop Computer Power Bus Switching
Protected SCSI Termination Power
The circuit shown in Figure 161 demonstrates how the
LTC1155 provides protected power to SCSI terminators.
The LTC1155 is initially triggered by the free-running 1Hz
oscillator (it could also be triggered by a pulse from the
microprocessor) and latches ON via the positive feedback
Application Note 66
as shown in Figure 162. The LTC1155 charge pump
boosts the gate voltage above the supply rail and continuously charges a 0.1µF reservoir capacitor. The LT1431
works against this capacitor and the 100k series resistor
to control the MOSFET gate voltage and maintain a constant 5V at the output.
provided by RFB. The power MOSFET gate is driven to 12V
and the MOSFET is fully enhanced.
The delay afforded by the two delay components, RDLY and
CDLY, ensures that the protection circuit is not triggered by
a high inrush-current load. If, however, the source of the
MOSFET is shorted to ground or if the output of LT1117 is
shorted, the delay will be exceeded and the MOSFET will be
held OFF until the pulse from the free-running oscillator
resets the input again. The drain sense resistor, RSENSE, is
selected to trip the LTC1155 protection circuitry when the
MOSFET current exceeds 1A. This current limit protects
both the LT1117 and any peripheral system powered by
the SCSI termination power line.
The regulator is switched ON and OFF by the control logic
or the microprocessor to conserve power in the standby
mode. The LTC1155 standby current drops to about 10µA
when the input is switched OFF. The total ON current,
including the LT1431 is less than 1mA.
5.5V TO 18V
+
The delay time afforded by RDLY and CDLY is chosen to be
considerably smaller than the reset time period (>100:1),
so that very little power is dissipated while the shortcircuit condition persists, i.e., the LTC1155 will deliver
small pulses of current during every reset time period until
the short-circuit condition is removed.
10µF
0.02Ω
0.1µF
VS
300k
DS2
5V
LTC1155
100k
CMOS
OR TTL
LOGIC
IN1
IRLZ24
GATE 1
GND
200pF
10µA
STANDBY
CURRENT
The LTC1155 and the LT1117, as well as the power
MOSFET shown, are available in surface mount packaging
and therefore consume very little board space.
10k
1
8
S
7
+
6
0.1µF
3
LT1431
4
5
5V/3A
+
470µF*
Extremely Low Voltage Drop Regulator
*CAPACITOR ESR SHOULD BE < 0.5Ω.
An extremely low voltage drop regulator can be built
around the LTC1155 and a low resistance power MOSFET
AN66 F162
Figure 162. 5V/3A Extremely Low Voltage Drop Regulator
VS = 4.75 TO 5.25V
+
1A MAX
CDLY
0.1µF
RDLY
30k
10µF
DS1
VS
DS2
G1
LTC1155
G2
IN1
GND
IN2
SIMILAR
CIRCUIT
RSENSE
0.1Ω
1N5817
IRLR024 OR
EQUIVALENT
RFB
100k
1 SEC
FROM
µP
1N4148
PROTECTED
TERM. POWER
1N4148
OR
+
+
510k
10µF
1µF
2.85V TO TERM.
RESISTORS
LT1117-2.85
1/6 74C14
47µF
GND
AN66 F161
Figure 161. SCSI Termination Power with Short-Circuit Protection
AN66-83
Application Note 66
A CIRCUIT THAT SMOOTHLY SWITCHES
BETWEEN 3.3V AND 5V
by Doug La Porte
be unable to react to counter the large positive voltage
step. This jump will cause damage to many low voltage
devices.
Many subsystems require supply switching between
3.3V and 5V to support both low power and high speed
modes. This back-and-forth voltage switching can cause
havoc to the main 3.3V and 5V supply buses. If done
improperly, switching the subsystem from 5V to 3.3V
can cause a momentary jump on the 3.3V bus, damaging
other 3.3V devices. When switching the subsystem from
3.3V to 5V, the 5V supply bus can be pulled down while
charging the subsystem’s capacitors and may inadvertently cause a reset.
The circuit in Figure 163 employs a comparator (IC2) and
utilizes the high impedance state of the LTC1470 to allow
switching with minimal effect on the supply. When the
3.3V output is selected, IC1’s output will go into a high
impedance state until its output falls below the 3.3V bus.
The output capacitors will slowly discharge to 3.3V, with
the rate of discharge depending on the current being
pulled by the subsystem and the size of the holding
capacitor. The example shown in Figure 163 is for a
250mA subsystem. The discharge time constant should
be about 4ms. Once the subsystem supply has dropped
below the 3.3V supply, the comparator will trip, turning on
the 3.3V switch. The comparator has some hysteresis to
avoid instabilities. The subsystem supply will reach a low
point of about 3V before the 3.3V switch is fully enhanced.
The circuit shown in Figure 163 allows smooth voltage
switching between 3.3V and 5V with added protection
features to ensure safe operation. IC1 is an LTC1470
switch-matrix device. This part has on-chip charge pumps
running from the 5V supply to fully enhance the internal
N-channel MOSFETs. The LTC1472 also has guaranteed
break-before-make switching to prevent cross conduction between buses. It also features current limiting and
thermal shutdown.
When switching the subsystem from 5V to 3.3V, the
holding capacitor and the load capacitance are initially
charged up to 5V. Connecting these capacitors directly to
the main 3.3V bus causes a momentary step to 5V. This
transient is so fast that the power supply cannot react in
time. Switching power supplies have a particularly difficult
time coping with this jump. Switching supplies source
current to raise the supply voltage and require the load to
sink current to lower the voltage. A switching supply will
1µF
+
When switching from 3.3V to 5V, IC1’s current limiting
prevents the main 5V bus from being dragged down while
charging the holding capacitor and the subsystem’s capacitance. Without current limiting, the inrush current to
charge these capacitors could cause a droop in the main
5V supply.
If done improperly, supply voltage switching leads to
disastrous system consequences. The voltage switch
should monitor the output voltage and have current limiting to prevent main supply transient problems. A correctly
designed supply voltage switch avoids the pitfalls and
results in a safe, reliable system.
5V 3.3V 1µF
5V
+
2
0 = 5V
1 = 3.3V
1k
3.3V
3
+
5VIN 3VIN 3VIN
5V
IC1
LTC1470
3.3V
0.1µF
8
IC2
LT1011
–
500mV/DIV
7
51k
3
2
6
4
7
4
5V
1
EN1
VOUT
EN0
VOUT
5V/DIV
TO
SUBSCRIBER
8
+
5
220µF
TANTALUM
HOLDING
CAPACITOR
AN66 F163
Figure 163. Schematic Diagram of 3.3V and 5V Switchover Circuit
AN66-84
0V
2ms/DIV
AN66 F164
Figure 164. Oscillograph of the
Switchover Waveform Showing
Smooth Transitions
Application Note 66
A FULLY ISOLATED QUAD 4A HIGH SIDE SWITCH
by Milton Wilcox
resulting in a total switch drop (including sense resistor)
of only 0.15V at 4A output current.
High side switching in hostile environments often requires
isolation to protect the controlling logic from transients on
the “dirty” power ground. The circuit shown in Figure 165
drives and protects four low RDS(ON) power MOSFET
switches over a wide operating supply range. The LT1161
drivers are protected from transients of up to 60V on the
supply pins and 75V on the gate pins. Fault indication is
provided by an inexpensive logic gate.
The LT1161 independently protects and restarts each
MOSFET. It senses drain current via the voltage drop
across a current shunt RS. When the current in one switch
exceeds approximately 6A (62mV/0.01Ω) the switch is
turned off without affecting the other switches. The switch
remains off for 50ms (set by external timing capacitor CT),
after which the LT1161 automatically attempts to restart it.
If the fault is still present this cycle repeats until the fault
is removed, thus protecting the MOSFET. Current shunts
are readily available in both through-hole and surface
mount case styles. AN53 has additional information on
shunts. Connect the LT1161 V + pins directly to the top of
the current shunts (see LT1161 data sheet).
Each of the four LT1161 switch channels has a completely
self-contained charge pump, which drives the gate of the
N-channel MOSFET switch 12V above the supply rail when
the corresponding Input pin is taken high. The specified
MOSFET device types have a maximum RDS(ON) of 0.028Ω,
24V
CT
0.33µF EA
5V
4.7k
NEC PS2501-4
RS
0.01Ω EA
+
V+
T1
V+
DS1
T2
DS2
T3
DS3
T4
10µF
50V
DS4
LT1161
INPUTS
4.7k
IN1
G1
4.7k
IN2
G2
4.7k
IN3
G3
MM74HC266A
2k
N-CHANNEL
MOSFETS:
IRFZ44
OR
MTP50N06E
OR
RFP50N05
IN4
G4
GND GND
ROL
2.2k EA
100k
CONNECT FOR
OPEN-LOAD
DETECTION
4N28
FAULT
OUTPUT
100k
100k
100k
AN66 F165
OUTPUTS
Figure 165. Protected Quad High Side Switch Has Isolated Inputs and Fault Output
AN66-85
Application Note 66
The highest MOSFET dissipation occurs with a “soft short”
(one in which the current is above the normal operating
level but still below the current limit threshold). This can
cause dissipation in Figure 165’s circuit to rise, in the
worst-case to 2W, requiring modest heatsinking. When an
output is directly shorted to ground the average dissipation is very low because the MOSFET conducts only during
brief restart attempts.
Fault indication is provided by a low cost exclusive NOR
gate. In normal operation a low on the LT1161 input forces
a low on the output and a high forces a high. If an input is
high and the corresponding output is low (i.e., short
circuited), the output of the exclusive NOR gate activates
the isolated fault output. Similarly, by adding resistor ROL
the low input/high output state can be used to diagnose an
open load condition. Adjusting the value of ROL sets the
output current at which the load is considered to be open.
For example, in Figure 165 with VSUPPLY = 24V, a fault
would be indicated if the load could not sink 10mA.
Figure 165’s circuit is ideal for driving resistive or inductive loads such as solenoids. However, the circuit can be
tailored for capacitive or high inrush loads as well. Consult
the LT1161 data sheet for information on programming
current limit, delay time and automatic restart period to
handle other loads. The LT1161 is available in both PDIP
and surface mount packaging.
THE LTC1153 ELECTRONIC CIRCUIT BREAKER
by Tim Skovmand
The LTC1153 electronic circuit breaker is designed to
work with a low cost, N-channel power MOSFET to interrupt power to a sensitive electronic load in the event of an
overcurrent condition. The breaker is tripped by an overcurrent condition and remains tripped for a period of time
programmed by an external timing capacitor, CT. The
switch is then automatically reset and the load momentarily retried. If the load current is still too high, the switch
is shut down again. This cycle continues until the overcurrent condition is removed, thereby protecting the sensitive load and the power MOSFET.
5V
ON/OFF
CT
0.22µF, Z5U
TO µP
The trip-delay time is set by the two delay components, RD
and CD, which establish an RC time constant in series with
the drain sense resistor, producing a trip delay that is
shorter for increasing breaker current (similar to that of a
mechanical circuit breaker). Figure 167 is a graph of the
trip-delay time versus the circuit breaker current for a 1ms
AN66-86
VS
CD
0.01µF
LTC1153
CT
DS
FLT
G
GND
SD
RSEN*
0.1Ω
RD
100k
51k
IRLR024
51k
5V
SENSITIVE
5V LOAD
70°C**
PTC
* ALL COMPONENTS SHOWN ARE SURFACE MOUNT.
** IMS026 INTERNATIONAL MANUFACTURING SERVICE, INC. (401) 683-9700
RL2006-100-70-30-PT1 KEYSTONE CARBON COMPANY (814) 781-1591
AN67 F166
Figure 166. LTC1153 5V/1A Circuit Breaker
with Thermal Shutdown
5V/1A Circuit Breaker with Thermal Shutdown
10
RSEN = 0.1Ω
RD = 100k
CD = 0.01µF
TRIP DELAY (ms)
The trip current, trip-delay time and autoreset period are
programmable over a wide range to accommodate a
variety of load impedances. Figure 166 demonstrates how
the LTC1153 is used in a typical circuit breaker application.
The DC trip current is set by a small valued resistor, RSEN,
in series with the drain lead, which drops 100mV when the
current limit is reached. In the circuit of Figure 166, the DC
trip current is set at 1A (RSEN = 0.1Ω).
IN
1
0.1
0.01
1
2
50
5
10
20
CIRCUIT BREAKER CURRENT (A)
100
AN66 F167
Figure 167. Trip Delay Time vs Circuit Breaker Current
(1ms RC Time Constant for the Circuit of Figure 166)
Application Note 66
RC time constant. Note that the trip time is 0.63ms at 2A,
but falls to 55µs at 20A. This characteristic ensures that
the load and the MOSFET switch are protected against a
wide range of overload conditions.
The autoreset time is typically set in the range of 10s of
milliseconds to a few seconds by selecting the timing
capacitor, CT. The autoreset period for the circuit in Figure
190 is 200ms, i.e., the circuit breaker is automatically
reset (retried) every 200ms until the overload condition is
removed.
An open-drain fault output is provided to warn the host
microprocessor whenever the circuit breaker has been
tripped. The microprocessor can either wait for the
autoreset function to reset the load, or shut the switch OFF
after a fixed number of retries.
The shutdown input interfaces directly with a PTC thermistor to sense overtemperature conditions and trip the
circuit breaker whenever the load temperature or the
MOSFET switch temperature exceeds a safe level. The
thermistor shown in Figure 166 trips the circuit breaker
when the load temperature exceeds approximately 70°C.
LTC1153: DC Motor Protector
5V
ON/OFF
CT
0.47µF, Z5U
TO µP
IN
VS
LTC1153
CT
DS
FLT
G
GND
SD
CD
0.22µF
RD
100k
51k
RSEN
0.02Ω
IRLR024
51k
5V
70°C
PTC
D.C.
1N5400
AN66 F168
Figure 168. DC Motor Driver with Overcurrent and
Overtemperature Protection
the motor is limited to 5A and a rather long trip delay is
used to ensure that the motor starts properly. The motor
temperature is also continuously monitored and the breaker
is tripped if the motor temperature exceeds 70°C. The fault
output of the LTC1153 informs the host microprocessor
whenever the breaker is tripped. The microprocessor can
disable the motor if a set number of faults occur or it can
initiate a retry after a much longer period of time has
elapsed. A rectifier diode across the motor returns the
motor current to ground and restricts the output of the
switch to less than 1V below ground.
A 5V DC motor can be powered and protected using the
circuit shown in Figure 168. The DC current delivered to
LTC1477: 0.07Ω PROTECTED HIGH SIDE SWITCH
ELIMINATES “HOT SWAP” GLITCHING
by Tim Skovmand
inadvertently diverted to sensitive (and expensive) integrated circuits that cannot tolerate either overvoltage or
overcurrent conditions even for short periods of time.
When a printed circuit board is “hot swapped” into a live
5V socket, a number of bad things can happen.
Third, if the card is removed and then reinserted in a few
milliseconds, the glitching of the supply may “confuse”
the microprocessor or peripheral ICs on the card, generating erroneous data in memory or forcing the card into an
inappropriate state.
First, the instantaneous connection of a large, discharged
supply bypass capacitor may cause a glitch to appear on
the power bus. The current flowing into the capacitor is
limited only by the socket resistance, the card trace
resistance, and the equivalent series resistance (ESR) of
the supply bypass capacitor. This supply glitch can create
real havoc if the other boards in the system have poweron RESET circuitry with thresholds set at 4.65V.
Second, the card itself may be damaged due to the large
inrush of current into the card. This current is sometimes
Fourth, a card may be shorted, and insertion may either
grossly glitch the 5V supply or cause severe physical
damage to the card.
Figure 169 is a schematic diagram showing how an
LTC1477 protected high side switch and an LTC699
power-on RESET circuit reduce the chance of glitching or
damaging the socket or card during “hot swapping.”
AN66-87
Application Note 66
5V
C1
1µF
1
2
R1
510k
NC
6
VOUT
VIN1
VIN2
+
CLOAD
100µF
LTC1477CS8
7
LTC699CS8
VOUT
VOUT,
ISC = 2A
VIN2
VIN3
EN
GND
DISABLE
3
4
5
8
Q1
2N7002
AN66 F169
Figure 169. “Hot Swap” Circuit Featuring LTC1477 and LTC699
The LTC1477 protected high side switch provides extremely low RDS(ON) switching (typically 0.07Ω) with
built-in 2A current limiting and thermal shutdown, all in an
8-pin SO package.
100µF) is used, the LTC1477 will limit the inrush current
to 2A and ramp the capacitor at an even slower rate.
Further, the board is protected against short-circuit conditions by limiting the switch current to 2A.
As the card is inserted, the LTC699 power-on RESET
circuit holds the Enable pin of the LTC1477 low for
approximately 200ms. When the Enable pin is asserted
high, the output is ramped on in approximately 1ms. Even
if a very large supply bypass capacitor (for example, over
The 5V card supply can be disabled via Q1. The only
current flowing is the standby quiescent current of the
LTC1477, which drops below 1µA, the 600µA quiescent
current of the LTC699 and the 10µA consumed by R1.
Miscellaneous
grounding the OSC pin (Pin 7). When off the LTC1044
draws only 2µA.
PROTECTED BIAS FOR GaAs POWER AMPLIFERS
by Mitchell Lee
Portable communications devices such as cellular telephones and answer-back pagers rely on small GaAsFETbased 0.1W to 1.0W RF amplifiers as the transmitter output
stage. The main power device requires a negative gate bias
supply, which is not readily available in a battery-operated
product. The circuit shown in Figure 170 not only develops
a regulated negative gate bias, it also switches the positive
supply, protects against the loss of gate bias, limits power
dissipation in the amplifier under high standing-wave ratio
(SWR) conditions and protects against amplifier failures
that might otherwise short circuit the battery pack.
Negative bias is supplied by an LTC1044 charge pump
inverter and the amplifier’s positive supply is switched by
an LTC1153 electronic circuit breaker. An open-collector
switch can be used to turn the LTC1044 inverter off by
AN66-88
The negative output from the LTC1044 is sensed by a 2.5V
reference diode (IC2) and Q2. With no negative bias available, Q2 is off and Q3 turns on, pulling the LTC1153’s
control input low. This shuts off the GaAs amplifier. Total
standby power, including the LTC1044, is approximately
25µA.
If the LTC1044’s OSC pin (Pin 7) is released, a negative
output nearly equal in magnitude to the battery input voltage
appears at VOUT (Pin 5). The negative bias is regulated by
R1, IC2 and Q2’s base-emitter junction. Q2 saturates,
shutting Q3 off and thereby turning the LTC1153 on.
The LTC1153 charges the N-channel MOSFET (Q4) gate to
10V above the battery potential, switching Q4 fully on.
Power is thus applied to the GaAs amplifier.
The nominal negative bias is – 3.2V, comfortably assuring
the – 2.5V minimum specified for the amplifier. Total
Application Note 66
7.2V
(6 NiCd CELLS)
R3
1M
+
R4
1M
1
V+
BOOST
8
C2
100nF
2
+
4
CAP–
VOUT
OFF
ON
5
Q1
C3
1µF
2
C7
1nF
C6
220nF
4
R6
5.1M
GND
R2
10k
IC2
LT1004CS8-2.5
R1
3.3k
+
C1
1µF
7
CAP+ IC1 OSC
LTC1044CS8
3
6
GND
LV
+
1
C5
100nF
C4
1µF
VS
7
CT
IC3 DS
LTC1153CS8
3
6
STATUS GATE
Q3*
Q2*
IN
8
SHDN
5
C8
10µF
GaAsFET
AMPLIFIER
FAULT
R7
0.05Ω
R5
1k
Q4
IRFR024
VDD
RF
OUT
GATE BIAS
R8
1M
* ZETEX ZTX 384
ZETEX (516) 543-7100
OR MOTOROLA MMBT3904
AN66 F170
Figure 170. Schematic Diagram
quiescent current, exclusive of the GaAs amplifier drain
supply, is approximately 1.5mA in the ON state.
Short circuits or overcurrent conditions in the GaAs amplifier can damage the circuit board, the batteries or both. The
LTC1153 senses the amplifier’s supply current and turns
Q4 off if it is over 2A. After a timeout period set by C6
LT1158 H-BRIDGE USES GROUND REFERENCED
CURRENT SENSING FOR SYSTEM PROTECTION
by Peter Schwartz
The LT1158 half-bridge motor driver incorporates a number of powerful protection features. Some of these, such
as its adaptive gate drive, are dedicated in function. Others
are open to a variety of uses depending upon application
requirements. The circuit shown in Figure 171 takes
advantage of the wide common mode input range of the
LT1158’s fault comparator to perform ground referenced
current sensing in an H-bridge motor driver. By using
ground referenced sensing, protection can easily be provided against overloaded, stalled or shorted motors. For
overloads and stalls the circuit becomes a constant current chopper, regulating the motor’s armature current to
a preset maximum value. For shorted loads the circuit
protects itself by operating at a very low duty cycle until the
short is cleared.
(200ms) the LTC1153 tries again, turning Q4 on. If the
amplifier’s supply current is still too high the LTC1153 trips
off again. This cycle continues until the fault condition is
cleared. Under fault conditions the LTC1153’s Status pin
(Pin 3) is low. As soon as the fault is cleared the LTC1153
resets and normal operation is restored.
Setting Up for Ground Referenced Sensing
The circuit of Figure 171 is essentially a straightforward
LT1158 H-bridge of the “sign/magnitude” type. (See the
LT1158 data sheet for a description of component functions.) In many LT1158 applications, a current sense
resistor is placed in each upper MOSFET source lead. This
circuit, however, senses the IR drop across one resistor
(R1) common to the sources of both lower MOSFETs. In
Figure 171, U1’s FAULT output activates the constant
current protection mode (for overloads and stalls) and
U2’s FAULT output indicates a shorted load. Hence, given
a maximum continuous motor current of 15A, R1’s value
is easily determined: VSENSE+ of U1 must exceed VSENSE –
by the LT1158’s internal threshold of 110mV in order for
FAULT to go low. 15A • R1 = 0.110V, so R1 = (0.110V/15A)
at 0.0075Ω. The FAULT pin of U2 should go low when IR1
is 24A, so a 1.6:1 voltage divider is added at U2’s Sense+
input. R2, R3, C1 and C2 filter any switching spikes that
appear across R1.
AN66-89
Application Note 66
24V
470µF*
2
5V
V+
10
+
BOOST DR
V+
BOOST
10µF
47k
5
FAULT
4
ENABLE
6
INPUT
3
BIAS
1
1N4148
1N4148
16
15
T GATE DR
14
T GATE FB
13
T SOURCE
0.1µF
Q1**
IRFZ44
33Ω
1N5819
7
9
B GATE DR
8
B GATE FB
GND
Q3**
IRFZ44
0.1µF
33Ω
15
T GATE DR
14
T GATE FB
13
T SOURCE
1N5819
M
SENSE+
12
C1
0.01µF
11
SENSE–
Q2**
IRFZ44
33Ω
33Ω
9
B GATE DR
8
B GATE FB
160Ω
R1
0.0075Ω
3W
TWISTED PAIR
12
5V
+
10µF
47k
5
FAULT
4
ENABLE
6
INPUT
3
BIAS
U2
LT1158
Q4**
IRFZ44
R3
100Ω
R2
100Ω
2
V+
10
V+
1
BOOST DR
16
BOOST
DC MOTOR
(15A CONT)
U1
LT1158
0.01µF
+
+
470µF*
0.01µF
7
GND
SENSE+
270Ω
C2
0.01µF
TWISTED PAIR
11
SENSE–
100Ω
39k
5V
220k
U3A
15
1000pF
5V
REXT/CEXT
74HC221 Q 13
14
CEXT
1
A
4
2
Q
B
3
CLR
1N4148
5
6
5V
U4B
74HC02
7
4
0.047µF
Q1 TO Q4 MOUNTED ON HEAT SINK
2
3
PWM (“MAGNITUDE”)
4.7k
1
8
9
1N4148
U4C
74HC02
4.7k
U4A
74HC02
10
11
12
DIRECTION (“SIGN”)
4.7k
* LOW ESR CAPACITORS (SPRAGUE 673D, ETC.)
5V
REXT/CEXT
6
CEXT
9
A
10
B
11
CLR
Q
Q
5
12
10k***
U3B
74HC221
** DIODE SHOWN IS THE MOSFET’S INTEGRAL
DRAIN-BODY DIODE.
13
*** PULLDOWN FOR “ENABLE” LINE IN CASE
5V IS NOT PRESENT.
AN66 F171
U4D
74HC02
Figure 171. H-Bridge Motor Driver with Ground Referenced Current Sensing
Closing the Loop on Overloads
If the motor is overloaded or stalled, its back EMF will drop,
causing the armature current to increase at a rate determined primarily by the motor’s inductance. Without protection this current could rise to a value limited only by
supply voltage and circuit resistance. The necessary protection is provided via the feedback loop formed by U1’s
FAULT output, U3A, U4B and U4D. When IR1 exceeds 15A,
the FAULT pin of U1 conducts, triggering the 40µs
monostable U3A. The Q output of U3A in turn forces the
outputs of U4B and U4D to a logic low state, turning off Q1
or Q3, and turning on both Q2 and Q4. For the time during
which U3A’s Q output is high, the motor current decays
through the path formed by the motor’s resistance, plus
AN66-90
the on-resistance of Q2 and Q4 in series. In this application, turning both lower MOSFETs on is preferable to
forcing all four MOSFETs off, as it provides a low resistance recirculation path for the motor current. This reduces motor and supply ripple currents, as well as MOSFET
dissipation. At the end of U3A’s 40ms timeout the Hbridge turns on again. If the overload still exists, the
current quickly builds up to the U1 FAULT trip point again
and the 40ms timeout repeats. This feedback loop holds
the motor current approximately constant at 15A for any
combination of supply voltage and duty cycle that would
otherwise cause an excess current condition. When the
motor’s current draw falls below 15A, the circuit resumes
normal operation.
Application Note 66
Opening the Loop on Shorts
A Final Note
In the event of a short across the motor terminals the
current through the H-bridge rises faster than the U1/U3A
loop can regulate it. This could easily exceed the safe
operating area limits of the MOSFETs. The solution is
simple: when the fault comparator of U2 detects that IR1 ≥
24A, monostable U3B is triggered. The Q output of U3B
will then hold the enable line of the two LT1158s low for
10ms, resulting in a rapid shutdown and a very low duty
cycle. After the 10ms shutdown interval, U3B’s Q output
will return high and the bridge will be reenabled. If the
motor remains shorted, U3B is triggered again, causing
another 10ms shutdown. When the short is cleared, circuit
operation returns to that described above.
As a class, sign/magnitude H-bridge systems are susceptible to MOSFET and/or motor damage if the motor velocity is accelerated rapidly, or the state of the DIRECTION line
is switched while the motor is rotating. This is especially
true if the motor/load system has high inertia. The circuit
of Figure 171 is designed to provide protection under
these conditions: the motor may be commanded to accelerate and to change direction with no precautions. For the
case of deceleration, however, it’s generally best to use a
controlled velocity profile. If a specific application requires
the ability to operate with no restrictions upon the rate of
change of duty cycle, there are straightforward modifications to Figure 171 that allow this. Please contact the
factory for more information.
LT1158 ALLOWS EASY 10A
LOCKED ANTIPHASE MOTOR CONTROL
by Milton Wilcox
pitfalls encountered in the design of high efficiency motor
control and switching regulator circuits.
Allowing synchronous control of two N-channel power
MOSFETs operating from 5V to 30V, the LT1158 halfbridge driver effectively deals with the many problems and
0.1µF
0.1µF
1N4148
BAT82
15Ω
+
IRFZ34
T DR
1N4148
BAT82
10V TO 30V
BOOST
BOOST
DR
Figures 172a and 172b illustrate a locked antiphase motor
drive in which the motor stops if either side is shorted to
ground (since a 50% input duty cycle is used to stop the
motor in locked antiphase operation, the motor would
BOOST
(2) 500µF
LOW ESR
IRFZ34
T DR
T FB
T FB
SRC
SRC
LT1158
SENSE+
LT1158
SENSE+
0.015Ω
0.015Ω
SENSE–
SENSE–
15Ω
B DR
IRFZ34
2.4k
2.4k
IRFZ34
15Ω
B DR
B FB
B FB
IN
PWM
INPUT
BOOST
DR
15Ω
IN
AN66 F172a
1/2
74HC132
Figure 172a. 10A Locked Antiphase Full-Bridge Circuit Operates Over Wide Supply Range
AN66-91
Application Note 66
5V
1/2 74HC132
5k
0.01µF
FROM LT1158
FAULT PINS
TO LT1158
ENABLE PINS
RT
150k
CT
0.1µF
1N4148
AN66 F172b
Figure 172b. Protection Logic Stops Motor if Either Side Is
Shorted to Ground
normally accelerate to half speed with one side shorted).
When a fault is detected by either LT1158, the Figure 172b
latch is set, disabling both LT1158s. The circuit periodically tries restarting the motor at a time interval determined by RT and CT. If the short still exists, the disabled
state is resumed within 20µs, far too short a time to move
the motor.
ALL SURFACE MOUNT PROGRAMMABLE
0V, 3.3V, 5V AND 12V VPP GENERATOR FOR PCMCIA
by Jon A. Dutra
Generating the VPP voltage for a PCMCIA port in laptop
computers has become more complicated with PCMCIA
standard 2.0. The VPP line must come up to 5V initially
until the card “tuple” tells the card its type and VPP
voltage. For example, a 3.3V SRAM card must have VPP
adjusted to 3.3V. If it is a flash memory card, 12V must be
supplied during programming. During card insertion, 0V
is desirable to unconditionally prevent latch-up. Shutdown supply current must be as low as possible and the
supply must not overshoot. This design idea presents a
circuit (Figure 173) that meets these specifications. The
same topology could be useful for generating other programmable supplies.
The circuit uses the LT1107 micropower DC/DC converter
with a single surface mount transformer. The LT1107
features an ILIM pin that enables direct control of maximum inductor current. This allows use of a smaller transformer without risk of saturation. The LT1111 could also
be used with a reduction in output power.
AN66-92
The LT1158 can be used with virtually any N-channel
power MOSFET, including 5-lead current sensing
MOSFETs. This configuration offers the benefit of no-loss
current sensing, since a current shunt is no longer needed
in the source. In addition, RSENSE increases by a factor of
1000 or more: from milliohms to ohms. The LT1158 can
also be used with logic level MOSFETs for operation as low
as 4.5V if a Schottky boost diode is used and connected
directly to the supply.
The LT1158 N-channel power MOSFET driver anticipates
all of the major pitfalls associated with the design of high
efficiency bridge circuits. The designed-in ruggedness
and numerous protection features make the LT1158 the
best solution for 5V to 30V medium-to-high current synchronous switching applications.
VIN
5V ±
10%
+
1/4 CD4066
C
“ENABLE”
CIN
10µF
16V
VO
0V, 3.3V,
5V, 12V
0mA TO
60mA
1N5819
1/4 CD4066
T1*
CTX33-4
+
C1
1µF
16V
COUT
56µF
35V
30Ω
100k
VIN
+
165k
1%
ILIM
FB
SW1
LT1107
(1.25V)
AO
SET
SW2
GND
29.4k
1%
1/4 CD4066
1N5819 OR
MBRS140
*COILTRONICS (407) 241-7876
B
121k
1%
100k
1%
1/4 CD4066
A
AN66 F173
Figure 173. Schematic Diagram for VPP Generator
Application Note 66
Circuit Operation
The circuit is basically a gated-oscillator flyback topology.
The SET pin of the LT1107 is held at 1.25V by negative
feedback. Summing currents into the SET pin to zero for
the three different output states yields three equations
with three unknown resistor values. The resistor values
are easily solved for using Mathametica, MathCad or
classical techniques. Table 1 shows the output voltage
truth table.
Table 1
INPUTS
OUTPUTS
A
B
ENABLE
VO
NOTE
X
X
0
0V
Off
1
1
1
12V
12V
1
0
1
5V
5V
0
1
1
10.3V
Not Used
0
0
1
3.3V
3.3V
Output noise is reduced by using the auxiliary gain block
(AGB) in the feedback path. This added gain effectively
reduces the hysteresis of the comparator and tends to
randomize output noise. With a low ESR capacitor for C1,
output noise is below 30mV over the output load range.
Output power increases with VBATTERY from about 1.4W
out with 5V in to about 2W out with 8V or more. Efficiency
is 62% to 76% over a broad output power range. No
minimum load is required.
Component Selection
Substantial current flows through CIN and COUT. Most
tantalum capacitors are not rated for current flow and can
result in field failures. Using a rated tantalum or rated
electrolytic will result in longer system life.
Shutdown
The circuit is shut down by using two sections of the
CD4066 in parallel as a high side switch. Alternatively,
simply disabling the logic supply to the VIN and ILIM nodes
of the LT1107 will shut it down. This drops quiescent
current from the VBATTERY input below 2µA. When the
device is shut down VOUT drops to 0V.
VTERMINAL
A TACHLESS MOTOR SPEED REGULATOR
by Mitchell Lee
A common requirement in many motor applications is a
means of maintaining constant speed with variable loading or variable supply voltage. Speed control is easily
implemented using tachometer feedback, but the cost of
a tach may be prohibitive in many situations and adds
mechanical complications to the product. A lower cost
solution with no moving parts is presented here.
Motor speed changes under conditions of varying loads
because of the effects of series loss terms in the motor.
The effects of the predominant contributors to loss, copper and brush/commutator resistance (collectively known
as RM), are best understood by considering the circuit
model for a motor (see Figure 174). A motor’s back EMF
(VM) is proportional to speed (n) and the motor current
(IM) is proportional to the load torque (T). The following
equation predicts the speed of the motor for any given
condition of loading:
–R2 (RS)
R3
RM
IM = T/KT
+
–
VREF
(
R1 + R2
R1
)
+
–
SPEED
REGULATOR
VM = K V (n)
MOTOR
AN66 F174
Figure 174. On the Right is Shown an Equivalent Circuit for a
Motor. On the Left is the Model for a Circuit Which Will Stablize
the Motor’s Speed Against Changes in Supply Voltage and
Loading
)
RM
V
n = TERMINAL – T
KV
(KT)(KV)
)
(1)
where KV and KT are constants of proportionality for
rotational velocity and torque. For a fixed terminal voltage,
the speed of the motor must decrease as increasing load
AN66-93
Application Note 66
terminal voltage by an amount equal to (IM)(RM). Depending on the value of R3, the speed can be made to increase,
decrease or stay the same under load. If R3 is just right, the
motor speed will remain constant until the LT1170 reaches
full power and the circuit runs out of steam.
torque is applied to the shaft. For a fixed load, the speed of
the motor will also change if the supply (terminal) voltage
is changed.
A voltage regulator fixes the problem of a varying terminal
voltage, but the only way to eliminate torque from Equation (1) is by reducing RM to zero. Physically this is
impractical, but an electrical solution exists.
Many small motors in the 1W to 10W class are not well
characterized. In order to choose proper component values for a given motor, figures for RM and VM are necessary. Fortunately, these are easily measured using a DVM
and a motor characterization test stand. If you don’t have
a motor characterization test stand, it is also possible to
use a lathe or drill press to do the job.
If a motor is driven from a regulated source whose output
impedance is opposite in sign and equal in magnitude to
RM (see Figure 174), the result is a motor that runs at a
constant speed—regardless of loading and power source
variations. Figure 175 shows a circuit that does it all. The
LT1170 is configured as a buck/boost converter, which
can take a wide ranging 3V to 20V input source and
produce a regulated output of, say, 6V. The circuit shown
can deliver 1A at 6V with a 5V input, adequate for many
small permanent-magnet DC motors.
Chuck up the candidate motor’s shaft in a variable speed
drill press or lathe, which is set to run at the same speed
you’re intending to operate the motor. Clamp down the
motor frame so it won’t spin. Turn on the big machine, and
measure the open-circuit motor terminals with a DVM.
This is the motor voltage, VM, as shown in Figure 174.
Switch the meter to measure the motor’s short circuit
current, ISC. Motor resistance RM = VM/ISC. With these
figures the other component values can be calculated:
To cancel the effects of the motor resistance, a negative
output impedance is introduced with an op amp and a
current sense resistor (RS). As the motor current increases, the LT1006 responds by increasing the motor
3V TO 20V
INPUT
L1
50µH
+ C1
C2
+330µF
1000µF
VIN
R2
6490Ω
1%
VSW
LT1170
R3
309Ω
1%
L2
50µH
FB
R1A
619Ω
1%
GND
MBRD340CT
VC
R5
1k
+
LT1006
C5
2.2µF
–
R4
1k
C4
1µF
10k
1%
R1B
619Ω
1%
AN66-94
1000µF
RM
+
M VM
–
10k
1%
L1 = L2 = 50-2-52 COILTRONICS (407-241-7876). CAN BE COUPLED AS INDICATED BY PHASING DOTS.
LT1006 POWER SUPPLY PINS CONNECTED TO INPUT SUPPLY.
Figure 175. Tachless Motor Speed Regulator
+ C3
RS
0.1Ω
COPPER
AN67 F175
Application Note 66
IMAX = motor current at full load
VREF = 1.244V
R1 = series combination of 619Ω + 619Ω = 1238Ω
RS ≤ 1/IMAX (drops less than 1V at maximum load)
R2 = (VM • R1/VREF) – R1
(2)
R3 = (R2 • RS)/(RM + RS)
(3)
The component values shown in Figure 175 are for a small
motor with the following characteristics measured at
360RPM: VM = 7.8V, ISC = 3.7A, RM = 2.1Ω, IMAX ≈ 1A.
RS, a copper resistor, is either located close to or wound
around the motor to assist in tracking changes in armature
resistance with temperature. Copper has a strong,
3930ppm/°C temperature coefficient, matching the TC of
the motor winding.
Setup Procedure
Initial tests should be performed with a potentiometer in
place of, and twice the value of, R3. R5 and C5 should be
disconnected; remove all loading from the motor. Check
the motor’s unloaded speed, and adjust R2 if necessary to
set it precisely.
With the motor driving a nominal load, decrease R3 until
the motor commences “hunting.” R3 will be near the
nominal calculated value. This threshold is very close to
optimum motor resistance cancellation. R5 and C5 offer a
convenient means of compensating for frictional and
inertial effects in the mechanical system, eliminating instabilities. System stability should be evaluated under a
variety of loading conditions. The effect of R5 is to reduce
the negative output impedance of the circuit at high
frequencies. Systems with a net positive impedance are
inherently stable.
When the system stability is satisfactory, a final adjustment of R3 can be made to achieve the desired speed
regulation under conditions of varying loads. These final
values can be used in production. Note that R2 defines the
regulated speed value and may be production trimmed in
precision applications.
LT1161: … AND BACK AND STOP AND FORWARD
AND REST — ALL WITH NO WORRIES AT ALL
by Peter Schwartz and Milt Wilcox
A Complete, Six-Part Plan
Many applications of DC motors require not only the ability
to turn the motor on and off, but also to control its direction
of rotation. When directional control is involved, the need
for rapid deceleration (electronic braking) can also be
assumed. A microcontroller interface (logic-level control)
is a necessity in modern systems, as is protection of both
the motor controller and the motor itself. With the advent
of high power, logic-level N-channel MOSFETs, it is a
straightforward matter to build the lower half of an
H-bridge suitable for the versatile control of DC motor
loads. Equivalent performance P-channel MOSFETs, however, are still expensive devices of limited availability, even
without logic-level capability. Therefore, motor control
circuits commonly use N-channel devices for the upper
half of the H-bridge as well. The trick is to do this without
requiring an additional power supply to provide bias for
the upper MOSFET gates, while ensuring the necessary
system protections.
❏ Motor Forward Rotation—In this mode, Q1 and Q4 are
on, and Q2 and Q3 are off.
The circuit shown in Figure 176 is a complete H-bridge
motor driver, with six distinct modes of operation:
❏ Motor Reverse Rotation—In this mode, Q2 and Q3 are
on, and Q1 and Q4 are off.
❏ Motor Stop—Here, a rapid stop is performed by using
“plugging braking,” wherein the motor acts as a generator to dissipate mechanical energy as heat in the
braking circuit’s resistance.
❏ Motor Idle—All four MOSFETs are turned off. The
motor is, in effect, disconnected from the H-bridge
driver.
❏ Load Protect—If the motor is overloaded or stalled for
an excessive period, the on-chip fault detection and
protection circuitry of the LT1161 will shut the motor
off for programmed interval, then turn it back on.
AN66-95
Application Note 66
❏ Short-Circuit Protect—If a source-to-ground short is
detected on either Q1 or Q2, the on-chip fault detection
and protection circuitry of the LT1161 will shut off the
MOSFET at risk for the programmed interval and then
attempt to turn the circuit back on.
terminals of Q1. This could otherwise happen under
certain conditions of “motor-idle” operation. D4 serves
the same function for Q2.
Figure 176 shows a straightforward H-bridge using four
N-channel MOSFETs (Q1 to Q4). The lower MOSFETs (Q3
and Q4) are logic-level devices to allow direct drive from
5V logic. The upper MOSFETs (Q1 and Q2) are driven via
level translation circuitry integral to the LT1161. Input 1 of
the LT1161 controls a charge pump in the IC, whose
output is developed on Gate 1. Similarly, Input 2 controls
a charge pump whose output is available on Gate 2. The
Gate outputs have voltage swings from 0V to (VCC + 12V),
which is more than sufficient to enhance a standard
threshold N-channel MOSFET, such as the IRFZ34. D3 is
added to Q1 as a gate-source protection diode to prevent
excessive voltage from appearing across the gate-source
The logic of the circuit is straightforward and could be
replaced by a microcontroller in many applications. CMOS
inverters U1 and U2 drive the lower MOSFETs directly
from a 5V supply, with the RCD networks on their inputs
providing the necessary timing to prevent shoot-through
currents in the MOSFET switches. Inverter U3 and NOR
gate U5 work together to turn Gate 1 and hence Q1 on
when point A is at a logical high. This also ensures that C3
is charged to a logical high to take U2’s output low and turn
Q3 off. Under these conditions, with point B low (or left
floating), U1 will turn Q4 on and U6 will hold Gate 2 and
hence Q2 off. If point A is now immediately taken low (or
left floating), and point B is taken high, the symmetry of the
+
C1
10µF
6.3V
C2
10µF
6.3V
+
U5
74HC02
24V
+
10µF
35V
+
U3
74HC14
The Logic Behind It All
11
20
V+
2
4
3
C5
1µF
6.3V
SENSE1
TIMER2
SENSE2
INPUT1
GATE1
19
5
U6
74HC02
NC
NC
6
8
7
9
INPUT2
GATE2
TIMER3
SENSE3
TIMER4
SENSE4
INPUT3
GATE3
INPUT4
GATE4
GND
1
D1
BAT85
18
D2
BAT85
16
R2
0.01Ω
Q1
IRFZ34
Q2
IRFZ34
D3
1N4148
15
+
470µF
35V
RETURN
D4
1N4148
13
14
12
NC
M
NC
GND
10
U2
74HC14
1N4148
Q3
IRLZ34
“POINT A”
10k
R1
0.01Ω
R4
10k
17
LT1161
U4
74HC14
C6
1µF
6.3V
R3
10k
V+
TIMER1
+
1M
C3
1nF
1M
C4
1nF
1N4148
Q4
IRLZ34
U1
74HC14
“POINT B”
10k
Figure 176. LT1161-Based H-Bridge Motor Driver Schematic Diagram
AN66-96
AN66 F176
Application Note 66
logic will reverse these conditions—but only after C3 has
discharged to the point where the output of U2 can go high
to turn Q3 on. This is the shoot-through prevention
mentioned previously.
There are two exceptions to the symmetry of the logic: if
both point A and point B are low, both upper MOSFETs are
turned off while both lower MOSFETs are turned on. Under
these conditions, the kinetic energy stored in the motor
and its load is used to drive the motor as a generator. This
produces a current through the motor winding, Q3 and Q4.
In this “plugging braking” mode, the motor’s energy is
largely dissipated as I2R losses and a rapid stop occurs. If
point A and point B are both high, all four MOSFETs will be
turned off and the motor is essentially disconnected from
the electrical circuit. Although primarily included as a
cross-conduction interlock in the event that both inputs
should ever be high at the same time (things do happen on
the test bench), this can also be useful in situations where
it is desirable that the motor coast down from a higher
velocity to a lower one.
Just a Few Grams… But Lots of Protection
In addition to its level translation and charge pump features, the LT1161 also provides comprehensive protection features via its Sense 1 and Sense 2 pins. Each Sense
pin is the (–) input to an on-chip comparator, with the (+)
input to that driver’s comparator fixed at a level 65mV
(nominal, 50mV minimum) below the LT1161’s V + input.
If a Sense pin goes more than 65mV below V +, several
things happen: the corresponding Gate output is rapidly
pulled to ground, the capacitor on the Timer pin is dumped
to ground and the charge pump is shut off. The charge
pump will remain shut off, and the Gate pin will remain
clamped to ground until the Timer capacitor has charged
back up to 3V from an on-chip 14µA current source. When
the capacitor reaches this 3V threshold, the internal charge
pump starts up again and the clamp from the Gate pin to
ground is removed. The net effect of this is that, if one of
the Sense pins is pulled 65mV below V +, the MOSFET
turns off for a period that is set by the value of the capacitor
connected to the Timer pin. At the end of this programmed
interval the circuit will automatically restart. If the fault has
been cleared, the protection circuitry then becomes transparent to the system. This shutdown/retry cycle will repeat
until the fault is cleared.
The fault scenarios for which protection is required are, as
mentioned above, an overloaded or stalled motor or a
source-to-ground short on Q1 or Q2. In each case such a
fault will cause excessive current to flow through the
affected upper MOSFET; this current is readily transformed into a voltage by a current shunt resistor. Allowing
for a 5A motor current under load, this yields a resistor
value of [5A/50mV (min)] = 0.01Ω for R1 and R2. To allow
for inrush current when the motor starts up or changes
direction, delay networks (R3/C5 and R4/C6) have been
added to each half of the H-bridge. At a 20A startup
current, the values shown give a 3ms delay. The value of
the capacitor can be changed to affect longer or shorter
delays as needed (the resistor value should not be raised
above 10k). A short-to-ground fault, however, requires a
shutdown in microseconds, not milliseconds. This is
accomplished by adding two BAT85 signal level Schottky
diodes (D1 and D2) in parallel with the 10k delay resistors.
At a fault current of approximately 45A, which is easily
attained in the short-circuit case, VSHUNT = 0.45V. At this
voltage the appropriate diode conducts to temporarily
bypass the delay resistor, allowing the LT1161 to turn off
the imperiled MOSFET within 20µs (typical). In each case,
the retry interval is programmed by C1 and C2; the 10µF
shown gives a time-out of about 1.8 seconds.
The LT1161 is a quad driver IC, capable of providing drive
and protection for two additional MOSFETs beyond those
shown in Figure 176.
AN66-97
Application Note 66
SIMPLE THERMAL ANALYSIS — A REAL COOL
SUBJECT FOR LTC REGULATORS
by Alan Rich
VIN
VOUT
ILOAD
As the temperatures go up... so go the problems with
voltage regulators.
PDISS = (VIN – VOUT) ILOAD (WATTS)
Introduction
Linear Technology Corporation applications engineers get
lots of calls saying, “that $X%#@& voltage regulator is so
hot I can’t touch it!” The purpose of the article is to show
you, the design engineer, how to perform simple thermal
calculations to determine regulator temperature and select the proper package style and/or heat sink. In addition,
it will show an alternate method of specifying thermal
parameters on LTC voltage regulators.
Figure 177. Typical Linear Regulator Circuit
The last two terms are determined by how a regulator is
mounted to the heat sink and by the properties of the heat
sink. Heat sinks are used to decrease the thermal resistance
and therefore lower the temperature rise of the regulator.
Temperature is a term with which we are all very familiar.
All thermal calculations will use the Centigrade scale or °C.
TJ — temperature of the junction of the regulator die
TC — temperature of the case of the regulator
TA — ambient temperature
Definition of Terms
Power dissipation is the parameter that causes a regulator
to heat up; the unit for power is watts. Power is the product
of the voltage across a linear regulator times the load
current (see Figure 177).
Thermal resistance is a measure of the flow of heat from
one surface to another surface; the unit of thermal resistance is °C/watt. Common terms for thermal resistance
that show up on most LTC data sheets are:
θJC —thermal resistance from the junction of the die to
the case of the package
θJA —thermal resistance from the junction of the die to
the ambient temperature
Some typical LTC regulators and their thermal characteristics are shown in Table 1.
Table 1. θJC and θJA for Three LTC Regulators
DEVICE
θJC (°C/W)
θJA (°C/W)
LT1005CT
5.0
—
LT1083MK
1.6
—
LT1129CT
5.0
50
The maximum operating junction temperature, TJ MAX for
LTC regulators is shown on the device data sheet.
What is Thermal Analysis?
The goal of any thermal analysis is to determine the
regulator junction temperature, TJ, to ensure that this
temperature is less than either the regulator rating or a
design specification. In the simplest case, temperature
rise is calculated by multiplying the power times the total
of all thermal resistance:
TR = P(θTOTAL)
θTOTAL includes the thermal resistance junction-to-case
(θJC), thermal resistance case-to-heat sink (θCS), and
thermal resistance heat sink-to-ambient (θSA).
TR represents the temperature rise above the ambient
temperature; therefore, to determine the actual junction
temperature of the regulator, the ambient temperature
must be added to TR:
Regulator junction temperature =
Ambient Temperature + TR
There are several other common thermal resistance terms:
θCS — thermal resistance from the case of the package
to the heat sink
θSA — thermal resistance from a heat sink surface to
the ambient temperature
AN66-98
AN66 F177
For example, consider a circuit using an LT1129CT operating in a 50°C enclosure with an input voltage of 8VDC, an
output voltage of 5VDC and a load current of 1A1.
1
The LT1129CT is guaranteed for 700mA, but could be selected to output 1A.
Application Note 66
The power dissipated by the LT1129CT is:
P = (VIN – VOUT)(ILOAD) = (8V – 5V)(1a) = 3W
The first question is, does this circuit need a heat sink?
Since we have assumed no heat sink on the LT1129CT for
the purpose of this calculation, we must use thermal
resistance from junction to ambient, θJA = 50°C/W.
TJ = P(θJA) + TA = 3W(50°C/W) + 50°C
= 150°C + 50°C = 200°C
The junction temperature, TJ, that we just calculated is
greater than the LT1129CT’s maximum junction temperature specification of 125°C; therefore this circuit must use
a heat sink.
Now the task at hand is to calculate the correct heat sink
to use. The selected heat sink must hold the junction
temperature at less than 125°C for the LT1129CT.
TJ = P(θTOTAL) + TA
125°C = 3W(θTOTAL) + 50°C
θTOTAL = 25°C/W and
θTOTAL = θJC + θCS + θSA
For this configuration:
θJC = 5°C/W (LT1129CT data sheet)
θCS = 0.2°C/W (typical for heat sink mounting)
θSA = heat sink specification
Plugging in these numbers:
25°C/W = 5°C/W + 0.2°C/W + θSA
θSA = 19.8°C/W
Therefore, the heat sink selected must have a thermal
resistance of less than 19.8°C/W to hold the LT1129CT
junction temperature at less than 125°C. Obviously, the
lower the heat sink thermal resistance, the lower the
LT1129CT junction temperature. A lower junction temperature will increase reliability.
Now, let’s consider a circuit using an LT1129CT operating
in a 50°C enclosure with an input voltage of only 6VDC, an
output voltage of 5VDC, and a load current of 1A.
The power dissipated by the LT1129CT is:
Does this circuit need a heat sink? Again, for the purposes
of the calculation, we must use thermal resistance from
junction to ambient, θJA= 50°C/W for the LT1129CT.
TJ = P(θJA) + TA = 1W(50°C/W) + 50°C
= 50°C + 50°C = 100°C
The junction temperature TJ that we just calculated is now
less than the LT1129CT’s maximum junction temperature
specification of 125°C. Therefore this circuit does not
need a heat sink. This illustrates the advantage of a low
dropout regulator like the LT1129CT.
An Alternative Method for
Specifying Thermal Parameters
Linear Technology Corp. has introduced an alternative
method to specify and calculate thermal parameters of
voltage regulators. Previous regulators, with a single
thermal resistance junction-to-case (θJC), used an average of temperature rise of the control and power sections.
This could easily allow excessive junction temperature
under certain conditions of ambient temperature and heat
sink thermal resistance.
Several LTC voltage regulators include thermal resistance
and maximum junction temperature specifications for
both the control and power sections, as shown in
Table 2. Two Examples Showing Thermal Resistance of Control and
Power Sections of LTC Regulators
CONTROL
POWER
θJC
TJMAX
θJC
TJMAX
LT1083MK
0.6°C/W
150°C
1.6°C/W
200°C
LT1085CT
0.7°C/W
125°C
3.0°C/W
150°C
DEVICE
As an example, let’s calculate the junction temperature for
the same application shown before, using an LT1085CT
instead of the LT1129CT. Once again, we are operating in
a 50°C enclosure; the input voltage is 8VDC, the output
voltage is 5VDC and the load current is 1A.
The power dissipated by the LT1085CT is the same as
before, 3W. We will assume we have selected a heat sink
with a thermal resistance, θSA of 10°C/W. First calculate
the control section of the LT1085CT:
P = (VIN – VOUT)(ILOAD) = (6V – 5V)1A = 1W
AN66-99
Application Note 66
θJC = 0.7°C/W (LT1085CT data sheet)
θCA = 0.2°C/W (typical)
θSA = 10°C/W
θTOTAL = θJC + θCA + θSA = 0.7°C/W + 0.2°C/W +
10°C/W = 10.9°C/W
To determine the control section junction temperature:
TJ = P(θTOTAL) + TA = 3W(10.9°C/W) + 50°C
= 82.7°C (TJ MAX = 125°C)
To calculate the power section of the LT1085CT:
θJC = 3°C/W (LT1085CT data sheet)
θTOTAL = θJC + θCA + θSA = 3°C/W + 0.2°C/W +
10°C/W = 13.2°C/W
To determine the power section junction temperature:
TJ = P(θTOTAL) + TA = 3W(13.2°C/W) + 50°C
= 89.6°C (TJ MAX = 150°C)
In both cases, the junction temperature is below the
maximum rating for the respective section; this ensures
reliable operation.
AN66-100
Conclusion
This article is an introduction to thermal analysis for
voltage regulators; however, the techniques also apply to
other devices, including operational amplifiers, voltage
references, resistors, and the like. For the more advanced
student of thermal analysis, it can be shown that there is
a direct analogy between electronic circuit analysis and
thermal analysis, as shown in Table 3.
Table 3. analogy Between Thermal Analysis
and Electronic Circuit Analysis
THERMAL WORLD
ELECTRICAL WORLD
Power
Current
Temperature Differences
Voltage
Thermal Resistance
Resistance
All standard electronic network analysis techniques
(Kirchhoff’s laws, Ohm’s law) and computer circuit analysis programs (SPICE) can be applied to complex thermal
systems.
Application Note 66
ALPHABETICAL INDEX (BY MAJOR CATEGORIES)
BATTERY CHARGERS
Charging NiMH/NiCd or Li-Ion with the LT1510 ............................................................................................... 70
Lithium-Ion Battery Charger ............................................................................................................................. 71
Simple Battery Charger Runs at 1MHz ............................................................................................................. 73
A Perfectly Temperature Compensated Battery Charger ................................................................................... 74
A Simple 300mA NiCd Battery Charger ............................................................................................................ 75
High Efficiency (>90%) NiCd Battery Charger Circuit Programmable for 1.3A Fast Charge
or 100mA Trickle Charge.................................................................................................................................. 76
MISCELLANEOUS
Protected Bias for GaAs Power Amplifiers ....................................................................................................... 88
LT1158 H-Bridge Uses Ground Referenced Current Sensing for System Protection........................................ 89
LT1158 Allows Easy 10A Locked Antiphase Motor Control .............................................................................. 91
All Surface Mount Programmable 0V, 3.3V, 5V and 12V VPP Generator for PCMCIA ...................................... 92
A Tachless Motor Speed Controller .................................................................................................................. 93
LT1161...And Back and Stop and Forward and Rest—All with No Worries at All ............................................ 95
Simple Thermal Analysis—A Real Cool Subject for LTC Regulators ............................................................... 98
POWER MANAGEMENT
LT1366 Rail-to-Rail Amplifier Controls Topside Current .................................................................................. 78
An Isolated High Side Driver ............................................................................................................................ 79
LTC1163: 2-Cell Power Management ............................................................................................................... 80
LTC1157 Switch for 3.3V PC Card Power ........................................................................................................ 81
The LTC1157 Dual 3.3V Micropower MOSFET Driver ...................................................................................... 82
The LTC1155 Does Laptop Computer Power Bus Switching, SCSI Termination Power or
5V/3A Extremely Low Dropout Regulator ......................................................................................................... 82
A Circuit That Smoothly Switches Between 3.3V and 5V.................................................................................. 84
A Fully Isolated Quad 4A High Side Switch ...................................................................................................... 85
The LTC1153 Electronic Circuit Breaker ........................................................................................................... 86
LTC1477: 0.07Ω Protected High Side Switch Eliminates “Hot Swap” Glitching ............................................... 87
REGULATORS—LINEAR
Low Noise Wireless Communications Power Supply ....................................................................................... 65
An LT1123 Ultralow Dropout 5V Regulator ...................................................................................................... 66
REGULATORS—LINEAR
Microprocessor Power
LT1580 Low Dropout Regulator Uses New Approach to Achieve High Performance ....................................... 67
LT1585: New Linear Regulator Solves Load Transients ................................................................................... 68
REGULATORS—SWITCHING (BOOST)
Medium Power (1A to 4A)
High Output Current Boost Regulator............................................................................................................... 24
Low Power (<1A)
Applications for the LT1372 500kHz Switching Regulator ............................................................................... 25
AN66-101
Application Note 66
REGULATORS—SWITCHING (BUCK)
High Power (>4A)
Big Power for Big Processors: The LTC1430 Synchronous Regulator ............................................................... 4
Applications for the LTC1266 Switching Regulator ............................................................................................ 5
A High Efficiency 5V to 3.3V/5A Converter ......................................................................................................... 7
High Current, Synchronous Step-Down Switching Regulator ............................................................................ 8
Medium Power (1A to 4A)
1MHz Step-Down Converter Ends 455kHz IF Woes ......................................................................................... 10
High Output Voltage Buck Regulator ................................................................................................................ 11
The LTC1267 Dual Switching Regulator Controller Operates from High Input Voltages................................... 12
High Efficiency 5V to 3.3V/1.25A Converter in 0.6 Square Inches .................................................................... 13
LT1074/LT1076 Adjustable 0V to 5V Power Supply ........................................................................................ 14
Triple Output 3.3V, 5V and 12V High Efficiency Notebook Power Supply ........................................................ 15
The New SO-8 LTC1147 Switching Regulator Controller Offers High Efficiency in a Small Footprint ............... 17
The LT1432: 5V Regulator Achieves 90% Efficiency ........................................................................................ 20
Low Power (<1A)
Applications for the LTC1265 High Efficiency Monolithic Buck Converter ........................................................ 22
REGULATORS—SWITCHING (BUCK/BOOST)
±5V Converter Uses Off-the-Shelf Surface Mount Coil..................................................................................... 27
Switching Regulator Provides Constant 5V Output from 3.5V to 40V Input Without a Transformer ................ 28
Switching Regulator Provides ±15V Output from an 8V to 40V Input Without a Transformer ......................... 29
REGULATORS—SWITCHING (DISCUSSION)
Adding Features to the Boost Topology............................................................................................................ 39
Sensing Negative Outputs ................................................................................................................................ 40
REGULATORS—SWITCHING (FLYBACK)
Applications for the LT1372 500kHz Switching Regulator ............................................................................... 25
REGULATORS—SWITCHING (INVERTING)
High Efficiency 12V to – 12V Converter ............................................................................................................ 32
Regulated Charge Pump Power Supply ............................................................................................................ 34
Applications for the LTC1265 High Efficiency Monolithic Buck Converter ........................................................ 22
LTC1174: A High Efficiency Buck Converter ..................................................................................................... 35
REGULATORS—SWITCHING (MICROPOWER)
Backlight
High Efficiency EL Driver Circuit....................................................................................................................... 58
A Low Power, Low Voltage CCFL Power Supply .............................................................................................. 60
All Surface Mount EL Panel Driver Operates from 1.8V to 8V Input ................................................................. 61
A Dual Output LCD Bias Voltage Generator ...................................................................................................... 62
LCD Bias Supply............................................................................................................................................... 63
AN66-102
Application Note 66
REGULATORS—SWITCHING (MICROPOWER)
Switched Capacitor
Regulated Charge Pump Power Supply ............................................................................................................ 34
REGULATORS—SWITCHING (MICROPOWER)
VPP Generator
LTC1262 Generates 12V for Programming Flash Memories Without Inductors ............................................... 64
Flash Memory VPP Generator Shuts Down with 0V Output ............................................................................. 64
REGULATORS—SWITCHING (POWER FACTOR CORRECTED)
The New LT1508/LT1509 Combines Power Factor Correction and a PWM in a Single Package ...................... 37
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
AN66-103
Application Note 66
U.S. Area Sales Offices
NORTHEAST REGION
Linear Technology Corporation
3220 Tillman Drive
Suite 120
Bensalem, PA 19020
Phone: (215) 638-9667
FAX: (215) 638-9764
SOUTHEAST REGION
Linear Technology Corporation
17000 Dallas Parkway
Suite 219
Dallas, TX 75248
Phone: (214) 733-3071
FAX: (214) 380-5138
SOUTHWEST REGION
Linear Technology Corporation
21243 Ventura Blvd.
Suite 227
Woodland Hills, CA 91364
Phone: (818) 703-0835
FAX: (818) 703-0517
Linear Technology Corporation
266 Lowell Street
Suite B-8
Wilmington, MA 01887
Phone: (508) 658-3881
FAX: (508) 658-2701
Linear Technology Corporation
5510 Six Forks Road
Suite 102
Raleigh, NC 27609
Phone: (919) 870-5106
FAX: (919) 870-8831
Linear Technology Corporation
15375 Barranca Parkway
Suite A-211
Irvine, CA 92718
Phone: (714) 453-4650
FAX: (714) 453-4765
NORTHWEST REGION
Linear Technology Corporation
1900 McCarthy Blvd.
Suite 205
Milpitas, CA 95035
Phone: (408) 428-2050
FAX: (408) 432-6331
CENTRAL REGION
Linear Technology Corporation
Chesapeake Square
229 Mitchell Court, Suite A-25
Addison, IL 60101
Phone: (708) 620-6910
FAX: (708) 620-6977
International Sales Offices
FRANCE
Linear Technology S.A.R.L.
Immeuble "Le Quartz"
58 Chemin de la Justice
92290 Chatenay Malabry
France
Phone: 33-1-41079555
FAX: 33-1-46314613
KOREA
Linear Technology Korea Co., Ltd
Namsong Building, #403
Itaewon-Dong 260-199
Yongsan-Ku, Seoul 140-200
Korea
Phone: 82-2-792-1617
FAX: 82-2-792-1619
GERMANY
Linear Technology GmbH
Oskar-Messter-Str. 24
85737 Ismaning
Germany
Phone: 49-89-962455-0
FAX: 49-89-963147
SINGAPORE
Linear Technology Pte. Ltd.
507 Yishun Industrial Park A
Singapore 2776
Phone: 65-753-2692
FAX: 65-754-4113
JAPAN
Linear Technology KK
5F NAO Bldg.
1-14 Shin-Ogawa-cho Shinjuku-ku
Tokyo, 162 Japan
Phone: 81-3-3267-7891
FAX: 81-3-3267-8510
TAIWAN
Linear Technology Corporation
Rm. 602, No. 46, Sec. 2
Chung Shan N. Rd.
Taipei, Taiwan, R.O.C.
Phone: 886-2-521-7575
FAX: 886-2-562-2285
UNITED KINGDOM
Linear Technology (UK) Ltd.
The Coliseum, Riverside Way
Camberley, Surrey GU15 3YL
United Kingdom
Phone: 44-1276-677676
FAX: 44-1276-64851
SWEDEN
Linear Technology AB
Sollentunavägen 63
S-191 40 Sollentuna
Sweden
Phone: 46-8-623-1600
FAX: 46-8-623-1650
World Headquarters
Linear Technology Corporation
1630 McCarthy Blvd.
Milpitas, CA 95035-7417
Phone: (408) 432-1900
FAX: (408) 434-0507
Linear Technology Corporation
McCarthy Blvd., Milpitas, CA 95035-7417
AN66-104 1630
(408) 432-1900
: (408) 434-0507
: 499-3977
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LT/GP 0896 5K • PRINTED IN USA
 LINEAR TECHNOLOGY CORPORATION 1996