Power Conditioning for Notebook and Palmtop Systems

Application Note 51
August 1992
Power Conditioning for Notebook and Palmtop Systems
Robert Dobkin
Carl Nelson
Dennis O’Neill
Steve Pietkiewicz
Tim Skovmand
Milt Wilcox
INTRODUCTION
Battery systems include NiCad, nickel-hydride, lead acid,
and rechargeable lithium, as well as throw-away alkaline
batteries. The ability to power condition a wide range
of batteries makes the ultimate product much more
attractive because power sources can be interchanged,
increasing overall system versatility.
a very high efficiency (Figure 1) 5V or 3.3V step-down
(buck) switching regulator. These regulators feature a
low-loss saturating NPN switch that is normally configured with the negative terminal (emitter) at ground. The
LT1432 allows the switch to be floated as required in a
step-down converter, yet still provides full switch saturation for highest efficiency.
Burst ModeTM is a trademark of Linear Technology Coporation.
100
NORMAL MODE
(USE AMPS SCALE)
90
EFFICIENCY (%)
Notebook and palmtop systems need a multiplicity of
regulated voltages developed from a single battery. Small
size, light weight, and high efficiency are mandatory for
competitive solutions in this area. Small increases in
efficiency extend battery life, making the final product
much more usable with no increase in weight. Additionally, high efficiency minimizes the heat sinks needed on
the power regulating components, further reducing system weight and size.
VOUT = 5V
80
BURST MODETM OPERATION
(USE mA SCALE)
70
A main rechargeable battery may be any of the four
secondary type cells, with a back-up or emergency ability
to operate off alkaline batteries. The higher energy density
available in non-rechargeable alkaline batteries allows the
systems to operate for extended time without battery
replacement.
LT1432 Driver for High Efficiency 5V and 3.3V Buck
Regulator
The LT1432 is a control chip designed to operate with the
LT1170 or LT1270 family of switching regulators to make
0
0
1A
20mA
3A
60mA
2A
40mA
LTAN51 • 01
100
90
EFFICIENCY (%)
The systems shown here provide power conditioning with
high efficiency and low parts count. Trade-offs between
complexity and efficiency have been made to maximize
manufacturability and minimize cost. All the supplies
operate over a wide range of input voltage allowing great
flexibility in the choice of battery configuration.
LT1271, L = 50µH
60
LT1271, L = 50µH
ILOAD = 1A
80
70
60
0
5
10
15
20
INPUT VOLTAGE (V)
25
30
LTAN51 • 02
Figure 1. LT1432 5V Efficiency
AN51-1
Application Note 51
Many other features have been incorporated into the
LT1432 to enhance operation in battery powered applications. An accurate current limit uses only 60mV sense
voltage, allows for foldback, and uses “free” PC board
trace material for the sense resistor. Logic controlled
shutdown mode draws only 15µA battery current to allow
for extremely long shutdown periods. The switching IC is
powered from the regulator output to enhance efficiency
and to allow input voltages as low as 6.5V.
The LT1432 has optional Burst ModeTM operation to
achieve high efficiency at very light load currents (0mA to
100mA). In normal switching mode, the standby power
loss is about 60mW, limiting efficiency at light loads. In
burst mode, standby loss is reduced to approximately
15mW. Output ripple is 150mVP-P in this mode, but this is
normally well within the requirements for digital logic
supplies. Burst ModeTM operation would typically be used
for “sleep” conditions where IC memory chips remain
powered for data retention, but the remainder of the
system is powered down. Load current in this mode is
typically in the 5mA-100mA range. The operating mode is
under logic control.
The LT1432 is available in 8-pin surface mount and DIP
packages. The LT1170 and LT1270 families are available
in a surface mount version of the 5-pin TO-220 package.
Circuit Description
The circuit shown in Figure 2 is a basic 5V positive buck
converter which can operate with input voltages from 6.5V
to 25V. The power switch is located between the VSW pin
and GND pin on the LT1271. Its current and duty cycle are
controlled by the voltage on the VC pin with respect to the
GND pin. This voltage ranges from 1V to 2V as switch
currents increase from zero to full scale. Correct output
voltage is maintained by the LT1432 which has an internal
reference and error amplifier. The amplifier output is level
shifted with an internal open collector NPN to drive the VC
pin of the switcher. The normal resistor divider feedback
to the switcher feedback pin cannot be used because the
feedback pin is referenced to the GND pin, which is
switching many volts. The feedback pin (FB) is simply
bypassed with a capacitor. This forces the switcher VC pin
AN51-2
to swing high with about 200µA sourcing capability. The
LT1432 VC pin then sinks this current to control the loop.
C4 forms the dominant loop pole with a loop zero added by
R1. C5 forms a higher frequency loop pole to control
switching ripple at the VC pin.
A floating 5V power supply for the switcher is generated by
D2 and C3 which peak detect the output voltage during
switch “off” time. This is a very efficient way of powering
the switcher because power drain does not increase with
regulator input voltage. However, the circuit is not selfstarting, so some means must be used to start the
regulator. This is performed by an internal current path in
the LT1432 which allows current to flow from the input
supply to the V+ pin during start-up.
In both the 5V and 3.3V regulators, D1, L1, and C2 act as
the conventional catch diode and output filter of the buck
converter. These components should be selected carefully
to maintain high efficiency and acceptable output ripple.
Current limiting is performed by R2. Sense voltage is only
60mV to maintain high efficiency. This also reduces the
value of the sense resistor enough to utilize a printed
circuit board trace as the sense resistor. The sense voltage
has a positive temperature coefficient to match the temperature coefficient of copper.
The basic regulator has three different operating modes,
defined by the mode pin drive. Normal operation occurs
when the mode pin is grounded. A low quiescent current
Burst ModeTM operation can be initiated by floating
the mode pin. Input supply current is typically 1.3mA
in this mode, and output ripple voltage is 100mVP-P.
Pulling the mode pin above 2.5V forces the entire regulator
into micropower shutdown where it typically draws less
than 20µA.
What are the benefits of using an active (synchronous)
switch to replace the catch diode? This is the trendy thing
to do, but calculations and actual breadboards show that
the improvement in efficiency is only a few percent at best.
This can be shown with the following simplified formulas:
Diode loss = Vf (VIN – VOUT)(IOUT)/ VIN
FET switch loss = (VIN – VOUT)(RSW)(IOUT)2/ VIN
Application Note 51
The change in efficiency is:
If Vf (diode forward voltage) = 0.45V, VIN = 10V, VOUT = 5V,
RFET = 0.1Ω, IOUT = 1A, and efficiency = 90%, the improvement in efficiency is only:
(Diode loss – FET loss)(Efficiency)2/(VOUT)(IOUT)
This is equal to:
(10V – 5V)(0.45V – 0.1Ω × 1A)(0.9)2/(10V)(5V) = 2.8%
(VIN – VOUT)(Vf – RFET ×
IOUT)(E)2/(VIN)(VOUT)
VIN
6V TO 25V
C1
200µF
35V
10µH
3A
VIN
VSW
LT1271/LT1269
+
FB
VC
GND
D2
1N4148
0.02µF
R1
680Ω
C5
0.03µF
C3
4.7µF
TANT
C4
0.1µF
+
L1
50µH
R2
0.015Ω
VOUT
5V, 3A
C2
390µF
16V
V+
DIODE
VIN
<0.3V = NORMAL RUN MODE
>2.5V = SHUTDOWN
OPEN = BURST MODETM
× ×
+
D1
MBR330p
VC
OPTIONAL
OUTPUT
FILTER
+
100µF
16V
VLIM
LT1432
VOUT
MODE
GND
220pF
LTAN51 • 03
Figure 2. High Efficiency 5V Regulator with Manual Burst ModeTM Operation
1N4148*
VIN
6V TO 25V
C1
200µF
35V
10µH
3A
VIN
VSW
LT1271/LT1269
+
FB
VC
+
GND
1N4148*
0.02µF
680Ω
0.03µF
4.7µF
TANT
L1
25µH**
DIODE
VIN
0.015Ω
+
MBR330p
VC
OPTIONAL
OUTPUT
FILTER
+
0.1µF
< 0.3V = NORMAL RUN MODE
>2.5V = SHUTDOWN
OPEN = BURST MODETM
100µF
16V
× ×
VOUT
3.3V, 3A
390µF
16V
V+
VLIM
LT1432-3.3
VOUT
MODE
220pF
GND
* CONNECT D1 ANODE TO OUTPUT FOR MINIMUM INPUT
VOLTAGE GREATER THAN 9V, AND TO INPUT IF MINIMUM
INPUT CAN FALL BELOW 9V. OUTPUT CONNECTION IS MORE
EFFICIENT AT LIGHT LOADS.
** USE MOLYPERMALLOY, KOOLMU OR FERRITE CORE.
LTAN51 • 04
Figure 3. High Efficiency 3.3V Regulator with Manual Burst ModeTM Operation
AN51-3
Application Note 51
This does not take FET gate drive losses into account,
which can easily reduce this figure to less than 2%. The
added cost, size, and complexity of a synchronous switch
configuration would be warranted only in the most extreme circumstances.
nous switching operation up to input voltages of 48V (60V
abs max) with a slight penalty in quiescent current.
The rated current level for all device types is set by the
external sense resistor according to the formula
IOUT = 100mV/RSENSE. The maximum peak inductor
current and Burst ModeTM current are also linked to
RSENSE. The peak current is limited to 150mV/RSENSE,
while Burst ModeTM operation automatically begins when
the output current drops below approximately 15mV/
RSENSE. In this mode, the external MOSFET(s) are held off
to reduce switching losses and the controller sleeps at
200µA supply current (600µA for the LTC1149), while the
output capacitor supports the load. When the output
capacitor discharges 50mV, the controller briefly turns
back on, or “bursts,” to recharge the capacitor. Complete
shutdown reduces the supply current to only 10µA (150µA
for the LTC1149).
Burst ModeTM efficiency is limited by quiescent current
drain in the LT1432 and the switching IC. The typical Burst
ModeTM zero-load input power is 17mW. This gives about
one month battery life for a 12V, 1.2AHr battery pack.
Increasing load power reduces discharge time proportionately. Full shutdown current is only about 15µA,
which is considerably less than the self-discharge rate of
typical batteries.
BiCMOS Switching Regulator Family Provides Highest
Step-Down Efficiencies
The LTC1148 family of single and dual step-down switching regulator controllers features automatic Burst ModeTM
operation to maintain high efficiencies at low output
currents. All members of the family use a constant offtime,
current mode architecture. This results in excellent line
and load transient response, constant inductor ripple
current, and well controlled start-up and short circuit
currents. The LTC1147/LTC1143 drive a single external
P-channel MOSFET, while the LTC1148/LTC1142 and
LTC1149 drive synchronous external power MOSFETs at
switching frequencies up to 250kHz.
The first application shown in Figure 4 converts 5V to 3.3V
at 1.5A output current. By choosing the LTC1147-3.3, a
minimum board space solution is achieved at a slight
penalty in peak efficiency (the LTC1148-3.3 driving
synchronous MOSFETs in this application would add
approximately 2.5% to the high current efficiency). Figure
5 shows how Burst ModeTM operation maintains high
operating efficiencies at low output currents.
VIN
5V
Table 1 gives an overview of the family with applicability to
common notebook DC to DC converter requirements. The
LTC1147 is available in an 8-pin SOIC and drives only a
single power MOSFET, giving it the smallest PC board
footprint at a slight penalty in efficiency. The LTC1148HV/
LTC1142HV offer synchronous switching capability at
input voltages from 4V to 18V (20V abs max) with a low
200µA quiescent current. The LTC1149 extends synchro-
0.1µF
0V = NORMAL
>1.5V = SHUTDOWN
6
3
3300pF
2
Si9433DY
8
1
VIN
SHUTDOWN
PDRIVE
LTC1147-3.3
5
SENSE +
ITH
+
0.01µF
SENSE –
CT
1k
120pF
47µF
16V
L
10µH
RS
0.068Ω
4
GND
7
+
MBRS130L
LTC1148
LTAN51 • 25
Figure 4. High Efficiency Surface Mount 5V to 3.3V Converter
Delivers 1.5A in Minimum Board Area
LTC1143
LTC1142
LTC1148HV
LTC1142HV
Continuous VIN < 48V
Continuous VIN < 18V
Continuous VIN < 13.5V
X
X
X
X
Low Dropout 5V
X
X
X
X
X
X
Adjustable
AN51-4
LTC1149
X
Dual 5V and 3.3V
X
220µF
10V
L = SUMIDA CDR74-100LC
RS = IRC LRC-LR2010-01-R068-F
Table 1. LTC1148 Family Applications
LTC1147
VOUT
3.3V/
1.5A
X
X
X
X
X
X
X
X
Application Note 51
In the second application (Figure 6), an LTC1148HV-5 is
used as the controller for a 10W high efficiency regulator.
This circuit can be used with as few as 5 NiCad or NiMH
cells thanks to its excellent low dropout performance. Like
other members of the family, the LTC1148HV goes to
100% duty cycle (P-channel MOSFET turned on DC) in
dropout. The input to output voltage differential required
to maintain regulation then simply becomes the product of
the load current and total resistance of the MOSFET,
inductor, and current sense resistor. In the Figure 6 circuit,
this total resistance is less than 0.2Ω. For operation at low
input voltages, logic-level MOSFETs must be used.
VIN
8V TO 30V
+
2
VIN
16
PGATE
CAP
1
IRFR9024
0.068µF
+
3
3.3µF
L
62µH
0.047µF
VCC
LTC1149-5
5
10
0V = NORMAL
>2V = SHUTDOWN
100µF
50V
1N4148
VCC
PDRIVE
7
1N5819
SHUTDOWN 1
SENSE
15
4
100Ω
+ 9
1000pF
SHUTDOWN 2
ITH
8
SENSE –
CT
NGATE
RS
0.04Ω
VOUT
5V/
2.5A
100Ω
3300pF
6
1k
IRFR024
13
SGND PGND RGND
680pF
11
100
LTC1147-3.3
VIN = 5V
12
+
14
L = COILTRONICS CTX62-2-MP
RS = IRC LR2512-01-R040J
150µF,
OS-CON
LTAN51 • 28
EFFICIENCY (%)
90
Figure 7. High Efficiency 5V/2.5A Regulator Operates from
AC Wall Adapters as High as 30V
80
AUTOMATIC
BURST MODETM
OPERATION
100
LTC1149-5
VIN = 12V
70
EFFICIENCY (%)
90
60
1
10
100
LOAD CURRENT (mA)
1000
LTAN51 • 26
Figure 5. High Operating Efficiency for Figure 4 Circuit Spans
Three Decades of Output Current
80
AUTOMATIC
BURST MODETM
OPERATION
70
60
20
VIN
5.2V TO 18V
+
0V = NORMAL
>1.5V = SHUTDOWN
+
3
1µF
VIN
10
PDRIVE
SHUTDOWN
LTC1148HV-5
SENSE +
6
3300pF
510Ω
4
Si9430DY
1
ITH
SENSE –
CT
NDRIVE
SGND
11
220pF
PGND
12
LTAN51 • 29
Figure 8. Operating Efficiency for LTC1149-5 High Efficiency
Converter
RS
0.05Ω
VOUT
5V/2A
7
+
440µF
10V
MBRS140T3
L = COILTRONICS CTX25-4
RS = KRL BANTRY SL-1-C1-0R050J
2000
44µF
35V
L
25µH
8 1000pF
14
Si9410DY
200
LOAD CURRENT (mA)
LTAN51 • 27
Figure 6. High Efficiency Low Dropout 5V Switching Regulator
Needs Only 200mV Headroom at 1A Output
While the 18V input voltage rating of the LTC1148HV and
LTC1142HV can generally accommodate most battery
packs, the AC wall adapters used in conjunction with
notebook systems often dictate significantly higher input
voltages. This is the primary home for the LTC1149,
shown in the Figure 7 application. This 2.5A regulator can
operate at input voltages from 8V (limited by the standard
MOSFET threshold voltages) to 30V, while still providing
excellent efficiency as shown in Figure 8. The synchronous switch plays an increasing role at high input voltages
due to the low duty cycle of the main switch.
AN51-5
Application Note 51
Board layout of the Figures 4, 6, and 7 circuits is critical for
proper transition between Burst ModeTM operation and
continuous operation. The timing capacitor pin and inductor current are the two most important waveforms to
monitor while checking an LTC1148 family regulator. The
timing capacitor pin only goes to 0V during sleep intervals,
which should only happen when the load current is less
than approximately 20% of the rated output current.
Consult the appropriate data sheet for information on
proper component location and ground routing.
and Zetex ZBD949 are specified for this application. The
dropout voltage of this regulator depends on the PNP
transistor saturation and can be in the range of 0.25V at 3A
output current — lower at lower current. The simplicity of
this system is attractive for notebook applications and
efficiency is good since little power is lost across the linear
regulator at low input voltages.
Surface Mount Capacitors for Switching Regulator
Applications
For input voltages of 5.2V* and above, the output is
regulated at 5V. As the battery voltage decreases below
5.2V*, the transistor saturates and the output voltage
follows the input voltage down with the saturation voltage
of the transistor subtracted from the input voltage.
A good rule of thumb for the output capacitor selection in
all LTC1148 family circuits is that it must have an ESR less
than or equal to the sense resistor value (for example,
0.05Ω for the Figure 6 circuit). In surface mount applications multiple capacitors may have to be paralleled to meet
the capacitance, ESR, or RMS current handling requirements of the application. Aluminum electrolytic and dry
tantalum capacitors are both available in surface mount
configurations.
The LT1123 low dropout driver can supply up to 125mA of
base current to the pass transistor. At dropout this current
is supplied continuously into the base of the pass transistor as the transistor remains in saturation. If lower drive
current is desired an optional resistor (R2) can be inserted
in series with the base of the transistor to minimize the
drive current and decrease the power dissipation in the IC.
An N-channel FET can be inserted in series with the drive
lead of the LT1123 to electrically shutdown the system.
High Efficiency Linear Supplies
The switching supplies operate over a wide input range
while maintaining high efficiency. Alternative notebook
systems have been developed for narrow supply operation
using for example, four NiCad batteries and a linear
regulator to provide the 5V output. At full charge, four
NiCad batteries can be as high as 6V and are allowed to
discharge down to 4.5V while directly powering the system. A high efficiency low dropout linear regulator suited
for this technique is shown in Figure 9.
This is a complete IC in a very low cost TO-92 3-pin
package driving a low saturation PNP transistor. Many
power PNP transistors can be used. The Motorola MJE1123
AN51-6
* Actual voltage depends on load current.
0.5
0.4
DROPOUT VOLTAGE (V)
In the case of tantalum, it is critical that the capacitors are
surge tested for use in switching power supplies. An
excellent choice is the AVX TPS series of surface mount
tantalum capacitors, available in case heights ranging
from 2mm to 4mm. For example, if 440µF/10V is called
for in an application, (2) AVX 220µF/10V (P/N
TPSE227K010) could be used. Consult the manufacturer
for other specific recommendations.
0.3
0.2
0.1
0
0
1
3
4
2
OUTPUT CURRENT (A)
5
LTAN51 • 05
5V
600Ω
R2
10µF
DRIVE
BATTERY
LT1123
FB
GND
LTAN51 • 06
Figure 9. LT1123 Dropout Voltage
Application Note 51
5V
3.3V
CDLY
0.1µF
100mV
RSEN =
ILIM
+
10µF
VS
G1
VS
DS
DS
G
G
IN
µP OR
CONTROL
LOGIC
RDLY
100k
LTC1155
DUAL
8-PIN SO
IN1
IRLR024
3.3V
LOAD
LTC1157
100k
1k
IRLR024
G2
IN2
GND
NFET
3.3V
LOAD
LARGE
SUPPLY
CAPACITOR
+
0.1µF
GND
LOAD
LOGIC
LTAN51 • 30
Figure 11. LTC1157 Dual 3.3V MOSFET Driver
LTAN51 • 07
Figure 10. LTC1155 Dual Micropower N-Channel
MOSFET Driver
250
VIN 5.5V TO 20V
5V, 150mA
VIN
+
LT1121
SHUTDOWN
VOUT
GND
+
COUT
1µF
TANT
INPUT CURRENT (µA)
200
ILOAD = 100µA
150
100
50
ILOAD = 0
SHUTDOWN
0
0
1
2
3
4
5
6
7
8
9
10
INPUT VOLTAGE (V)
LTAN51 • 08
Figure 12. LT1121 Micropower Low Dropout Regulator
Power Switching with Dual High Side Micropower
N-Channel MOSFET Drivers
The LT1155 dual high side N-channel FET gate driver
allows using low cost N-channel FETs for high side switching applications. No external components are needed
since an internal charge pump boosts the gate above the
positive rail, fully enhancing an N-channel MOSFET. Micropower operation, with 8µA standby current and 85µA
operating current, allows use in virtually all battery powered systems even for main power switching.
Included on the chip is over-current sensing to provide
automatic shutdown in case of short circuits. A time delay
can be added in series with the current sense to prevent
false triggering on high in-rush loads such as capacitors
or lamps.
The LTC1155 operates off a 4.5V to 18V supply input and
safely drives the gates of virtually all FETs. It is particularly
LTAN51 • 09
Figure 13. LT1121 Input Current
well-suited for portable applications where micropower
operation is critical. The device is available in 8-pin SO and
DIP packages.
The LTC1157 is a dual driver for 3.3V supplies. The
LTC1157 internal charge pump boosts the gate drive
voltage 5.4V above the positive rail (8.7V above ground),
fully enhancing a logic-level N-channel MOSFET for 3.3V
high side switching applications. The charge pump is
completely on-chip and therefore requires no external
components to generate the higher gate voltage. The
charge pump has been designed to be very efficient,
requiring only 3µA in the standby mode and 80µA while
delivering 8.7V to the power MOSFET gate.
Figure 11 demonstrates how two surface mount MOSFETs
and the LTC1157 can be used to switch two 3.3V loads.
The gate rise and fall time is typically in the tens of
microseconds, but can slowed by adding two resistors
and a capacitor as shown on the second channel. Slower
AN51-7
Application Note 51
rise and fall times are sometimes required to reduce the
start-up current demands of large supply capacitors.
with fixed output voltages of 3.3V and 5V and in an
adjustable version with an output range of 3.75V to 5V.
The device is available in a 5-pin DD package.
LT1121 Micropower 150mA Regulator with Shutdown
The LT1121 is a low dropout regulator designed for
applications where quiescent current must be very low
when output current is low. It draws only 30µA input
current at zero load current. Ground pin current increases
with load current, but the ratio is about 1:25, so the
efficiency of the regulator is only about 4% below theoretical maximum for a linear regulator. More importantly, the
ground pin current does not increase significantly when
the input voltage falls below the minimum required to
maintain a regulated output.
These characteristics allow the LT1121 to be used in
situations where it is desirable to have the output track the
input when the input falls below its normal range. Previous
regulators drew such high input current in this condition
that micropower operation was not possible.
Cold Cathode Fluorescent Display Driver
New backlight systems seem universally to use cold
cathode fluorescent tubes. Electroluminescent backlights
have limited light output and limited life for notebook
systems, and have limited usage among notebook and
notebook manufacturers. The cold cathode fluorescent,
on the other hand, has high efficiency, long life, and high
light output. Typically the cold cathode fluorescent wants
to be driven with 1mA to 5mA at 30kHz to 50kHz. The
driving voltage and current are a function of the manufacturer and tube geometry.
C2
15pF
3kV
5
1
The LT1129, a 700mA version of the LT1121 is also
available. The LT1129 includes all of the protection features of the LT1121. No load quiescent current is slightly
higher at 50µA and the LT1129 requires a minimum of
3.3µF of output capacitance. The LT1129 is also available
AN51-8
2
D1
1N4148
3
4
+
+VIN
10µF
D2
1N4148
C1
0.033µF
Q1
MPS650
1k
562Ω
L1
300µH
5
6
VIN
R1
VSW
E1
7
LT1172
8
Q2
MPS650
1N5818
+VIN
4.5V TO 20V
E2
VFB
GND
1
The LT1121 is available with a fixed output voltage of 3.3V
or 5V and as an adjustable device with an output voltage
range of 3.75V to 30V. Fixed voltage devices are available
in 3-pin SOT-223, and 8-pin SO packages. Adjustable
devices are available in an 8-pin SO package.
7
L2
Extra effort was taken to make the LT1121 stable with
small output capacitors that have high ESR. A 1µF
tantalum output capacitor is suggested, as compared to
10µF for previous designs. Larger output capacitors can
be used without fear of instabilities.
The LT1121 is ideal for the backup and/or suspend mode
power supply in notebook computers. A shutdown pin
allows the regulator to be fully turned off, with input
current dropping to only 16µA. Careful design of the IC
circuitry connected to the input and output pins allows the
output to be held high while the input is pulled to ground
or reversed, without current flowing from the output back
to the input. The input pin can be reversed up to 20V.
LAMP
9
VC
+
R3
10k
50k
INTENSITY
ADJUST
3
2
+
2µF
C1 = MUST BE A LOW LOSS CAPACITOR.
METALIZED POLYCARB
WIMA FKP2 (GERMAN) RECOMMENDED.
L1 = SUMIDA-6345-020 OR COILTRONICS-CTX110092-1.
PIN NUMBERS SHOWN FOR COILTRONICS UNIT
L2 = COILTRONICS-CTX300-4
Q1, Q2 = AS SHOWN OR BCP 56 (PHILIPS SO PACKAGE)
DO NOT SUBSTITUTE COMPONENTS
SUMIDA (708) 956-0666
COILTRONICS (305) 781-8900
Figure 14. CCFL Inverter
C6
1µF
LTAN51 • 10
Application Note 51
Optimally the current through the tube should be regulated
to control its brightness.
To understand the operation of the cold cathode fluorescent display driver in Figure 14, the circuit should be
looked at as two sections; 1. The regulating loop, 2. The
high voltage oscillator/driver.
The regulating loop consists of an LT1172 switching
regulator in a buck mode configuration driving constant
current into a self-oscillating converter coupled to a high
voltage transformer. The architecture of the driver allows
a wide input range of battery voltage while maintaining
fluorescent tube current constant. In negative buck mode,
the LT1172 periodically connects inductor L1 to ground
via the switch pin. This creates a flow of current in L1
which is steered by self-oscillating transistors Q1 and Q2
to the primary of transformer L2. The output of L2 is a high
voltage AC waveform that is partially ballasted by the 15pF
capacitor. To achieve the desired regulation of actual bulb
current, D1 and D2 rectify bulb current and pass one phase
through R1. This rectified current is converted to a voltage
by R1 and filtered by R3 and C6. The filtered signal
becomes a feedback signal to the LT1172, which maintains it at 1.25V.
Enclosing the cold cathode fluorescent bulb in a feedback
loop allows precise control of its operating current and
allows microprocessor control of its brightness. Voltage
fed through a resistor to the top of C6, either from a D/A
converter or from logic, will control the current through
the fluorescent tube, allowing brightness to be varied
from a keyboard input.
BALLAST
CAPACITORS
CCFLs
TO DRIVE
CIRCUITRY
TO D1,
D2 JUNCTION
COILTRONICS
CTX110459-1
LTAN51 • 11
Figure 15. Two Bulb Adaption for Color
This architecture of a buck converter driving a selfoscillating inverter was chosen because it allows a wide
range of input voltages. It is also tolerant of winding ratios
on the cold cathode fluorescent transformer. One caution
with this circuit is the voltage applied to the bulb terminals
is not limited if the feedback loop is broken, so care must
be taken to minimize the possibility of power being applied
to this circuit with the fluorescent tube removed. Cures for
this problem and much more detail on backlight engineering and circuits appear in LTC Application Note 55, “Techniques for 92% Efficient LCD Illumination.”
BATTERY CHARGING
Lead Acid Battery Charger
Though not as popular as NiCad, Lead acid rechargeable
gel cells are attractive because of their high energy density
per unit volume. These cells have a long life expectancy
when treated properly, but often suffer premature failure
because of improper charging. The circuit shown in Figure
16 provides a near ideal charging system for lead-acid
cells. It has precise nonlinear temperature compensation,
constant voltage charging with constant current override,
and high efficiency over a wide range of input and battery
voltages.
The basic charger is a flyback design to allow operation
with input voltages above or below battery voltage. The
LT1171 IC switcher operates at 100kHz and can deliver up
to 15W into the battery. A dual op amp is used to control
constant voltage and constant current modes. A1 activates as a current limiter when charging current through
R7 exceeds a preset limit determined by R3, R6, and R7.
This current limit is included to prevent excess charge
current for heavily discharged batteries. Losses in R7 are
kept low because the voltage drop across R7 is kept to
several hundred millivolts.
Lead acid batteries have a nonlinear negative temperature
coefficient which must be accurately compensated to
ensure long battery life and full charge capacity. R5 is a
positive temperature coefficient thermistor (tempsistor)
whose +0.7%/°C linear TC is converted to the required
nonlinear characteristic by the parallel connection with
R2. The combination of R2, R3, and R4 multiply the
1.244V feedback level of the LT1171 to the proper 2.35V
AN51-9
Application Note 51
D4
MBR350
INPUT
VOLTAGE
10V TO 25V
D1
6V
1/4W
R9 = 200k × (CELLS –1)
= 1M FOR 6 CELLS
D2
35V
1/2W
T1
100µH
1:1 BIFILAR
2.5A PEAK
1.25ARMS
D3
1N5818
+
C4
200µF
35V
0A TO 1A
100pF
R9
VIN
R8
120k
VSW
LT1171
–
R4
1.33k*
FB
1/2 A1
VC
GND
D5
1N4148
C2
0.2µF
R1
100k
R2
3k*
+
R5
3k**
+
A1 = LT1013
BATTERY
1 TO 10
CELLS
R3
18k
C3
1000pF
R10
200k*
R11
470Ω
+
+
C1
100µF
35V
C5
0.1µF
V+
1/2 A1
V–
R6
2k
ILIM =
1.8V (R6)
R7•R3
R7
0.2Ω
–
* 1%
** TEMPSISTOR, +0.7%/°C, MIDWEST COMPONENT SALES. R5 DOES
A NEAR PERFECT TEMPERATURE COMPENSATION FOR LEAD ACID.
LTAN51 • 12
Figure 16. Lead Acid Battery Charger
level required by one cell at 25°C. A2 is used as a buffer to
drive the resistor network. This allows large resistors to be
used for the cell multiplier string, R9 and R10. R9 is set at
200k for each series cell over one. R9 current is only 12µA,
so it can be left permanently connected to the battery. R1
is added to give the charger a finite output resistance
(≈ 0.025Ω/cell) in constant voltage mode to prevent low
frequency hunting.
NiCAD Charging
Battery charging is a very important section of any notebook system. The battery charging circuits shown here for
nickel cadmium or nickel metal hydride batteries control
AN51-10
the current into the battery but do not detect when full
battery charge is reached.
The first circuit, Constant Current Battery Charger (Figure
17), is built around a flyback configuration. This allows the
battery voltage to be lower or higher than the input voltage.
For example, a 16V battery stack may be charged off of a
12V automobile battery. The charge current is sensed by
R4, a 1.2Ω resistor and set at approximately 600mA.
Resistors R5 and R6 limit the peak output voltage when no
battery is connected. Diode D3 prevents the battery discharging through the divider network when the charger is
off, while transistor Q1 allows electronic shutdown of
the charger.
Application Note 51
D2
MBR350
VIN
10V TO 25V*
VR1
P6KE36A
D4
6V
+
600mA
L1
1:1
50µH
D1
1N5818
C1
100µF
35V
D3
1N5818
+
C4
200µF
35V
VIN
R5
16.2k
1%
VSW
+
LT1171
BATTERY
4V TO 20V
FB
GND
VC
SHUTDOWN
Q1
VN2222LL
Q2
2N2222
R2
470Ω
C3
0.1µF
R3
470Ω
R4
1.2Ω
R6
1.21k
1%
LTAN51 • 13
* LOWER INPUT VOLTAGES MAY BE
USED BY REDUCING D4 VOLTAGE.
Figure 17. Constant Current Battery Charger
The next two chargers are a high efficiency buck charger
configuration. The input voltage must be higher than the
battery voltage for charging to occur. These chargers are
90% efficient when charging at maximum output current.
No heat sinks are needed on either the switching regulator
or diodes because the efficiency is so high.
The dual rate battery charger in Figure 18 uses a logic
signal to toggle between a high charge rate, up to 2A, or a
trickle rate for keep alive. An LT1006 amplifier senses the
current into the battery and drives the feedback pin of an
LT1171 switching regulator. The entire control circuit is
bootstrapped to the LT1171 and floats at the switching
frequency, so stray capacitance must me minimized.
A gain setting transistor changes the gain on the LT1006
by shorting or opening resistor R1. This changes the
charge rate, for the value shown, between 0.1A and 1A.
The charger in Figure 19 is programmable with a voltage
from D-A converters. The charging current is directly
proportional to the program voltage. A small sense resistor in the bottom side of the battery senses the battery
charging current. This is compared with the program
voltage and a feedback signal is developed to drive the
LT1171 VC pin. This controls the charging current from the
LT1171 and with appropriate control circuits any battery
current may be programmed. Efficiency during high charge
currents is 90%.
AN51-11
Application Note 51
DC INPUT
VOLTAGE
MIN = VBATT + 3V
MAX = 30V
MBR330
TO RAW MAIN SUPPLY
+
100µF
35V
1N5819
VIN
100k
VSW
HI RATE =
R2
12k
VN2222
S
2.2µF
LT1171
D
R6
6.6k
TRICKLE =
R1
110k
VC
GND
1.24V (R3 + R5)
= 100mA
(R4) (R1 + R2)
–
LT1006
FB
1.24V (R3 + R5)
= 1A
(R4) (R2)
R3
900Ω
+
0.047µF
0.01µF
0.33µF
0.01µF
R5
100Ω
30µH
1A
R7
110k
R4, 0.1Ω
KELVIN CONTACTS
220µF
25V
VN2222
1N5819
HI RATE = 0V
TRICKLE = 5V
D
TO SYSTEM
POWER
SWITCH
+
BATTERY
3V TO 20V
S
LTAN51 • 14
Figure 18. High Efficiency Dual Rate Battery Charger (Up to 2A)
LCD Display Contrast Power Supply
A 4-Cell NiCad Regulator/Charger
LCD display typically requires between –18V and –24V to
set the contrast of the display. Usually, a switching regulator is needed in the system to generate this voltage
although it runs at low power. The LT1172 generates the
voltage with a minimum parts count.
The new LTC1155 Dual Power MOSFET Driver delivers
12V of gate drive to two N-channel power MOSFETs when
powered from a 5V supply with no external components
required. This ability, coupled with its micropower current
demands and protection features, makes it an excellent
choice for high side switching applications which previously required more expensive P-channel MOSFETs.
The circuit in Figure 20 works by generating +18V to +24V
in a boost configuration and then inverting the voltage by
charge pumping. This allows the use of a small inductor
for the converter rather than a transformer.
AN51-12
Application Note 51
1N5819
DC INPUT VOLTAGE
MINIMUM TO START = VBATT + 3V VIN+
MINIMUM TO RUN = VBATT + 2V
MAXIMUM = 35V
1N5819
+
100µF
50V
C1
1µF
35V
TANT
EFFICIENCY = 90% AT 1 CHARGE = 1A
WITH VBATT = 12V
VSW
VIN
LT1171
FB
VC
GND
C3
0.1µF
R7
470Ω
C4
0.01µF
L1
50µH, 1A
KOOL Mµ® OR
FERRITE CORE
TO LOAD
+
C5
100µF
35V
R2
2k
VIN
1N5819
+
R3
2k*
LT1006
2N3904
C6
0.01µF
–
C8
500pF
R7
8.2k
1N914
BATTERY
4V TO 20V
R1
0.1Ω*
KELVIN
SENSED
R4
97.6k*
C7
0.01µF
R6
1k
VIN– (GND)
PROGRAM VOLTAGE
I CHARGE = 0.2A/V
* = 1%
LTAN51 • 15
Figure 19. High Efficiency Programmable Buck-Mode Battery Charger (Input Voltage Must be Higher than Battery Voltage)
VBATT = 3V TO VOUT + 1V
L1
47µH
VIN = 3V-5V
+
C5
2µF
VIN
D1
1N914
VSW
FB
LT1172
C1
2µF
+
R2
100k
D2
1N5817
VOUT =
–10V TO –30V
R1
200k
GND
OPTIONAL
SHUTDOWN
Q1
VN2222LL
VC
D3
1N5817
R3
15k
C4
0.047µF
C3
4.7nF
+
C2
4.7µF
LTAN51 • 16
L1 = SUMIDA CD75-470M
C1, C2 = TANTALUM
Figure 20. LCD Bias Circuit Generates –24V
Kool Mµ® is a registered trademark of Magnetics, Inc.
AN51-13
Application Note 51
D1
MBR330
9V, 2A
CURRENT
LIMITED
LED1
FAST CHG
MODE
R2
10k
1%
R9
20Ω
2W
R8
510Ω
Q1
VN2222L
3
RT1 5k NTC
KEYSTONE
RL1004-2910-97-D1
OR EQUIVALENT
+
1
2
–
R3
10k
1%
5
+
7
6
–
C3
0.01µF
R7
1k
R5
30k
4
2
R6
100k
8
LTC1155
7
C4
0.1µF
5
4
3
R4
10k
1%
R10
100k
6
1
C1
0.1µF
Q3
IRLZ24
ON/OFF
FROM µP
C2
1µF
LT1018
R1
5.11k
1%
Q5
VN2222L
Q2
IRLZ24
8
R12
100k
R11
100k
RAW
BATTERY TO
STANDBY
REGULATOR
R14
0.03Ω
D2
1N4148
+
D3
1N4148
D4
1N4148
Q4
IRLZ24
4 NiCAD
BATTERY
PACK
C5
220pF
R13
10k
1
8
ON/OFF
FROM µP
3
LT1431
7
4
5
6
5V, 2A
SWITCHED
C6
470µF
10V
ESR <0.5Ω
LTAN51 • 17
Figure 21. The LTC1155 Dual MOSFET Driver Provides Gate Drive and Protection for a 4-Cell NiCad Charger and Regulator
A notebook computer power supply system in Figure 21 is
a good example of an application which benefits directly
from this high side driving scheme. A 4-cell, NiCad battery
pack can be used to power a 5V notebook computer
system. Inexpensive N-channel power MOSFETs have
very low ON resistance and can be used to switch power
with low voltage drop between the battery pack and the 5V
logic circuits.
Figure 21 shows how a battery charger and an extremely
low voltage drop 5V regulator can be built using the
LTC1155 and three inexpensive power MOSFETs. One half
of the LTC1155 Dual MOSFET Driver controls the charging
of the battery pack. The 9V, 2A current limited wall unit is
switched directly into the battery pack through an extremely low resistance MOSFET switch, Q2. The gate drive
AN51-14
output, pin 2, generates about 13V of gate drive to fully
enhance Q1 and Q2. The voltage drop across Q2 is only
0.17V at 2A and, therefore, can be surface mounted to save
board space.
An inexpensive thermistor, RT1, measures the battery
temperature and latches the LTC1155 off when the temperature rises to 40°C by pulling low on pin 1, the Drain
Sense Input. The window comparator also ensures that
battery packs which are very cold (<10°C) are not quick
charged.
Q1 drives an indicator lamp during quick charge to let the
computer operator know that the battery pack is being
charged properly. When the battery temperature rises to
40°C, the LTC1155 latches off and the battery charge
current flowing through R9 drops to 150mA.
Application Note 51
A 4-cell NiCad battery pack produces about 6V when fully
charged. This voltage will drop to about 4.5V when the
batteries are nearly discharged. The second half of the
LTC1155 provides gate voltage drive, pin 7, for an extremely low voltage drop MOSFET regulator. The LT1431
controls the gate of Q4 and provides a regulated 5V output
when the battery is above 5V. When the battery voltage
drops below 5V, Q4 acts as a low resistance switch
between the battery and the regulator output.
A second power MOSFET, Q3, connected between the 9V
supply and the regulator output “bypasses” the main
regulator when the 9V supply is connected. This means
that the computer power is taken directly from the AC line
while the charger wall unit is connected. The LT1431
provides regulation for both Q3 and Q4, and maintains a
constant 5V at the regulator output. The diode string made
up of diodes D2-D4 ensure that Q3 conducts all the
regulator current when the wall unit is plugged in by
separating the two gate voltages by about 2V.
R14 acts as a current sense for the regulator. The regulator
latches off at 3A when the voltage drop between the
second Drain Sense Input, pin-8, and the supply, pin-6,
rises above 100mV. R10 and C3 provide a short delay. The
µP can restart the regulator by turning the second input,
pin-5, off and then back on.
The regulator is switched off by the µP when the battery
voltage drops below 4.6V. The standby current for the 5V,
2A regulator is less than 10µA. The regulator is switched
on again when the battery voltage rises during charging.
Power dissipation in the notebook computer itself is
generally quite low. The current limited wall unit dissipates
the bulk of the power created by quick charging the battery
pack. Q2 dissipates less than 0.5W. R9 dissipates about
0.7W. Q4 dissipates about 2W for a very short period of
time when the batteries are fully charged and dissipates
less than 0.5W as soon as the battery voltage drops to 5V.
The three integrated circuits shown are micropower and
dissipate virtually no power. Q3, however, can dissipate as
much as 7W if the full 2A output current is required while
powered from the wall unit.
The circuit shown in Figure 21 consumes very little board
space. The LTC1155 is available in an 8-pin SO package
and the three power MOSFETs can also be housed in SO
packaging. Q3 and Q4 must be heat sinked properly
however. (Consult the MOSFET manufacturer data sheet
for surface mount heat sink recommendations).
The LTC1155 allows the use of inexpensive N-channel
MOSFET switches to directly connect power from a 4-cell
NiCad battery pack to the charger and the load. This
technique is very cost effective and is also very efficient.
Nearly all the battery power is delivered directly to the load
to ensure maximum operating time from the batteries.
POWER SUPPLIES FOR PALMTOP COMPUTERS
Palmtop computer power supply designs present an entirely separate set of problems from notebook computers.
Notebook machines typically use a 9V to 15V NiCad stack
for the power source. Palmtop machines, due to their
extremely small size, have room for only two or four AA
cells. The palmtop machines require much longer operating time in sleep mode, since they presently do not have
disk drives. A typical palmtop system may have several
hours of operating life with the processor at full activity,
tens of hours of quiescent operation with the processor
shutdown but the display active, and up to two months life
in sleep mode where all memory is retained but no
computation takes place. Palmtop machines also use a
lithium battery for backup power when the AA cells are
dead or being replaced.
The power source for palmtops are usually disposable AA
alkaline cells. The use of these disposable batteries generates a separate set of problems from notebook computers.
Unlike power supply systems powered by rechargeable
NiCad or NiMH batteries, high efficiency power converter
circuits are not necessarily optimum for use with disposable batteries. Since rechargeable batteries have very low
output impedance, the most efficient converter circuits
result in maximum operating time.
Disposable cells, on the other hand, have relatively high
internal impedance, so maximum battery life results when
the battery load is low and relatively constant. Power
supply converters that minimize both the loss in the
AN51-15
Application Note 51
converter circuit and minimize the effect of battery internal
resistance will give longest system operating life. Some of
the 4-cell designs presented here are optimized for low
peak battery current to lengthen the disposable battery
life. Other configurations, while they may have higher
efficiency, require higher peak energy demands on the
battery and consequently shorten the battery life. The
converter circuits shown here have been tested using
alkaline AA cells and provide long battery life.
2-Cell Input Palmtop Power Supply Circuits
A regulated 5V supply can be generated from two AA cells
using the circuit shown in Figure 22. U1, an LT1108-5
micropower DC to DC converter, is arranged as a step-up,
or “boost” converter. The 5V output, monitored by U1’s
SENSE pin, is internally divided down and compared to a
1.25V reference voltage inside the device. U1’s oscillator
turns on when the output drops below 5V, cycling the
switching transistor at a 19kHz rate. This action alternately causes current to build up in L1, then dump into C1
through D1, increasing the output voltage. When the
output reaches 5V, the oscillator turns off.
L1*
100µH
R1
47Ω
ILIM
3V
2 × AA
CELL
D1
1N5818
VIN
voltage range, allowing minimum energy transfer to occur
at low battery voltage without exceeding L1’s maximum
current rating at high battery voltage. Maximum current
demands should be carefully considered, with R1 tailored
to the individual application to obtain longest possible
battery life. For example, if only 75mA maximum is required, R1 can be increased to 100Ω. This will limit switch
current to approximately 650mA which has the effect of
increasing converter efficiency and lowering peak current
demands, considerably extending battery life.
The circuit delivers 5V at up to 150mA from an input range
of 3.5V to 2.0V. Efficiency measures 80% at 3.0V,
decreasing to 70% at 2.0V for load currents in the 15mA
to 150mA range. Output ripple measures 75mVP-P and noload quiescent current is just 135µA.
LCD Bias from 2 AA Cells
A –24V LCD bias generator is shown in Figure 23. In this
circuit U1 is an LT1173 micropower DC to DC converter.
The 3V input is converted to +24V by U1’s switch, L1, D1,
and C1. The switch pin (SW1) then drives a charge pump
consisting of C2, C3, D2, and D3 to level shift the +24V
output to -24V. Line regulation is less than 0.2% from 3.3V
to 2.0V inputs. Load regulation measures 2% from a 1mA
to 7mA load. The circuit delivers up to 7mA from a 2.0V
input at 73% efficiency.
SW1
SENSE
GND
SW2
D1
1N4148
L1*
100 µ H
U1
LT1108-5
+
5V OUTPUT
150mA
R1
100Ω
C1
100µF ✝
SW1
LTAN51 • 18
U1
LT1173
Figure 22. 2 AA Cells to 5V Deliver 150mA
The gated oscillator provides the mechanism to keep the
output at a constant 5V. R1 invokes the current limit
feature of the LT1108, limiting peak switch current to
approximately 1A. U1 limits switch current by turning off
the switch when the current reaches the programmed limit
set by R1. Switch “on” time, therefore, decreases as VIN is
increased. Switch “off” time is not affected. This scheme
keeps peak switch current constant over the entire input
2.21M**
VIN
I LIM
*COILTRONICS CTX100-2
✝
SANYO OS-CON 16SA100
+
C2
4.7µ F
C1
0.1µF
FB
GND
SW2
D3
1N5818
3V
2 × AA
CELL
D4
1N4148
OPERATE SHUTDOWN
*TOKO 262LYF-0092K
**1% METAL FILM
118k**
D2
1N5818
+
C3
22 µ F
–24V OUTPUT
LTAN51 • 19
Figure 23. 2-Cell LCD Supply Generates –24 at 7mA
AN51-16
Application Note 51
MAIN
BATTERY
OUTPUT
L1*
100µH
1N5818
SI9405DY
VLOGIC
+
+
120Ω
150µF
150µF
1
2
ILIM
4 × AA
ALKALINE
7
U1
LT1173
SET
470k
VIN
3
SW1
AO
FB
SW2
GND
5
75k
+
330µF
6
8
240Ω
4
62k
39k
5V/3.6V
VN2222
L2**
220µH
1N4933
+
910k
1k
3
3V
LITHIUM
470k
1
ILIM
SW1
+
33µF
360k
2
VIN
6
AO
BL4
2.4V
U2
LT1173
10µF
7
1M
SET
GND
5
* L1 = TOKO 262LYF-0092K
** L2 = TOKO 262LYF-0096K
FB
SW2
4
8
200k
VN2222
BKUP/NORM
LTAN51 • 20
Figure 24. Main Logic Converter Generates 3.6V/5V at 200mA; Backup Converter Generates 3.4V
when Main Battery is Dead or Removed
4-Cell Input Palmtop Power Supply Circuits
Newer, more powerful palmtop machines using 386SX
processors require more power than two AA cells can
deliver for reasonable operating life. The circuits shown
here provide a switchable 3.6V/5V output for main logic,
a –23V output for LCD display bias, a +12V output for Flash
memory VPP generation, and an automatic backup supply
using a 3V lithium cell. Under no-load conditions, the
quiescent current required by the entire system is 380µA.
The main converter circuit shown in Figure 24 is a combination step-up/step-down converter. When the 4 AA cells
are fresh, the circuit behaves as a linear regulator. While
this may seem to be inefficient, note that the battery
voltage normally quickly drops from 6V to 5V. At 5V input,
the efficiency is 3.6V/5V or 72%. As battery voltage drops
further, efficiency increases, reaching over 90% at 4.2V
input. When the battery drops below 4V, the circuit switches
over to step-up mode, squeezing every bit of available
energy out of the battery.
AN51-17
Application Note 51
until the main logic supply voltage drops to 3.4V. This
converter is capable of supplying 3.6V at 10mA. If the
BACKUP/NORMAL signal is driven from Figure 27’s circuit, the backup converter will automatically kick in when
the main AA cells are removed or dead. A low-battery
detector function is provided using the gain block inside
the LT1173. The 910k/1M divider set the BL4 output to go
low when VBATT equals 2.4V.
The converter delivers 200mA at 3.6V with as little as 2.5V
input. In step-up mode, efficiency runs between 83% and
73% (at 2.5VIN). The linear regulator has no current
spikes. AA alkaline cells have a fairly high internal impedance, and the current spikes that switching regulators
demand from the battery reduces battery life. A 4-cell AA
alkaline battery has an impedance of about 0.5Ω when
fresh, increasing to 2Ω at end-of-life. This topology delivers over 9.3 hours of 3.6V, 200mA output power, compared to just 7 hours using a flyback topology.
The –24V LCD bias generator, shown in Figure 25, uses
the LT1173 as a controller driving the FZT749, a 2A PNP
in a SOT-223 package. The LT1173 maintains 1.25V
between its FB pin and its GND pin. Current must flow
through the 3M resistor to force 1.25V across R1. This
forces the “GND” pin negative. The 220µH inductor limits
switch current to 500mA from a fresh battery and 300mA
from a dead (3.6V) battery. Efficiency of this converter is
in the 70% range. Higher efficiency can be obtained
merely by decreasing the value of the inductor; however,
this will actually DECREASE battery life due to the higher
current spikes drawn from the battery.
A backup function is implemented with another LT1173
circuit also shown in Figure 24. Power for the LT1173
comes from the main logic output. The lithium battery
sees a load consisting of the 10µF capacitor leakage,
switch leakage, and about 1.5µA due to the 910k/1M
resistor divider. The total load is less than 5µA. The
LT1173 requires 110µA quiescent current, taken from the
main logic supply line.
When the BACKUP/NORMAL input goes high, the feedback string is connected, but the converter does not cycle
MAIN
BATTERY
INPUT
BSS315
SND/LCD
220Ω
1
ILIM
2
VIN
3
SW1
180Ω
FZT749
U3
LT1173
SW2
4
FB
GND
5
1N4148
+
3M
8
100µF
25V
R1
160k
1N5818
220µH
–24V LCD
BIAS
LTAN51 • 21
Figure 25. LCD Bias Generator Delivers –24V at 10mA
AN51-18
Application Note 51
A Flash memory VPP generator is shown in Figure 26.
Up to 40mA at 12V is available from the output. The
converter is switched on and off via the small N-channel
MOSFET connected to the 124k feedback resistor. When
the MOSFET is turned on, the resistor is connected
to ground and the converter generates 12V. When the
MOSFET is off, the 124k resistor is disconnected and
the feedback pin floats high, turning off the converter.
When off, output voltage sits at battery voltage minus a
diode drop. This condition is approved for Flash
memory. Inadvertent programming cannot occur as the
Flash chip contains a level detector. When the VPP pin
voltage is less than 11.4V, the Flash chip itself will not
allow programming to take place. Another low-battery
Finally, a micropower two-terminal reference and dual
comparator form a pair of battery detectors. The upper
comparator in Figure 27 senses the main battery directly.
When the battery voltage falls below 2.5V (a very dead
battery!) or the battery is removed, BL3 will go high. If
connected to the BACKUP/NORMAL signal of the lithium
backup converter, the backup will take over the main logic
supply line automatically. The other comparator goes low
when the battery voltage falls below 3.6V.
100µH
MAIN
BATTERY
INPUT
443k
6
BL1
4.0V
detect function is provided using the LT1173’s gain
block. The main alkaline battery is being sensed here,
and the AO pin goes low when the battery voltage
falls below 4.0V.
1N5818
12V
VPP
1.07M
1%
100Ω
1
ILIM
AO
2
VIN
3
SW1
+
U4
LT1173
1M
VLOGIC
7
SET
GND
5
FB
SW2
47µF
16V
8
124k
1%
4
200k
VN2222
PGM/SND
LTAN51 • 22
Figure 26. Flash Memory VPP Generator Delivers 12V, 40mA from 4 AA Input
VLOGIC
47k
MAIN
BATTERY
INPUT
2
3
TO BKUP/NORMAL
–
8
1/2
LT1017
1
BL3
2.5V
7
BL2
3.6V
+
430k
5
LT1004-2.5
6
+
1/2
LT1017
–
4
1M
LTAN51 • 23
Figure 27. Battery Detectors Sense Removal of Main Battery, Indicate VBATT < 3.6V
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
AN51-19
Application Note 51
A CCFL Backlight Driver for Palmtop Machines
L1, Q1, and Q2 comprise a current driven Royer class
converter which oscillates at a frequency primarily set by
L1’s characteristics (including its load) and the 0.01µF
capacitor. This entire converter is gated on and off by the
burst mode operation of the LT1173. The 1M/0.01µF RC
at the LT1173 feedback pin filters the half-sine appearing
at the 3.3k-1M potentiometer chain. This signal represents 1/2 the lamp current. The LT1173 servos the energy
in the lamp to maintain 1.25V at its feedback pin, closing
a loop. For low bulb currents, the LT1173 idles most of the
time, drawing only 110µA quiescent current. At the 1mA
maximum bulb current, the circuit draws less than 100mA.
A substantial amount of light is emitted by the bulb at an
input current drain of less than 5mA.
Backlit displays have greatly enhanced user acceptance of
portable computers. Palmtop machines have not used
backlit displays because of the high power required by the
inverter circuit used to drive the bulb. Figure 28’s circuit,
a micropower CCFL supply, overcomes this problem. A
typical notebook CCFL supply drives the bulb at 5mA. This
circuit, using an LT1173 micropower DC to DC converter,
operates over an input range of 2.0V to 6V. Maximum bulb
current is limited to 1mA. Control over bulb current is
maintained down to 1µA, a very dim light! It is intended for
palmtop applications where the longest possible battery
life is desired.
LAMP
15pF
3kV
9
7
L1
5
1
2
3
4
+
+VIN
10µF
D2
1N4148
C1
0.01µF
Q1
MPS650
330Ω
+VIN
2V TO 6V
47Ω
1N5818
L2
82µH
AO
SW1
LT1173
NC
Q2
MPS650
3.3k
VIN
ILIM
NC
D1
1N4148
SET
INTENSITY
ADJUST
1M
1N5818
1M
FB
SW2
GND
1N4148
0.01µF
SHUTDOWN
C1 = MUST BE A LOW LOSS CAPACITOR.
METALIZED POLYCARB
WIMA FKP2 (GERMAN) RECOMMENDED.
L1 = SUMIDA-6345-020 OR COILTRONICS-CTX110092-1.
PIN NUMBERS SHOWN FOR COILTRONICS UNIT
L2 = TOKO 262LYF-0091K
DO NOT SUBSTITUTE COMPONENTS
LTAN51 • 24
Figure 28. Micropower CCFL Driver Delivers Up to 1mA of Bulb Current from 2 AA Cells
Linear Technology Corporation
McCarthy Blvd., Milpitas, CA 95035-7487
AN51-20 1630
(408) 432-1900
: (408) 434-0507
: 499-3977
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FAX
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TELEX
LT/GP 0194 5K REV B • PRINTED IN USA
 LINEAR TECHNOLOGY CORPORATION 1994