HIP6005B TM Data Sheet February 1999 FN4568.2 Buck Pulse-Width Modulator (PWM) Controller and Output Voltage Monitor Features The HIP6005B provides complete control and protection for a DC-DC converter optimized for high-performance microprocessor applications. It is designed to drive an N-Channel MOSFET in a standard buck topology. The HIP6005B integrates all of the control, output adjustment, monitoring and protection functions into a single package. • Operates from +5V or +12V Input The output voltage of the converter is easily adjusted and precisely regulated. The HIP6005B includes a fully TTL-compatible 5-input digital-to-analog converter (DAC) that adjusts the output voltage from 1.3VDC to 2.05VDC in 0.05V increments and from 2.1V DC to 3.5VDC in 0.1V steps. The precision reference and voltage-mode regulator hold the selected output voltage to within ±1% over temperature and line voltage variations. • Excellent Output Voltage Regulation - ±1% Over Line Voltage and Temperature The HIP6005B provides simple, single feedback loop, voltagemode control with fast transient response. It includes a 200kHz free-running triangle-wave oscillator that is adjustable from below 50kHz to over 1MHz. The error amplifier features a 15MHz gain-bandwidth product and 6V/µs slew rate which enables high converter bandwidth for fast transient performance. The resulting PWM duty ratio ranges from 0% to 100%. • Over-Voltage and Over-Current Fault Monitors - Does Not Require Extra Current Sensing Element, Uses MOSFET’s rDS(ON) • Drives N-Channel MOSFET The HIP6005B monitors the output voltage with a window comparator that tracks the DAC output and issues a Power Good signal when the output is within ±10%. The HIP6005B protects against over-current and over-voltage conditions by inhibiting PWM operation. Additional built-in over-voltage protection triggers an external SCR to crowbar the input supply. The HIP6005B monitors the current by using the rDS(ON) of the upper MOSFET which eliminates the need for a current sensing resistor. Ordering Information PART NUMBER TEMP. RANGE ( oC) PACKAGE PKG. NO. HIP6005BCB 0 to 70 20 Ld SOIC M20.3 HIP6005BCV 0 to 70 20 Ld TSSOP M20.173 NOTE: When ordering, use the entire part number. Add the suffix T to obtain the TSSOP variant in tape and reel, e.g., HIP6005BCV-T. • Simple Single-Loop Control Design - Voltage-Mode PWM Control • Fast Transient Response - High-Bandwidth Error Amplifier - Full 0% to 100% Duty Ratio • TTL-Compatible 5-Bit Digital-to-Analog Output Voltage Selection - Wide Range . . . . . . . . . . . . . . . . . . . 1.3V DC to 3.5VDC - 0.1V Binary Steps . . . . . . . . . . . . . . 2.1V DC to 3.5VDC - 0.05V Binary Steps . . . . . . . . . . . . 1.3VDC to 2.05VDC • Power-Good Output Voltage Monitor • Small Converter Size - Constant Frequency Operation - 200kHz Free-Running Oscillator Programmable from 50kHz to over 1MHz Applications • Power Supply for Pentium®, Pentium Pro, Pentium II, PowerPC™, K6™, 6X86™ and Alpha™ Microprocessors • High-Power 5V to 3.xV DC-DC Regulators • Low-Voltage Distributed Power Supplies Pinout HIP6005B (SOIC, TSSOP) TOP VIEW VSEN 1 20 RT OCSET 2 19 OVP SS 3 18 VCC VID0 4 17 NC VID1 5 16 NC VID2 6 15 BOOT VID3 7 14 UGATE VID4 8 13 PHASE COMP 9 12 PGOOD FB 10 11 GND 6X86™ is a trademark of Cyrix Corporation. Alpha Micro™ is a trademark of Digital Computer Equipment Corporation K6™ is a trademark of Advanced Micro Devices. Pentium® is a registered trademark of Intel Corporation. PowerPC™ is a registered trademark of IBM. 110 CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 321-724-7143 | Intersil (and design) is a trademark of Intersil Americas Inc. Copyright © Intersil Americas Inc. 2002. All Rights Reserved HIP6005B Typical Application +12V VIN = +5V OR +12V VCC PGOOD OCSET MONITOR AND PROTECTION SS OVP BOOT RT OSC UGATE VID0 VID1 VID2 VID3 VID4 PHASE HIP6005B D/A +VOUT - FB + + - COMP GND VSEN Block Diagram VCC VSEN POWER-ON RESET (POR) 110% + - PGOOD 90% + OVERVOLTAGE 115% 10µA + OVP SOFTSTART + - OCSET REFERENCE 200µA SS OVERCURRENT BOOT 4V UGATE PHASE VID0 VID1 VID2 VID3 VID4 FB D/A CONVERTER (DAC) PWM COMPARATOR DACOUT + - + - ERROR AMP INHIBIT PWM GATE CONTROL LOGIC COMP GND RT OSCILLATOR HIP6005B Absolute Maximum Ratings Thermal Information Supply Voltage, V CC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .+15V Boot Voltage, VBOOT - VPHASE . . . . . . . . . . . . . . . . . . . . . . . .+15V Input, Output or I/O Voltage . . . . . . . . . . . . GND -0.3V to VCC +0.3V ESD Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Class 2 Thermal Resistance (Typical, Note 1) θJA (oC/W) SOIC Package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 110 86 SOIC Package (with 3in2 of Copper) . . . . . . . . . . . . TSSOP Package . . . . . . . . . . . . . . . . . . . . . . . . . . . 140 Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . . 150oC Maximum Storage Temperature Range . . . . . . . . . -65oC to 150oC Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . 300oC (Lead Tips Only) Operating Conditions Supply Voltage, V CC . . . . . . . . . . . . . . . . . . . . . . . . . . . +12V ±10% Ambient Temperature Range. . . . . . . . . . . . . . . . . . . . . 0oC to 70oC CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. NOTE: 1. θJA is measured with the component mounted on an evaluation PC board in free air. Electrical Specifications Recommended Operating Conditions, Unless Otherwise Noted PARAMETER SYMBOL TEST CONDITIONS MIN TYP MAX UNITS UGATE Open - 5 - mA Rising VCC Threshold VOCSET = 4.5V - - 10.4 V Falling VCC Threshold VOCSET = 4.5V 8.2 - - V - 1.26 - V RT = Open 185 200 215 kHz 6kΩ < RT to GND < 200kΩ -15 - +15 % - 1.9 - VP-P - - 0.8 V V CC SUPPLY CURRENT Nominal Supply ICC POWER-ON RESET Rising VOCSET Threshold OSCILLATOR Free Running Frequency Total Variation Ramp Amplitude ∆VOSC RT = Open REFERENCE AND DAC DAC (VID0-VID4) Input Low Voltage DAC (VID0-VID4) Input High Voltage 2.0 - - V DACOUT Voltage Accuracy -1.0 - +1.0 % - 88 - dB - 15 - MHz - 6 - V/µs 350 500 - mA - 5.5 10 W - 115 120 % VOCSET = 4.5V 170 200 230 µA VSEN = 5.5V; VOVP = 0V 60 - - mA - 10 - µA VSEN Rising 106 - 111 % Lower Threshold (VSEN /DACOUT) VSEN Falling 89 - 94 % Hysteresis (V SEN /DACOUT) Upper and Lower Threshold - 2 - % IPGOOD = -5mA - 0.5 - V ERROR AMPLIFIER DC Gain Gain-Bandwidth Product Slew Rate GBW SR COMP = 10pF GATE DRIVER Upper Gate Source IUGATE Upper Gate Sink RUGATE VBOOT - VPHASE = 12V, VUGATE = 6V PROTECTION Over-Voltage Trip (VSEN/DACOUT) OCSET Current Source IOCSET OVP Sourcing Current IOVP Soft Start Current ISS POWER GOOD Upper Threshold (VSEN /DACOUT) PGOOD Voltage Low VPGOOD HIP6005B Typical Performance Curves 40 1000 CUGATE = 3300pF 30 RT PULLUP TO +12V ICC (mA) RESISTANCE (kΩ) 35 100 25 CUGATE = 1000pF 20 15 10 10 RT PULLDOWN TO VSS CUGATE = 10pF 5 10 100 1000 0 100 200 300 SWITCHING FREQUENCY (kHz) FIGURE 1. RT RESISTANCE vs FREQUENCY 400 500 600 700 800 900 1000 SWITCHING FREQUENCY (kHz) FIGURE 2. BIAS SUPPLY CURRENT vs FREQUENCY Functional Pin Description VID0-4 (Pins 4-8) VSEN 1 20 RT OCSET 2 19 OVP SS 3 18 VCC VID0 4 17 NC VID1 5 16 NC VID2 6 15 BOOT VID3 7 14 UGATE VID4 8 13 PHASE COMP 9 12 PGOOD FB 10 11 GND VID0-4 are the input pins to the 5-bit DAC. The states of these five pins program the internal voltage reference (DACOUT). The level of DACOUT sets the converter output voltage. It also sets the PGOOD and OVP thresholds. Table 1 specifies DACOUT for the 32 combinations of DAC inputs. COMP (Pin 9) and FB (Pin 10) COMP and FB are the available external pins of the error amplifier. The FB pin is the inverting input of the error amplifier and the COMP pin is the error amplifier output. These pins are used to compensate the voltage-control feedback loop of the converter. VSEN (Pin 1) GND (Pin 11) This pin is connected to the converters output voltage. The PGOOD and OVP comparator circuits use this signal to report output voltage status and for overvoltage protection. Signal ground for the IC. All voltage levels are measured with respect to this pin. OCSET (Pin 2) PGOOD is an open collector output used to indicate the status of the converter output voltage. This pin is pulled low when the converter output is not within ±10% of the DACOUT reference voltage. Exception to this behavior is the ‘11111’ VID pin combination which disables the converter; in this case PGOOD asserts a high level. Connect a resistor (ROCSET) from this pin to the drain of the upper MOSFET. ROCSET , an internal 200µA current source (IOCS), and the upper MOSFET on-resistance (rDS(ON)) set the converter over-current (OC) trip point according to the following equation: IOCSET x R OCSET I PEAK = ----------------------------------------------------r DS ( ON ) An over-current trip cycles the soft-start function. SS (Pin 3) Connect a capacitor from this pin to ground. This capacitor, along with an internal 10µA current source, sets the soft-start interval of the converter. PGOOD (Pin 12) PHASE (Pin 13) Connect the PHASE pin to the upper MOSFET source. This pin is used to monitor the voltage drop across the MOSFET for over-current protection. This pin also provides the return path for the upper gate drive. UGATE (Pin 14) Connect UGATE to the upper MOSFET gate. This pin provides the gate drive for the upper MOSFET. HIP6005B BOOT (Pin 15) This pin provides bias voltage to the upper MOSFET driver. A bootstrap circuit may be used to create a BOOT voltage suitable to drive a standard N-Channel MOSFET. NC (Pin 16) No connection. NC (Pin 17) No connection. VCC (Pin 18) Provide a 12V bias supply for the chip to this pin. OVP (Pin 19) The OVP pin can be used to drive an external SCR in the event of an overvoltage condition. Output rising 15% more than the DAC-set voltage triggers a high output on this pin and disables PWM gate drive circuitry. RT (Pin 20) the SS voltage reaches the valley of the oscillator’s triangle wave. The oscillator’s triangular waveform is compared to the ramping error amplifier voltage. This generates PHASE pulses of increasing width that charge the output capacitor(s). This interval of increasing pulse width continues to t2. With sufficient output voltage, the clamp on the reference input controls the output voltage. This is the interval between t2 and t3 in Figure 3. At t3 the SS voltage exceeds the DACOUT voltage and the output voltage is in regulation. This method provides a rapid and controlled output voltage rise. The PGOOD signal toggles ‘high’ when the output voltage (VSEN pin) is within ±5% of DACOUT. The 2% hysteresis built into the power good comparators prevents PGOOD oscillation due to nominal output voltage ripple. PGOOD (2V/DIV.) 0V This pin provides oscillator switching frequency adjustment. By placing a resistor (RT) from this pin to GND, the nominal 200kHz switching frequency is increased according to the following equation: SOFT-START (1V/DIV.) OUTPUT VOLTAGE (1V/DIV.) 6 5 x 10 Fs ≈ 200kHz + --------------------R T ( kΩ ) (RT to GND) Conversely, connecting a pull-up resistor (RT) from this pin to V CC reduces the switching frequency according to the following equation: 0V 0V t3 TIME (5ms/DIV.) FIGURE 3. SOFT START INTERVAL 7 (RT to 12V) Initialization The HIP6005B automatically initializes upon receipt of power. Special sequencing of the input supplies is not necessary. The Power-On Reset (POR) function continually monitors the input supply voltages. The POR monitors the bias voltage at the VCC pin and the input voltage (VIN) on the OCSET pin. The level on OCSET is equal to VIN less a fixed voltage drop (see overcurrent protection). The POR function initiates soft start operation after both input supply voltages exceed their POR thresholds. For operation with a single +12V power source, VIN and VCC are equivalent and the +12V power source must exceed the rising VCC threshold before POR initiates operation. Soft Start The POR function initiates the soft start sequence. An internal 10µA current source charges an external capacitor (CSS) on the SS pin to 4V. Soft start clamps the error amplifier output (COMP pin) and reference input (+ terminal of error amp) to the SS pin voltage. Figure 3 shows the soft start interval with CSS = 0.1µF. Initially the clamp on the error amplifier (COMP pin) controls the converter’s output voltage. At t1 in Figure 3, Over-Current Protection The over-current function protects the converter from a shorted output by using the upper MOSFET’s on-resistance, rDS(ON) to monitor the current. This method enhances the converter’s efficiency and reduces cost by eliminating a current sensing resistor. SOFT-START Functional Description 4V 2V 0V OUTPUT INDUCTOR 4 x 10 Fs ≈ 200kHz – --------------------R T (k Ω) t2 t1 15A 10A 5A 0A TIME (20ms/DIV.) FIGURE 4. OVER-CURRENT OPERATION HIP6005B The over-current function cycles the soft-start function in a hiccup mode to provide fault protection. A resistor (ROCSET) programs the over-current trip level. An internal 200µA current sink develops a voltage across ROCSET that is referenced to VIN . When the voltage across the upper MOSFET (also referenced to VIN) exceeds the voltage across ROCSET , the over-current function initiates a soft-start sequence. The softstart function discharges CSS with a 10µA current sink and inhibits PWM operation. The soft-start function recharges CSS , and PWM operation resumes with the error amplifier clamped to the SS voltage. Should an overload occur while recharging CSS , the soft start function inhibits PWM operation while fully charging CSS to 4V to complete its cycle. Figure 4 shows this operation with an overload condition. Note that the inductor current increases to over 15A during the CSS charging interval and causes an over-current trip. The converter dissipates very little power with this method. The measured input power for the conditions of Figure 4 is 2.5W. The over-current function will trip at a peak inductor current (IPEAK) determined by: I OCSET x R OCSET I PEAK = ----------------------------------------------------r DS ( ON ) where IOCSET is the internal OCSET current source (200µA typical). The OC trip point varies mainly due to the MOSFET’s rDS(ON) variations. To avoid over-current tripping in the normal operating load range, find the R OCSET resistor from the equation above with: 1. The maximum rDS(ON) at the highest junction temperature. 2. The minimum IOCSET from the specification table. 3. Determine IPEAK for I PEAK > IOUT ( MAX ) + ( ∆I ) ⁄ 2 , where ∆I is the output inductor ripple current. For an equation for the ripple current see the section under component guidelines titled “Output Inductor Selection.” A small ceramic capacitor should be placed in parallel with ROCSET to smooth the voltage across ROCSET in the presence of switching noise on the input voltage. Output Voltage Program The output voltage of a HIP6005B converter is programmed to discrete levels between 1.8V DC and 3.5VDC . The voltage identification (VID) pins program an internal voltage reference (DACOUT) with a TTL-compatible, 5-bit digital-to-analog converter (DAC). The level of DACOUT also sets the PGOOD and OVP thresholds. Table 1 specifies the DACOUT voltage for the 32 different combinations of connections on the VID pins. The output voltage should not be adjusted while the converter is delivering power. Remove input power before changing the output voltage. Adjusting the output voltage during operation could toggle the PGOOD signal and exercise the overvoltage protection. ‘11111’ VID pin combination resulting in a 0V output setting activates the Power-On Reset function and disables the gate drive circuitry. For this specific VID combination, though, PGOOD asserts a high level. This unusual behavior has been implemented in order to allow for operation in dual-microprocessor systems where AND-ing of the PGOOD signals from two individual power converters is implemented. Application Guidelines Layout Considerations As in any high frequency switching converter, layout is very important. Switching current from one power device to another can generate voltage transients across the impedances of the interconnecting bond wires and circuit traces. These interconnecting impedances should be minimized by using wide, short printed circuit traces. The critical components should be located as close together as possible using ground plane construction or single point grounding. TABLE 1. OUTPUT VOLTAGE PROGRAM PIN NAME PIN NAME NOMINAL OUTPUT VID4 VID3 VID2 VID1 VID0 VOLTAGE DACOUT VID4 VID3 VID2 VID1 0 1 1 1 1 1.30 1 1 1 1 0 1 1 1 0 1.35 1 1 1 1 0 1 1 0 1 1.40 1 1 1 0 0 1 1 0 0 1.45 1 1 1 0 0 1 0 1 1 1.50 1 1 0 1 0 1 0 1 0 1.55 1 1 0 1 0 1 0 0 1 1.60 1 1 0 0 0 1 0 0 0 1.65 1 1 0 0 0 0 1 1 1 1.70 1 0 1 1 0 0 1 1 0 1.75 1 0 1 1 0 0 1 0 1 1.80 1 0 1 0 0 0 1 0 0 1.85 1 0 1 0 0 0 0 1 1 1.90 1 0 0 1 0 0 0 1 0 1.95 1 0 0 1 0 0 0 0 1 2.00 1 0 0 0 0 0 0 0 0 2.05 1 0 0 0 NOTE: 0 = connected to GND or VSS , 1 = connected to VDD through pull-up resistors. VID0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 NOMINAL OUTPUT VOLTAGE DACOUT 0 2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.9 3.0 3.1 3.2 3.3 3.4 3.5 HIP6005B The modulator transfer function is the small-signal transfer function of VOUT/VE/A . This function is dominated by a DC Gain and the output filter (LO and CO ), with a double pole break frequency at F LC and a zero at FESR . The DC Gain of the modulator is simply the input voltage (VIN ) divided by the peak-to-peak oscillator voltage ∆VOSC . VIN HIP6005B Q1 LO CIN VOUT CO D2 VIN LOAD UGATE PHASE OSC DRIVER PWM COMPARATOR ∆VOSC RETURN - Figure 7 highlights the voltage-mode control loop for a buck converter. The output voltage (VOUT) is regulated to the Reference voltage level. The error amplifier (Error Amp) output (VE/A) is compared with the oscillator (OSC) triangular wave to provide a pulse-width modulated (PWM) wave with an amplitude of VIN at the PHASE node. The PWM wave is smoothed by the output filter (LO and CO). +VIN D1 LO VOUT PHASE VCC SS Q1 +12V D2 CO CVCC CSS GND FIGURE 6. PRINTED CIRCUIT BOARD SMALL SIGNAL LAYOUT GUIDELINES LOAD HIP6005B CO ESR (PARASITIC) VE/A ZIN - + REFERENCE ERROR AMP DETAILED COMPENSATION COMPONENTS ZFB C2 C1 VOUT ZIN R2 C3 R3 R1 COMP - FB + HIP6005B DACOUT FIGURE 7. VOLTAGE-MODE BUCK CONVERTER COMPENSATION DESIGN Modulator Break Frequency Equations Feedback Compensation CBOOT PHASE ZFB Figure 5 shows the critical power components of the converter. To minimize the voltage overshoot the interconnecting wires indicated by heavy lines should be part of ground or power plane in a printed circuit board. The components shown in Figure 6 should be located as close together as possible. Please note that the capacitors CIN and CO each represent numerous physical capacitors. Locate the HIP6005B within 3 inches of the MOSFET, Q1 . The circuit traces for the MOSFET’s gate and source connections from the HIP6005B must be sized to handle up to 1A peak current. BOOT VOUT + FIGURE 5. PRINTED CIRCUIT BOARD POWER AND GROUND PLANES OR ISLANDS Figure 6 shows the circuit traces that require additional layout consideration. Use single point and ground plane construction for the circuits shown. Minimize any leakage current paths on the SS PIN and locate the capacitor, Css close to the SS pin because the internal current source is only 10µA. Provide local VCC decoupling between VCC and GND pins. Locate the capacitor, CBOOT as close as practical to the BOOT and PHASE pins. LO 1 F LC = -----------------------------------------2π x LO x C O 1 F ESR = -----------------------------------------------2π x (ESR x C O ) The compensation network consists of the error amplifier (internal to the HIP6005B) and the impedance networks ZIN and ZFB . The goal of the compensation network is to provide a closed loop transfer function with the highest 0dB crossing frequency (f0dB) and adequate phase margin. Phase margin is the difference between the closed loop phase at f0dB and 180 degrees. The equations below relate the compensation network’s poles, zeros and gain to the components (R1, R2 , R3 , C1, C2 , and C3) in Figure 8. Use these guidelines for locating the poles and zeros of the compensation network: 1. 2. 3. 4. 5. 6. 7. Pick Gain (R2/R1) for desired converter bandwidth. Place 1ST Zero Below Filter’s Double Pole (~75% FLC). Place 2ND Zero at Filter’s Double Pole. Place 1ST Pole at the ESR Zero. Place 2ND Pole at Half the Switching Frequency. Check Gain against Error Amplifier’s Open-Loop Gain. Estimate Phase Margin - Repeat if Necessary. HIP6005B Compensation Break Frequency Equations 1 F Z1 = -----------------------------------2π x R 2 x C 1 1 FP1 = -------------------------------------------------------- C1 x C 2 2π x R 2 x ---------------------- C 1 + C2 1 F Z2 = ------------------------------------------------------2π x (R 1 + R 3 ) x C 3 1 FP2 = -----------------------------------2π x R 3 x C 3 Figure 8 shows an asymptotic plot of the DC-DC converter’s gain vs frequency. The actual Modulator Gain has a high gain peak due to the high Q factor of the output filter and is not shown in Figure 8. Using the above guidelines should give a Compensation Gain similar to the curve plotted. The open loop error amplifier gain bounds the compensation gain. Check the compensation gain at FP2 with the capabilities of the error amplifier. The Closed Loop Gain is constructed on the log-log graph of Figure 8 by adding the Modulator Gain (in dB) to the Compensation Gain (in dB). This is equivalent to multiplying the modulator transfer function to the compensation transfer function and plotting the gain. The compensation gain uses external impedance networks ZFB and ZIN to provide a stable, high bandwidth (BW) overall loop. A stable control loop has a gain crossing with -20dB/decade slope and a phase margin greater than 45 degrees. Include worst case component variations when determining phase margin. 100 FZ1 FZ2 FP1 FP2 80 OPEN LOOP ERROR AMP GAIN GAIN (dB) 60 40 20 20LOG (R2/R1) 20LOG (VIN/∆VOSC) 0 COMPENSATION GAIN MODULATOR GAIN -20 CLOSED LOOP GAIN -40 FLC -60 10 100 1K FESR 10K 100K 1M 10M Component Selection Guidelines Output Capacitor Selection An output capacitor is required to filter the output and supply the load transient current. The filtering requirements are a function of the switching frequency and the ripple current. The load transient requirements are a function of the slew rate (di/dt) and the magnitude of the transient load current. These requirements are generally met with a mix of capacitors and careful layout. Modern microprocessors produce transient load rates above 1A/ns. High frequency capacitors initially supply the transient and slow the current load rate seen by the bulk capacitors. The bulk filter capacitor values are generally determined by the ESR (Effective Series Resistance) and voltage rating requirements rather than actual capacitance requirements. High frequency decoupling capacitors should be placed as close to the power pins of the load as physically possible. Be careful not to add inductance in the circuit board wiring that could cancel the usefulness of these low inductance components. Consult with the manufacturer of the load on specific decoupling requirements. For example, Intel recommends that the high frequency decoupling for the Pentium Pro be composed of at least forty (40) 1µF ceramic capacitors in the 1206 surface-mount package. Use only specialized low-ESR capacitors intended for switching-regulator applications for the bulk capacitors. The bulk capacitor’s ESR will determine the output ripple voltage and the initial voltage drop after a high slew-rate transient. An aluminum electrolytic capacitor's ESR value is related to the case size with lower ESR available in larger case sizes. However, the Equivalent Series Inductance (ESL) of these capacitors increases with case size and can reduce the usefulness of the capacitor to high slew-rate transient loading. Unfortunately, ESL is not a specified parameter. Work with your capacitor supplier and measure the capacitor’s impedance with frequency to select a suitable component. In most cases, multiple electrolytic capacitors of small case size perform better than a single large case capacitor. FREQUENCY (Hz) FIGURE 8. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN Output Inductor Selection The output inductor is selected to meet the output voltage ripple requirements and minimize the converter’s response time to the load transient. The inductor value determines the converter’s ripple current and the ripple voltage is a function of the ripple current. The ripple voltage and current are approximated by the following equations: V I N – V OU T V OUT ∆I = -------------------------------- x ---------------VIN FS x L ∆V OU T = ∆I x ESR Increasing the value of inductance reduces the ripple current and voltage. However, the large inductance values reduce the converter’s response time to a load transient. HIP6005B One of the parameters limiting the converter’s response to a load transient is the time required to change the inductor current. Given a sufficiently fast control loop design, the HIP6005B will provide either 0% or 100% duty cycle in response to a load transient. The response time is the time required to slew the inductor current from an initial current value to the transient current level. During this interval the difference between the inductor current and the transient current level must be supplied by the output capacitor. Minimizing the response time can minimize the output capacitance required. The response time to a transient is different for the application of load and the removal of load. The following equations give the approximate response time interval for application and removal of a transient load: L x I TRAN t RISE = -------------------------------V IN – V OUT L x ITRAN t FALL = ---------------------------VOUT where: ITRAN is the transient load current step, tRISE is the response time to the application of load, and tFALL is the response time to the removal of load. With a +5V input source, the worst case response time can be either at the application or removal of load and dependent upon the DACOUT setting. Be sure to check both of these equations at the minimum and maximum output levels for the worst case response time. With a +12V input, and output voltage level equal to DACOUT, tFALL is the longest response time. Input Capacitor Selection Use a mix of input bypass capacitors to control the voltage overshoot across the MOSFETs. Use small ceramic capacitors for high frequency decoupling and bulk capacitors to supply the current needed each time Q1 turns on. Place the small ceramic capacitors physically close to the MOSFETs and between the drain of Q1 and the anode of Schottky diode D2 . The important parameters for the bulk input capacitor are the voltage rating and the RMS current rating. For reliable operation, select the bulk capacitor with voltage and current ratings above the maximum input voltage and largest RMS current required by the circuit. The capacitor voltage rating should be at least 1.25 times greater than the maximum input voltage and a voltage rating of 1.5 times is a conservative guideline. The RMS current rating requirement for the input capacitor of a buck regulator is approximately 1/2 the DC load current. For a through hole design, several electrolytic capacitors (Panasonic HFQ series or Nichicon PL series or Sanyo MV-GX or equivalent) may be needed. For surface mount designs, solid tantalum capacitors can be used, but caution must be exercised with regard to the capacitor surge current rating. These capacitors must be capable of handling the surgecurrent at power-up. The TPS series available from AVX, and the 593D series from Sprague are both surge current tested. MOSFET Selection/Considerations The HIP6005B requires an N-Channel power MOSFET. It should be selected based upon rDS(ON) , gate supply requirements, and thermal management requirements. In high-current applications, the MOSFET power dissipation, package selection and heatsink are the dominant design factors. The power dissipation includes two loss components; conduction loss and switching loss. The conduction losses are the largest component of power dissipation for the MOSFET. Switching losses also contribute to the overall MOSFET power loss (see the equations below). These equations assume linear voltage-current transitions and are approximations. The gatecharge losses are dissipated by the HIP6005B and do not heat the MOSFET. However, large gate-charge increases the switching interval, tSW , which increases the upper MOSFET switching losses. Ensure that the MOSFET is within its maximum junction temperature at high ambient temperature by calculating the temperature rise according to package thermalresistance specifications. A separate heatsink may be necessary depending upon MOSFET power, package type, ambient temperature and air flow. PCOND = IO 2 x r DS(ON) x D P SW = 1/2 I O x VIN x t SW x F S Where: D is the duty cycle = V OUT /VIN , tSW is the switching interval, and FS is the switching frequency Standard-gate MOSFETs are normally recommended for use with the HIP6005B. However, logic-level gate MOSFETs can be used under special circumstances. The input voltage, upper gate drive level, and the MOSFETs absolute gate-to-source voltage rating determine whether logic-level MOSFETs are appropriate. Figure 9 shows the upper gate drive (BOOT pin) supplied by a bootstrap circuit from VCC . The boot capacitor, CBOOT , develops a floating supply voltage referenced to the PHASE pin. This supply is refreshed each cycle to a voltage of V CC less the boot diode drop (V D) when the Schottky diode, D2, conducts. Logic-level MOSFETs can only be used if the MOSFETs absolute gate-to-source voltage rating exceeds the maximum voltage applied to VCC . Figure 10 shows the upper gate drive supplied by a direct connection to VCC . This option should only be used in converter systems where the main input voltage is +5VDC or less. The peak upper gate-to-source voltage is approximately VCC less the input supply. For +5V main power and +12VDC for the bias, the gate-to-source voltage of Q1 is 7V. A logic-level MOSFET is a good choice for Q1 under these conditions. HIP6005B Schottky Selection +12V DBOOT VCC +5V OR +12 + VD BOOT HIP6005B CBOOT UGATE PCOND = I0 x Vf x (1 - D) Q1 (NOTE) PHASE D2 - + GND NOTE: VG-S ≈ VCC - VD . FIGURE 9. UPPER GATE DRIVE - BOOTSTRAP OPTION +12V +5V OR LESS VCC HIP6005B Rectifier D2 conducts when the upper MOSFET Q1 is off. The diode should be a Schottky type for low power losses. The power dissipation in the Schottky rectifier is approximated by: BOOT UGATE Q1 PHASE NOTE: VG-S ≈ VCC -5V - + D2 GND FIGURE 10. UPPER GATE DRIVE - DIRECT V CC DRIVE OPTION Where: D is the duty cycle = VOUT / VIN , and Vf is the Schottky forward voltage drop. In addition to power dissipation, package selection and heat-sink requirements are the main design tradeoffs in choosing the schottky rectifier. Since the three factors are interrelated, the selection process is an iterative procedure. The maximum junction temperature of the rectifier must remain below the manufacturer’s specified value, typically 125oC. By using the package thermal resistance specification and the schottky power dissipation equation (shown above), the junction temperature of the rectifier can be estimated. Be sure to use the available airflow and ambient temperature to determine the junction temperature rise. HIP6005B HIP6005B DC-DC Converter Application Circuit Figure 11 shows an application circuit of a DC-DC Converter for an Intel Pentium Pro microprocessor. Detailed information on the circuit, including a complete Bill-ofMaterials and circuit board description, can be found in VIN = F1 +5V OR +12V application note AN9706. Although the Application Note details the HIP6005, the same evaluation platform can be used to evaluate the HIP6005B.. L1 - 1µH CIN 5x 1000µF 2x 1µF 2N6394 +12V 2K D1 0.1µF VSEN 1 RT 20 4 5 6 7 8 VID0 VID1 VID2 VID3 VID4 FB OVP 18 19 2 OCSET MONITOR AND PROTECTION SS 3 0.1µF 1000pF VCC 1.1K 12 PGOOD 15 BOOT OSC 14 UGATE HIP6005B 0.1µF Q1 13 PHASE L2 7µH D/A - + + 10 D2 - 9 COUT 9x 1000µF 11 COMP 2.2nF +VO GND 20K 8.2nF 0.082µF 1K 20 Component Selection Notes COUT - Each 1000µF 6.3WVDC, Sanyo MV-GX or Equivalent. CIN - Each 330µF 25WVDC, Sanyo MV-GX or Equivalent. L2 - Core: Micrometals T60-52; Each Winding: 14 Turns of 17AWG. L1 - Core: Micrometals T50-52; Winding: 6 Turns of 18AWG. D 1 - 1N4148 or Equivalent. D 2 - 25A, 35V Schottky, Motorola MBR2535CTL or Equivalent. Q1 - Intersil MOSFET; RFP70N03. FIGURE 11. PENTIUM PRO DC-DC CONVERTER All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems. Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries. 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