V22N2 - JULY

July 2012
I N
T H I S
I S S U E
pushbutton controller 10
15V, 2.5A monolithic buckboost with 95% efficiency
and low noise 17
easy 2-supply current
sharing 20
Volume 22 Number 2
Combined Voltage and Current
Control Loops Simplify LED Drivers,
High Capacity Battery/Supercap
Chargers & MPPT* Solar Applications
Xin (Shin) Qi
* Maximum Power Point Tracking
sub-1mm height 24V, 15A
monolithic regulator 26
deliver 25A at 12V from
inputs to 60V 28
The rapid expansion of constant-current/constant-voltage (CC-CV)
applications, especially in LED lighting and high capacity battery
and supercapacitor chargers challenges power supply designers to
keep pace with the increasingly complicated interplay of current and
voltage control loops. A switch-mode converter designed specifically
for CC-CV offers a clear advantage,
especially when the supply has limited
power, or its power is allocated
among several competing loads.
The LTC4155 is a monolithic switching battery charger that efficiently delivers
3.5A charge current in a compact PCB footprint. See page 13.
Caption
w w w. li n ea r.com
Consider, for instance, the challenge of charging a
supercapacitor in a minimum amount of time from
a power-limited supply. To maintain constant input
power, the controlled charging current must decrease
as the output (supercapacitor) voltage increases. The
LT®3796 solves the problem of power limited or constant current/constant voltage regulation by seamlessly
combining a current regulation loop and two voltage regulation loops to control an external N-channel
power switch. The inherent wired-OR behavior of its
three transconductance error amplifiers summed into
the compensation pin, VC , ensures that the correct loop
(that is, the one closest to regulation) dominates.
(continued on page 4)
Linear in the News
In this issue...
COVER STORY
Combined Voltage and Current Control Loops
Simplify LED Drivers, High Capacity Battery/
Supercap Chargers & MPPT* Solar Applications
Xin (Shin) Qi
1
Over the past several months, Linear Technology received several significant
awards for products, system solutions, and for manufacturing quality and delivery.
Here are the highlights:
DESIGN FEATURES
Pushbutton On/Off Controller Includes
Optional Automatic Turn-On When
Handheld Device is Plugged In
Vui Min Ho
Electronic Products 2011 Product of the Year Award: LTC5569 RF Mixer
10
I2C-Controlled Li-Ion Power Management IC with
Integrated Power Devices Charges High Capacity
Batteries from Any 5V Source While Keeping Cool
David Simmons
13
15V, 2.5A Monolithic Buck-Boost DC/DC Converter
with 95% Efficiency and Low Noise Operation
Eddy Wells
17
Novel Current-Sharing IC Balances
Two Supplies with Ease
Pinkesh Sachdev
20
DESIGN IDEAS
What’s New with LTspice IV?
Gabino Alonso
24
24V, 15A Step-Down Regulator in Sub-1mm Height
Package Pushes Monolithic Performance Limits
Stephanie Dai
26
Electronic Products honored Linear with the selection of the LTC®5569 dual
broadband RF mixer as Product of the Year. From thousands of products,
the editors of Electronic Products selected those they felt are among the
most outstanding. The selection is based on a significant advance in technology or its application, innovation in design, or gain in price/performance.
The editors stated, “In order to make LTE, a 4G (fourth-generation) high datarate wireless technology attractive to the carrier for deployment, base station manufacturers worldwide are trying to build a multiband, multimode
platform that is easily field configured for use in any frequency band and
standard to make the deployment cost attractive. The 26.8dBm IIP3 LTC5569
dual-broadband RF mixer accomplishes this by combining 26.8dBm IIP3,
300mW/ch, and a wide 300MHz to 4GHz frequency range. So a receiver such
as that based on the LTC5569, is capable of working in all of the 700MHz,
880MHz, 1.7GHz, 1.8GHz, 1.9GHz, 2.4GHz, and 2.6GHz bands. In addition,
within each band, the receiver must receive signals that are 60MHz wide,
compared to the previous 20MHz wide without sacrificing performance. The
LTC5569 meets both of these requirements with outstanding performance.”
EN-Genius Network Awards Linear for Products of the Year & the Decade
Step-Down Converter Delivers 25A at
12V Output from Inputs Up to 60V
Victor Khasiev
LINEAR RECEIVES PRESTIGIOUS AWARDS
28
1.5A Rail-to-Rail Output Synchronous
Step-Down Regulator Adjusts with a Single Resistor
Jeff Zhang
30
back page circuits
32
EN-Genius Network, formerly AnalogZone, selected Linear Technology’s
µModule® family as Best Product of Our Decade. According to the editor, “The concept of the µModules allows the product designer to
come to the decisions about power management quite late in the
development cycle—simplifying the final product choice.
“Linear has also extended the concept to a family of seventeen ‘System in a
Package’ (SiP) signal chain receiver modules, the first of which, the LTM®9001,
we reviewed here. The SiPs, which feature 12-/14-/16-bit solutions, not
only simplify circuit design for the less than sure RF engineer out there but
they also bypass export regulation control of some individual ADCs.
“All-in-all the µModule concept has been a spectacularly successful commercial and technological story for Linear. We are delighted to congratulate them
on their achievement and to recognize them as the Product of Our Decade.”
2 | July 2012 : LT Journal of Analog Innovation
Linear in the news
EN-Genius Network also selected Linear’s
LTC6946 373MHz to 5.79GHz Integer-N
synthesizer as Product of the Year for
the Best Integer-N Synthesizer. The publication commented, “The parts are a
welcome addition to Linear’s RF arsenal
and will prove to be equally attractive to
designers who do not have to cope with
large bandwidth issues straight away.”
China Electronics Awards
Several major electronics publications in
China presented Linear with awards:
• EDN China: Innovation Award, Excellent
Product Award for the LTC4000 high
voltage controller and power manager.
• China Electronic Market: Editor’s Choice
Award for the Most Competitive
Power Product: LTC4000
• EEPW Editor’s Choice Awards: Best Analog
Product: LTC6803 battery stack
monitor for hybrid/electric vehicles
Best Amplifier: LT1999 high voltage
bidirectional current sense monitor
• Electronic Products China: Annual
Award: LTM8047/8048 isolated
µModule DC/DC converter
Enics Manufacturing Award
Enics, one of the largest providers of
electronics manufacturing services,
named the best suppliers of the year, and
honored Linear as the Best Component
Manufacturer at the annual Enics Fair in
Zurich, Switzerland.
LINEAR’S STRONG PATENT
PORTFOLIO RECOGNIZED
Ocean Tomo announced the leading
companies in patent assets. According
to Ocean Tomo’s ranking, Linear
Technology’s patent portfolio received an
IPQ score of 123—the highest of any major
analog semiconductor company. Ocean
Tomo claims to rate patent assets objectively based on proven statistical methodology. This reinforces the value of Linear’s
growing patent portfolio and the company’s strong analog intellectual property.
CONFERENCES & EVENTS
Power Systems Show 2012, Tokyo Big Sight,
Tokyo, Japan, July 11–13, Booth 6B-301—Linear
will showcase power products, including µModule products and FPGA power
management solutions. More info at www.
jma.or.jp/tf/en11/electronics/index.html.
EN-GENIUS NETWORK AWARDS LINEAR FOR PRODUCTS OF THE YEAR & THE DECADE
EN-Genius Network, formerly AnalogZone, selected Linear’s µModule products as Best Product of Our Decade. According to the editor, “The concept of the µModules
allows the product designer to come to the decisions about power management quite late in the development cycle—simplifying the final product choice.”
July 2012 : LT Journal of Analog Innovation | 3
The LT3796’s wide VIN range (6V to 100V) and rail-to-rail
(0V to 100V) output current monitoring and regulation allow
it to be used in a wide variety of applications from solar
battery chargers to high power LED lighting systems.
(LT3796, continued from page 1)
HIGH POWER LED DRIVER WITH
ROBUST OUTPUT SHORT CIRCUIT
PROTECTION
The additional, standalone current sense
amplifier can be configured for any
number of functions, including input current limit and input voltage regulation.
Figure 1 shows the LT3796 configured as a
boost converter to drive a 34W LED string
from a wide input range. The LED current is derated at low input voltages
to prevent external power components
from overheating. The front-end current
sense amplifier monitors the input current by converting the input current to
a voltage signal at the CSOUT pin with
The LT3796’s wide VIN range (6V to
100V) and rail-to-rail (0V to 100V) output current monitoring and regulation
allow it to be used in a wide variety of
applications from solar battery chargers to high power LED lighting systems. The fixed switching frequency,
current-mode architecture results in
stable operation over a wide range of
supply and output voltages. The LT3796
incorporates a high side current sense,
enabling its use in boost, buck, buckboost or SEPIC and flyback topologies.
Figure 1. A 34W LED driver with robust
output short-circuit protection.
VCSOUT = IIN • RSNS1 •
The LT3796 includes short-circuit protection independent of the LED current sense. The short-circuit protection
feature prevents the development of
excessive switching currents and protects the power components. The
protection threshold (375mV, typ) is
designed to be 50% higher than the
default LED current sense threshold.
R6
R5
The resistor network at the FB1 pin
provides OPENLED protection, which
limits the output voltage and prevents
the ISP pin, ISN pin and several external
VIN
9V TO 60V
100V (TRANSIENT)
R1
1M
R2
118k
R3
499k
OPTIONAL INPUT
CURRENT REPORTING
R4
97.6k
VS
CSN
CSP
EN/UVLO
GATE
R6
40.2k
PWM
SYNC
LED CURRENT REPORTING
INTVCC
R10
100k
R9
100k
FAULT
VMODE
RSNS
15mΩ
FB1
LT3796
SYNC
ISP
ISMON
Q1
M2
TG
FAULT
INTVCC
VMODE
R11 402k (OPT)
RLED
620mΩ
ISN
C4
0.1µF
R11 OPTIONAL
FOR FAULT LATCHOFF
UP TO
400mA
GND
PWM
VREF
M1: INFINEON BCS160N10NS3-G
M2: VISHAY SILICONIX Si7113DN
L1: COILTRONICS DR127-220
D1: DIODES INC PDS5100
D2: VISHAY ES1C
Q1: ZETEX FMMT589
LED: CREE XLAMP XR-E
R8
13.7k
M1
SENSE
CSOUT
C3
10nF
C2
2.2µF
×4
100V
R7
1M
CTRL
CSOUT
D1
R5
2k
VIN
circuits.linear.com/558
L1 22µH
RSNS1 50mΩ
IIN
C1
2.2µF
×3
LTspice IV
4 | July 2012 : LT Journal of Analog Innovation
components from exceeding their maximum rating. If an LED fails open or if
the LED string is removed from the high
power driver, the FB constant voltage
loop takes over and regulates the output to 92.5V. The VMODE flag is also
asserted to indicate an OPENLED event.
SS
FB2
RC
10k
C6
0.1µF
CC
10nF
D2
C5
4.7µF
RT
VC
INTVCC
85V LED
RT
31.6k
250kHz
design features
The LT3796 solves the problem of power limited,
or constant-current/constant-voltage regulation
by seamlessly combining a current regulation
loop and two voltage regulation loops to control
an external N-channel power switch.
SS
2V/DIV
SS
2V/DIV
LED+
50V/DIV
LED+
50V/DIV
FAULT
10V/DIV
FAULT
10V/DIV
IM2
1A/DIV
IM2
1A/DIV
If there is no resistor between the SS pin
and VREF pin, the converter enters hiccup
mode and periodically retries as shown
in the Figure 2. If a resistor is placed
between VREF and SS pin to hold SS pin
higher than 0.2V during LED short, then
the LT3796 enters latchoff mode with
GATE pin low and TG pin high, as shown
in Figure 3. To exit latchoff mode, the
EN/UVLO pin must be toggled low to high.
5ms/DIV
5ms/DIV
Figure 2. Short LED protection: hiccup mode
(without R11 in Figure 1)
Figure 3. Short LED protection: latchoff mode
(with R11 in Figure 1)
LED DRIVER WITH HIGH PWM
DIMMING RATIO
Once the LED overcurrent is detected, the
GATE pin drives to GND to stop switching, the TG pin is pulled high to disconnect the LED array from the power
path and the FAULT pin is asserted. The
Schottky diode D2 is added to protect
the drain of PMOS M2 from swinging well below ground when shorting
to ground through a long cable. The
PNP helper Q1 is included to further limit
the transient short-circuit current.
Using an input referred LED string
allows the LT3796 to act as a buck
mode controller as shown in Figure 4.
The 1MHz operating frequency enables
a high PWM dimming ratio. The
OPENLED regulation voltage is set to
1.25V •
VIN
16V TO 36V
RLED 250mΩ
1A
Figure 4. A buck
mode LED driver
with 3000:1 PWM
dimming ratio
R3
100k
R1
1M
VIN
R2
100k
VS
VREF
CSN
LT3796
R4
100k
8V
LED
TG
CSP
PWM
PWM
ISN
ISP
EN/UVLO
CTRL
CSOUT
FB1
LED CURRENT REPORTING
C5
0.1µF
R6
124k
L1
10µH
SENSE
INTVCC
RSNS
33mΩ
R9
100k
FAULT
VMODE
GND
FAULT
VMODE
VC
M1: VISHAY SILICONIX Si73430DV
M2: VISHAY SILICONIX Si7113DN
D1: ZETEX ZLLS2000TA
L1: WÜRTH 744066100
LED: CREE XLAMP XM-L
M1
GATE
ISMON
INTVCC
SYNC
RT
RC
10k
CC
4.7nF
RT
6.65k
1MHz
SS
C4
0.1µF
C2
10µF
×4
25V
R5 1M
FB2
R8
100k
through the independent current sense
amplifier at CSP, CSN and CSOUT pins.
During the PWM off phase, the LT3796
disables all internal loads to the VC pin
LED+
M2
C3
4.7µF
R3  R5 
•  + 1
R6  R4 
Figure 5. 3000:1 PWM dimming ratio of the circuit in
Figure 4 at VIN = 24V and PWM = 100Hz
D1
C1
2.2µF
×2
50V
PWM
5V/DIV
IL
1A/DIV
INTVCC
IL
1A/DIV
1µs/DIV
July 2012 : LT Journal of Analog Innovation | 5
Voltage drops in wiring and cables can cause load regulation errors. These errors can
be corrected by adding remote sensing wires, but adding wires is not an option in
some applications. As an alternative, the LT3796 can adjust for wiring drops, regardless
of load current, provided that the parasitic wiring or cable impedance is known.
OUT
•
1:1
C1
10µF
M1
C3
10µF
L1B
+
C4
100µF
25V
RWIRE
VLOAD
12V, 1A CURRENT LIMIT
RSNS
33mΩ
VIN
C8
0.1µF
VREF
GATE
SENSE
GND
EN/UVLO
ISP
ISMON
ISN
PWM
LT3796
SYNC
C5
0.1µF
R7
100k
SS
VS
CTRL
FAULT
FAULT
R4
287k
FB1
VMODE
TG
VC
RT
19.6k
400kHz
and preserves the charge state. It also
turns off the PMOS switch M2 to disconnect the LED string from the power path
and prevent the output capacitor from
discharging. These features combine to
greatly improve the LED current recovery
time when PWM signal goes high. Even
with a 100Hz PWM input signal, this buck
mode LED driver can achieve a 3000:1
dimming ratio as illustrated in Figure 5.
OUT
CSOUT
RT
L1: WÜRTH 744871220
D1: ZETEX ZLLS2000TA
M1: VISHAY SILICONIX Si4840DY
C7
1µF
CSP
R6
100k
VMODE
R2
38.3k
R3
154k
VREF
INTVCC
R1
38.3k
CSN
FB2
INTVCC
RC
24.9k
CC
10nF
R5
12.4k
INTVCC
C6
4.7µF
SEPIC CONVERTER WITH R WIRE
COMPENSATION
Voltage drops in wiring and cables can
cause load regulation errors. These errors
can be corrected by adding remote sensing
wires, but adding wires is not an option in
some applications. As an alternative, the
LT3796 can adjust for wiring drops, regardless of load current, provided that the parasitic wiring or cable impedance is known.
Figure 6 shows a 12V SEPIC converter that
uses the RWIRE compensation feature.
RSNS1 is selected to have 1A load current
6 | July 2012 : LT Journal of Analog Innovation
RSNS1
250mΩ
D1
•
Figure 6. This SEPIC converter
compensates for voltage drops
in the wire between the controller
and the load (RWIRE)
C2
10µF
L1A 22µH
VIN
12V
limit controlled by the ISP, ISN pins. The
resistor network R1–R5, along with the
LT3796’s integrated current sense amplifier
(CSAMP in Figure 7), adjusts the OUT node
voltage (VOUT) to account for voltage
drops with respect to the load current.
This ensures that VLOAD remains constant at 12V throughout the load range.
Figure 7 shows how the LT3796’s internal
CSAMP circuit plays into the operation. The
LT3796’s voltage loop regulates the FB1 pin
at 1.25V so that I3 stays fixed at 100µ A for
R5 = 12.4k. In Figure 7, VOUT changes
design features
The LT3796 in a 28-lead TSSOP package performs
tasks that would otherwise require a number of control
ICs and systems. It offers a reliable power system with
simplicity, reduced cost and small solution size.
ILOAD
R3 2 • (R WIRE )
=
R1
R SNS1
RWIRE
CSN
CSP
–
I1
CSAMP
R4
RWIRE = 0.5Ω
VOUT
800mA
12.5
VLOAD
LT3796
+
R3
12V
R2
R1
13.0
I2
VOUT/VLOAD (V)
VOUT
RSNS1
IOUT
500mA/DIV
12.0
11.5
11.0
VOUT
500mV/DIV
(AC-COUPLED)
10.5
10.0
CSOUT
FB1 = 1.25V
R5
200mA
0
200
400
600
800
1000
1200
500µs/DIV
ILOAD (mA)
I3
Figure 7. RWIRE voltage drops are compensated for
via the LT3796’s CSAMP circuit
Figure 8. Measured VLOAD and VOUT with respect to
ILOAD
Figure 9. Load step response of the circuit in
Figure 6
with current I2 as VOUT = 1.25V + I2 • R4.
If the change of I2 • R4 can offset
the change of ILOAD • (RSNS1 + RWIRE),
then VLOAD will stay constant.
output current is what gives VOUT the
positive load regulation characteristic.
The positive load regulation is just what is
needed to compensate for the cable drop.
SOLAR PANEL BATTERY CHARGER
Referring to Figure 7, the divider
R1/R3 from VOUT sets the voltage
regulated at CSP by the current I1 flowing in R2. I1 is conveyed to the FB1
node where it sums with I2 .
The measured VLOAD and VOUT with respect
to ILOAD are shown in Figure 8. Clearly,
VLOAD is independent of ILOAD when
ILOAD is less than the 1A current limit.
When ILOAD approaches 1A, the current
loop at ISP and ISN pins begins to interfere
with the voltage loop and drags the output
voltage down correspondingly. The load
transient response is shown in Figure 9.
As the output current increases, I1
decreases due to the increasing voltage
drop across RSNS ; its decrease must be
compensated by a matching increase in
the current I2 to maintain the constant
100µ A into FB2. This increase in I2 with
Solar powered devices rely on a highly
variable energy source, so for a device to
be useful at all times, energy from solar
cells must be stored in a rechargeable
battery. Solar panels have a maximum
power point, a relatively fixed voltage at
which the panel can produce the most
power. Maximum power point tracking
(MPPT) is usually achieved by limiting a
converter’s output current to keep the
panel voltage from straying from this
value. The LT3796’s unique combination
of current and voltage loops make it an
ideal MPPT battery charger solution.
July 2012 : LT Journal of Analog Innovation | 7
WÜRTH SOLAR PANEL
VOC = 37.5V
VMPP = 28V
C6
2.2µF 100V
OUT
BAT
RSNS1
250mΩ
•
D1
D2
15V
1:1
C1
4.7µF
50V
•
VIN
L1A 33µH
M2
R4
301k
INTVCC
R1
10k
C2
10µF
L1B
R10
30.1k
R9
10k
NTC
VCHARGE = 14.6V
VFLOAT = 13.5V
AT 25°C
+
BAT
R5
137k
R2
475k
VIN
VS
CSN
CSP
M1
GATE
EN/UVLO
R3
20k
SENSE
RSNS
15mΩ
CTRL
CSOUT
C3
0.1µF
GND
CSOUT
R6
100k
R11
93.1k
FB1
FB2
PWM
LT3796
VREF
ISP
OUT
ISN
BAT
R8
113k
M3
ISMON
C6
0.1µF
SS
C4
0.1µF
VMODE
SYNC
TG
M1: VISHAY SILICONIX Si7456DP
M2: VISHAY SUD19P06-60-E3
M3: ZETEX ZXM61N03F
L1: COILCRAFT MSD1260-333
D1: ON SEMI MBRS260T3G
D2: CENTRAL SEMI CMDZ15L
R9: MURATA NCP18XH103F03RB
R12
10.2k
INTVCC
VC
RT
RC
499Ω
CC
22nF
FAULT
RT
19.6k
400kHz
VMODE
R7
49.9k
C5
4.7µF
INTVCC
R13
49.9k
FAULT
Figure 10. A solar panel battery charger maximum power point tracking (MPPT)
Figure 10 shows a solar panel to sealed
lead acid (SLA) battery charger driven by
the LT3796. The charger uses a three-stage
charging scheme. The first stage is a constant current charge. Once the battery is
charged up to 14.35V, the charging current
The charging current is programmed
by the resistor network at the CSP and
CSOUT (CTRL) pins as follows,
1.2
1.0
V −V

V
VCTRL = R6 •  IN INTVCC − INTVCC  ,

R4
R5 
 R4 
FOR VIN ≥ VINTVCC 1+ 
 R5 
VCTRL = 0 V,
ICHARGE (A)
0.8
0.6
0.4
 R4 
FOR VIN < VINTVCC 1+ 
 R5 
0.2
0
begins to decrease. Finally, when the
required battery charge current falls below
100m A, the built-in C/10 termination
disables the charge circuit by pulling down
VMODE, and the charger enters float charge
stage with VFLOAT = 13.5V to compensate
for the loss caused by self-discharge.
20
25
30
35
VIN (V)
Figure 11. ICHARGE vs VIN for the solar
charger in Figure 10
8 | July 2012 : LT Journal of Analog Innovation
40
Maximum power point tracking is
implemented by controlling the maximum output charge current. Charge
current is reduced as the voltage on
the solar panel output falls toward
28V, which corresponds to 1.1V on the
CTRL pin and full charging current, as
shown in Figure 11. This servo loop thus
acts to dynamically reduce the power
requirements of the charger system to
the maximum power that the panel
can provide, maintaining solar panel
power utilization close to 100%.
SUPERCAPACITOR CHARGER WITH
INPUT CURRENT LIMIT
Supercapacitors are rapidly replacing
batteries in a number of applications
from rapid-charge power cells for cordless tools to short term backup systems
for microprocessors. Supercapacitors are
longer lasting, greener, higher performance
and less expensive over the long run, but
charging supercapacitors requires precise
control of charging current and voltage
design features
RSNS1 150mΩ
1.33A MAX
VS
L1B
CSP
CSN
EN/UVLO
OUTPUT CURRENT REPORTING
GATE
ISMON
PWM
VREF
INPUT CURRENT
REPORTING AND LIMIT
CSOUT
SYNC
R2
124k
C4
0.1µF
VOUT = 0V TO 28V
D1
R1
20k
VIN
C3
0.1µF
C6
10µF
•
C1
10µF
C7
0.1µF
L1A 33µH
•
VIN
28V
C2
4.7µF
×2
50V
R8
536k
R9
24.9k
M1
SENSE
RSNS
33mΩ
LT3796
1.67A
MAX
GND
CSOUT
FB1
FB2
ISP
RSNS2
150mΩ
SS
ISN
SUPERCAP
TG
INTVCC
R7
100k
R6
100k
INTVCC
FAULT
FAULT
CHGDONE
C6
4.7µF
VMODE
CTRL VREF
RT
VC
RC
499Ω
VOUT
L1: COILCRAFT MSD1260-333
D1: ON SEMI MBRS260T3G
M1: VISHAY SILICONIX Si7850
Q1: ZETEX FMMT591A
R3
499k
C5
0.1µF
R10
499k
R5
1M
R4
30.1k
CC
22nF
Q1
RT
19.6k
400kHz
Figure 12. A 28V/1.67A supercapacitor charger with input current limit
regulation until the input current moves
close to the 1.33A input current limit.
Some applications require that the
input current is limited to prevent the
input supply from crashing. Figure 12
shows a 1.67A supercapacitor charger
with 28V regulated output voltage and
1.33A input current limit. The input
current is sensed by RSNS1, converted
to a voltage signal and fed to the FB2
pin to provide input current limit.
CONCLUSION
In each charging cycle, the supercapacitor is charged from 0V. The feedback loop
from VOUT to the RT pin through R3, C5, R5,
R10, R4, and Q1 to RT works as frequency
foldback to keep regulation under control.
In Figure 13, the input current and output
charging current are plotted against
output voltage for this charger, showing
the LT3796 maintaining the output current
The LT3796 is a versatile step-up
DC/DC controller that combines accurate
current and voltage regulation loops. Its
unique combination of a single current
loop and two voltage loops makes it easy
to solve the problems posed by applications that require multiple control loops,
such as LED drivers, battery or supercapacitor chargers, MPPT solar battery
chargers, and step-up or SEPIC converters
with input and output current limit. It
also includes a number of fault protection and reporting functions, a top gate
driver and current loop reporting.
The LT3796 in a 28-lead TSSOP package performs tasks that would other­
wise require a number of control ICs
and systems. It offers a reliable power
system with simplicity, reduced
cost and small solution size. n
1800
IOUT
1600
INPUT/OUTPUT CURRENT (mA)
limiting to prevent any system-wide damage or damage to the supercapacitor.
1400
1200
IIN
1000
800
600
400
200
0
0
5
10
15
20
25
30
VOUT (V)
Figure 13. Input/output current vs output voltage for
28V/1.67A supercapacitor charger in Figure 12
July 2012 : LT Journal of Analog Innovation | 9
Pushbutton On/Off Controller Includes Optional Automatic
Turn-On When Handheld Device is Plugged In
Vui Min Ho
It is well-known that most mechanical pushbutton
switches bounce when pressed, and that a debounce
circuit is required to produce a clean, usable signal from
the pushbutton. There are many debounce solutions
available—common ones use flip-flops or R-S latches—but
designing and implementing a debounce circuit is not as
trivial as it might seem, especially for handheld devices.
Because a pushbutton debounce circuit
must remain on all the time, a low supply
current is critical for battery-powered
handheld devices. Additionally, the circuit
should be capable of accepting power
from any available standby supply voltage without requiring a linear regulator.
Furthermore, the pushbutton input should
be able to withstand high ESD levels during
operation since it is usually connected
where contact with the human finger is
possible. And finally, the circuit must be
small enough to fit into whatever little
space is left on the printed circuit board.
The LTC2955 pushbutton controller covers
all of these requirements. It generates a
latched enable output from the noisy pushbutton input. The enable output comes in
both active high (LTC2955-1) or active low
(LTC2955-2) options, allowing it to drive
the on/off input of any system or regulator.
The LTC2955 features a voltage monitor pin (ON) that can be used for automatic system turn-on when the device
is plugged into a secondary supply such
as a wall adapter or car battery. This is
a common feature found in handheld
devices where, if you plug in the wall
adapter or charger cable, the device
automatically powers up by itself without a press of the on/off switch.
The LTC2955 is also designed with blanking times after each pushbutton event,
during which all inputs are ignored. This
prevents the EN output from turning on
and off continuously if the pushbutton
is held down or stuck low. These blanking times ensure sufficient time for the
voltage regulator to fully charge up or
discharge its output and allow the system or microprocessor time to perform
power on/off tasks. In addition, the
power-down debounce time is adjustable
using an external capacitor. This allows
the designer to extend the power-down
The LTC2955 is designed to interface
with a microprocessor via the LTC2955’s
INT (interrupt) output pin and KILL input
pin. The LTC2955’s INT output alerts the
microprocessor that the pushbutton is
pressed, allowing the microprocessor to
perform any power-down tasks. Once
these tasks are complete, the microprocessor can communicate—via the
KILL pin—that the system is ready to be
switched off. The user can also force the
Figure 1. Pushbutton on/off control for battery-powered device, with automatic turn-on when the
device is plugged in
LT3060
WALL ADAPTER INPUT
12V
P-CHANNEL
MOSFET
BATTERY INPUT
3.6V
SHDN
LTC4412
VIN
GATE
GND SENSE
VIN
CTL
SEL
STAT
ON
9.6V THRESHOLD
200k
1M
PGD
PB
TMR
LTspice IV
EN
LTC2955-1
2.21M
circuits.linear.com/559
10 | July 2012 : LT Journal of Analog Innovation
system to power-down if the microprocessor fails to respond to the interrupt
signal (KILL pin remains high). This is
the familiar user-holds-the-button-down
for a duration greater than the defined
power-off period. The power-off period
is adjustable through the capacitor at the
TMR pin and it can be made as long as
required to prevent accidental turn-off.
INT
GND
KILL
µP
design features
The LTC2955 pushbutton controller generates a latched enable
output from the noisy pushbutton input. The enable output
comes in both active high or active low options, allowing
it to drive the on/off input of any system or regulator.
time in cases where some systems need
more time to perform power-down tasks.
MOSFET to reduce the voltage drop across
TOP OFto
BOARD
the diode connected
the 3.6V supply.
The LTC2955 minimizes components by
operating directly from inputs as low as
a 1.5V single-cell battery up to a 36V multicell stack—with no additional boost
regulator or LDO required. The low quiescent current of 1.2μA extends the battery
life. The device is available in a spacesaving 10-lead 3mm × 2mm DFN package and 8-lead ThinSOT™ package.
The LTC2955-1 ON pin monitors the
12V input through the resistive divider R1
and R2. When the user plugs in the wall
adapter, the 12V supply becomes present.
The LTC2955-1 detects that the ON pin is
high and pulls the EN (enable) pin high
after a 32ms debounce time and turns on
the voltage regulator, applying power to
the system. This allows automatic system
turn-on when the user plugs in the wall
adapter. The system can also be turned on
by pressing the pushbutton. The LTC2955
alerts the microprocessor that the 12V supply is present or absent by pulling the
PGD output pin high or low, respectively.
HANDHELD WITH AUTOMATIC
TURN-ON WHEN PLUGGED IN
Figure 1 shows a typical LTC2955-1
application for a handheld device. The
3.6V supply is produced directly from the
handheld’s battery; the 12V secondary
supply comes from a wall adapter. Both
the 3.6V and 12V inputs are connected
to the LT3060 regulator input via diodeOR so that either supply can power the
system. The LTC4412 is an ideal diode
controller that controls the P-channel
PUSHBUTTON PIN ESD PROTECTION
The LTC2955 PB (pushbutton) input
is protected to ESD levels of up to
±25kV HBM with respect to ground. This
protection level exists during all modes
Figure 2. LTC2955-1 waveforms
TURN ON PULSE
SHORT PULSE
VERSATILE PUSHBUTTON INPUT
The LTC2955 requires only a few external components for most applications,
as shown in Figure 1. Except for the
logic-level pins used to interface with
the microprocessor, most of the pins can
withstand a maximum voltage of 36V,
precluding the need for external supplies or resistor dividers. Designs remain
flexible in the face of high input supply voltages, especially when no typical
board-level supply (e.g., 5V) is available.
The PB input is designed to operate in
harsh and noisy environments. The pin
Figure 3. Pushbutton input
LONG PULSE
LTC2955
EXTERNAL
SUPPLY
PB
EN
of operation including power-down,
BOTTOM
BOARD is dispower-up or when
theOFsupply
connected from the battery. When the
pushbutton pin is hit with an ESD strike
during operation, the part remains in
its current logic state. The device does
not reset or latch up and there is no
need to cycle the supply to recover.
TURNS ON
STAYS ON
TURNS OFF
VLDO
OPTIONAL
PULLUP
RESISTOR
D1
900k
PB
0.8V
–
+
INT
INTERRUPT
INTERRUPT
July 2012 : LT Journal of Analog Innovation | 11
The PB input is designed to operate in harsh and noisy environments.
The pin can withstand both positive and negative voltages up to ±36V.
This allows for long cable runs between the pushbutton switch and the
LTC2955, where even if the input rings, it will not cause damage to the part.
VIN
LTC2955-1
VLDO
LTC2955-2
2µA
D2
EN
VOUT
OPTIONAL
PULLUP
RESISTOR
VIN
M1
EXTERNAL
SUPPLY
OPTIONAL VIN
VOUT
PULLUP
VOLTAGE
RESISTOR
REGULATOR
EN
CONTROL
900k
EN
SHDN
EN
CONTROL
LTC2955-2
Figure 4. LTC2955-1 EN output
can withstand both positive and negative
voltages up to ±36V. This allows for long
cable runs between the pushbutton switch
and the LTC2955, where even if the input
rings, it will not cause damage to the part.
Figure 3 shows the pushbutton pin connection and internal circuitry. The internal
900k pull-up resistor allows connection
of the pin directly to the pushbutton
switch (with the other terminal grounded)
without requiring an external pull-up
resistor. If an external pull-up resistor is
desired in applications where the pushbutton switch is leaky, this optional pull-up
resistor can be tied to any voltage up to
the 36V maximum as shown. The internal
diode D1 blocks the external supply current from flowing into the device, preventing unnecessary current consumption.
12 | July 2012 : LT Journal of Analog Innovation
Figure 5. LTC2955-2 EN output
VERSATILE ENABLE OUTPUT
Figure 4 shows the LTC2955-1 active
high EN pin driving the SHUTDOWN input
of a voltage regulator. The LTC2955-1
EN pin pulls high to 4.3V with an internal
2µ A pull-up current in active mode. If a
higher VOH voltage is required, an optional
external pull-up resistor can be added
to pull this pin above 4.3V as shown.
The diode D2 blocks the external supply current from flowing into the device.
The EN pin can be pulled high up to 36V.
Figure 5 shows the LTC2955-2 active low
EN pin driving a P-channel MOSFET to
control the system supply. The LTC2955-2
EN pin pulls high through an internal 900k
resistor during the inactive mode. In the
active mode with the pin low, this 900k
resistor is disconnected from the supply to
minimize the quiescent current consumed
by the 900k resistor. If a VOH lower than
the supply voltage is required, this pin
can be tied to the external supply through
an optional pull-up resistor as shown.
The ON input and SEL inputs can withstand
voltages up to 36V. This makes it easy to
connect these pins directly to the high voltage supply without requiring a resistive
divider, and thus minimize the quiescent
current consumed by the resistive divider.
CONCLUSION
The LTC2955 is a family of micropower
(1.2µ A), wide input voltage range (1.5V to
36V) pushbutton controllers. These parts
lower system cost and preserve battery life by integrating a rugged pushbutton input, a flexible enable output
and a simple microprocessor interface
that provides intelligent power-up and
power-down. The device is available
in space-saving 10-lead 3mm × 2mm
DFN and 8-lead ThinSOT packages. n
design features
I2C-Controlled Li-Ion Power Management IC with
Integrated Power Devices Charges High Capacity Batteries
from Any 5V Source While Keeping Cool
David Simmons
Designers of portable electronics are challenged to create
devices that do everything while running endlessly on a
single battery charge. While it is impossible to fully meet
this challenge, each successive generation of batteries at
least comes closer to that goal. With devices now sporting
large vibrant touch-sensitive displays, multicore CPU
and graphics processors, and an assortment of wireless
modems for high speed communications anywhere on
the planet, high battery capacity is essential. Battery
manufacturers have met the demand with light weight,
compact cells with capacities to over 30 watt-hours.
While USB has become the dominant standard for device interconnect, synchronization and data exchange, its power delivery
capabilities have not kept pace with battery demands. USB 2.0 allows a maximum
2.5W load, while USB 3.0 extends the limit
to 4.5W. Even with perfect efficiency and
all power going directly to the battery,
a full charge cycle via USB would require
overnight and then some. Though USB is
not suitable as a primary power source for
large capacity batteries, it still has great
value as an opportunistic power source
to charge when and where possible, and
to prevent battery drain when the device
is tethered to a traditional computer.
Figure 1. I2C controlled high power battery charger/USB power manager
L1
1µH
WALLSNS
WALLGT
5V INPUT
VOUTSNS
VOUT
VBUS
10µF
TO µC
TO µC
3
CHGSNS
LTC4155
USBGT
3.6k
TO
SYSTEM
22µF LOAD
SW
BATGATE
USBSNS
MP1
BATSNS
NTCBIAS
ID
I2C
100k
IRQ
OVGCAP
CLPROG1 CLPROG2 GND VC
1.21k
NTC
2.4A
PROG
0.047µF
499Ω
L1: COILCRAFT XFL4020-102ME
MP1: VISHAY Si5481DU-T1-GE3
Li-ION
BATTERY
THE BEST OF BOTH WORLDS
The LTC4155 is a monolithic switching
battery charger that delivers 3.5A charge
current efficiently in a compact PCB footprint. Figure 1 shows the required components in a typical application. The
2.25MHz switching frequency allows for
a small inductor and bypass capacitors
to minimize the overall PCB footprint.
High efficiency (Figure 2) even at multiamp charge rates is critical not only to
make optimal use of available input
power, but also to control power dissipation inside the portable device. The
combination of high power dissipation and marginal thermal performance
in a tightly enclosed space can make
a device with a less efficient charging
solution too warm to hold comfortably.
To help keep things cool, the LTC4155’s
integrated power switches feature an
on-resistance well under 100mΩ.
While the LTC4155’s power switches are
sized to handle higher currents than
available from USB, the LTC4155 remains
fully USB compatible for opportunistic
charging. Input current is automatically
measured internally and limited to any
of sixteen I2C user-selectable values. Of
these settings, three correspond to guaranteed maximum limits of 100m A and
500m A for USB 2.0 and 900m A for USB 3.0.
Automatic input current limiting can also
be used with AC adapters or other sources
by choosing any of the other current
limit settings up to a maximum of 3A.
July 2012 : LT Journal of Analog Innovation | 13
High efficiency, even at multi-amp charge rates, is critical
not only to make optimal use of available input power, but
also to control power dissipation inside the portable device.
To help keep things cool the LTC4155’s integrated power
switches feature an on resistance well under 100mΩ.
100
1.75
90
1.50
70
LOAD CURRENT (A)
EFFICIENCY (%)
80
60
50
40
30
20
The LTC4155 supports a pin-programmable
power-on default input current. For high
power applications that do not require
USB compatibility, a single resistor connected to the CLPROG1 pin programs a
default power-on input current. This
resistor is chosen to correspond to an
initial current limit most appropriate to
the particular application, intended power
source capability, etc. After power-up,
the input current limit can be modified
under I2C control to any of the sixteen
other available settings up to 3A.
For USB applications, the CLPROG1 and
CLPROG2 pins can be connected together
to program the LTC4155 to enforce
USB current limit rules. The input current
limit will default to 100m A upon application of external power. After successful
enumeration with the USB host controller, the input current limit setting can be
increased under I2C control to 500m A or
900m A as appropriate. Figure 3 shows
the available current to the system load
14 | July 2012 : LT Journal of Analog Innovation
900mA MODE
1.25
1.00
500mA MODE
0.75
10
0
VFLOAT = 4.05V
VBAT = 3.9V
0
0.5
1.0 1.5 2.0
2.5
LOAD CURRENT (A)
3.0
3.5
0.50
2.4
2.7
3.0
3.3
3.6
BATTERY VOLTAGE (V)
3.9
4.2
Figure 2. Switching regulator efficiency
Figure 3. USB-compliant load current available
before discharging battery
and battery charger. Note that the switching regulator output current is higher
than the USB-limited input current. If the
system detects that the power source is
an AC adapter, dedicated USB charger,
or other non-USB source, the input current limit setting can be increased under
I2C control to any other setting up to 3A.
is removed and a 5V/3A AC adapter is
connected to the same port, the input
source priority can be modified over I2C to
switch to the new higher power source.
SEAMLESS HANDLING OF MULTIPLE
INPUT CONNECTORS
The LTC4155 optionally accepts input from
two power sources, solving the challenge of intelligently routing power from
two different physical connectors to the
product. When both input sources are
connected simultaneously, the decision of
which source to use is based on a userprogrammable priority. As long as each
input voltage is within the valid operating
range, either one may be selected without concern for which voltage is higher
than the other. This allows, for instance,
a 4.5V/2A AC adapter to be favored over a
5V/500m A USB port. If the USB connection
The LTC4155 supports independent
I2C programmable input current limits for each of its two power inputs.
When the higher priority input source
is disconnected, charging can continue
uninterrupted, with automatic reduction to the new lower maximum input
current limit. No immediate attention is
required from the system microcontroller.
Depending on the external components selected for the input multiplexer,
overvoltage and reverse voltage protection up to ±77V can be easily implemented if required for the application.
Additionally, the LTC4155 can produce
a USB On-The-Go 5V current-limited
supply to the USB connector using no
additional external components.
design features
The LTC4155 optionally accepts input from two power sources,
solving the challenge of intelligently routing power from two
different physical connectors into the product. When both input
sources are connected simultaneously, the decision of which
source to use is based on a user-programmable priority.
4.4
84
4.3
72
HOT FAULT
48
TOO WARM
36
NOMINAL
24
IVOUT = 0A, VFLOAT = 4.2V
100% CHARGE
CURRENT MODE
50% CHARGE
CURRENT MODE
12.5% CHARGE
CURRENT MODE
CHARGER DISABLED
4.2
VOUT VOLTAGE (V)
TEMPERATURE (°C)
60
12
TOO COLD
0
4.1
4.0
3.9
3.8
3.7
3.6
–12
3.5
–24
3.4
0
16
32
48 64 80
ADC CODE
96
112
2.4
2.7
3.0
3.3
3.6
BATTERY VOLTAGE (V)
3.9
4.2
Figure 4. Transfer function of the LTC4155 battery
temperature data converter, with the autonomous
charger cut-out temperature thresholds highlighted.
Figure 5. VOUT voltage vs battery voltage
EXTENSIVE PROGRAMMABILITY
AND TELEMETRY FOR ADVANCED
CHARGING ALGORITHMS
Continuous battery temperature data is
available to system software to dynamically adapt system or charger behavior
to manage extreme operational corners. For instance, float voltage and/or
charge current may be reduced under
I2C control to increase the battery safety
margin at high ambient temperatures.
Similarly, charge current or total system
load current can be reduced in response
to high temperature to reduce additional
heating within the product enclosure.
The LTC4155 provides continuous I2C status reporting, allowing system software
to have a complete view of the state of
input power sources, fault conditions,
battery charge cycle state, battery temperature, and several other parameters.
Key charge parameters can be changed
under I2C control to implement customized
charge algorithms. Unlike microcontrollerbased or other programmable charge
algorithms, all possible LTC4155 settings
available under software I2C control are
intrinsically safe for the battery. Float
voltage can never be programmed above
4.2V or below 4.05V. Similarly, battery
charge current is programmble to one
of 15 possible settings, but software may
never increase the limit above the level
set by the designer—via a programming
resistor chosen to match the battery
capacity and maximum charge rate.
Like all other aspects of battery charger
programmability, the LTC4155 implements an intrinsically safe charging
solution without (or despite) any software intervention. Battery charging is
always paused when the cell temperature falls below 0°C or rises above 40°C.
Additionally, a fault interrupt may be
optionally generated whenever cell
temperature rises above 60°C. Figure 4
shows the transfer function of the LTC4155
battery temperature data converter,
with the autonomous charger cut-out
temperature thresholds highlighted.
POWERPATH INSTANT-ON
OPERATION
Dead batteries can be especially troublesome in a traditional power architecture
where most of the portable product is
connected directly to the battery. When
the battery voltage is too low for the
system to run, the product may appear
to be unresponsive even minutes after
being connected to a source of input
power—possibly generating unnecessary support phone calls. The problem
is further compounded when the battery
capacity is very large relative to the available charging current (e.g., a USB-powered
system with a large capacity battery).
Linear Technology PowerPath™ products such as the LTC4155 decouple the
system power rail from the battery to
enable instant-on operation and solve
July 2012 : LT Journal of Analog Innovation | 15
The LTC4155 features automatic reduction of input current when the input voltage
begins to drop to an unacceptable level. At high charge current levels, this can happen
when connections are made through undersized wire, to an undersized adapter, through
connectors with mild corrosion, or other conditions outside the usual design envelope.
the two most vexing problems caused
by deeply discharged batteries.
The first problem is that charge current
and system load become indistinguishable when the system power rail is
connected directly to the battery. When
the battery is deeply discharged, battery manufacturers recommend a greatly
reduced initial charge current until the cell
voltage reaches a safer level. This trickle
charge current must be programmed
to a safe level for the battery assuming
minimum or no system load current.
Secondly, in a direct-connect battery
system, if the system is operational during trickle charging, a significant portion
of the charge current intended for the
battery is shunted to the system rail. The
resulting reduced battery charge current
extends recovery time proportionately.
A sizable system load can cause the net
battery current to reverse, further discharging the battery. For the duration of
this low battery condition, the portable
system may not be able to respond to
the user due to insufficient voltage on
the system power supply rail. The duration of unresponsiveness is multiplied
by at least a factor of 10 because of the
reduced power available to the commonconnected battery and system power rail.
The LTC4155 delivers 3.5V to the system
rail when the battery is deeply discharged
to enable instant start-up. As the battery voltage rises during the precharging phase, the LTC4155 seamlessly and
automatically transitions to a higher
efficiency mode to speed charging and
16 | July 2012 : LT Journal of Analog Innovation
minimize heat production. Figure 5 shows
the voltage available to the system power
rail as a function of battery voltage.
The LTC4155 battery charge current is
programmed independently from the input
current limit to decouple battery charge
current constraints from input power
constraints. The input current limit can
be programmed based only on the limitations of the input supply. Similarly, the
battery charge current can be programmed
based only on the battery capacity. The
LTC4155 always enforces input current
limit and prioritizes power to the system
load over battery charging if necessary.
ROBUST IN THE FACE OF NON-IDEAL
SOURCES
The LTC4155 features automatic reduction
of input current when the input voltage begins to drop to an unacceptable
level. At high charge current levels, this
can happen when connections are made
through undersized wire, to an undersized adapter, through connectors with
mild corrosion, or any number of conditions outside the usual design envelope.
Without intervention, the input voltage to the IC would continue to drop,
eventually falling below the undervoltage lockout threshold. The IC would then
shut down, allowing the input voltage to
recover and restart the whole cycle. The
LTC4155 makes the best of a bad situation. As the input voltage falls to 4.3V,
the LTC4155 smoothly reduces its input
power by whatever amount is necessary to
prevent further decay of the input voltage. In this mode the current delivered to
the system load and battery is less than
the programmed amount, but more than
would be available if the input voltage
oscillation were allowed to continue.
Additionally, the LTC4155 produces an
I2C status report and optional interrupt
signal to alert the system that corrective or diagnostic action may need to be
undertaken by the end user to restore
maximum charge current capability.
CONCLUSION
The LTC4155 combines high current capability and efficiency with a small monolithic PCB footprint, ideal for portable
devices with large lithium batteries where
board space is at a premium, and heat and
charge time are the enemy. USB-compatible
input current limit settings further extend
versatility to allow opportunistic charging from ubiquitous but lower power
sources. Extensive telemetry allows for
custom behavior based on changing
environmental or application conditions
without compromising autonomous
battery safety. Uninterrupted power is
delivered to the system rail despite common problems such as a deeply discharged
battery or a resistive undersized input
power cable. The LTC4155 is available in
a 28-lead 4mm × 5mm QFN package. n
design features
15V, 2.5A Monolithic Buck-Boost DC/DC Converter with
95% Efficiency and Low Noise Operation
Eddy Wells
Power-hungry handheld devices and industrial instruments often require multicell or high
capacity batteries to support their ever-increasing processing needs. A wide voltage range,
high efficiency buck-boost DC/DC converter is the ideal solution for longer battery run
times and handling multiple input sources. The LTC3112 is a 2.2V to 15V input capable
2.5A buck-boost converter. The extended voltage range allows conversion from a variety of
power sources such as one, two or three Li-ion cells, lead acid batteries, supercapacitors,
USB cables and wall adapters to output voltages programmed between 2.5V and 14V.
The LTC3112 features the latest generation
buck-boost PWM control scheme, effectively eliminating jitter when crossing the
barrier between buck and boost operation. Safeguards such as current limit,
overvoltage protection, thermal shutdown,
and short-circuit protection provide
robust operation in harsh environments.
For demanding applications where component size or conversion efficiency is critical, the LTC3112’s 750kHz default switching
frequency can be synchronized between
300kHz and 1.5MHz. For designs where
output current needs to be controlled
or measured, an output current monitor
pin is available. Selectable Burst Mode®
operation extends the operating life when
the battery-powered device is idle.
The LTC3112-based converter shown in
Figure 1 can generate 30W of power with
a 12V output. The solution footprint
is less than 200mm2, which cannot be
matched by a controller-based buck-boost
or complex dual-inductor SEPIC design
at similar power levels. The main external components are limited to the input
and output filter caps and the power
inductor. The LTC3112 is offered in a
thermally enhanced 16-lead 4mm × 5mm
DFN or 20-lead TSSOP package.
extends high efficiency operation for
more than two decades of load current.
Figure 1. LTC3112 based 30W solution
OPERATION FROM MULTIPLE INPUT
SOURCES
The LTC3112’s wide operating range allows
devices to be powered from multiple input
sources. Figure 2 shows an application
where the LTC4412 PowerPath controller (TSOT-23 package) provides a low loss
selection between two input sources. The
LTC4412 maintains a 20mV forward voltage
across the selected P-channel MOSFET, keeping losses to a minimum. In this circuit,
the LTC4412 automatically switches the
greater of a single Li-ion cell or 12V wall
adapter to the input of the LTC3112.
Efficiency curves based on the two
input sources are given in Figure 3. Peak
efficiencies of greater than 90% are
achieved with either input. Selectable
Burst Mode operation (dashed lines)
with 50µ A of typical sleep current
A feedforward network (CFF, RFF of
Figure 2) connected between the VIN and
FB pins provides improved transient
response when the wall adapter voltage is
applied. Feedforward values were selected
by first measuring the voltage change in
voltage at COMP as VIN transitions from
3.6V to 12V. A 380mV change at COMP was
observed, optimal values for VIN and
RFF can now be calculated as follows:
CFF =
∆VCOMP
• (CFB + CP ) = 33pF
∆VIN
RFF =
RFB • CFB
= 681k
CFF
VOUT regulation is maintained within
300mV or 6% during the 15µs transition with a 47µF output cap (Figure 4)
and 500m A load. A falling VIN edge is
about 10-times slower, resulting in
an even smaller transient.
A 3.6V input, 5V output load step response
using the compensation components of
Figure 2 is shown in Figure 5. In this
case, a 250m A to 1A load step results in
only a 250mV transient on VOUT with a
47µF output capacitor. Figures 4 and
5 illustrate how the LTC3112’s loop
July 2012 : LT Journal of Analog Innovation | 17
The LTC3112 features the latest generation buck-boost
PWM control scheme, effectively eliminating jitter when
crossing the barrier between buck and boost operation.
AUXILIARY
P-CHANNEL
MOSFET
12V
WALL
ADAPTER
CFF
33pF
RFF
681k
0.1µF
VIN
Figure 2. LTC4412
PowerPath controller
selects highest voltage
input to power the
LTC3112 converter
BURST PWM
GATE
CTL
STAT
response can be configured to provide
excellent response to both input voltage and output current load steps.
SW2
BST1
VIN
BST2
VOUT
LTC3112
47µF
OFF ON
1µF
470k
PWM/SYNC
COMP
IOUT
GND
OVP
To protect data, some data systems require
a short period of time to backup data
when the primary power source fails. A
bank of supercapacitors is often used to
provide the required burst of energy. The
LTC3112’s wide input voltage range and
In this circuit, a stack of supercapacitors
totaling 22mF is charged to 15V while
the primary power source is active. A
lower ESR electrolytic or ceramic cap is
placed in parallel to minimize VIN ripple.
The VCC supply pin is back-driven from
the 5V output in this example, allowing the LTC3112’s gate drive circuits to
Figure 3. 5V output efficiency from a single Li-ion
cell (3.6V) or wall plug (12V)
Figure 4. 3.6V to 12V input step and resulting VOUT
transient
CFB
680pF
FB
RUN
ability to buck or boost make it ideal for
such an application, as shown in Figure 6.
5V BACKUP SUPPLY
0.1µF
22pF
LTC4412
VIN
SENSE
GND
SW1
VCC
PRIMARY
P-CHANNEL
MOSFET
Li-ION
BATTERY
CELL
4.7µH
845k
RFB
33k
47pF
10k
TO ADC
1V PER AMP
100pF
42.2k
VIN
5V/DIV
EFFICIENCY (%)
90
operate efficiently with an input voltage from 15V down to 2.2V. Available
energy at the input is given by:
1
2
2

EIN = • CIN • ( VINITIAL ) − ( VFINAL ) 


2
22mF  2
2
=
• 15 − 2.2 


2
= 2.4J
The results of the backup event are
shown in Figure 7. A resistive network
from VIN, VOUT and GND is used to drive
Figure 5. 250mA to 1A load step and resulting VOUT
transient
12V
3.6V
Burst Mode Operation
85
75
70
VOUT
500mV/DIV
VOUT
1V/DIV
80
VIN = 3.6V
VIN = 12V
0.1
1
10
100
ILOAD (mA)
1A
18 | July 2012 : LT Journal of Analog Innovation
10A
IL
1A/DIV
IL
1A/DIV
20µs/DIV
47µF
158k
95
PWM
VOUT
5V
1.5A
200µs/DIV
design features
The LTC3112’s ability to support large load currents make it ideal
for handheld devices with increased processing power. Solution
size and conversion efficiency benefit from 50mΩ N-channel
MOSFET switches and thermally enhanced packages.
VOUT
4.7µH
499k
BAT54
0.1µF
VIN
15V TO 2.2V
+
22mF
SUPERCAP
STACK
+
+
= 250mA • 5V • 1.7s
= 2.1J
The prior example can be easily scaled
depending on the voltage rating of the
Figure 7. Supercap discharge performance during
power supply backup event
BST2
VOUT
LTC3112
PWM/SYNC
499k
RUN
5V/DIV
ILOAD
500mA/DIV
500ms/DIV
680pF
VOUT
5V/500mA
33k
845k
IOUT
GND
OVP
1µF
Figure 8. Maximum output current versus VIN with
VOUT = 5V and VCC back-fed
4
3.5
TO ADC
1V PER AMP
100pF
47µF
47pF
FB
RUN
supercapacitors and the energy required
for backup. The IOUT pin (Figure 6) can
be monitored by an ADC to measure
load current during the backup event.
An important consideration in design is
the maximum output current capability
of the buck-boost converter. As shown
in Figure 8, the LTC3112 is able to support up to 4A of load current when
VIN >> VOUT. As the converter transitions
from buck to boost mode, the maximum load current drops accordingly.
MAXIMUM OUTPUT CURRENT (A)
VOUT
5V/DIV
0.1µF
22pF
4.5
VIN
5V/DIV
COMP
220µF
Figure 6. Backup Supply for 5V
rail runs down to VIN = 2.2V
EOUT = IOUT • VOUT • t
SW2
BST1
VIN
VCC
1M
+
the RUN pin to provide a clean shutdown
of VOUT. In this example, a constant
250m A load is drawn from the LTC3112
resulting in the VIN capacitors maintaining regulation for 1.7 seconds, and an
average conversion efficiency of 88%
including the supercapacitor losses.
SW1
42.2k
10k
158k
SUMMARY
The LTC3112 produces low noise buckboost conversion in a range of applications requiring an extended input or
output voltage range. The LTC3112’s
ability to support large load currents
make it ideal for handheld devices with
increased processing power. Solution size
and conversion efficiency benefit from
50mΩ N-Channel MOSFET switches and
thermally enhanced packages. To provide longer run times, a low Burst Mode
quiescent current extends high efficiency
over several decades of load current.
Features such as synchronized switching
frequency, programmable output voltage, a load current monitor and external
loop compensation allow the LTC3112 to
be tailored for a specific application. n
3
2.5
2
1.5
1
0.5
0
2 3 4 5 6 7 8 9 10 11 12 13 14 15
VIN (V)
July 2012 : LT Journal of Analog Innovation | 19
Novel Current-Sharing IC Balances Two Supplies with Ease
Pinkesh Sachdev
Failure is not an option. That’s the likely motto for the architects of today’s alwaysup electrical infrastructure—think telecommunications networks, the Internet and
the electrical grid. The problem is that the bricks of this infrastructure, from the
humble capacitor to the brainy blade-servers, have a limited lifetime usually ending
at the most Murphy of moments. The usual workaround to the mortality problem is
redundancy—backup systems ready to take over whenever a critical component fails.
For instance, high availability computer
servers typically ship with two similar
DC supplies feeding power to each individual board. Each supply is capable of
taking on the entire load by itself, with
the two supplies diode-ORed together
via power diodes to create a single 1 + 1
redundant supply. That is, the higher
voltage supply delivers power to the load,
while the other supply idly stands by. If the
active supply voltage drops or disappears,
due to failure or removal, the once lowervoltage supply becomes the higher voltage
supply, so it takes over the load. The
diodes prevent back-feeding and crossconduction between supplies while protecting the system from a supply failure.
5A
12.2V
NC
•Supply lifetimes are extended if each
takes on half the load, spreading the
supply heat and reducing thermal
stresses on supply components. A rule
of thumb for the lifetime of electronics
is that the failure rate of components
halves for every 10°C fall in temperature.
That’s a significant dependability gain.
+ 325mV –
SUM85N03-06P
39nF*
EN1 CPO1
0.1µF
The diode-OR is a simple winner-takeall system where the highest voltage
supply sources the entire load current.
The lower voltage supply remains idle
until called into action. Although easy
to implement, the 1 + 1 solution is inefficient, wasting resources that could be
better used to improve overall operating efficiency and lifetime. It is far better for the supplies to share the load in
tandem, offering several advantages:
VIN1
GATE1
OUT1
VCC
LTC4370
GND
RANGE
FETON1
2mΩ
FETON2
2mΩ
OUT2
EN2 CPO2
VIN2
*OPTIONAL, FOR FAST TURN-ON
10A
11.875V
GATE2 COMP
39nF*
11.9V
11.875V
0.18µF
SUM85N03-06P
+ 25mV –
5A
20 | July 2012 : LT Journal of Analog Innovation
Figure 1. The LTC4370 balancing
a 10A load current between two
diode-ORed 12V supplies. Sharing
is achieved by modulating the
MOSFET voltage drops to offset the
mismatch in the supply voltages.
•Because the lower voltage supply is
always operational, there is no surprise
when transitioning to a backup supply
that might have already silently failed—
a possibility in a simple diode-OR system.
•It is possible in a load-sharing system to parallel smaller at-hand
supplies to build a larger one.
•The recovery dynamics on supply
failure are smoother and faster, since
the supply changes are on the order
of less and more, not off and on.
•A DC/DC converter formed by two supplies running at half capacity has better
overall conversion efficiency than a
single supply running near full capacity.
METHODS OF CURRENT SHARING
Connecting the outputs of multiple
power supplies allows them to share
a common load current. The division
of the load current among the supplies
depends on the individual supply output voltages and supply path resistances
to the common load. This is known as
droop sharing. To prevent back-feeding
of a supply and to isolate the system
from a faulting supply, diodes can be
inserted in series with each supply. Of
course, this added diode voltage drop
affects the balance of the load sharing.
design features
The LTC4370 introduces a new paradigm for current sharing, where the contributions
from individual supplies are under full active control, but no share bus, with its extra
wires, is required. Complete control is as easy to implement as a simple diode-OR droop
sharing system, but the traditional passive diodes are replaced with adjustable diodes,
with turn-on voltages that can be adjusted to achieve actively balanced current sharing.
THE CURRENT SHARING
CONTROLLER
The LTC4370 features Linear Technology’s
proprietary adjustable-diode current
sharing technique. It balances the load
between two supplies using external
N-channel MOSFETs that act as adjustable diodes whose turn-on voltage can
be modulated to achieve balanced sharing. Figure 1 shows the LTC4370 sharing
a 10A load between two 12V supplies
Figure 2 shows the device internals
that affect load sharing. Error amplifier EA monitors the differential voltage
between the OUT1 and OUT2 pins. It sets
the forward regulation voltage VFR of two
servo amplifiers (SA1, SA2), one for each
supply. The servo amplifier modulates
the gate of the external MOSFET (hence its
resistance) such that the forward drop
across the MOSFET is equal to the forward
regulation voltage. The error amplifier sets
Figure 2. Load-sharing-related internals of the LTC4370
I1
R1
M1
VSUPPLY1
C1
This article introduces a new method of
current sharing, allowing active control
of individual supply contributions, but
with the simplicity of droop sharing. In
this system, the diodes are replaced with
adjustable diodes with turn-on voltages
that can be adjusted to achieve balanced
current sharing. This produces better
sharing accuracy than droop sharing
and the power spent in the adjustable
diodes is the minimum required to
achieve sharing, far less than that lost in
a traditional diode. Because no sharing
bus is required, it offers simpler supplyindependent compensation and portable
design. Supplies with difficult or no
access to their trim pins and feedback
networks are ideal for this technique.
VIN1
GATE1
CPO1
+
OUT1
CHARGE
PUMP1
SA1
–
VFR1
+
–
COMP
CC
+
OUT1
–
OUT2
EA
SERVO
ADJUST
gm = 150µS
+
–
10µA
+
VIN2
GATE2
CHARGE
PUMP2
CPO2
VCC
RANGE
R3
0.3V
DISABLE
LOAD SHARE
–
+
–
SA2
TO
LOAD
+
VFR2
–
Droop sharing is simple but sharing
accuracy is poorly controlled, and the
series diodes present a voltage and power
loss. A more controlled way of current
sharing is to monitor the supply current,
compare it to an average current required
from each supply, then adjust the supply
voltage (through its trim pin or feedback
network) until the supply current matches
the required value. This method requires
wires to every supply—a share bus—to
signal the current contribution required
from each. The current sharing loop
compensation is customized to accommodate the power supply loop dynamics. Controlled current sharing requires
careful design and access to all of the
supplies—not possible in some systems.
VCC
OUT2
C2
R2
VSUPPLY2
I2
M2
July 2012 : LT Journal of Analog Innovation | 21
NORMALIZED
CURRENT
I2
VRANGE = 500mV
1
I1
1
= 2RS
SLOPE
0.5
I1
I2
0
–500mV
0
500mV
SHARING CAPTURE RANGE ±∆VIN(SH)
Figure 3. Current sharing
characteristic of the LTC4370
method as the supply voltage
difference varies.
MAXIMUM M2
MOSFET POWER
DISSIPATION
VIN1 – VIN2 = ∆VIN
IL • RS
MAXIMUM M1
MOSFET POWER
DISSIPATION
MOSFET
FORWARD
DROP
VFR(MAX)
525mV
VFWD2
VFWD1
VFR(MIN)
–500mV
25mV
0
500mV
VIN1 – VIN2 = ∆VIN
DRAWING IS NOT TO SCALE!
the VFR on the lower voltage supply to a
minimum value of 25mV. The servo on the
higher voltage supply is set to 25mV plus
the difference in the two supply voltages. In this way both the OUT pin voltages are equalized. OUT1 = OUT2 implies
I1 • R1 = I2 • R2 . Hence, I1 = I2 if R1 = R2 .
A simple adjustment to different-valued
sense resistors can be used to set up ratiometric sharing, i.e., I1 /I2 = R2 /R1. Note
that the load voltage tracks 25mV below
the lowest supply voltage.
The MOSFET in conjunction with the servo
amplifier behaves like a diode whose
turn-on voltage is the forward regulation
voltage. The MOSFET is turned off when its
forward drop falls below the regulation
voltage. With increasing MOSFET current,
the gate voltage rises to reduce the onresistance to maintain the forward drop
at VFR . This happens until the gate voltage
rails out at 12V above the source. Further
rise in current increases the drop across
the MOSFET linearly as IFET • RDS(ON).
Given the above, when the error amplifier sets the forward regulation voltage
of the servo amplifier, it is functionally
equivalent to adjusting the turn-on voltage
22 | July 2012 : LT Journal of Analog Innovation
of the (MOSFET-based) diode. The adjustment range runs from a minimum of
25mV to a maximum set by the RANGE pin
(see “Design Considerations” below).
The controller can load share supplies
from 0V to 18V. When both supplies are
below 2.9V, an external supply in the range
2.9V to 6V is required at the VCC pin to
power the LTC4370. Under reverse current
conditions the gate of the MOSFET is turned
off within 1µs. The gate is also turned
on in under a microsecond for a large
forward drop. The fast turn-on, important
for low voltage supplies, is achieved with
a reservoir capacitor on the integrated
charge pump output. It stores charge at
device power-up and delivers 1.4A of gate
pull-up current during a fast turn-on event.
The EN1 and EN2 pins can be used to
turn off their respective MOSFETs. Note
that current can still flow through the
body diode of the MOSFET. When both
channels are off, the device current
consumption is reduced to 80µ A per supply. The FETON outputs indicate whether
the respective MOSFET is on or off.
THE CURRENT SHARING
CHARACTERISTIC
Figure 3 shows the current sharing
characteristic of the LTC4370, adjustablediode method. There are two plots,
both with the supply voltage difference,
ΔVIN = VIN1 – VIN2 , on the x-axis. The
top plot shows the two supply currents normalized to the load current; the
lower shows the forward voltage drops,
VFWDx, across the MOSFETs. When both
supply voltages are equal (ΔVIN = 0V),
the supply currents are equal, and both
forward voltages are at the minimum
servo voltage of 25mV. As VIN1 increases
above VIN2 (positive ΔVIN), VFWD2 stays
at 25mV, while VFWD1 increases exactly
with ΔVIN to maintain OUT1 = OUT2.
This is turn keeps I1 = I2 = 0.5ILOAD.
There is an upper limit to the adjustment on VFWD set by the RANGE pin. For
the example in Figure 3, that limit is
525mV, set by the RANGE pin at 500mV.
Once VFWD1 hits this limit, sharing
becomes imbalanced and any further
rise in VIN1 pushes OUT1 above OUT2.
The break point is VFR(MAX) – VFR(MIN),
where more of the load current comes
from the higher voltage supply. When
OUT1 – OUT2 = ILOAD • RSENSE , the entire
load current transfers over to I1. This is
the operating point with the maximum
power dissipation in MOSFET M1, since
the entire load current flows through
it with the maximum forward drop.
For example, a 10A load current causes
5.3W (= 10A • 525mV) dissipated in the
MOSFET. For any further rise in ΔVIN,
the controller ramps down the forward
design features
The LTC4370’s novel approach to load-sharing power
supplies results in easy design, especially with supplies
that don’t lend themselves to on-the-fly tweaking.
Inherent diode behavior protects supplies from reverse
currents and the system from faulting supplies.
drop across M1 to the minimum 25mV.
This minimizes power dissipation in
the MOSFET for large VIN when the load
current is not being shared. The behavior is symmetric for negative ΔVIN .
The sharing capture range in this example
is 500mV and is set by the RANGE pin
voltage. With this range the controller
can share supplies that have a tolerance
of ±250mV. This translates to the following: ±7.5% tolerance on a 3.3V supply,
±5% on a 5V, and ±2% on a 12V supply.
DESIGN CONSIDERATIONS
These are some of the high level considerations for a load share design.
MOSFET Choice — Ideally the MOSFET’s RDS(ON)
should be small enough that the controller
can servo the minimum forward regulation
voltage of 25mV across the MOSFET with
half of the load current flowing through
it. A higher RDS(ON) prevents the controller from regulating 25mV. In this case, the
unregulated drop is 0.5IL • RDS(ON). As this
drop rises, the sharing break point (now
defined by VFR(MAX) – 0.5IL • RDS(ON)) occurs
earlier, shrinking the capture range.
Since the MOSFET dissipates power, up to
IL • VFR(MAX) as in Figure 3, its package and
heat sink should be chosen appropriately.
The only way to dissipate less power in
the MOSFET is by using more accurate
supplies or by forgoing sharing range.
RANGE Pin — The RANGE pin sets the shar-
ing capture range of the application,
which in turn depends on the accuracy of the supplies. For example, a
5V system with ±3% tolerance supplies
Figure 4. 5V diode-OR load share with
status light. Red LED D1 lights up
whenever any MOSFET is off, indicating
a break in sharing.
SUM85N03-06P
VINA
5V
39nF
EN1
0.1µF
CPO1
VIN1 GATE1
VCC
GND
LTC4370
OUT1
FETON1
FETON2
D1
820Ω
SHARE
OFF
2.5mΩ
2.5mΩ
OUT
10A
RANGE
would need a sharing range of
OUT2
30.1k
2 • 5V • 3% or 300mV (higher
EN2 CPO2 VIN2 GATE2 COMP
0.18µF
supply is 5.15V while lower
39nF
D1: RED LED
LN1251C
VINB
is 4.85V). The RANGE pin has
5V
SUM85N03-06P
a precise internal pull-up current of
10µ A. Placing a 30.1k resistor on the
RANGE pin sets its voltage to 301mV and
CONCLUSION
now the controller can compensate for the
Balancing load currents between supplies
300mV supply difference (see Figure 4).
is a historically difficult problem, conjuring visions of juggling on a tightrope.
Leaving the RANGE pin open (as shown
When power modules or bricks don’t offer
in Figure 1) gives the maximum posbuilt-in support, some designers will spend
sible sharing range of 600mV. But when
significant time designing a well-controlled
servo voltages approach the diode voltsystem (and redesigning it whenever the
age, currents can flow through the body
supply type changes); others will settle for
diode of the MOSFET causing loss of
crude resistance-based droop sharing.
sharing. Connecting RANGE to VCC disables load share to transform the device
The LTC4370 takes a completely differinto a dual ideal-diode controller.
ent approach to load-sharing supplies
Compensation — The load share loop is
compensated by a single capacitor from
the COMP pin to ground. This capacitor
must be 50× the input (gate) capacitance
of the MOSFET, CISS . If fast gate turn-on is
not being used (CPO capacitors absent)
then the capacitor can be just 10× CISS .
Sense Resistors — The sense resistors deter-
than any other controller. It eases design,
especially with supplies that don’t lend
themselves to on-the-fly tweaking, and it
can be ported to various types of supplies. Inherent diode behavior protects
supplies from reverse currents and the
system from faulting supplies. The LTC4370
provides a simple, elegant and compact
solution to a complicated problem. n
mine the load sharing accuracy. Accuracy
improves as resistor voltage drops
increase. The maximum error amplifier
offset is 2mV. Therefore, a 25mV sense resistor drop yields a 4% sharing error. The
resistance can be lowered if power dissipation is more important than accuracy.
July 2012 : LT Journal of Analog Innovation | 23
What’s New with LTspice IV?
Gabino Alonso
New Video: Evaluating
Electrical Quantities
NEW HOW-TO VIDEOS
NEW DEMO CIRCUITS
Evaluating Electrical Quantities
video.linear.com/115
µModule Regulators
The LTspice IV waveform viewer provides
visual analysis of circuit performance and
performs basic measurements. Sometimes,
though, you need a more in depth numerical analysis of circuit performance. For
this, .MEASURE statements allow you to
perform direct measurements such as:
•rise, fall and time delay
•average, RMS, min, max
and peak-to-peak
•find X when Y occurs
•derivative and integral evaluations
This new video shows an example
of how to use .MEASURE statements for numerical analysis.
What is LTspice IV?
LTspice® IV is a high performance SPICE
simulator, schematic capture and waveform
viewer designed to speed the process of power
supply design. LTspice IV adds enhancements
and models to SPICE, significantly reducing
simulation time compared to typical SPICE
simulators, allowing one to view waveforms for
most switching regulators in minutes compared
to hours for other SPICE simulators.
LTspice IV is available free from Linear
Technology at www.linear.com/LTspice. Included
in the download is a complete working version of
LTspice IV, macro models for Linear Technology’s
power products, over 200 op amp models, as
well as models for resistors, transistors and
MOSFETs.
24 | July 2012 : LT Journal of Analog Innovation
• LTM8047: 725V DC isolated low noise
µModule regulator (3.1–29V to 5V at
280m A) www.linear.com/LTM8047
• LTC3866: High efficiency,
1.5V/30A step-down converter with
very low DCR Sensing (4.5V–20V to
1.5V at 30A) www.linear.com/LTC3866
LED Drivers
• LTM8048: 725V DC isolated low
noise µModule regulator with
LDO post regulator (3.5V–30V to
5V at 120m A & 5.7V at 120m A)
• LT3799-1: Offline isolated flyback
LED controller with active power
factor correction (PFC) (277VAC to
3A/36V) www.linear.com/LTC3799-1
Monolithic Switching Regulators
MACRO MODELS
• LTC3103: Solar-powered buck supply with
Li battery backup (3.2V–15V to 2.2V at
300m A) www.linear.com/LTC3103
• LTC3115-1: 12V 1MHz buck-boost regulator
with undervoltage lockout (10V–40V to
12V at 1.4A) www.linear.com/LTC3115-1
Switching Regulator Controllers
• LT3798: Isolated no opto-coupler flyback
controller with active power factor
correction (PFC) (90VAC –265VAC to
24V at 2A) www.linear.com/LTC3798
µModule Regulators
• LTM8026: 36V input, 5A CVCC
step-down µModule regulator
www.linear.com/LTM8026
• LTM8029: 36V input, 600m A step-down
µModule converter with 5µ A quiescent
current www.linear.com/LTM8029
Monolithic Switching Regulators
• LT3692A: Monolithic dual tracking
3.5A step-down switching regulator
www.linear.com/LT3692A
• LT8582: 1.5MHz +5V to ±12V dual
converter (5V to ±12V at 550m A)
www.linear.com/LT8582
• LT3973: 42V, 750m A step-down regulator
with 2.5µ A quiescent current and integrated diodes www.linear.com/LT3973
• LTC3765/LTC3766: Active clamp
forward converter (18V–72V to
12V at 12.5A) www.linear.com/LTC3765,
www.linear.com/LTC3766
• LT3988: Dual 60V monolithic
1A step-down switching regulator
www.linear.com/LT3988
• LTC3838: Dual output, 350kHz, step-down
converter with differential DCR output
sensing (4.5V–38V to 1.2V at 15A &
1.5V & 15A) www.linear.com/LTC3838
• LTC3839: 2MHz, 2-phase, step‑down
converter with differential
RSENSE sensing (4.5-14V to 3.3V at 25A)
www.linear.com/LTC3839
• LT3992: Monolithic dual tracking
3A step-down switching regulator
www.linear.com/LT3992
• LT8610: 42V, 2.5A synchronous stepdown regulator with 2.5µ A quiescent
current www.linear.com/LT8610
• LT8611: 42V, 2.5A synchronous
step-down regulator with current
sense and 2.5µ A quiescent current
www.linear.com/LT8611
design ideas
Follow @LTspice on Twitter for
up-to-date information on models, demo circuits,
events and user tips: www.twitter.com/LTspice
• LTC3600: 15V, 1.5A synchronous
rail-to-rail single resistor step-down
regulator www.linear.com/LTC3600
• LTC3630: High efficiency,
65V 500m A synchronous step-down
converter www.linear.com/LTC3630
Switching Regulator Controllers
LED Drivers
Differential Amplifiers & ADC Drivers
• LT3791: 60V 4-switch synchronous
buck-boost LED driver controller
www.linear.com/LT3791
• LTC6362: Precision, low power rail-to-rail
input/output differential op amp/
SAR ADC driver www.linear.com/LTC6362
• LT3796: 100V constant-current and
constant-voltage controller with dual
current sense www.linear.com/LT3796
System Supervisor, Monitor and Control
• LT4363-2: High voltage surge stopper with
current limit www.linear.com/LT4363
• LT3798: Isolated no opto-coupler flyback
controller with active power factor
correction (PFC) www.linear.com/LT3798
Inductorless Converters
• LTC3260: Low noise dual supply inverting
charge pump www.linear.com/LTC3260
• LTC2960: 36V Nano-current
two input voltage monitor
www.linear.com/LTC2960
• LTC3861: Dual, multiphase step-down
voltage mode DC/DC controller
with accurate current sharing
www.linear.com/LTC3861
• LTC3261: High voltage, low quiescent
current inverting charge pump
www.linear.com/LTC3261
• LTC2955-1/-2: Pushbutton on/off
controller with automatic turn-on
www.linear.com/LTC2955 n
Power User Tip
ANNOTATING SCHEMATIC AND WAVEFORM PLOTS
Adding informative comments to a schematic using text is very useful. However,
there are times when adding a line, rectangle, circle or arc to a schematic can
better illustrate a circuit. A classic example is highlighting a transformer core using
two lines. These graphical annotations are available under Draw in the Edit menu. If
you do not want these graphical annotations to snap to the grid, you can hold down
the Ctrl key while positioning.
Waveform plots can be annotated with text, arrows, lines, boxes and circles.
These annotations are effective for illustrating a particular result in your plot to a
colleague. Plot annotations along with Move and Drag can be found under Notes &
Annotations under Plot Setting menu. Note that if you annotate a plot you will need
to save your annotations via a Plot Setting file (available under Plot Setting menu);
otherwise they will not be saved.
Happy simulations!
Examples of graphical
annotations added
to a schematic and a
waveform plot.
July 2012 : LT Journal of Analog Innovation | 25
24V, 15A Step-Down Regulator in Sub-1mm Height Package
Pushes Monolithic Performance Limits
Stephanie Dai
Monolithic switching regulators and switching controllers together dominate the DC/DC
converter market. Generally, there is little overlap in their respective applications.
Controller-based solutions are favored for high performance, higher power applications
where minimal power loss and top thermal performance are priorities. In contrast,
monolithic regulators are favored in lower power applications where compact size
is the main requirement. Controllers typically offer more features than monolithic
solutions, but are at a significant solution-size disadvantage. The light footprint of
monolithic regulators usually comes at a cost of features and increased power loss,
and their reliance on integrated MOSFETs places practical limits on power.
The LTC3613 monolithic regulator
blurs the line drawn between applications for controllers and monolithic
regulators by combining a high performance fully featured controller
with onboard low RDS(ON) MOSFETs.
FEATURES
The LTC3613 can accept an input voltage between 4.5V to 24V and supports
output voltages between 0.6V to 5.5V. The
onboard top and bottom MOSFETs feature low RDS(ON), around 7mΩ and 5mΩ,
respectively, keeping power dissipation
low and allowing the LTC3613 to deliver
up to 15A of adjustable load current.
The LTC3613 features true remote differential output voltage sensing. This allows
for accurate regulation of the output
with maximum load currents and shared
ground planes. This feature is critical for
low output voltage applications, where
even small voltage offsets caused by
parasitic IR drops on board traces can
cost several percentage points in regulation accuracy. Remote differential output
sensing and the LTC3613’s accurate internal
reference combine to offer excellent output regulation accuracy over line, load and
26 | July 2012 : LT Journal of Analog Innovation
temperature: ±0.25% at 25°C, ±0.67% from
0°C to 85°C and 1% from –40°C to 125°C.
frequency is constant over steady state
conditions under line and load. It also
allows the LTC3613 to recover from a large
load step in only a few short cycles. This
architecture yields well balanced current
sharing among multiple LTC3613s, which
can be easily paralleled for high power
applications. It also includes a phaselock loop (PLL) for synchronization to an
The LTC3613 has a low minimum on-time
of 60ns, allowing for high step-down
ratios at high switching frequencies.
Because of its sophisticated controlled ontime, valley current mode architecture, the
on-time is controlled so that the switching
Figure 1. 24V input to 1.2V output using inductor DCR sensing to minimize solution size and cost and to
maximize efficiency
LTspice IV
INTVCC
RPGD
100k
RDIV1
52.3k
PVIN
350kHz
CSS
0.1µF
CITH1
220pF RITH
28k
CIN2
10µF
VOUT
PGOOD
LTC3613
VRNG
RDIV2
10k
circuits.linear.com/560
SVIN
RUN
SENSE–
SENSE+
MODE/PLLIN
EXTVCC
SW
CDCR RDCR
0.1µF 3.09k
L1
0.56µH
VOUT
1.2V
15A
CB 0.1µF
BOOST
TRACK/SS
DB
INTVCC
ITH
INTVCC
CVCC
4.7µF
CITH2 100pF
RT
115k
VIN
CIN1 6V TO 24V
82µF
25V
+
RFB2
20k
RFB1
20k
COUT2
100µF
×2
+
PGND
RT
SGND
VOSNS+
VOSNS–
CIN1: SANYO 25SVPD82M
COUT1: SANYO 2R5TPE330M9
3613 F10
DB: CENTRAL CMDSH-3
L1: VISHAY IHLP4040DZ-056µH
COUT1
330µF
2.5V
×2
design ideas
LTC3608
LTC3609
LTC3610
LTC3611
LTC3613
PV IN(MAX)
18V
32V
24V
32V
24V
I LOAD(MAX)
8A
6A
12A
10A
15A

Frequency Sync
Precise Differential
Output Sensing
±1%
±1%
±1%
±1%
±0.67%
Accurate Current
Sensing
Bottom FET R DS(ON)
Bottom FET R DS(ON)
Bottom FET R DS(ON)
Bottom FET R DS(ON)
R SENSE or DCR sensing
MOSFET R DS(ON)
Top/Bottom
10mΩ/8mΩ
18mΩ/13mΩ
12mΩ/6.5mΩ
15mΩ/9mΩ
7.5mΩ/5.5mΩ
Package
7mm × 8mm × 0.9mm
64-pin
7mm × 8mm × 0.9mm
64-pin
9mm × 9mm × 0.9mm
52-pin
9mm × 9mm × 0.9mm
52-pin
7mm × 9mm × 0.9mm
56-pin
Table 1. High power monolithic regulator family
external clock, especially beneficial for
high current, low output voltage applications where interleaving of paralleled
phases can minimize output voltage ripple.
50%, then the maximum sense voltage
is reduced to about one-fourth of its full
value, limiting the inductor current level
to one-fourth of its maximum value.
The LTC3613 includes several safety and
protection features including overvoltage protection and current-limit foldback. If the output exceeds 7.5% of the
programmed value, then it is considered
an overvoltage (OV) condition, the top
MOSFET is immediately turned off and
the bottom MOSFET is turned on indefinitely until the OV condition is cleared.
A power good output monitor is also
available which flags if the part is outside
a ±7.5% window of the 0.6V reference
voltage. In the case of an output short
circuit, if the output fails by more than
The LTC3613 offers precise control of
the output during start-up and shutdown sequencing though its output
voltage tracking and soft-start features. An external VCC input pin is
also available, allowing for bypassing of its internal LDO for an efficiency
benefit in high power applications.
Figure 2. Efficiency of the regulator in Figure 1
Figure 3. Load transient response of the circuit in
Figure 1
The LTC3613 can be configured to sense
the inductor current through a series
sense resistor, RSENSE , or via an inductor DCR sensing network. The tradeoffs between the two current sensing
schemes are largely matters of cost, power
100
90
EFFICIENCY (%)
80
PULSE-SKIPPING
MODE
VOUT
100mV/DIV
70
60
50
FORCED
CONTINUOUS
MODE
IL
10A/DIV
40
30
20
VIN = 12V
VOUT = 1.2V
10
0
0.1
1
10
LOAD CURRENT (A)
100
ILOAD
10A/DIV
40µs/DIV
LOAD TRANSIENT = 0A TO 15A
VIN = 12V, VOUT = 1.5V
FIGURE 10 CIRCUIT
consumption and accuracy. DCR sensing
owes its increasing popularity to its lower
expense and power loss compared to a
sense resistor scheme. Even so, the tight
tolerances of current-sensing resistors
provide the most accurate current limit.
Figure 1 shows a typical application of
the LTC3613, configured for DCR sensing
in a high step-down solution, 24V to 1.2V,
and synchronized to a 350kHz external
clock. Figure 2 shows the efficiency and
Figure 3 shows transient performance.
CONCLUSION
The LTC3613 offers far more design flexibility than a typical monolithic switching
regulator by including a variety of userprogrammable features such as soft-start,
programmable frequency, external clock
synchronization, adjustable current limit
and selectable light load operating modes.
Its critical safety features such as overvoltage protection and programmable current limit with foldback current limiting
further improve the robustness of the
part. An external VCC input is provided
for high power applications. Its compact
solution size, extensive feature set and
high performance capabilities extend
its range of use compared to traditional
monolithics, making it suitable for an
an expanding range of applications. n
July 2012 : LT Journal of Analog Innovation | 27
Step-Down Converter Delivers 25A at 12V Output from
Inputs Up to 60V
Victor Khasiev
The LTC3890 (dual outputs) and LTC3891 (single output) step-down DC/DC controllers
directly accept inputs from 4V to 60V. This wide input range covers input voltages for
single or double battery automotive environments, eliminating the need for snubbers and
voltage suppression circuitry typically required to protect ICs during load dumps. This
range also encompasses 48V telecom applications. If no galvanic isolation is required
between the input and output voltages, the LTC3890 and LTC3891 can replace expensive
and bulky transformer-based converters. Compared to a transformer-based solution,
an LTC3890 or LTC3891 step-down converter increases efficiency, reduces power loss
in the supply lines, simplifies layout and significantly reduces the bill of materials.
HIGH EFFICIENCY 2-PHASE
CONVERTER PRODUCES 12V AT 25A
Figure 1 shows the LTC3890 in a 2-phase
single output step-down converter configuration that delivers 25A at 12V, which
can be scaled up to 75A by adding more
Figure 1. High efficiency
converter produces 25A at
12VOUT from inputs up to 60V
LTC3890 ICs to increase the number of
power phases. For lower output current,
the single-phase LTC3891 can be used.
Implementing a 2-phase converter simply
requires tying together the independent
channel pins of the LTC3890, namely,
30.1k
0.1µF
47pF
4.7nF
9.76k
47pF
35.7k
VOUT
PLLIN
LTC3890
FREQ
10pF
RUN2
2.2pF
SENSE1+
SS1
TG1
SS2
SW1
VIN
100Ω
RJK0651DPB
0.1µF DFLS1100
ITH1
ITH2
2.2µF/100V
×4
1µF/100V
SENSE1–
RUN1
VFB2
PGND
3m
10µF
×2
BOOST1
+
150µF
RJK0653DPB
INT
VOUT
12V AT 25A
4.7µF
INTVCC
VIN
RJK0651DPB
TG2
L2 10µH
SW2
0.1µF DFLS1100
PGOOD2
ILIM
L1
10µH
INT
BG1
PGOOD1
1µF
2.2Ω
VIN
VFB1
499k
Although the ITH pins are connected
together, each is terminated to a separate
47pF capacitor to compensate
VIN, 16V TO 60V
1M
57.6k
FB1 and FB2, TRACK/SS1 and TRACK/SS2,
RUN1 and RUN2, ITH1 and ITH2.
10µF
×2
INT
BOOST2
3m
+
EXTVCC
RJK0653DPB
BG2
SENSE2+
100Ω
2.2pF
SENSE2–
28 | July 2012 : LT Journal of Analog Innovation
L1, L2: WÜRTH 7443631000
150µF
design ideas
The LTC3890 dual output, synchronous step-down
converter can be easily configured as a single output,
dual phase converter for high input voltage, high output
current automotive and telecom applications.
for possible noise from interconnecting
traces. A relatively low switching frequency, around 150kHz, and a relatively
high phase inductance of 10µ H are used
to reduce switching losses at high input
voltages. The output voltage is fed to
the EXTVCC pin to reduce losses associated with biasing the chip and internal
gate drivers at high input voltages.
CIRCUIT PERFORMANCE
Efficiency is shown in Figure 2, measured without cooling air flow. Efficiency
peaks close to 98% in the middle of
the load range and declines to 96% at
the 25A maximum load. Figure 3 shows
the average input current vs input voltage at no load in Burst Mode operation. The value of this current is below
0.5m A. Figure 4 shows a thermal map
of the board with no air flow present at
VIN = 20V and VOUT = 12V at 25A (300W).
CONCLUSION
The LTC3890 dual output, synchronous step-down converter can be
easily configured as a single output,
dual phase converter for high input
voltage, high output current automotive and telecom applications. n
where f is the switching frequency
and k is a coefficient defined by
the current imbalance between the
phases. For converters based on the
LTC3890, k = 1.08, assuming current
sense resistors with a 1% tolerance.
97.5
97.0
96.5
96.0
Figure 4. Temperature hot spots with no air flow
0.22
0.20
INPUT CURRENT (mA)
EFFICIENCY (%)
98.0
0.18
0.16
0.14
0.12
95.5
95.0
∆I 2
12
(VIN – VOUT) • D
∆I =
L•f
VOUT
D=
VIN
I
I PH = k • OUT
2
∆I
I PK = I PH +
2
I RMS = (IPH)2 +
20V
36V
50V
98.5
Selection of power MOSFETs and input/
output capacitors is described in detail in
the LTC3890 data sheet. It is important to
note that the typical internal VCC voltage and, consequently, the MOSFET gate
voltage is 5.1V. This means that logic level
MOSFETs must be used in the design.
Two values define selection of
the inductor: RMS current (IRMS)
and saturation current (IPK):
Figure 3. Average input current vs input voltage at
no load. VOUT is 12V.
Figure 2. Efficiency at VIN = 20V, 36V and 50V
99.0
COMPONENT SELECTION
1
6
11
16
LOAD (A)
21
26
0.10
20
25
30
35
40
45
50
VIN
July 2012 : LT Journal of Analog Innovation | 29
1.5A Rail-to-Rail Output Synchronous Step-Down Regulator
Adjusts with a Single Resistor
Jeff Zhang
The LTC3600 features a
wide output range with
tight regulation over the
entire range. In most
regulators, the lowest
output voltage is limited
to the reference voltage.
The LTC3600, however,
uses a novel regulation
scheme with a precision
50µA current source and
a voltage follower, creating
an adjustable output from
“0V” to close to VIN. It
also features constant
loop gain, independent of
the output voltage, giving
excellent regulation at any
output and allowing multiple
regulators to be paralleled
for high output currents.
LTC3600
VIN
VIN
12V
RUN
50µA
10µF
BOOST
+
–
Figure 1. High efficiency,
12V to 3.3V 1MHz stepdown regulator with
programmable reference
2.2µH
SW
PWM CONTROL
AND SWITCH
DRIVER
ERROR
AMP
0.1µF
VOUT
3.3V
22µF
GND
VOUT
ISET
0.1µF
MODE/
SYNC INTVCC RT
RSET
66.5k
OPERATION
The LTC3600 is a current mode monolithic
step-down buck regulator with excellent
line and load transient responses. The
200kHz to 4MHz operating frequency can
be set by a resistor or synchronized to an
external clock. The LTC3600 internally
generates an accurate 50μA current source,
allowing the use of a single external
resistor to program the reference voltage
PGFB
ITH PGOOD
1µF
from 0V to 0.5V below VIN . As shown in
Figure 1, the output feeds directly back
to the error amplifier with unity gain.
The output equals the reference voltage
at the ISET pin. A capacitor can be paralleled with RSET for soft-start or to improve
noise while an external voltage applied
to the ISET pin is tracked by the output.
Figure 3. Efficiency of 12V input to 3.3V output
regulator in CCM and DCM mode
Figure 2. 0A to 1.5A load step response of the Figure 1 schematic
100
90
80
VOUT
100mV/DIV
IOUT
1A/DIV
EFFICIENCY (%)
VOUT
100mV/DIV
IOUT
1A/DIV
70
60
50
40
30
20
2VIN to 3.3VOUT (DCM)
2VIN to 3.3VOUT (CCM)
10
50µs/DIV
12VIN TO 3.3VOUT
INTERNAL COMPENSATION
ITH TIED TO INTVCC
30 | July 2012 : LT Journal of Analog Innovation
50µs/DIV
12VIN TO 3.3VOUT
ITH TIED TO EXTERNAL
RC COMPENSATION (R=56kΩ, C=68pF)
0
–0.1 0.1
0.3
0.5 0.7 0.9
ILOAD (A)
1.1
1.3
1.5
design ideas
VIN
BOOST
LTC3600
12V
RUN
50µA
+
10µF
0.1µF
PWM
CONTROL
AND
SWITCH
DRIVER
ERROR
AMP
–
2.2µH
SW
0.1Ω
GND
IOUT =
0A TO 1.5A
22µF
VOUT
Internal loop compensation stabilizes
the output voltage in most applications, though the design can be customized with external RC components.
The device also features a power good
output, adjustable soft-start or voltage tracking and selectable continuous/
discontinuous mode operation. These
features, combined with less than 1μA supply current in shutdown, VIN overvoltage protection and output overcurrent
protection, make this regulator suitable
for a wide range of power applications.
ISET
RT
PGFB
ITH
PGOOD
1µF
RSET
0k TO 3k
Figure 4. The LTC3600 as a programmable 0A to 1.5A current source
Figure 5. 15V to 2.5V, 3A dual phase regulator
VIN
4V
TO
15V
LTC3600
VIN
RUN
100k
50µA
10µF
BOOST
0.1µF
+
PWM CONTROL
AND SWITCH
DRIVER
ERROR
AMP
–
APPLICATIONS
Figure 1 shows the complete LTC3600
schematic in a typical application that
generates a 3.3V output voltage from
12V input. Figure 2 shows the load step
transient response using internal compensation and with external compensation. Figure 3 shows the efficiency in
CCM and DCM modes. Furthermore, the
LTC3600 can be easily configured to be
a current source, as shown in Figure 4.
By changing the RSET resistance from
0Ω to 3kΩ, the output current can
be programmed from 0A to 1.5A.
MODE/
SYNC INTVCC
22µF
VOUT
ISET
MODE/
SYNC INTVCC
RT
GND PGFB
VOUT = 2.5V
3A
ITH PGOOD
10pF
1µF
27k
VIN
4V
TO
15V
9
LTC3600
VIN
100k
RUN
50µA
5
10µF
150pF
BOOST
8
0.1µF
+
PWM CONTROL
AND SWITCH
DRIVER
ERROR
AMP
–
2.2µH
SW
7
VOUT
CONCLUSION
The LTC3600 uses an accurate internal
current source to generate a programmable reference, expanding the range of
output voltages. This unique feature gives
the LTC3600 great flexibility, making it
possible to dynamically change the output
voltage, generate current sources, and
parallel regulators for applications that
would be difficult to implement using a
standard DC/DC regulator configuration. n
2.2µH
SW
ISET
MODE/
SYNC INTVCC
6
1
10
24.9k
0.1µF
100k
RT
3
GND PGFB
13
1µF
V+
INTVCC
11
GND
SET
LTC6908-1*
4
ITH PGOOD
2
22µF
12
10pF
OUT1
OUT2
MOD
*EXTERNAL CLOCK FOR FREQUENCY SYNCHRONIZATION IS RECOMMENDED
July 2012 : LT Journal of Analog Innovation | 31
highlights from circuits.linear.com
LOW NOISE ±12V POWER SUPPLY FROM A
SINGLE-ENDED 15V INPUT SUPPLY
The LTC3260 is a low noise dual polarity output power supply that
includes an inverting charge pump with both positive and negative
LDO regulators. The charge pump operates over a wide 4.5V to 32V
input range and can deliver up to 100mA of output current. Each LDO
regulator can provide up to 50mA of output current. The negative LDO
post regulator is powered from the charge pump output. The LDO
output voltages can be adjusted using external resistor dividers.
circuits.linear.com/556
15V
LDO+
VIN
10µF
12V
10µF
LTC3260
1µF
EN+
ADJ+
EN–
BYP+
MODE
GND
C+
BYP–
C–
ADJ–
VOUT
LDO–
10nF
10nF
909k
100k
100k
909k
LTspice IV
10µF
circuits.linear.com/556
–12V
10µF
RT
200k
OVERVOLTAGE REGULATOR WITH 250V SURGE PROTECTION
The LT4363 surge stopper protects loads from high voltage transients. It
regulates the output during an overvoltage event, such as load dump in
vehicles, by controlling the gate of an external N-channel MOSFET. The
output is limited to a safe value, allowing the loads to continue functioning.
The LT4363 also monitors the voltage drop between the SNS and OUT
pins to protect against overcurrent faults. An internal amplifier limits the
voltage across the current sense resistor to 50mV. In either fault condition,
a timer is started inversely proportional to MOSFET stress. Before the
timer expires, the FLT pin pulls low to warn of an impending power-down.
If the condition persists, the MOSFET is turned off. The LT4363-1 remains
off until reset, whereas the LT4363-2 restarts after a cooldown period.
circuits.linear.com/529
VIN
12V
Q1
FDB33N25
RSNS
10mΩ
MPS-A42
D1*
SMAJ58A
0.1µF
10Ω
57.6k
47nF
GATE SNS
VCC
127k
OUT
FB
SHDN
4.99k
UV
SHDN GND
ENOUT
circuits.linear.com/529
OV
GND
FLT
TMR
FAULT
0.1µF
*DIODES INC.
VIN
7V TO 36V
VCC
DC/DC
CONVERTER
LT4363DE-2
49.9k
LTspice IV
36V INPUT, 5.6A TWO 2.5V SERIES SUPERCAPACITOR CHARGER
The LTM8026 is a 36V input, 5A constant-voltage, constant-current (CVCC)
step-down µModule regulator. Included in the package are the switching
controller, power switches, inductor and support components. Operating
over an input voltage range of 6V to 36V, the LTM8026 supports an output
voltage range of 1.2V to 24V. CVCC operation allows the LTM8026 to
accurately regulate its output current up to 5A over the entire output range.
The output current can be set by a control voltage, a single resistor or a
thermistor. Only resistors to set the output voltage and frequency and the
bulk input and output filter capacitors are needed to complete the design.
circuits.linear.com/543
OUTPUT
CLAMP
AT 16V
22µF
49.9k
10µF
510k
VIN
LTM8026
VOUT
VOUT
5V
RUN
SS
SYNC
CTL_I
COMP
CTL_T
GND ADJ
RT
2.5V
2.2F
VREF
68.1k
47µF
2.5V
2.2F
3.09k
LTspice IV
circuits.linear.com/543
L, LT, LTC, LTM, Linear Technology, the Linear logo, LTspice, Burst Mode, and µModule are registered trademarks, and PowerPath and ThinSOT are trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
© 2012 Linear Technology Corporation/Printed in U.S.A./55.8K
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(408) 432-1900
www.linear.com
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